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  • IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO. 3, JUNE 1991 203

    25-kW / 5 0 - k ~ z Generator for Induction Heating Enrique J. Dede, Member, IEEE, Josk V. Gonzilez, Juan A. Linares, Josc Jordan, Diego Ramirez,

    Student Member, IEEE, and Pablo Rueda, Student Member, IEEE

    Abstract-This report presents the features, technology, and construction of a transistorized generator for induction heating operating over the 4-50-kHz frequency range, which was devel- oped in the Research and Development Department of G. H. Industrial S.A.

    This new type of 25-kW-output-power generator, allows re- placement of the electronic tube generators for most of their applications. The most outstanding advantages of this new generator with respect to tube generators are more energy effi- ciency, extended life, reduced size, separated heating station of the generator, and connection by flexible cable.

    In addition, the generator has incorporated a frequency auto- matic tracking system that allows one to operate without any adjustments over a wide frequency range.

    I. INTRODUCTION T THE MOMENT, two basic types of generators for A induction heating applications are on the market: thyris-

    tors static inverters and high-frequency electron tube genera- tors.

    The thyristor static inverters are used at frequencies from 500 Hz to 10 kHz. They are machines with great electrical and mechanical robustness and satisfactory energy efficiency. On the other hand, the most important problem is that the frequency range of a thyristor static inverter is relatively narrow so that in case of frequent inductor changes, it is necessary to make difficult adjustments.

    The thyristor technology is limited to use at frequencies less than 10 kHz. This is due mainly to the recovery time needed, which is very short, and the thyristor basic operating principle does not allow such short recovering times.

    Electronics tube generators, unlike thyristor static invert- ers, do not have any limit related to the frequencies used. There are, at this moment, two type of electronic tube generators: conventional or classic ones and aperiodic ones. Both types of generators have in common the use of the electronic tube. The most important disadvantages of both types of generators are produced by the unavoidable use of the electronic tube.

    Electronic tubes have a limited life that, in practice, lies between 4000 and 6OOO operating hours, depending on the working conditions. On the other hand, the tube is a rela- tively inefficient element itself (never more than 75 %),

    Manuscript received lune 6, 1990; revised December 3, 1990. E. J. Dede, I . V. Gonzalez, I. A. Linares, and I. Jordan are with G.H.

    D. Ramirez is with the Departmento de Electr6nica e Infodt ica , Univer-

    P. Rueda is with the European Space Technology Center, Noordwijk, the

    IEEE Log Number 9144759.

    Industrial, S.A., Valencia, Spain.

    sidad de Valencia, Valencia, Spain.

    Netherlands.

    which conditions the generator global efficiency. Therefore, the efficiency to be achieved in practice is 50 % in classic generators and 70 % in the aperiodic one.

    We present a bipolar transistor generator for induction heating, which was designed by the Research and Develop- ment Department of G.H. Industrial S.A. in Valencia, Spain, and, now having ended the development stage, is in the commercialized phase in Europe and the United States.

    11. MAIN FEATURES OF THE DEVELOPED TRANSISTORIZED GENERATOR

    The efficiency of the generator is higher than 90 %. It operates in very wide frequency range with the same genera- tor. The frequency ratio range with the same generator is 1:3. The change of the inductors is simple because of the wide range of frequencies to be used. It is approximately one tenth the size of comparable electronic tube generators.

    III. TOPOLOGICAL FUNDAMENTALS Basically, the load of an induction heating generator is an

    inductor in which we find the piece to be heated. A direct feed of the heating coil would result in apparent to real power ratio that is too high; therefore, compensation of the heating coil is needed. Compensation of the power factor is carried out by a capacitor dimensioned so that this factor will be close to the unity at the working frequency.

    The compensation capacitor can be placed in series or in parallel with the inductor. In the first case, the load acts like a current source (inductance in series), and therefore, it has to be fed by the voltage source (voltage-fed inverter). In the second case, when the load is a parallel resonant circuit, it will react like a voltage source (capacitor in parallel), and therefore, it has to be fed by a current source (current-fed inverter).

    On the other hand, when driving a resonant load by an inverter, there will always be a phase-shift between output voltage and current. In the case of a series load, in some time intervals, the current will flow from the load to the power source. This implies that in an inverter with a series resonant load (series inverter), the switches must be bidirectional in current and unidirectional in voltage.

    In an inverter with parallel resonant load (parallel inverter), there will be some time intervals in which the output voltage is opposite to the output current. In this case, the switches must be bidirectional in voltage and unidirectional in current (dual to the former case). These above theoretical differences are summarized in Fig. 1.

    Basically, the two load structures mentioned above adapt correctly to the induction heating. However, there are some

    0278-0046/91/0600-0203$01.00 0 1991 IEEE

  • 204 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO. 3, JUNE 1991

    series resonance Parallel resonance The power delivered by the generator is controlled by a feedback loop comparing the nominal dc current with the actual one. The output of the loop regulates the duty cycle of the chopper. The inverter is controlled in such a way that the driving pulses Q and its complementary a have at each moment a frequency lower than the resonant frequency of the load (which is variable with the chosen inductor and also varies during the heating cycle, especially when the piece

    This power structure is controlled by four european dou- ble-size control circuit boards. The chopper, the starter, and inverter bridge consist of plug-in european sized boards, where the power transistors are water cooled.

    A . Chopper Description The chopper structure together with switching aid net-

    works can be seen in Fig. 3 Coil L, is designed to limit di / dt to less than 50 A/ps and capacitor C, for limiting the transistor V,, to less than 200 V at the end of the collector current fall time. The switch to be used in the chopper, for a generator rated 25 kW/50 kHz, is a Darlington whose struc-

    L

    Load hC+ S w i t c h crosses the Curie temperature.) + + Source +-- +-*

    Fig. 1. Topological differences between series and parallel resonance.

    '

    TABLE I DIFFERENCES BETWEEN PARALLEL AND SERIES INVERTER

    Series Inverter

    (+) Simpler structure and, consequently, much cheaper,

    ( + ) control of the unit normally simpler (+) power source from a noncontrolled rectifier ( - ) unloaded operation only possible with a sophisticated control ( - ) no short circuit capability ( - ) power control by frequency shift can give problems in some

    Parallel Inverter

    ( + ) Unloaded operation possible (+) short circuit in the working coil possible (+) high-voltage capacitors are not required (+) transistors conduct only the active current ( - ) logic control of the generator more sophisticated ( - ) power source by a controlled recifier or a chopper ( - ) need of a smoothing choke

    appGcations ( - ) Transistors have to conduct the whole current of the inductor

    ( - ) bigger size

    differences between one configuration and the other. Table I gives the advantages (+ ) and disadvantages ( - ) of both circuits.

    IV. DESCRIPTION OF POWER SECTION Making a balance between the advantages and disadvan-

    tages of the two basic topologies and based on the type of application of those generators, we choose the parallel reso- nant inverter with a full bridge configuration. Fig. 2 shows the block diagram of the developed generator.

    The practical implementation of Ql-Q4 is a switch bidi- rectional in voltage and unidirectional in current (Fig. 1). C represents the compensation capacitor bank and L the heat- ing inductor.

    The generator consists of a complete noncontrolled recti- fied bridge to transform the three-phase current into dc, a chopper to control the delivered power to the load, a starter

    ture is shown in Fig. 4. The transistors used are ISOTOP packaged (Thomson) and water cooled.

    Fig. 5 shows the experimental waveforms of the collector current (lower trace) and collector-emiter voltage (upper trace) for the chopper switch (Fig. 4).

    B. Starter Circuit Description This circuit limits the input current to the inverter when

    inverter resonant frequency and drive frequencies are not the same. This occurs during the generator starting phase as well in case of short circuit in the load.

    The power structure of this starter circuit can be seen in Fig. 6, where the switch Q has the identical structure as the chopper switch (Fig. 4). When the load resonant circuit and inverter frequencies are not the same, the switch Q is cut off, and the R resistor limits the current. When such frequencies are the same, the switch Q is saturated, and there is no current limit, apart from that imposed by the current regula- tion circuit of logic control. The duty of capacitor C2 is to absorb the magnetic energy stored in the smoothing coils.

    This capacitor is designed such that the maximum collec- tor-emitter voltage in the switch Q does not exceed 700 V in the worst case.

    C. Inverter Description This structure is a complete bridge current-fed inverter. In

    Fig. 7, an inverter switch for the 25-kW/5O-kHz generator, together with its switching aid network, is shown. The diode in series with the transistors is necessary to block negative voltage between switch A-B poles. The driving circuit is optocoupled to the logic control due to the inverter structure itself (full-bridge inverter).

    circuit whose duty is to limit current during the starting phase of the generator, or in the case of a short circuit and a v. PRocEss AND SwrrCHINGArDNETWoRKS

    w

    current-fed transistorized bridge inverter, which transforms the dc into a high-frequency current (up to 50 kHz), which powers to the load consisting of a parallel resonant circuit.

    The purpose of the present section is to analyze the behav- ior of the inverter during the conmutation process, when 4 (phase-shift angle between output current and voltage) is

  • 205 DEDE et al.: 25-KW/SO-KHZ GENERATOR FOR INDUCTION HEATING

    u u

    L

    c1 1 1

    UADOI F.A.P. TRANSISTORIZED GENERATOR BLOCK DIAGRAM I G.H. INDUSTRIAL S . k 1yi : I DE : M A I 11/4/90 89.7511

    Fig. 2. Parallel resonant inverter generator block diagram.

    A = 10mV 0= 2 V 0.Ims

    Fig. 4. Chopper switch.

    positive and when 4 is negative, with the aim of choosing the best form of operation of the inverter.

    The theoretical waveforms of the inverter are shown in Fig. 8. VQi is the voltage across the switch Qi, V,, is the output

    voltage, and IMF is the output current of the inverter.

    RETURN Fig. 5. Zc, V,, waveforms for chopper switch. Lower trace: I , (50

    A/div); hpper trace: V,, (200 v/div); time scale: 0, 1 ms/div.

    On the other hand, as

    4 = a r g [ Z ( J w ) ] = (-)arctgQ(w/w,- w o / w )

    where Q is the Q factor of the resonant circuit and

    w0 = 27rf0 = l /SQRT(LC)

  • 206 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO. 3, JUNE 1991

    Fig. 6. Starter circuit.

    U m

    I LLI I m T *

    c > L

    E

    L (c

    I

    i w B I

    Fig. 7. Switch of inverter bridge with switching aid network.

    # (0 (phase Lag) #>0 (phase Lead) VMF I

    V Q 2 ! J q I n v Q 2 , v 7 p v

    f > f, f < f, Fig. 8. Inverter theoretical waveforms.

    results in

    f > f O = * 4 = (-) = =. phaselag f < f , = * 4 = (+) =*phaselead.

    Therefore, to control the inverter with a lagging phase angle, the frequency of the driving pulses Q and Q has to be higher than the resonant frequency of the load, and with a leading phase angle, the frequency has to be lower than the resonant circuit one.

    The sign of the phase-sift angle 4 has great repercussions on the conmutation process of the switches Ql-Q4 (Fig. 2), and consequently, on the kind of switching-aid network to be used.

    A . Conmutation Process with 4 < 0 Let us think that the frequency of the inverter is higher

    than the corresponding of the load. In this case, 4 is nega- tive.

    L

    Fig. 9. Equivalent circuit during commutation when $J < 0.

    I I

    I Q4

    I I I I I I I I I I

    i = i Dl

    . . I 1 I I I 1 I 1

    I I I 1 D1 I

    I I I I 1

    TI- T4 T I Fig. 10. Switching process waveforms when $J < 0.

    The equivalent circuit during the conmutation is given in Fig. 9 (only one mesh is shown).

    At the moment at which the base signal of T1 goes to 0 and the signal of T4 goes to 1, the output capacitor voltage is positive (as is shown in Fig. 9). The switching process waveforms can be seen in Fig. 10. L, represents the parasitic wiring inductance.

    Observe that the turn off of branch 1 does not depend on the turn off of T1 but on the turn on of T4.

    The advantages ( + ) and disadvantages ( - ) of the conmu- tation process with 4 < 0 (f > f,) include the following:

    (+) No conmutation losses at turn off ( - ) conmutation losses at turn on (- ) recovery current from series diode (- ) choice of series diode (-) negative voltage in the series diode (-) a good layout is worsening the problem of the series

    diode recovery (higher di / dt means higher IRRM)

  • DEDE er al.: ZS-KW/SO-KHZ GENERATOR FOR INDUCTION HEATING 207

    L Equivalent circuit during conmutation when $J > 0. Fig. 11.

    ( - ) possible problems in the control logic due to the turn

    ( - ) EM1 and RFI interference problems. To reduce the former problems a turn-on switching-aid

    network that decreases the di ld t of the turn on of T4, is required consequently reducing the turn-off speed of D1. However, magnetics, like saturable inductances, are gener- ally not practical at high frequency.

    B. Conmutation Process with t$ > 0 In this case, the frequency of the inverter must be lower

    than the load resonant frequency. The equivalent circuit (one mesh) during the conmutation is shown in Fig. 11.

    When Q goes to 0 and e goes to 1, the polarity of the voltage in the output capacitor is shown in Fig. 11. Note that the conmutation process is, for this case, controlled by the turn off of T1. Only when T1 is switched off, 04 can begin to conduct. Now, the series diode is cut off by a positive voltage (directly polarized), and therefore, there are no recovery problems in the series diode.

    The switching process waveforms can be seen in Fig. 12. The advantages ( + ) and disadvantages ( - ) of the conmu-

    (+ ) No losses at turn on (+ ) no recovery current in the series diode. This is very

    (- ) losses at turn off ( - ) voltage peak across T1 ( -) layout must be good.

    Making a balance of the conmutation process with t$ > 0 and t$ < 0 it results that for higher frequencies, it is more suitable to work with a leading phase angle (4 > 0). That means that the inverter frequency must be kept lower than the resonant load frequency. In this case, the switching-aid net- works must be turn-off switching-aid networks.

    Fig. 13 shows the V,, and Vce wave forms for the inverter switch.

    Fig. 14 shows the collector current IC and the voltage VQj across on switch Qj.

    off of the series diode

    tation process with t$ > 0 include the following:

    important fact when working at high frequencies.

    VI. DESCRIPTION OF THE CONTROL LOGIC OF THE CONVERTER

    The block diagram of the control circuit for the inverter is shown in Fig. 15.

    Fig.

    Y--Q L -L - I iL I I I

    I I I I I I I I I

    I I I I I

    I I I

    I 1 I I I I

    I I I I I I I

    I I I I

    Fig. 12. Switching process waveforms when 6 > 0.

    A= 10 v B= 2 v 5us

    V

    RETURN

    , 13. V,, and V,, waveforms for the inverter switch. Lower trace: V,, (10 V/div); upper trace: V,, (200 V/div); time scale: 5 psldiv.

  • 208

    S1 S

    IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO. 3, JUNE 1991

    A=O. 1 V 8- 2 V 5us

    COMPENSATOG- L INTEGRATOR - v c o - - MODULE AND

    RETURN Fig. 14. Zc and VQj waveforms. Lower trace: Zc (50 A/div); upper trace:

    Vaj (200 v/div); time scale: 5 ps/div.

    -316

    rlL zERo-cRG v M F N L VMF \ - Qcc

    PWASE-SIUFT - AMjlESIGNUs COMPARATOR DETECTOR MPD

    Fig. 15. Block diagram of the inverter control circuit.

    R-S BIESTABLE

    Let us omit for a moment the time-compensation circuit, which will be explained later, so that we can presume S = S1. The basic circuit consists of a VCO oscillator con- trolled by an integrator. This one is controlled by the signal SIG in such a way that when the phase-shift between the pulses S and V,, is negative (inverter frequency higher than the load resonance frequency), the signal SIG is 1, and a rising slope is generated at the output of the integrator, which makes the VCO frequency decrease.

    On the other hand, if the phase shift between S and VMF is positive (inverter frequency lower than the load resonance frequency), the signal SIG is 0 and generates a decreasing slope at the output of the intregator, consequently increasing the VCO frequency. Obviously, the balance is established when the pulses S and VMF are in phase.

    This would be totally right if there were no delay between S and the real activating pulses of the inverter transistors. In the real circuit, this delay always exists and makes the output current lag behind the output voltage (4 > 0). Obviously, this effect tends to increase as the frequency does.

    The time-compensation circuit works in such a way that the puses S1 (which are the phase-comparator pulses) are

    -Q - Q

    -

    delayed with respect to the driving pulses S . In the steady state, the signal V,, is in phase with the pulses S1. Because the pulses S lead the pulses S1 by AT, with a well adjusted AT, we obtain a compensation of the phase-shift A 4 pro- duced by the wiring, switching time of the transistors, logic delays, etc.

    VII. CONCLUSIONS The paper presents a medium power converter designed

    for high-frequency induction heating applications. The inverter bridge, implemented with bipolar transistors

    in a Darlington configuration, can work at frequencies of up to 50 IrHz. The inverter is controlled by an automatic track- ing circuit, which means that resonant and inverter frequen- cies are almost the same at every moment. The power delivered to the load is regulated by a feedback loop that controls the duty cycle of the input chopper.

    REFERENCES [l] W. E. Frank and C. F. Der, Solid state RF generators for induction

    heating applications, in Proc. ZEEE-ZAS, 1982, pp. 939-944. [2] D. Naunapper and H. J. Eckhardt, Power MOSFETS, thyristors and

  • DEDE et al.: 25-KW/5O-KHZ GENERATOR FOR INDUCTION HEATING 209

    transmitting tubes in converters for hardening, in Proc. PCIM Conf., 1989.

    [3] J. Nuns, Transistors are replacing valves in induction heating gener- ators,in Proc. EPE Conf., 1987.

    141 L. Malesani and P. Tenti, Medium frequency GTO inverter for induction heating applications, in Proc. EPE Conf., 1987.

    [5] D. Tebb, An induckion heating power supply using high voltage MOSFETS, in Proc. PCIMConf., 1987.

    [6] F. Z. Peng, H. Akagi and A. Nabae, High current source inverters using SiTh for induction heating applications, in Proc. IEEE-IAS Conf., 1987.

    [7] L. Hobson, D. Tebb and C. Pudney, A current fed inverter with inherent short circuit protection and suppression ringing, Znt. J . Electron. vol. 60, no. 4, 1986.

    [8] L. Hobson and D. Tebb, Transistorized power supplies for induc- tion heating, Int. J . Electron., vol. 59, no. 5, 1985.

    [9] J. M. Peter, The power transistor and its environment, Thomson Semicond., 1979.

    [lo] -, Transistors and diodes in power processing, Thomson Semicond.. 1985.