A High Power Density Drivetrain-IntegratedElectric Vehicle Charger

Usama Anwar, Hyeokjin Kim, Hua Chen, Robert Erickson,Dragan Maksimovic and Khurram K. Afridi

Colorado Power Electronics CenterDepartment of Electrical, Computer and Energy Engineering

University of Colorado, Boulder, CO 80309-425, USAEmail: usama.anwar@colorado.edu

Abstract—This paper presents a new architecture for an iso-lated level 2 on-board electric vehicle (EV) battery charger whichis integrated with the EV’s drivetrain dc-dc boost converter. Theproposed charger leverages many of the existing stages of a highlyefficient and power-dense composite-architecture-based drive-train boost converter. This composite boost converter comprisesa buck, a boost and a dc transformer (DCX) stage. In selectingthe proposed charging architecture, four alternative approachesto drivetrain integration are identified, explored and comparedquantitatively in terms of added weight and charging losses. Outof the considered approaches, the proposed charging architectureprovides an effective tradeoff between additional weight andcharging losses. This drivetrain-integrated charger adds only abridgeless-boost-based power factor correction (PFC) ac-dc stage,plus an H-bridge and a single winding to the composite boostconverter, to achieve high-power on-board charging functionalitywithout substantial additional weight. A 6.6 kW prototype of theproposed charger has been designed and its PFC stage builtand tested. The PFC stage uses a digital implementation of acurrent sensor-less control strategy and employs effective waysof mitigating zero crossing distortion. The proposed chargerarchitecture reduces the additional weight required for the on-board charging functionality, while achieving greater than 97%peak efficiency for the added charger module PFC stage.

Index terms— drivetrain integrated charger, EV on-boardcharger, battery charger, PFC boost DCM, PFC zero crossingdistortion, power regulation stage.

I. INTRODUCTION

A high power on-board charger provides convenience,reduces range-anxiety and is considered vital to acceleratethe adoption of electric vehicles (EVs) [1]. However, theadditional weight of a high power on-board charger canreduce the range of the vehicle and its volume can imposeunfavorable design constraints. Hence, reducing the weight ofon-board chargers is an important goal for vehicle manufac-turers. Different approaches to implementing on-board electricvehicle chargers have been proposed [2], [3]. However, inall of these approaches, the on-board charger is considereda separate power electronic converter. Some attempts havealso been made to integrate on-board charging capabilitywith the electric drivetrain by leveraging the windings of theelectric motor [1], [5], [6]. However, this approach introduces

complexity into the design of the electric motor and can impactits cost and reliability. Hence, there is a need to explorealternative ways of integrating on-board charging functionalitythat minimize added weight and maintain high efficiency.

Recently a new drivetrain architecture for electric vehiclesthat utilizes a composite boost dc-dc converter between thevehicle battery and the motor drive has been introduced [7].The proposed composite boost dc-dc converter comprises dis-similar converter modules combined into a power conversionsystem with superior performance in terms of efficiency andpower density compared to the traditional drivetrain boostconverter. The architecture of the composite boost dc-dc con-verter is shown in Fig. 1. This modular approach to boost thebattery voltage provides an opportunity to leverage some of itspower conversion stages as part of the charging function. Suchcharger integration can result in minimal additional weight,while providing high power on-board charging capability,increasing the on-board charger power density. This paperidentifies, explores and quantitatively compares alternativearchitectures for integrating the charging function with thecomposite boost converter. The architecture among these thatprovides an effective trade-off between added weight andcharging losses is selected for detailed design and testing.This drivetrain-integrated charger only adds a bridgeless boostbased power factor correction (PFC) ac-dc stage, an H-bridgeand a single winding to the composite boost dc-dc converter.Hence, it significantly reduces the additional weight requiredfor the on-board charging functionality. A 6.6 kW prototypeof this proposed charger has been designed and its PFC stagetested. It achieves greater than 97% peak efficiency for theadded charger module PFC stage.

This paper is organized as follows. A brief review of thecomposite boost dc-dc converter is presented in Section II.Multiple approaches to integrate the charging functionalitywith the composite dc-dc converter are discussed in Sec-tion III, with a quantitative comparison presented in Sec-tion IV. Based upon the quantitative results, a drive-train-integrated charger architecture is identified and discussed indetail in Section V. Section VI presents prototype design and

Figure 1: Architecture of the dc-dc composite boost converter used in an electric vehicle drivetrain between the vehicle battery and the motor drive [7]. Theconverter is comprised of multiple power modules: a buck module, a boost module and a dc transformer (DCX) module.

simulation and experimental results of the proposed chargerarchitecture. Finally, Section VII summarizes key results andconclusions.

II. DC-DC COMPOSITE BOOST CONVERTERARCHITECTURE

The architecture of the composite boost dc-dc converter [7]is shown in Fig. 1. This architecture comprises of a buckmodule, a boost module and a dual active bridge operatedas a dc transformer (DCX). The composite boost converterachieves a high voltage gain by connecting the outputs ofthe boost and the DCX modules in series. The buck moduleforms a cascade with the DCX stage, and its input is parallel-connected to the input of the boost module. This architectureallows the composite boost converter to achieve high outputvoltages with relatively low breakdown voltage switches hav-ing low on-resistance and fast switching speed. Each moduleof the composite converter processes only a fraction of the totalpower processed by the converter. This approach enables effi-ciency and power density optimization of the overall converterby utilizing a control strategy that maximizes the amount ofdirect power processed by the converter across a wide range ofoperating conditions. These factors enable the composite boost

converter to achieve greater efficiency and power density thana traditional boost converter. In addition to these benefits, themodular nature of the composite boost converter opens up thepossibility of integrating the charging functionality within thedrivetrain, as discussed in detail in the next section.

III. ALTERNATIVE DRIVETRAIN-INTEGRATED ELECTRICVEHICLE CHARGER ARCHITECTURES

The charging functionality can be integrated with thecomposite boost architecture in multiple ways. Four of themost promising approaches to integrate an on-board chargerinto the composite boost converter are shown in Fig. 2.Figure 2(a) shows architecture A, a bridgeless boost stagefollowed by a phase-shifted full-bridge isolated dc-dc con-verter. The bridgeless boost stage is operated as a powerfactor correction (PFC) rectifier, and the phase-shifted full-bridge isolated dc-dc converter provides isolation and alsoregulates the battery charging current. Although this archi-tecture involves only two cascaded power stages, resulting inrelatively low losses, the added weight of this architecture isthe highest among the four considered charging architectures.This is due to its relatively weak integration with the compositeboost converter, as it only utilizes the composite converter’s

(a) Architecture A (b) Architecture B

(c) Architecture C (d) Architecture D

Figure 2: Four proposed architectures suitable for integration of charging functionality with the composite boost converter of Fig. 1.

Figure 3: Drivetrain integrated charger architecture comprising a bridgeless boost converter, followed by a dual active bridge (DAB) and a buck converter.

boost stage filter. Figure 2(b) shows architecture B, whichcomprises a diode rectifier followed by an isolated full-bridgeboost stage to achieve power factor correction and isolation.The full-bridge boost converter integrates into the existingdrivetrain by adding an additional winding to the transformerof the DCX and utilizing its secondary side H-bridge forrectification. These stages are then followed by the drivetrainbuck converter that regulates the battery current. While thisarchitecture provides strong integration with the compositeboost converter, increased losses in the isolated boost stagedegrades its efficiency. Figure 2(c) shows architecture C,which comprises a bridgeless boost converter followed by adual active bridge (DAB) converter and the drivetrain buckconverter. The bridgeless boost converter at the front end isoperated as a PFC stage. The DAB is used to provide isolationand the drivetrain buck converter is then used to regulate thebattery current. The DAB is implemented by adding a primaryside H-bridge and an additional transformer to the existingdrivetrain DCX. The secondary side H-bridge of the drivetrainDCX is reused in the charging operation. This architecture canbe further integrated with the composite boost converter byintegrating the two DAB transformers. This can be achievedby utilizing a three winding transformer in the DAB stage, asshown in architecture D of Fig. 2(d). Architecture D achievesstrong integration by effectively utilizing the existing drivetrainstages and appears highly promising.

IV. QUANTITATIVE COMPARISON OF THE ALTERNATIVEDRIVETRAIN-INTEGRATED CHARGER ARCHITECTURES

In order to quantitatively compare the proposed integratedcharger architectures shown in Fig. 2, loss models are devel-oped to estimate their losses over a line cycle. These lossmodels include switch conduction and switching losses, andinductor and transformer core and winding losses. In addition,models to compute the weight of each architecture are alsodeveloped. The weight models consider the weight of themagnetic and capacitive components, as well as the heat

sinks, since they constitute the largest fraction of the converterweight. The computed weights and losses from these modelsfor the four considered architectures are shown in Table I.It can be observed that architecture A has the least lossesbut contributes the most additional weight. On the other handarchitecture B adds substantially less weight but has highlosses. Note that architecture B adds least components but notthe least weight due to increased size of the heat sink. Sincein an EV application both weight and efficiency are importantconsiderations, architecture D appears to provide an excellenttradeoff between these attributes as it effectively leveragesthe existing modules of the composite boost converter. Thusarchitecture D is selected for detailed design, fabrication andtesting.

Table I: Comparison of proposed integrated charger architec-tures in terms of added weight and full-power losses for a6.6 kW on-board EV charger.

Architecture Added Weight [kg] Losses [W]A 2.67 238B 1.97 322C 2.07 268D 1.90 265

V. DRIVETRAIN-INTEGRATED CHARGER DESIGN ANDCONTROL

The schematic of the proposed drivetrain-integrated chargeris shown in Fig. 3. The first stage of the architecture isa bridgeless boost PFC. The second stage is an H-bridgeforming a dual-active bridge with the existing H-bridge ofthe composite converter’s DCX stage. This is then followedby the drivetrain buck converter to regulate the battery power.

A. AC-DC Bridgeless Boost Converter

The bridgeless boost topology is well suited for high effi-ciency ac-dc power conversion since it eliminates the passiverectifier bridge used in traditional PFC rectifiers, resulting in

reduced component count and decreased conduction losses [4].An additional advantage of this topology compared to a totem-pole bridgeless boost is that the active switches are referencedto the same node, simplifying gate driver requirements.

A detailed design of PFC stage of the proposed drivetrain-integrated charger is conducted and optimizations were per-formed to minimize losses over the line cycle. More detailson the optimizations are presented in Section VI. As a resultof optimizations, discontinuous conduction mode (DCM) ofoperation of the circuit was selected. DCM operation resultsin reduced inductor size and switching losses.

Operating the bridgeless boost converter in DCM providesadditional benefits related to control implementation. In DCMoperation, PFC functionality can be implemented withoutsensing the inductor current which typically requires a high-bandwidth sensor. Thus, only the input and output voltages ofthe converter need to be sensed to implement the control. InDCM, the average inductor current can be expressed as [8]:

〈il〉Ts= (

Ts2L

d2

1− vinvbus

)vin. (1)

Here 〈il〉Tsis the average inductor current, vin is the

input line voltage, vbus is the dc bus voltage, Ts is theswitching period, d is the duty ratio and L is the bridgelessboost inductance. The control objective for achieving PFCfunctionality is:

〈il〉Ts=vinRe

. (2)

It can be observed from (1) that the average input (inductor)current can be made proportional to input voltage, as requiredto achieve the control objective ( 2), by adjusting the dutyratio such that the term in parenthesis becomes constant.

Ts2L

d2

1− vinvbus

=1

Re. (3)

This can be manipulated to yield the following control law[9], [10] :

d =

√2L

ReTs(1− vin

vout). (4)

Here Re is the emulated input resistance of the converter.The value of this emulated resistance can be adjusted by a PIcompensated outer voltage loop.

For the bridgeless boost converter, since the converter hastwo active switches, the control strategy can be implementedas:

d1 =

d, vin > 0

1, vin < 0(5)

d2 =

1, vin > 0

d, vin < 0(6)

Here d1 and d2 correspond to duty cycles of the twoactive switches of the bridgeless boost converter. The con-

Figure 4: Power factor correction functionality (PFC) implemented usingcombined hybrid feedforward-feedback control.

trol architecture falls in the category of hybrid feedforwardcontrol architecture embedded inside a feedback loop [11]. Adigital implementation of this control strategy using a micro-controller is selected for the PFC stage. The micro-controllersenses the input voltage and output voltage of the converterand processes them as given by (5) and (6) to modulate theduty ratio of the converter in each switching interval. Ascan be observed from the duty ratio modulation equation,the operations of square root, division, multiplication andsubtraction are required to implement the control. This canbe easily performed in a modern micro-controller.

The converter outer voltage loop is designed to adjust theemulated input resistance of the converter. The outer voltageloop is designed, as is a traditional practice, by assuming thatthe input current control loop works ideally at low frequenciesand the ideal rectifier model adequately represents the low-frequency system behavior [8]. When the converter outervoltage loop is designed properly, it regulates the converteroutput voltage vbus. Any tolerance in the converter inductancevalue, which may appear to affect the performance of theinner current control loop (as the converter duty ratio givenby (4) depends on the inductance value), is automaticallycorrected by the outer voltage loop. Since the output of theouter voltage loop compensator appears as a multiplicativefactor with the circuit parameters in (4), inaccuracy in anyother circuit parameters also get corrected by the action ofouter feedback loop.

Zero crossing distortion is a common problem in PFCconverters [12]. The fundamental reason that the distortionappears around input voltage zero crossing in the operationof the converter operating in DCM is because of inaccurateinput voltage sensing and delays in the control loop and thegate driver. Figure 5(a) shows when the actual input voltagecrosses zero volts and Fig. 5(b) shows that the sensor sensesinput voltage zero crossing before it actually happens. If thecontroller is implemented as given by (5) and (6), then dueto inaccurate input voltage sensing, the controller causes twolegs of the H-bridge to switch their operation before it shouldactually happen. This causes the two switches of the converterto remain on for interval II as marked in Fig. 5. Notice that

Figure 5: Sensor error in sensing input voltage zero crossing: (a) Actual zero-crossing; (b) zero crossing sensed when the input voltage sensor detects thezero crossing before it actually happens; (c) zero crossing sensed when theinput voltage sensor detects the zero crossing after it actually happens; and(d) proposed solution of overlapping the switching time of the two half-bridgelegs around input-voltage zero crossing solves the problem.

even though the controller commands switch Q2 to switchin interval II, the transistor remains on due to conduction ofbody diode, as the current only changes polarity when theinput voltage changes polarity in DCM. This causes inputvoltage to appear across the inductor for the time intervalII. Since, small inductors are normally employed in DCMconverters, even a small input voltage around zero crossingcan cause spikes the in inductor current. In order to mitigatethis issue, both transistors of the bridgeless boost converterare switched on and off at the same time around the zerocrossing. This prevents a large build-up in the inductor current,as the current is periodically interrupted by the switching ofthe two transistors and enables smooth commutation of thecurrent from one leg of the converter to the other, as shown inFig. 5(d). Notice that in the operation of the bridgeless boostconverter, one of the two switches of the converter remainson for half line cycle. Thus normal operation of the converterremains unchanged by overlapping the switching time of thetwo switches around input voltage zero-crossing. This zero-crossing mitigation can be expressed as:

d1 =

d, vin > −δV

1, vin < −δV(7)

d2 =

1, vin > δV

d, vin < δV.(8)

Here δV is the estimated inaccuracy in the input voltagesensing. The two transistors are switched from −δV to δV ,allowing current to naturally commute between the two legsof the half bridge. This sensing inaccuracy when accounted inthe micro-controller can cause switching of the two legs of theconverter to overlap around the input voltage zero crossing,preventing inductor current spikes. Furthermore, the controlstrategy has benefits of simple implementation and improvedconverter performance. In case when the input voltage sensorsenses the voltage zero crossing after it actually happens,as shown in Fig. 5(c), the effects observed are similar andthe proposed control strategy solves the problem. Finally, itis interesting to note that the current commutation is callednatural as the commutation happens when the control schemeis implemented without any effort on the part of controller todetect the input voltage zero crossing accurately.

B. Dual-Active Bridge Converter

The second stage of the charger is a dual-active bridge(DAB) converter which is integrated with the drivetrain usinga three winding transformer, implemented by simply addingone extra winding to the existing drivetrain DCX. One side ofthe transformer, which corresponds to the output of the DCXis turned off while the charger is charging the battery. Thepower transfer takes place from the grid side DAB full bridgeto the battery side DAB full bridge. For charging purposes, theDAB converter provides isolation and regulates the voltage atits output.

The control of the DAB converter is done by switchingthe primary and secondary H-bridges at approximately 50%duty cycle and controlling the phase shift between two H-bridges [13]. The control is based on sensing the converteroutput voltage, comparing it with the reference and passingthe error through compensator to generate the converter phaseshift. This control loop is designed to have much lowerbandwidth than the twice the line frequency. The twice linefrequency ripple present on the grid side of the DAB isbuffered by both the grid side and the battery side capacitors(C1 and C2) in Fig. 3.

C. Power Regulation Boost Converter

The final stage of the charger is the drivetrain bi-directionalbuck, which functions as a boost converter during the chargingoperation. The converter regulates the current flowing intothe battery. To realize the controller on this boost converter,inductor current is sensed and used in a standard feedbackarchitecture to generate the duty ratio command. Since theinductor is present at the input of the converter, current controlis not done in a standard fashion by sensing the battery currentand comparing it with the reference current. Instead, as thecontrol objective is to regulate the battery power, the controlleris synthesized to make the converter behave as a power sink

at the input port [8]. Since in steady state the converter has nonet energy storage, the input power comes out of the converteroutput port and the output port behaves as a power source,which charges the battery at constant power. The schematic,the large signal model and the control architecture of thispower regulating boost stage is shown in Fig. 6.

Figure 6: Boost converter acting as a power regulation stage to regulate thebattery power: (a) Boost converter topology (b) Large signal model and (c)Controller implemented to achieve input power sink characteristics.

To make the converter act as a power sink at the input port,the input voltage of the boost converter is sensed and used togenerate the converter current reference. The converter currentreference can be expressed as:

iref =Pref

vbst(9)

The converter feedback loop is designed to achieve inputcurrent regulation of the converter, which in case of idealoperation of current control loop can be expressed as:

〈ibst〉Ts= iref (10)

Thus in steady state, the converter input power can beexpressed as:

Pin = vbstiref = Pref (11)

Ignoring the losses in the converter, as they are smallcompared to the input power, the converter output power canbe expressed as:

Pout = Pref (12)

When this control is implemented, the output power of theconverter can be controlled without directly sensing the batterycurrent. It can be noted here that the converter input voltagehas twice line frequency ripple. The ideal power regulationstage will get rid of this twice line frequency ripple andcharge the battery with constant current. Thus bandwidth of theinner current loop is designed to be much larger than twiceline frequency to track the ripple in input current referenceappearing because of ripple in input voltage of the converter.

VI. PROTOTYPE DESIGN AND EXPERIMENTAL RESULTS

A 6.6 kW electric vehicle charger integrated with a 30 kWcomposite boost dc-dc converter has been designed and itsPFC stage is tested.

A detailed design of the PFC stage is conducted. Thedesign variations considered span a large design space, in-cluding continuous and discontinuous conduction modes ofoperation, switching frequencies spanning 20 kHz to 80 kHz,wide range of component values and different componenttypes. For example, within each system level design iteration,optimal inductors are designed for the PFC boost stage byiterating over various core materials, core geometries andwinding parameters. The optimization is performed using anexhaustive search based numerical approach that computes theoverall converter weight and line-cycle average losses for eachconsidered design. The final design is selected by searching forsolutions that minimize losses while staying within specifiedweight limits. As a result of this optimization, a switchingfrequency of 20 kHz and discontinuous conduction mode(DCM) for the bridgeless boost converter are selected for theprototype design. The design details of the optimized inductorfor the bridgeless boost stage are given in Table II. The activeswitches in the bridgeless boost stage are implemented usingsilicon super-junction MOSFETs while the rectifier diodesare realized using silicon-carbide Schottky diodes. The detailsof the components used in bridgeless boost converter arepresented in Table II.

Table II: Design Details of the Bridgeless Boost Stage

MOSFETs IPW65R041CFD2650 V, 68.5 A

Silicon Super Junction FETDiodes C5D50065D SiCOutput capacitor 40 µF, 500 VCoupled Inductor 23 µH

Amorphous alloy10 turns of 1000-strands

38 AWG litz wireMicrocontroller TI-TMS320F28069

32 Bit CPU, 90 MHz

The PFC stage was tested at full power. Figures 7 and 8show waveforms of the bridgeless boost stage operating at apower level of 950 W and 6.7 kW, respectively. Lag in inputcurrent with respect to input voltage is due to capacitive filtersemployed at the converter input to filter out switching currentfrom line current. It can be seen that the input current is fairly

sinusoidal. The peak-to-peak dc bus voltage ripple is designedto be 30%, as can be seen from Fig. 8. Figure 9 shows thewaveforms for the switch-node voltage, the inductor currentand the MOSFETs gate-to-source voltage.

Figure 7: Input voltage vin, input current iin and output voltage voutwaveforms of PFC converter with Vin = 80 Vrms , Iin = 11.8 Arms andVOUT = 144 V .

Figure 8: Input voltage vin, input current iin and output voltage voutwaveforms of PFC converter with Vin = 240 Vrms , Iin = 28 Arms andVOUT = 378 V .

Figure 9: Switching waveforms of the bridgeless boost ac-dc converter.

The effectiveness of the zero-crossing distortion mitigationstrategy is shown in Fig. 10. As can be seen in Fig. 10 theline current transitions across the line zero crossing with novisible distortion. Both active switches of the converter switcharound zero crossing to allow for seemless zero crossing.

The efficiency of the PFC stage across a more than 10:1output power range with an input line voltage of 240 Vrmshas been measured and is shown in Fig. 11. It can be seenthat the PFC stage achieves a peak efficiency of 97.7% andmaintains efficiencies above 97% across a 6:1 output powerrange. The worst case measured efficiency is 96.3%.

Simulated input and output voltage waveforms of the dual-active bridge converter are shown in Fig. 12. Slow outputvoltage (vbst) regulating loop allows twice line frequencyripple present at the input energy buffering capacitor of theDAB converter to appear at the output. Thus the capacitorconnected at the output of DAB converter plays a role in the

Figure 10: AC-DC bridgeless boost converter behavior around input voltagezero crossing. The waveforms depict natural commutation of inductor currentbetween legs of bridgeless boost converter by the improved switching schemeof the converter.

Figure 11: Measured efficiency of the power factor correction stage of thedrivetrain integrated electric vehicle charger.

energy buffering as well. Inductor current waveforms of theDAB converter are shown in Fig. 13 when the converter isoperated at a switching frequency of 120 kHz. The waveformsshow that the converter achieves ZVS on both primary andsecondary H-bridges.

Figure 12: Dual-active bridge converter input (vbus) and output (vbst) voltagewaveforms with Vbus = 375 Vnom and Vbst = 200 Vnom.

Simulated input and output voltage and current waveformsof the boost stage are shown in Fig. 14. It can be observedthat the input voltage and input current of the boost converterhas twice line frequency ripple, while the output current isalmost constant. The fast current regulating loop of the boostconverter allows the converter input port to act as a constantpower sink which appear at the converter output port as aconstant power source, thus eliminating twice line frequencycurrent ripple from the output current.

VII. CONCLUSIONS

A new architecture is presented for an isolated level 2 on-board electric vehicle (EV) battery charger which is integrated

Figure 13: Dual-active bridge converter inductor current waveforms. Thewaveforms show that the converter achieves zero voltage switching on bothprimary and secondary H-bridges and switching frequency fsw = 120 kHz.

Figure 14: Boost converter input voltage (vbst), output voltage (vbatt),average input current (〈ibst〉Ts ) and average output current (〈ibatt〉Ts )waveforms with Vbst = 375 Vnom, Vbatt = 200 V , Ibst = 33 Anom andIbatt = 36.4 Anom.

with the EV’s drivetrain dc-dc boost converter. The proposedcharger leverages many of the existing stages of a highly effi-cient and power dense composite-architecture based drivetrainboost converter. This composite boost converter comprisesa buck, a boost and a dc transformer (DCX) stage andprovides opportunity to integrate charging functionality withthe drivetrain. To determine effective means of integrating thecharging functionality with the composite dc-dc converter, fouralternative approaches to drivetrain integration are identified,explored and compared quantitatively in terms of added weightand charging losses. Out of the considered approaches, theproposed charging architecture provides an effective trade offbetween additional weight and charging losses. This drivetrain-integrated charger adds only a bridgeless boost based powerfactor correction (PFC) ac-dc stage, an H-bridge and a singlewinding to the composite boost converter to achieve high-power on-board charging functionality without substantialadditional weight.

A 6.6 kW prototype of the proposed charger’s PFC has beendesigned, built and tested. The front-end power factor correc-tion stage of the proposed architecture uses a current sensor-less control strategy and employs effective ways of mitigatingzero crossing distortion in the PFC stage. The current-sensor-less control strategy implemented on the PFC stage achievesgood performance and simpler controller implementation. Adigital controller implementation is selected to realize the

control on the converter. Furthermore, effects of zero crossingdistortion observed in PFC boost converter operating in DCMare analyzed and a simple controller is presented that canmitigate the problem and allows current to naturally commutebetween the two legs of the bridgeless boost converter.

Experimental results of the proposed charger’s PFC stageare also presented. The proposed charger architecture reducesthe additional weight required for the on-board chargingfunctionality, while achieving greater than 97% peak efficiencyfor the added PFC stage.

ACKNOWLEDGEMENT

The authors wish to acknowledge the financial supportreceived from the Office of Energy Efficiency and RenewableEnergy (EERE), U.S. Department of Energy, under AwardNumber DE-EE0006921.

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