a new family of zero-voltage-transition nonisolated...

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 3, MARCH 2016 1519

A New Family of Zero-Voltage-TransitionNonisolated Bidirectional Converters With

Simple Auxiliary CircuitMohammad Reza Mohammadi and Hosein Farzanehfard

Abstract—In this paper, a new family of zero-voltage-transition (ZVT) bidirectional converters are introduced.In the proposed converters, soft-switching condition forall semiconductor elements is provided regardless of thepower flow direction and without any extra voltage and cur-rent stress on the main switches. The auxiliary circuit iscomposed of a coupled inductor with the converter maininductor and two auxiliary switches. The auxiliary switchesbenefit from significantly reduced voltage stress and with-out requiring floating gate drive circuit. Also, by apply-ing the synchronous rectification to the auxiliary switchesbody diodes, conduction losses of the auxiliary circuit arereduced. In the auxiliary circuit, the leakage inductor isused as the resonant inductor and all the magnetic compo-nents are implemented on a single core which has resultedin significant reduction of the converter volume. In theproposed converters, the reverse recovery losses of theconverter-rectifying diodes are completely eliminated andhence, using the low-speed body diode of the power switchas the converter-rectifying diode is feasible. The theoreti-cal analysis for a bidirectional buck and boost converter ispresented in detail and the validity of the theoretical anal-ysis is justified using the experimental results of a 250-Wprototype converter.

Index Terms—Bidirectional dc–dc converter (BDC),soft-switching technique, zero-voltage switching (ZVS),zero-voltage transition (ZVT).

I. INTRODUCTION

I N RECENT years, bidirectional dc–dc converters (BDCs)have received major attention due to increasing growth of

systems in which energy recovery or energy storage systems arerequired. BDCs are commonly used in a wide variety of appli-cations such as hybrid/electric vehicles (HEVs/EVs) [1], [2],uninterruptable power supplies (UPSs) [3], photovoltaic andfuel cell power systems [4], [5], and dual-voltage automotivesystems [6]. The overall roles of BDCs in such applications aremanaging power flow, converting voltage level of the energystorage devices, and maintaining energy storage devices healthby controlling the charge and discharge current of the energystorage devices [7]. Various BDCs can be divided into non-isolated and isolated types. In applications which isolation

Manuscript received June 22, 2015; revised September 23, 2015;accepted October 19, 2015. Date of publication November 11, 2015;date of current version February 8, 2016.

The authors are with the Department of Electrical and ComputerEngineering, Isfahan University of Technology, Isfahan 84156-83111,Iran (e-mail: [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2015.2498907

and high-voltage ratio are not essential, nonisolated BDCs arealways applied due to their simple structure and control scheme[1]–[7].

As with other dc–dc converters, in order to achieve highpower density in BDCs, high operating frequency is desirable.High switching frequency can significantly reduce the con-verter reactive components size [8]. However, in hard switch-ing converters, increasing switching frequency leads to higherswitching losses and electromagnetic interference (EMI). Inhard switching BDCs, the rectifying diode reverse recoveryis the major cause of the mentioned problems since the low-speed antiparallel body diode of the power switch is used asthe converter-rectifying diode [9]. This shortcoming limits thedesired high-frequency power conversion of BDC. To solvethese problems, soft-switching techniques are developed forthe BDCs. The task is especially challenging in the BDCs,as soft switching must be ensured for the both forward andreverse operating modes [10]. Based on the soft-switching tech-niques applied to the existing soft-switching BDCs, they can beroughly classified as follows.

1) The conventional method to achieve soft-switching con-dition in BDCs is to make the converter main inductorcurrent flow in the negative direction. This negative cur-rent is used to discharge the snubber capacitor and so,zero-voltage switching (ZVS) is achieved with no auxil-iary elements [11]–[15]. Besides, the diode reverse recov-ery losses are completely eliminated. However, using thismethod, the converter efficiency is greatly reduced atlight loads due to the large constant peak-to-peak currentswing of the main inductor and almost constant conduc-tion losses [11]. Besides, the converter suffers from highturn-off losses and high current ripple at the input voltagesource. In order to minimize the circulating current for awide converter operating region, nonlinear inductors [11]or variable frequency control [12]–[14] is adopted whichincreases the converter complexity and cost. Besides, toreduce the input current ripple, this method is generallyused in multiphase interleaved BDCs [12]–[15] whichsuffer from high component count.

2) In the converters proposed in [16] and [17], ZVS con-dition and reduced diode reverse recovery losses areachieved by applying passive elements. In contrast tothe previous method, the main inductor can be designedas a conventional bidirectional converter which operatesin continuous conduction mode (CCM). Thus, the cur-rent ripple of the input voltage source would be reduced

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1520 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 3, MARCH 2016

or even be pure dc-like in [17]. However, the auxiliarycircuit imposes high circulating current in the converterwhich reduces the efficiency especially at light loads[17]; otherwise, a complex variable frequency controlmethod should be adopted [16]. Another soft-switchedBDC is proposed in [18] where two auxiliary switchesare used. Since, in each converter-operating mode (for-ward or reverse mode), one of the auxiliary switches isalways on and the other is always off, it can be assumedthat the soft-switching condition is achieved using pas-sive elements. However, this converter still suffers fromthe same drawbacks of the converters in [16] and [17].

3) In [19]–[21], active clamping technique using an aux-iliary switch is applied for the BDCs to provide softswitching. In this method, the conduction losses of theconverter main elements are minimized and the conven-tional fixed-frequency PWM controller could be applied.However, the auxiliary circuit of [19] still suffers fromcirculating current since the auxiliary switch need to con-duct in almost the whole converter-operating modes. In[20] and [21], the auxiliary circuit-circulating current isminimized by the reduction of auxiliary switch conduc-tion time. However, when the auxiliary switch is off, theauxiliary resonance inductors are placed in series withthe snubber capacitors resulting in undesirable oscillationwhich reduces efficiency and increases the voltage stressof the semiconductor elements. Besides, at the auxiliaryswitch turn-on instant, the remained charge of the reso-nance capacitor discharges through the auxiliary switch.In [22], the same soft-switching cell of [20] and [21] isemployed for a three-level BDC which suffers from thesame drawbacks.

4) Another soft-switching technique which is widely appliedto BDCs is zero-voltage-transition (ZVT) technique.These converters benefit from inherent features such aslow circulating current, retaining PWM operation, andwide load soft-switching range [23]. The ZVT BDCsproposed in [24] and [25] are basically the ZVT cellintroduced previously in [23]. The main drawback of theconverters in [24] and [25] is hard switching of the aux-iliary switches at turn-off instant. Besides, the voltagestress of the auxiliary switches is at the same voltagelevel as the high voltage side. This results in high capaci-tive turn-on losses and conduction losses of the auxiliaryswitches. In [26] and [27], coupled inductors are added toprovide soft-switching of the auxiliary switches at turn-off. In addition, by using the leakage inductor as theresonant inductor, all the magnetic components can beimplemented on a single core. However, in these convert-ers, the auxiliary switches voltage stress exceeds 120% ofthe high voltage side. Also, the auxiliary switches must beunidirectional which is typically implemented by placingadditional diode in series with the power switch resultingin additional components and conduction losses. Besides,in all the converters presented in [24]–[27], one of theauxiliary switches needs floating gate drive since thesource terminal of an auxiliary switch is not in commonwith the input voltage source ground. Another ZVT BDC

is proposed in [28] where the soft-switching condition isprovided for the auxiliary switches at both turn-off andturn-on instants with the same auxiliary components in[24] and [25]. Besides, the voltage stress of the auxiliaryswitches is reduced to the voltage difference of the high-voltage side and low-voltage side in boost mode or tothe low-voltage side in buck mode which is significantlylower than the converters in [24]–[27]. However, the maindrawback of this converter is that the soft-switching con-dition of the main switches is lost at operating duty cycleslower than 0.5 [29]. This causes the soft-switching condi-tion to be lost in one of the converter operational modes(boost or buck modes), since in the bidirectional con-verters, the duty cycles of the boost and buck modes arecomplement of each other. To solve this problem, vari-ous approaches such as using the diode reverse recoverycurrent [29], applying the idea of synchronous rectifiersto store more energy in the resonance inductor [30], [31],and adding an auxiliary dc supply by means of insertingan additional capacitor [32] are proposed. However, allthe converters in [29]–[32] require auxiliary switches withfloating gate drive. Additionally, in [32], the auxiliary cir-cuit is applied two times in a switching cycle, resulting inmore complex control circuit and losses.

In this paper, a new family of nonisolated ZVT BDCs isintroduced. In the proposed converters, soft-switching condi-tion for all semiconductor elements is provided without anyextra voltage and current stresses on the main switches. In com-parison with the previously proposed ZVT BDCs, the voltagestress of the auxiliary switches are reduced significantly and so,it is feasible to select switches with low voltage rating and lowon-state resistance (RDS(on)) as the auxiliary switches. As aresult, both the capacitive turn-on losses and conduction lossesof the auxiliary switches are reduced. In addition, by applyingthe synchronous rectification to the auxiliary switches, conduc-tion losses of the auxiliary switches body diodes are reducedsignificantly. In the proposed converters, the auxiliary circuitbenefits from not requiring floating gate drive circuit and anyadditional diodes. Also, by using the leakage inductor as theresonance inductor, all the converter inductors could be imple-mented on a single core. By applying the idea of synchronousrectifiers to store more energy in the resonance inductor [30],[31], [33], the diode reverse recovery losses are completelyeliminated and soft-switching condition is provided for thewhole converter operating region. Furthermore, the auxiliarycircuit is applied only once in a switching cycle.

The analysis and operation of the proposed ZVT bidirec-tional buck and boost converters are described in Section II.Design considerations are presented in Section III. To con-firm the theoretical analysis, experimental results are shown inSection IV. Other converter family members of the proposedZVT BDCs are introduced in Section V. Finally, the drawnconclusions are presented in Section VI.

II. CIRCUIT DESCRIPTION AND OPERATION

As shown in Fig. 1(a), the proposed converter is composedof two main switches S1 and S2, two auxiliary switches Sa1

MOHAMMADI AND FARZANEHFARD: NEW FAMILY OF ZVT NONISOLATED BIDIRECTIONAL CONVERTERS 1521

Fig. 1. Proposed ZVT bidirectional buck and boost converter and itsequivalent circuit. (a) Proposed converter. (b) Equivalent circuit.

and Sa2, a snubber capacitor CS , and two coupled inductors L1

and L2 with turns ratio of n(L2 = n2L1

). As seen, the source

terminals of both auxiliary switches are in common with theinput voltage ground and thus, no floating gate drive is requiredfor the auxiliary circuit. The coupled inductors can be modeledas a combination of a magnetizing inductance (LM ), an idealtransformer with corresponding turns ratio (n), and a leakageinductance (Llk). The equivalent circuit of the proposed circuitis shown in Fig. 1(b). Note that the magnetizing inductance LM

is employed as the converter filter inductor.The converter has two modes of operation and eight operat-

ing intervals in each power flow direction mode. The equivalentcircuits of each operating interval in the boost and buck modesare shown in Figs. 2 and 3, respectively, where the currentarrows refer to the actual direction of the current. Since theoperation of the auxiliary circuit and the theoretical equationsin both boost and buck modes are similar, only the boost modeis discussed. In order for the theoretical equations to be applica-ble for both modes, the theoretical equations are written basedon the operating duty cycle (D). As a result, the theoreticalequations of each interval in the boost mode for S1, Sa1, andS2 are true in the buck mode for S2, Sa2, and S1, respectively[26], [29]. The key waveforms of the converter are illustratedin Fig. 4. In order to simplify the theoretical analysis, thefollowing assumptions are made.

(1) All elements are ideal, and the converter is operating atsteady-state condition.

(2) Magnetizing inductor LM is large enough to assume thatits current (ILM ) is constant in a switching cycle.

(3) The input and output voltages are constant and modeledas two voltage sources VL and VH , respectively.

A. Boost Mode of Operation

In this mode, S1 and Sa1 are the main and auxiliary switches,respectively. Besides, S2 and Sa2 act as synchronous rectifier(SR) switches. Before the first interval, it is assumed that S1,Sa1, and Sa2 are off and S2 is on and the magnetizing induc-tor current (ILM ) is flowing to VH through S2. Besides, it

is assumed that no current flows in the windings of the idealtransformer in the model and the voltage across the primaryand secondary windings of the ideal transformer in the modelare VHD and nVHD, respectively.

1) Interval 1: (t0 − t1) [see Fig. 2(a)] At t0, Sa1 and Sa2

turn on simultaneously. Due to the series inductor Llk, Sa1 andSa2 turn-on is under zero-current switching (ZCS). By turningSa1 and Sa2 on, the voltage on the secondary winding of theideal transformer (nVHD) is placed across Llk and thus, Sa1

current starts to increase linearly as follows:

ISa1 =nVHD

Llk(t− t0). (1)

During this interval, ISa1 flows out the dotted terminal ofthe ideal transformer secondary side, and thus, nISa1 entersthe dotted terminal of the primary side. Therefore, IS2 can beobtained as follows:

IS2 = −ILM + nISa1. (2)

By substituting (1) in (2),

IS2 = −ILM +n2VHD

Llk(t− t0). (3)

According to (3), in this interval, IS2 decreases from −ILM

to zero and then increases in the opposite direction for a shorttime through S2. As seen, the reverse recovery of the S2 bodydiode is prevented to occur and hence, the losses of the rec-tifying diode reverse recovery are eliminated completely. Atthe end of this interval, S2 current is defined as IR and con-sequently, Sa1 current is (ILM + IR) /n. The duration of thisinterval is

t1 − t0 =(ILM + IR)Llk

n2VHD. (4)

Note that the value of IR is effective in providing ZVS condi-tion when the converter operates at operating duty cycles below0.5. This point is discussed in the Section III.

2) Interval 2: (t1 − t2) [see Fig. 2(b)] In this interval, theSR main switch S2 turns off under ZVS condition due to thecapacitor CS . By turning S2 off, a resonance starts between Llk

and CS . During this resonance, CS discharges from VH to zeroto provide ZVS condition for S1 at turn-on. S1 voltage is

VS1 = VH(1−D) + VHD cos(ω0(t− t1))

− IRZ0 sin(ω0(t− t1)) (5)

where

ω0 =n√

LlkCS

Z0 =

√LlkCS

n. (6)

At the end of this interval, Sa1 current is I0.3) Interval 3: (t2 − t3) [see Fig. 2(c)] This interval starts

when the S1 body diode starts conducting and the main switchS1 can be turned on under ZVS condition. Also, the volt-age across the primary and secondary windings of the ideal


Fig. 2. Equivalent circuit for each operating interval of the proposed converter in boost mode of operation. (a) Interval 1. (b) Interval 2. (c) Interval3. (d) Interval 4. (e) Interval 5. (f) Interval 6. (g) Interval 7. (h) Interval 8.

Fig. 3. Equivalent circuit for each operating interval of the proposed converter in buck mode of operation. (a) Interval 1. (b) Interval 2. (c) Interval 3.(d) Interval 4. (e) Interval 5. (f) Interval 6. (g) Interval 7. (h) Interval 8.

transformer in the model are changed to −VH(1−D) and−nVH(1−D), respectively. Therefore, a negative voltage isplaced across Llk and Sa1 current reduces linearly as follows:

ISa1 = I0 − nVH(1−D)

Llk(t− t2). (7)

Consequently, S1 current is

IS1 = ILM − nI0 +n2VH(1−D)

Llk(t− t2). (8)

At the end of this interval, IS1 and ISa1 currents reach zeroand ILM/n, respectively, and the body diode of S1 is turned offunder ZCS condition. The duration of this interval is

t3 − t2 =(nI0 − ILM )Llk

n2VH(1−D). (9)

4) Interval 4: (t3 − t4) [see Fig. 2(d)] In this interval, theSR auxiliary switch Sa2 turns off and so, the body diode of Sa2

conducts ISa1. Besides, Sa1 current reduces from ILM/n tozero and IS1 increases from zero to ILM . Sa1 and S1 currentequations are as follows:

ISa1 =ILM

n− nVH(1−D)

Llk(t− t3) (10)

IS1 =n2VH(1−D)

Llk(t− t3). (11)

At the end of this interval, the body diode of Sa2 is turned offunder ZCS condition. The duration of this interval is

t4 − t3 =ILMLlk

n2VH(1−D). (12)

5) Interval 5: (t4 − t5) [see Fig. 2(e)] This interval is iden-tical to a conventional PWM boost converter when the mainswitch is on. The magnetizing inductor current ILM flowsthrough S1. In this interval, since any current is not flowingthrough the auxiliary circuit, the auxiliary switch Sa1 can beturned off under ZCS condition.

6) Interval 6: (t5 − t6) [see Fig. 2(f)] At t5, the mainswitch S1 turns off under ZVS condition due to capacitor CS .By turning S1 off, CS is charged linearly by the magnetizinginductor current ILM . At the end of this interval, CS is chargedto VH and the S2 body diode begins to conduct.

7) Interval 7: (t6 − t7) [see Fig. 2(g)] This operating inter-val is identical to a conventional PWM boost converter whenthe main switch is off. The magnetizing inductor current ILM

flows to the output through the S2 body diode.8) Interval 8: (t7 − t0 + T ) [see Fig. 2(h)] Since the

S2 body diode is conducting, the main switch S2 can beturned on under ZVS condition. So, ILM flows to the outputthrough S2.


Fig. 4. Converter theoretical waveforms in the boost mode (buck mode)of operation.

III. DESIGN CONSIDERATIONS

A. Design of the Proposed Converter Components

The main buck and boost converter is designed like a regu-lar PWM buck and boost converter. Note that in the proposedconverter, LM is employed as the converter filter inductor, andthus, it is designed as the inductor filter in a conventional PWMbuck and boost converter.CS provides ZVS condition for the main switches at turn-

off instant. Therefore, its value can be selected similar to anysnubber capacitor [34]. Similarly, Llk provides ZCS conditionfor the auxiliary switches at turn-on instant and its value can beselected similar to any snubber inductor [34]. The value of Llk

on the secondary side is obtained as follows:

Llk = (1− k2)L2 (13)

where k is coupling coefficient of the coupled inductors and L2

is the inductance of the secondary windings which is equal tothe square of n multiplied by LM (L2 = n2LM ). So, Llk on thesecondary side can be obtained as

Llk = (1− k2)n2LM . (14)

Finally, it is important to select the value of n. Note that from(6) and (14), the values of ω0 and Z0 are obtained as follows:

ω0 =1√

(1− k2)LMCS

Z0 =

√(1− k2)LM

CS. (15)

As a result, the values of ω0 and Z0 and also, the duration ofthe interval 2 are independent of n. Besides, if the value of the

Fig. 5. Normalized value of IR_Min_Req versus D

Llk obtained by (14) is substituted in (4), (9), and (12), it canbe seen that the duration of intervals 1, 3, and 4 are also inde-pendent of n. Thus, there is flexibility in choosing the valueof n. The voltage stress of the auxiliary switches is equal tonVHD. Thus, selecting smaller value for n results in lowervoltage stress for auxiliary switches. However, as observedfrom (14), a small value of n would reduce the value of Llk

which plays the role of snubber inductor for auxiliary switches.Consequently, the value of n is can be selected in the range1/4–1/2.

B. Soft Switching Condition

As discussed in the previous section, during interval 1, inorder to eliminate the rectifying diode reverse recovery losses,the current of the SR main switch is reduced to zero by theauxiliary circuit and then flows in the opposite direction fora short time. The final value of the reverse current in the SRmain switch is defined as IR. So, if IR is equal or greater thanzero, reverse recovery losses are eliminated completely. On theother hand, in order to achieve ZVS condition for the mainswitches at turn-on, at the end of the operating interval 2, themain switch voltage must be zero. So, from (5), the followingequation should be satisfied:

VH(1−D) + VHD cos (ω0(t2 − t1))

− IRZ0 sin (ω0(t2 − t1)) = 0. (16)

The preceding equation is true in both boost and buck modesof operations. Note that for a specific voltage of the input andoutput, the duty cycles of the boost and buck modes are com-plement of each other. As seen from (16), the value of IR iseffective for the ZVS condition. The minimum required valueof IR which satisfies (16) is defined as IR_Min_Req. Using (16),the normalized value of IR_Min_Req versus D is plotted in Fig. 5.As seen, for operating duty cycles higher than 0.5, the valueof IR_Min_Req is zero. On the other hand, for operating dutycycles below 0.5, as the duty cycle is decreased, the value ofIR_Min_Req is increased. Hence, if the value of IR is greaterthan IR_Min_Req , ZVS condition for the main switch at turn-on and elimination of the rectifying diode reverse recoveryare provided, simultaneously. So, the following soft-switchingcondition can be formulated:

IR > IR_Min_Req. (17)


Fig. 6. Experimental waveforms for VL = 80V, VH = 200V, and Po = 250W in (a) boost mode and (b) buck mode (time scale is 1µs/div).


Now, it is important to provide IR which satisfied the pre-ceding condition. From (4), the value of IR is obtained asfollows:

IR = −ILM +(t1 − t0)n

2VHD

Llk(18)

where (t1 − t0) is the duration time of interval 1. Since, thistime is the duration time between the auxiliary switch turn-onand the SR main switch turn-off, it is feasible to adjust this dura-tion. So, from (17) and (18), and by substituting Llk from (14),the following soft-switching condition is achieved:

(t1 − t0) >(ILM + IR_Min_Req)(1− k2)LM

VHD. (19)

Consequently, by tuning the duration time between the aux-iliary switch turn-on and the SR main switch turn-off as

formulated in (19), ZVS condition of the main switches at turn-on and elimination of the rectifying diodes reverse recovery areprovided. Note that to achieve ZVS condition over the inputvoltage variations and various load conditions, the duration time(t1 − t0) should be adjusted for the worst-case converter oper-ation. In this case, IR_Min_Req, ILM , and D should be selectedso that the right side of (19) is maximized.

In order to achieve ZCS condition for the auxiliary switchat turn-off, the turn-on period of the auxiliary switch should besufficient, so that the auxiliary switch current can be reducedto zero. Besides, the SR auxiliary switch should be turnedoff before resetting the auxiliary switch current to zero. Theturn-on period of the auxiliary switch and SR auxiliary switchare defined as Ton_aux and Ton_SR_aux, respectively. So, Ton_aux

should be greater than the duration time of the operating inter-vals 1, 2, 3, and 4. In this case, the auxiliary switch currentduring the interval 2 is assumed to be constant and for the worth



case condition, the duration time of the interval 2 is estimatedat half of the resonance period time. So, from (4), (6), (9), (12),and (14), Ton_aux is obtained as follows:

Ton_aux > (t1 − t0)1

1−D+

π

ω0. (20)

In addition, to make sure that the SR auxiliary switchturns off before resetting the auxiliary switch current to zero,the duration time of the interval 2 is assumed as the deadtime between the auxiliary and SR auxiliary switches turn-offinstants. So, from (20), Ton_SR_aux is obtained as follows:

Ton_SR_aux < (t1 − t0)1

1−D. (21)

In (20) and (21), the value of (t1 − t0) is set as discussedbefore and ω0 is obtained from (6). Besides, in (20) and(21), the value of D is designed for the maximum and mini-mum values of the converter-operating duty cycle, respectively.Hence, by tuning Ton_aux and Ton_SR_aux as formulated in (20)and (21), ZCS condition and synchronous rectification of theauxiliary switches are provided. Note that (20) and (21) aretrue in both boost and buck modes of operations. As dis-cussed, in the boost mode, Sa1 is the auxiliary switch andSa2 acts as the SR auxiliary switch and vice versa for thebuck mode.

IV. EXPERIMENTAL RESULTS

A prototype of the proposed bidirectional buck and boostconverter is implemented. The high-voltage side (VH) is 200 Vand the low-voltage side (VL) range is 80–100 V. The converteroperates at 100 kHz and an output power (Po) of 250 W.

According to the discussions in the previous section, theselected values of n, LM , and CS are 1/3, 270μH, and2.2 nF, respectively. Also, the value of k is obtained about0.98 and so, from (14), the value of Llk is about 1.2μH. For

TABLE 1COMPARISON OF THE AUXILIARY SWITCHES VOLTAGE STRESS

IN THE PROPOSED CONVERTER AND THE PREVIOUSLYPROPOSED ZVT BDCS

the main switches IRF644 (VDS = 250V, RDS(on) = 280mΩ)and for the auxiliary switches IRF3205 (VDS = 55V,RDS(on) = 8mΩ) are used. In boost mode, the range of oper-ating duty cycle is 0.5–0.6. So, from (19), the worst-caseoperating point is when the converter operates at full load andoperating duty cycle of 0.6. By using (19), the duration timeof (t1 − t0) for the boost mode is obtained as about 0.278μs.This time is set at 0.3μs. Similarly, in buck mode, the operat-ing duty cycle range is 0.4–0.5. From (19), the duration time of(t1 − t0) is obtained as about 0.58μs for the full load conditionand operating duty cycle of 0.4. This time is set at 0.6μs for thebuck mode. Besides, from (20) and (21), the values of Ton_aux

and Ton_SR_aux for the boost mode are set at

Ton_aux = 1.3μs, Ton_SR_aux = 0.6μs.

For the buck mode, Ton_aux and Ton_SR_aux are set at

Ton_aux = 1.7μs,Ton_SR_aux = 1μs.

The experimental waveforms of the proposed converter forVL = 80V, VH = 200V at full load (Po = 250W) and 50%load (Po = 125W) are shown in Figs. 6 and 7, respectively.Besides, Fig. 8 shows the experimental waveforms of theproposed converter for VL = 100V, VH = 200V at full load(Po = 250W). It can be observed that ZVS condition for


Fig. 9. Efficiency versus output power of the proposed converter atoperating point of VL = 80V and VH = 200V and its hard switchingcounterpart in (a) boost and (b) buck modes.

the main switches as well as ZCS condition for the auxiliaryswitches is provided for both turn on and turn off instants. Asseen, the auxiliary switches voltage stress in comparison withthe high-voltage side (VH) is reduced significantly. For exam-ple, in the operating point of VL = 80V and VH = 200V, theauxiliary switches voltage stress in the boost and buck modesare about 40 V (20% of high voltage side) and 30 V (15%of high voltage side), respectively. This is in contrast with thepreviously coupled inductor ZVT BDC in [26], where the auxil-iary switch voltage stresses exceed the output voltage by about120%. The comparison of the auxiliary switch voltage stressin the proposed converter and the previously proposed ZVTBDCs is presented in Table I. The converter efficiency curvesfor operating point of VL = 80V and VH = 200V are shownin Fig. 9 for both buck and boost modes. Due to ZCS con-dition and low-voltage stress of the auxiliary switches, usingswitches with low-voltage rating and low on-state resistance(RDS(on)) as the auxiliary switches and synchronous rectifi-cation of the auxiliary switches body diodes, the total auxiliarycircuit losses are significantly low. Hence, high efficiency overa wide operating range is obtained. Note that the efficiency ofthe hard-switching counterpart converter is for a regular buckand boost converter with the same parameters using IRF644as their switches. Fig. 10 shows the implemented prototypeconverter.

V. OTHER ZVT BDCS

Similar to the bidirectional buck and boost converter, thisZVT technique can be applied to other basic nonisolated bidi-rectional converters to improve their efficiency, as shown inFig. 11 In all topology variations, the theoretical operatingmodes are very similar to the operation of the bidirectional buck

Fig. 10. Implemented prototype converter.

Fig. 11. The proposed family of ZVT bidirectional converters:(a) buckboost/buck-boost, (b) Cuk/Cuk, (c) SEPIC/Zeta

and boost converter explained in Section II and thus furtherexplanation is neglected.

VI. CONCLUSION

A new family of ZVT bidirectional converters with a simpleauxiliary circuit has been introduced. In these converters, soft-switching condition for all semiconductor elements is achievedonly by adding two auxiliary switches and coupled inductors.The theoretical analysis for a bidirectional buck and boostconverter was presented in detail. The theoretical analysisshows that by tuning the duration time between the auxiliaryswitch turn-on and the synchronous rectifier switch turn-off,as formulated, ZVS condition of the main switches at turn-onand elimination of the rectifying diodes reverse recovery areprovided. In addition, ZCS condition for the auxiliary switches


at both turn-on and turn-off instants is achieved. Consequently,high efficiency over a wide operating range is obtained. In addi-tion, the auxiliary switches benefits from very low voltage stress(nDVH) and there is no need for a floating gate drive circuit. Tovalidate the theoretical analysis, a 250-W prototype of the con-verter at 100 kHz is implemented. Also, other family membersof the proposed bidirectional converters were presented.

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Mohammad Reza Mohammadi was born inIsfahan, Iran. He received the B.S. degree fromAmir Kabir University of Technology, Tehran,Iran, in 2007, and the M.S. degree fromIsfahan University of Technology, Isfahan, Iran,in 2011, both in electrical engineering. He iscurrently working toward the Ph.D. degree inthe Department of Electrical and ComputerEngineering, Isfahan University of Technology.

His research interests include soft-switchingtechniques in bidirectional converters.

Hosein Farzanehfard was born in Isfahan, Iran,in 1961. He received the B.S. and M.S. degreesin electrical engineering from the University ofMissouri, Columbia, MO, USA, in 1983 and1985, respectively, and the Ph.D. degree inelectrical engineering from Virginia PolytechnicInstitute and State University, Blacksburg, VA,USA, in 1992.

Since 1993, he has been a Faculty Memberwith the Department of Electrical and ComputerEngineering, Isfahan University of Technology,

Isfahan, Iran. He is the author or coauthor of more than 100 tech-nical papers published in journals and conference proceedings. Hisresearch interests include high-frequency soft-switching converters,pulse power applications, power factor correction, active power filters,and high-frequency electronic ballasts.

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