a novel dc-ac single phase resonant inverter usingsoft switching boost converte

A Novel DC-AC Single Phase Resonant Inverter using

Soft switching boost converter

ABSTRACT

In this paper, a novel DC-AC single phase inverter is proposed. When the switches

are turned on and off, a conventional inverter generates switching loss because of the hard

switching. Thus, the inverter loss is increased. Proposed system contains auxiliary circuit.

The converter stage switches perform soft-switching because of the auxiliary circuit. Also

inverter stage switches perform ZVS when the dc-link voltage is zero. Therefore all switches

perform soft-switching when the switches are turned on and off. Thus the proposed system

reduces switching loss and voltage stress.

Literature survey of the Project

[I] F. C. Schwarz and J. Benklassens, “A controllable 45 kW currentsource for dc machines,” IEEE Trans. Ind. Appl., vol. IA-15, no. 4. The development of zero-switching-loss inverters has attracted much interest for industrial

applications. The resonant de link inverter (RDCLI) provides a simple and robust approach

for realizing switching frequencies > 20 kHz in multi-kilowatt systems but impresses

substantial voltage stress (- 2.5 supply voltage) across the devices. Two alternate topologies

for realizing zero switching losses in high-power converters are proposed. The actively

clamped resonant dc-link inverter (ACRLI) uses the concept of a lossless active clamp to

restrict voltage stresses to only 1.3-1.5 supply voltage (V,) while maintaining a mode of

operation similar to the RDCLI. For applications demanding substantially superior spectral

performance, the resonant pole inverter (RPI), also called the quasi-resonant current mode

inverter (QRCMI), is proposed as a viable topology. Using only six deices rated at supply

voltage ( Vs), this circuit transfers the resonant components to the ac side of each phase and

thus requires additional inductor and capacitor (LC) components. On the other hand, the RPI

is capable of true pulse width modulation (PWM) operation at high frequency as opposed 10

discrete pulse modulation (DPM) in the RDCLI. Detailed analysis, simulation, and

experimental results are presented to verify- operation principles of both power converters.

Cl audio Manoel C. Duarte, Student Member, IEEE, and Ivo Barbi, Senior Member, IEEE

This paper presents a technique to generate a complete family of two-switch pulsewidth-

modulated (PWM) with active clamping dc/dc converters, featuring soft commutation of the

semiconductors at zero-voltage (ZVS). The main purpose of this technique is to integrate

these converters under a same theoretical principle to derive the topologies in a

comprehensive form and generate new circuits. All the converters have the advantage of soft

commutation (ZVS) with minimum switch voltage stress due to the clamping action. Besides

operating at constant frequency and with reduced commutation losses, these converters have

output characteristics similar to the PWM hardswitching counterpart, which means that there

is no circulating

reactive energy that would cause large conduction losses. Principle of operation, theoretical

analysis, simulation and experimental results of one particular converter taken as an example,

are provided in this paper.

Jung-Goo Cho, Member, IEEE, Ju-Won Baek, Chang-Yong Jeong, and Geun-Hie Rim,

Member, IEEE A novel zero-voltage and zero-current- switching (ZVZCS) full-bridge

pulsewidth modulation converter is presented to simplify the circuits of the previously

presented ZVSCS converters. A simple auxiliary circuit, which consists of one small

capacitor and two small diodes, is added in the secondary to provides ZVZCS conditions to

primary switches, as well as to clamp secondary rectifier voltage. The additional clamp

circuit for the secondary rectifier is not necessary. The auxiliary circuit includes neither lossy

components nor additional active switches, which makes the proposed converter efficient and

cost effective. The principle of operation, features, and design considerations are illustrated

and verified on a 2.5-kW 100–kHz insulated-gate-bipolar-transistor-based experimental

circuit

CHAPTER- I

1.1 INTRODUCTION OF PROJECT:

Nowadays, the power electronics are required to develop smaller, lighter, less expensive and

reliable system. In order to operate these systems, a switching frequency has to be increased.

But, increasing the inverter switching frequencies is dependent on the advances in device

technology and makes higher switching losses. To solve this problem, the soft switching

techniques have been adopted in the inverter circuit. By the soft-switching techniques, the

switching losses are ideally zero and the switching frequencies can be increased to above the

audible range. In this paper, a novel DC-AC single phase resonant inverter using soft

switching boost converter is proposed. This proposed inverter consists of soft-switching boost

converter and H-bridge inverter. The soft-switching boost converter in proposed inverter

additionally has resonant inductor Lr, resonant capacitor Cr, bridge diode and auxiliary

switch Q2. When the resonance between resonant inductor and capacitor is generated, the

converter switches are turned on and off with soft-switching.

Also H-bridge inverter switches are turned on and off with ZVS when the auxiliary

switch is turned off. So all of the switches are turned on and off with soft-switching.

Therefore the novel DC-AC single phase resonant inverter using soft-switching boost

converter can reduce the switching loss and voltage stress. In this paper, we have analyzed

the operational principle of the proposed resonant inverter. Simulation results presented to

confirm the theoretical analysis. Fig. 1 shows the proposed a novel DC-AC single phase

resonant inverter using soft-switching boost converter. The auxiliary circuit in proposed

inverter consists of an auxiliary switch, resonant inductor, resonant capacitor, and bridge

diode. So, the main switch is turned on with ZCS and turned off with ZVS. Also the auxiliary

switch is turned on and off with ZVS. Therefore, the converter stage switches perform the

soft-switching. The H-bridge inverter performs the soft-switching under the influence of the

dc-link voltage. When the auxiliary switch is turned off with ZVS, the dc-link voltage is zero.

The inverter switches are turned on and off with ZVS while the dc-link voltage becomes zero.

So all of switches in proposed inverter are turned on and off with soft-switching. Therefore,

the proposed soft-switching inverter has many advantages like as improved efficiency, low

switching losses, low voltage stress, reduced acoustic noise and EMI. Another significant

advantage of the proposed topology is an excellent PWM capability due to not only variable

link pulse but also variable pulse position. The proposed inverter operation mode analysis can

be divided into six modes, as shown in Fig shows the proposed waveforms for the novel DC-

AC single phase resonant inverter using soft-switching boost converter.

1.2 INTRODUCTION TO INVERTERS

1.2.1 INVERTER:

Inverter is power electronic circuit that converts a direct current into an alternative current

power of desired magnitude and frequency. The inverters find their application in modern ac

motor and uninterruptible power supplies.

Classification of inverters:

Based on the source used

Voltage source inverter

Current source inverter

Based on switching methods

Pulse width modulation inverters

Square wave inverters

Based on switching devices used

Transistorized inverter

Thyristorised inverter

Based on the inversion principle

Resonant inverter

Non- Resonant inverter

Semiconductor devices:

The electronic semiconductor device acts as a switching device in the power

electronic converters. In general, the characteristics of the device are utilized in such a way

that it acts as a short circuit when closed. In addition to, an ideal switch also consumes less

power to switch from one state to other.

Semiconductor is defined as the material whose conductivity depends on the energy

(light, heat, etc.,) falling on it. They don’t conduct at absolute zero temperature. But, as the

temperature increases, the current conducted by the semi conductor increases as it gets energy

in the form of heat. The increase in current is proportional to the temperature rise.

Semiconductor switches are diodes, SCR, MOSFET, IGBT, BJT, TRIAC etc.,

1.2.2 MOSFET-Assisted Soft-Switching of IGBTs

A power MOSFET is almost invariably used in today's high-frequency power converter

applications being a voltage controlled, fast switching and majority-carrier device. However,

MOSFET's major limitation is that the on-state drain-to-source resistance (RDSon) is high and

increases rapidly with the device's voltage rating. As a consequence the conduction losses are

high, particularly for high power applications, limiting their application.

An Insulated Gate Bipolar Transistor (IGBT) is a more attractive device for high power and

high voltage applications. The device combines the advantages of BJT which has fixed

voltage drop in the on-state, high voltage, and high current ratings and a MOSFET which is a

voltage controlled majority carrier device. The major problem of IGBT for operation at high

frequency is the "current tailing". At turn-off the device current does not fall rapidly but a

considerable portion of the current lingers or tails for a longer time. The co-existence of tail

current and high collector-to-emitter voltage of IGBT cause high turn-off switching losses.

This sets the upper limit on the switching frequency of an IGBT.

In order to increase the switching frequency of operation with IGBT, one must use the switch

in soft-switching mode. That is, the switching transitions must take place either under zero-

voltage-switching (ZVS) or zero-current-switching (ZCS) conditions. In ZVS, turn-on losses

are eliminated but the turn-off losses are reduced by slowing down the rise of voltage across

the device. With the presence of tail current, ZVS does not completely eliminate the turn-off

losses. In ZCS, collector current is forced to zero before turn-off. Therefore the effect of tail

current on the turn-off losses is eliminated. However, with ZCS the loss due to discharge of

device's output capacitance is not prevented.

Resonant, quasi-resonant, and resonant-transition converters can create favorable ZVS or

ZCS conditions for an IGBT in a circuit. This is achieved with the help of auxiliary switches

and/or additional inductors and capacitors. Apart form the increased complexity, these

techniques can increase the peak voltage and current stresses. A simple and effective method

of reducing turn-off loss in an IGBT with the help of MOSFET is proposed in Reference 9

of Qian et al. In the said technique a MOSFET is either operated in series or parallel with the

IGBT and with the proper sequencing of the turn-on and turn-off of the MOSFET, soft-

switching of IGBT is achieved.

In this article the MOSFET-assisted soft-switching of IGBT is re-investigated. With

simulation models, it was noticed that although the total power dissipation in MOSFET and

IGBT is considerably reduced, the loss in MOSFET is almost equal to that in the IGBT under

the most favorable operating conditions. This contradicts with the intention: since a MOSFET

is used only to assist the soft-switching of the IGBT, we intend that the MOSFET ratings

should be small as compared to the main switch—IGBT. If we have a MOSFET that can

dissipate half of the total switch power dissipation, we would rather put two MOSFETs in

parallel instead of using an IGBT.

An alternative switching method suggested in this article can reduce the ratings of MOSFET.

MOSFET-Assisted Soft-Switching of IGBTs: An Overview

It has been shown in Qian et al that a MOSFET connected in series and parallel with the

IGBT can create soft-switching conditions for the IGBT during turn-off. MOSFET and IGBT

operated in series, is termed as the emitter-open operation mode. The parallel operation of

MOSFET and IGBT has been termed as the mixed parallel operation mode.

The insulated-gate bipolar transistor or IGBT

The insulated-gate bipolar transistor or IGBT is a three-terminal power semiconductor

device, noted for high efficiency and fast switching. It switches electric power in many

modern appliances: electric cars, variable speed refrigerators, air-conditioners, and even

stereo systems with digital amplifiers. Since it is designed to rapidly turn on and off,

amplifiers that use it often synthesize complex waveforms with pulse width modulation and

low-pass filters.

The IGBT combines the simple gate-drive characteristics of the MOSFETs with the high-

current and low–saturation-voltage capability of bipolar transistors by combining an isolated-

gate FET for the control input, and a bipolar power transistor as a switch, in a single device.

The IGBT is used in medium- to high-power applications such as switched-mode power

supply, traction motor control and induction heating. Large IGBT modules typically consist

of many devices in parallel and can have very high current handling capabilities in the order

of hundreds of amps with blocking voltages of 6,000 V.

The IGBT is a fairly recent invention. The first-generation devices of the 1980s and early

1990s were relatively slow in switching, and prone to failure through such modes as latch up

and secondary breakdown. Second-generation devices were much improved, and the current

third-generation ones are even better, with speed rivaling MOSFETs, and excellent

ruggedness and tolerance of overloads.

The extremely high pulse ratings of second- and third-generation devices also make them

useful for generating large power pulses in areas like particle and plasma physics, where they

are starting to supersede older devices like thyratrons and triggered spark gaps.

Their high pulse ratings, and low prices on the surplus market, also make them attractive to

the high-voltage hobbyist for generating large amounts of high-frequency power to drive

experiments like Tesla coils. Availability of affordable, reliable IGBTs is a key enabler for

electric vehicles and hybrid cars. Toyota's second generation hybrid Prius has a 50 kW IGBT

inverter controlling two AC motor/generators connected to the DC battery pack.

In addition, for about 20 years Toshiba has applied IGBTs as audio amplifiers

1.2.3 History of the IGBT

The IGBT is a semiconductor device with four alternating layers (P-N-P-N) that are

controlled by a metal-oxide-semiconductor (MOS) gate structure without regenerative action.

This mode of operation was first proposed by Yamagami in his Japanese patent S47-21739,

which was filed in 1968. This mode of operation was first experimentally discovered by B. J.

Baliga in vertical device structures with a V-groove gate region and reported in the literature

in 1979. The device structure was referred to as a ‘V-groove MOSFET device with the drain

region replaced by a p-type Anode Region’ in this paper and subsequently as the insulated-

gate rectifier (IGR), the insulated-gate transistor (IGT), the conductivity-modulated field-

effect transistor (COMFET). and "bipolar-mode MOSFET".

Plummer found the same IGBT mode of operation in the four layer device(SCR) and he first

filed a patent application for the device structure in 1978. USP No.4199774 was issued in

1980 and B1 Re33209 was reissued in 1995 for the IGBT mode operation in the four layer

device(SCR.)

Hans W. Beck and Carl F. Wheatley invented a similar device for which they filed a patent

application in 1980, and which they referred to as "power MOSFET with an anode region".

This patent has been called "the seminal patent of the Insulated Gate Bipolar Transistor." The

patent claimed "no thyristor action occurs under any device operating conditions." This

substantially means the non-latch-up IGBT operation for the entire device opeation range.

Baliga had filed for a patent on his similar device the same year, but it took much longer to

issue (1990, compared to Beck and Wheatley's 1982 issue date).

Devices capable of operating over an extended current range for use in applications were first

reported by Baliga et al. in 1982. A similar paper was also submitted by J.P. Russel et al. to

IEEE Electron Device Letter in 1982. The applications for the device were initially regarded

by the power electronics community to be severely restricted by its slow switching speed and

latch-up of the parasitic thyristor structure inherent within the device. However, it was

demonstrated by Baliga and also by A.M. Goodman et al. in 1983 that the switching speed

could be adjusted over a broad range by using electron irradiation. This was followed by

demonstration of operation of the device at elevated temperatures by Baliga in 1985.

Successful efforts to suppress the latch-up of the parasitic thyristor and the scaling of the

voltage rating of the devices at GE allowed the introduction of commercial devices in 1983,

which could be utilized for a wide variety of applications.

Complete suppression of the parasitic thyristor action and the resultant non-latch-up IGBT

operation for the entire device operation range was achieved by A. Nakagawa et al. in 1984.

The non-latch-up design concept was filed for US patents. The developed 1200V IGBTs were

directly connected without any loads to a 600V constant voltage source and were switched-on

for 25micro-seconds. The device forward voltage became the same as the source voltage and

a large current as much as possible flowed. The developed devices successfully withstood the

severe condition. This was the first demonstration of so-called "short-circuit-withstanding-

capability" in IGBTs. Non-latch-up IGBT operation was ensured, for the first time, for the

entire device operation range. In this sense, the non-latch-up IGBT proposed by Hans W.

Beck and Carl F. Wheatley was realized by A. Nakagawa et al. in 1984. Products of non-

latch-up IGBTs were first commercialized by Toshiba in 1985.

Structure of IGBT

An IGBT cell is constructed similarly to a n-channel vertical construction power MOSFET

except the n+ drain is replaced with a p+ collector layer, thus forming a vertical PNP bipolar

junction transistor.

http://en.wikipedia.org/wiki/Image:IGBT_Cross_Section.jpg

Cross section of a typical IGBT showing internal connection of MOSFET

and Bipolar Device

This additional p+ region creates a cascade connection of a PNP bipolar junction transistor

with the surface n-channel MOSFET. This connection results in a significantly lower forward

voltage drop compared to a conventional MOSFET in higher blocking voltage rated devices.

As the blocking voltage rating of both MOSFET and IGBT devices increases, the depth of the

n- drift region must increase and the doping must decrease, resulting in roughly square

relationship increase in forward conduction loss compared to blocking voltage capability of

the device. By injecting minority carriers (holes) from the collector p+ region into the n- drift

region during forward conduction, the resistance of the n- drift region is considerably

reduced. However, this resultant reduction in on-state forward voltage comes with several

penalties:

The additional PN junction blocks reverse current flow. This means that IGBTs cannot

conduct in the reverse direction, unlike a MOSFET. In bridge circuits where reverse current

flow is needed an additional diode (called a freewheeling diode) is placed in parallel with the

IGBT to conduct current in the opposite direction. The penalty isn't as severe as first assumed

though, at the higher voltages where IGBT usage dominates; discrete diodes are of

significantly higher performance than the body diode of a MOSFET.

The reverse bias rating of the N- drift region to collector P+ diode is usually only of 10's of

volts, so if the circuit application applies a reverse voltage to the IGBT, an additional series

diode must be used. The minority carriers injected into the n- drift region take time to enter

and exit or recombine at turn on and turn off. This results in longer switching time and hence

higher switching loss compared to a power MOSFET.

The additional PN junction adds a diode-like voltage drop to the device. At lower blocking

voltage ratings, this additional drop means that an IGBT would have a higher on-state voltage

drop. As the voltage rating of the device increases, the advantage of the reduced N- drift

region resistance overcomes the penalty of this diode drop and the overall on-state voltage

drop is lower (the crossover is around 400 V blocking rating). Thus IGBTs are rarely used

where the blocking voltage requirement is below 600 V

Classification of semiconductor device

Based on controllability:

Uncontrol switching device (SCR)

Semicontrol switching device

Fully control switching device

Based on control modes:

Current control devices(SCR ,BJT)

Voltage control device(MOSFET ,IGBT)

Based on current direction

Unidirectional device (SCR,MOSFET ,IGBT)

Bi- Unidirection device(TRIAC)

Advantages of inverters:

Small leakage current during off stage

Low voltage drop during ON stage

Faster turn ON and turn OFF

Small control power to switch from one state to other

High forward current and blocking voltage capabilities.

High dv/dt and di/dt ratings

Application of Inverters:

Adjustable speed ac drives,

UPS static VAR compensators

Active filters

Flexible AC transmission system

In all vehicle for lightning

Now also used for driving electric vehicle

CHAPTER – II

2.1 TRADITIONAL SOURCE INVERTERS

Traditional source inverters are Voltage Source Inverter and Current Source

Inverter. The input of Voltage Source Inverter is a stiff dc voltage supply, which can be a

battery or a controlled rectifier both single phase and three phase voltage source inverter are

used in industry. The switching device can be a conventional MOSFET, Thyristor, or a power

transistor.

Voltage source inverter is one which the dc source has small or negligible

impedance. In other words a voltage source inverter has stiff dc source voltage at its input

terminals. A current-fed inverter or current source inverter is fed with adjustable dc current

source. In current source inverter output current waves are not affected by the load.

2.2 VOLTAGE SOURCE INVERTER

When the power requirement is high, three phase inverters are used. When three

single phase inverters are connected in parallel, we can get the three phase inverter. The

gating signals for the three phase inverters have a phase difference of 120o. These inverters

take their dc supply from a battery or from a rectifier and can be called as six-step bridge

inverter. Fig.1.1 shows the three phase inverter using six MOSFET’s and with diodes.

A large capacitor is connected at the input terminals tends to make the input dc voltage

constant. This capacitor also suppresses the harmonics fed back to the source. The Voltage

Source Inverter is widely used. However, it has the some conceptual and theoretical barriers

and limitations. The AC output voltage is limited and cannot exceed the AC input voltage.

Therefore the Voltage Source Inverter is only buck (step down) inverter operation for DC to

AC power conversion or boost (step-up) operation for AC to DC power conversion.

Fig. Voltage Source Inverter

For applications where over drive is desirable and the available dc voltage is

limited, an additional dc-dc boost converter is needed to obtain a desired ac output. The

additional power converter stage increases system cost and lowers efficiency. The upper and

lower devices of each phase leg cannot be gate on simultaneously either by purpose or by

EMI noise. Otherwise a shoot through problem by Electromagnetic interference noise’s

misgating–on is major killer to the inverter reliability. Dead time to block both upper and

lower devices has to provide in the Voltage Source Inverter which causes the waveform

distortion, etc. An output LC filters needed for providing a sinusoidal voltage compared with

Current Source Inverter which causes additional power loss and control complexity.

2.3 CURRENT SOURCE INVERTER

A Current Source Inverter is fed from a constant current source. Therefore

load current remains constant irrespective of the load on the inverter. The load voltage

changes as per the magnitude of load impedance. When a voltage source has a large

inductance in series with it, it behaves as a Current Source .The large inductance maintains

the current constant.

Fig. Current Source Inverter

The traditional three phase Current Source Inverter structure is shown in Fig. A

dc current source feeds the three phase main inverter circuit. The dc current source can be a

relatively large dc inductor fed by a Voltage Source such as a battery or a rectifier. It consists

of six switches and with anti parallel diodes. This diode provides the bidirectional current

flow and unidirectional voltage blocking capability.

Current Source Inverter has the following conceptual and theoretical barriers and

limitations. The ac output voltage has to be greater than the original dc voltage that feeds the

dc inductor or the dc voltage produced is always smaller than the ac input voltage. Therefore

this inverter is a boost inverter for dc to ac power conversion. For applications where a wide

voltage range is desirable, an additional dc to dc buck converter is needed. The additional

power conversion stage increases system cost and lowest efficiency.

At least one of the upper devices and one of the lower devices have to be gated on

and maintained on at any time. Other wise, an open circuit of the DC inductor would occur

and destroy the devices. The open circuit problem by EMI noise’s misgatting-off is a major

concern of the converters reliability. A current source inverter is fed from a constant current

source. Therefore load current remains constant irrespective of the load on the Inverter. The

load voltage changes as per the magnitude of load impedance. When a voltage source has a

large inductance in series with it, it behaves as a current source .The large inductance

maintains the current constant.

2.4 MODE OF INVERTERS OPERATION

The mode of inverter operation is mainly classified based on the thyristor

conduction period as

180ºconduction

120ºconduction

Three phase inverters are normally used for high power applications. Three single

phase half or full bridge inverters can be connected in parallel to form the configuration of a

three phase inverter. The gating signals of single phase inverters should be advanced or

delayed by 120º with respect to each other in order to obtain three phase balanced voltages.

The three phase output can be obtained from a configuration of six switches and

six diodes. Two types of control signals can be applied to the switches: 180ºconduction or

120ºconduction.

2.4.1 180° CONDUCTION

Each switch conducts for 180°. Three switches remain on at any instant of time.

When switch 1 is switched on, terminal ‘a’ is connected to the positive terminal of the dc

input voltage. When switch 4 is switched on, terminal ‘a’ is connected to the negative

terminal of the dc source. There are six modes of operation in a cycle and the duration of

each mode is 60°.

The switches are numbered in the sequence of gating the switches 123, 234, 345, 456, 561,

612. The gating signals are shifted from each other by 60° to obtain three phase balanced

voltages [7,8].

During mode 1 for 0 ≤ ωt ≤ π/3, switches 1 and 6 conducts.

van = Vs/2 vbn = -Vs/2 vcn = 0

During mode 2 for π/3 ≤ ωt ≤ 2π/3, switches 1 and 2 conduct.

van = Vs/2 vbn = 0 vcn = -Vs/2

During mode 3 for 2π/3 ≤ ωt ≤ 3π/3, switches 2 and 3 conduct.

van = 0 vbn = Vs/2 vcn = -Vs/2







2.4.2. 120° CONDUCTION

Each switch conducts for 120°. Only two switches remain on at any instant of

time. The conduction sequence of switches is 61, 12, 23, 34, 45, 56, and 61. There are three

modes of operation in a half cycle and the equivalent circuits for wye connected load are

shown in Fig. 3.3.







The a to b line voltage is vab =√3 van with a phase advance of 30°. There is a delay of π/6

between the turning off switch 1 and turning on of switch 4. Thus there should be no short

circuit of the dc supply through one upper and lower switch. At any time, two load terminals

are connected to the dc supply and the third one remains open. The potential of this open

terminal will depend on the load characteristics and would be unpredictable. Since one switch

conducts for 120°, the switches are less utilized as compared to that of 180° conduction for

the load condition.

2.5 Resonant conversion: advantages

The chief advantage of resonant converters: reduced switching loss

Zero-current switching

Zero-voltage switching

Turn-on or turn-off transitions of semiconductor devices can occur at

zero crossings of tank voltage or current waveforms, thereby reducing

or eliminating some of the switching loss mechanisms. Hence

resonant converters can operate at higher switching frequencies than

comparable PWM converters

Zero-voltage switching also reduces converter-generated EMI

Zero-current switching can be used to commutate SCRs

In specialized applications, resonant networks may be unavoidable

High voltage converters: significant transformer leakage

inductance and winding capacitance leads to resonant network

CHAPTER - III

3.1 PROPOSED INVERTER

Proposed inverter circuit

3.1.1. Hard switching and Soft Switching Techniques

In the 1970’s, conventional PWM power converters were operated in a switched mode

operation. Power switches have to cut off the load current within the turn-on and turn-off

times under the hard switching conditions. Hard switching refers to the stressful switching

behavior of the power electronic devices. The switching trajectory of a hard-switched power

device is shown in Fig.1. During the turn-on and turn-off processes, the power device has to

withstand high voltage and current simultaneously, resulting in high switching losses and

stress. Dissipative passive snubbers are usually added to the power circuits so that the dv/dt

and di/dt of the power devices could be reduced, and the switching loss and stress be diverted

to the passive snubber circuits. However, the switching loss is proportional to the switching

frequency, thus limiting the maximum switching frequency of the power converters. Typical

converter switching frequency was limited to a few tens of kilo-Hertz (typically 20 kHz to 50

kHz) in early 1980’s. The stray inductive and capacitive components in the power circuits

and power devices still cause considerable transient effects, which in turn give rise to

electromagnetic interference (EMI) problems. Fig.2 shows ideal switching waveforms and

typical practical waveforms of the switch voltage. The transient ringing effects are major

causes of EMI.

In the 1980’s, lots of research efforts were diverted towards the use of resonant

converters. The concept was to incorporate resonant tanks in the converters to create

oscillatory (usually sinusoidal) voltage and/or current waveforms so that zero voltage

switching (ZVS) or zero current switching (ZCS) conditions can be created for the power

switches. The reduction of switching loss and the continual improvement of power switches

allow the switching frequency of the resonant converters to reach hundreds of kilo-Hertz

(typically 100kHz to 500kHz). Consequently, magnetic sizes can be reduced and the power

density of the converters increased. Various forms of resonant converters have been proposed

and developed. However, most of the resonant converters suffer several problems. When

compared with the conventional PWM converters, the resonant current and voltage of

resonant converters have high peak values, leading to higher conduction loss and higher V

and I ratings requirements for the power devices. Also, many resonant converters require

frequency modulation (FM) for output regulation. Variable switching frequency operation

makes the filter design and control more complicated.

In late 1980’s and throughout 1990’s, further improvements have been made in

converter technology. New generations of soft-switched converters that combine the

advantages of conventional PWM converters and resonant converters have been developed.

These soft-switched converters have switching waveforms similar to those of conventional

PWM converters except that the rising and falling edges of the waveforms are ‘smoothed’

with no transient spikes. Unlike the resonant converters, new soft-switched converters usually

utilize the resonance in a controlled manner. Resonance is allowed to occur just before and

during the turn-on and turn-off processes so as to create ZVS and ZCS conditions. Other than

that, they behave just like conventional PWM converters. With simple modifications, many

customized control integrated control (IC) circuits designed for conventional converters can

be employed for soft-switched converters. Because the switching loss and stress have been

reduced, soft-switched converter can be operated at the very high frequency (typically

500kHz to a few Mega-Hertz). Soft-switching converters also provide an effective solution to

suppress EMI and have been applied to DC-DC, AC-DC and DC-AC converters. This

chapter covers the basic technology of resonant and soft-switching converters. Various forms

of soft-switching techniques such as ZVS, ZCS, voltage clamping, zero transition methods

etc. are addressed. The emphasis is placed on the basic operating principle and practicality of

the converters without using much mathematical analysis.

Fig.Typical switching trajectories of power switches.

Fig.. Typical switching waveforms of (a) hard-switched and (b) soft-switched devices

3.2. Soft switching

Soft switching can mitigate some of the mechanisms of switching loss

and possibly reduce the generation of EMI

Semiconductor devices are switched on or off at the zero crossing of

their voltage or current waveforms:

Zero-current switching: transistor turn-off transition occurs at zero

current. Zero-current switching eliminates the switching loss

caused by IGBT current tailing and by stray inductances. It can

also be used to commutate SCR’s.

Zero-voltage switching: transistor turn-on transition occurs at

zero voltage. Diodes may also operate with zero-voltage

switching. Zero-voltage switching eliminates the switching loss

induced by diode stored charge and device output capacitances.

Zero-voltage switching is usually preferred in modern converters.

Zero-voltage transition converters are modified PWM converters, in

which an inductor charges and discharges the device capacitances.

Zero-voltage switching is then obtained.

ZCS turn-on transition: hard switching

3.3. switched PWM, and switching loss occurs.

Resonant Switch

Prior to the availability of fully controllable power switches, thyristors were the major power

devices used in power electronic circuits. Each thyristor requires a commutation circuit,

which usually consists of a LC resonant circuit, for forcing the current to zero in the turn-off

process. This mechanism is in fact a type of zero-current turn-off process. With the recent

advancement in semiconductor technology, the voltage and current handling capability, and

the switching speed of fully controllable switches have significantly been improved. In many

high power applications, controllable switches such as GTOs and IGBTs have replaced

thyristors. However, the use of resonant circuit for achieving zero-current-switching (ZCS)

and/or zero-voltage-switching (ZVS) has also emerged as a new technology for power

converters. The concept of resonant switch that replaces conventional power switch is

introduced in this section.

A resonant switch is a sub-circuit comprising a semiconductor switch S and resonant

elements, Lr and Cr. The switch S can be implemented by a unidirectional or bidirectional

switch, which determines the operation mode of the resonant switch. Two types of resonant

switches, including zero-current (ZC) resonant switch and zero-voltage (ZV) resonant

switches, are shown in Fig.3 and Fig.4, respectively.

Fig. Zero-current (ZC) resonant switch.

Fig.4 Zero-voltage (ZV) resonant switch.

3.4. ZC resonant switch

In a ZC resonant switch, an inductor Lr is connected in series with a power switch S in

order to achieve zero-current-switching (ZCS). If the switch S is a unidirectional switch, the

switch current is allowed to resonate in the positive half cycle only. The resonant switch is

said to operate in half-wave mode. If a diode is connected in anti-parallel with the

unidirectional switch, the switch current can flow in both directions. In this case, the resonant

switch can operate in full-wave mode. At turn-on, the switch current will rise slowly from

zero. It will then oscillate, because of the resonance between Lr and Cr. Finally, the switch

can be commutated at the next zero current duration. The objective of this type of switch is to

shape the switch current waveform during conduction time in order to create a zero-current

condition for the switch to turn off.

3.5. ZV resonant switch

In a ZV resonant switch, a capacitor Cr is connected in parallel with the switch S for

achieving zero-voltage-switching (ZVS). If the switch S is a unidirectional switch, the voltage

across the capacitor Cr can oscillate freely in both positive and negative half-cycle. Thus, the

resonant switch can operate in full-wave mode. If a diode is connected in anti-parallel with

the unidirectional switch, the resonant capacitor voltage is clamped by the diode to zero

during the negative half-cycle. The resonant switch will then operate in half-wave mode. The

objective of a ZV switch is to use the resonant circuit to shape the switch voltage waveform

during the off time in order to create a zero-voltage condition for the switch to turn on.

CHAPTER – IV

4.1. RESONANT CONVERTER TOPOLOGIES

4.1.1. INTRODUCTION

Increasing the frequency of operation of power converters is desirable, as it allows the size of

circuit magnetics and capacitors to be reduced, leading to cheaper and more compact circuits.

However, increasing the frequency of operation also increases switching losses and hence

reduces system efficiency. One solution to this problem is to replace the "chopper" switch of

a standard SMPS topology (Buck, Boost etc.) with a "resonant" switch, which uses the

resonances of circuit capacitances and inductances to shape the waveform of either the

current or the voltage across the switching element, such that when switching takes place,

there is no current through or voltage across it, and hence no power dissipation - see figure 1.

A circuit employing this technique is known as a resonant converter (or, more accurately, a

quasi-resonant converter, as only part of the resonant sinusoid is utilized).

A Zero Current Switching (ZCS) circuit shapes the current waveform, while a Zero Voltage

Switching (ZVS) circuit shapes the voltage waveform.

4.2. ZERO CURRENT SWITCH

A typical Zero Current Switch consists of a switch, S, in series with the resonant inductor

LRES, and the

resonant capacitor CRES connected in parallel. Energy is supplied by a current source.The

circuit and waveforms are shown in figure 2.

If an output transformer is used, in certain cases its parasitic inductance can be used as the

resonant inductance (in both this and the zero voltage topology). However, as its value is

generally not known, the resonant frequency will not be fixed, which may cause problems in

the circuit design.

When the switch S is off, the resonant capcitor is charged up with a more or less constant

current, and so the voltage across it rises linearly.

When the switch is turned on, the energy stored in the capacitor is transferred to the inductor,

causing a sinusoidal current to flow in the switch. During the negative half wave, the current

flows through the anti-paralleled diode, and so in this period there is no current through or

voltage across the switch; and it can be turned off without losses.

This type of switching is also known as thyristor mode, as it is one of the more suitable ways

of using thyristors; these devices will only turn off if the current through them is forced to

zero, which occurs naturally in this topology.

4.3. ZERO VOLTAGE SWITCH

A typical Zero Voltage Switch consists of a switch in series with a diode. The resonant

capacitor is connected in parallel, and the resonant inductor is connected in series with this

configuration. A voltage source connected in parallel injects the energy into this system. The

circuit and waveforms are shown in figure 3.

When the switch is turned on, a linear current flows through the inductor. When the switch

turns off, the energy that is stored in the inductor flows into the resonant capacitor. The

resulting voltage across the capacitor and the switch is sinusoidal. The negative half-wave of

the voltage is blocked by the diode. During this negative half wave, the current and voltage in

the switch are zero, and so it can be turned on without losses.

4.4. POWER SEMICONDUCTORS IN RESONANT CONVERTERS

Because they require a substantial drive current, Bipolar transistors are not generally used in

resonant converters, unless the base drive is provided by the resonant circuit itself (for

example in TV deflection circuits and fluorescent lamp ballasts). Power MOSFETs and

IGBTs, with their effectively capacitive inputs and low drive energy requirements, are the

most frequently used types.

The graph in figure 4 shows the die size of Power MOSFETs and IGBTs required to conduct

1 amp with a voltage drop of 2 volts, against the maximum rated voltage. For low voltage

applications, the MOSFET is interesting, as the die size is very small (and so the device is

cheap). However for higher breakdown voltages, the IGBT is more suitable, as the die size

required is almost constant approaching the maximum rated voltage.

4.4.1. MOSFETs

The MOSFET has a resistive behavior in its on state, and the output characteristic passes

through zero. It can conduct a small current with a very low voltage drop.

4.4.2. Zero Current Switch

A MOSFET can be modelled as an ideal switch with a series resistance, and a capacitor

connected in parallel - see figure 5. Every time it is turned on, the parallel capacitor is

discharged through the resistance and (0.5 x Cout x VDS) units are lost. A MOSFET in a Zero

Current Switch will have to turn on with a high drain-source voltage, and there will be

capacitive switching losses. Additionally, the reactive overcurrent in the switch is very high,

and as the MOSFET does not perform well in overcurrent conditions, the conduction

losses will be very high. Therefore the MOSFET is not very suitable as a Zero Current

Switch.

4.4.3. Zero Voltage Switch

In a Zero Voltage Switch, the MOSFET turns on without any voltage between drain and

source, and so there are no capacitive switching losses. There is no reactive overcurrent and

the conduction losses are not very important. The MOSFET does have to turn off a current,

but as the switching times of a MOSFET are small, the turn off losses will not be excessive.

4.4.4. IGBTs

The IGBT has a threshold voltage of around 0.7V; a voltage drop lower than this value is not

possible. The “resistive part” of the output characteristics of an IGBT is very low, and so it

can conduct large currents with a low voltage drop. It is thus most suitable for use at high

current densities.

An IGBT can be simply modelled as a pnp-transistor driven by a MOSFET. The

disadvantage of this structure is the turn off. If a pnp transistor is to be turned off quickly, a

positive base current must be supplied, to force the carriers in the base to recombine and stop

the device conducting. In the IGBT, the base of the pnp stage cannot be accessed directly,

and so this current cannot be delivered at turn off, meaning that the device continues to

conduct while the carriers recombine "naturally". During this time, a current tail appears.

4.5. Zero Voltage Switch

In a Zero Voltage Switch, the IGBT must turn off a current. Even if the voltage across the

switch rises with a limited dV/dt (sinusoidal waveform), the current tail phenomenon means

that turn off losses will be important. Therefore the IGBT is not very suitable for zero voltage

switching.

4.6. Zero Current Switch

In a Zero Current Switch, the external circuit defines the current in the switch. This current

tends to zero, and hence the IGBT does not turn off current, so no tail appears. Another

problem that can occur with the IGBT, latching, does not occur in this mode. Even if the

IGBT latches at the maximum current, it can turn off later because the current is defined by

the external circuit. The carriers that remained in the base of the pnp-transistor can be

recovered by a positive current into the base. In a Zero Current Switch, the negative half

wave of the resonant current flows through the antiparallel diode. During that time, a negative

voltage is applied to the IGBT. A current flows through the body diode of the internal

MOSFET into the base of the pnp-transistor.

CHAPTER - V

5.1 Modes of operations

5.1.1. Mode 1 (t0t<t1):

The resonant capacitor is discharged through resonant path Cr and Lr. The resonant inductor

current begins to increase linearly from zero. Therefore, the main switch is turned on with

ZCS influenced by resonant inductor. The energy of the main inductor is delivered to the load

through the switches (S1, S4). The next mode is started as soon as the resonant capacitor has

fully discharged. In this mode, the main inductor current is given by

5.1.2. Mode 2 (t1t<t2):

When the resonant capacitor is fully discharged to Mode1, the anti-parallel diodes in inverter

switches constitute current path. At this time, the inverter stage is zero. When the inverter

stage is in the zero voltage condition, the inverter switches are given to PWM signal (S1, S4).

So, inverter switches are turned on and off with ZVS. This mode isG maintained when the

main switch is turned-off. The main inductor current is given by

5.1.3. Mode 3 (t2˺t<t3):

When the switch is turned off with ZVS, the resonant inductor releases energy. Thus, the

bridge diode is turned on. The dc-link capacitor is transferred to main inductor and resonant

inductor energy through the bridge diode. The resonant capacitor starts to charge main

inductor energy. At that time, the resonant inductor current and resonant capacitor voltage are

given by

5.1.4. Mode 4 (t3˺t<t4):

When the resonant inductor energy is fully released, the bridge diode is turned-off. This mode

is maintained until the resonant capacitor voltage becomes 400[V]. The main inductor

current flows continuously through the inverter switches (S1, S4). In this mode, resonant

capacitor voltage is given by

5.1.5. Mode 5 (t4 ื t<t5):

This mode is started when the resonant capacitor is fully charged. After that, the auxiliary

switch is turned on with ZVS because the switch voltage is zero. When the main inductor

current decreases linearly, the dc-link capacitor is charged from the main inductor energy.

Because the main inductor current flows through the anti-parallel diode, the auxiliary switch

voltage is zero voltage. When the auxiliary switch current path is changed, the next mode

starts. In this mode, the main inductor current can be expressed as

5.1.6. Mode 6 (t5 ื t<t6):

In this mode, the auxiliary switch current path is changed because the dc-link capacitor starts

to discharge. Therefore, the load is supplied the energy by the dc-link capacitor and main

inductor. This mode maintains that the main inductor current equal to the resonant current.

The main inductor current in this mode is expressed as

Output waveforms

5.2. Applications

This type of inverter produces an approximately sinusoidal waveform at a high output

frequency, ranging from 200 to 100 kHz, and is commonly used in relatively fixed ouput

applications, for example, induction eating, sonar transmitter, fluorescent lighting, or

ultrasonic generators. Due to the high switching frequency, the size of the resonating

components is small.

CONCLUSION

In this project, we proposed a novel DC-AC single phase resonant inverter using soft-

switching boost converter. In this topology, all switches perform a soft switching by

resonance between the resonant inductor and capacitor. So, the proposed topology can reduce

the switching loss and voltage stress. The proposed inverter is analyzed through the operation

mode, and its validity is proven through simulation.

REFERENCES

D. M. Divan, G. Skibinski, ˈ Zero switching loss inverters for high

power applications,ˉ IEEE IAS Rec., pp. 627-634, 1987.

S.R. Park, S.H. Park, C.Y. Won, Y.C. Jung, "Low loss soft switching

boost converter", 2008 13th International Power Eletronics and Motion

Control Conference (EPE-PEMC 2008) pp. 181 - 186. 2008

K. Wang, G. Hua, and F. C. Lee, ˈ "Analysis, design and ZCS-PWM

boost converters” in Proc. IEEJ Int. Power Electronics Conf.,

pp.1202-1207, 1995

G. Hua, X. Yang, Y. Jiang, and F.C. Lee, “Novel zero-current-transition

PWM converter”, IEEE Power Electronics Specialist Conf. Rec., pp.

538~544, 1993.

H. Ohashi, ‘‘Power Electronics Inovation with Next Generation

Advanced Power Devices’’, Proceeding of INTELEC'03, Oct.2003,

Yokohama (Japan), pp.9-13

D. M. Divan and G. Skibinski, “Zero switching loss inverters for high

power applications,” IEEE IAS Rec., pp. 627-634, 1987.

a novel dc-ac single phase resonant inverter usingsoft switching boost converte

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