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Date of publication xxxx 00, 0000, date of current version xxxx 00, 0000.

Digital Object Identifier 10.1109/ACCESS.2017.Doi Number

A Wearable Textile RFID Tag Based on an Eighth-Mode Substrate Integrated Waveguide Cavity

Giovanni A. Casula1, Member, IEEE, Giorgio Montisci1, Senior Member, IEEE, and Hendrik Rogier2, Senior Member, IEEE 1Dipartimento di Ingegneria Elettrica ed Elettronica, Università degli Studi di Cagliari, Cagliari, 09123 Italy 2IDLab-EM group, imec-Ghent University, B-9052 Gent, Belgium

Corresponding author: Giorgio Montisci (e-mail: giorgio.montisci@unica.it).

ABSTRACT A novel wearable textile Radio Frequency Identification (RFID) tag based on an eighth-mode

substrate integrated waveguide cavity is presented. Antenna size reduction for effective operation in the [865-

870]-MHz RFID UHF band is obtained by exploiting the H-field symmetry planes of a cylindrical Substrate

Integrated Waveguide (SIW) cavity. High isolation from the human body and excellent robustness with

respect to variations in antenna-body distance are achieved using an energy-based design strategy, aiming to

reduce ground plane size. The resulting tag exhibits very low manufacturing complexity and may be produced

at low-cost. Design and simulations were performed using CST Microwave Studio, and a prototype of the

tag has been manufactured and tested in a real environment.

INDEX TERMS Eighth-mode substrate integrated waveguide (EMSIW), RFID, substrate integrated

waveguide (SIW), textile antennas, wearable antennas.

I. INTRODUCTION

Mechanical robustness, easy manufacturing, low-cost,

reduced size, light weight, flexibility, reliability in the

proximity of the human body, and a low Specific Absorption

Rate (SAR) are the main features that distinguish antennas

for wearable devices from “conventional” antennas and that

allow them to be efficiently integrated into a garment.

Though the most common applications of Radio

Frequency Identification (RFID) technology are within

logistics, retail, transportation, inventory management,

manufacturing and security [3], [4], recently interest is also

growing for sensor networks [5]-[7], personal healthcare [8]-

[10] and entertainment [11], [12]. Passive RFID tags are

particularly suitable for these applications, since they do not

require maintenance or regular recharging. Moreover, they

usually have a long life and low production cost [3]. The tag

device consists of a radio-frequency antenna with integrated

microchip transponder, and, potentially, with additional

appropriate sensors. In several applications, such as in body

area networks or personal area networks, it must operate in

proximity of, or attached to, the human body. In this regard,

UHF wearable RFID systems are critical, since tag antennas

must backscatter the power received from an interrogator

while being deployed close to an extremely lossy platform,

the human body. In addition, the tag antenna size must be

kept as small as possible in the RFID frequency bands (from

865 to 970 MHz including European, US and Asia sub-

bands), to ensure the comfort of the wearer, as required by

on-body devices.

Several wearable RFID tag antennas have been proposed

in recent literature [13]-[23]. Many of them have been

designed using textile substrates, thus easing integration into

clothes and garments [13], [16], [17], [22], [23]. These

antennas exhibit low efficiency and gain due to the strong

proximity of the human body. In addition, the robustness

with respect to the distance from the human body [18], [20],

[22] and the read range exhibit high variability, depending

on the deployment conditions on the human body.

Recently, to increase the antenna-body isolation, wearable

textile antennas based on Substrate Integrated Waveguide

(SIW) cavities have been proposed in the 2.4 GHz ISM band

and at higher frequencies [24]-[36]. These structures are

particularly suitable for wearable applications since, in

addition to the good isolation provided by the SIW

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technology, they ensure high flexibility [24], [26], [27],

wideband/multiband operation [26]-[27], a low-profile

planar structure and good radiation characteristics.

Furthermore, antennas based on SIW cavities can be easily

miniaturized by exploiting the symmetry of the field

distributions of their resonant modes [24]-[36].

SIW antennas have widely demonstrated their potential for

several applications at 2.4 GHz and beyond, using either

coaxial cable or microstrip line feeding [24]-[47], leading to

compact footprints. However, to the best of the authors’

knowledge, up to now SIW technology has never been

exploited for the design of a RFID passive wearable tag,

especially in the lower part of UHF band (around 900 MHz),

at which miniaturization remains a key challenge.

In this work, we propose a wearable textile RFID tag

antenna based on an eighth-mode substrate integrated

waveguide circular cavity (EMSIW), operating in the

European UHF band (865-870 MHz). The EMSIW

configuration minimizes the antenna size, while its ground

plane is optimized, using the energy-based design strategy

proposed in [37]-[41], to mitigate the deterioration in antenna

performance due to the body coupling.

The Impinj Monza 4 chip has been connected to the

EMSIW cavity antenna. This microchip was appropriately

positioned to minimize the perturbation of the electromagnetic

field within the cavity, and to maximize the overall antenna

efficiency.

A common closed-cell rubber foam is selected as a

substrate, and adhesive copper coated non-woven PET fabric

is employed for the metallization. The resulting tag antenna

has a compact size (8.73 × 7.78 × 0.4 cm3, which is 0.24 × 0.22

× 0.011 λ03, with λ0 the free-space wavelength at 868 MHz),

yielding significantly smaller electrical dimensions compared

to similar wearable SIW antennas described in the literature

[24]-[36] (see Table I for a comparison), realizing a volume

reduction by at least a factor of 7, and an area reduction by at

least a factor of 2.5, in terms of free-space wavelength.

The antenna design and all the electromagnetic field

simulations have been performed using CST Microwave

Studio. Measurements have been performed in a real

environment using a commercial RFID reader, showing high

body-antenna isolation and good robustness with respect to

variations of the antenna-body distance.

II. ANTENNA TOPOLOGY AND MATERIALS

The proposed SIW textile wearable antenna is designed for

RFID Applications in the European UHF band, with a center

frequency of 868 MHz. A 4 mm-thick closed-cell rubber

foam, typically used in firefighter suits, has been selected as

dielectric substrate for the SIW antenna. Its dielectric

permittivity equals 1.3, with a loss tangent tanδ = 0.03

(measured by means of the microstrip T-resonator method

[48]). The metallization has been implemented using adhesive

copper coated non-woven PET fabric, with a sheet resistivity

of 0.04 Ω/square and a thickness of 0.11 mm. By following

the formulas for SIW circular cavities specified in [49] as a

starting point, we have designed a SIW cylindrical resonant

cavity, operating in the TM010 mode at 868 MHz (Fig. 1a). The

cavity diameter equals AB = 180 mm (Fig.1a). The radius of

the vias is RH = 5 mm, with a spacing Sy = 16 mm (Fig. 2).

Next, we have taken advantage of the magnetic field

symmetry of the TM010 mode to reduce the size of the cavity.

First, a virtual magnetic wall has been placed along the

horizontal symmetry plane of the SIW cavity, indicated by the

line AOB in Fig. 1a. Then, the half of the cavity below the line

AOB has been cut off, keeping a suitable ground plane

extension Ge.

TABLE I

COMPARISON BETWEEN THE WEARABLE SIW PROPOSED IN THE

LITERATURE AND OUR RIFD TAG

Ref.

Operating

Frequency (GHz)

Antenna Size

Body

effect

Feeding

[24] 2.45 1.03 × 1.07 × 0.032 λ03 Yes Microstrip

[25] 4.45 1.67 × 1.25 × 0.053 λ03 No Coax

[26] 2.45 1.02 × 0.76 × 0.032 λ03 Yes Microstrip

[27] 2.45 0.53 × 0.44 × 0.030 λ03 Yes Coax

[28] 5.20 0.61 × 0.69 × 0.027 λ03 No Microstrip

[29] 2.45 0.49 × 0.49 × 0.030 λ03 Yes Coax

[30] 2.45 0.49 × 0.33 × 0.027 λ03 No Microstrip

[31] 2.45 0.61 × 0.39 × 0.027 λ03 Yes Coax

[32] 5.8 1.45 × 0.81 × 0.019 λ03 Yes Microstrip

[33] 5.8 0.37 × 0.37 × 0.030 λ03 Yes Microstrip

[34] 5.8 1.26 × 0.77 × 0.030 λ03 Yes CPW

[35] 5.8 0.91 × 0.80 × 0.030 λ03 Yes Microstrip

[36] 2.45 0.41 × 0.41 × 0.029 λ03 Yes Coax

This work 0.868 0.24 × 0.22 × 0.011 λ03 Yes MicroCHIP

FIGURE 1. Design evolution: the symmetry of the first resonant mode is exploited to obtain the EMSIW structure. Full-Mode SIW (a); Half-Mode SIW (b); Quarter-Mode SIW (c); Eighth-Mode SIW (d); Electric field distribution of the TM010 mode for the Full-Mode circular SIW cavity as a function of applied field (e).

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In this way, a semi-circular cavity, resonating in half-

mode SIW (HMSIW) operation, is obtained (Fig. 1b). Then,

the same procedure has been applied to the semi-circular

cavity, which has been halved by exploiting the vertical

magnetic field symmetry plane indicated by the line CO in

Fig. 1b. The result is the quarter-mode SIW resonator

(QMSIW) shown in Fig. 1c. Finally, the latter has been

further halved, cutting the QMSIW along the symmetry

plane OD, to generate the eighth-mode SIW resonator

(EMSIW) of Fig. 1d. The circular EMSIW operates as an

effective broadside antenna, radiating an electric field

through the open side walls with a maximum directed along

the z-axis, propagating away from the wearer. III. ANTENNA DESIGN AND SIMULATIONS

Following the design process described in Fig. 1, we obtain

the antenna topology shown in Fig. 2, in which all the

relevant geometrical parameters are clearly indicated.

FIGURE 2. Layout of the designed SIW antenna.

The radiation characteristics of wearable antennas are

strongly influenced by the proximity of the human body,

which is a lossy and non-homogeneous material. Since the

distance d between the body and the antenna changes

randomly during actual operating conditions, it is important

to maintain a stable performance regardless of the distance

from the human body.

A numerical phantom has been added to the simulation

scenario (Fig. 3), in order to investigate the effect of the

body-antenna coupling. We have chosen a simplified

phantom, consisting of a single layer with muscle-like

dielectric properties at 868 MHz (εr = 56.6, σ = 1.33 S/m)

and with a size of 250 × 200 × 100 mm3 [41].

FIGURE 3. CST Microwave Studio model of the designed antenna with the single layer body phantom: a) 3D view; b) side view.

The design has been performed using CST Microwave

Studio through an optimization procedure with the antenna

attached to the body phantom, being for d = 0 (see Fig. 3).

The main points of the design procedure, leading to the

configuration of Fig. 2, are:

i) The addition of a slot of length LS, cut out in the top

patch. The slot lengthens the current path, further

reducing the antenna size. Increasing LS reduces the

operating frequency, while the slot offset XS can be used

to fine tune the frequency. The slot width WS is set to 2.5

mm. The optimized values at 868 MHz are LO = 77.8

mm, LV = 87.3 mm, LS = 21.7 m, and XS = 19.9 mm.

ii) The selection of a suitable feeding point for the

connection of the microchip. The microchip has been

placed along the diagonal side of the SIW cavity, as

indicated in Fig. 2, since this location does not affect the

electromagnetic field inside the cavity and allows easy

matching of the antenna to the input impedance of the

microchip (in our case a Impinj Monza 4 with an input

impedance Zchip equal to 13-j151 Ω at 868 MHz).

Conjugate matching to Zchip is achieved for ∆Lx_chip =

24.3 mm, ∆Ly_chip = 14.4 mm.

iii) The choice of the ground plane and dielectric substrate

extensions Ge (Fig. 1d). Since the ground plane shields

the antenna from the human body, a large ground plane

improves the antenna robustness with respect to the

human body proximity but increases the antenna

dimensions. Therefore, an appropriate extension Ge of

both the ground plane and the dielectric substrate should

be kept in proximity of each virtual magnetic wall (see

Fig. 1d) to achieve a trade-off between antenna

robustness and its dimensions [37-41]. Table II reports

the antenna input impedance Zin, the power reflection

coefficient τ = 4 Re (Zchip) Re (Zin)/|Zchip* + Zin|2, and its

percentage variation with respect to d = 0 for three cases

(see Fig. 4):

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a) No ground plane extension (Ge = 0 mm): LG1 = LG2

= LG3 = 0 (Fig. 4a);

b) Ground plane extension of Ge = 15 mm along all the

antenna sides: LG1 = LG2 = LG3 = 15 mm (Fig. 4b);

c) Ground plane extension of Ge = 10 mm along the

vertical side and no ground plane extension along

the diagonal side: LG1 = LG2 = 10 mm, LG3 =0 (Fig.

4c).

FIGURE 4. Different ground plane extensions for the proposed RDIF tag: a) No ground plane extension; b) Ground plane extension of Ge = 15 mm along all the antenna sides; c) Ground plane extension of Ge = 10 mm along the vertical side and no ground plane extension along the diagonal side.

From the results of Table II, it appears that a reasonable

trade-off between antenna dimensions and robustness is

achieved in case c), resulting in the layout in Fig. 2, with

LG1 = 10 mm and LG2 = 10 mm.

TABLE II

COMPARISON BETWEEN THE RFID TAG PERFORMANCE FOR

DIFFERENT GROUND PLANE EXTENSIONS

case d [mm] Zin [Ω] τ ∆τ [%]

a)

0 17 + j150 0.988 0

10 8.2 + j142.1 0.81 18.2

20 7.2 + j141.2 0.73 26.1

30 6.5 + j140.7 0.7 29.25

b)

0 9.9 + j151.5 0.981 0

10 9.3 + j151.2 0.973 0.81

20 8.5 + j151.1 0.96 2.1

30 8.2 + j151.2 0.952 2.9

c)

0 14.5 + j151 0.996 0

10 10.5 + j150.9 0.981 1.51

20 9 + j150.6 0.96 3.61

30 8.5 + j150.5 0.94 5.62

The simulated frequency response of the designed antenna

is shown in Fig. 5.

FIGURE 5. Simulated frequency response of the antenna in Fig. 2 for d = 0 (antenna attached to the body).

To assess the antenna robustness according to the energy-

based design consideration of [37]-[41], in Fig. 6 the electric

energy density distributions in the antenna substrate for d =

0 (antenna attached to the body) and d = 30 mm are reported

at 868 MHz. The energy distributions are quite similar,

confirming the low sensitivity of the designed antenna with

respect to the human body proximity.

In Fig. 7, the simulated power transmission coefficient τ

vs. frequency is shown for different spacings d between the

antenna and the body phantom and, in Fig. 8, the variation of

τ and the total efficiency τ × η (η being the radiation

efficiency), as a function of the antenna-body separation d,

are reported at 868 MHz. From Figs. 7 and 8, is apparent that

the transmission coefficient is quite stable with respect to

different antenna-body separations. The antenna bandwidth

is quite large: τ > 0.8 in the range 850-890 MHz for d = 0,

and in the range 859-885 MHz for d = 30 mm. The total

efficiency is about 16% for d ∈ [0 - 30 mm], which is a good

value for a textile wearable RFID tag, due to the high

dissipation in the body tissues and in the low-cost materials

used to manufacture the tag (being the rubber foam and the

textile conductive fiber). Wearable antennas with textile

substrates available in the literature [13], [16], [17], [22],

[23] provide lower values of efficiency (around 14% or less).

However, a comparison with our configuration is not

consistent since the efficiency strongly depends on the

materials used. For this reason, we have focused on the

ability of the proposed structure to provide high robustness

and isolation with respect to human body coupling, which is

a feature depending only on the proposed SIW architecture,

and it is confirmed by the results reported in Figs. 7 and 8.

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FIGURE 6. Simulated Electric energy density distribution in the antenna substrate: RFID tag attached to the body (d = 0) (a); RFID tag at d = 30 mm from the body (b).

FIGURE 7. Simulated power transmission coefficient τ for different

spacings of the antenna from the body phantom.

The CST Microwave Studio simulations confirm that the

tag antenna’s polarization is nearly linear along the y-axis

when it is rotated by α = 20° in the xy-plane (see Fig. 9). The

simulated surface current for d = 0 is also depicted in Fig. 9.

In Fig. 10, the normalized radiation pattern is reported for

α = 20°, for the antenna attached to the body (d = 0) and for

a distance of d = 30 mm from the phantom, showing a cross-

polarization in the broadside direction below -15 dB in both

the principal planes.

FIGURE 8. Simulated power reflection coefficient τ, radiation efficiency

η and total efficiency τ × η, as a function of body-antenna separation d, at 868 MHz.

FIGURE 9. Surface current of the designed SIW antenna at 868 MHz for d = 0.

FIGURE 10. Simulated radiation pattern of the designed SIW antenna at

868 MHz for α = 20° (see Fig. 8).

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Finally, table III reports the simulated Directivity and

Gain in the broadside direction, at 868 MHz, of the SIW

antenna fed by a discrete port instead of the microchip, for

different body-antenna separations d.

TABLE III

SIMULATED DIRECTIVITY AND GAIN OF THE SIW

ANTENNA AT 868 MHZ

d [mm] Directivity

[dB] Gain [dBi]

0 6.15 -2.60

10 5.70 -2.85

20 5.40 -3.05

30 5.65 -2.80

IV. EXPERIMENTAL VERIFICATION

A prototype of the designed SIW RFID tag of Fig. 2 has

been manufactured. The commercial chip Impinji Monza 4

has been connected to the antenna, soldering it to the patch

at one side, and to a via connected to the ground plane at the

other side. The antenna prototype is depicted in Fig. 11.

To assess the robustness of this RFID tag, an artificial

model of the human tissue has been synthesized. The human

tissue has been experimentally simulated by using a

simplified phantom, consisting of a PVC tank of dimension

25 × 20 × 10 cm3, having a thickness of 1 mm, filled with a

tissue-simulating liquid with muscle-like parameters at 870

MHz (εr = 56.6, σ = 1.33 S/m), consisting of deionized water

(53%), saccharose (45.6%) and sodium chloride (1.4%) [41].

The measurement setup has been deployed in a classroom

at the University of Cagliari (Fig. 11). The floor of the

premises is made of tiles over precast concrete and,

therefore, we can assume as refractive index of the floor a

value around 2.5 [3]. The ceiling is 3.5-meter high and the

side walls are 4-meter far from the tag. In our case, their

effect has been neglected.

A vertically polarized field incident on a horizontal floor

will experience no reflection at the Brewster’s Angle, which

is around 68° in our case (measured w.r.t. the vertical). In

this condition, reflection loss is above 15 dB for an incidence

between 58° and 75°. For a reader antenna placed 0.8 m

above the floor, the Brewster’s angle reflection point is 2 m

away, so the specularly reflected location is 4 m away. The

reflection point for incident angles of 58° and 75° is 1.3 m

and 3 m away from the reader, respectively. Therefore, for

RFID interrogator/tag separations between 2.6 m and 6 m the

reflection loss remains above 15 dB. As we will see in the

following, these values are within the measured read range

of our RFID tag.

Based on the above considerations, the antenna is oriented

in order to receive a nearly vertical polarization along the y-

axis (α = 20° in Fig. 9), with the floor lying in the xz-plane.

The tag is attached to the body phantom and placed in front

of the commercial UHF reader Zebra RFD8500 Handheld

[39], fixed on a mobile mount, and remote-controlled by a

smartphone application via Bluetooth® wireless technology.

The spacing between the antenna and the phantom has been

modified by using pads of different thickness of expanded

polystyrene (EPS), with dielectric permittivity close to 1.

The read range has been measured for different spacings

between the antenna and the body phantom, and the results

of the experimental verification, shown in Fig. 11, have been

compared with the theoretical read range computed using the

expression [50]:

= λπ ∙∙∙ (1)

The transmitter power is set to PCP = 30 dBm (the reader

antenna radiates circular polarization) and Gt = 5.15 dB,

whereas the read sensitivity of the IC Monza 4 equals Pchip=

-17.4 dBm. Gtag and τ have been computed using CST

Microwave Studio.

FIGURE 11. Prototype of the designed SIW antenna and measurement set up.

The average measured read range is about 20% lower than

the theoretical one (4.62 m vs. 5.82 m). This difference is

probably due to the uncertainty in the estimation of the floor

dielectric permittivity and to the path loss caused by the

multipath fading generated by reflections on the ceiling and

side walls. However, as expected, the read range remains

stable when varying the antenna body-distance, which is the

key feature of the proposed configuration, providing a

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validation of the simulated results in Figs. 7 and 8.

FIGURE 12. Simulated and measured read range of the RFID tag in Fig. 10.

V. CONCLUSION

The SIW technology has been employed for the design of a

wearable RFID tag operating in the European UHF band

(865-870 MHz). Miniaturization is a key challenge at these

frequencies. Then, exploiting the magnetic field symmetry

of the TM010 mode of the circular SIW cavity, we have

designed an Eighth-Mode SIW cavity antenna that allows to

achieve a compact size of about 3900 mm2 × 4 mm.

Moreover, thanks to the SIW architecture, the resulting

RFID tag provides high isolation from the human body, and

high robustness with respect to the antenna-body distance

variation. Experimental assessment of the tag performance is

obtained by measurement of the read range when the tag is

housed in a classroom of our University.

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GIOVANNI A. CASULA (M’04) received the

M.S. degree in electronic engineering and the

Ph.D. degree in electronic engineering and computer science from the University of Cagliari,

Cagliari, Italy, in 2000 and 2004, respectively.

Since December 2017, he has been an Associate Professor of electromagnetic fields at the

University of Cagliari, teaching courses in

electromagnetics and antenna engineering. He has authored or coauthored about 50 papers in

international journals. His current research

interests include the analysis and design of waveguide slot arrays, RFID Antennas, wearable antennas, numerical methods in electromagnetics.

Dr. Casula is an associate editor of IET Microwaves, Antennas &

Propagation.

GIORGIO MONTISCI (M’08–SM’19) received the M.S. degree in electronic engineering and the

Ph.D. degree in electronic engineering and

computer science from the University of Cagliari, Cagliari, Italy, in 1997 and 2000, respectively.

Since November 2015, he has been an Associate

Professor of electromagnetic fields at the University of Cagliari, teaching courses in

electromagnetics and microwave engineering. He

has authored or coauthored about 70 papers in international journals. His current research

interests include the analysis and design of waveguide slot arrays, RFID

Antennas, wearable antennas, numerical methods in electromagnetics, and microwave circuits and systems.

Dr. Montisci is an associate editor of IEEE ACCESS, IET Microwaves,

Antennas & Propagation, and an academic editor of the International Journal of Antennas and Propagation.

VOLUME XX, 2017 9

HENDRIK ROGIER (SM’06) received the M.Sc. and Ph.D. degrees in electrical engineering

from Ghent University, Ghent, Belgium, in 1994

and 1999, respectively. From 2003 to 2004, he was a Visiting Scientist with the Mobile

Communications Group, Vienna University of

Technology, Vienna, Austria. He is a currently a Full Professor with the Department of

Information Technology, Ghent University, a

Guest Professor with the Interuniversity Microelectronics Centre, Ghent, and a Visiting

Professor with the University of Buckingham,

Buckingham, U.K. He has authored or co-authored over 160 papers in international journals and over 180 contributions in conference proceedings.

His current research interests include antenna systems, radio wave

propagation, body-centric communication, numerical electromagnetics, electromagnetic compatibility, and power/signal integrity.

Dr. Rogier is a member of Technical Committee 24 on RFID Technology

with the IEEE Microwave Theory and Techniques Society (MTT-S) and a member of the Governing Board of Topical Group MAGEO on Microwaves

in Agriculture, Environment and Earth Observation with the European Microwave Association, Leuven, Belgium. He was a recipient of the URSI

Young Scientist Award (twice) at the 2001 URSI Symposium on

Electromagnetic Theory and at the 2002 URSI General Assembly, the 2014 Premium Award for Best Paper in the IET Electronics Letters, the Best

Paper Award First Place in the 2016 IEEE MTT-S Topical Conference on

Wireless Sensors and Sensor Networks, the Best Poster Paper Award at the 2012 IEEE Electrical Design of Advanced Packaging and Systems

Symposium, the Best Paper Award at the 2013 IEEE Workshop on Signal

and Power Integrity, and the Joseph Morrissey Memorial Award for the First Best Scientific Paper at BioEM 2013. He is an Associate Editor of IET

Electronics Letters, IET Microwaves, Antennas and Propagation, and the

IEEE Transactions on Microwave Theory and Techniques. He acts as the URSI Commission B representative for Belgium.