a simplified approach to multi-carrier modulation
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A SIMPLIFIED APPROACH TO MULTI-CARRIER MODULATION
A Thesis
Submitted to the Faculty
of
Purdue University
by
Andrew C. Marcum
In Partial Fulfillment of the
Requirements for the Degree
of
Master of Science in Engineering
May 2010
Purdue University
Fort Wayne, Indiana
ii
For my grandfather Alan and my grandmothers Dorothy and Rita.
iii
ACKNOWLEDGMENTS
I first thank Dr. Todor Cooklev for the support and leadership he provided me
throughout the last two semesters. I am very thankful for the opportunity to work under
the guidance and teachings of Dr. Cooklev, and I am better engineer for it. Next, I thank
my graduate committee; Dr. Steven Walter, Dr. Carlos Raez and Dr. Tim Grove and
thesis format director, Barbara Lloyd for the time and effort expended on my behalf. I
thank Raytheon Company for supporting my efforts and desires to further my education.
At times, managing the requirements of graduate school in conjunction with a demanding
job can be very stressful and difficult to balance. As such, I am thankful to work for a
company that fosters an environment where education is valued and the goals of its
employees are supported. I thank my family for supporting my dreams and providing me
every possible opportunity to reach this milestone. Last but not least, I thank my fiancรฉ,
Rebecca. Rebecca sacrificed a lot to be with me and has been nothing but supportive
during the many hours I have put into graduate school above and beyond the duties of
work.
iv
TABLE OF CONTENTS
Page
LIST OF TABLES ........................................................................................................................... v
LIST OF FIGURES ........................................................................................................................ vi
LIST OF ABBREVIATIONS ......................................................................................................... ix
ABSTRACT ..................................................................................................................................... x
1. INTRODUCTION ...................................................................................................................... 1
2. CONVENTIONAL SYSTEM .................................................................................................... 3
2.1 SISO System Description .............................................................................................. 3
2.2 SISO OFDM Description ............................................................................................... 5
2.3 MIMO System Description .......................................................................................... 10
2.4 V-Blast MIMO OFDM Description ............................................................................. 14
3. SIMPLIFIED SYSTEM ............................................................................................................ 26
3.1 Simple System Description ........................................................................................... 26
3.2 Simple Discrete Fourier Transform Matrix .................................................................. 27
3.3 Simple Inverse Discrete Fourier Transform Matrix ...................................................... 32
3.4 Simple Fast Fourier Transform Algorithm ................................................................... 34
3.5 Simple Inverse Fast Fourier Transform Algorithm ....................................................... 38
3.6 Simple SISO OFDM ..................................................................................................... 43
3.7 Simple MIMO OFDM .................................................................................................. 47
4. SIMULATION RESULTS ....................................................................................................... 54
4.1 SISO OFDM Architecture ............................................................................................ 54
4.2 MIMO OFDM Architecture .......................................................................................... 70
5. CONCLUSIONS....................................................................................................................... 79
BIBLIOGRAPHY .......................................................................................................................... 83
v
LIST OF TABLES
Table Page
4.1 Rounded FFT/IFFT Twiddle Factor Quantization ......................................................54
5.1 FFT and Rounded FFT Complexity ............................................................................80
5.2 Complexity Reduction Provided by Rounded FFT ....................................................81
vi
LIST OF FIGURES
Figure Page
2.1. SISO OFDM System....................................................................................................4
2.2. SISO OFDM Transceiver Block Diagram ...................................................................5
2.3. MIMO OFDM System ...............................................................................................11
2.4. SISO Capacity vs. MIMO Capacity...........................................................................13
2.5. V-Blast MIMO OFDM Receiver/Transmitter Block Diagram ..................................14
3.1. Response of Five Level Rounded Sine (k=2) ............................................................29
3.2. Response of Seventeen Level Rounded Sine (k=8) ...................................................30
3.3. Four Point Radix-4 FFT Butterfly Diagram ..............................................................36
3.4. Four Point Rounded Radix-4 FFT Butterfly Diagram ...............................................38
3.5. Four Point Radix-4 IFFT Butterfly Diagram .............................................................40
3.6. Four Point Rounded Radix-4 IFFT Butterfly Diagram..............................................43
3.7. Simplified SISO OFDM Transceiver Block Diagram ...............................................44
3.8. Simplified V-Blast MIMO OFDM Receiver/Transmitter Block Diagram ................48
4.1. Flat Fading Channel Frequency Response (Channel 1) .............................................55
4.2. Typical Office Channel Frequency Response (Channel 2)........................................56
4.3. Large Open Area Channel Frequency Response (Channel 3) ...................................57
4.4. SISO OFDM with QPSK BER, k=2, Channel 1 ........................................................58
vii
Figure Page
4.5. SISO OFDM with QPSK BER, k=4, Channel 1 ........................................................58
4.6. SISO OFDM with QPSK BER, k=8, Channel 1 ........................................................59
4.7. SISO OFDM with QPSK BER, k=16, Channel 1 ......................................................59
4.8. SISO OFDM with 16QAM BER, k=2, Channel 1 ....................................................60
4.9. SISO OFDM with 16QAM BER, k=4, Channel 1 ....................................................60
4.10. SISO OFDM with 16QAM BER, k=8, Channel 1 ..................................................61
4.11. SISO OFDM with 16QAM BER, k=16, Channel 1 ................................................61
4.12. SISO OFDM with QPSK BER, k=2, Channel 2 ......................................................62
4.13. SISO OFDM with QPSK BER, k=4, Channel 2 ......................................................62
4.14. SISO OFDM with QPSK BER, k=8, Channel 2 ......................................................63
4.15. SISO OFDM with QPSK BER, k=16, Channel 2 ....................................................63
4.16. SISO OFDM with 16QAM BER, k=2, Channel 2 ..................................................64
4.17. SISO OFDM with 16QAM BER, k=4, Channel 2 ..................................................64
4.18. SISO OFDM with 16QAM BER, k=8, Channel 2 ..................................................65
4.19. SISO OFDM with 16QAM BER, k=16, Channel 2 ................................................65
4.20. SISO OFDM with QPSK BER, k=2, Channel 3 ......................................................66
4.21. SISO OFDM with QPSK BER, k=4, Channel 3 ......................................................66
4.22. SISO OFDM with QPSK BER, k=8, Channel 3 ......................................................67
4.23. SISO OFDM with QPSK BER, k=16, Channel 3 ....................................................67
4.24. SISO OFDM with 16QAM BER, k=2, Channel 3 ..................................................68
4.25. SISO OFDM with 16QAM BER, k=4, Channel 3 ..................................................68
4.26. SISO OFDM with 16QAM BER, k=8, Channel 3 ..................................................69
viii
Figure Page
4.27. SISO OFDM with 16QAM BER, k=16, Channel 3 ................................................69
4.28. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=2 .....................71
4.29. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=4 .....................71
4.30. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=8 .....................72
4.31. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=16 ...................72
4.32. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=2 ..............73
4.33. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=4 ..............73
4.34. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=8 ..............74
4.35. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=16 ............74
4.36. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=2 ..................75
4.37. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=4 ..................75
4.38. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=8 ..................76
4.39. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=16 ................76
4.40. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=2 ...........77
4.41. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=4 ...........77
4.42. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=8 ...........78
4.43. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=16 .........78
ix
LIST OF ABBREVIATIONS
AWGN Additive White Gaussian Noise
BER Bit Error Rate
CP Cyclic Prefix
DFT Discrete Fourier Transform
FFT Fast Fourier Transform
ICI Inter-Carrier Interference
IDFT Inverse Discrete Fourier Transform
IFFT Inverse Fast Fourier Transform
ISI Inter-Symbol Interference
MIMO Multiple Input, Multiple Output
MMSE Minimum Mean Square Error
MRC Maximal Ratio Combining
OFDM Orthogonal Frequency Division Multiplexing
QAM Quadrature Amplitude Modulation
QPSK Quadrature Phase Shift Keying
SIC Successive Interference Cancellation
SISO Single Input, Single Output
SWaP Size Weight and Power
V-Blast Vertical-Bell Laboratories-Layered-Space-Time
ZF Zero Forcing
x
ABSTRACT
Marcum, Andrew C. M.S.E., Purdue University, May 2010. A Simplified Approach to
Multi-Carrier Modulation. Major Professor: Steven Walter.
There is a significant demand for a decrease in the size, weight and power
(SWaP) associated with wireless systems. In recent years, multiple-input, multiple-
output (MIMO) wireless systems have received considerable attention due to the high
data rates they provide. Orthogonal frequency division multiplexing (OFDM), a digital
multi-carrier modulation technique, is well suited to be used in MIMO systems as it
provides the ability to operate in frequency-selective channel environments. When
OFDM is combined with the capacity increase provided by MIMO systems, the result is a
very successful communication system. In this research, a reduced-complexity MIMO
OFDM system is advanced. The proposed system is multiplier-less and thus requires a
simpler digital hardware implementation. As a result, the chip area, power consumption
and cost associated with the MIMO OFDM system can be significantly reduced.
The reduction in complexity is obtained via modification to conventional Fast
Fourier Transform (FFT) and Inverse Fast Fourier Transform (IFFT) algorithms
necessary to implement OFDM multi-carrier modulation. System computational
complexity is reduced by quantizing what are known as โtwiddle factorsโ in traditional
xi
FFT algorithms such as the Radix-2 and Radix-4. The quantization allows for all
multiplications to be done with a value of one, negative one, zero or a power of two.
Ensuring that all multiplications are performed with any of the aforementioned values
results in a transform where all multiplications are considered trivial. Replacing standard
multiplications with trivial multiplications significantly reduces system computational
complexity. As an example, the complexity associated with the implementation of the
rounded FFT as compared to a conventional Radix-4 FFT is reduced by 47% when
numerical values are represented with 16 bits. Depending on the application, different
quantization levels can be utilized in order to obtain the necessary performance
characteristics. As the number of quantization levels grows, the system capability
increasingly approaches the performance of a system that uses the conventional
transforms. When applied to MIMO OFDM systems, the computational savings are
significant as the combination of the IFFT and FFT algorithms are implemented for every
spatial stream (i.e. antenna). As such, the simplified approach provides a system that is a
lower-cost, practical alternative to the MIMO OFDM systems used today.
1
1. INTRODUCTION
The motivation for this research is to determine a solution that allows for a
reduction of computational complexity when applied to the implementations of existing
wireless communication technologies. Specifically, an investigation is performed to
simplify both single-input, single-output (SISO) and multiple-input, multiple-output
(MIMO) multi-carrier modulation systems with Orthogonal Frequency Division
Multiplexing (OFDM). One such method to reduce the complexity associated with the
implementation of SISO and MIMO systems is to simplify the processing of the Fast
Fourier Transform (FFT) and Inverse Fast Fourier Transform (IFFT) as required by
OFDM modulation. As with every simplification, there is a trade associated with system
performance that must occur. In this context, inclusion of the simplified FFT and IFFT
algorithms will result in some reduction of overall system performance when compared
to conventional systems, with the critical parameter affected by the simplification being
bit error rate (BER). In order to assess the impact, this analysis describes the
performance delta between the conventional system and the simplified system derived
from a comprehensive set of computer simulations [1]. The simulations model several
different system configurations, including multiple OFDM bit-to-symbol mapping
techniques. Utilizing an approach that considers many different implementations permits
this research to identify the schemes that provide optimal performance. Furthermore, the
2
results of the computer simulations, when coupled with the analysis of computational
complexity, provide the critical information necessary to determine whether or not the
simplified design can be considered a practical and viable communication system.
3
2. CONVENTIONAL SYSTEM
2.1 SISO System Description
The first system considered in this research utilizes an architecture that consists of
a single transmitter and single receiver, known as SISO. The modulation technique
utilized in conjunction with the SISO architecture is OFDM. OFDM can be characterized
as a digital modulation scheme that multiplexes complex data symbols and transmits the
symbols on multiple carriers that are closely spaced in frequency and orthogonal to one
another (considered a single OFDM symbol) [2, 3, 4]. In this configuration of multi-
carrier modulation, orthogonality between the closely spaced carriers is essential in order
to eliminate crosstalk and cancellation otherwise known as inter-carrier interference (ICI)
[2, 3]. Conventional bit-to-symbol mappings such as Quadrature Phase Shift Keying
(QPSK) and Quadrature Amplitude Modulation (QAM) are employed to obtain the
information transmitted by each carrier. In OFDM modulation, symbols are transmitted
by the carriers at a low rate, thus simplifying the hardware implementation of both the
transmitter and receiver. The combination of the set of low symbol rate carriers (OFDM
symbol) transmitted and received in parallel results in high a data rate system of modest
complexity.
As previously stated, orthogonality between carriers must be ensured in order to
realize the advantages of OFDM modulation. It is well known that the result of
4
computing the Fourier Transform with respect to a set of complex values results in an
orthogonal data set. As such, standard configurations of OFDM utilize the IFFT
algorithm in the transmitter and corresponding FFT in the receiver [1, 2, 3]. The IFFT is
performed for each carrier transmission and provides a time domain representation of the
complex symbols generated as a result of bit-to-symbol mapping schemes. The FFT
algorithm is utilized in the receiver in order to reverse the effects of the IFFT that is
implemented in the transmitter by converting the data into a frequency domain
representation. The frequency domain depiction of the data contains the original
complex symbol data with additional effects of the channel and noise. Because the
symbols are represented in the frequency domain, the removal of the channel
characteristics, known as equalization, is simplified as channel de-convolution can be
implemented by dividing the channel frequency response from each carrier. This method
of equalization is commonly referred to as zero-forcing (ZF) as the original data (with
additive noise) can be obtained simply through one division per OFDM carrier. To
further describe the explanation of SISO OFDM, Figure 2.1 is provided to illustrate the
system.
IFFT P/S
ChannelX(0)
X(1)
X(N-1)
.
.
.
.
.
.
S/P FFT.
.
.
Equalizer.
.
.
.
.
.
Y(0)
Y(1)
Y(N-1)
Fig. 2.1. SISO OFDM System
5
Another advantage of OFDM modulation is its ability to operate in frequency
selective channel conditions that can be harmful to the reliability of high data rate single-
carrier systems. Because of the slow data rate and thus small bandwidth associated with
each carrier, OFDM modulation can operate successfully in frequency selective
environments as the channel response can be considered flat with respect to a specific
carrier. In general, OFDM can be viewed as a set of slowly-modulated narrowband
signals as opposed to one rapidly modulated wideband signal associated with single-
carrier systems.
2.2 SISO OFDM Description
To begin a detailed discussion of the SISO system, consider an OFDM multi-
carrier modulation system with single transmit and receive antennas, as illustrated in
Figure 2.2.
IFFT P/S
X(0)
X(1)
X(N-1)
.
.
.
S/P FFT
Channel
Estimation
.
.
.
.
.
.
Y(0)
Y(1)
Y(N-1)
Add
CP
.
.
.
h +
w
Remove
CP
.
.
.
FFT
\
.
.
.
\
\
.
.
.
Fig. 2.2. SISO OFDM Transceiver Block Diagram
6
It can be quickly observed that the system represented in Figure 2.2 seems significantly
more complex than that in Figure 2.1. The added complexity is due to the fact that
Figure 2.2 includes a reference to the cyclic prefix (CP), channel impulse response model
as well as an expanded view of equalization. The complex information symbols are
denoted in Figure 2.2 by ๐(๐), where ๐ = 0,1, โฆ , ๐ โ 1 and ๐ is the total number of
carriers. The values of the complex symbols are derived from bit-to-symbol mapping
techniques such as M-ary Quadrature Amplitude Modulation (QAM). The IFFT block
represented in Figure 2.2 provides the capability to transform complex information
symbols, represented by ๐ฟ, into a time domain representation via a standard algorithm
[2]. Execution of the IFFT algorithm ensures carrier orthogonality during transmission,
which is a necessary requirement to successfully implement OFDM communications. In
this particular description, the IFFT length is equal to the number of carriers associated
with ๐ฟ, defined as ๐. In order to provide a mathematical representation of the IFFT, the
notation ๐ญ๐ตโ1 is introduced in Equation 2.1 to represent the Inverse Discrete Fourier
Transform (IDFT) matrix of size ๐๐ฅ๐.
๐ = ๐ญ๐ตโ1๐ฟ (2.1)
Vector ๐ is the result of performing the IDFT, which is the length ๐ time domain
representation of ๐ฟ. Proceeding through Figure 2.2, a cyclic prefix (CP) of length ๐พ is
applied to vector ๐. Inclusion of the CP results in a data packet with length ๐ + ๐พ. The
CP is a necessary component of OFDM modulation as it prevents inter-symbol
interference (ISI) that occurs as a result of multi-path. The process of applying the CP to
vector ๐ is described in Equation 2.2.
7
๐๐๐ = ๐ฅ ๐ โ ๐พ , ๐ฅ ๐ โ ๐พ โ 1 , โฆ๐ฅ ๐ โ 1 , ๐ฅ 0 , ๐ฅ 1 , โฆ , ๐ฅ ๐ โ 1 (2.2)
In order to determine the appropriate CP length ๐พ, the multi-path components of the
wireless channel must be understood. The wireless channel can be modeled
mathematically as a finite impulse response (FIR) transfer function with ๐ฟ taps or channel
coefficients. The wireless channel FIR transfer function is defined in Equation 2.3.
๐ป ๐ง = ๐0 + ๐1๐งโ1 + โฏ + ๐๐ฟโ1๐ง
โ๐ฟ+1 (2.3)
In this analysis, it is assumed that the channel can be characterized by slow fading
and thus the channel impulse response does not change within one OFDM symbol. In
order for the CP to be effective in eliminating the effects of ISI, the CP length must
exceed the duration of the channel impulse response or more specifically, the number of
multi-path channel components as defined by Equation 2.4.
๐พ โฅ ๐ฟ โ 1 (2.4)
Once the CP is incorporated into ๐, ๐๐๐ is transmitted through the wireless channel. At
the receiver, signal ๐๐๐ can be mathematically represented by the linear convolution
between transmitted signal ๐๐๐ and the channel impulse response. Channel noise ๐ is
also added to the received signal as specified in Equation 2.5 and Equation 2.6.
๐๐๐ = ๐๐๐ โ ๐๐ + ๐ (2.5)
๐ฆ๐๐ ๐ = ๐๐๐ฅ๐๐ ๐ โ ๐ + ๐ค ๐ , ๐ = 0,1, โฆ๐ + ๐พ + ๐ฟ โ 2๐ฟโ1๐=0 (2.6)
Equation 2.5 generically describes the linear convolution, whereas Equation 2.6
characterizes the linear convolution by its mathematical definition. Channel noise ๐,
referenced in both Equation 2.5 and Equation 2.6, is Additive White Gaussian Noise
(AWGN) with zero mean and variance ๐2 =๐0
2, where ๐0 is the single-sided power
8
spectral density [1]. In OFDM modulated systems, the assumption is made that ISI
occurs for the first ๐พ received symbols (received CP symbols) and thus these symbols
contained in received signal ๐๐๐ are discarded by the receiver. The process of removing
the CP to define signal ๐ is specified in Equation 2.7.
๐ = ๐๐๐ ๐พ : (๐ + ๐พ โ 1) (2.7)
The removal of the CP eliminates ISI, but there is another observation that can be made.
Removal of the CP converts the linear convolution between the transmission and the
channel impulse response as defined in Equation 2.6, into a cyclic convolution. To
further explore the observation of the cyclic convolution, consider representing the
channel in the format as indicated by Equation 2.8 and received signal ๐ with cyclic
prefix removed as indicated by Equation 2.9, where ๐ โ is the conjugate transpose
operation.
๐ = [๐0 , ๐1 , โฆ , ๐๐ฟโ1, 0, โฆ 0]๐โ (2.8)
๐ = [๐ฆ ๐ , ๐ฆ ๐ + 1 , โฆ , ๐ฆ[๐ + ๐พ โ 1]]๐โ (2.9)
In order to mathematically describe the cyclic convolution of channel ๐ and ๐, a cyclic
matrix representation of channel ๐, notated as ๐ is defined in Equation 2.10.
๐ =
0 1 2 1
1 0 1 2
1 2 1 0
0 ... 0 ...
0 ... 0 ...
...
0 ... 0 ...
L L
L
L L
h h h h
h h h h
h h h h
H
(2.10)
Therefore, the received signal ๐ can be expressed as follows.
๐ = ๐ ๐ + ๐ (2.11)
9
Through consideration of the relationship defined by Equation 2.1, signal ๐ can further
be specified as indicated in Equation 2.12.
๐ = ๐ ๐ญ๐ตโ๐๐ฟ + ๐ (2.12)
The next step in the receive chain of the SISO system is to apply the Fast Fourier
Transform (FFT) with respect to received signal ๐. Similar to the systemโs definition of
the IFFT algorithm in the transmitter, the FFT length is equal to ๐. In order to provide a
mathematical representation of the FFT, notation ๐ญ๐ต is introduced as in Equation 2.13 in
order to represent the Discrete Fourier Transform (DFT) matrix of size ๐๐ฅ๐.
๐ = ๐ญ๐ต๐ (2.13)
Furthermore, Equation 2.13 can be substituted into Equation 2.12 in order to define the
relationship presented in Equation 2.14.
๐ = ๐ญ๐ต๐ ๐ญ๐ตโ๐๐ฟ + ๐ญ๐ต๐ (2.14)
In order for the equations defined in this chapter to successfully represent a
communication system, the complex information symbols that originated as ๐ฟ must be
recovered in some manner from ๐. The process of recovering the original symbols from
the received signal is known as equalization. Equalization can be accomplished in SISO
OFDM by multiplying the inverse of relationship ๐ญ๐ต๐ ๐ญ๐ตโ๐ with ๐, as noted in Equation
2.15.
๐ฟ = ๐ญ๐ต๐ ๐ญ๐ตโ๐
โ1๐ (2.15)
Typically, matrix inversion is extremely process intensive and in many cases, can only be
approximated; however, matrix inversion of ๐ญ๐ต๐ ๐ญ๐ตโ๐ is greatly simplified numerically
due to the fact that DFT multiplication diagonalizes circular matrices.
10
๐ญ๐ต๐ ๐ญ ๐ต = ๐ป[1] 0 0
0 โฑ 00 0 ๐ป[๐ โ 1]
(2.16)
In Equation 2.16, ๐ฏ represents the ๐ - point DFT of the channel impulse response ๐. As
such, it can be observed that the diagonalization effectively decomposes the channel into
parallel, ISI-free sub-channels. In other words, the frequency-selective channel is
transformed into a channel with flat fading per carrier. From the perspective of the
physical implementation, an estimate of ๐ฟ can also be obtained with a simple zero-
forcing (ZF) detector that requires one division per carrier as defined in Equation 2.17.
๐ ๐ = ๐ ๐
๐ป ๐ ๐ค๐๐๐๐ ๐ = 0,1, โฆ , ๐ โ 1 (2.17)
Once an estimate of ๐ฟ is determined (noted as ๐ฟ ), the probability of bit error ๐๐ or BER,
is computed in order to measure the communication system performance. As an
example, for a ๐ carrier OFDM system with cyclic prefix, QAM bit-to-symbol mapping
scheme, AWGN channel and theoretical probability of QAM bit error ๐๐๐ด๐ ๐ธ๐
๐0 where
๐ธ๐ is the energy per bit, the BER is defined by Equation 2.18.
๐๐ =1
๐ ๐๐๐ด๐
๐ป[๐] 2๐๐ธ๐
๐+๐พ ๐0 ๐โ1
๐=0 (2.18)
2.3 MIMO System Description
The second system considered in this research utilizes an architecture that consists
of multiple transmitters and multiple receivers, known as MIMO. In this particular
system, OFDM modulation is supported by each transmit and receive chain in a manner
similar to the system introduced for the SISO architecture. OFDM modulation is utilized
11
in conjunction with MIMO to obtain all of the benefits OFDM provides for a
communication system. To start, consider the MIMO OFDM system shown in Figure
2.3.
IFFT P/S
Channel
X1(0)
X1(1)
X1(N-1)
.
.
.
.
.
.
S/P FFT.
.
.
V-Blast
Symbol
Detection
.
.
.
.
.
.
Y1(0)
Y1(1)
Y1(N-1)
IFFT P/S
Xm(0)
Xm(1)
Xm(N-1)
.
.
.
.
.
.
S/P FFT.
.
.
.
.
.
.
.
.
Yn(0)
Yn(1)
Yn(N-1)
Fig. 2.3. MIMO OFDM System
When assessing the description of the MIMO OFDM system contained in Figure 2.3, it
appears to be very similar in construction to the high level SISO architecture defined in
Figure 2.1. The key difference between the two systems, other than the inclusion of
multiple transmit and receive antennas, is the receiverโs method used to estimate the
transmitted signal. In the SISO architecture, an equalizer is used in accordance with the
process as described in the SISO OFDM description. In the MIMO case, an architecture
developed by Bell Laboratories known as Vertical-Bell Laboratories-Layered-Space-
Time (V-Blast) [5] is utilized to estimate the transmitted signal. Before diving into the
details of V-Blast, the reason for considering MIMO systems must be introduced. The
12
obvious advantage to MIMO systems is the fact that system throughput increases as the
total number of transmitters and receivers increases, while occupying an amount of
bandwidth consistent with SISO OFDM systems. For example, in a simple two
transmitter, two receiver system, different data is transmitted by the first antenna and the
second antenna in the same time slot and at the same frequency. Thus, in this simple
example, the data rate is doubled with respect to a traditional SISO system. In general, it
has been proven that the channel capacity of MIMO systems is greater than that of SISO
systems [6]. The capacity of the SISO system in AWGN is defined by Equation 2.19 and
the capacity of a MIMO system is defined by Equation 2.20
๐ถ๐๐ผ๐๐ = log2 1 + ๐๐๐ (2.19)
๐ถ๐๐ผ๐๐ = ๐๐๐2 ๐๐๐ก ๐ฐ +๐๐๐
๐๐๐๐โ (2.20)
Where:
๐ถ๐๐ผ๐๐ = SISO Capacity (bits/s/Hz)
๐ถ๐๐ผ๐๐ = MIMO Capacity (bits/s/Hz)
๐๐๐ = Signal-to-Noise Ratio (Linear)
๐ = Number of Receive Antennas
๐ = Number of Transmit Antennas
๐ฐ = NxM Identity Matrix
๐ = NxM MIMO Fading Channel
In order to visualize the difference in capacity expressed by Equation 2.19 and Equation
2.20, Figure 2.4 has been constructed to show both the SISO and MIMO capacities for a
random complex fading channel.
13
Fig. 2.4. SISO Capacity vs. MIMO Capacity
The V-Blast MIMO architecture offers many benefits to which advantage can be
taken. The primary benefit employed by the MIMO architecture utilized in this analysis
is spatial multiplexing gain. Spatial multiplexing gain is achieved through utilization of a
rich scattering/fading environment that allows for each transmitter to utilize the same
carrier frequency and transmission power, or in the case of OFDM, the same carrier
frequencies [5]. In this design, maximization of throughput can be achieved if the
channel environment is dynamic enough to allow the receiver to discern between signals
received from each transmitter. V-Blast is a specific approach for MIMO systems that
aims to take advantage of spatial multiplexing gain and maximize throughput [5]. This is
-5 0 5 10 15 200
1
2
3
4
5
6
SNR (dB)
Capacity (
bits/s
/Hz)
SISO Capacity
MIMO Capacity
14
achieved via an algorithm that resides in the receiver and utilizes the signals received
from both antennas in order to determine an estimate for the transmitted signal.
2.4 V-Blast MIMO OFDM Description
To begin a detailed discussion, consider the MIMO OFDM multi-carrier
modulation system with ๐ transmit and ๐ receive antennas, as illustrated in Figure 2.5.
IFFT P/S
X1(0)
X1(1)
X1(N-1)
.
.
.
.
.
.
S/P FFT.
.
.
VBLAST
Symbol
Detection
.
.
.
.
.
.
Y1(0)
Y1(1)
Y1(N-1)
IFFT
Xm(0)
Xm(1)
Xm(N-1)
.
.
.
.
.
.
Yn(0)
Yn(1)
Yn(N-1)
Add
CP
h11 +
Remove
CP
P/S.
.
.
S/P FFT.
.
.
.
.
.
Add
CP
hnm +
Remove
CP
hn1 +
h1m +
w
w
Fig. 2.5. V-Blast MIMO OFDM Receiver/Transmitter Block Diagram
It can be observed that the system represented in Figure 2.5 appears to be more
complicated than the MIMO OFDM system illustrated in Figure 2.3. The added
complication is due to the fact that the system modeled in Figure 2.5 includes a reference
to the OFDM CP (see Chapter 2.2 for more information) as well as the MIMO channel
model. The complex information symbols ๐ฟ๐ associated with the ๐๐ก๐ transmitter are
shown in Equation 2.21.
15
๐ฟ๐ =
๐๐ 0
๐๐ 1 โฎ
๐๐(๐ โ 1)
(2.21)
Each set of complex symbols ๐ฟ๐ is derived from symbol array ๐ฟ, which is defined as
follows.
๐ฟ =
๐ฟ๐
๐ฟ๐
โฎ๐ฟ๐
(2.22)
If the total number of transmitters is equal to three, then ๐ equals three and ๐ฟ has a
vector length 3๐, where ๐ is the total number of carriers associated with any transmitter.
In order to define each ๐ฟ๐, ๐ฟ is parsed into ๐ data vectors of equal length, such that
different sets of complex symbols can be transmitted in parallel. The values associated
with the complex symbols are derived from bit-to-symbol mapping techniques such as
QPSK or QAM. Similar to the SISO case, the IFFT blocks represented in Figure 2.5
provide the capability to transform the complex information symbols associated with a
specific transmitter, into a time domain representation via standard algorithm. Inclusion
of the IFFT algorithm ensures orthogonality between the carriers of a specific transmitter.
In this particular description, the IFFT length ๐ is equal to the number of carriers
associated with any ๐ฟ๐. In order to provide a mathematical representation of the IFFT,
the notation ๐ญ๐ตโ1 is introduced to represent the IDFT matrix of size ๐๐ฅ๐. As such, for ๐
transmit antennas, the following is declared where notation โจ is the Kronecker Product
and ๐ฐ๐ is an Identity matrix with a size of ๐๐ฅ๐ [7].
๐ฑ = (๐ญ๐ตโ1โจ๐ฐ๐)๐ฟ (2.23)
16
Each ๐๐ is the length ๐ time domain representation of ๐ฟ๐. Progressing through Figure
2.5, a cyclic prefix (CP) of length ๐พ is applied to each ๐๐. In this particular analysis, it is
assumed that the channel impulse response duration associated with each permutation of
transmitter and receiver are the same. Taking this into account, the inclusion of the CP
results in an ๐๐ length equal to ๐ + ๐พ. The process of applying the CP to each vector ๐๐
is described as follows where ๐ is the transpose operation.
๐๐๐๐๐ =
๐ฅ๐ ๐ โ ๐พ , ๐ฅ๐ ๐ โ ๐พ + 1 , โฆ๐ฅ๐ ๐ โ 1 , ๐ฅ๐ 0 , ๐ฅ๐ 1 , โฆ , ๐ฅ๐ ๐ โ 1 (2.24)
The CP length, previously defined as ๐พ, is determined by the channel characterization
associated with every possible spatial combination of transmit and receive antennas. In
order to determine an optimum value of ๐พ, the MIMO wireless channel must be
estimated. The channel can be modeled as a matrix of coefficients in accordance with
every possible permutation of transmit and receive antennas. Equation 2.25 depicts the
MIMO channel generically for ๐ transmitters and ๐ receivers. Each specific channel
coefficient ๐๐๐ , where ๐ identifies the receiver and ๐ identifies the transmitter, is a complex
Gaussian random variable that provides the fading gain for every spatial path of
transmission.
๐ =
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
(2.25)
Similar to the SISO analysis, it is assumed that the MIMO channel can be characterized
by slow fading and thus the channel impulse response does not change within one OFDM
symbol. This analysis also does not consider the multi-path associated with each specific
17
transmitter and receiver combination and thus, CP is not actually required. In general
though, the same principles used to define the length of the CP for the SISO case also
apply to MIMO. Therefore, the CP length must exceed the duration of the channel
impulse response, with ๐ฟ channel coefficients as seen by each combination of transmit
and receive antennas, defined in Equation 2.26.
๐พ โฅ ๐ฟ โ 1 (2.26)
Once the CP is incorporated into ๐ to define ๐๐๐, the transmission of data into the MIMO
channel occurs. At each receiver in the MIMO system, it is assumed that ISI occurs for
the first ๐พ received symbols (the CP symbols) and thus these symbols included in
received signal ๐๐๐ are discarded. The procedure for removing the CP from ๐๐๐ to
define signal ๐ is presented in Equation 2.27.
๐ =
๐๐๐๐ ๐พ โ 1 : ๐ + ๐พ โ 1
โฎ๐๐๐๐ ๐พ โ 1 : (๐ + ๐พ โ 1)
(2.27)
With the elimination of the CP by the MIMO receiver, signal ๐ can be mathematically
represented as the linear convolution between the transmitted signal ๐ and associated
MIMO channel coefficient, plus channel noise ๐, as specified in Equation 2.28.
๐๐
๐
๐๐๐
โฎ๐๐
๐
=
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
โ
๐๐
๐
๐๐๐
โฎ๐๐
๐
+
๐๐
๐
๐๐๐
โฎ๐๐
๐
(2.28)
Channel noise ๐ included in Equation 2.28 exists for each spatial path and is Additive
White Gaussian Noise (AWGN) with zero mean and variance ๐2 =๐0
2, where ๐0 is the
18
single-sided power spectral density [1]. The channel noise associated with the ๐๐ก๐ receive
antenna can be represented as follows.
๐๐ =
๐ค๐ 0
๐ค๐ 1
โฎ๐ค๐ (๐ โ 1)
(2.29)
At this point, the FFT is applied to the received signal ๐ to reverse the IFFT modulation
in the transmitter. Similar to the systemโs implementation of the IFFT in the transmitter,
the FFT is also of length ๐. In order to provide a mathematical representation of the
FFT, the notation ๐ญ๐ต is introduced in Equation 2.30 in order to represent the Discrete
Fourier Transform (DFT) matrix of size ๐๐ฅ๐.
๐ = (๐ญ๐ตโจ๐ฐ๐)๐ (2.30)
After computing the DFT to define ๐, an approach is subsequently determined to recover
an estimate of ๐ฟ from ๐. Analyzing the assumptions made with respect to the MIMO
OFDM system, it can be concluded that the linear convolution between the MIMO
channel matrix ๐ and transmitted signal ๐ can be rewritten with multiplication as follows
due to the singular duration of channel impulse response.
๐๐
๐๐
โฎ๐๐
=
๐11๐๐ + ๐12๐๐ + โฏ + ๐1๐๐๐
๐21๐๐ + ๐22๐๐ + โฏ + ๐2๐๐๐
โฎ๐๐1๐๐ + ๐๐2๐๐ + โฏ + ๐๐๐ ๐๐
+
๐๐
๐๐
โฎ๐๐
(2.31)
With this arrangement, the relationship expressed in Equation 2.30 can be substituted into
Equation 2.31 as follows.
๐๐
๐๐
โฎ๐๐
= (๐ญ๐ตโจ๐ฐ๐)
๐11๐๐ + ๐12๐๐ + โฏ + ๐1๐๐๐ + ๐๐
๐21๐๐ + ๐22๐๐ + โฏ + ๐2๐๐๐ + ๐๐
โฎ๐๐1๐๐ + ๐๐2๐๐ + โฏ + ๐๐๐ ๐๐ + ๐๐
(2.32)
19
Furthermore, each signal ๐๐ is equivalent to the IDFT of corresponding vector ๐ฟ๐ and as
such, can be utilized as indicated in Equation 2.33.
๐๐
๐๐
โฎ๐๐
=
(๐ญ๐ตโจ๐ฐ๐)
๐11(๐ญ๐ต
โ1๐ฟ๐) + ๐12(๐ญ๐ตโ1๐ฟ๐) + โฏ + ๐1๐(๐ญ๐ต
โ1๐ฟ๐) + ๐๐
๐21(๐ญ๐ตโ1๐ฟ๐) + ๐22(๐ญ๐ต
โ1๐ฟ๐) + โฏ + ๐2๐ (๐ญ๐ตโ1๐ฟ๐) + ๐๐
โฎ๐๐1(๐ญ๐ต
โ1๐ฟ๐) + ๐๐2(๐ญ๐ตโ1๐ฟ๐) + โฏ + ๐๐๐ (๐ญ๐ต
โ1๐ฟ๐) + ๐๐
(2.33)
Using the properties of the matrix formed by the Kronecker Product, the DFT matrix ๐ญ๐ต
can be transitioned into Equation 2.33 as follows.
๐๐
๐๐
โฎ๐๐
=
๐11(๐ญ๐ต๐ญ๐ต
โ1๐ฟ๐) + ๐12(๐ญ๐ต๐ญ๐ตโ1๐ฟ๐) + โฏ + ๐1๐ (๐ญ๐ต๐ญ๐ต
โ1๐ฟ๐) + (๐ญ๐ต๐๐)
๐21(๐ญ๐ต๐ญ๐ตโ1๐ฟ๐) + ๐22(๐ญ๐ต๐ญ๐ต
โ1๐ฟ๐) + โฏ + ๐2๐ (๐ญ๐ต๐ญ๐ตโ1๐ฟ๐) + (๐ญ๐ต๐๐)
โฎ๐๐1(๐ญ๐ต๐ญ๐ต
โ1๐ฟ๐) + ๐๐2(๐ญ๐ต๐ญ๐ตโ1๐ฟ๐) + โฏ + ๐๐๐ (๐ญ๐ต๐ญ๐ต
โ1๐ฟ๐) + (๐ญ๐ต๐๐)
(2.34)
Additional rearrangement can be made given the fact that a matrix multiplied by its
inverse results in an Identity matrix. Utilizing said property, Equation 2.34 can be
reduced to the following.
๐๐
๐๐
โฎ๐๐
=
๐11๐ฟ๐ + ๐12๐ฟ๐ + โฏ + ๐1๐๐ฟ๐
๐21๐ฟ๐ + ๐22๐ฟ๐ + โฏ + ๐2๐๐ฟ๐
โฎ๐๐1๐ฟ๐ + ๐๐2๐ฟ๐ + โฏ + ๐๐๐๐ฟ๐
+ (๐ญ๐ตโจ๐ฐ๐)
๐๐
๐๐
โฎ๐๐
(2.35)
At this point, symbol ๐พ is defined as follows to represent the frequency representation of
AWGN.
20
๐พ =
๐พ๐
๐พ๐
โฎ๐พ๐
= (๐ญ๐ตโจ๐ฐ๐)
๐๐
๐๐
โฎ๐๐
(2.36)
For the purposes of simplicity with the explanation going forward, vectors ๐, ๐ฟ and ๐พ
are redefined as follows.
๐ =
๐๐
๐
๐๐๐
โฎ๐๐
๐
, ๐ฟ =
๐ฟ๐
๐
๐ฟ2๐
โฎ๐ฟ๐
๐
, ๐พ =
๐พ๐
๐
๐พ๐๐
โฎ๐พ๐
๐
(2.37)
With the definition of Equation 2.37, the MIMO OFDM communication system can be
represented in a format easily extended for subsequent processing as shown in Equation
2.38.
๐ =
๐๐
๐
๐๐๐
โฎ๐๐
๐
=
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
๐ฟ๐
๐
๐ฟ๐๐
โฎ๐ฟ๐
๐
+
๐พ๐
๐
๐พ๐๐
โฎ๐พ๐
๐
= ๐๐ฟ + ๐พ (2.38)
With Equation 2.38, the communications model has be simplified to the point where an
estimate of ๐ฟ can be determined using a standard V-Blast approach. MIMO V-Blast
offers a few different alternatives that can be employed in order to determine a viable
estimate of ๐ฟ. The focus of this research has been specific to the V-Blast algorithms of
Successive Interference Cancellation (SIC) with Optimal Ordering using both Zero-
Forcing (ZF) and Minimum Mean Square Error (MMSE) equalization. The following
sections provide a technical overview of the equalization techniques of ZF and MMSE as
well as a description of optimal ordered SIC. Once an estimate of ๐ฟ is determined using
one of the aforementioned V-Blast techniques, the BER is computed in order to measure
the communication system performance.
21
2.4.1 V-Blast ZF equalization
V-Blast ZF equalization provides a simple approach to determine a realistic
estimate of transmitted signal ๐ฟ. In order to successfully solve for an estimate of ๐ฟ,
spatial filtering matrix ๐พ๐๐ญ is computed as follows through utilization of the MIMO
channel model where notation ๐ โ is the conjugate transpose [8].
๐พ๐๐ญ = (๐๐โ๐)โ1 ๐๐โ (2.39)
After matrix ๐พ๐๐ญ is determined, it is applied to ๐ to define the following.
๐ฟ = ๐พ๐๐ญ๐ = (๐๐โ๐)โ1 ๐๐โ ๐๐ฟ + ๐พ (2.40)
Looking in detail at the relationship contained in Equation 2.40, it can noted that the
estimate of ๐ฟ, defined as ๐ฟ , contains an additive ratio of noise applied to equalizing
matrix ๐พ๐๐ญ as clarified by Equation 2.41.
๐ฟ = ๐ฟ + (๐๐โ๐)โ1 ๐๐โ๐พ (2.41)
In order to reduce the impact of the additive ratio of noise to equalization and improve
the estimate of ๐ฟ, optimal ordered successive interference cancellation is employed.
2.4.2 V-Blast MMSE equalization
Similar to the algorithm based on ZF equalization as described in the previous
section, MMSE equalization is applied to the received signal ๐ via a spatial filtering
matrix defined as ๐พ๐ด๐ด๐บ๐ฌ. MMSE equalization provides an approach that is more
accurate than ZF and thus allows for computation of a more realistic estimate of
transmitted signal ๐ฟ. In order to successfully solve for an estimate of ๐ฟ, spatial filtering
22
matrix ๐พ๐ด๐ด๐บ๐ฌ is computed as follows through utilization of the MIMO channel model
where ๐0 is the single-sided noise power spectral density and ๐ฐ is the identity matrix [8].
๐พ๐ด๐ด๐บ๐ฌ = (๐๐โ๐+๐0๐ฐ)โ1 ๐๐โ (2.42)
When comparing the MMSE equalizing matrix to ZF equalization, it is important to note
that ๐พ๐ด๐ด๐บ๐ฌ contains an additive component dependent on noise. The goal of MMSE is
to develop matrix ๐พ๐ด๐ด๐บ๐ฌ to minimize the error between transmitted signal ๐ฟ and
received signal ๐ as follows.
๐ธ๐๐๐๐ = ๐พ๐ด๐ด๐บ๐ฌ๐ โ ๐ฟ ๐พ๐ด๐ด๐บ๐ฌ๐ โ ๐ฟ ๐โ (2.43)
After matrix ๐พ๐ด๐ด๐บ๐ฌ is computed, it is applied to ๐ as described in Equation 2.44.
๐ฟ = ๐พ๐ด๐ด๐บ๐ฌ๐ = (๐๐โ๐+๐0๐ฐ)โ1 ๐๐โ ๐๐ฟ + ๐พ (2.44)
The result of MMSE is declared as in Equation 2.45 where the cumulative error
associated with ๐ฟ due to the channel and AWGN is minimized.
๐ฟ = (๐๐โ๐+๐0๐ฐ)โ1 ๐๐โ๐๐ฟ + (๐๐โ๐+๐0๐ฐ)โ1 ๐๐โ๐พ (2.45)
Similar to ZF equalization the remaining error associated with ๐ฟ can be further reduced
using optimal ordered successive interference cancellation.
2.4.3 SIC with optimal ordering
The first step in the implementation of optimal ordered successive interference
cancellation is to determine the transmitted array ๐ฟ๐ that most likely was received with
the minimum collective power across all receiver antennas [5]. This is determined by
assessing the magnitude of each MIMO channel coefficient with respect to a specific
transmitted signal ๐ฟ๐ as shown in Equation 2.46.
23
๐ท๐ = ๐๐ฅ1
โฎ๐๐ฅ๐
= ๐11
2 + ๐21 2 + โฏ + ๐๐1
2
โฎ ๐1๐ 2 + ๐2๐ 2 + โฏ + ๐๐๐ 2
(2.46)
After the computation of ๐ท๐, the ๐ฟ๐ associated with the ๐๐ฅ๐ that is the minimum of vector
๐ท๐, will be estimated first using traditional SIC. The SIC algorithm with optimal
ordering ensures that the first estimate of ๐ฟ will have a lower probability of error than
any other symbol estimate. As the error probability associated with a symbol estimate
decreases, the likelihood of making incorrect decisions in the receiver decreases. For the
purposes of this description, the estimate of ๐ฟ๐ with associated minimum ๐๐ฅ๐ is declared
as ๐ฟ ๐_๐๐๐. The process for estimating ๐ฟ ๐_๐๐๐ with SIC requires the subtraction of all
other values of ๐ฟ , multiplied by the appropriate channel coefficient as indicated by
Equation 2.47.
๐น =
๐น๐
๐น๐
โฎ๐น๐
=
๐๐
๐
๐๐๐
โฎ๐๐
๐
+
โ๐11๐ฟ ๐
๐ โ โฏโ ๐1๐๐ฟ ๐๐ + ๐1๐๐ฟ ๐_๐๐๐
๐
โ๐21๐ฟ ๐๐ โ โฏโ ๐2๐๐ฟ ๐
๐ + ๐2๐๐ฟ ๐_๐๐๐๐
โฎโ๐๐1๐ฟ ๐
๐ โ โฏโ ๐๐๐๐ฟ ๐๐ + ๐๐๐๐ฟ ๐_๐๐๐
๐
(2.47)
It is important to note that Equation 2.47 includes the additive term ๐ฟ ๐_๐๐๐ multiplied by
its associated channel coefficient in order to clearly show that it is not subtracted from ๐
like all other vectors of ๐ฟ . Substituting the definition of ๐ into Equation 2.47 results in
the following.
24
๐น๐
๐น๐
โฎ๐น๐
=
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
๐ฟ๐
๐
๐ฟ๐๐
โฎ๐ฟ๐
๐
+ ๐พ +
โ๐11๐ฟ ๐
๐ โ โฏโ ๐1๐๐ฟ ๐๐ + ๐1๐๐ฟ ๐_๐๐๐
๐
โ๐21๐ฟ ๐๐ โ โฏโ ๐2๐๐ฟ ๐
๐ + ๐2๐๐ฟ ๐_๐๐๐๐
โฎโ๐๐1๐ฟ ๐
๐ โ โฏโ ๐๐๐๐ฟ ๐๐ + ๐๐๐๐ฟ ๐_๐๐๐
๐
(2.48)
It can be assumed that the cumulative error included in all values of ๐ฟ approximately
accounts for the additive AWGN present in received signal ๐ as shown in Equation 2.49
and thus, ๐ฟ ๐_๐๐๐ can be determined in an iterative fashion using the format expressed in
Equation 2.50.
๐11๐ฟ๐
๐ + ๐12๐ฟ๐๐ + โฏ + ๐1๐๐ฟ๐
๐
๐21๐ฟ๐๐ + ๐22๐ฟ๐
๐ + โฏ + ๐2๐๐ฟ๐๐
โฎ๐๐1๐ฟ๐
๐ + ๐๐2๐ฟ๐๐ + โฏ + ๐๐๐๐ฟ๐
๐
+ ๐ +
โ๐11๐ฟ ๐
๐ โ โฏโ ๐1๐๐ฟ ๐๐ + ๐1๐๐ฟ ๐_๐๐๐
๐
โ๐21๐ฟ ๐๐ โ โฏโ ๐2๐๐ฟ ๐
๐ + ๐2๐๐ฟ ๐_๐๐๐๐
โฎโ๐๐1๐ฟ ๐
๐ โ โฏโ ๐๐๐๐ฟ ๐๐ + ๐๐๐๐ฟ ๐_๐๐๐
๐
โ
๐1๐๐ฟ ๐_๐๐๐
๐
๐2๐๐ฟ ๐_๐๐๐๐
โฎ๐๐๐๐ฟ ๐_๐๐๐
๐
(2.49)
๐ 1(๐)
โฎ๐ ๐(๐)
= ๐1๐
โฎ๐๐๐
๐ ๐_๐๐๐ ๐ , ๐ค๐๐๐๐ ๐ = 0. . ๐ โ 1 (2.50)
Specifically, complex symbols ๐ฟ ๐_๐๐๐ can be estimated by Maximal Ratio Combining
(MRC) through rearrangement of Equation 2.50 as indicated by Equation 2.51.
๐ ๐_๐๐๐ (๐) =
๐1๐โฎ
๐๐๐
๐โ
๐ 1(๐)
โฎ๐ ๐ (๐)
๐1๐โฎ
๐๐๐
๐โ
๐1๐โฎ
๐๐๐
(2.51)
25
Once all values of array ๐ฟ ๐_๐๐๐ are computed, the remaining estimates of ๐ฟ can be
determined by repeating the process defined in this section for each transmitted ๐ฟ๐. The
process re-initiates after each estimate by determining the next value of ๐ฟ to be computed
based on the smallest value of ๐ท๐ for which an associated estimate of ๐ฟ has not already
been determined. For each subsequent estimate of ๐ฟ, all ๐ฟ๐ that have already been
computed via SIC are utilized in place of the original estimates determined with ZF and
MMSE equalization.
26
3. SIMPLIFIED SYSTEM
3.1 Simple System Description
With the conventional system clearly defined, this chapter introduces the
approach necessary to describe the simplified system. The use of the word โsimplifiedโ
in this context directly pertains to a reduction in computational complexity associated
with multi-carrier systems, such as OFDM. The reduction of computational complexity
is derived from the application of a new approach in performing the Fourier Transform
and Inverse Fourier Transform. As indicated in the previous chapter, the OFDM
implementation of each transmitter and receiver requires the use of the transform in order
to ensure carrier orthogonality and to simplify equalization. Extending the simplification
to MIMO systems that utilize OFDM modulation, where each receiver utilizes an FFT
and each transmitter utilizes an IFFT, the simplified approach can provide significant
savings in complexity. The approach utilized to execute both the Fourier Transform and
Inverse Fourier Transform introduces the capability to do so by performing all
multiplications with values of negative one, zero, one and powers of two. Such
multiplications are very simple to implement and are considered trivial. In doing so, the
simplification results in multiplier-less versions of Fourier Transform and Inverse Fourier
Transform. The multiplier-less transforms are derived from the process of intelligently
quantizing functions sin(๐ฅ) and cos(๐ฅ) included in the general equation necessary to
27
describe the DFT. Proper utilization of the rounded functions allow for simpler
algorithms in terms of multiplicative complexity.
3.2 Simple Discrete Fourier Transform Matrix
The general equation for determining the DFT of array ๐ is defined by Equation
3.1 [9].
๐๐ = ๐ฅ๐๐โ2๐๐๐๐
๐๐โ1๐=0 ๐ค๐๐๐๐ ๐ = 0, โฆ , ๐ โ 1 (3.1)
The process for determining the DFT can also be represented in a matrix format in
accordance with Equation 3.2 and Equation 3.3, where ๐ญ๐ต represents the conventional
DFT matrix and ๐๐ is commonly referred to as the twiddle factor.
๐ญ๐ต = (๐๐๐๐ )๐ ,๐=0,โฆ,๐โ1 (3.2)
๐๐ = ๐โ๐2๐
๐ (3.3)
With the definition of matrix ๐ญ๐ต, it can be used to determine the DFT of vector ๐ as
indicated by Equation 3.4.
๐ฟ = ๐ญ๐ต๐ (3.4)
In order to derive the simplified version of matrix ๐ญ๐ต, first recall Eulerโs Identity as
shown in Equation 3.5.
๐๐๐ฅ = cos ๐ฅ + ๐ sin(๐ฅ) (3.5)
Substituting Eulerโs Identity into Equation 3.3, the twiddle factor can be represented as
shown in Equation 3.6.
๐๐ = cos 2๐
๐ โ ๐ sin
2๐
๐ (3.6)
28
Furthermore, the relationship defined in Equation 3.6 can be substituted into Equation 3.2
as shown by Equation 3.7, to represent the DFT.
๐ญ๐ต = cos 2๐๐๐
๐ โ ๐ sin
2๐๐๐
๐
๐ ,๐=0,โฆ,๐โ1 (3.7)
With the definition of Equation 3.7, the DFT matrix ๐ญ๐ต is in the proper format to apply
the simplification.
In order to represent the rounded sin(๐ฅ) and cos(๐ฅ) functions, the following
syntax is introduced, where ๐๐๐ข๐๐() is the round-off operation and ๐ indicates the level
of quantization [10].
๐๐๐๐ ๐ ๐ฅ =๐๐๐ข๐๐ (๐ cos ๐ฅ )
๐ (3.8)
๐๐ ๐๐๐ ๐ฅ =๐๐๐ข๐๐ (๐ sin ๐ฅ )
๐ (3.9)
As the value of ๐ is increased, the response of functions ๐๐ ๐๐๐ ๐ฅ and ๐๐๐๐ ๐ ๐ฅ approach
the behavior of the conventional sin(๐ฅ) and cos(๐ฅ) functions [10]. Figure 3.1 displays
the response of ๐๐ ๐๐๐ ๐ฅ with respect to the conventional sin(๐ฅ), where k = 2.
29
Fig. 3.1. Response of Five Level Rounded Sine (k=2)
Increasing the value of ๐ to eight and comparing the response of ๐๐ ๐๐๐ ๐ฅ with sin(๐ฅ) as
shown in Figure 3.2, it is observed that the response of ๐๐ ๐๐๐ ๐ฅ more closely
approximates sin(๐ฅ).
0 1 2 3 4 5 6 7-1
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
Am
plit
ude
Phase (radians)
Conventional Sine
Quantized Sine
30
Fig. 3.2. Response of Seventeen Level Rounded Sine (k=8)
In general, the number of quantization steps represented in the result of computing
functions ๐๐ ๐๐๐ ๐ฅ and ๐๐๐๐ ๐ ๐ฅ can be described by the relationship defined in
Equation 3.10.
๐๐ ๐ก๐๐๐ = 2๐ + 1 (3.10)
With an understanding of the performance associated with the rounded functions,
it is time to apply the quantization directly to the DFT matrix. To do so, the ๐๐ ๐๐๐ ๐ฅ
and ๐๐๐๐ ๐ ๐ฅ functions are substituted into Equation 3.7 as shown by Equation 3.11
where notation ๐ญ ๐ต represents the rounded DFT [10].
0 1 2 3 4 5 6 7-1
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
Am
plit
ude
Phase (radians)
Conventional Sine
Quantized Sine
31
๐ญ ๐ต = rcosk 2๐๐๐
๐ โ ๐ rsink
2๐๐๐
๐
๐ ,๐=0,โฆ,๐โ1 (3.11)
In order to achieve the system level simplification expected of this research, the values of
๐ that can be selected must be done intelligently. In binary digital arithmetic,
multiplications with values of negative one, zero, one and powers of two are very simply
computed, implemented and considered trivial. As such, the simplification proposed by
Equation 3.11 can be optimized by selecting values of ๐ that will result in a DFT matrix
that consists entirely of values that provide trivial multiplications and thus permitting the
matrix to be considered multiplier-less. More specifically, values of ๐ are desired such
that the responses of ๐๐ ๐๐๐ ๐ฅ and ๐๐๐๐ ๐ ๐ฅ include as many powers of two as possible.
This is achieved by utilizing values of ๐ that are in fact a power of two. Equation 3.12
provides the guideline for selecting ๐.
๐ = 2๐ ๐ค๐๐๐๐ ๐ = 1,2, โฆ ,โ (3.12)
Simply utilizing values of ๐ in accordance with Equation 3.12 will not completely result
in a matrix that consists of values that support trivial multiplications. For example, when
๐ = 4, the DFT matrix will contain values of 0.75, which of course are not a power of
two. There is a method though that will permit DFT matrices of all possible values of ๐,
to be considered multiplier-less. Reconsidering the rounded DFT matrix with ๐ = 4,
even though a value of 0.75 is not a power of two, it can be obtained via the addition or
subtraction two values that are in fact powers of two such as one minus 0.25 or 0.25 plus
0.50. With this observation, it can be stated that a multiplier-less DFT matrix for all
values of ๐ can be developed by increasing the total number of additions. In the
example where ๐ = 4, every instance of 0.75 in the DFT matrix will result in two
32
additions as opposed to one non-trivial multiplication. A few subsequent additions can
be tolerated as in general, binary multiplication is more difficult to implement than binary
addition. Consider the terminology (๐) ๐ด๐2๐ผ ๐ and (๐) ๐ด๐2๐ผ ๐ด to represent the
minimum area-time digital hardware complexity for ๐-bit multiplication and addition
respectively with the following notation [11].
๐ด = ๐ถ๐๐๐ ๐ด๐๐๐
๐ = ๐๐๐๐๐๐๐๐๐๐๐ ๐๐๐๐
๐ผ โ 0, 1
When comparing (๐) ๐ด๐2๐ผ ๐ with (๐) ๐ด๐2๐ผ ๐ด, as shown in Equation 3.13, the result is
a multiplicative complexity that is on the order of ๐ greater than additive complexity.
(๐) ๐ด๐2๐ผ ๐
(๐) ๐ด๐2๐ผ ๐ด= ฮฉ ๐ ๐๐๐ ๐๐๐ ๐ผ (3.13)
3.3 Simple Inverse Discrete Fourier Transform Matrix
Because the transmitter in an OFDM system utilizes the IDFT, simplification of
the IDFT matrix is also be considered. Utilizing an approach similar to the determination
of the rounded DFT, consider the general equation for the IDFT of an array ๐ฟ as defined
by Equation 3.14.
๐ฅ๐ =1
๐ ๐๐๐
2๐๐๐๐
๐๐โ1๐=0 ๐ค๐๐๐๐ ๐ = 0, โฆ , ๐ โ 1 (3.14)
As in the case of the DFT, the process for determining the IDFT can be represented in a
matrix format in accordance with Equations 3.3.2 and 3.3.3 where ๐ญ๐ตโ1 represents the
conventional IDFT matrix and ๐๐ is the twiddle factor.
33
๐ญ๐ตโ๐ =
1
๐(๐๐
โ๐๐ )๐ ,๐=0,โฆ,๐โ1 (3.15)
๐๐ = ๐โ๐2๐
๐ (3.16)
Similar to the derivation for the rounded DFT, the rounded IDFT matrix, with notation
๐ญ ๐ตโ1, can be determined through utilization of the ๐๐ ๐๐๐ ๐ฅ and ๐๐๐๐ ๐ ๐ฅ functions to
define the matrix as in Equation 3.17.
๐ญ ๐ตโ1 =
1
๐ rcosk
2๐๐๐
๐ + ๐ rsink
2๐๐๐
๐
๐ ,๐=0,โฆ,๐โ1 (3.17)
Another method that can be utilized to determine the rounded DFT is to simply
compute the inverse of the rounded DFT matrix. Given the fact that a true inverse of the
rounded DFT cannot be easily computed for all values of ๐, another approach must be
utilized in order to determine the rounded IDFT matrix. To start, consider the fact that
any square matrix ๐จ when applied to its inverse will result in an Identity matrix as shown
in Equation 3.18.
๐จ โ ๐จโ1 = ๐ฐ (3.18)
In order to determine the rounded IDFT, a matrix can be computed such that when
applied to the rounded DFT, the result is an approximate identity function as shown in
Equation 3.19.
๐ญ ๐ต โ ๐ญ ๐ตโ1 โ ๐ฐ (3.19)
The approximate rounded IDFT matrix can be computed by taking the conjugate
transpose of the rounded DFT matrix as indicated by Equation 3.20.
๐ญ ๐ตโ1 =
1
๐๐ญ ๐ต
๐โ (3.20)
As such, an approximate, multiplier-less inverse matrix, can be determined directly from
the simplified implementation of the DFT [10]. Since the inverse is an approximation,
34
use of the IDFT computed in this manner will result in additional error in system
performance above and beyond the error injected due to quantization.
3.4 Simple Fast Fourier Transform Algorithm
Current system implementations do not simply perform matrix multiplication as
described in the previous two sections in order to utilize the Fourier Transform. As the
demand for improved performance increases, efficient algorithms have been introduced
to allow for fast and simple transformation. One such implementation of DFT is the
Radix-4 Fast Fourier Transform. The Radix-4 is derived by breaking up the original DFT
equation into four separate summations by providing ๐
4 consecutive samples of ๐ in each
sum as dictated by Decimation In Frequency (DIF) [9]. With some simplification, the
four summations can be recombined into the construct of a single summation as shown in
Equation 3.21, where ๐๐ represents the twiddle factor.
๐ ๐ =
๐ฅ ๐ + (โ๐)๐๐ฅ ๐ +๐
4 + (โ1)๐๐ฅ ๐ +
๐
2 + (๐)๐๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐๐
๐๐ (3.21)
In its current form, Equation 3.21 cannot be used to determine an FFT as the array length
is not consistently defined to be ๐/4 due to the definition of the twiddle factor that
depends on a length of ๐. In order to rearrange Equation 3.21 into an FFT of length ๐/
4, the sequence ๐ ๐ is again divided into four separate summations for the cases of
๐ = 4๐, ๐ = 4๐ + 1, ๐ = 4๐ + 2 and ๐ = 4๐ + 3. With the property ๐๐4๐๐ = ๐๐/4
๐๐ , the
four sequences that comprise the Radix-4 FFT can defined by Equation 3.22, Equation
3.23, Equation 3.24 and Equation 3.25 [9].
35
๐ 4๐ =
๐ฅ ๐ + ๐ฅ ๐ +๐
4 + ๐ฅ ๐ +
๐
2 + ๐ฅ ๐ +
3๐
4 ๐๐
0 ๐/4 โ1๐=0 ๐๐/4
๐๐ (3.22)
๐ 4๐ + 1 =
๐ฅ ๐ โ ๐๐ฅ ๐ +๐
4 โ ๐ฅ ๐ +
๐
2 + ๐๐ฅ ๐ +
3๐
4 ๐๐
๐ ๐/4 โ1๐=0 ๐๐/4
๐๐ (3.23)
๐ 4๐ + 2 =
๐ฅ ๐ โ ๐ฅ ๐ +๐
4 + ๐ฅ ๐ +
๐
2 โ ๐ฅ ๐ +
3๐
4 ๐๐
2๐ ๐/4 โ1๐=0 ๐๐/4
๐๐ (3.24)
๐ 4๐ + 3 =
๐ฅ ๐ + ๐๐ฅ ๐ +๐
4 โ ๐ฅ ๐ +
๐
2 โ ๐๐ฅ ๐ +
3๐
4 ๐๐
3๐ ๐/4 โ1๐=0 ๐๐/4
๐๐ (3.25)
Each of the four equations listed above, collectively represent the Radix-4 FFT and can
be described as four length ๐/4 FFTs. Another method commonly used to visualize the
implementation of an FFT algorithm is a butterfly chart. A butterfly chart attempts to
visualize the processing that occurs in order to compute an FFT. As an example,
consider a length four Radix-4 FFT as shown in Figure 3.3.
36
j
j
-1
-1
-1
j
-1
j
x(n)
x(n + N/4)
x(n + N/2)
x(n + 3N/4)
X(k)
X(k + N/4)
X(k + N/2)
X(k + 3N/4)
+
+
+
+
ฯ0Nฯ
nkN/4
ฯnkN/4ฯn
N
ฯnkN/4ฯ2n
N
ฯnkN/4ฯ3n
N
Fig. 3.3. Four Point Radix-4 FFT Butterfly Diagram
Figure 3.3 clearly shows the dependency between each value of ๐ฅ ๐ on one another to
determine the FFT array ๐ฟ.
Utilizing concepts developed to simplify the DFT matrix, the same approach can
be applied to the implementation of the Radix-4 algorithm. Reviewing each of the four
equations that comprise the Radix-4 FFT, it can be observed that the only multiplications
are with respect to the twiddle factors associated with each ๐/4 FFT. Focusing
specifically on Equation 3.22, the collective twiddle factor is represented as in Equation
3.26.
๐ก๐1 = ๐๐0 ๐๐/4
๐๐ (3.26)
To start, ๐ก๐1 can be further reduced as shown in Equation 3.27 as ๐๐0 = 1.
๐ก๐1 = ๐๐/4๐๐ (3.27)
37
Using the properties defined to create the rounded DFT matrix, a quantized version of ๐ก๐1
can be defined as follows.
๐ก๐ 1 = rcosk 8๐๐๐
๐ โ ๐ rsink
8๐๐๐
๐ (3.28)
With the definition of ๐ก๐ 1, it can be applied to Equation 3.22 to result in the multiplier-
less sequence presented in Equation 3.29.
๐ 4๐ = ๐ฅ ๐ + ๐ฅ ๐ +๐
4 + ๐ฅ ๐ +
๐
2 + ๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 1 (3.29)
The remaining three equations that comprise the Radix-4 are not as simply quantized.
Looking in detail with respect to Equation 3.23, the associated twiddle factor is defined
as in Equation 3.30.
๐ก๐2 = ๐๐๐๐๐/4
๐๐ (3.30)
Furthermore, twiddle factor ๐ก๐2 can be expanded as indicated in Equation 3.31.
๐ก๐2 = ๐๐๐๐๐/4
๐๐ = ๐โ๐2๐๐
๐ ๐โ๐8๐๐๐
๐ = ๐โ๐2๐๐
๐+
โ๐8๐๐๐
๐ = ๐ โ๐2๐๐
๐ 1+4๐
(3.31)
Using the properties defined to create the rounded DFT matrix, a quantized version of ๐ก๐2
can be constructed as follows.
๐ก๐ 2 = rcosk 2๐๐
๐ 1 + 4๐ โ ๐ rsink
2๐๐
๐ 1 + 4๐ (3.32)
The definition of ๐ก๐ 2 can be applied to Equation 3.23 resulting in the multiplier-less
sequence presented in Equation 3.33.
๐ 4๐ + 1 = ๐ฅ ๐ โ ๐๐ฅ ๐ +๐
4 โ ๐ฅ ๐ +
๐
2 + ๐๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 2 (3.33)
Progressing with the same procedure, twiddle factors ๐ก๐ 3 and ๐ก๐ 4 can be represented as
shown in Equation 3.34 and Equation 3.35 and are used to define Equation 3.36 and
Equation 3.37 to fully represent the rounded FFT.
38
๐ก๐ 3 = rcosk 4๐๐
๐ 1 + 2๐ โ ๐ rsink
4๐๐
๐ 1 + 2๐ (3.34)
๐ก๐ 4 = rcosk 2๐๐
๐ 3 + 4๐ โ ๐ rsink
2๐๐
๐ 3 + 4๐ (3.35)
๐ 4๐ + 2 = ๐ฅ ๐ โ ๐ฅ ๐ +๐
4 + ๐ฅ ๐ +
๐
2 โ ๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 3 (3.36)
๐ 4๐ + 3 = ๐ฅ ๐ + ๐๐ฅ ๐ +๐
4 โ ๐ฅ ๐ +
๐
2 โ ๐๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 4 (3.37)
To conclude the definition of the rounded FFT, the butterfly diagram of Figure 3.3 has
been updated as shown in Figure 3.4, to clearly show the quantized twiddle factors.
j
j
-1
-1
-1
j
-1
j
x(n)
x(n + N/4)
x(n + N/2)
x(n + 3N/4)
X(k)
X(k + N/4)
X(k + N/2)
X(k + 3N/4)
+
+
+
+
Fig. 3.4. Four Point Rounded Radix-4 FFT Butterfly Diagram
3.5 Simple Inverse Fast Fourier Transform Algorithm
Similar to DFT, a โfastโ version of the IDFT, referred to as the IFFT, can be
developed. In this description, the Radix-4 concept also provides the framework for
derivation of the rounded IFFT. The original definition of IDFT is broken into four
39
separate summations to have ๐/4 consecutive samples of ๐ฟ. With some simplification,
the four summations can be recombined into the construct of a single summation as
shown in Equation 3.38 where ๐๐ represents the twiddle factor.
๐ฅ ๐ =
๐ ๐ + (๐)๐๐ ๐ +๐
4 + (โ1)๐๐ฅ ๐ +
๐
2 + (โ๐)๐๐ฅ ๐ +
3๐
4
๐/4 โ1๐=0 ๐๐
โ๐๐ (3.38)
In its current form, Equation 3.38 cannot be used to determine an IFFT as the array length
is not consistently defined to be ๐/4 as the twiddle factor depends on a length of ๐. In
order to rearrange Equation 3.38 into an IFFT of length ๐/4, the sequence x ๐ is
divided into four separate summations for the cases of ๐ = 4๐, ๐ = 4๐ + 1, ๐ = 4๐ + 2
and ๐ = 4๐ + 3. Noting the property ๐๐โ4๐๐ = ๐๐/4
โ๐๐ , the four sequences that comprise
the Radix-4 IFFT are defined by Equation 3.39, Equation 3.40, Equation 3.41 and
Equation 3.42.
๐ฅ 4๐ =
๐ ๐ + ๐ ๐ +๐
4 + ๐ ๐ +
๐
2 + ๐ ๐ +
3๐
4 ๐๐
โ0 ๐/4 โ1๐=0 ๐๐/4
โ๐๐ (3.39)
๐ฅ 4๐ + 1 =
๐ ๐ + ๐๐ ๐ +๐
4 โ ๐ ๐ +
๐
2 โ ๐๐ ๐ +
3๐
4 ๐๐
โ๐ ๐/4 โ1๐=0 ๐๐/4
โ๐๐ (3.40)
๐ฅ 4๐ + 2 =
๐ ๐ โ ๐ ๐ +๐
4 + ๐ ๐ +
๐
2 โ ๐ ๐ +
3๐
4 ๐๐
โ2๐ ๐/4 โ1๐=0 ๐๐/4
โ๐๐ (3.41)
๐ฅ 4๐ + 3 =
๐ ๐ โ ๐๐ ๐ +๐
4 โ ๐ ๐ +
๐
2 + ๐๐ ๐ +
3๐
4 ๐๐
โ3๐ ๐/4 โ1๐=0 ๐๐/4
โ๐๐ (3.42)
40
Each of the four equations listed above represent the Radix-4 IFFT and can be described
as four length ๐/4 IFFTs. As with any FFT, a butterfly chart can be used to visualize the
processing that occurs in order to compute an IFFT. As an example, consider a length
four Radix-4 IFFT as shown in Figure 3.5.
j
-j
-1
-1
-1
-j
-1
j
x(n)
x(n + N/4)
x(n + N/2)
x(n + 3N/4)
X(k)
X(k + N/4)
X(k + N/2)
X(k + 3N/4)
+
+
+
+
ฯ0Nฯ
nkN/4
ฯnkN/4ฯn
N
ฯnkN/4ฯ2n
N
ฯnkN/4ฯ3n
N
Fig. 3.5. Four Point Radix-4 IFFT Butterfly Diagram
Utilizing the concepts developed to simplify the FFT, the same approach can be
applied to the implementation of the Radix-4 IFFT algorithm. Reviewing each of the
four equations that comprise the Radix-4 IFFT, it can be observed that the only
multiplications are with respect to the twiddle factors associated with each ๐/4 IFFT.
Focusing specifically on Equation 3.39, the collective twiddle factor is represented as in
Equation 3.43.
๐ก๐1 = ๐๐โ0๐๐/4
โ๐๐ (3.43)
41
To start, ๐ก๐1 can be further reduced as shown in Equation 3.44.
๐ก๐1 = ๐๐/4โ๐๐ (3.44)
Using the properties defined to create the rounded IDFT matrix, a quantized version of
๐ก๐1 can be defined as follows.
๐ก๐ 1 = ๐๐๐๐ ๐ 8๐๐๐
๐ + ๐ ๐๐ ๐๐๐
8๐๐๐
๐ (3.45)
Twiddle factor ๐ก๐ 1 can then be applied to Equation 3.39, resulting in a multiplier-less
IFFT as shown in Equation 3.46.
๐ฅ 4๐ = ๐ ๐ + ๐ ๐ +๐
4 + ๐ ๐ +
๐
2 + ๐ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 1 (3.46)
The remaining three equations that comprise the Radix-4 are not as simply quantized.
Looking in detail with respect to Equation 3.40, the associated twiddle factor is defined
as in Equation 3.47.
๐ก๐2 = ๐๐โ๐๐๐/4
โ๐๐ (3.47)
Furthermore, twiddle factor ๐ก๐2 can be expanded as shown in Equation 3.48.
๐ก๐2 = ๐๐โ๐๐๐/4
โ๐๐ = ๐๐2๐๐
๐ ๐๐8๐๐๐
๐ = ๐๐2๐๐
๐+
๐8๐๐๐
๐ = ๐ ๐2๐๐
๐ 1+4๐
(3.48)
Using the properties defined to create the rounded IDFT matrix, a quantized version of
๐ก๐2 can be defined as follows.
๐ก๐ 2 = ๐๐๐๐ ๐ 2๐๐
๐ 1 + 4๐ + ๐ ๐๐ ๐๐๐
2๐๐
๐ 1 + 4๐ (3.49)
The definition of ๐ก๐ 2 can be applied to Equation 3.40, resulting in the multiplier-less
sequence presented in Equation 3.50.
๐ฅ 4๐ + 1 = ๐ ๐ + ๐๐ ๐ +๐
4 โ ๐ ๐ +
๐
2 โ ๐๐ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 2 (3.50)
42
Furthermore, twiddle factors ๐ก๐ 3 and ๐ก๐ 4 can be represented as shown in Equation 3.51
and Equation 3.52 and applied to define Equation 3.53 and Equation 3.54 in order to fully
represent the rounded IFFT.
๐ก๐ 3 = ๐๐๐๐ ๐ 4๐๐
๐ 1 + 2๐ + ๐ ๐๐ ๐๐๐
4๐๐
๐ 1 + 2๐ (3.51)
๐ก๐ 4 = ๐๐๐๐ ๐ 2๐๐
๐ 3 + 4๐ โ ๐ ๐๐ ๐๐๐
2๐๐
๐ 3 + 4๐ (3.52)
๐ฅ 4๐ + 2 = ๐ ๐ โ ๐ ๐ +๐
4 + ๐ ๐ +
๐
2 โ ๐ ๐ +
3๐
4
๐/4 โ1๐=0 ๐ก๐ 3 (3.53)
๐ฅ 4๐ + 3 = ๐ ๐ โ ๐๐ ๐ +๐
4 โ ๐ ๐ +
๐
2 + ๐๐ ๐ +
3๐
4 ๐ก๐ 4
๐/4 โ1๐=0 (3.54)
To conclude the definition of the rounded IFFT, the butterfly diagram of Figure 3.5 has
been updated as shown in Figure 3.6, to clearly display the quantized twiddle factors.
j
-j
-1
-1
-1
-j
-1
j
x(n)
x(n + N/4)
x(n + N/2)
x(n + 3N/4)
X(k)
X(k + N/4)
X(k + N/2)
X(k + 3N/4)
+
+
+
+
Fig. 3.6. Four Point Rounded Radix-4 IFFT Butterfly Diagram
43
3.6 Simple SISO OFDM
Now that simplified versions of both the FFT and IFFT algorithms have been
defined, it is time to apply both to the SISO OFDM system. To start, consider the
simplified OFDM multi-carrier modulation system with single transmit and receive
antennas as illustrated in Figure 3.7.
Simp
IFFTP/S
X(0)
X(1)
X(N-1)
.
.
.
S/PSimp
FFT
Channel
Estimation
.
.
.
.
.
.
Y(0)
Y(1)
Y(N-1)
Add
CP
.
.
.
h +
w
Remove
CP
.
.
.
Simp
FFT
\
.
.
.
\
\
.
.
.
Fig. 3.7. Simplified SISO OFDM Transceiver Block Diagram
It is important to the note that the difference between Figure 2.2 and Figure 3.7 is specific
to the simplification of the FFT and IFFT blocks. The complex information symbols are
denoted in Figure 3.7 by ๐(๐), where ๐ = 0,1, โฆ , ๐ โ 1, where ๐ is the total number of
carriers. The values associated with the complex symbols are derived from bit-to-symbol
mapping techniques such as QPSK and QAM. The simplified IFFT block provides the
capability to transform complex information symbols, represented by ๐ฟ into an
approximate time domain representation via the rounded IFFT algorithm. Depending on
the level of quantization, execution of the rounded IFFT algorithm provides near carrier
44
orthogonality, still allowing for successful OFDM communications. Error associated
with carrier orthogonality, is injected into the system due to the characteristics of the
simplified IFFT algorithm. As the level of quantization specified for the rounded IFFT
increases, errors due to approximate orthogonality reduce. In this particular description,
the rounded IFFT length is equal to the number of carriers associated with ๐ฟ, defined as
๐. In order to provide a mathematical representation of the rounded IFFT, the notation
๐ญ ๐ตโ1 is introduced in Equation 3.55 in order to represent the rounded IDFT matrix of size
๐๐ฅ๐.
๐ = ๐ญ ๐ตโ1๐ฟ (3.55)
Vector ๐ is the result of evaluating Equation 3.55, which is the length ๐ approximate
time domain representation of ๐ฟ. Stepping through Figure 3.7 from left to right, a cyclic
prefix (CP) of length ๐พ must be applied to vector ๐. In this description, the CP is
included in signal ๐ via the identical procedure defined for the conventional SISO OFDM
system, resulting in Equation 3.56.
๐ ๐๐ =
๐ฅ ๐ โ ๐พ , ๐ฅ ๐ โ ๐พ โ 2 , โฆ๐ฅ ๐ โ 1 , ๐ฅ 0 , ๐ฅ 1 , โฆ , ๐ฅ ๐ โ 1 (3.56)
As in the standard system, it is assumed that the channel can be characterized by slow
fading and thus the channel impulse response does not change within one OFDM symbol.
Once the CP is incorporated into ๐ , ๐ ๐๐ is transmitted through the wireless channel. At
the receiver, signal ๐ is mathematically represented by the linear convolution between
transmitted signal ๐ ๐๐ and the channel impulse response, plus channel noise ๐, as
specified in Equation 3.57 and Equation 3.58.
45
๐ ๐๐ = ๐ ๐๐ โ ๐๐ + ๐ (3.57)
๐ฆ ๐๐ ๐ = ๐๐๐ฅ ๐๐ ๐ โ ๐ + ๐ค ๐ , ๐ = 0,1, โฆ๐ + ๐พ + ๐ฟ โ 2๐ฟโ1๐=0 (3.58)
Equation 3.57 generically describes the convolution, where as Equation 3.58 represents
the convolution by its mathematical definition. Channel noise ๐ is defined as Additive
White Gaussian Noise (AWGN) with zero mean, variance ๐2 =๐0
2 and ๐0 is the single-
sided power spectral density. In simple OFDM, the CP is removed from received signal
๐ ๐๐ in the same manner as in the conventional system per Equation 3.59.
๐ = ๐ ๐๐ ๐พ : (๐ + ๐พ โ 1) (3.59)
The removal of the CP converts the linear convolution between the transmitted symbols
and the channel impulse response into a cyclic convolution. The result of the cyclic
convolution is a wireless system defined in accordance with Equation 3.60.
๐ = ๐ ๐ + ๐ (3.60)
Incorporating the relationship specified in Equation 3.55, signal ๐ can further be
expressed as defined in Equation 3.61.
๐ = ๐ ๐ญ ๐ตโ๐๐ฟ + ๐ (3.61)
The next step in the receive chain is to apply the rounded FFT to signal ๐ . Similar to the
systemโs use of the rounded IFFT contained in the transmitter, the simple FFT length is
equal to ๐. In order to provide a mathematical representation of the rounded FFT, the
notation ๐ญ ๐ต is introduced in Equation 3.62 to represent the rounded Discrete Fourier
Transform (DFT) matrix of size ๐๐ฅ๐.
๐ = ๐ญ ๐ต๐ (3.62)
46
The relationship defined in Equation 3.62 can be substituted into Equation 3.61 to define
Equation 3.63.
๐ = ๐ญ ๐ต๐ ๐ญ ๐ตโ1๐ฟ + ๐ญ ๐ต๐ (3.63)
In order for the relationships defined in this section to successfully represent a
communication system, the information symbols that originated as ๐ฟ must be recovered
from ๐ through equalization. Equalization can be accomplished by multiplying the
inverse of the relationship ๐ญ ๐ต๐ ๐ญ ๐ตโ1 with ๐ . Due to the quantization utilized in the
development of matrix ๐ญ ๐ต and ๐ญ ๐ตโ1, the resultant matrix formed by the relationship
๐ญ ๐ต๐ ๐ญ ๐ตโ1 will not result in a completely diagonal matrix as in the conventional system.
As such, the processing necessary to compute ๐ญ ๐ต๐ ๐ญ ๐ตโ1 is more complicated than in
standard OFDM. As a result, an estimate of the transmitted information ๐ฟ can still be
obtained via multiplication of the inverse matrix of ๐ญ ๐ต๐ ๐ญ ๐ตโ1 with ๐, however a simple
zero-forcing equalizer is preferred as it provides identical complexity with respect to the
standard system.
๐ฟ = ๐ญ ๐ต๐ ๐ญ ๐ตโ1
โ1๐ (3.64)
A simple zero-forcing (ZF) detector, that requires one division per carrier as defined in
Equation 3.65 can be implemented to determine an estimate of ๐ฟ.
๐ ๐ = ๐ ๐
๐ป ๐ ๐ค๐๐๐๐ ๐ = 0,1, โฆ , ๐ โ 1 (3.65)
Figures 4.1 through 4.27, contained in Chapter 4, provide the BER curves
necessary to evaluate the performance of the conventional SISO OFDM system described
in Chapter 2.2 with respect to the simplified SISO OFDM system.
47
3.7 Simple MIMO OFDM
In order to study the application of the rounded FFT and IFFT algorithms to
MIMO OFDM multi-carrier modulation, consider a system with ๐ transmit and ๐ receive
antennas as illustrated in Figure 3.8.
Simp
IFFTP/S
X1(0)
X1(1)
X1(N-1)
.
.
.
.
.
.
S/P Simp FFT.
.
.
VBLAST
Symbol
Detection
.
.
.
.
.
.
Y1(0)
Y1(1)
Y1(N-1)
Simp
IFFT
Xm(0)
Xm(1)
Xm(N-1)
.
.
.
.
.
.
Yn(0)
Yn(1)
Yn(N-1)
Add
CP
h11 +
Remove
CP
P/S.
.
.
S/P Simp FFT.
.
.
.
.
.
Add
CP
hnm +
Remove
CP
hn1 +
h1m +
w
w
Fig. 3.8. Simplified V-Blast MIMO OFDM Receiver/Transmitter Block Diagram
In this system, the complex information symbols associated with the ๐๐ก๐ transmitter are
identified as ๐ฟ๐. Each set of complex symbols ๐ฟ๐ is derived from a data set defined as ๐ฟ.
For example, if the number of transmitters is equal to three, then ๐ equals three and ๐ฟ
would have a vector length 3๐ where ๐ is the total number of carriers associated with a
single transmitter. In order to define each ๐ฟ๐, ๐ฟ is parsed into ๐ data vectors of equal
length such that different sets of complex symbols can be transmitted in parallel. The
values associated with the complex symbols of ๐ฟ are derived from bit-to-symbol
mapping techniques such as QPSK and QAM. Similar to the SISO case, the rounded
48
IFFT blocks represented in Figure 3.8 provide the capability to transform the complex
information symbols associated with a specific transmitter, into a near time domain
representation. The evaluation of the rounded IFFT algorithm provides near
orthogonality between carriers specific to a transmitter such that successful OFDM
communications can still be obtained. It is important to note that as the quantization level
associated with the rounded IFFT increases, errors due to approximate orthogonality are
reduced. Each rounded IFFT is of length ๐, which is equal to the number of carriers
associated with each ๐ฟ๐. In order to provide a mathematical representation of the
rounded IFFT with respect to ๐ number of transmitters, the notation ๐ญ ๐ตโ1 is introduced in
Equation 3.66 to represent the rounded Inverse Discrete Fourier Transform (IDFT) matrix
of size ๐๐ฅ๐.
๐ = (๐ญ ๐ตโ1โจ๐ฐ๐)๐ฟ (3.66)
Progressing through Figure 3.8 from left to right, a cyclic prefix (CP) of length ๐พ must be
applied to each vector ๐ ๐. In this description of the simplified MIMO OFDM system, the
CP is applied to each ๐ ๐ in the same manner as in the conventional system with the result
indicated in Equation 3.67.
๐ ๐๐๐๐ =
๐ฅ ๐ ๐ โ ๐พ , ๐ฅ ๐ ๐ โ ๐พ + 1 , โฆ๐ฅ ๐ ๐ โ 1 , ๐ฅ ๐ 0 , ๐ฅ ๐ 1 , โฆ , ๐ฅ ๐ ๐ โ 1 (3.67)
The MIMO wireless channel can be modeled as a matrix of channel coefficients in
accordance with every possible combination of transmit and receive antennas. Equation
3.68 provides a generic representation of the MIMO channel for ๐ transmitters and ๐
receivers. Each specific channel coefficient ๐๐๐ , where ๐ identifies the receiver and ๐
49
identifies the transmitter, is a complex Gaussian random variable that provides the fading
gain between each variation of transmitter and receiver data path.
๐ =
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
(3.68)
As in previous descriptions, it is assumed that the MIMO channel can be characterized by
slow fading and thus the channel impulse response does not change within one OFDM
symbol. This analysis also does not account for the multi-path associated with each
specific combination of transmit and receive antenna and thus no CP is actually required.
In general though, the same principles used to define the length of the CP for the
previously discussed systems also apply to simple MIMO. Once ๐ ๐๐ is defined as
indicated in Equation 3.67, the corresponding data is transmitted through the MIMO
wireless channel. At each receiver in the simple MIMO system, the CP associated with
received signal ๐ ๐๐ is discarded as indicated by Equation 3.69.
๐ =
๐ ๐๐๐ ๐พ โ 1 : (๐ + ๐พ โ 1)
โฎ๐ ๐๐๐ ๐พ โ 1 : (๐ + ๐พ โ 1)
(3.69)
Once the CP is removed, signal ๐ can be mathematically represented as the linear
convolution between the transmitted signal ๐ and associated MIMO channel coefficient,
plus channel noise ๐, as specified in Equation 3.70.
๐ ๐
๐
๐ ๐๐
โฎ๐ ๐
๐
=
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
โ
๐ ๐
๐
๐ ๐๐
โฎ๐ ๐
๐
+
๐๐
๐
๐๐๐
โฎ๐๐
๐
(3.70)
50
Channel noise ๐ค included in Equation 3.70, exists for each spatial path and is Additive
White Gaussian Noise (AWGN) with zero mean and variance ๐2 =๐0
2, where ๐0 is the
single-sided power spectral density. The channel noise associated with the ๐๐ก๐ receive
antenna can be represented as follows.
๐๐ =
๐ค๐ 0
๐ค๐ 1
โฎ๐ค๐ (๐ โ 1)
(3.71)
At this point, the rounded DFT of length ๐ is applied to received signal ๐ in order to
reverse the impact of the rounded IDFT modulation in the transmitter. In order to
provide a mathematical representation of the rounded DFT, the notation ๐ญ ๐ต is introduced
in Equation 3.72.
๐ = (๐ญ ๐ตโจ๐ฐ๐)๐ (3.72)
After computing ๐ , the next step in the receiver is to determine an approach for
recovering an estimate of ๐ฟ from ๐ . Analyzing the assumptions made with respect to the
simplified MIMO OFDM system, it can be concluded that the linear convolution between
the MIMO channel matrix ๐ and transmitted signal ๐, as indicated in Equation 3.70, can
be rewritten as follows with multiplication due to the singular duration of channel
impulse response.
๐ ๐
๐ ๐
โฎ๐ ๐
=
๐11๐ ๐ + ๐12๐ ๐ + โฏ + ๐1๐๐ ๐
๐21๐ ๐ + ๐22๐ ๐ + โฏ + ๐2๐๐ ๐
โฎ๐๐1๐ ๐ + ๐๐2๐ ๐ + โฏ + ๐๐๐ ๐ ๐
+
๐๐
๐๐
โฎ๐๐
(3.73)
With this arrangement, the relationship expressed in Equation 3.72 can be substituted into
Equation 3.73 as follows.
51
๐ 1๐ 2
โฎ๐ ๐
= (๐ญ ๐ตโจ๐ฐ๐)
๐11๐ ๐ + ๐12๐ ๐ + โฏ + ๐1๐๐ ๐ + ๐๐
๐21๐ ๐ + ๐22๐ ๐ + โฏ + ๐2๐๐ ๐ + ๐๐
โฎ๐๐1๐ ๐ + ๐๐2๐ ๐ + โฏ + ๐๐๐ ๐ ๐ + ๐๐
(3.74)
Furthermore, each signal ๐ ๐ is approximately equal to the rounded IDFT of
corresponding vector ๐ฟ๐ and as such, can be utilized as specified in Equation 3.75.
๐ ๐
๐ ๐
โฎ๐ ๐
=
(๐ญ ๐ตโจ๐ฐ๐)
๐11(๐ญ ๐ต
โ๐๐ฟ๐) + ๐12(๐ญ ๐ตโ๐๐ฟ๐) + โฏ + ๐1๐ (๐ญ ๐ต
โ๐๐ฟ๐) + ๐๐
๐21(๐ญ ๐ตโ๐๐ฟ๐) + ๐22(๐ญ ๐ต
โ๐๐ฟ๐) + โฏ + ๐2๐ (๐ญ ๐ตโ๐๐ฟ๐) + ๐๐
โฎ๐๐1(๐ญ ๐ต
โ๐๐ฟ๐) + ๐๐2(๐ญ ๐ตโ๐๐ฟ๐) + โฏ + ๐๐๐ (๐ญ ๐ต
โ๐๐ฟ๐) + ๐๐
(3.75)
Using the properties of the matrix formed by the Kronecker Product, the rounded DFT
matrix ๐น ๐ can be transitioned into Equation 3.75 as follows.
๐ ๐
๐ ๐
โฎ๐ ๐
=
๐11(๐ญ ๐ต๐ญ ๐ต
โ๐๐ฟ๐) + ๐12(๐ญ ๐ต๐ญ ๐ตโ๐๐ฟ๐) + โฏ + ๐1๐ (๐ญ ๐ต๐ญ ๐ต
โ๐๐ฟ๐) + (๐ญ ๐ต๐๐)
๐21(๐ญ ๐ต๐ญ ๐ตโ๐๐ฟ๐) + ๐22(๐ญ ๐ต๐ญ ๐ต
โ๐๐ฟ๐) + โฏ + ๐2๐ (๐ญ ๐ต๐ญ ๐ตโ๐๐ฟ๐) + (๐ญ ๐ต๐๐)
โฎ๐๐1(๐ญ ๐ต๐ญ ๐ต
โ๐๐ฟ๐) + ๐๐2(๐ญ ๐ต๐ญ ๐ตโ๐๐ฟ๐) + โฏ + ๐๐๐ (๐ญ ๐ต๐ญ ๐ต
โ๐๐ฟ๐) + (๐ญ ๐ต๐๐)
(3.76)
Additional reduction can be performed given the fact that a matrix multiplied by its
inverse results in an Identity matrix. Utilizing said property, Equation 3.76 can be
simplified to the following.
๐ ๐
๐ ๐
โฎ๐ ๐
=
๐11๐ฟ๐ + ๐12๐ฟ๐ + โฏ + ๐1๐๐ฟ๐
๐21๐ฟ๐ + ๐22๐ฟ๐ + โฏ + ๐2๐๐ฟ๐
โฎ๐๐1๐ฟ๐ + ๐๐2๐ฟ๐ + โฏ + ๐๐๐๐ฟ๐
+ (๐ญ ๐ตโจ๐ฐ๐)
๐๐
๐๐
โฎ๐๐
(3.77)
52
At this point, symbol ๐พ is defined as follows to represent the frequency representation of
AWGN.
๐พ =
๐พ ๐
๐พ ๐
โฎ๐พ ๐
= (๐ญ ๐ตโจ๐ฐ๐)
๐๐
๐๐
โฎ๐๐
(3.78)
For the purposes of simplicity going forward with the explanation, vectors ๐ , ๐ฟ and ๐พ
are redefined as follows.
๐ =
๐ ๐
๐
๐ ๐๐
โฎ๐ ๐
๐
, ๐ฟ =
๐ฟ๐
๐
๐ฟ๐๐
โฎ๐ฟ๐
๐
, ๐พ =
๐พ ๐
๐
๐พ ๐๐
โฎ๐พ ๐
๐
(3.79)
With the definition of Equation 3.79, the simple MIMO OFDM communication system
can be represented in a format easily extended for additional processing as shown in
Equation 3.80.
๐ =
๐ ๐
๐
๐ ๐๐
โฎ๐ ๐
๐
=
๐11 ๐12
๐21 ๐22
โฆ ๐1๐
โฆ ๐2๐
โฎ โฎ๐๐1 ๐๐2
โฆ โฎโฆ ๐๐๐
๐ฟ๐
๐
๐ฟ๐๐
โฎ๐ฟ๐
๐
+
๐พ ๐
๐
๐พ ๐๐
โฎ๐พ ๐
๐
= ๐๐ฟ + ๐พ (3.80)
With the specification of Equation 3.80, the communications model has been defined to
the point where an estimate of ๐ฟ can be determined using a standard V-Blast approach.
All V-Blast approaches introduced for the conventional MIMO OFDM system also apply
to the simplified system. Once an estimate of ๐ฟ is determined using a standard V-Blast
technique, the BER is computed in order to evaluate the communication system
performance.
53
Figure 4.28 through Figure 4.43, contained in Chapter 4, provide the BER curves
necessary to compare the performance of the conventional MIMO OFDM system
described in Chapter 2.4 with the simplified MIMO OFDM system.
54
4. SIMULATION RESULTS
4.1 SISO OFDM Architecture
Using the relationships defined in Chapters 2 and 3, a MATLAB model has been
developed in order to represent the performance of the conventional and simplified SISO
OFDM modulated systems [1]. Simulations of the model have been executed to generate
the results presented in this section. The key parameter used to analyze the performance
of the system is bit error rate (BER). As such, BER curves with respect to the ratio of bit
energy to single-sided noise power spectral density are computed in order to evaluate the
performance of the conventional system with respect to the simplified system. With
regards to the model developed to represent the simplified system, the levels of
quantization used to implement the rounded FFT and rounded IFFT are as follows.
Table 4.1
Rounded FFT/IFFT Twiddle Factor Quantization
๐ Quantization
Steps
2 5
4 9
8 17
16 33
55
Other degrees of freedom considered in the simulations are bit-to-symbol mappings of
QPSK and 16QAM as well as three different wireless channel models. The first channel
model represents a flat fading channel with the following frequency response.
Fig. 4.1. Flat Fading Channel Frequency Response (Channel 1)
The flat fading channel will attenuate the magnitude of the transmission by slightly less
than -3 dB; however, there is no effect on the transmission phase. The second channel
model included in this analysis represents a typical office environment with 50 nano
second root mean square (RMS) delay spread and Rayleigh fading. The frequency
response of the second channel is as follows.
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-1
-0.5
0
0.5
1
Normalized Frequency ( rad/sample)
Phase (
degre
es)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-5
-4
-3
-2
Normalized Frequency ( rad/sample)
Magnitude (
dB
)
56
Fig. 4.2. Typical Office Channel Frequency Response (Channel 2)
The last channel included in this research characterizes a large open area with 100 nano
second RMS delay spread and Rayleigh fading. The frequency response of channel
number three is as follows.
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10
50
100
150
200
Normalized Frequency ( rad/sample)
Phase (
degre
es)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-20
-15
-10
-5
0
Normalized Frequency ( rad/sample)
Magnitude (
dB
)
57
Fig. 4.3. Large Open Area Channel Frequency Response (Channel 3)
In conjunction with the degrees of freedom included in the SISO OFDM model, static
parameters such as a symbol rate equal to 250 KHz, presence of AWGN and FFT/IFFT
and rounded FFT/IFFT Length of 64 are employed. The following figures are plots
generated to describe the performance of the conventional SISO OFDM system versus
the rounded SISO OFDM system.
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200
-100
0
100
200
Normalized Frequency ( rad/sample)
Phase (
degre
es)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-30
-20
-10
0
10
Normalized Frequency ( rad/sample)
Magnitude (
dB
)
58
Fig. 4.4. SISO OFDM with QPSK BER, k=2, Channel 1
Fig. 4.5. SISO OFDM with QPSK BER, k=4, Channel 1
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
59
Fig. 4.6. SISO OFDM with QPSK BER, k=8, Channel 1
Fig. 4.7. SISO OFDM with QPSK BER, k=16, Channel 1
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
60
Fig. 4.8. SISO OFDM with 16QAM BER, k=2, Channel 1
Fig. 4.9. SISO OFDM with 16QAM BER, k=4, Channel 1
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
61
Fig. 4.10. SISO OFDM with 16QAM BER, k=8, Channel 1
Fig. 4.11. SISO OFDM with 16QAM BER, k=16, Channel 1
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
62
Fig. 4.12. SISO OFDM with QPSK BER, k=2, Channel 2
Fig. 4.13. SISO OFDM with QPSK BER, k=4, Channel 2
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
63
Fig. 4.14. SISO OFDM with QPSK BER, k=8, Channel 2
Fig. 4.15. SISO OFDM with QPSK BER, k=16, Channel 2
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
64
Fig. 4.16. SISO OFDM with 16QAM BER, k=2, Channel 2
Fig. 4.17. SISO OFDM with 16QAM BER, k=4, Channel 2
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
65
Fig. 4.18. SISO OFDM with 16QAM BER, k=8, Channel 2
Fig. 4.19. SISO OFDM with 16QAM BER, k=16, Channel 2
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
66
Fig. 4.20. SISO OFDM with QPSK BER, k=2, Channel 3
Fig. 4.21. SISO OFDM with QPSK BER, k=4, Channel 3
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
67
Fig. 4.22. SISO OFDM with QPSK BER, k=8, Channel 3
Fig. 4.23. SISO OFDM with QPSK BER, k=16, Channel 3
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM QPSK
Rounded OFDM QPSK
68
Fig. 4.24. SISO OFDM with 16QAM BER, k=2, Channel 3
Fig. 4.25. SISO OFDM with 16QAM BER, k=4, Channel 3
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
69
Fig. 4.26. SISO OFDM with 16QAM BER, k=8, Channel 3
Fig. 4.27. SISO OFDM with 16QAM BER, k=16, Channel 3
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
5 10 15 20 25 30 35 40
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
OFDM 16QAM
Rounded OFDM 16QAM
70
4.2 MIMO OFDM Architecture
Similar to the SISO system, a computer simulation has been developed in
MATLAB to provide the performance of both the conventional and simplified MIMO
OFDM modulated systems [1]. Simulations of the MIMO model have been performed in
order to generate the results presented in this section. As in the SISO system, the key
parameter used to analyze the performance of the MIMO architecture is BER. As such,
several BER curves are computed in order to evaluate the performance of the
conventional system with respect to the simplified system. The levels of quantization
used to implement the rounded FFT and rounded IFFT for the simplified MIMO
architecture are the same as in the simulations for the simplified SISO model. The
quantization levels used to execute the rounded FFT and rounded IFFT are indicated in
Table 4.1. Other parameters included in this analysis are bit-to-symbol mappings of
QPSK and 16QAM as well as the symbol detection technique of Optimal Ordered SIC
coupled with ZF and MMSE equalization. Additionally, results are generated for
randomly generated flat fading complex channel. Static parameters used in the
simulation are a symbol rate equal to 250 KHz, presence of AWGN, FFT/IFFT and
rounded FFT/IFFT Length of 64, two transmit antennas and two receive antennas. The
following is a series of plots to describe the performance of the conventional SISO
OFDM system versus the rounded SISO OFDM system.
71
Fig. 4.28. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=2
Fig. 4.29. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=4
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
72
Fig. 4.30. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=8
Fig. 4.31. MIMO OFDM with Optimal Ordered ZF-SIC and QPSK BER, k=16
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
73
Fig. 4.32. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=2
Fig. 4.33. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=4
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
74
Fig. 4.34. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=8
Fig. 4.35. MIMO OFDM with Optimal Ordered MMSE-SIC and QPSK BER, k=16
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
75
Fig. 4.36. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=2
Fig. 4.37. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=4
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
76
Fig. 4.38. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=8
Fig. 4.39. MIMO OFDM with Optimal Ordered ZF-SIC and 16QAM BER, k=16
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
77
Fig. 4.40. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=2
Fig. 4.41. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=4
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
78
Fig. 4.42. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=8
Fig. 4.43. MIMO OFDM with Optimal Ordered MMSE-SIC and 16QAM BER, k=16
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
0 5 10 15 20 25 30
10-4
10-3
10-2
10-1
Bit E
rror
Rate
Eb/N0
Rounded MIMO-OFDM
MIMO-OFDM
79
5. CONCLUSIONS
A new concept has been developed that can be utilized in conjunction with
existing systems such that they can be considered cutting-edge and innovative. The new
concept introduced in this research is the application of the rounded Fast Fourier
Transform (FFT) and rounded Inverse Fast Fourier Transform (IFFT) to both SISO and
MIMO OFDM modulated systems. As the descriptions and associated results presented
in this research confirm, inclusion of the rounded FFT and rounded IFFT into both SISO
and MIMO OFDM systems provide performance that approaches the conventional
system, while eliminating all non-trivial multiplications. Furthermore, the results prove
that the approach introduced for simple OFDM leads to viable communication systems.
Considering the results associated with the simple SISO architecture presented in
Chapter 4.1, it is clear that the performance of systems that include QPSK bit-to-symbol
mappings as opposed to QAM, more closely resemble the performance of the
conventional. It is also evident that the low end of โtwiddle factorโ quantization (i.e. k =
2, k = 4) performs poorly in the SISO system as error is introduced both in the multi-
carrier modulation process as well as in the equalizer due to the use of the rounded FFT
to generate a frequency domain representation of the channel impulse response. With
regards to the MIMO system, the simulation results presented in Chapter 4.2 clearly
indicate that the performance degradation is smaller than what has been concluded for the
80
SISO system. This observation is significant for systems that implement MIMO as the
IFFT and FFT algorithms are required for every spatial stream (i.e. antenna) and thus the
system-wide computational complexity is further reduced. Therefore, the proposed
approach is particularly suitable for modern high-data rate MIMO systems.
The elimination of non-trivial multiplications provided by the rounded FFT and
IFFT will allow for simpler hardware implementation due to the reduction in
computational complexity. The reduction in computational complexity is quantified by
comparing the total number of actual multiplications and additions necessary to
implement the conventional Radix-2 and Radix-4 FFT versus the rounded algorithm as
described by Table 5.1.
Table 5.1
FFT and Rounded FFT Complexity
Transform
Size
Multiplications Additions
Radix-2
FFT
Radix-4
FFT
Rounded
FFT
Radix-2
FFT
Radix-4
FFT
Rounded
FFT
64 264 208 k = 2: 0 1032 976 k = 2: 976
k = 4: 0 k = 4: 1008
k = 8: 0 k = 8: 1032
k = 16: 0 k = 16: 1072
256 1800 1392 k = 2: 0 5896 5488 k = 2: 5488
k = 4: 0 k = 4: 5616
k = 8: 0 k = 8: 5736
k = 16: 0 k = 16: 5844
The values provided for additions and multiplications as represented in Table 5.1 are the
actual number of non-trivial real multiplications and real additions. As previously
described, different levels of quantization can be utilized to develop the rounded FFT and
IFFT. As the level of quantization increases, the overall system performance approaches
81
that of the conventional system. Depending on the performance required by a specific
application, different variations of the rounded FFT and IFFT can be utilized in order to
obtain minimum complexity.
Table 5.1 clearly describes that the implementation rounded FFT requires a slight
increase of additions along with zero non-trivial multiplications when compared to the
Radix-2 and Radix-4 FFT; however, this result must be further quantified. Using the
comparison of additive complexity versus multiplicative complexity provided in
Equation 3.13, a ratio can be developed in order to scale the complexity associated with
the implementation of a multiplication to be consistent with an addition [11].
Furthermore, an estimate of overall complexity can be computed and compared for the
Radix-2 FFT, Radix-4 FFT and rounded FFT with the results contained in Table 5.2.
Table 5.2
Complexity Reduction Provided by Rounded FFT
Number
of Bits
Order of
Multiplicative
Complexity
vs. Additive
Complexity
Estimated Complexity Reduction in
Complexity
256
Length
Radix-2
FFT
256
Length
Radix-4
FFT
256 Length
Rounded
FFT (k=16)
Rounded
FFT vs.
Radix-2
Rounded
FFT vs.
Radix-4
7 2.65 10666 9177 5844 45% 36%
8 2.83 10990 9427 5844 47% 38%
9 3.00 11296 9664 5844 48% 40%
10 3.16 11584 9887 5844 50% 41%
11 3.32 11872 10109 5844 51% 42%
12 3.46 12124 10304 5844 52% 43%
13 3.61 12394 10513 5844 53% 44%
14 3.74 12628 10694 5844 54% 45%
15 3.87 12862 10875 5844 55% 46%
16 4.00 13096 11056 5844 55% 47%
82
To summarize the results included in Table 5.2, the rounded FFT provides an estimated
minimum reduction in complexity of 45% with respect to the Radix-2 FFT and a
minimum reduction in complexity of 36% with respect to the Radix-4 FFT for systems
that represent numerical values with a number of bits greater than six. Table 5.2 provides
the critical result necessary to quantify the reduction in complexity for the new rounded
system.
BIBLIOGRAPHY
83
BIBLIOGRAPHY
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