analysis and practical considerations in implementing multiple transmitters for wireless power...

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Analysis and Practical Considerations inImplementing Multiple Transmitters for WirelessPower Transfer via Coupled Magnetic Resonance

Rizal Johari, Student Member, IEEE, James Krogmeier, Member, IEEE, and David Love, Senior Member, IEEE

AbstractMultiple transmitters can be used to simultaneouslytransmit power wirelessly to a single receiver via stronglycoupled magnetic resonance. A simple circuit model is used tohelp explain the multiple transmitter wireless power transfersystem. Through this particular scheme, there is an increasein gain and diversity of the transmitted power according tothe number of transmit coils. The effect of transmitter resonantcoil coupling is also shown. Resonant frequency detuning dueto nearby metallic objects is observed and the extent of howmuch tuning can be done is demonstrated. A practical power linesynchronization technique is proposed to synchronize all transmitcoils which reduces additional dedicated synchronization wiringor the addition of an RF front end module to send the referencedriving signal.

Index TermsCircuit model, multiple transmitters, resonantcoupling, wireless power transfer.

I. INTRODUCTION

RECENTLY, there has been significant interest in effi-cient medium-range wireless power transfer for poweringand/or charging future personal electronic devices. Systemsthat allow short-range powering and charging are alreadycommercially available and research challenges remain inextending the range and improving the power transfer effi-ciency. Researchers have demonstrated that inductive couplingbetween low-loss resonant coils allows significant power to betransmitted with high efficiency over distances on the orderof a few times the radius of the transmit coil [2], [3]. Thesingle transmitter (TX) and single receiver (RX) demonstrationsystem consists of four coils, two at the TX and two at theRX. The two coils at the power transmitter consist of a sourcecoil and a resonant coil. Similarly, the two coils at the powerreceiver consist of a resonant coil and a load coil.

Applications for medium range wireless power transfercould include a wide range of areas, among them are wirelesscontrolled robots, RFID based systems [7], electric vehiclecharging [9], charging mobile devices and biomedical implants[7],[14]. Different system configurations such as a multiplereceiver setup where a single transmit coil powers severalloads have been discussed in [5],[8]. The coupling coefficients

Manuscript received November 30, 2012; revised February 13, 2013;accepted April 22, 2013.

Copyright (c) 2013 IEEE. Personal use of this material is permitted.However, permission to use this material for any other purposes must beobtained from the IEEE by sending a request to [email protected].

R. Johari, J. V. Krogmeier and D. J. Love are with the School of Electricaland Computer Engineering, Purdue University, West Lafayette, IN 47907USA. Emails: {rjohari, jvk, djlove}@purdue.edu.

linking each resonant coil, which corresponds to the geometry,angle and distance between coils play an important role intransmission efficiency. Reference [15] introduces an adaptivefrequency technique to ensure maximum power transfer effi-ciency within an overcoupled region where frequency splittingoccurs. In [16], the distance between TX/RX coil pairs areadjusted to keep an effective matching condition. Powertransfer efficiency for multiple transmitters in a fixed positionsurrounding the load is investigated by [13]. Their test caseshows a theoretical bound on power transfer efficiency for the2 TX and 1 RX case. Effect of coupling between multipletransmitters or receivers for the single resonant source/loadcoil configuration is discussed in [10]. Multiple transmittersusing an optimized structure is used by [24] to maximizecoupling between TX and RX for a free-positioning pla-nar system. Coupled magnetic resonance can further be ex-tended to increase the operating distance by introducing relays[4],[6],[11]. An increased gain is seen at further distancesdue to efficient wireless energy transfer between relay coils.References [2]-[6] use coupled mode theory (CMT) as ananalytical framework to model resonant energy exchange. Thispaper relies on basic circuit theory to model the resonantenergy transfer, as also done in [8]-[18]. As proved in [20],both frameworks result in the same set of equations in steadystate and are applicable for both short and midrange couplingconditions.

Different transmitter and receiver settings can be developedfor various purposes and benefits. The multiple transmittercase is considered for added gain and diversity effects. Asthe number of transmitters increase, the gain of the systemis enhanced while also regulating the magnetic field strength.Diversity benefits occur when foreign metal objects come inclose proximity to any of the resonant coils. This interferencecauses a detuning effect on the resonant coil and sufficientlydegrades power transfer efficiency. Retuning the resonant coilsis necessary although some power loss due to excess heatgeneration is unavoidable. The use of multiple transmittersrequires proper synchronization. A unique way of doing so isby using the ubiquitous power line network.

This paper is organized as follows. Section II describesthe basic principles and framework for a two transmitterand one receiver case. In Section III, The transfer functionfor the multiple transmitter case is simulated and comparedwith actual measurements. Effects of transmitter resonantto resonant coupling are analyzed in Section IV. SectionV explains the diversity effect and is shown by incidental



Fig. 1. Two transmitter and one receiver experimental setup.

resonant frequency shifts due to nearby metallic objects.Lastly, a practical synchronization technique via power linecommunications including its benefits are presented in SectionVI.

II. MULTIPLE TRANSMITTER SYSTEM OVERVIEW

Different scenarios constitute different setups in a wirelesspower transfer scheme. The focus in this paper is to utilizemore than one TX coil pair for added gain and diversity ben-efits. There are certain challenges when using multiple coilswhich include signal synchronization and coupling betweenmultiple transmitters. By increasing the number of transmit-ters, power transfer reliability and gain can be improved whilealso regulating the amount of power being sent through freespace. Metal object interference or the ability to uniformlysend power over a wide area can be supported by havingsynchronized transmitters.

Fig. 1 illustrates the experimental setup for 2 TX pairsand one RX pair. This setup assumes loose coupling betweenboth TX coils and also the RX pair coils. If the TX and RXcoils are tightly coupled, frequency splitting occurs and woulddegrade the efficiency of the system. Frequency adaptation[15], variable coupling between coil pairs [16], matchingnetworks [25] or antiparallel resonant loops [26] could beused to improve efficiency at these distances. These techniquescan be used in the multiple transmitter case considering loosecoupling between transmitter resonant coils and equal distancebetween both TX and RX coils.

Source voltages VS1 and VS2 in Fig. 1 are sinusoidal signalswith equal magnitude and phase. A total of 6 inductor coilslabelled L1 through L6 with radii of 0.057 m were constructedusing AWG14 copper magnet wire. The resonator coils inthis setup have 5 turns each. For simplicity, the source andload coils consist of only 1 turn. The distances, d12, d34, andd56 between coil pairs were set at 0.04 m. The transmitterswere placed at a distance, d25 = d45 = 0.35 m away fromthe receiver with angular separation of 45. Resistances R1through R6 are the coils ohmic resistance at resonance. RS1and RS2 represent the source resistances and RL is the loadresistance. The resonant coils L2, L4, and L5 were terminatedwith lumped capacitors C2, C4 and C5, whose values were

Fig. 2. A picture of the 2-TX and 1-RX wireless power transfer systemexperimental setup.

chosen such that the resonant frequencies

f0 =1

2piLC

(1)

were all equal. It is important that the driving signals at eachsource be synchronized with the same frequency and phasein order to avoid severe power fluctuations at the receiver.Capacitances C2, C4 and C5 are the sum of parasitic capaci-tance (CP ) between the turns of the multi-turn resonant coilsand lumped capacitance (CL) included for tuning the resonantfrequency (Ci = Ci,P + Ci,L, for i = 2, 4, and 5). Since theresonant coils were designed to have high Q, it is importantto account for parasitic capacitance. Small offsets of 1 2 pF(parasitics) can cause a resonant shift of approximately 70 150 kHz and with a very narrow bandwidth even the slightestshift can degrade performance. The parasitic capacitance forcoils L1, L3 and L6 are neglected since they have but a singleturn.The resonator coils L2, L4, and L5 have the same nominalinductances since they are identically fabricated. The coilswere tuned to the selected driving frequency of 8.4 MHz byadjusting the lumped capacitors Ci,L.

Assuming sinusoidal steady state excitation the phasor volt-age across the k-th coil can be written as Vcoil,k = jkwhere

k = LkIk l 6=k

Mk,lIl (2)

is the total flux linking the turns of the k-th coil, Mk,lrepresents the mutual inductance between coils k and l, andIl is the phasor current in the l-th coil. Mk,l can be expressedin terms of the coupling coefficient kk,l and self inductanceof the corresponding loops Lk and Ll as such

Mk,l = kk,lLkLl (3)

With the above constitutive equations and Kirchoffs voltagelaw taken at each loop as depicted in Fig.1, one can solve forphasor currents as the product of the inverse of the impedancematrix times a source voltage column vector:

I1I2I3I4I5I6

= [{Zk,l}1k,l6]1VS10VS2000

(4)



7.4 7.6 7.8 8 8.2 8.4 8.6 8.8x 106

70

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40

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20

10

Frequency, MHz

Mag

nitu

de, d

B

Simulation2TXSimulation1TX (coil 1&2)Simulation1TX (coil 3&4)Exp2TXExp1TX (coil 1&2)Exp1TX (coil 3&4)Simplified Sim 2TXSimplified Sim 1TX

Fig. 3. Transfer Function |VL/Vs| (dB) for simulated (complete andsimplified) and experimental measurements for two and one transmitter setups.

The individual impedances (Zk,l) are given by

Z11 = jL1 +RS1 +R1

Z22 = jL2 +R2 +1

jC2Z33 = jL3 +RS2 +R3

Z44 = jL4 +R4 +1

jC4

Z55 = jL5 +R5 +1

jC5Z66 = jL6 +R6 +RL (5)

with Zk,l = Zl,k = jMk,l for k 6= l with the exception ofZx,y = Zy,x = jMx,y for {x, y} = {1, 3}, {1, 4}, {2, 3}and {2, 4}.

The magnitude of the load voltage is |VL| = |RLI6| and wewish to compute and compare transfer functions from inputsto the load for both the SISO and MISO case. For simplicity,we assume VS1 = VS2 = VS for the MISO case. For the SISOcase we can assume without loss of generality that VS1 = VSand VS2 = 0. In either case we plot transfer functions |VL/VS |(dB) vs. frequency.

Solving for I6 explicitly from (4) results in a very com-plicated expression. To simplify things, with respect to theexperimental setup conditions, we can set Z11 = Z33 due toboth source coils equal properties. The resonator coils alsoshare the same characteristics resulting in Z22 = Z44 = Z55.Cross coupling coefficients are neglected and are set to zero.This results in a simplified impedance matrix as stated in (7),

Z11 jwM12 0 0 0 0jwM12 Z22 0 0 jwM25 0

0 0 Z11 jwM12 0 00 0 jwM12 Z22 jwM25 00 jwM25 0 jwM25 Z22 jwM120 0 0 0 jwM12 Z66

(7)

The above simplification gives us a reasonable model ofthe experimental setup in Fig. 1 where (6) is the simplified

0 20 40 60 80 100 120 140 160 18080

70

60

50

40

30

20

10

0

(Degrees) Phase difference between TX1 and TX2

Mag

nitu

de, d

B

Simulation ResultsSimplified Simulation (Eq.6)Experimental Results

Fig. 4. Transfer Function |VL/Vs| (dB) with phase differences at TX (1,1 = 0, f = 8.4MHz).

equation for I6. The transfer function for the MISO casecan be obtained through superposition of the two powersources. At resonance, simulation and theoretical results showno substantial difference in value for I6 between the simplifiedequation, I6(a) in (6) and the exact one, I6, in (4) when crosscouplings are set to zero.

Further analysis of (6) shows a voltage gain of 2 and apower gain of 4 when the phasor voltage VS1 = AS1ejS1 andVS2 = AS2e

jS2 are equal in both magnitude (AS1 = AS2)and phase (S1 = S2). This is a direct result of |VL| = |I6RL|and |PL| = |I26RL| when the current of I6 is doubled. The gainshown above is only true if the transmitters are synchronizedin frequency, phase and amplitude.

In a practical system, input sources could be out of phase.The difference in phase results in a lower transfer functionvalue with the worst case at 180. Fig. 4 shows the results fordifferent phases at the transmitter and is discussed further inSection III. If the input signals are completely out of phase,theoretically no power should be transferred due to completedestructive combining at the receiver. Experiments conductedby [21] investigates different receiver angles for two casesinvolving in-phase and out-of-phase input signals also showthe destructive case at a 0 receiver angle.

III. EXPERIMENTAL AND SIMULATION RESULTS ANDANALYSIS

Three different measurements were taken with both transmitcoils turned on and two measurements taken with the transmitcoils turned on individually. This was done to experimentallydetermine the gain available from using additional transmitcoils. The experimental setup assumes static transmitter coilpositions with distances far enough to avoid coupling betweentransmit resonant coils. Fig. 2 is an image of how the experi-ment was set up.

The experimental results agree well with the theoreticalresults as shown in Fig. 3. As explained in the previous section,there is a 6 dB gain in theory provided that both signals ideally

I6(a) =jw3M212M25 (AS1e

jS1 +AS2ejS2)

w4M412 + Z11Z66Z222 + w

2Z11M212Z22 + w2Z66M212Z22 + 2w

2Z11Z66M225(6)



TABLE IPARAMETER VALUES

Par. Value Par. Value Par. Value Par. ValueRS1 50 L1 .30 H k12 .2 k34 .2RS2 50 L2 6.10 H k13 .0001 k35 .0006R1 .053 L3 .30 H k14 .0001 k36 .0005R2 .265 L4 6.11 H k15 .0006 k45 .00064R3 .053 L5 6.12 H k16 .0005 k46 .0006R4 .265 L6 .31 H k23 .0001 k56 .2R5 .265 C2 58.9 pF k24 .0001R6 .053 C4 58.8 pF k25 .00064RL 100 C5 58.7 pF k26 .0006

combine coherently at the receiver. Actual transfer functionmeasurements showed a gain of approximately 5.3dB. This0.7 dB difference could be due to small matching errors inthe coil resonant frequencies, minor phase delay differencesbetween reference signals, and imperfect geometric alignmentbetween transmitter and receiver.

The benefits of having multiple transmitters consideringhardware limitations (limited power or size of transmitter) isthat one could increase gain or power transfer area by simplyadding extra transmitters. If a larger power transfer area ispreferred rather than gain, extra transmitters could be addedsuch that the power transfer is combined at the edges to ensureuniform gain distribution. A number of difficulties arise withadded transmitters including reference signal synchronizationand phase delays due to differing distances between thetransmit power coils and the receiver coil. It is important tomake sure that the shared frequency is locked.

In this paper, a master reference signal is shared with the TXcoil (slave). This will ensure a locked frequency with slightphase delays depending on the signal wavelength, distance andchannel characteristics between TX coils. As seen in Fig. 4,at 8.4MHz, a phase difference of 10 between transmittersdegrades the voltage transfer function by only 0.38% or0.03dB while a 90 phase difference results in a 30% or 2.4dBdegradation experimentally. In theory, if the phase differenceis 120, the results would be the same as if only one TX wastransmitting. For the 180 out-of-phase case, the experimentalresults show a 99.9% reduction in received voltage which intheory is 100% reduction at zero volts. In practice, completedestructive combining is hard to occur due to minor differencesof the coils Q-factor or resonance, angle and distance betweenboth TX to the RX. The simulation results for the simplifiedcase (6) and the complete case (I6 in (4)) are the same andagree well with the experimental results.

The parameter values used in the simulations consist ofmeasured and theoretically calculated values. The self induc-tances of each coil (L1 to L6) were measured using a SencoreLC102 Capacitor-Inductor Analyzer. The AC resistances ofthe coils were based on the standard formula (which accountsfor skin effect),

R =rN

a

02

(8)

where r is the coil radius, N is the number of turns, ais the cross sectional radius of the wire, = 2pif is theangular frequency (here corresponding to the nominal resonantfrequency of 8.4 MHz), 0 is the magnetic permeability offree space (4pi 107 H/m) and is the conductivity of

Fig. 5. Experimental setup for TX resonant to resonant coupling. Anglebetween TX and RX is reduced from 45 to 10 without changing its distance.

copper (5.813107 S/m). Total capacitance values C2, C4 andC5 for the corresponding resonant loop were computed usingmeasured inductance values and the formula for resonance in(1). The neglected cross coupling coefficients in the previoussection were included in the simulation results for a moreaccurate analysis within a wider frequency range. Couplingcoefficient values were chosen to fit the experimental resultsas was done in [8] and were based on the characteristics ofthe coils distance between each other and also their angularorientation. Table I shows the parameter values used in theexperiment and simulations. Simulated SISO conditions foreach transmitter were shown in Fig. 3 to corroborate thecoupling coefficient values. Another method by [12] measuredvarious S-Parameter configuration values and were used inAdvanced Design System (ADS) to extract the coupling coef-ficient parameters.

The parameter values in Table I can then be used to cal-culate the efficiency of the experimental setup. At resonance,the power transfer efficiency for the two transmitter case is10% while the single transmitter case had a 5% efficiency.These low power transfer efficiencies are primarily due to thedistance between TX and RX operating in the undercoupled re-gion. If operating in the critically coupled mode (less distance),the experimental setup can achieve up to 70% efficiency.

IV. EFFECTS OF TRANSMITTER RESONANT COILCOUPLING

The experimental setup explained previously assumes in-significant coupling between transmitter resonant coils. Thisassumption is acceptable since having tightly spaced trans-mitters reduces diversity benefits where a single interferingmetal object could cause simultaneous resonant shifts at bothtransmitter resonant coils. Besides diversity degradation, it canbe shown that there is a decreasing effect on the gain as thetransmit resonant coils move closer to each other (increasingk24).

Simplified theoretical calculations of I6 shown in (6) ne-glects the k24 coupling coefficient which signifies the amountof coupling between transmit resonant coils. To gauge theeffects of having transmitter coupling, the cross couplingcoefficient, k24, of the resonant transmit coils is added to



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x 106

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60

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45

40

35

30

25

Frequency, MHz

Magnitude, dB

Simulation

Experimental

Simplified Sim

Fig. 6. (A) Transfer function |VL/VS | (dB) as a function of frequency andthe resonant to resonant transmit coupling coefficient k24 with regards to thesimplified I6(B) equation in (12). (B) Simulated (complete and simplified)and measured values for the transfer function |VL/VS | (dB) with significantresonator to resonator transmitter coupling (k24 = 0.061)

the impedance matrix (7) corresponding to Z24 and Z42. Forsimplicity, similarly to (6), cross coupling terms are neglectedwith the exception of k24. Fig. 6(A) shows the transferfunction as a function of frequency and coupling coefficientk24. Simulations indicate a splitting effect as the transmittercoils are brought closer to each other. The behavior of thesplitting occurs differently when compared to the multiplereceiver case [8] where one of the two peaks remains atthe resonant frequency with the other diverging at a higherfrequency. Simulation parameters were based on the valuesused in the previous section with a varying k24 coupling term.An increase in the k24 coefficient represents a closer distancebetween TX resonant coils. Assuming the magnitude of (6) as

|I6(a)| = |A||B| (9)

where A is the numerator and B the denominator of (6) .With the addition of the k24 term, the magnitude of I6 can beexpressed as

|I6(b)| = |A||B + C| (10)

where C = k24L2L4j(M212Z11w3 Z11Z22Z66w) is an

added imaginary term at the denominator. (12) represents thesimplified equation with the added term of k24. Since all theterms are non-negative, in order to maximize |I6(b)|, the C termshould approach zero (k24 0). A closer look at Fig. 6(A)shows that as C becomes much smaller than B (C B), thesplitting peaks of the transfer function converge towards theresonant frequency and approaches the maximum value withrespect to (6). An experiment was conducted to see if thetheoretical and simulation results would match experimentaldata. The setup in Fig. 2 was slightly modified by movingthe transmitter resonant coils closer while maintaining thesame distance between TX and RX as shown in Fig. 5. Onlythe angle between both TX and RX coils were minimizedfrom 45 to approximately 10 without changing any distance

TABLE IICOUPLING COEFFICIENTS FOR THE TX-TX COUPLING EXPERIMENT

Par. Value Par. Value Par. Valuek12 .2 k23 .055 k35 .00084k13 .001 k24 .061 k36 .0007k14 .055 k25 .00089 k45 .00089k15 .00084 k26 .00084 k46 .00084k16 .0007 k34 .2 k56 .2

between coils. Coupling coefficient values between TX andRX differ by multiplying an = cos()/cos(45) factor giventhe known k(tx,rx,45) values used in Table I. The followingequation is used for the TX and RX coupling coefficients:

k(tx,rx,) = k(tx,rx,45) =

(cos()

cos(45)

)k(tx,rx,45) (11)

Coupling coefficients k12, k34, and k56 remain the same whiletransmitter to transmitter coupling coefficients k13, k24, k23and k14 has values of .001, .061, .055 and .055 respectively.Table II shows the coupling coefficient values for the TXresonant to resonant coupling experiment.

Fig. 6(B) indicates two peaks at resonance, 8.4 MHz,and 8.67 MHz which coincides with experimental results. Intheory, any increase in the k24 coefficient will reduce powertransfer. If the A and B coefficient of the setup is operatingat maximum efficiency, there is no obvious way of improvingpower transfer when there is significant coupling between TXresonant coils. Time multiplexing between transmitters couldbe implemented where one of the coil is detuned or turnedoff while the other is transmitting [22]. This provides similarpower transfer levels as the SISO case. Transmitting with onlyone coil would appear to be more efficient. A 12dB differenceis seen when comparing the results in Fig. 3 and Fig. 6(B).The placement of the TX coils should be at a distance suchthat there is insignificant interaction between TX coils but at aminimum angle orientation to increase TX and RX coupling.The optimum position for such a case for the two TX and oneRX scenario is by placing all three coils in a single axis. TheTX coils are placed at two opposite ends with the RX coilpositioned in the middle [23].

V. DIVERSITY EFFECT OF THE MULTIPLE TRANSMITTERWIRELESS POWER TRANSFER SETUP

Besides gain advantage, power transfer reliability is en-hanced through multiple transmitters. In certain practical sce-narios, foreign metal can interact with the TX resonant coilsand cause the coils self inductance to change. A shift inresonance can cause huge power losses due to the resonantcoils high Q factor. Therefore, readjusting the resonance tothe correct value is crucial. There are limitations to how muchtuning can be done due to reduced magnetic flux coupling andpower losses from eddy current formation.

I6(b) =jw3M212M25 (AS1e

jS1 +AS2ejS2)

w4M412 + Z11Z66Z222 + w

2Z11M212Z22 + w2Z66M212Z22 + 2w

2Z11Z66M225 +M24j(M212Z11w3 Z11Z22Z66w)(12)



A B

Fig. 7. (A) Experimental setup for the metal sheet covering half the resonantcoils area. (B) Metal sheet covering a significant portion of the resonant coilsarea

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Frequency, MHz

Mag

nitu

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B

FreespaceMetalHalfMetalHalf AdjustedMetalFullMetalFull Adjusted

Fig. 8. Resonant frequency shift due to metal objects present. Two differentscenarios are present where an aluminum plate is covering half the coils areaand one that fully covers the area

A good conductor will allow circulating eddy currentswhen exposed to a changing magnetic field. This phenomenonproduces an opposing magnetic field that reduces the coilsmagnetic field and thus reduces its inductance. From (1),a decrease in inductance will result in a higher resonantfrequency. The Q-factor of the resonant coil can be calculatedby

Q =L

R=

1

R

L

C(13)

at its resonant angular frequency = 1/LC. A decrease in

the coils inductance will also result in a lower Q. To achieveacceptable power transfer efficiencies, the lumped capacitoris readjusted to a higher value to help realign the resonantfrequency. Equation (1) is used to estimate the retuned lumpedcapacitor value.

An experiment (see Fig. 7) consisting of a TX pair coil anda single pick up coil receiver was conducted to understand theeffects of having a metal interferer within close proximity ofthe resonant coil. An aluminum metal plate with dimension17.3 12.2 0.1 cm was used. The separation between eachcoil was 0.04 m with axes aligned. The resonant frequencyand corresponding peak output voltage were observed with anoscilloscope. Fig. 8 shows results for three cases: 1) no plate,2) aluminum plate covering half the coil area, and 3) coveringthe entire coil area.

Using the Sencore LC meter, the inductance of the coilwhen the metal sheet covered half and the entire area wasmeasured at 5.35H and 4.27H respectively. Adjusted ca-pacitance values of 67.1pF and 83.9pF were needed to reshiftthe resonance. The freespace condition had a theoretical totalcapacitance of 58.85pF. It is important to remember that duringpractical tuning of the lumped capacitor, parasitic capacitance

Coupling Circ.

Coupling Circ.Phase

& Gain Adj.

Reference

Signal

Wall Outlet

Wall Outlet

Amp.

Amp.

Master TX Coil

Slave TX Coil

Coupling Circuit

Gain & Phase Adj.

Power Line Connection

Fig. 9. (A) Block Diagram, (B) Picture of the power line communicationsynchronization technique for multiple transmitter wireless power transfer viacoupled magnetic resonance.

should be taken into account.As seen in Fig. 8, for the full metal covering case, it is

impractical to retune the resonance as there is still a significantamount of power degradation (18 dB) after resonant tuning.This is primarily due to significant reduction in the couplingcoefficient between the TX and RX. One can think of themetal as a magnetic shield where the majority of the magneticfield lines flow within the metal and back to the TX coil. Forthe half metal covered case, an initial 26dB power loss isreduced to an acceptable 2.5dB loss after retuning. In theory,if only resonance detuning occurred without loss of coupling,adjusting the capacitance value would regain original results.Multiple transmitters introduce a diversity effect that reducesthe probability of outage and also maintains a certain qualityof standard with regards to power transfer when comparedwith the SISO case.

The method of measuring and retuning the resonance issuitable for permanent internal interference such as placingcoils within an electronic device containing metal content. Forexternal type interference, detaching the system and retuningthe system using an LCR analyzer is not feasible. Therefore,an automatic system to automatically detect if the resonance isbelow or above the optimum frequency is needed. Monitoringthe resonant frequency of the coil in real time is necessary andsuch a system for automatic capacitance tuning is subject ofour ongoing research. Also, there needs to be a set thresholdvalue for which the system should regulate or stop transmittingpower due to power losses incurred at the metal object, as thiswill also save power consumption and reduce potential safetyhazards due to unintentional heating.

VI. PRACTICAL IMPLEMENTATION FOR THE MULTIPLETRANSMITTER SETUP VIA POWER LINE

The multiple transmitter experiment conducted in Section IIused a dedicated wire to establish synchronization. One uniqueway of synchronizing the driving signals is to utilize the powerline infrastructure. This reduces the need of additional wiringor an RF front end module. Another useful reason for utilizingthe power line network is the flexibility of sending differentdriving frequencies if automatic frequency tuning is needed incase of frequency splitting in the overcoupled region [15].

Power lines were built primarily for power transmissionand were not optimized for efficient wireline communications.



R2

-+R1 R3

C3

R5

-+

Input

Output

-+ R4

Amp. 1Amp. 2

Amp. 3

Fig. 10. Gain and phase adjustment circuit diagram.

Vs(8.4MHz)

50 Ohms 100pF

1:140uH : 40uH

100pF

1:140uH : 40uH

TransmitCoilPhase

& Gain Adj.

Wall Outlet

Wall Outlet

Fig. 11. Coupling circuit diagram connecting the reference signal (8.4MHz)to the slave transmit coil via power line communications.

Some of the issues associated with PLC communicationsinclude impedance mismatch, absence of EMI shielding andthe existence of detrimental noise sources varying with bothfrequency and time. The main objective of this experiment isto use the already available infrastructure to send a referencesignal in order to synchronize the magnetic field transmittedfrom both transmit coils. The overall setup for the PLCsynchronization technique is shown in Fig. 9.

A. Coupling Circuit & Gain and Phase Adjustment

A coupling circuit is needed to block the power lines120 Vrms 60 Hz sinusoidal signal and to inject the intendeddriving signal of 8.4 MHz. A high pass filter with a reasonablecutoff frequency was used as a medium to channel signalsthrough the power line network. A high voltage capacitorvalued at 100 pF, 1000 V rating together with a 1:1 highfrequency transformer (CoilCraft, PWB-1-AL, 0.08450 MHzBandwidth) with an inductance of 40 H was utilized to passsignals above 2.5 MHz. The high pass filter was configured asan LC filter. See Fig. 11. This circuit configuration is knownas a transformer-capacitor coupler design and is used in manyPLC products. The advantage of using a transformer is due toits ability to provide galvanic isolation, impedance matching,and also to act as a limiter in case of high voltage transients[27]. Due to the frequency response of the coupling circuitand the power line itself, the signal transmitted will havedistortion in terms of both magnitude and phase. Therefore,a gain and phase adjustment circuit was needed (Fig. 10).An LM6171BIN National Semiconductor amplifier with unitygain bandwidth of 100 MHz was used to drive the powertransmitter coils with an external 50 resistor in series withthe output. This particular amplifier was used for theoreticalpurposes due to the flexibility of modifying or modulatingspecific signals for future research and is not capable ofhandling high output current. If higher power is needed, aclass-E amplifier setup can be used [19].

(A)

(B)

Original Signal

Attenuated Signal -

PLC output

Original Signal

Adjusted Signal

Fig. 12. (A) 8.4MHz reference signal before phase and gain adjustment. (B)after gain and phase adjustment.

The reference signal after traversing the coupling circuitand power line has a peak to peak voltage of 674 mV, areduction of 17.4 dB from its original 5 V peak-to-peak value.In addition, there is a phase difference of 49 correspondingto 16.3 nsec of delay. The design of the gain and phaseadjustment circuit is given in the equations below. For phaseadjustment, ideally, the values for components R3 and C3 asin Fig. 10 can be chosen using the transfer function of thebasic RC setup

HRC =VoutVin

=1/(jC3)

1/(jC3) +R3=

1 jR3C31 + 2R23C

23

(14)

with the magnitude and phase to be

|HRC | = 1/

1 + 2R23C23 (15)

6 HRC = tan1(R3C3) (16)Values R3 = 11.01 k and C3 = 2 pF provides the necessaryphase shift of approximately 6 HRC = 310.7 but causes adecrease in magnitude of |HRC | = 0.6522 or -3.71 dB. Thetotal gain, G, of 21.11 dB is now needed to increase the Vppfrom 440mV to 5V. Each inverting amplifier 1 and 3 in Fig.10 has a gain , G1 = R2/R1 and G3 = R5/R4, where thetotal gain is the multiplication of G1 and G3. Gains, G1 andG3 can be set arbitrarily as long as the total gain equals 11.36or 21.11 dB. Experimentally, instead of using fixed resistors,variable resistors R2, R3, and R5 were used to manually tweakthe gain and phase. This is due to the amplifiers internal delaywhich also varies according to the gain setting. Fig. 12(B)shows the corrected reference signal for the slave transmittercoil. Only a slight phase offset of 5 is seen after phase andgain tweaking.

If the positions of the transmitters are static with a relativelystable power line channel condition, manually tweaking thegain and phase of the synchronization signal would suffice. Fora more dynamic solution with varying transmitter positions,an automatic gain and phase tuning system would be morepractical. This automatic solution could incorporate a feedbackchannel between receiver and transmitter to automaticallyadjust its driving signal at the slave transmitter for optimumpower transfer. Communications through the magnetic fielditself, for example, an RFID type communication system couldbe implemented. If a low frequency driving signal is used,



7.6 7.8 8 8.2 8.4 8.6x 106

65

60

55

50

45

40

35

30

25

20

15

10

Frequency, MHz

Mag

nitu

de, d

B

2TXDedicated2TXPLC

Fig. 13. Transfer Function |VL/Vs| (dB) for experimental measurementsfor dedicated and PLC synchronization techniques.

for example the Qi wireless power standard frequency range(100kHz - 200kHz), phase delays become inconsequential.This reduces the need of an automatic adjustment systemand synchronization through PLC proves to be a very simplemethod when compared to a wireless solution.

B. Experimental Results

Fig. 13 shows the measured transfer function of the multipletransmitter setup using the power line communication synchro-nization technique together with the dedicated wire technique.The phase and gain adjustment was tuned to work specificallyat 8.4 MHz resonant frequency. The frequency response ofthe power line channel could vary in both magnitude andphase at different frequencies. However, for this experimentalsetup, it is acceptable since the wireless power transfer systemis designed to work at resonance. At 8.4 MHz specifically,the transfer function of the PLC synchronization techniqueperforms equally with the dedicated wire technique.

VII. CONCLUSION

A multiple transmitter wireless power transfer scheme viacoupled magnetic resonance is analyzed using electrical circuittheory. For the case of two multiple transmit coils, the gainand diversity effect is presented. Negative effect of transmitterresonant coupling is shown theoretically and experimentally.Practical synchronization issues with regard to frequency andphase is presented for the multiple transmitter case. Exper-iments were also conducted to gauge the effect of resonantfrequency shifting due to nearby metallic objects. Frequencyshifts can be readjusted to the correct frequency by performingcapacitance tuning. A practical synchronization technique ispresented to ensure proper magnetic field combining at thereceiver coil. This was done via power line communicationswith appropriate gain and phase tuning. Future work includesintelligent transmit coils that communicate between each otherto enhance power efficiency depending on each coils environ-ment. This includes automatic capacitive tuning for incidentalresonance shifts for both transmitter and receiver coils andautomatic gain and phase tuning for synchronizing referencesignals between transmitters.

ACKNOWLEDGMENT

The authors would like to thank Barrett F. Robinson for histechnical insight and assistance in setting up the experimentalmeasurements.

REFERENCES

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[2] A. Kurs, A. Karalis, R. Moffatt, J.D. Joannopoulus, P. Fisher andM. Soljacic ,Wireless Power Transfer via Strongly Coupled MagneticResonances, Science, vol. 317, pp. 83-86, July 6 2007.

[3] A. Karalis, J.D. Joannopoulus, and M. Soljacic, Efficient wireless non-radiative mid-range energy transfer, Ann. Phys., vol. 323, pp. 34-48, Jan.2008.

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[6] J. Kim, H. Son, K. Kim, and Y. Park Efficiency Analysis of MagneticResonance Wireless Power Transfer with Intermediate Resonant Coil,IEEE Antennas and Wireless Prop. Letters, Vol.10, 2011.

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Rizal Johari (S12) received his B.S. and M.S.degree from Purdue University, West Lafayette, in2006 and 2012, respectively. He is currently workingtoward the Ph.D. degree in electrical and computerengineering at Purdue University.

In 2006 and 2008, he was an RF Hardware Engi-neer with Motorola Mobile Devices at Libertyville,IL. and Singapore AMK, where he played an im-portant role in the successful launch of the Motorola3G RAZR and Motorola Tundra mobile phones. Hisresearch interests include wireless power transfer,

software defined radio and power line communications. He is a member ofEta Kappa Nu and Tau Beta Pi.

James V. Krogmeier (M89) received the B.S.E.E.degree from the University of Colorado, Boulder,in 1981 and the M.S. and Ph.D. degrees from theUniversity of Illinois, Urbana-Champaign, in 1983and 1990, respectively.

He is currently a Professor with the School ofElectrical and Computer Engineering, Purdue Uni-versity, West Lafayette, IN. His research interests in-clude the application of signal processing in wirelesscommunications, adaptive filtering, channel equal-ization, synchronization, and intelligent transporta-

tion systems.Dr. Krogmeier has served on a number of IEEE technical program

committees, as an Associate Editor for the IEEE TRANSACTIONS ONWIRELESS COMMUNICATIONS, and as the Representative of the IEEESignal Processing Society to the IEEE Intelligent Transportation SystemsCommittee.

David J. Love (S98, M05, SM09) received theB.S. (with highest honors), M.S.E., and Ph.D. de-grees in electrical engineering from the Universityof Texas at Austin, in 2000, 2002, and 2004, re-spectively. During the summers of 2000 and 2002,he was with Texas Instruments, Dallas, TX. SinceAugust 2004, he is with the School of Electricaland Computer Engineering, Purdue University, WestLafayette, IN, where he is now a Professor andUniversity Faculty Scholar. He served as an Asso-ciate Editor for the IEEE Transactions on Signal

Processing and IEEE Transactions on Communications. He has also servedas a guest editor for special issues of the IEEE Journal on Selected Areasin Communications and the EURASIP Journal on Wireless Communicationsand Networking. His research interests are in the design and analysis ofcommunication systems, MIMO array processing, and array processing formedical imaging.

Dr. Love has been inducted into Tau Beta Pi and Eta Kappa Nu. Alongwith co-authors, he was awarded the 2009 IEEE Transactions on VehicularTechnology Jack Neubauer Memorial Award for the best systems paperpublished in the IEEE Transactions on Vehicular Technology in that year. Hewas the recipient of the Fall 2010 Purdue HKN Outstanding Teacher Awardand was an invited participant to the 2011 NAE Frontiers of EngineeringEducation Symposium. In 2003, Dr. Love was awarded the IEEE VehicularTechnology Society Daniel Noble Fellowship.

analysis and practical considerations in implementing multiple transmitters for wireless power...

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