awap 2016 - 전자파학회 · 2016-07-01 · awap 2016 asian workshop on ... koji suizu chiba...

Korean Instituteof ElectromagneticEngineering Society

AWAP 2016Asian Workshop on Antennas and Propagation

http://www.kiees.or.kr/awap2016

January 27-29, 2016, Centum Hotel, Busan, Korea

Organized by

Sponsored by

Korean Instituteof ElectromagneticEngineering Society

Korea Advanced Institute of Science & Technology

Technical Group on Antennas and Propagation of the Korean Institute of Electromagnetic Engineering and Science

Technical Committee on Antennas and Propagation of the Institute of Electronics, Information and Communication Engineers

Electromagnetic Group of the Electrical Engineering/Electronics, Computer, Telecommunications and Information Technology Association

IEEE Antennas and Propagation Society, Seoul Chapter

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Message from General Chairmen ........................................................ ii

Organizing Committee .......................................................................... iii

PROGRAM .............................................................................................. iv

Regular Session 27th January, 2016, Wednesday .................................................1

Plenary Talk and Invited Session 28th January, 2016, Thursday .......................23

Map ........................................................................................................47

Banquet .................................................................................................48

Transportation ........................................................................................49

Contents

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Welcome to AWAP2016 in Busan!

It is a great pleasure and a distinct honor to host the 3th 2016 Asian Workshop on Antennas and Propagation

(AWAP2016) which will be held at Centum Hotel in Busan, Korea, on January 27-29, 2016.

It has been a pleasure to work with our colleagues who have organized the program. Prof. Jae-Young Chung,

Prof. Minseok Kim, and Prof. Akkarat Boonpoonga helped put together a very strong and technical program

for comprising 34 papers. We do hope that these papers will be found to be intriguing and of enhanced scientific

quality by the participants. In this workshop, it will be organized with the single oral session delivered by invited

researchers and professors to encourage intimate professional discussion and the two parallel sessions for the

award contest.

AWAP2016 is a continuation of a series of annual antenna workshops held in Kanazawa, Japan (2014) and

Bangkok, Thailand (2015). The AWAP2016 is an annual forum for the exchange of information on the research

and development in antennas technologies, propagation, and related fields. AWAP2014 was the first AWAP and

was grown from the joint conference KJAP of two countries to the Asian Workshop in which all researchers and

students were welcomed from any countries from all over the world. The 2014 Asian Workshop on Antennas and

Propagation was held at the Kanazawa Theatre in Kanazawa, Japan from May 14 to 16, 2014. The 2015 Asian

Workshop on Antennas and Propagation (AWAP2015) was held at Swissotel Le Concorde in Bangkok, Thailand,

from June 17 (Wednesday) to 18 (Thursday), 2015.

The AWAP2016 is jointly organized by the Technical Group on Antennas and Propagation society of the Korean

Institute of Electromagnetic Engineering and Science (KIEES), Korea, the Technical Committee on Antennas

and Propagation of the Institute of Electronics, Information and Communication Engineers (IEICE), Japan, and

Electromagnetic Group of the Electrical Engineering/Electronics, Computer, Telecommunications and Information

Technology Association (ECTI), Thailand.

We would like to express our sincere gratitude to every author and participant whose high-level contributions

guaranteed the success of the AWAP2016. Moreover, we warmly thank Prof. Kyeong-Sik Min and the members

of the International Steering Committee as well as the Technical Program Committee for their constant help and

constructive advice.

Finally, thanks to the members of the Organizing Committee and the Secretariat for their support in the

completion of the Conference.

We hope that AWAP2016 will be stimulating, enjoyable, and fulfilling experience to all who attend it.

Wishing all the members a happy and prosperous 2016!

General Co-Chair of AWAP2016

Professor Seong-Ook Park (Korea)

Professor Qiang Chen (Japan)

Professor Titipong Lertwiriyaprapa (Thailand)

Message from General Chairmen of AWAP2016

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International Advisory Committee Members

Prof. Toshikazu Hori (University of Fukui, Japan)

Prof. Hiroyuki Arai (Yokohama National University, Japan)

Prof. Jaehoon Choi (Hanyang University, Korea)

Prof. Kin-Lu Wong (National Sun Yat-Sen University, Kaohsiung, Taiwan)

Prof. Monai Krairiksh (King Mongkut’s Institute of Technology Ladkrabang, Thailand)

Prof. Young Joong Yoon (Yonsei University, Korea)

Prof. Sangwook Nam (Seoul National University, Korea)

Prof. Kyeong-Sik Min (Korea Maritime and Ocean University, Korea)

Dr. Jin-Seob Kang (KRISS, Korea)

Prof. Keizo Cho (Chiba Institute of Technology, Japan)

Conference Co-Chairs

Prof. Seong-Ook Park (Korea Advanced Institute of Science and Technology, Korea)

Prof. Qiang Chen (Tohoku University, Japan)

Prof. Titipong Lertwiriyaprapa (King Mongkut's University of Technology, North Bangkok, Thailand)

Technical Program Committee Co-Chairs

Prof. Kangwook Kim (Gwangju Institute of Science and Technology, Korea)

Dr. Masayuki Nakano (KDDI R&D Laboratories Inc.)

Prof. Sarawuth Chaimool (Udon Thani Rajabhat University, Thailand)

Member: Prof. Kyung-Young Jung (Hanyang University, Korea)

Member: Prof. Keum Cheol Hwang (Sungkyunkwan University, Korea)

Member: Prof. Ick-Jae Yoon (Chungnam National University, Korea)

Local Arrangement

Prof. Dong-Kook Park (Korea Maritime and Ocean University, Korea)

Prof. Joong Han Yoon (Silla University, Korea)

Secretaries

Prof. Jae-Young Chung (Seoul Nat’l University of Science and Technology, Korea)

Prof. Minseok Kim (Niigata University, Japan)

Prof. Akkarat Boonpoonga, (King Mongkut's University of Techonolgy, North Bangkok, Thailand)

Organizing Committee

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PROGRAM

Number Start Min. 27th January, 2016, Wednesday, Regular Session

Session-R1 Session-R2

Authors Affilliation Title Chair Room Authors Title Affilliation Chair Room

1 15:00 15

Minseok Kim*, Tatsuki Iwata, Kento Umeki, Karma Wang-chuk, Jun-ichi Takada, and Shigenobu Sasaki

Niigata University, Tokyo In-stitute of Technology

Identification of Propagation Mechanism of Mm-Wave Outdoor Access Link

Keum Cheol Hwang

831(4th floor)

Nu Pham and Jae-Young Chung*

A dual-band GPS an-tenna integrated inside a military helmet

Seoul Nat'l Univer-sity of Science and Technology

Kyung-Young Jung

832(4th floor)

2 15:15 15Haewon Jung and Kangwook Kim*

Gwangju Institute of Science and Tech-nology

Subsurface GPR Imaging of Pavement Using MULSM Method

Minseok Kim

Naobumi Mich-ishita*, Naoto Nishiyama, Hisashi Morishita

Helmet Folded Dipole Antennas

National Defense Academy, Japan

Jae-Young Chung

3 15:30 15

Lakkhana Ban-nawat, Feaveya Kheawprae, and Akkarat Boonpoonga*

King Mongkut's University of Technol-ogy North Bankok

Improvement of Radar Target Identification with Near-field Calibra-tion Technique

Kang-wook Kim

Kunio Sakakibara*, Kei Firdaus, No-buyoshi Kikuma

Design of Frequency Selective Spiral Slot located in Near-Field of Wireless Power Trans-fer System by Eigen Mode Analysis

Nagoya Institute of Technology

Naobumi Michishita

4 15:45 15Jun Gi Jeong* and Young Joong Yoon

Yonsei Uni-versity

Gain enhanced compact bow-tie antenna with director

Akkarat Boon-poonga

Do-Gu Kang* and Jaehoon Choi

PIFA antenna for UWB applications

Hanyang Univer-sity

Kunio Sakaki-bara

16:00 15 Break

5 16:15 15Takeshi Fuku-sako*, Shohei Higashi

Kumamoto University

A Sensor Antenna for Non-destruc-tive Testing

Jae-Young Chung

831(4th floor)

Sarawuth Chaimool*, Prayoot Akkaraekthalin, Kwok L. Chung

Wideband Sequential-rotation arrays with circularly polarized Patch radiators using Anisotropic Metasur-face

Udon Thani Ra-jabhat University, King Mongkut's University of Technology North Bankok, Qingdao Technological Uni-versity

Seong-Ook Park

832(4th floor)

6 16:30 15Bo-Hee Choi and Jeong-Hae Lee*

Hongik University

4x4 Loop array for magnetic field control of wireless power transfer

Takeshi Fuku-sako

Myeongjun Kong, Geonyeong Shin, and Ick-Jae Yoon*

Electrically small spherical antennas using 3D printing tech-nology

Chungnam Nat'l University

Sarawuth Chaimool

7 16:45 15

Yuichi Kimura*, Fumihiko Nonaka, and Sakuyoshi Saito

Saitama University

Standing-wave and traveling-wave excitation of a microstrip antenna array fed by transverse slots on a broad wall of the rectangular waveguide for linear polarization parallel to the axis

Jeong-Hae Lee

Soon-Soo Oh*, Dong-Woo Kim, Tae-Hyung Kim, and Chi-Hyung Ahn

Beamwidth Reconfigu-rable Array Antenna Without Power Loss Using the Switched Coupler

Chosun University, Korea Railroad Research Institute

Ick-Jae Yoon

8 17:00 15

Jang-soon Park*, Jun-Bong Ko, and Dongho Kim

Sejong University

A Frequency-Reconfigurable Dipole Antenna Using a Tapered Impedance Match-ing Structure

Yuichi Kimura

Byeong-Yong Park, Tae-Wan Kim, and Seong-Ook Park*

Analysis of mode Splitting Behavior for Cylindrical Ferrite Resonator Antenna

Korean Advanced Institute of Science and Technology

Soon-Soo Oh

17:15 45 Break

18:00 150Welcome Reception

20:30

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number Start Min. 28th January, 2016, Thursday, Plenary Talk and Invited Session

8:30 40 Registration

9:10 15 Opening Ceremony, 831(4th floor)

9:25 30Plenary Talk, 831(4th floor) Chair

Toshikazu Hori* University of Fukui Low-Profile Design of Meta-Surface with Frequency Selective Surface and Its Application

Seong-Ook Park

Invited Session, 831(4th floor)

Authors Affilliation Title Chair

1 9:55 20 C. Rienthong, C. Kittiyanpunya, and M.Krairiksh*

King Mongkut’s Institute of Technology Ladkrabang

Surface moisture content sensor detecting mutual coupling magnitude between parallel and perpen-dicular dipole antennas (Invited paper)

Toshikazu Hori

2 10:15 20 Jinpil Tak, Eun Jeong, and Jaehoon Choi* Hanyang University Design of a Metamaterial Absorber for 24 GHz Auto-

motive Radar System (Invited paper)Monai Krai-riksh

10:35 15 Break

3 10:50 20 Ikmo Park* and Son Xuat Ta Ajou University Cavity-Backed Printed-Dipole Antenna for Millime-ter-Wave Applications (Invited paper) Jaehoon Choi

4 11:10 20 Jiro Hirokawa*, Dong-Hun Kim Tokyo Institute of Technology Waveguide Short-slot 2D-plane Coupler for 2D Beam-switching Butler Matrix (Invited paper) Ikmo Park

5 11:30 20 Ji Hwan Yoon and Young Joong Yoon* Yonsei University Millimeter-wave Reflectarray Antennas with Dual-reflector Configurations (Invited paper) Jiro Hirokawa

6 11:50 20 Yoshio Inasawa*, Takashi Tomura, Michio Takikawa, Hiroaki Miyashita

Mitsubishi Electric Corpora-tion

Gain Improvement of Shaped-beam Reflector Using Simultaneous Design of a Multimode Horn and Shap-ing Functions (Invited paper)

Young Joong Yoon

12:10 90 Lunch, Steering Committee meeting (18th floor meeting room)

7 13:40 20 Kin-Lu Wong* National Sun Yat-sen Univer-sity

Introduction to 5G Communications and its Smart-phone Antenna Design Perspectives (Invited paper) Yoshio Inasawa

8 14:00 20 Seungtae Ko*, Youngju Lee, Kwanghyun Baek, Yoongun Kim and Wonbin Hong Samsung Electronics Low Profile PCB Integrated mmWave Array Antenna

Solutions for 5G Mobile Communication (Invited paper) Kin-Lu Wong

9 14:20 20 Kentaro Nishimori* Niigata University Multi-beam massive MIMO using analog-digital hybrid configuration (Invited paper)

Jae-Young Chung

14:40 15 Break

10 14:55 20Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho*, Hiroaki Nakabayas-hi, Koji Suizu

Chiba Institute of Technology Measurement of Antenna Substrate by Collimated THz Waves (Invited paper)

Kentaro Nishi-mori

11 15:15 20 Kyeong-Sik Min* Korea Maritime and Ocean University

High-gain Multiband Spiral Antenna Design History for NLJD System (Invited paper) Keizo Cho

12 15:35 20 T. Imai*, K. Kitao, N. Tran, N. Omaki, Y. Okumura, and K. Nishimori

NTT DOCOMO, Niigata Uni-versity

A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band (Invited paper)

Kyeong-Sik Min

13 15:55 20 Jin-Seob Kang*, Jeong-Hwan Kim, and Jeong-Il Park

Korea Research Institute of Standards and Science

Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-Band (Invited paper) Tetsuro Imai

16:15 15 Break

14 16:30 20 Titipong Lertwiriyaprapa* and Montree Saowadee

King Mongkut’s University of Technology North Bangkok, Anunda Technology

Development of an Approximate UTD Ray Solution for EM Diffraction by a Planar Material Junction on PEC Ground Plane (Invited paper)

Jin-Seob Kang

15 16:50 20 Il-Suk Ko* Inha UniversityDirect Derivation of Closed-form Expression of Sommerfeld Integral for Impedance Half-plane from Exact Image Formulation (Invited paper)

Titipong Ler-twiriyaprapa

16 17:10 20 Hiroyuki Arai* Yokohama National Univer-sity

Optical beam scanning antenna for ultra high speed short range communication system (Invited paper) Il-Suk Ko

17 17:30 20 Sangwook Nam* Seoul National University An Electrically Small Isotropic Antenna Using Folded Split Ring Resonator (Invited paper) Hiroyuki Arai

17:50 10 Break

18:00 180Banquet

21:00

Start Min. 29th January, 2016, Friday, Technical Tour and Discussion

9:00 180Technical Tour and Discussion

12:00

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27th January, 2016Wednesday

Regular Session

The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016

Identification of Mm-Wave Radio Propagation Mechanism inOutdoor Access Links

Minseok Kim†, Tatsuki Iwata†, Kento Umeki†, Karma Wangchuk‡, Jun-ichi Takada‡, Shigenobu Sasaki†,†Graduate School of Science and Technology, Niigata University, Niigata, Japan

‡Graduate School of Science and Engineering, Tokyo Institute of Technology, Tokyo, JapanEmail: [email protected]

Abstract—This paper discusses the dominant propagationmechanisms in an outdoor environment through the channelmeasurement and ray tracing simulation at a mm-wave band of58.5 GHz where the measurements were conducted in an outdoorenvironment in Niigata university campus assuming the open areaoutdoor hotspot access scenario in 5G mobile systems.

I. INTRODUCTION

In future mobile systems, it will be necessary to operatein densely populated areas with increasing capacity and datarates to a much greater degree, and hence conventional cellularnetworks covering as large area as possible cannot be expectedto provide sufficient performance any longer. To develop superhigh bit-rate systems beyond 4G, there is a general consensusthat signal bandwidth should be significantly increased inaddition to utilize recent powerful transmission techniquessuch as MIMO (multiple-input-multiple-output technology),coordinated multipoint (CoMP), heterogeneous networks (Het-Nets), and carrier aggregation (CA). However, because ofserious congestion of the frequency spectrum of lower mi-crowave bands below 6 GHz, developing new frequency bandsshould be inevitable choices. Obviously, it is well knownthat free space propagation loss and shadowing loss are bothsignificantly increasing with frequency increase, which limitscommunication range. That is a reason why a small cellcommunication using high frequency bands within a confinedarea is currently gaining much attention [1], [2].

Currently, lots of studies argue the use of microwave andmillimeter wave (mm-wave) spectrum for cellular networkssuch as 28 and 38 GHz [1] which are allocated for localmultipoint distribution service (LMDS) and currently availablewith spectrum allocations of over 1 GHz bandwidth as wellas 60 GHz which offers 5 ∼ 9 GHz of unlicensed bandwidthin most countries [2]. However, radio propagation channelproperties at high frequency bands in such small cell mobileapplications have not been sufficiently studied. Moreover,there are few reports on the propagation properties in outdoorenvironments and the dominant propagation mechanisms havenot been thoroughly investigated by measurements.

In small cell environments, site-specific property of thepropagation mechanism is very important to develop a betterchannel model. As an initial step, in this study, the mm-wave propagation mechanism in an outdoor environment isidentified through the channel measurement and ray tracing

BS

MS

Fig. 1. Measurement campaign (topview).

(RT) simulation at a mm-wave band of 58.5 GHz where themeasurements were conducted in an outdoor environment inNiigata university campus assuming the open area outdoorhotspot access scenario in 5G mobile systems.

II. MEASUREMENT CAMPAIGN

In the measurement, the developed costom channelsounder has been used, which employs a commercialproduct of mm-wave Tx and Rx which integrate waveg-uide module with standard WR15/WG25 flange interfaces(V60TXWG1/V60RXWG1, VubIQ) [3]. It is configured in2 × 2 MIMO to measure full polarimetric channel responsesimultaneously. The RF transceivers employ a heterodyneIF architecture with variable frequency IF and RF mixersfor different RF channel selection, which requires a singlecommon synthesizer for IF and RF LO signal generation. Inthe typical setup, the baseband signal input power is adjustedby approximately −13 dBm, so that the transmit power ofapproximately 10 dBm is achieved by the power amplificationof 23 dB. We exclude the influence of the measurement systemfrom the measured channel responses by full MIMO back-to-back calibration (direct connection between Tx and Rx antennaports with a waveguide and an attenuator). The measurementdynamic range is limited to approximately less than 40 dB.

The measurement campaign was conducted in an outdooropen area as shown in Fig. 1 where Tx which was assumedto be the base station (BS) was located at around the centerof the area and the channel transfer functions were measuredat three MS positions. MS pos1 and MS pos2 were in line-of-sight (LoS) condition and MS pos3 was in obstructed-LoS(OLoS) condition. The antenna heights were 3 m for BS and1.5 m for MS. The area is surrounded by some buildings which

3


(a) MS pos1 (VV) (b) MS pos2 (VV)

(c) MS pos3 (VV) (d) MS pos3 (HH)

Fig. 2. Synthetic PDPs; (a)(b):LoS, and (c)(d):OLoS.

is 20 ∼ 30m far away from the BS antenna. The distancebetween BS and MS antennas was approximately 30 m.

For directional channel acquisition, high gain horn antennaswere rotated. The gains and half power beam-width (HPBW)are 15 dBi and 30 degrees for MS, and 24 dBi and 12 degreesfor BS, respectively. The BS and MS antenna were rotatedfrom 0 to 360 degrees in azimuth, and from 60 to 120 degreesin co-elevation. Azimuth and co-elevation at BS and MS werevaried in 12 and 30 degree steps, respectively.

III. RESULTS

From the measurement data, we obtained double directionalangle delay power spectrum (DDADPS). Then, the angularpower spectrum (APS) at both sides of the BS and MS,and the omni-directional power delay profile (PDP) weresynthesized from the DDADPS. For precise interpretation ofthe measurement results, the RT simulation was used wherethe maximum order of reflection were set to be three for LoSand two for OLoS conditions, respectively. The RT simulationemploys the image method. The first order diffraction wasfurther calculated only for OLoS condition based on uniformtheory of diffraction (UTD). This simulation calculated theray parameters of the received power, time delay of arrival,angles of departure and arrival for each path. For comparisonwith the measurement results, the simulation based DDADPSwas reconstructed from those parameters using the antennapatterns, then the APS and PDP were calculated in the samemanner as the measurement.

From the synthetic omni-directional PDPs of Fig.2, it canbe seen that a few significant multi-paths are observed besidesLoS path in the limited measurement dynamic range, and thedominant paths in the RT results are well matched to thosein the measurement results. On the other hand, using themeasured and simulated APS at both sides of BS and MS

(a) APS@BS

(b) APS@MS

Fig. 3. Synthetic APSs at 107.5 ns for MS pos3 (OLoS condition).

for the individual delay tap, the propagation mechanism wasidentified. The APS at delay tap of 107.5 ns for MS pos3 areshown in Figs.3. It illustrates that the propagation mechanismof that path is the edge diffraction on the vertical metallicpillar, which is supported by the PDP of MS pos3 showingHH-pol has a larger gain than VV-pol in Figs.2(c) and (d). Inthe same manner, all dominant propagation mechanisms wereidentified up to the third order specular reflection from wallsand the corresponding ground reflection, the penetration intoglass, and the first order diffraction. Some other observationsare summarized as follows.

• Only a few significant multi-paths were observed. Thesecond largest path powers for MS pos1 and MS pos2were 7 dB and 15 dB below LoS power, respectively.

• In OLoS condition, the path gain in HH-pol is sig-nificantly larger than VV-pol where the power of thediffracted path on the vertical edge was less than 5 dBfrom that of the largest reflected path in HH-pol.

• The detected paths include the corresponding groundreflected path due to the low measurement resolution. Theground reflection should be appropriately considered inthe channel model as a shadowing factor.

ACKNOWLEDGMENT

This work was partly supported by “The Strategic In-formation and Communications R&D Promotion Program(SCOPE: No.145004102)” and JSPS KAKENHI Grant Num-ber 15H04003.

REFERENCES

[1] T. Rappaport, et al., “Millimeter Wave Mobile Communications for 5GCellular: It Will Work!,” IEEE Access, Vol. 1, 2013.

[2] MiWEBA, FP7 ICT-2013-EU-Japan, http://www.miweba.eu[3] M. Kim, K. Umeki, K. Wangchuk, J. Takada, S. Sasaki, “Polarimet-

ric Mm-Wave Channel Measurement and Characterization in a SmallOffice,” Proc. PIMRC 2015, Aug. 2015.

4


Subsurface GPR Imaging of Pavement Using MULSM Methodo Haewon Jung and Kangwook Kim

Gwangju Institute of Science and Technology

[email protected] and [email protected]

Ⅰ. IntroductionIn pavement inspection, the ground penetrating radar

(GPR) is frequently used to inspect the subsurface

structures, such as pavement thickness, reinforcement bars

(rebars), water pipes, etc. It can also be used to detect

subsurface cavities nondestructively with high resolution

features.

Generally, the pavement is a multilayer geometry with

asphalt, concrete, sand, and etc. In a multilayer geometry,

multilayer Stolt migration (MULSM) method can be used

to produce migrated images [1]. The MULSM is a hybrid

method in such a way that the phase-shift migration [2] is

recursively applied to each boundary between layers, and

then the Stolt migration [3] is used within the

homogeneous layer under each boundary.

In this work, GPR survey was conducted in experimental

pavement geometry, and the collected data was imaged to

identify subsurface targets using the MULSM method.

Ⅱ. Experiment and ImagingThe pavement geometry that contains cavities, brick,

metal sheets, and rebar is illustrated in Fig. 1, where the

cavities are modeled as a block of Styrofoam. The top and

bottom layers are asphalt and sand with relative

permittivity of approximately 6 and 4, respectively.

The antenna array data is obtained from a synthesized

aperture of monostatic radar that is composed of a vector

network analyzer, resistive vee dipole antenna [4]. The

input waveform is a differentiated Gaussian pulse with a

peak frequency at 2.25 GHz.

The migrated result under pavement is depicted in Fig. 2

as an isosurface image. The targets are seen to be well

migrated under multilayer pavement geometry.

Ⅲ. ConclusionIn this work, the GPR is used to collect data under

pavement that contains objects. The data is migrated using

MULSM method and the results are well matched to the

actual target positions.

Figure 1. Experiment setup.

Figure 2. Isosurface image of migrated result.

Acknowledgment

This work was supported by ICT R&D program of

MSIP/IITP [10041950, Development of mobile safety-

inspection systems using high resolution penetration

imaging technology for transportation infrastructure].

References[1] Y. C. Kim, R. Gonzalez, and J. R. Berryhill,

“Recursive wavenumber-frequency migration,” Geophysics, vol. 54, no. 3, pp. 319-329, Mar. 1989.

[2] J. Gazdag, “Wave equation migration with the phase-shift method,” Geophysics, vol. 43, no. 7, pp. 1342-1351, Dec. 1978.

[3] R. H. Stolt, “Migration by Fourier transform,” Geophysics, vol. 43, no. 1, pp. 23-48, Feb. 1978.

[4] K. Kim and W. R. Scott, “Design of a resistively loaded vee dipole for ultrawide-band ground-penetrating radar applications,” IEEE Trans. Antennas Propagat., vol. 53, no. 8, pp. 2525-2532, Aug. 2005.

5


Improvement of Radar Target Identification

with Near-field Calibration Technique Lakkhana Bannawat, Feaveya Kheawprae, and Akkarat Boonpoonga

Department of Electrical and Computer Engineering, Faculty of Engineering, King Mongkut’s University of Technology North Bangkok, Thailand

[email protected], [email protected] and [email protected]

Ⅰ. Introduction Radar target identification has been extensively studied

in research area of electromagnetic wave propagation.

After transmitting an impulse signal to a target,

electromagnetic field scattered from the target is utilized to

detect and characterize objects of various shapes and

constitutions. The singularity expansion method (SEM)

was introduced to model the late- time portion of the

electromagnetic transient response of a target irradiated by

an electromagnetic pulse as a sum of damped exponentials

with complex natural frequency [1]. A complex frequency

which is often referred as a pole extracted from late time

response is a tool for aspect-independent target

identification. A Matrix pencil method (MPM) is one of the

most popular techniques which are widely employed to

extract the poles [2]. Recently the MPM was slightly

modified as the short-time matrix pencil method (STMPM)

which can resolve the problem of finding the

commencement of late-time portion [3].

In the paper, we show another problem of applying the

SEM to identify the object in the practical situation. In the

radar system, the antenna is an essential component to

transmit and receive the EM pulse. However, this

component impact on the accuracy of pole. This paper

presents a calibration technique to reduce the degradation

of pole due to the antenna response.

Ⅱ. Simulations and Results Simulations are conducted to verity the proposed

technique by using electromagnetic software simulator.

The PEC cube with dimension of 20 cm and sphere with

radius of 20 cm are modeled as radar targets. The bow-tie

antenna is employed as both transmitting and receiving

antennas. The distance between the antennas is 60 cm. The

calibration technique proposed in [4] is applied to reduce

the effect of the antenna response on the pole extracted by

using STMPM. Figure 1 and 2 show the natural frequency

extracted by using STMPM without and with the proposed

technique, respectively. At the late-time portion, note that

the natural frequencies of PEC sphere and cube, extracted

by using STMPM without the calibration technique are

almost identical. These frequencies cannot be used to

identify the targets. To resolve this problem, the calibration

technique proposed in [4] is applied before extracting poles.

Figure 2 clearly reveals that the nature frequencies of PEC

sphere and cube, extracted by using STMPM without the

calibration technique is separate.

Figure 1. Natural frequency extracted by STMPM without

the proposed technique.

Figure 2. Natural frequency extracted by STMPM with the

proposed technique.

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Ⅲ. Conclusion This paper has presented an improvement of a radar

target identification by using a near-field calibration

technique. In the radar system, a target can be identified

based on a singularity expansion method (SEM) principle.

Poles representing the signature of a target is extracted by

using a short-time matrix pencil method (STMP).

Transmitting and receiving antennas are the important parts

of the radar system for transmitting and receiving the radar

pulse. In the practical situation, the response of the antenna

impacts on the accuracy of extracted poles. To resolve the

underlying problem, we introduce an efficient technique of

the antenna calibration technique in order to reduce its

effect. Simulations were conducted to verify the proposed

technique. The results show the increase of accuracy of

extracted pole.

References [1] C. E. Baum, E. J. Rothwell, K. Chen, and D. P.

Nyquist, “The Singularity Expansion Method and Its Application to Target Identification,” Proceedings of the IEEE, Vol. 79, No. 10, Oct. 1991

[2] T. K. Sarkar and O. Pereira, “Using the matrix pencil method to estimate the parameters of a sum of

complex exponentials,” IEEE Antenna and Wireless propagation magazine, vol. 37, pp. 44-55, 1995.

[3] R. Rezaiesarlak and M. Manteghi, “Short-Time

Matrix Pencil Method for Chipless RFID Detection

Applications,” IEEE Trans. on Antennas and Propagation, Vol. 61, No. 5, May 2013.

[4] V. A. Mikhnev and P. Vainikainen, "Single-

Reference Near-Field Calibration Procedure for Step-

Frequency Ground Penetrating Radar," IEEE Trans.

on Geoscience and Remote Sensing, vol. 41, No. 1

pp. 75-80, Jan. 2003.

7


Gain Enhanced Compact Bow-tie Slot Antenna with Director oJun Gi Jeong, Young Joong Yoon

Department of electrical and electronic engineering, Yonsei University

[email protected]

.Ⅰ IntroductionUltra-wide Band (UWB) applications are used in many

modern systems, and several types of antennas are used. In

many cases, bow-tie type antenna is used for the UWB

applications because of the dimensional compactness and

the broadside pattern. Typically, reflector is used at the

back side of antenna to make the unidirectional pattern. In

addition, several methods are proposed to enhance the gain

of bow-tie antenna[1] for high gain applications recently.

However, previous cases are occupy a large area.

In this paper, gain enhanced compact bow-tie antenna is

proposed. It has a director to enhance the gain of antenna

and also it has compact size. Thus, this characteristics are

useful for compact UWB systems for high gain.

.Ⅱ Design and resultDesigned antenna is shown in Fig. 1. The proposed

antenna has three layers and theses are designed in the

same planar area. Director is composed of short lines and it

is arrayed in both sides (top, bottom) of substrate.

The Electric field from the bow-tie antenna is induce the

current at the short lines of the director, and electric field is

re-radiated from the short lines. It is similar with the

operation principle of quasi Yagi-Uda antenna.

Designed antenna is operating at the upper band of

UWB (6.09 ~ 11.28 GHz) as shown in Fig. 2. Also,

antenna gain is enhanced about 0.5 ~ 1.6 dB at the overall

operating band as shown in Table 1.

Antenna parameters are determined as L=31mm,

W=13mm, h1=11mm, h2=7mm, l1=13mm, l2=4.2mm,

l3=6.35mm, respectively.

Figure 1. Configuration of proposed antenna.

Figure 2. S-parameter of proposed antenna.

Table 1. Comparison of antenna gain [dB]

7GHz 8GHz 9GHz 10GHz

Without director 6.2 7.8 8.5 8.5

With director 6.7 8.6 9.5 10.1

.Ⅲ ConclusionA gain enhanced compact bow-tie slot antenna is

proposed. It is operating at the upper band of UWB. Also,

the gain of antenna is enhanced in overall operation band.

Thus, the proposed antenna can be used to high gain UWB

applications.

Acknowledgement

This research was supported by the MSIP(Ministry of

Science, ICT and Future Planning), Korea, under the

ITRC(Information Technology Research Center) support

program(IITP-2015-H8501-15-1019) supervised by the

IITP(Institute for Information & communications

Technology Promotion)

References

[1] Shi-Wei Q., Chi-Hou C. and Quan X.,

“Ultrawideband composite cavity-backed folded

sectorial bowtie antenna with stable pattern and high

gain," IEEE Trans. Antennas and Propagation, vol.

57, no. 8, Aug. 2009.

8


A Sensor Antenna for Non-Destructive Testingo Takeshi Fukusako and Shohei Higashi, Kumamoto University

[email protected]

1. IntroductionSensor antennas have been widely used recently in several fields. A sensor antenna using RFID Chiphas been proposed for detecting the relative permittivity of material[1][2]. Although there is a variety of sensing method for concrete, we proposed a sensor antenna which can detect the relative permittivity of concrete by measuring the resonant frequency and transmitted power. 2 The sensing system

A sensor antenna and a reader antenna are used in the proposed sensing system. The sensor antennaequipped with RFID is pasted on a concrete block.The sensor antenna is provided with the power through the reader antenna. When concrete is changed from dry state to wet one, the relative permittivity of concrete becomes high. In this case, the resonant frequency of sensor antenna is shifted lower. The readercan detect the shifted frequency with scanning. The frequency shift indicates the wed degree of concrete.3 Antenna StructureFig.1 shows the proposed structure. The proposed antenna consists of four parts consisting of ground plane, feed part, meander-line antenna and additional microstrip elements. There is a slit in the ground plane in order to be sensitive for the permittivity shift of concrete. One tip of the meander line antenna is shorted to feed part. In addition, four microstrip elements along a slit nearby the shorting part. As shown in Fig. 2, this sensor antenna is pasted on the concrete, and a dipole antenna is used as the reader antenna. The distance between the sensor antenna and the reader antenna is 1 m.4 Simulation resultsFig.3 shows simulation results, where S11 and S21 characteristics are analyzed when the relative permittivity of the concrete is between 4 and 10.S11 characteristics are shifted low with an increase of the relative permittivity. In the S21 characteristics, the reader antenna can obtain high transmitted power with a keep peak at the resonant frequency when the relative permittivity of concrete is 4,5,8 and 10. However, the keen peak is not clear for other values of relative permittivity.5 ConclusionA sensor antenna for non-destractive testing has been proposed. The proposed antenna has a narrow band characteristics so as to make a keen peak in transmitted power characteristics. This contribute to a precise measurement of permittivity.

Fig.1 Proposed sensor antenna structure

References [1]R.Suwalak , C.Phongcharoenpanich , D.Torrungrueng, and M.Krairiksh,“DETERMINATION OF DIELECTRIC PROPERTY

OF CONSTRUCTION MATERIAL PRODUCTS USING A NOVEL RFID SENSOR " Progress In Electromagnetics Research , Vol.130, 601-617, 2012[2] F. Yang, Q. Qiao, and A. Z. Elsherbeni, “Reconfigurable sensing antennas: concept, design, and applications,” Antennas

and Propagation in Wireless Communication (APWC2013), pp. 748-752, Torino, Italy, September 2013.

Fig.2 Simulation model

Fig.3 Simulated results

9


4x4 Loop array for magnetic field control of wireless power transfer

Bo-Hee Choi and o Jeong-Hae Lee

Department of Electronic Information and Communication Engineering, Hongik University,

Seoul 121-791, Korea

[email protected]

Ⅰ. IntroductionAngular misalignment of a receiver is sensitive issue of

wireless power transfer (WPT) because it reduces power

transfer efficiency (PTE). Three-dimensional structure with

multi-sources was researched for magnetic-field control [1].

Later on, planar-type loop array controlling magnetic-field

with a source was presented [2]. It has an advantage for

practical space in usage and it can be a key technology to

improve a reduced PTE of a misaligned receiver. In this

paper, a 4x4 loop array for magnetic-field control will be

presented to improve a PTE of a misaligned receiver in

wireless power transfer.

Ⅱ. Magnetic-field control loop arrayFig. 1 shows the configuration and dimension of 4x4

loop array with a source at the first loop. A receiver is

50cm away from the center of array and can be rotated 0°,

45°, and 90°. The receiver size is the same as that of one

array element. The operating frequency is 6.78MHz.

The loop array with a source and a load can be expressed

in an equivalent circuit [2]. The optimum C2, …, Cn, and

RL are determined concurrently using genetic algorithm

(GA) to obtain the maximum efficiency. The C1 is given by

Im(Zin)=0 and RS is set to be Re(Zin), where Zin is input

impedance at the source.

Figure 1. Configuration of 4x4 loop array

Fig. 2 shows the PTE according to a receiver angle

compared with one large loop which has the same size as

the loop array. In the cases of the receiver angle is 0° and

45°, one large loop has the higher efficiencies than the loop

array. However, when the receiver angle becomes 90°, the

one large loop has zero efficiency while the loop array PTE

is still high. The results show that the power is transferred

to an orthogonal receiver by controlling magnetic-field of

loop array.

0 45 90

0

20

40

60

80

100

Effic

ienc

y (%

)

Rotation angle (˚)

Rotated forx-axisy-axisx,y-axisy-axis(One Large Loop)

Figure 2. Power transfer efficiency vs. receiver angle.

Ⅲ. ConclusionThe control of magnetic-field is demonstrated by a 4x4

loop array. By designing the proper values of capacitance

of loop, the magnitude and phase of loop currents can be

controlled and, thus, magnetic-field control is possible.

Therefore, the loop array can transfer power efficiently to

an orthogonal receiver while the conventional orthogonal

loops have zero efficiency. The appropriate capacitances of

loop array are determined using GA for maximum

efficiency.

References

[1] Y. Lim and J. Park, “A Novel Phase-Control-Based

Energy Beamforning Techniques in Nonradiative

Wireless Power Transfer,” IEEE Trans. Power

Electron., vol. 30, no. 11, pp. 6274-6287, Nov. 2015.

[2] B-H Choi, B-C Park, and J-H Lee, “Near-field

Beamforming Loop Array for Selective Wireless

Power Transfer,” IEEE Microw. Wireless Compon.

Lett., vol. 25, no. 11, pp. 748-750, Nov. 2015.

10


Standing-wave and traveling-wave excitation of a microstrip antenna array

fed by transverse slots on a broad wall of the rectangular waveguide for

linear polarization parallel to the axis o Yuichi Kimura, Fumihiko Nonaka, and Sakuyoshi Saito

Dept. Electrical and Electronic Systems, Graduate School of Science and Engineering, Saitama University

[email protected]

I. Introduction Waveguide slot arrays are commonly used for various

applications such as radars and communication systems. A

longitudinal slot array on a broad wall of the rectangular

waveguide is one of the typical designs for polarization

perpendicular to the waveguide axis. For transverse slot

arrays on a broad wall of the waveguide for polarization

parallel to the axis, reduction of grating lobes becomes a

significant problem because transverse slots are arranged at

a spacing of one guided wavelength that is usually longer

than a wavelength in free space. In order to shorten the slot

spacing, a dielectric-filled waveguide or slow wave

structures in the waveguide are utilized for the transverse

slot arrays [1]. Another solution that uses T-slots arranged

on a ridged waveguide is reported [2].

For that purpose, the authors have proposed a novel

planar array antenna on a broad wall of the rectangular

waveguide for linear polarization parallel to the axis [3]. A

two-element series-fed microstrip antenna (MSA) array

placed on a dielectric substrate, which is excited by a

transverse slot on the broad wall of the waveguide, is used

as an element of the proposed array. It is revealed that a

coupling power ratio of the array element can be controlled

from a few % to around 50% by tuning the dimensions of

the array element [4]. In this paper, array designs of the

proposed array antenna with standing-wave excitation and

traveling-wave excitation are presented.

II. Design of standing-wave excitation array

Figure 1 presents a configuration of the proposed

microstrip antenna array on a broad wall of the rectangular

waveguide with standing-wave excitation. Microstrip patch

antennas on a dielectric substrate are arranged on the broad

wall and two patches are connected by a microstrip line.

The series fed patch array is excited by a transverse slot on

Figure 1. The proposed array with standing-wave

excitation.

(a) E-plane (b) H-plane

Figure 2. Radiation patterns.

the broad wall. In this design, the relative dielectric

constant of the substrate is 2.6. Then, the spacing of the

two patches is approximately 0.62 wavelengths in free

space, which is corresponding to around a half of the

guided wavelength in the waveguide. Thus, the grating

lobes of the transverse slots can be suppressed. Polarization

of the proposed array is parallel to the axis of the

waveguide. In this example, six patches and three

transverse slots are arranged, where inset-feeding method

is used for the patches. One of the ends of the waveguide is

set to the feeding port and the other is terminated by a short.

The coupling power ratio required for the three array

1.2

10.2

22.9

1.2

a

a

a

a

a

a

b

bb

bb

b

lf

lf

lf

33.4

33.4

33.0

Port 1

Short

25.3

126

wls

wls

wls

cd

c d

cd

c d

cd

c d

a = 8.4b = 8.4c = 2.0d = 2.0w = 1.0ls = 10.4lf = 13.2unit:[mm]

θ [deg]

E 2 [d

B]

030

60

90 0

120

150180

30

60

90

120

150

0

-20

-20 -20 0

-20

0

Exp. Co-pol. Sim. Co-pol. Exp. X-pol. Sim. X-pol.

θ [deg]

E 2 [d

B]

030

60

90 0

120

150180

30

60

90

120

150

0

-20

-20 -20 0

-20

0

11


elements is 33% for standing-wave excitation array. The

design frequency is 11.2 GHz.

Figure 2 presents the simulated and measured radiation

patterns in E- and H-planes at 11.2 GHz, where the

simulated result is obtained by Ansys HFSS. It is found

that the grating lobes at around 50 degree directions in the

E-plane pattern produced by the transverse slot array

arranged at one guided wavelength are well suppressed.

III. Design of traveling-wave excitation

array Figure 3 presents the proposed microstrip antenna array

on a broad wall of the rectangular waveguide with

traveling-wave excitation. In this design, sixteen patches

and eight transverse slots are arranged. The inset-feeding is

used for the first five elements near the feeding port and

the edge feeding is used for the last three elements. In order

to create uniform excitation distribution, the coupling

power ratio for the element #n numbered from the last is

set to 1/n. Furthermore, the forward beam tilting design is

introduced to suppress the reflection at the feeding port.

The spacing of the transverse slots are slightly apart from

one guided wavelength.

Figure 4 presents the simulated and measured radiation

patterns in E-plane at 11.2 GHz. The grating lobes at

around 50 degree directions produced by the transverse slot

array arranged at one guided wavelength are not observed.

Figure 3. The proposed array with traveling-wave

excitation.

Figure 4. E-plane radiation patterns.

IV. Conclusion A microstrip antenna array fed by transverse slots on a

broad wall of the rectangular waveguide for linear

polarization parallel to the axis is presented. Validity of the

proposed array with standing wave excitation and

traveling-wave excitation is confirmed by simulation and

measurement.

References [1] S. Yamaguchi, et al., “A slotted waveguide array

antenna covered by a dielectric slab with a post-wall

cavity,” IEICE Tech. Rep., vol. 112, no. 7, AP2012-5,

pp. 21-26, Apr. 2012.

[2] S. Mihara and N. Kuga, “T-slot antenna on the ridged

plane of a ridged waveguide,” IEICE Trans. (B), vol.

J95-B, no. 9, pp. 1052-1059, Sep. 2012.

[3] Y. Kimura and F. Nonaka, “A Microstrip Antenna

Array on a Broad Wall of the Rectangular

Waveguide with Polarization Parallel to the Axis,”

Proc. 2013 Korea-Japan Workshop on Antennas and

Propagation, p. 8, Jan. 2013.

[4] F. Nonaka, S. Sakuyoshi Saito, and Y. Kimura,

“Design of a planar array antenna on a broad wall of

the rectangular waveguide for polarization parallel to

the axis with standing-wave excitation,” IEICE Tech.

Rep., vol. 114, no. 354, AP2014-156, pp. 31-36, Dec.

2014. 22.9

1.2

10.21.2

35.0

Port 1

Short25.3

305

35.0

35.0

35.0

16.5

35.0

35.0

35.0

lf1

lf1

lf2

lf3

lf3

lf3

lf3

lf4

ls1

ls1

ls2

ls3

ls4

ls5

ls6

ls7

a

a

a

a

a

a

a

a

a

a

a

a

a

a

a

a

b1

b1b1

b1b2

b2b3

b3b4

b4b2

b2b5

b5b2

b2

c d

c d

c d

c d

c d

c d

c d

c d

c d

c d

w1

w1

w2

w2

w2

w2

w2

w2

2.0

2.0

2.0

2.0

2.0

2.0

2.0

2.0

#1

#2

#3

#4

#5

#6

#7

#8a = 8.4, b1 = 8.4, b2 = 10.8, b3 = 9.0, b4 = 10.4, b5 = 11.2,c = 2.0, d = 2.0, lf1 = 12.0, lf2 = 15.2, lf3 = 13.2, lf4 = 13.6, w1 = 4.4, w2 = 1.0, ls1 = 18.6, ls2 = 9.8, ls3 = 9.4, ls4 = 9.0, ls5 = 8.6, ls6 = 7.9, ls7 = 7.8 unit :[mm]

-90 -60 -30 0 30 60 90-40

-30

-20

-10

0

Angle [deg.]

Rel

ativ

e A

mpl

itude

[dB

]

Exp. Co-pol. Sim. Co-pol. Exp. X-pol. Sim. X-pol.

12


A Frequency-Reconfigurable Dipole Antenna Using a Tapered Impedance

Matching Structure

Jang-soon Park, Jun-Bong Ko, and o Dongho Kim

Department of Electronic Engineering, Sejong University

[email protected]

I. IntroductionRapid appearance of diverse wireless communication

services has continuously been increasing the demand for

frequency reconfigurable antennas. For wideband operation,

good impedance matching between an input port and an

antenna’s radiating part is necessary. Accordingly, we

propose a versatile wideband balun structure enables good

impedance matching in a broad frequency range.

II. Design and experimentThe geometry of a proposed antenna is given in Fig. 1.

The antenna consists of a microstrip line, a parallel plate

line, a tapered balun, and two radiating arms with inserted

varactor diodes, which is fed by a 50 • coaxial connector.

The lines and arms have been etched on each side of a 1.52

mm thick commercial Taconic RF-35 dielectric substrate.

In order for frequency scan in a wide frequency range,

good impedance between the input port and the dipole

antenna is necessary. To do that, the tapered parallel plate

line has been introduced as show in Fig. 1, which not only

transfers waves to the dipole antennas (arms) with low

reflection but changes field distribution suitable for dipole

radiation. In fact, the input impedance varies from 50 • to

196 • .

The varactor diodes (SMV-1405) from Skyworks have

been used to electrically scan the resonant frequency of the

antenna. A 5 pF DC blocking capacitor and a 20 nH RF

choke inductor have been used to prevent the undesirable

influence of DC and RF signals, respectively.

x y

z

Varactor diode

RF choke

DC blolck

Parallel plate line

Figure 1. Fabricated antenna

2.5 3.0 3.5 4.0 4.5-35

-30

-25

-20

-15

-10

-5

0

S11

[dB

]Frequency [GHz]

0v Mea. 30v Mea. 0v Sim. 30v Sim.

Figure 2. Comparison of the simulated and measured

reflection coefficient with different values of bias voltage.

The simulated and measured reflection coefficient is

shown in Fig. 2, which proves our antenna successfully

hops from 3.35 GHz to 3.78 GHz when the bias voltage

switches from 0 V to 30 V.

III. ConclusionWe have proposed the tapered balun structure which

shows good impedance matching in a wide frequency

range and can be used in various antenna applications.

AcknowledgementThis work was supported by Institute for Information &

Communications Technology Promotion (IITP) grant

funded by the Korea government (MSIP). [R-20150224-

000291, Development on Semi-conductor based Smart

Antenna for Future Mobile Communications]

References

[1] P. Y. Qin, A. R. Weily, Y. J. Guo, T. S. Bird, and C.

H. Liang, “Frequency reconfigurable quasi-Yagi

folded dipole antenna,” IEEE Trans. Antennas

Propag., vol. 58, no. 8, pp. 2742-2747, Aug. 2010.

13


A Dual-Band GPS Antenna Integrated Inside a Military Helmet

Nu Pham and oJae-Young Chung

Seoul National University of Science and Technology

[email protected]

Ⅰ. IntroductionRecently, the capability of precisely locating and

tracking a device, vehicle or human has received more and

more attention. However, the positioning Error of a

conventional Global Navigation Satellite system (GNSS) is

limited to several meters due to multipath, ionospheric and

trospheric delays. One method to mitigate such errors is

using a dual-band operation system implemented with a

dual-band, dual-circularly polarized (CP) antenna.

Here we present a design of dual-band antenna

integrated in a military helmet for tracking a soldier in the

field. The standalone antenna was presented in AWAP2015,

Bangkok, Thailand [1]. The contents of this paper focus on

the implementation of the antenna inside an helmet in a

limited installation area and with possible head effects.

Ⅱ. Antenna Installation and SimulationThe standalone dual-band antenna was designed to

operate at L1 (1.57 GHz) and L2 (1.23 GHz) bands of

Global Positioning System (GPS). Figure 1 depicts the

antenna integrated in the military helmet. This microstrip

GPS antenna has a single feed fed by the side of the PCB

to ease the connection between the antenna and an external

receiver. Also, it is compact and low-profile suitable for

installation. The antenna is as small as 73mm×73mm×6.4

mm, corresponding to 0.29• ×0.29• ×0.026• at 1.227 GHz.

In the full-wave simulation model, a 3D model of

helmet was imported from a CAD file. In addition, a

spherical head phantom was designed to investigate the

head effect on the antenna performance. The head phantom

consists of three materials: skin, skull and brain tissue, and

their material properties are assigned based on [2].

Figure 2 shows the comparisons of antenna reflection

coefficient (S11) and axial ratio (AR) when the antenna is

in the free-space, with helmet, and with helmet and head

phantom. It can be seen that the S11 and AR are shifted to

the higher frequency as the helmet exists. On the other

hand, the effect of phantom is trivial due to the large

ground isolating the high loss head phantom.

Figure 1. GPS antenna mounted in helmet simulation.

Figure 2. Comparisons S11 and axial ratio.

Ⅲ. ConclusionThe proposed method of attaching antenna inside the top

of helmet takes an important role on remaining operation

of standalone antenna, adapt with wearable requirement in

military equipment.

Acknowledgement

This work was supported by the Basic Science Research

Program through the NRF Korea funded by the Ministry of

Science, ICT & Future Planning (No.

2013R1A1A1005735).

References

[1] N. Pham and J.-Y. Chung, "A miniaturized circular

patch antenna for dual-band GPS application,"

AWAP2015, Bangkok, Thailand, 2015.

[2] http://niremf.ifac.cnr.it/tissprop/htmlclie/htmlclie.php

1.1 1.2 1.3 1.4 1.5 1.6 1.7-50

-40

-30

-20

-10

0

Frequency [GHz]

S11

(dB

)

StandaloneHelmetHelmet+Phantom

1.1 1.2 1.3 1.4 1.5 1.6 1.70

3

6

9

12

Frequency [GHz]

AR

(dB

)

StandaloneHelmetHelmet+Phantom

Helmet

Patch Antenna Large ground

BrainSkin

Skull

14


Helmet Folded Dipole Antennas

Naobumi Michishita, Naoto Nishiyama, and Hisashi Morishita

National Defense Academy

[email protected]

I. IntroductionHelmet antennas have been developed for integrating

into the helmet for disaster prevention [1]. To reduce the

degradation of antenna characteristics due to the human

head, an inverted-F antenna with a copper ring structure

has been proposed [2]. However, the inverted-F antenna

has the narrow bandwidth. This paper presents the folded

dipole antenna with the slit-loaded copper ring structure.

II. Antenna CharacteristicsFigure 1 shows the simulation models of the proposed

folded dipole antenna with human head. The antenna

element is arranged at 5 mm from the edge of the

hemispherical dielectric shell with a diameter of 125 mm.

To achieve the impedance matching, the widths of the feed

and non-fed arms of the dipole are 3 mm and 12 mm,

respectively. Fig. 1(b) shows the slit-loaded copper ring

structure.

Figure 2 shows the simulated VSWR characteristics. The

relative bandwidth at VSWR = 3 becomes 2.7% and 1.9%

with and without the slit, respectively. Figure 3 shows the

radiation efficiencies. The efficiency of 8.6% can be

improved by loading the slit. Figure 4 shows the simulated

10 g average local SAR distributions. The unwanted

radiation toward the human head can be suppressed and

SAR value is reduced.

III. ConclusionThe proposed helmet folded dipole antenna with slit-

loaded copper ring structure has high radiation efficiency

and low SAR value.

References

[1] T. Nakao, H.T. Nguyen, M. Nagatoshi, and H.

Morishita,“Fundamental study on curved folded

dipole antenna,” IEEE AP-S Int. Symp., Chicago, IL,

pp.1-2, July 2012.

[2] N. Nishiyama, N. Michishita, and H. Morishita,

“Low-frequency inverted-F antenna on annular

ground plane,” IMWS-Bio, Taipei, Taiwan, pp.143-

144, Sept. 2015.

z

yx

111

22

Human head

[Unit: mm]

15

Dielectric shell

Slit

Copper

(a) (b)

Figure. 1 (a) Helmet folded dipole antenna with human

head. (b) Slit-loaded copper ring structure.

140 145 150 155 1601

2

3

4

56

7

8

9

10

w/o Slitw/ Slit

V

SWR

Frequency [MHz]

Figure 2 VSWR characteristics.

140 145 150 155 1600.0

0.2

0.4

0.6

0.8

1.0

w/o Slitw/ Slit

Rad

iati

on e

ffic

ienc

y

Frequency [MHz] Figure 3 Radiation efficiencies.

1[W/kg]

0

(a) (b)

Figure 4 10 g average local SAR distributions of (a) with

and (b) without slit.

15


Design of Frequency Selective Spiral Slot located in Near-Field

of Wireless Power Transfer System by Eigenmode Analysiso Kunio Sakakibara, Kei Firdaus, Nobuyoshi Kikuma

Nagoya Institute of Technology

[email protected]

Ⅰ. IntroductionA transmitting circuit of a wireless power transfer

system is shielded by a metal case to prevent noise

emission. On the other hand, a receiving circuit equipped

in cars or mobile devices are located outside of the shield

case. Therefore, the transmitting and receiving antennas are

separated by the metal plate. A frequency selective spiral

slot is proposed to be cut on the metal plate. Only the

transmitting power in the operating frequency of the spiral

slot can transmit to the receiving antenna. The spiral slot is

designed by eigenmode analysis individually from the

wireless power transfer system. The simulated and

measured performances are demonstrated in this paper.

Ⅱ. AnalysisTransmitting and receiving helical antennas are

separated by an infinite ground plane with the frequency

selective spiral slot as shown in Fig. 1. The two areas of the

transmitting and receiving antennas are electrically

connected only through the spiral slot. The operating

frequency is 50MHz band. To reduce the physical size of

this system, only one spiral slot is used, although popular

frequency selective surfaces (FSS) are composed of

periodic structure. The diameters of the helical and the

spiral slot are 120mm and 112mm, respectively. The

ground plane with the spiral slot is at the center between

the helical antennas separated by 40mm, where the stored

electromagnetic field of the wireless power transfer system

distributes. The resonant frequency of the spiral slot is

designed to be the same as the wireless power transfer

system by eigenmode analysis of finite element method.

The simulated resonant frequency was 59.75MHz.

To demonstrate the performance of the proposed system,

the analysis model in Fig. 1 was fabricated for

measurements. The photograph of the spiral slot on the

copper ground plane is shown in Fig. 2. The size of the

ground plane was 300mm square. The simulated and

measured transmission property is shown in Fig. 3. The

simulated resonant frequency was 59.5MHz which is

almost the same with the eigenmode analysis. The

measured transmission was −4.7dB (34%).

Ⅲ. ConclusionThe frequency selective spiral slot is designed by

eigenmode analysis independent on the wireless power

transfer system, therefore, the resonant frequencies of the

wireless power transfer systems and the spiral slot may be

perturbed. However, the transmission property was still

conserved even in the near field of the wireless power

transfer system.

Figure 1. Analysis model.

Figure 2. Frequency selective spiral slot. (25 windings)

Figure 3. Transmitting property.

16


PIFA antenna for UWB applicationso Do-Gu Kang and Jaehoon Choi

Department of Electronics and Computer Engineering, Hanyang University

[email protected] (corresponding author)

Ⅰ. IntroductionRecently, ultrawideband (UWB) communication

systems have received much attention because of supporting high data rate and low power consumption [1].Due to the limited area available for an antenna, UWB antenna should have compact size and low height [2]. Although the planar inverted-F antenna (PIFA) has compact size and low height, the PIFA is not suitable for UWB antenna due to its narrow bandwidth.

To overcome this problem, PIFA antenna with slotted ground plane for UWB applications is proposed. The proposed antenna has stable and near omnidirectional radiation characteristic.

Ⅱ. Antenna design and simulation resultFigure 1 shows the geometry of the proposed antenna.

The proposed antenna consists of a PIFA and ground plane with a slot. The PIFA is placed on the top of an FR4 substrate (•r = 4.4) with 1 mm thickness. The wideband impedance matching of the PIFA is realized by adding aslot on the ground plane. The ground plane is located on the bottom of the substrate and has a total size of 30 mm ×50 mm.

As shown in Figure 2, a -10 dB reflection coefficient bandwidth of the proposed antenna satisfies the full UWBfrequency range (3.1 GHz - 10.6 GHz).

Figure 3 illustrates the simulated radiation patterns. The proposed antenna has stable and near omnidirectional radiation patterns. Peak gains are 4.53 dBi, 3.55 dBi, 5.14 dBi, 6.08 dBi, 5.16 dBi at 3.4 GHz, 4.19 GHz, 4.88 GHz, 7.39 GHz, 10.2 GHz, respectively.

Ⅲ. ConclusionIn this paper, a PIFA antenna for UWB applications is

proposed. The proposed antenna has the wide -10 dB reflection coefficient bandwidth (3.08 GHz - 10.67 GHz)

Figure 1. Geometry of the proposed antenna.

Figure 2. Simulated reflection coefficient.

(a) (b)Figure 3. Simulated radiation patterns (a) xy-plane, (b) yz-plane.

satisfying the UWB frequency range by adding a slot on the ground plane. The antenna provides stable and near omnidirectional radiation for UWB applications.

References[1] G.-P. Gao, B. Hu, and J.-S. Zhang, “Design of a

miniaturization printed circular-slot UWB antenna by the half-cutting method,” IEEE Antennas and Wireless Propagation Letter, vol. 12, pp. 567-570,2013.

[2] A. Foudazi, H. R. Hassani, and S. M. A. Nezhad, “Small UWB planar monopole antenna with addedGPS/GSM/WLAN bands,” IEEE Transactions on Antennas and Propagation, vol. 60, no. 6, pp. 2987-2992, 2012.

AcknowledgementThis work (Grants No. C0331675) was supported by

Business for Cooperative R&D between Industry, Academy, and Research Institute funded Korea Small and Medium Business Administration in 2015.

17


Wideband Sequential-Rotation Arrays with Circularly Polarized Patch

Radiators using Anisotropic Metasurface Sarawuth Chaimool1, Prayoot Akkaraekthalin2, and Kwok L. Chung3

1Udon Thani Rajabhat University, Thailand 2King Mongkut’s University of Technology North Bangkok, Thailand

3Qingdao Technological University, China

[email protected]

I. Introduction Recently metasurface used for microwave antenna

applications becomes popular after supporting theories [1].

Novel types of metasurface for various applications with

significant improvements on microwave antennas were

reported [2-8]. The high-gain CP antenna was successfully

produced by using a grounded anisotropic metasurface in [2].

Simultaneous enhancement of gain and bandwidths of LP

and CP patch antennas were demonstrated in [3-5].

Polarization conversion using novel metasurfaces was

presented in [6-7]. Metasurfaces used for improving front-to-

back radiation ratio of slot/aperture antennas were illustrated

in [8-9]. More recently, performance enhancement and

sidelobes suppression of a conventional low-cost CP patch

array using a thin dielectric layer with metallic patterned cells

designated as the anisotropic metasurface (AMTS) have been

proposed in [10]. In this paper, we further study the sidelobe

levels suppression of the CP array by using various types of

thin AMTS.

II. Performance Enhancement of CP Array

A. Geometry of the original array and AMTS Figure 1 shows the geometry of the low-cost CP patch

array with the addition of the AMTS [10], where a layered structure was used for the wideband feed-network, 2-by-2 CP patches and the thin (0.007λo) metasurface. The metasurfaced CP array has an overall height of only about 0.073 λo.

B. Sidelobes Suppression In addition to the simultaneous enhancement of boresight

gain, gain bandwidth, axial-ratio and impedance bandwidths, suppression of sidelobe levels from both the co-polar (LHCP) and x-polar (RHCP) patterns were also achieved as evident in Figs. 2 and 3. The CP patches have an element spacing of 0.8 λo, the maximum sidelobe level (SLL) was recorded as high as -9.1 dB (average of max SLLs at two principal planes) at 2.45 GHz, as shown in Fig. 2. After mounting the AMST, the radiation patterns shown in Fig. 3 demonstrate an average reduction of SLLs by 4.4 dB. This is one of the reasons that explain the boresight gain enhancement of about an octave.

Figure 1. Geometry of CP patch array with AMTS.

Figure 2. Radiation patterns of CP array without AMTS.

Figure 3. Radiation patterns of CP array with AMTS-1.

III. Further Sidelobe Levels Suppression In this study, we aim to further suppress the sidelobe

levels of the CP array by simply reconfiguring the metasurface, rather than a complicate feed-network and or the element spacing. Two types of anisotropic metasurface have been investigated. Their performances are compared

-30dB

-20dB-10dB

0dB

90-90

180

= 0

xz-plane 2.45 GHz

LHCP RHCP

z

x

HPBW = 31.2Max SLL = -9.39dB at -62

yz-plane 2.45 GHz

LHCP RHCP

z

y -30dB

-20dB

-10dB0dB

90-90

180

= 0

HPBW = 30.8Max SLL = -8.76dB at 58

-30dB

-20dB-10dB

0dB

90-90

=180

0

xz-plane 2.45 GHz

LHCP RHCP

z

x

simulation measurement LHCPRHCP

HPBW = 28.6Max SLL = -13.4dBat -56

yz-plane 2.45 GHz

LHCP RHCP

z

y

simulation measurement LHCPRHCP

-30dB

-20dB

-10dB0dB

90-90

=180

0

HPBW = 29Max SLL = -13.6dBat -56

18


with AMTS-1, viz., the original one presented in [10]. Fig. 4 shows the geometries whereas Fig. 5 presents their SLL suppression performances. As can be seen, the inner 4 cells of each AMTS were modified. In AMTS-2, we replaced the diagonal strips by the 10-mm square rings whereas AMTS-3 has detached diagonal strips.

(a) (b)

Figure 4. Geometries of anisotropic metasurfaces: (a) AMTS-2, (b) AMTS-3.

(a) xz-plane

(b) D-plane

Figure 5. SLL performances of CP array after mounting different AMTS.

IV. Concluding Remarks New types of anisotropic metasurface for performance

enhancement of CP patch array are proposed in this paper. Further suppression of sidelobe levels without sacrificing the wideband and high-gain performance is achieved. The new metasurfaces can suppress up to 3 dB more in the principal planes but up to 10 dB in the diagonal plane. More results will be presented and discussed in the conference.

References [1] C. L. Holloway, E.F.Kuester, J.A.Gordon, J.O’Hara, J.Booth,

and D.R.Smith, “An overview of the theory and applications of metasurfaces: The two-dimensional equivalents of metamaterials,” IEEE Antennas Propag. Mag., vol.54, (2), pp.10–35, Apr.2012.

[2] G. Minatti, S. Maci, P. De Vita, A. Freni and M. Sabbadini, “A circularly-polarized isoflux antenna based on anisotropic metasurface,” IEEE Trans. Antennas Propag., vol.60, (11), pp.4998-5009, 2012.

[3] K. L. Chung and S. Chaimool, “Diamagnetic metasurfaces for performance enhancement of microstrip patch antenna,” 5th European Conference on Antenna and Propagation, pp. 55-60, EuCAP 2011, Apr 11-15, 2011, Rome, Italy.

[4] S. Chaimool, K. L. Chung, and Prayoot Akkaraekthalin, “Simultaneous gain and bandwidths enhancement of a single-feed circularly polarized patch antenna using a metamaterial reflective surface,” Progress In Electromagnetics Research B, Vol. 22, pp. 23-37, 2010.

[5] M. H. Ullah and M. T. Islam, “A new metasurface reflective structure for simultaneous enhancement of antenna bandwidth and gain,” Smart materials and Structures, Vol. 23, (8), 085015, 2014.

[6] H. L. Zhu, K. L. Chung, X. L. Sun, S. W. Cheung and T. I. Yuk, “CP Metasurfaced Antennas Excited by LP Sources,” IEEE Antennas and Propagat., Society Intern Symp 2012.

[7] H. L. Zhu, S. W. Cheung, K. L. Chung, and T. I. Yuk, “Linear-to-circular polarization conversion using metasurface,” IEEE Trans Antennas and Propagat., Vol. 62, (9), pp. 4615-4623, 2013.

[8] S. Sarawuth, C. Rakluea and P. Akkaraekthalin, “Mu-near-zero metasurface for microstrip-fed slot antennas,” Appl Phys A, vol. 112(3), pp.669-675, 2013.

[9] K. L. Chung and S. Kharkovsky, “Metasurface-loaded circularly-polarized slot antenna with high front-to-back ratio,” Electronics Letters, Vol. 49, no. 16, pp. 979-981, Aug. 2013.

[10] K. L. Chung, S. Chaimool and C. Zhang, “Wideband subwavelength-profile circularly-polarized array antenna using anisotropic metasurface,” Electronics Letters, Vol. 51, (18), pp. 1403-1405, Sep. 2015.

19


Electrically small spherical antennas using 3D printing technology Myeongjun Kong, Geonyeong Shin, and Ick-Jae Yoon

Dept. of Electrical Engineering, Chungnam National University, Daejeon, Korea

[email protected]

Ⅰ. Introduction For the past decades, there has been much interest on the

theoretical analysis of the radiation properties of electrically

small antennas and practical realization of them. The

theoretical bound is represented by the radiation quality

factor Q versus the electrical size of the antenna ka, where k

is the free space propagation constant and a is the radius of

the imaginary sphere enclosing the antenna. Among the

many practical antenna designs, a folded spherical helix

(FSH) dipole antenna is one of the well-known designs

approaching the low Q bound [1]. However, such the design

has a limit in miniaturization since the wires are physically

overlapped each other as the ka becomes smaller. Under this

background, this paper designs an FSH dipole antenna made

of thin and wide metal strips instead of thick wires, which

enables much smaller designs approaching the Q bound.

Ⅱ. FSH dipole antenna made of thin and

wide metal strip At high frequency, a thick wire can be electrically

replaced by a thin and wide metal strip with the same

circumference. It becomes practically available to design an

antenna with very small ka then when such the metal strip is

utilized for the excitation and folding arms in an FSH

structure.

To verify the proposed concept, we first design an FSH

antenna with ka=0.38 using a thick wire [1] and a thin and

wide metal strip with the same circumference, respectively.

It is found that the two antennas show about the same

radiation properties such as the Q value, input impedance

and radiation efficiency. Consequently, we design an FSH

antenna with ka=0.21 using the thin and wide metal strip. It

is worth to note that that the multiple folding arms should be

overlapped when the wires with the same circumference are

used. One may use thinner wire to avoid the overlapping,

but it will result in low radiation efficiency. The increased

number of arms with thinner wire for stepped-up radiation

efficiency will also cause physical overlapping. Thus, even

smaller FSH antenna can be designed without the physical

limitation using the proposed method of utilizing metal strip.

In Fig. 1, it is observed from the full wave EM simulation

that the designed antenna with ka=0.21 resonates at 301

MHz with radiation efficiency of 89.7%. It is also found that

it approaches the low Q bound. The proposed antenna is

fabricated using the 3-D printing technology and copper

coating process. The radiation characteristics together with

the measurement results will be reported at the conference.

(a)

(b)

Fig. 1. Simulated results of the designed FSH antenna

made of thin and wide metal strip (ka=0.21). (a) Reflection

coefficient. (b) Radiation efficiency.

Ⅲ. Conclusion This paper shows that even smaller FSH dipole antenna

approaching the low Q bound can be designed using the

proposed thin and wide metal strip structure which replaces

the thick wires in the original design [1]. The proposed

design can be fabricated using the commercialized 3-D

printing technology and copper coating process.

References [1] S. R. Best, “The radiation properties of electrically

small folded spherical helix antennas,” IEEE Trans.

Antennas Propag., vol. 52, no. 4, pp. 953-960, April,

2004.

20


Beamwidth Reconfigurable Array Antenna

Without Power Loss Using the Switched Couplero Soon-Soo OH, o Dong-Woo Kim, o Tae-Hyung Kim, and * Chi-Hyung Ahn

o Chosun University, * Korea Railroad Research Institute

[email protected]

Ⅰ. IntroductionThe RFID was proposed to be a good candidate for train

position detection where the beamwidth was controlled for

the optimum detection depending on the speed of train [1].

In this paper, the beamwidth reconfigurable array antenna

was proposed. Compare with the previous technique [1],

the beamwidth was controlled in three-way switched

coupler.

Ⅱ. Coupler and Antenna DesignThe proposed coupler is shown in Fig. 1. The input

port is Port 1 and the output port is Port2, Port3, and

Port 4. The Port 5 and Port 6 is isolated port. The

central frequency is 915 MHz. The dielectric

constant and height of the substrate is 4.5 and 1.6mm.

Fig. 1. Proposed coupler using the three-way output.

The four switches connected to the ground is place

on the vertical branch of the coupler as shown in Fig.

1. When they all are open, the transmission

coefficient was S21 = -4.69 dB, S31 = -5.28 dB, S41 = -

4.69 dB as shown in Fig.2. Meanwhile, when two

switches are open, the transmission coefficients are

S21 = -20.2 dB, S31 = -3.11 dB, S41 = -3.77 dB. When

all switch are closed, the transmission coefficients

are S21 = -17.4 dB, S31 = -0.34 dB, S41 = -17.4 dB.

Therefore, depending on the switch status, the power

could be delivered to three ports, two ports or only

one port. Compared with the previous technique of

cutting the branch, the proposed technique has the

almost zero loss.

Fig. 2. Reflection and transmission coefficients for

three-way coupler.

This coupler has been connected to the antenna

array with three radiating elements. The beamwidth

was controlled by switching the connection to the

ground, which will be presented in the conference.

Ⅲ. ConclusionIn this paper, the beamwidth controlled array antenna

was proposed with the three-way reconfigurable coupler.

The proposed array antenna could be used for the

application of the train position detection.

Acknowledgement

This research was supported by a grant from the Advanced

Technology Center R&D Program funded by the Ministry

of Trade, Industry & Energy of Korea (10048475).

References

[1] C.-H. Ahn, B.-K. Cho, S.-H. Ryu, S.-S. Oh, “Design

of beam-forming reader antenna for train position

detection using RFID,” Journal of the Korean

Society for Railway, vol. 18, no. 2, pp. 105-110, Apr.

2015.

21


Ⅰ. Introduction With the development of wireless communications,

antennas capable of tunable operation have been required

for various applications. A ferrite-loaded antenna is a good

candidate for this requirement. It is well-known that the

electromagnetic properties of the ferrites can be varied by

applying a static magnetic field. This feature can be

utilized to develop multifunctional antennas. In this talk,

mode splitting behavior of biased ferrite resonator antennas

will first be addressed [1]. Then, three tuning performances

(frequency, bandwidth, and polarization) of the antenna

will be discussed.

Ⅱ. Mode Splitting Behavior of Ferrite-

Loaded Resonator Antennas Theoretical models of cylindrical resonator

antennas with a static magnetic field along the z-axis are shown in Fig. 1.

It is known that for the analysis of cylindrical resonators, it

is possible to introduce a simplification which leads to

conventional eigenvalue equations [2]. Mode splitting

behavior of the cylindrical resonator antennas is verified by

mode matching technique. This technique will be

addressed. In addition, mode splitting behavior (see Figs. 2

and 3) enables three tuning performances: frequency,

bandwidth, and polarization. These features will be also

discussed.

III. Conclusion Theoretical models of ferrite-loaded resonator antennas

are found to be efficient to explain the HE11δ mode splitting

behavior. In addition, this unique feature provides various

functions: frequency and bandwidth tuning, and

polarization switching performances. This indicates that

the proposed antennas are suitable for multifunctional

antennas.

References [1] B.Y. Park, T.W. Kim and S.O. Park, “Analysis of

HE11δ mode splitting behavior of cylindrical ferrite

resonator antenna,” Korean Institute of Electro-

magnetic Engineering and Science Summer Sympo-

sium, Jeju, Korea, 2015.

[2] D. Kajfez and P. Guillon, Dielectric Resonators,

NOBLE, Tucker, Georgia, USA, 2007

Byeong-Yong Park, Tae-Wan Kim, and Seong-Ook Park*, Senior Member, IEEE School of Electrical Engineering, Korea Advanced Institute of Science and Technology, Daejeon, Korea

[email protected]

(a) (b)

Figure. 1. Theoretical models (a) ferrite resonator

antenna, (b) hybrid resonator antenna.

Figure. 2. Frequency response of cylindrical ferrite

resonator antenna for various DC magnetic bias.

Figure. 3. Frequency response of cylindrical hybrid

resonator antenna for various DC magnetic bias.

Analysis of Mode Splitting Behavior for Cylindrical Ferrite Resonator

Antenna

22

- 23 -


Plenary Talk and Invited Session

28th January, 2016Thursday

Plenary Speaker Biography

Toshikazu Hori received the B.E., M.E. and Dr. Eng. degrees in electrical engineering from Kanazawa University, Japan, in 1974, 1976 and 1993, respectively. In 1976, he joined the Electrical Communications Laboratories, Nippon Telegraph and Telephone Public Corporation (now, Nippon Telegraph and Telephone Corporation, NTT Corp.). Since then, he has been engaged in the research and development of antennas for satellite, cellular and microcellular mobile, and broadband wireless communication systems. In 2001, he moved to the University of Fukui, and is currently a Professor at the Graduate School of Engineering. His current research interests lie in the area of antennas and propagation for wireless broadband systems, especially, broadband antennas and meta-surfaces. Prof. Hori served as the Honorary Chair of AWAP2014 in Kanazawa, Japan. He also served as the Vice-Chair and the Chair of the IEEE AP-S Japan Chapter from 1999 to 2000, the Vice-Chair and the Chair of the IEEE AP-S Nagoya Chapter from 2007 to 2010, the Vice-Chair of IEEE Nagoya Section from 2013 to 2014, respectively. He was also the Guest Editor-in-Chief of four special issues in IEICE Trans., the Vice-Chair of ISAP2004 in Sendai, the TPC Chair of ISAP2007 in Niigata, the Vice-Chair and the Chair of the Technical Committee on Antennas and Propagation (TC-AP) of IEICE Japan from 2005 to 2009, and the Representative of IEICE Hokuriku Section, respectively. He is now the Advisory Committee Member of the TC-AP of IEICE Japan. He is a Fellow of the IEEE, and is also a Fellow of the IEICE Japan.


Low-Profile Design of Meta-Surface

with Frequency Selective Surface and Its Application Toshikazu Hori

University of Fukui, Japan

[email protected], [email protected]

Abstract An artificial magnetic conductor (AMC) and a high

impedance surface (HIS) have the perfect magnetic

conductor (PMC) characteristics at the specific frequency.

Meta-surfaces with an arbitrary reflection phase have been

studied extensively. And, these meta-surface technologies

have also applied to the antennas and propagation area.

The meta-surface composed of both frequency selective

surface (FSS) and the ground plane was proposed for an

easy configuration. By utilizing the spatial filter

characteristics of FSS well, a meta-surface with desired

reflection characteristics can be realized simply.

In this plenary talk, a low-profile design method using

both FSS and the ground plane is presented, and its

application to the antennas and propagation area are

introduced. When the spatial filter characteristics of FSS is

given, the relational equation between the reflection phase

θs and the thickness h of the meta-surface is derived by

using approximate optical ray theory. For an arbitrary

frequency f and thickness h, the meta-surface with the

desired reflection phase can be designed using this

relational equation. Here, the design method of a meta-

surface is made clear by using this relational equation, and

several design examples are introduced.

(1) Low-profile design of AMC

A loop slot type FSS which is a band-pass filter, a loop-

type FSS which is a band-rejection filter and a patch type

FSS with low-pass filter characteristics in the similar size

are considered as the FSS for an AMC with PMC

characteristics at the specific frequency. The thickness h

between FSS and the ground plane with PMC

characteristics at the arbitrary frequency f is derived from

the above-mentioned relational equation. As a result, the

low-profile design method with wider PMC relative

bandwidth is made clear. And, it is shown that the meta-

surface using loop-type FSS (as shown in Fig.1) is

excellent in PMC relative bandwidth, a low profile, and the

directive gain when employing as the reflector of a dipole

antenna.

(2) Design of meta-surface with the PMC characteristics

Here, a meta-surface with the frequency independent

PMC characteristics is considered while an AMC has PMC

characteristics at the specific frequency. In order to realize

a meta-surface with the frequency independent PMC

characteristics, the spatial filter characteristics of FSS is

derived from the above-mentioned relational equation. And,

the achieved possibility is verified.

(3) Design of meta-surface for polarization conversion

The principle of the meta-surface for polarization

conversion from linearly polarized wave to circularly

polarized wave is shown. The design method of the meta-

surface for polarization conversion using the patch type

FSS is also introduced. As a result, it is shown that the

meta-surface for polarization conversion has broadband

axial ratio characteristics. Next, the ideal filter

characteristics of FSS is derived from the above-mentioned

relational equation. And, the configuration of FSS suitable

to the meta-surface for polarization conversion is also

made clear.

Figure 1. Configuration of meta-surface with loop-type

FSS.

(Plenary Talk)

25


Surface moisture content sensor detecting mutual coupling magnitude

between parallel and perpendicular meander line dipole antennas

Chattapon RIENTHONG1, Chainarong KITTIYANPUNYA 2, and Monai KRAIRIKSH1**

1Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, 1 Chalongkrung Rd., Ladkrabang,

Bangkok10520, Thailand

E-mail: [email protected], [email protected]

**[email protected].

Abstract This research presents a surface moisture content sensor detecting mutual coupling magnitude between parallel and perpendicular

meander line dipole antennas. It operates at 245 MHz by detecting mutual coupling magnitude when dielectric properties of soil changes

corresponding to moisture content. The result can be used for creating a graph that can predict a moisture content of soil by using difference

of mutual coupling of parallel and perpendicular meander line dipole antennas. From this technique, an accurate result can be obtained and it

is essential to the effective irrigation system.

Keyword: Soil moisture content sensor, mutual coupling, parallel and perpendicular coupling, meander line dipole antennas.

1. Introduction

Di e l ec t r i c p ro p e r t i e s i s an i n d i c a to r t o

ch a r ac t e r i z e a ma t e r i a l . So i l mo i s tu re co n t en t i s

i mp o r t an t t o t h e y i e ld o f p l an t a t i o n an d i t i s

n ec e s sa r y t o kn o w a c cu r a t e l y . M an y r e se a rch h a v e

b een co n d u c t ed to d e v e l o p so i l mo i s tu re co n t en t

sen so r , e . g . [ 1 ] wh ich ad o p ted u l t r a - wid eb an d r ad ar .

Th e ad van ta g e an d d i sad v an t a ge o f ea ch t e ch n iq u e

i s c r i t i c a l l y r e v i e wed in [ 2 ] . Fo r a wid e a r e a , a

sen so r t h a t c an me a su re i n r ea l t i me i s d es i r ab l e . I t

mu s t p ro v id e f a s t r e su l t s an d d o n o t n e ed to p lu n g e

in to t h e so i l . Wh i l e u s in g o n l y o n e an t en n a i s

e f f e c t i ve [ 3 ] , i t n e ed s a c o mp l e x me a su r e men t . Th e

co s t e f f e c t i v e f r e e sp a c e t e ch n iq u e , i . e . u s in g

co u p l ed an t en n a s an d r e f l e c t i o n co e f f i c i en t [ 4 ]

wh i ch r eq u i r e s a d i r e c t i o n a l co u p l e r wa s u s ed fo r

d e t e r min in g d i e l ec t r i c p r o p er t i e s a t t h e su r f a c e o f a

ma t e r i a l . Th e wo r k in [ 5 ] wa s d e v e lo p ed fo r i n - s i t umo n i to r in g mo i s tu r e co n t en t o f p ad d y wi th o u t u s in g

a d i r e c t i o n a l co u p l e r b y me a su r in g co u p l ed s i gn a l s

f ro m p a r a l l e l an d p e rp e n d icu l a r d ip o l e an t en n a s .

Th e d i e l e c t r i c p ro p er t i e s o f p ad d y wa s d e t e r min e d

f ro m in t e r s ec t io n o f mu tu a l i mp ed an c e f ro m p a ra l l e l

an d p e rp en d i cu l a r d ip o l e a n t en n as o n a su r f a c e cu r v e .

Fo r t h e d i e l e c t r i c p ro p e r t i e s d e t e r min a t io n a t t h e

su r f a c e wi th o u t u s in g a d i r ec t i o n a l co u p l e r , t h e

co u p l ed s i gn a l s f ro m p ar a l l e l an d p e rp en d icu l a r

d ip o l e an t en n as c an b e ap p l i ed . Th i s can b e

a cco mp l i sh ed b y mo d i fy i n g e xp r es s io n s fo r mu tu a l

i mp ed an c e b e t we en th e d ip o le an t en n a s i n [ 5 ] . B y

p ro p e r l y mo d i fy in g th e e xp r es s io n s , t h e su r f a c e

cu r v e fo r d e t e r min a t io n o f 𝜀𝜀𝜀𝜀𝑟𝑟𝑟𝑟′ an d 𝜀𝜀𝜀𝜀𝑟𝑟𝑟𝑟" f ro m me asu r ed

co u p l ed s i gn a l f ro m p a ra l l e l an d p e rp en d i cu l a r

d ip o l e an t en n a s can b e u t i l i zed .

2. Principle of operation The work in [5] was proposed for plunging the sensor

in the material of interest. However, for convenience in

measuring large amount of data in wide area, this work

placed the sensor on the surface of the material of interest

(Invited paper)

26


which is soil moisture content. By using the data of soil

from [6], and modifying the calculation for half space (air

and soil) from full space (soil) of the mutual coupling from

[5], the mutual coupling can be calculated. Furthermore, it

was found that low mutual coupling was obtained at

microwave frequency. Hence, lower frequency like 245

MHz (citizen band) was used. In this regard, meander line

dipole antenna [7] was used instead of straight dipole

antenna to miniature the size of the sensor. Note that one

was for parallel polarization whereas the other one was for

perpendicular polarization. The width of the sensor is 30

cm.

3. Results An oscillator from a transmitter of 245 MHz was used

for transmitting signal via a transmitting antenna. The

receiving antennas consisting of parallel and perpendicular

polarized meander line dipole antennas which an RF switch

was used for selecting the polarization state of the

receiving antennas. A Schottky diode was used as a power

detector and the output D.C. signal was fed to a micro-

ammeter for displaying the mutual coupling. A DM-5 soil

tester was fixed in the soil for measuring moisture content

where moisture content can be increased by increasing

amount of water.

The mutual coupling from only parallel polarization,

which has level about ten times higher level than the

perpendicular counterpart may provide good correlation

with moisture content but comparing with the difference

from the parallel and perpendicular ones, the difference

provides better result. Hence, we selected to find moisture

content from the measure difference of both polarizations.

Although other polynomials can provide better

approximation to the measured results, the power

approximation has sufficiently accuracy and the moisture

content can be quickly estimated. This is important for real

time moisture content measurement in the wide area.

The performance of the sensor showing the predict-

measure relation from the experiment is much better than

the measured results using only one polarization. The

accuracy is 98.6%.

4. SummaryAccording to the requirement of a sensor for measuring

moisture content of soil in a wide area, a sensor that can

provide real time results is proposed. The sensor can

measure moisture content by placing it on the surface of

soil and the good result is obtained from difference of

mutual coupling from parallel and perpendicular polarized

dipole antennas. Since the frequency at VHF band of 245

MHz can penetrate deep in the soil and a low cost device

can be obtained from this frequency band, the sensor was

designed at this frequency. The sensor is miniaturized by

designing meander line dipole antennas. The measurement

results comparing to the soil tester is pretty good. The

result is from the fixed kind of soil and the different kind

of soil is in the further study. This sensor can be useful for

irrigation system.

References[1] A.E-C Tan, S.Richards, L.Sarrabezolles, I.Platt, and

I.Woodhead, “Proximal soil moisture sensing of dairy pasture,” IEEE APS/URSI 2015, Vancouver, 2015.

[2] S. Lekshmi, D.N. Sigh, and M.S. Baghini, “A critical review of soil moisture measurement,” Measurement , vol. 54, pp.92-105, 2014.

[3] J.L. Nicole, “The input impedance of horizontal antennas above an imperfect earth,” Radio Science, Vol.15, pp.471-477, 1980.

[4] J. Mearnchu, T. Limpiti, D. Torrungrueng, P. Akkaraekthalin, and M. Krairiksh, “A handheld moisture content sensor using coupled-dipole antennas,” Latin Amer. Appl. Res., vol.40, No.3, pp.199-206, 2010.

[5] T.Limpiti and M.Krairiksh, “In Situ moisture content monitoring sensor detecting mutual coupling magnitude between parallel and perpendicuar dipole antennas,” IEEE Trans. Instrumentation and Measurement, vol.61, no.8, pp.2230-2241, Aug. 2012.

[6] C.M.K. Gardner, T.J. Dean, and J.D. Cooper, “Soil

water content measurement with a high-frequency

capacitance sensor,” J. Agric. Eng. Res., Vol. 71 ,

pp.395-403, 1998.[7] H.Nakano, H.Tagami, A.Yoshizawa, and J.Yamauchi,

“Shortening of modified dipole antennas,” IEEE Trans. Antennas and Propagation, vol.AP-32, pp.385-386, Apr.1984.

27


Design of a Metamaterial Absorber for 24 GHz Automotive Radar System

Jinpil Tak, Eun Jeong, and Jaehoon Choi

Department of Electronics & Computer Engineering, Hanyang University, Seoul, Republic of Korea

[email protected] (corresponding author)

Ⅰ. IntroductionElectromagnetic (EM) absorbers are generally classified

into three types. The most common EM absorber is an

attenuation type which has pyramid-shaped array. It is

commonly used in an anechoic chamber. However, the

absorber is bulky, easily damaged, and expensive. Another

type is conductive loss or magnetic loss type [1]. However

it is quite expensive. The third type is resonance type. A

metamaterial (MTM) absorber is an electric resonant type

absorber, in general, which is also called frequency

selective surface (FSS) absorber. Many researchers pay

great attention to the dual band MTM absorber because it

inherently has narrow bandwidth [2].

In this paper, a bandwidth-enhanced double resonance

absorber is proposed. The proposed FSS absorber operates

at 24 GHz with dual absorption peaks for broad absorption

band.

Ⅱ. Geometry and Simulated resultsThe unit cell of MTM absorber consists of resonant

patch, ground, and FR4 substrate, and has a

pinwheel-like slot having different slot arms (long

arms and short arms) to achieve polarization

insensitivity and broad bandwidth characteristic as

shown in Figure 1(a). Figure 1(b) shows 90°

clockwise rotated unit cell. The sub-array is

composed of two unit cells and two rotated unit cells

with offset locations as shown in Figure 1(c).

Through simulation with an infinite array of sub-

array unit, the proposed absorber has two absorption

peaks, placed at 24.1GHz with 97.8% of absorptivity

and at 25.2 GHz with 74.5% of absorptivity in Figure

1(d). The proposed MTM absorber has 1.9 GHz of

full-width at half-maximum (FWHM).

Ⅲ. ConclusionIn vehicular environments, 24 GHz radar is used

for a core sensor of the safety system such as

collision avoidance or blind spot detection. The

proposed absorber can be used to false image signals

detected by an automotive radar.

(a) (b)

(c)

(d)

Figure 1. Geometry of the MTM absorber and simulated

results: (a) the unit cell, (b) 90° clockwise rotated unit cell,

(c) the sub-array unit, (d) simulated absorptivity and

reflectance characteristics (In infinite array).

References

[1] C.P. Nep, “Optimization of carbon fiber composite

for microwave absorber”, IEEE Transactions on

Electromagnetic Compatibility, vol. 46, no. 1,

pp.102-106, 2004.

[2] P. V. Tuong, J. W. Park, J. Y. Rhee, K. W. Kim, W.

H. Jang, H. Cheong, and Y. P. Lee, “Polarization-

insensitive and polarization-controlled dual-band

absorption in metamaterials,” Applied Physics Letters,

vol. 102, no. 8, 2013.

Acknowledgements

This research was supported by the Korea Ministry

of Land, Infrastructure, and Transport. It was also

supported by the Korea Agency for Infrastructure

Technology Advancement (Project No.: 15PTSI-

C054118-07)

28


Cavity-Backed Printed-Dipole Antenna for Millimeter-Wave Applications

Ikmo Park and Son Xuat Ta

Department of Electrical and Computer Engineering

Ajou University, Suwon, Republic of Korea

[email protected]

Ⅰ. IntroductionIn recent years, printed antennas have attracted much

interest for millimeter-wave applications because of low

cost, ease of fabrication, wide bandwidth, and high-

efficiency operation [1], [2]. Printed T-dipole antennas fed

by integrated balun have been widely developed for

wireless communications [3]. The T-dipole antennas can

achieve wideband or multiband operations, but they have a

relatively low high-gain.

In this paper, we present a planar printed-dipole antenna

for use in millimeter-wave applications. We utilized a

cavity to improve the radiation characteristics of the

printed dipole in terms of its gain and similar beamwidths

in the E- and H-planes. The electromagnetic simulator of

CST Microwave Studio was used for this work.

Ⅱ. Antenna GeometryThe geometry of the proposed antenna, which is

composed of a cavity and an angled dipole, is shown in Fig.

1. The angled dipole, which is the primary radiation

element of, was printed on an RT/Duroid 5880 substrate

with a dielectric constant of 2.2 and a thickness of 0.254

mm. The radiator consists of two identical 45° angled arms,

with one on the top side and the other on the bottom side of

the substrate. The antenna was designed to match a

microstrip line. The antenna was fed by a microstrip line,

which transits to a parallel-plate transmission line of the

angled dipole. The cavity is divided into the front and rear

parts for ease of installation. The substrate having an

angled dipole was clamped between the two parts of the

cavity. The antenna, with a fixed-cavity aperture of 0.5• ×

0.5• , was optimized in terms of good impedance matching

and high gain at 28 GHz using CST Microwave Studio.

Aluminum cavity

Substrate

Hole for nut

x y

z

Angled dipole with feedline

Figure 1. Geometry of the antenna.

Ⅲ. ConclusionA cavity-backed printed-dipole antenna has been

described for millimeter-wave applications. The use of the

cavity enhanced the radiation characteristics of the angled

dipole antenna including the gain, back radiation, and

beamwidths in the E- and H-planes.

References

[1] Y. Suh and K. Chang, “A new millimeter-wave

printed dipole phased array antenna using microstrip-

fed coplanar stripline tee junction,” IEEE Trans.

Antennas Propag., 52(8), pp. 2019–2026, Aug. 2008.

[2] R. Alhalabi and G. Rebeiz, “High-efficiency angled-

dipole antennas for millimeter-wave phased array

applications” IEEE Trans. Antennas Propag., 56(10),

pp. 3136–3142, Oct. 2008.

[3] B. Edward and D. Rees, “A broadband printed dipole

with integrated balun,” Microw. J., pp. 339–344, May

1987.

(Invited paper)

29


Waveguide Short-slot 2D-plane Coupler

for 2D Beam-switching Butler Matrix

Jiro Hirokawa and Dong-Hun Kim

Tokyo Institute of Technology

[email protected]

Ⅰ. IntroductionThe conventional short-slot one-plane coupler [1] is used

as hybrid or cross coupler in the one-dimensional beam-

switching Butler matrix [2]. This paper presents the

waveguide short-slot two-plane coupler as shown in Fig.1

for the two-dimensional beam-switching Butler matrix.

Ⅱ. StructureThe short slot two-plane coupler has 2x2 ports at each end

of the coupled region. The cross section shape of the coupled

region is changed from a rectangular. It has concaves at the

center of the top and the bottom sides and at the corners to

keep the symmetry in both the horizontal and the vertical

directions. The modes considered in the coupled region are

TE10, TE01, TE20, TM11, TE11, TM21, TE21 and TE30

modes of the corresponding multimode rectangular

waveguide. The cross section shape of the coupled region

should satisfy the following five conditions.

(1) TE10-like mode does not couple with the dominant

mode of the ports.

(2) TE21-like and TE30-like modes should be attenuated.

(3) The propagation constants of TE20-like, TM11-like

and TE11-like modes should be equal. Because TM11-like

and TE11-like modes can be dealt as one mode, it is called

as TM/TE11-like mode thereafter. (4) When the propagation constants of TE01-like, TE20-

like and TM21-like modes are 10β , 20β and 21β ,

respectively, 10 2120 2

β ββ

+= should be satisfied.

(5) TE10-like, TE20-like, TM/TE11-like and TM21

modes should have equal coupling with the dominant modes

of the ports.

The ideal operation of the hybrid is explained as follows.

For an incidence from Port 1 as an example, Ports 1-4 have

no outputs and Ports 5-8 have equal division in amplitude.

Ports 6 and 7 have 90-degree delay and Port 8 has 180-

degree delay in comparison with Port 5. The ideal operation

of the cross coupler is described as follows. For an incidence

from Port 1 as an example, only Port 8 have output and Ports

1-7 have no output in amplitude.The length l of the coupled region should satisfy

( )10 20 2 4

l πβ β− = for the hybrid and ( )10 20 2 2

l πβ β− =

for the cross coupler.

Ⅲ. ConclusionThe five conditions on the modes in the coupled region of

the short slot two-plane coupler has been shown.

The two-plane hybrid act as the 2x2-way two-

dimensional beam-switching Butler matrix. The measured

radiation patterns of the hybrid will be shown in the

workshop.

References

[1] H.J.Riblet, “The short-slot hybrid junction,” Proc. IRE,

vol. 40, no. 2, pp.180-184, Feb. 1952.

[2] J.Butler and R. Lowe, “Beam-forming matrix

simplifies design of electronically scanned antennas,”

Electron. Des., vol. 9, no. 8, pp. 170-173, Apr. 1961

Fig. 1. Waveguide Short-slot 2-plane Coupler.

(Invited paper)

30


Millimeter-wave Reflectarray Antennas with

Dual-reflector Configurations

Ji Hwan Yoon and Young Joong Yoon

Yonsei University

[email protected]

Ⅰ. IntroductionReflector antennas are useful for the applications where

highly directive radiation patterns are required. A most

common type reflector antenna to achieve pencil beam is a

parabolic antenna which consists of a reflector with

paraboloidal surface and a feeder at the focal point of the

paraboloid [1]. Since the feeder has to be located at the

face of the reflector surface, the antenna suffers many

disadvantages such as large antenna height, additional loss

from long transmission line connecting the source at the

back of the reflector and the feeder at the front of the

reflector. By adopting dual-reflector configurations, such

as Cassegrain and Gregorian antennas, these disadvantages

can be avoided [2].

The reflector antennas can be designed with more

compact size by using microstrip reflectarrays [3].

Reflectarrays are flat reflector that consists of a number of

reflective elements. Each element provides required

reflection phase in order to compensate the different path

delays which are originated from substituting curved

reflector surface into flat surface.

In [4], microstrip reflectarray antennas with on-axis

dual-reflector configurations have been proposed, and the

design process of the dual-reflectarray that mimics the

equivalent Cassegrain or Gregorian antenna was described.

In this communication, the detailed design process and

analysis of the microstrip reflectarrays with on-axis dual-

reflector configurations are reviewed, and the design

results at millimeter wave band are presented.

Ⅱ. Design ProcessIn case of Cassegrain/Gregorian antenna, the sub-

reflector is designed as hyperboloidal/ellipsoidal surface

with two foci. The feeder is located at one focus (F1) and

the main-reflector is designed so that its focus overlaps

with the other focus (F2) of the hyperboloid. For replacing

the curved reflectors with microstrip reflectarrays, the

required reflection phase at a point S on the sub-

reflectarray is given as follows.

( ) πnSFSFkψsh

221

±−= (1)

( ) πnSFSFkψse

221

±+= (2)

where ψsh and ψse are the reflection phases for

hyperboloidal and ellipsoidal sub-reflectors, respectively.

For main-reflectarray, the required reflection phase is at a

point M is

( ) πnMFkψm 22 ±= (3)

An intrinsic problem of the on-axis dual-reflector

configuration is the blockage from the sub-reflector. By

adopting offset feeding configuration, it is possible to

avoid the blockage but the volume of the antenna has to be

increased. As a future work, it is desirable to adopt axially

displaced configuration to avoid blockage from the sub-

reflector. The detailed design process will be presented at

the conference.

References

[1] P. J. B. Clarricoats and G. T. Poulton, “High

efficiency microwave reflector antennas-A review,”

Proc. IEEE, vol. 65, pp. 1470–1504, Oct. 1977.

[2] P. W. Hannan, “Microwave antennas derived from

the Cassegrain telescope,” IRE Trans. Antennas

Propag., vol. AP-9, pp. 140–153, Mar. 1961.

[3] J. Huang and J. A. Encinar, Reflectarray Antennas.

Hoboken, NJ: Wiley–IEEE, 2008.

[4] J. H. Yoon, Y. J. Yoon, W. Lee, and J. So, “Axially

symmetric dual-reflectarray antennas,” Electron.

Lett., vol. 50, no. 13, pp. 908–910, Jun. 2014..

(Invited paper)

31


Gain Improvement of Shaped-beam Reflector Using Simultaneous

Design of a Multimode Horn and Shaping Functions (Invited paper)oYoshio Inasawa, Takashi Tomura, Michio Takikawa, and Hiroaki Miyashita

Mitsubishi Electric Corporation

[email protected]

1. Introduction

A shaped beam reflector antenna can realize complex

coverage such as Japan area and is widely used for satellite

onboard antennas. The gain enhancement in shaped beam

antenna is desirable for the improvement of quality in the

service area. We investigate a design method to improve

the performance of shaped beam reflector and verify the

effectiveness of the design method.

2. Design Method

In general, the primary radiator and the main reflector

would be separately optimized in the design of shaped

beam reflector. We investigate a design method for the

shaped beam reflector by simultaneously optimizing the

primary radiator and shaping functions of main reflector.

The primary radiator is a multimode horn antenna defined

by a combination of waveguide modes. The mode

excitation ratio and the aperture diameter of the primary

radiator are optimized by PSO (Particle Swarm

Optimization). Shaping functions of the main reflector are

defined by the Fourier-Bessel series and the coefficients of

the shaping functions are optimized by CG (Conjugate

Gradient) method.

3. Verification

The simultaneous design method is implemented with

Japan coverage area. The frequency is in the Ku-band and

the initial surface of the main reflector is a paraboloid. For

the excitation modes of the primary radiator, three cases

are investigated: One mode (EH11), Two modes (TE11,

TM11) and Six modes (TE11, TM11, TE12, TM12, TE13,

TM13). The variables optimized by PSO are the horn

aperture diameter and the mode excitation ratio. The target

gain in the objective function is set to the same value for all

evaluation points. Here the ideal gain is 44.8 dBi for the

solid angle (1.35 deg^2) of the coverage.

The specifications of the optimized primary radiator and

the obtained minimum gain are shown in Table 1. The

result for a primary radiator of EH11 mode with typical

edge level of -20 dB is also shown for comparison. The

gain for all evaluation points is shown in Fig. 1. The gain

enhancement compared to the result of EH11 is also shown.

The EOC gain is highest in the case of 6 modes as the

primary radiator. Figure 1 shows the EOC gain is improved

at many evaluation points. It is confirmed that the

minimum EOC gain is improved by 0.3dB.

4. Conclusion

We have verified the effectiveness of the proposed

method for simultaneously optimizing the primary radiator

and the primary radiator.

Table 1. Design Results Primary radiator

EH11 EH11TE11,TM11

6 Modes

Parameters optimized by

PSO

HornDiameter

Horn Diameter

Mode Ratio

Mode Excitation

Ratio- - 1:0.49

1.00:0.61:0.40:0.42:0.38:-0.08

Horn Aperture Diameter(mm)

100 110.1 102.2 204.5

Edge Level(dB)

E-Plane -20 -28.2 -20.7 -29.4

H-Plane -20 -28.2 -23.5 -22.1Minimum Gain(dBi)

39.1 39.3 39.4 39.6

-0.4

-0.2

0

0.2

0.4

0.6

0.8

1

1.2

1.4

1.6

37

38

39

40

41

42

0 5 10 15 20 25 30 35 40 45 50 55 60 65

Dir

ecti

vity

enh

ance

men

t (dB

)

Dir

ecti

vity

(dB

i)

Evaluation point

EH11TE11+TM116-mode

Figure 1. Directivity and its Enhancement

(Invited paper)

32


Introduction to 5G Communications and its Smartphone Antenna

Design Perspectives (Invited Paper)Kin-Lu Wong

Department of Electrical Engineering, National Sun Yat-sen University, Kaohsiung, Taiwan

[email protected]

.Ⅰ IntroductionOver the past two decades, the smartphone antenna has

evolved from the external antenna before the year 2000 to

the internal antenna or the casing-integrated antenna for

2G/3G/4G communications till now. Then, what will be the

next for the evolution of the smartphone antenna? It is

expected that the Massive MIMO antenna will be

perspective for the B4G/5G terminal device antenna. In

this talk, the visions of 5G mobile communications will

first be addressed. Promising 4-antenna, 8-antenna, and 16-

antenna MIMO arrays in the smartphone [1] and their

achievable MIMO channel capacities will be discussed.

.Ⅱ Multiple MIMO Smartphone AntennasTypical Massive MIMO systems for multi-users with

multi-antennas thereof are shown in Fig. 1.

Figure 1. Massive MIMO systems for multi-users.

It is known that for the LTE MIMO operation, the

achievable channel capacity increases with an increasing

number of the transmiting and receiving antennas therein.

However, owing to limited space inside the smartphone, it

has been a challenge in disposing multiple MIMO antennas

therein. Some promising multiple MIMO antennas

including four LTE low-band (824~960 MHz) antennas

(see Fig. 2) and 8-antenna and 16-antenna arrays in the

3.4~3.6 GHz (see Fig. 3) based on the open-slot antenna [2]

are presented. Their MIMO performance is also discussed.

Figure 2. Four LTE low-band (824~960 MHz) antennas.

8-antenna array 16-antenna array

Figure 3. 8-antenna and 16-antenna MIMO arrays.

.Ⅲ ConclusionMultiple MIMO antennas embedded in the smartphone

have been shown to be promising to implement. Enhanced

MIMO channel capacities can be obtained when multiple

MIMO antennas are applied.

References [1] K. L. Wong, J. Y. Lu, L. Y. Chen, W. Y. Li, and Y. L.

Ban, “8-antenna and 16-antenna arrays using the quad-antenna linear array as a building block for the 3.5-GHz LTE MIMO operation in the smartphone,” Microwave Opt. Technol. Lett., vol. 58, pp. 174-181, Jan. 2016.

[2] K. L. Wong and C. Y. Huang, “Triple-wideband open-slot antenna for the LTE metal-framed tablet device,” IEEE Trans. Antennas Propag., vol. 63, pp. 5966-5971, Dec. 2015.

(Invited paper)

33


Low Profile PCB Integrated mmWave Array Antenna Solutions

for 5G Mobile Communication o Seungtae Ko, Youngju Lee, Kwanghyun Baek, Yoongun Kim and Wonbin Hong

Samsung Electronics, Mobile Communications Business

[email protected]

Ⅰ. Introduction Recently, 5G mobile communication service using

mmWave has been received attention worldwide for the

commercialization [1]. The phased array antenna has been

nominated as one of the candidates for reliable

communication service.

In this paper, several mmWave phased array antenna

solutions based on a low profile PCB substrate are

presented in order to obtain higher gain, polarization

robustness, and wide coverage.

Ⅱ. Guidelines Figure 1 shows Samsung’s antenna solution in a handset

for the 5G mobile communication. Two phased array antenna modules are inserted at corner of the mobile device and each one covers approximately 180 in order to remove the shadow region in the real time, as shown in Fig. 1(a). In addition, for stable performance under roll, pitch, and yaw motions between a transmitter (Tx) and a receiver (Rx), the interleaved array configuration is designed using a vertically polarized (VP) antenna and a horizontally polarized (HP) antenna, as shown in Fig. 1(b). The antenna elements are shown in Fig. 1(c) and a novel VP antenna having an extremely low profile is designed, especially. By

using proposed interleaved array, we can obtain the any polarization.

Since a small space will be allowed for the antenna module in the mobile device like Fig. 1(a), it is difficult to design the higher gain antenna. Currently, we expect that the limited number of antenna will be 8 or 16 elements. Because of this, the insufficient gain should be compensated from a base station for reliable communication. However, increasing the number of antenna in order to obtain higher gain is not a clear method, because feed line loss would be a critical problem. Samsung have researched the another solution to obtain highest gain as well as beam-steering property. Fig. 2 shows the proto-type planar lens antenna having an ultra-high gain for a base station. Although most lenses require large area, it will be suitable application in in the base station. Ⅲ. Conclusion In this paper, phased array antenna solutions are

presented for higher gain, polarization robustness, and wide coverage. Other valuable antenna designs and actual test results will be shown in presentation.

References [1] W. Hong, K. Baek, Y. Lee, Y. Kim, and S. Ko, “Study and

Prototyping of Practically Large-Scale mmWave Antenna

System for 5G Cellular Devices,” IEEE Communications

Magazine, vol. 52, no. 9, pp. 63-69, 2014.

Lens

Ant.

Fig. 2. Planar lens with 4X1 array antenna.

3mm

RFIC

(a) (b)

Horizontal Pol. Antenna Vertical Pol. Antenna (c)

Fig. 1. Phased array antenna module for 5G mobile communication, (a) antenna module in mobile device, (b) antenna module, (c) antenna elements.

(Invited paper)

34


Multi-beam massive MIMO using analog-digitalhybrid configuration

Kentaro NishimoriFaculty of Engineering, Niigata University

Ikarashi 2-nocho 8050, Nishi-ku Niigata-shi, 950-2181 JapanEmail : [email protected]

I. INTRODUCTION

Recently, the concept of massive MIMO has been proposed,because massive MIMO realizes simple signal processing inMulti-user MIMO (MU-MIMO) transmission [1]. However,when the Channel State Information (CSI) feedback is em-ployed from the user terminals (UTs) to an access point (AP),this procedure gives a very large overhead compared with thecommunication data.

To solve this problem, an implicit beamforming methodwhich eliminates the CSI feedback was proposed [2]. How-ever, even if implicit beamforming is applied for the massiveMIMO system, the CSI estimation itself is still large overheadwhen considering the short packet communications such asWireless LAN systems [2].

In this paper, we propose analog-digital hybrid configura-tion using analog multi-beams with dielectric line array andlens and blind algorithm called Constant Modulus Algorithm(CMA) [3] which does not need the CSI estimation. Via acomputer simulation, the effectiveness of proposed configura-tion is verified.

II. PROPOSED METHOD AND CONFIGURATION

Fig.1 shows the configuration by the proposed method.In the proposed method, M orthogonal multiple beams areprepared at analog part. Fig. 2 shows an example of multi-beam patterns. The received powers for all the users aremeasured at the output of multiple beams. Selected numberof beams is less than number of users (K). The user trackingis realized by the beam selection without CSI estimation.

Key question is how to realize the hardware of multi-beamforing network in the analog part. It is well known thatbutler matrix realizes multi-beam pattern [4]. Fig.3 shows anexample which realizes multiple beams in the analog part.Because this configuration consists of dielectric line array andlens, multi-beam forming with low loss is expected. Moreover,because reduction effect in the antenna and circuit size due todielectric is obtained, this circuit can be applicable for not onlyhigh frequency band but also micro frequency bands which areused in latest wireless communication systems. In addition,feeder circuit and antenna can be combined inside one circuit,because the production is possible by an injection molding.

The MU-MIMO transmission without CSI estimation foreach user is realized by the circuit in Fig.3. However, whendirection of arrivals (DoAs) are closed among users, same

#1 #2

CMA (Digital)

#M

Beam forming network (Analog)

Down.conv.

A/D

Down.conv.

A/D

Beam selector#K#1

UT#1UT#2

BS

Fig. 1. Proposed configuration.

-90 -60 -30 0 30 60 90-40

-35

-30

-25

-20

-15

-10

-5

0

Angle [deg.]

Rel

ativ

e po

wer

[dB

]

Fig. 2. Multi-beam patterns (16-beam).

beam must be utilized for each user. Moreover, the signalsby users except intended user are actually received at sidelobe and the interference cannot be realized by only multi-beamforing network. In order to realize the perfect interferencerejection, digital beam forming (DBF) based blind adaptivearray is introduced at the output of selected multi beams. Inthis paper, constant modulus algorithm (CMA) is used as theblind adaptive algorithm [3]. The CMA works only receivedsignals and does not need CSI. In addition, CMA reduce the

(Invited paper)

35


Dielectric rod

Dielectric lens

Dielectric lens

Dielectric rod

Fig. 3. Hardware configuration realizing multi-beam.

interference with environment where carrier and timing offsetexist. Hence, hybrid configuration with multi beams in theanalog part and blind algorithm in the digital part is suitablefor efficient transmission in massive MIMO system.

III. SINR CHARACTERISTICS BY PROPOSED METHOD

To verify the basic performance of proposed method, thecomputer simulation is carried out. When two user existin multi-path environment, the signal to interference plusnoise power ratio (SINR) is evaluated. In this simulation,single cluster model is assumed and angular spread at the BSassumed to be 10. The signal to noise power ratio (SNR) perantenna at the BS is 20 dB. The number of multi beams is 64.Least square method is adopted as the optimization algorithmregarding the CMA [3]. The propagation condition is changedand the number of trials is 10,000.

Cumulative Density Function (CDF) of SINR is plotted inFig. 4 when the difference of DoA (∆θ) is set to be 5◦ and30◦. Selected number of beams for each user is changed fromone to three. As can be seen in Fig. 4, the SINR is almost0 dB with CDF=1% when ∆θ = 5◦. On the other hand, theSINR with 2 beams is greater than 20 dB even in CDF=1%. Asshown in Fig. 4, although the SINR by only 1 beam withoutCMA is greater than 20 dB with CDF=1% when ∆θ = 30◦,the further improvement in the SINR is observed thanks to thecombination of CMA.

The SINR versus difference of DoAs between two users(∆θ) with CDF=10% is plotted in Fig. 5. Selected numberof beams for each user is changed from one to three. As canbe seen in Fig. 5, the SINR is greatly decreased when ∆θis less than 10◦. On the other hand, it is observed that theSINR is greater than 25 dB when the number of beams isgreater equal to two. Although the results are not plotted inthis figure, we confirmed that the SINR is not improved even ifthe number of beams is greater than three. From these results,it is shown that our multi-beam massive MIMO is effectivefor high transmission quality.

0 5 10 15 20 25 30 35 400

20

40

60

80

100

Output SINR [dB]

CD

F [%

]

1 beam

∆θ = 5 deg.

2 beams

∆θ = 30 deg.

K = 2, M = 64

Fig. 4. CDF of SINR (∆θ = 5, 30◦).

0

5

10

15

20

25

30

35

0 10 20 30 40 50 60

SIN

R [

dB]

Difference of DoA [deg.]

CDF = 10 %

1-beam2-beam3-beam

Fig. 5. SINR Characteristics versus DoA of interference.

IV. CONCLUSION

In this paper, we have proposed analog-digital hybridconfiguration using analog multi-beams with dielectric linearray and lens and blind algorithm called Constant ModulusAlgorithm (CMA) which does not need the CSI estimation.Via computer simulation, although the SINR is almost 0 dBwith CDF=1% when the difference of DoAs is 5◦, the SINRwith 2 beams is greater than 20 dB even in CDF=1%.

ACKNOWLEDGMENTS

The part of this work was supported by KAKENHI, Grant-in-Aid forScientific Research (C) 25420362.

REFERENCES

[1] F. Rusek, D. Persson, B. K. Lau, E. G. Larsson, T. L. Marzetta,O. Edfors, and F. Tufvesson, “Scaling Up MIMO –Opportunities andchallenges with very large MIMO–,” IEEE Signal Processing Magazine,pp.40–60, Jan. 2013.

[2] Hiraguri and K. Nishimori, ”Survey of transmission methods and ef-ficiency using MIMO technologies for wireless LAN systems (InvitedSurvey Paper),” IEICE Trans. Commun. Vol.E98-B, No.7, pp.1250-1267,July 2015.

[3] B. G. Agee, “The Least-Squares CMA: A New Technique for RapidCorrection of Constant Modulus Signals,” Proc.IEEE ICASSP, pp.953–956, 1986.

[4] S. Yamamoto, J. Hirokawa, and M. Ando, ”Single-Layer Hollow-Waveguide 8-Way Butler Matrix with Modified Phase Shifters, ” Proc.of ISAP2015, Sept. 2015.

36


Measurement of Antenna Substrate by Collimated THz Waves Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho, Hiroaki Nakabayashi, and Koji Suizu

Dept. of Electrical, Electronics and Computer Engineering, Chiba Institute of Technology

[email protected]

I. Introduction Recent explosive communication traffic growth has been

pushing the use of high frequency band for wireless communication system. Thus higher accuracy is required for antennas than those using in low frequency bands.

THz radio waves have received much attention in sensing various materials because THz waves are situated in between radio and light waves and there are many materials showing specific reflection and diffraction characteristics for the THz waves [1].

In this paper, in order to investigate detailed configuration of substrate which is popularly used for planar antennas, reflection characteristic of the substrate measured by collimated THz time-domain spectroscopy is reported.

Ⅱ. Measurement setup and results

Measurement setup Figure 1 shows a photo of the measurement setup. The

feature of the measurement system is using collimated THz waves instead of focused THz wave. This makes the measurement area and depth increase. Collimated THz pulse waves are illuminated to the planar substrate (Rogers RO4533) [2] and reflected waves are received at the receiver. The beam diameter of both TX and RX equipments is about 12mm. The thickness of the substrate under test is 1.6mm, and the dielectric constant of the substrate is 3.3@10GHz (Figure 2). Measurement results

Figure 3(a) shows the received pulse when metal plate is placed on the sample stage. Figure 3(b) shows the amplitude spectrum of the pulse. The maximum frequency of the pulse is about 2THz, and the amplitude is maximum at around 0.4THz. Figure 4 shows the reflection characteristics when the substrate is placed on the sample stage. Figure 4(b) is the case when metal plate is placed behind the substrate. The delay time of the pulse reflected at the other side of the substrate is found by placing the metal plate behind the substrate as shown in Fig. 4(b).

Fig. 1. Measurement setup. Fig. 2. Substrate under test.

Fig. 3 Received pulse reflected by metal plate

Fig. 4 Received pulse reflected by substrate Seven reflected pulses are observed thus the substrate should be composed of seven layers. Microscopic image will be presented at the conference.

Acknowledgement This work was supported by MEXT Supported Program

for the Strategic Research Foundation at Private Universities, Grand Number S1311004.

References [1] J. B. Jackson, et.al, IEEE Trans. Terahertz Science

and Technology, vol.1, no.1, pp.220-231, 2011. [2] http://www.rogerscorp.com/documents/888/acs/RO4

500-Series-Cost-Performance-Antenna-Grade-Laminates.pdf

Transmitter Receiver

Sample stage

-1.5

-1

-0.5

0

0.5

1

1.5

0 10 20 30 40 50Relative time [ps]

Ampl

itude

[V]

0

0.01

0.02

0.03

0.04

0.05

0 0.5 1 1.5 2Frequency [THz]

Ampl

itude

spe

ctru

m [V

rms]

(a) Received pulse (b) Amplitude spectrum

(a) w/o metal plate (b) with metal plate

-0.15

-0.1

-0.05

0

0.05

0.1

0.15


Ampl

itude

[V]

-0.8-0.6-0.4-0.2

00.20.40.60.8


Ampl

itude

[V]

Measurement of Antenna Substrate by Collimated THz Waves Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho, Hiroaki Nakabayashi, and Koji Suizu

Dept. of Electrical, Electronics and Computer Engineering, Chiba Institute of Technology

[email protected]

I. Introduction Recent explosive communication traffic growth has been

pushing the use of high frequency band for wireless communication system. Thus higher accuracy is required for antennas than those using in low frequency bands.

THz radio waves have received much attention in sensing various materials because THz waves are situated in between radio and light waves and there are many materials showing specific reflection and diffraction characteristics for the THz waves [1].

In this paper, in order to investigate detailed configuration of substrate which is popularly used for planar antennas, reflection characteristic of the substrate measured by collimated THz time-domain spectroscopy is reported.

Ⅱ. Measurement setup and results

Measurement setup Figure 1 shows a photo of the measurement setup. The

feature of the measurement system is using collimated THz waves instead of focused THz wave. This makes the measurement area and depth increase. Collimated THz pulse waves are illuminated to the planar substrate (Rogers RO4533) [2] and reflected waves are received at the receiver. The beam diameter of both TX and RX equipments is about 12mm. The thickness of the substrate under test is 1.6mm, and the dielectric constant of the substrate is 3.3@10GHz (Figure 2). Measurement results

Figure 3(a) shows the received pulse when metal plate is placed on the sample stage. Figure 3(b) shows the amplitude spectrum of the pulse. The maximum frequency of the pulse is about 2THz, and the amplitude is maximum at around 0.4THz. Figure 4 shows the reflection characteristics when the substrate is placed on the sample stage. Figure 4(b) is the case when metal plate is placed behind the substrate. The delay time of the pulse reflected at the other side of the substrate is found by placing the metal plate behind the substrate as shown in Fig. 4(b).

Fig. 1. Measurement setup. Fig. 2. Substrate under test.

Fig. 3 Received pulse reflected by metal plate

Fig. 4 Received pulse reflected by substrate Seven reflected pulses are observed thus the substrate should be composed of seven layers. Microscopic image will be presented at the conference.

Acknowledgement This work was supported by MEXT Supported Program

for the Strategic Research Foundation at Private Universities, Grand Number S1311004.

References [1] J. B. Jackson, et.al, IEEE Trans. Terahertz Science

and Technology, vol.1, no.1, pp.220-231, 2011. [2] http://www.rogerscorp.com/documents/888/acs/RO4

500-Series-Cost-Performance-Antenna-Grade-Laminates.pdf

Transmitter Receiver

Sample stage

-1.5

-1

-0.5

0

0.5

1

1.5


Ampl

itude

[V]

0

0.01

0.02

0.03

0.04

0.05

0 0.5 1 1.5 2Frequency [THz]

Ampl

itude

spe

ctru

m [V

rms]

(a) Received pulse (b) Amplitude spectrum

(a) w/o metal plate (b) with metal plate

-0.15

-0.1

-0.05

0

0.05

0.1

0.15


Ampl

itude

[V]

-0.8-0.6-0.4-0.2

00.20.40.60.8


Ampl

itude

[V]

(Invited paper)

37


High-gain Multiband Spiral Antenna Design History for NLJD System

(Invited Paper) Kyeong-Sik Min

Department of Radio Communication Engineering, Korea Maritime and Ocean University727, Taejong-Ro, Youngdo-Ku, Busan, 606-791, Korea

[email protected]

Ⅰ. Introduction

This paper presents high-gain multiband spiral antenna

design history for NLJD (Non-Linear Junction Detector)

system. Detection of a tiny chip made by the semi-

conductor has been made possible by the development of

NLJD system [1]. The high-gain circular polarization

antenna has been mainly used for the NLJD system

application required high gain and high resolution.

The author has been proposed a high-gain spiral antenna

with a novel Archimedean spiral slit on the ground plane to

achieve circular polarization and designed a new cavity

added to the conical wall to realize the high gain. To realize

a higher gain and a better resolution, a biconvex dielectric

lens has been also designed and measured.

Ⅱ . Design HistoryFig. 1 shows design history of multiband spiral antenna.

Antenna structure design has been continuously conducted

to increase gain and to obtain good axial ratio.

(a) First model

(b) Cavity model

(c) Optimized slit on ground plane model

(d) Conical model

(e) Biconvex lens model

Fig. 1 Transition of spiral antenna structure design.

Ⅲ . ConclusionHigh-gain multiband spiral antenna design history for

NLJD system is explained in this paper. Biconvex lens

antenna model of Fig. 1 (e) was shown very high gain,

sharp beam and good axial ratio.

ACKNOWLEDGEMENT

This research was supported by Basic Science Research Program through the National Research Foundation of Korea (NRF) funded by the Ministry of Education (No. 2013R1A1A2059944.

References[1] Jae-Hwan Jeong, Kyeong-Sik Min and In-Hwan Kim,

“Design for High Gain Spiral Antenna by Added Conical Cavity Wall”, The journal of Korean Institute of Electromagnetic Engineering and Science, vol.13, no.3, pp. 165-172, Sep. 2013.

(1) Biconvex Lens (2) Expanded polystyrene support

(3) Spiral Antenna

(4) Conical wall

(5) Metal cap (Reflector)

38


B3 B2 B1

B4B5

A3 A2 A1

Building of measurement

50m

Tx positons

Tx antenna height :A1 – A3: 11.5 mB1 – B5: 2.5 m

Fig. 1 Tx positions and building of measurement

28.6

m

18.4 m

Mea

sure

men

t rou

te 1

3 m

6.2 m

1.6 m

XY

Balcony

Courtyard

1.6 m

Mea

sure

men

t rou

te 2

Fig. 2 Layout of building (2-8F) and measurement routes

A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band T. Imai1, K. Kitao1, N. Tran1, N. Omaki1, Y. Okumura1, and K. Nishimori2

1 (NTT DOCOMO. INC.): 5G Laboratory, Yokosuka, Japan, [email protected] 2 (Niigata University): Faculty of Engineering, Niigata, Japan

[email protected]

1. IntroductionCurrently, the next generation (5G) mobile

communication system has been actively investigated all

over the world, in order to satisfy the future expected

requirements, such as super high bit rate. Here, one of the

approaches is to utilize the high-SHF band (over 6GHz)

and EHF band (mainly 30 – 100 GHz) [1, 2]. Up to now,

the path loss characteristics of the high-SHF and EHF

bands have been reported [2]. However, characteristics of

“Outdoor-to-Indoor (O2I) propagation” and its model have

not been clarified enough, even though we reported

measurement results in [3]. In this paper, penetration loss

property is clarified based on the measurement from 0.8 to

3.7 GHz in UMi (urban microcell) scenario, and they are

modeled.

2. MeasurementMeasurement was conducted in the campus of

Niigata University, Japan. Measurement environment is illustrated in Fig. 1 and 2. Frequencies used for measurement were 0.8, 2.2, 4.7, 8.4, 26.3 and 37.0 GHz. Six sleeve antennas for each frequency were installed on a car roof (at B1 – B6) or roof of building (at A1 –A3) as BS with transmission of CW signal simultaneously. The antenna heights were 2.5 or 11.5 m. Measurement was conducted with two receiver units of hand trucks. Each one installed three different sleeve antennas on it with the interval of 1.5 m between floor and the antennas. Measurement was repeated on the floors of 1st, 2nd, 4th, 6th and 8th (namely, 1F, 2F, 4F 6Fand 8F).

Path loss was calculated by post-processing the measured data of received power.

3. Extraction of penetration lossIn order to simplify the discussion, data of LOS

between Tx and building face are used in Analysis. The number of samples is 74,798.

In conventional model [4], O2I path loss in dB can be basically modeled by,

intwb PLPLPLPL ++= . (1)

Here, PLb is “basic path loss” which represents loss in outdoor scenario, PLtw is penetration loss into building, and PLin is loss inside. In this paper, PLb is assumed as free space loss;

( ) 4.32log20log20 33 +++= −− GHzinDoutDb fddPL , (2)

and PLin is assumed as

inDin dPL −= 25.0 . (3)

B3 B2 B1

B4B5

A3 A2 A1

Building of measurement

50m

Tx positons

Tx antenna height :A1 – A3: 11.5 mB1 – B5: 2.5 m

Fig. 1 Tx positions and building of measurement

28.6

m

18.4 m

Mea

sure

men

t rou

te 1

3 m

6.2 m

1.6 m

XY

Balcony

Courtyard

1.6 m

Mea

sure

men

t rou

te 2

Fig. 2 Layout of building (2-8F) and measurement routes

A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band T. Imai1, K. Kitao1, N. Tran1, N. Omaki1, Y. Okumura1, and K. Nishimori2

1 (NTT DOCOMO. INC.): 5G Laboratory, Yokosuka, Japan, [email protected] 2 (Niigata University): Faculty of Engineering, Niigata, Japan

[email protected]

1. IntroductionCurrently, the next generation (5G) mobile

communication system has been actively investigated all

over the world, in order to satisfy the future expected

requirements, such as super high bit rate. Here, one of the

approaches is to utilize the high-SHF band (over 6GHz)

and EHF band (mainly 30 – 100 GHz) [1, 2]. Up to now,

the path loss characteristics of the high-SHF and EHF

bands have been reported [2]. However, characteristics of

“Outdoor-to-Indoor (O2I) propagation” and its model have

not been clarified enough, even though we reported

measurement results in [3]. In this paper, penetration loss

property is clarified based on the measurement from 0.8 to

3.7 GHz in UMi (urban microcell) scenario, and they are

modeled.

2. MeasurementMeasurement was conducted in the campus of

Niigata University, Japan. Measurement environment is illustrated in Fig. 1 and 2. Frequencies used for measurement were 0.8, 2.2, 4.7, 8.4, 26.3 and 37.0 GHz. Six sleeve antennas for each frequency were installed on a car roof (at B1 – B6) or roof of building (at A1 –A3) as BS with transmission of CW signal simultaneously. The antenna heights were 2.5 or 11.5 m. Measurement was conducted with two receiver units of hand trucks. Each one installed three different sleeve antennas on it with the interval of 1.5 m between floor and the antennas. Measurement was repeated on the floors of 1st, 2nd, 4th, 6th and 8th (namely, 1F, 2F, 4F 6Fand 8F).

Path loss was calculated by post-processing the measured data of received power.

3. Extraction of penetration lossIn order to simplify the discussion, data of LOS

between Tx and building face are used in Analysis. The number of samples is 74,798.

In conventional model [4], O2I path loss in dB can be basically modeled by,

intwb PLPLPLPL ++= . (1)

Here, PLb is “basic path loss” which represents loss in outdoor scenario, PLtw is penetration loss into building, and PLin is loss inside. In this paper, PLb is assumed as free space loss;

( ) 4.32log20log20 33 +++= −− GHzinDoutDb fddPL , (2)

and PLin is assumed as

inDin dPL −= 25.0 . (3)

(Invited paper)

39


BS

d2D-out

hBS

UT

d2D-in

hUT

θin

Fig. 3 Definitions of parameters

-10

0

10

20

30

40

0 15 30 45 60 75 90

Incident angle, θin [deg.]

Pen

etra

tion

loss

, PL t

w[d

B]

Measurement

Proposed model

Fig. 4 Penetration loss property

-30

-20

-10

0

10

20

30

1 10 100

Frequency, f [GHz]

Res

idua

l erro

r[dB

]

( )xy log484.1219.1 +−=

Fig. 5 Frequency dependent of residual error

Equations (2) and (3) are same as that in 3GPP_3D channel model [4], fGHz is frequency in GHz and definitions of other parameters are shown in the figure 3. The normalized measured data by calculated PLb and PLin are regarded as PLtw in this paper.

4. Analysis resultsFigure 4 shows the property of penetration loss,

PLtw, with respect to incident angle, θin. Note that all frequency data are plotted in this figure. We find that PLtw increases when θin becomes large and there are upper limit and lower limit. This means that this characteristic can be modeled as sigmoidal function. The proposed model shown in Fig. 4 is a regression result based on sigmoidal function, and it is expressed by

{ }19.4)-0.453(-exp16.8-21.98.6

intwPL

θ++= . (4)

Here, the values of 21.9 and 6.8 represent upper limit and lower limit, respectively. RMS value of residual error is 6.8 dB. And correlation coefficient between the residual error and incident angle is lower than 0.001. This means that the penetration loss, PLtw, is randomized by our proposed model with regard to the incident angle, enough. Figure 5 shows the frequency dependent of residual error in above mentioned regression analysis. From this figure, we can understand that frequency dependent of PLtw is very little.

5. ConclusionIn order to design the next generation (5G) mobile

communication system, it is necessary to clarify the propagation characteristics in high-SHF band (over 6GHz) and EHF band (mainly 30 – 100 GHz). In this paper, we evaluated the penetration loss property and modeled it based on measurement results from 0.8 to 3.7 GHz in UMi (urban microcell) scenario.

References [1] NTT DOCOMO, INC. “DOCOMO 5G White Paper,

5G Radio Access: Requirements, Concept and

Technologies,” July, 2014.

[2] METIS, Deliverable D1.4, METIS Channel Models,

Feb. 2015. (https://www.metis2020.com/.)

[3] T. Imai, et.al. “Study on Extension to Higher

Frequency Band of 3GPP Outdoor-to-Indoor Path

Loss Model,” ISAP2015, Nov. 2015.

[4] 3GPP TR 36.873 (V1.2.0), “Study on 3D channel

model for LTE,” Sep. 2013.

40


Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-BandoJin-Seob Kang, Jeong-Hwan Kim, and Jeong-Il Park

Center for Electromagnetic Wave, Korea Research Institute of Standards and Science, Daejeon, Korea

[email protected]

Ⅰ. IntroductionSince 2009, the Antenna Measurement Club organized

by KRISS has performed an annual antenna measurement

comparison to support antenna manufacturers and end

users in Korea as a proficiency test program of the club [1].

In 2015, a comparison of power gains, radiation patterns,

and reflection coefficients was performed for three R-/S-

/X-band standard gain horn antennas in a joint effort

between KRISS and seven domestic participants from

private companies and public institutions.

Ⅱ. Description of the ComparisonThe traveling standards consist of three R-/S-/X-band

pyramidal standard gain horn antennas. The input port of

the traveling standards is connected to an output port (#1)

of a T-adapter, a short is connected to the other output port

(#2) of the three-port device, and the input port (#3) is used

as the input port of the horn antenna. The three-port device

with a short is selected to increase the reflection coefficient

of the antennas, which worsens the impedance match at the

input port.

In this comparison, antenna gain was measured by all the

participants using the gain comparison method, whereas

the extrapolation method was used at KRISS. Each of the

participants used a commercial vector network analyzer

and a conventional pattern measurement method in the far-

field region of the transmitting and receiving antennas.

Ⅲ. Measurement ResultsFig. 1 shows that swept-frequency power gains of most

of the participants, measured at the finite distances, are

roughly within ±0.4 dB from the far-field ones obtained by

KRISS at an infinite distance, while some of the

participants show gain fluctuations and deviations that

presumably are due to the lack of correct compensation for

the impedance mismatch at the receiving part of the gain

measurement system.

If a two-pole type antenna mast, which is suitable for

radiation pattern measurements of squinted beam antennas

such as base station antennas, is used for the radiation

pattern measurement of horn antennas, the poles can scatter

and block the incident wave radiated from the transmitting

antenna in the angular region far away from the bore-sight

direction of the receiving antenna. The result is co-

polarized E-plane radiation patterns that roughly agree with

each other in the angular region smaller than 45 degrees

from the bore-sight direction, as shown in Fig. 2.

Fig. 1. Swept-frequency power gains for R-band.

Fig. 2. E-plane radiation pattern at 2.6 GHz for S-band.

Ⅳ. ConclusionThe measurements show that the results for the power

gain, radiation pattern, and reflection coefficient of the R-

/S-/X-band antennas roughly agree with each other. Greater

accuracy is required for the impedance measurements of

the antennas and the measuring instruments for better

impedance mismatch correction at the connection between

the antennas and the measuring instruments, which will

provide better agreement in the gain results.

References

[1] J. Kang, J. Kim, and J. Park, “Comparison of

Antenna Parameters of R-/S-Band Standard Gain

Horn Antennas,” Journal of Electromagnetic

Engineering and Science, vol. 15, no. 4, pp. 231, Oct.

2015.

Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-BandoJin-Seob Kang, Jeong-Hwan Kim, and Jeong-Il Park

Center for Electromagnetic Wave, Korea Research Institute of Standards and Science, Daejeon, Korea

[email protected]

Ⅰ. IntroductionSince 2009, the Antenna Measurement Club organized

by KRISS has performed an annual antenna measurement

comparison to support antenna manufacturers and end

users in Korea as a proficiency test program of the club [1].

In 2015, a comparison of power gains, radiation patterns,

and reflection coefficients was performed for three R-/S-

/X-band standard gain horn antennas in a joint effort

between KRISS and seven domestic participants from

private companies and public institutions.

Ⅱ. Description of the ComparisonThe traveling standards consist of three R-/S-/X-band

pyramidal standard gain horn antennas. The input port of

the traveling standards is connected to an output port (#1)

of a T-adapter, a short is connected to the other output port

(#2) of the three-port device, and the input port (#3) is used

as the input port of the horn antenna. The three-port device

with a short is selected to increase the reflection coefficient

of the antennas, which worsens the impedance match at the

input port.

In this comparison, antenna gain was measured by all the

participants using the gain comparison method, whereas

the extrapolation method was used at KRISS. Each of the

participants used a commercial vector network analyzer

and a conventional pattern measurement method in the far-

field region of the transmitting and receiving antennas.

Ⅲ. Measurement ResultsFig. 1 shows that swept-frequency power gains of most

of the participants, measured at the finite distances, are

roughly within ±0.4 dB from the far-field ones obtained by

KRISS at an infinite distance, while some of the

participants show gain fluctuations and deviations that

presumably are due to the lack of correct compensation for

the impedance mismatch at the receiving part of the gain

measurement system.

If a two-pole type antenna mast, which is suitable for

radiation pattern measurements of squinted beam antennas

such as base station antennas, is used for the radiation

pattern measurement of horn antennas, the poles can scatter

and block the incident wave radiated from the transmitting

antenna in the angular region far away from the bore-sight

direction of the receiving antenna. The result is co-

polarized E-plane radiation patterns that roughly agree with

each other in the angular region smaller than 45 degrees

from the bore-sight direction, as shown in Fig. 2.

Fig. 1. Swept-frequency power gains for R-band.

Fig. 2. E-plane radiation pattern at 2.6 GHz for S-band.

Ⅳ. ConclusionThe measurements show that the results for the power

gain, radiation pattern, and reflection coefficient of the R-

/S-/X-band antennas roughly agree with each other. Greater

accuracy is required for the impedance measurements of

the antennas and the measuring instruments for better

impedance mismatch correction at the connection between

the antennas and the measuring instruments, which will

provide better agreement in the gain results.

References

[1] J. Kang, J. Kim, and J. Park, “Comparison of

Antenna Parameters of R-/S-Band Standard Gain

Horn Antennas,” Journal of Electromagnetic

Engineering and Science, vol. 15, no. 4, pp. 231, Oct.

2015.

(Invited paper)

41


°

,

ε

°

,

ε

°

,

ε

(Invited paper)

42


50° 65°

5 60°

φ

∞ ∞

∞∞

∞ ∞ τ

ε

τ

ε

β′β

φ′

β

β

φ

=

φ

43


Direct Derivation of Closed-form Expression of Sommerfeld Integral for

Impedance Half-plane from Exact Image Formulation

Il-Suek Koh

Department of Electronic Engineering at Inha University

[email protected]

.Ⅰ IntroductionThe original derivation of the closed-form expression is

complicated of the Sommerfeld integral for an impedance

half-plane. This paper proposes a simple derivation.

.Ⅱ DerivationThe radiated field of an infinitesimal dipole can be

efficiently calculated by using the exact image

representation [1] given by

(1)

where 2 20zk k k , 0k is the free space

wavenumber, 2 2x x y y , and

0J is the Bessel function of the first kind. p is 0k or 0 /k , is the normalized impedance and

22R i z z . By substituting

sinhi z z i u , the integral in (1) can be

converted into

(2)

where 2 , 0/ 2i k p r z z and 0/ 2i k p r z z . When approaches infinite, u should approach / 2 i .

The integral path, C can be chosen to satisfy

Re cosh 0i u as shown in Figure 1.

Figure 1. Original integral path. The original integral path can be deformed into the path in

Figure 2. In Figure 1 & 2, 11 sinh /u z z

and 2 20 0/ve k p k p . By the Jordan’s

lemma, the integral on 4C becomes zero. Therefore,

2 53C C C . The integral on 5C can be

exactly evaluated as

5

cosh (1)00 2

i u

Ce du iH

where (1)0H is the Hankel function of the first kind.

Figure 2. Deformed integral path.

The integral on 1C can be represented in terms of

incomplete cylindrical function of Poisson form [2] as

1

1

cosh0 10

,2

u v i u

Ce du iE i u v .

The derived closed-form expression is identical to that in

[1].

.Ⅲ ConclusionThe closed-form expression of the Sommerfeld integral for

an impedance half-plane is directly formulated from the

exact image representation. The procedure is simple.

References

[1] I. Koh and J. Yook, “Exact Closed-Form

Expression of a Sommerfeld Integral for the

Impedance Plane Problem,” IEEE Trans. Antennas

Propag., Vol. 54, No. 9, PP 2568-2576, Sept. 2006.

[2] M. M. Agrest, M. S. Maksimov, Theory of

Incomplete Cylindrical Functions and Their

Applications, Springer-Verlag, New York, 1971.

0

00 0

1z

ik Rik z z p

z z

k eJ k e dk i e dk p k R

0

cosh

0

ik Rip z zp i u

C

e e d ie e duR

Re u

Im u

1u

/ 2

C

Im 0p Im 0p

Direct Derivation of Closed-form Expression of Sommerfeld Integral for

Impedance Half-plane from Exact Image Formulation

Il-Suek Koh

Department of Electronic Engineering at Inha University

[email protected]

.Ⅰ IntroductionThe original derivation of the closed-form expression is

complicated of the Sommerfeld integral for an impedance

half-plane. This paper proposes a simple derivation.

.Ⅱ DerivationThe radiated field of an infinitesimal dipole can be

efficiently calculated by using the exact image

representation [1] given by

(1)

where 2 20zk k k , 0k is the free space

wavenumber, 2 2x x y y , and

0J is the Bessel function of the first kind. p is 0k or 0 /k , is the normalized impedance and

22R i z z . By substituting

sinhi z z i u , the integral in (1) can be

converted into

(2)

where 2 , 0/ 2i k p r z z and 0/ 2i k p r z z . When approaches infinite, u should approach / 2 i .

The integral path, C can be chosen to satisfy

Re cosh 0i u as shown in Figure 1.

Figure 1. Original integral path. The original integral path can be deformed into the path in

Figure 2. In Figure 1 & 2, 11 sinh /u z z

and 2 20 0/ve k p k p . By the Jordan’s

lemma, the integral on 4C becomes zero. Therefore,

2 53C C C . The integral on 5C can be

exactly evaluated as

5

cosh (1)00 2

i u

Ce du iH

where (1)0H is the Hankel function of the first kind.

Figure 2. Deformed integral path.

The integral on 1C can be represented in terms of

incomplete cylindrical function of Poisson form [2] as

1

1

cosh0 10

,2

u v i u

Ce du iE i u v .

The derived closed-form expression is identical to that in

[1].

.Ⅲ ConclusionThe closed-form expression of the Sommerfeld integral for

an impedance half-plane is directly formulated from the

exact image representation. The procedure is simple.

References

[1] I. Koh and J. Yook, “Exact Closed-Form

Expression of a Sommerfeld Integral for the

Impedance Plane Problem,” IEEE Trans. Antennas

Propag., Vol. 54, No. 9, PP 2568-2576, Sept. 2006.

[2] M. M. Agrest, M. S. Maksimov, Theory of

Incomplete Cylindrical Functions and Their

Applications, Springer-Verlag, New York, 1971.

0

00 0

1z

ik Rik z z p

z z

k eJ k e dk i e dk p k R

0

cosh

0

ik Rip z zp i u

C

e e d ie e duR

Re u

Im u

1u

/ 2

C

Im 0p Im 0p

(Invited paper)

44


Optical beam scanning antenna for ultra high speed short range communication system

Hiroyuki Arai Dept. of Electrical and Computer Engineering

Yokohama National University 79-5, Tokiwadai, Hodogaya-ku, Yokohama-shi, Japan

[email protected]

Abstract— This paper presents an optical beam scanning antenna for ultra-high speed short range communication system. The antenna consists of switched beam waveguide and leaky antenna for high gain beam scanning. The beam direction is changed by switching feed waveguide and the beam scanning is given by sweeping the wavelength from 1500 to 1600 nm. The communication range is also discussed by this high gain transmission antenna and low gain receiving antenna with photo diode.

I. INTRODUCTION For 5G and beyond wireless communication systems, data

transmission rate is expected to be more than 100GBps [1] or 1TBps. To achieve ultra-high speed wireless data transmission, frequency band is required more than a tens of THz, which makes us to explore optical short range communication systems with small spot coverage area. A massive MIMO concept is expected to enhance channel capacity in frequency bands higher than current cellular systems, however it is necessary to use high speed digital processing with a large number of antenna elements. High speed beam scanning, equivalent to the massive MIMO, is given by conventional frequency scanned array, which is an attractive antenna in optical frequency range. Fiber optics technologies have been developed well, then the next frequency band to explore for the future system is optical range. To overcome the difficulty of large propagation loss in this frequency range, narrow beam and high gain antennas are required to be developed for small cell base station antennas.

A combination of frequency sweep and phased array technique was proposed for optical beam scanning arrays [2], however it is expected to use for sensor applications with narrow scanning range. A thermal switched phase shifter for two-dimensional (2D) beam scanner was also proposed for display devices [3], which is not for the application in communication system. As another beam scanning array, this paper presents leaky waffled waveguides fabricated on silicon wafer excited by beam switching circuits to scan in two orthogonal planes. Its beam scanning is given by frequency sweep and beam switching circuit is given by Mach-Zehnder (MZ) type optical phase shifter [4]. The next section describes the proposed antenna geometry and also discusses commination range.

II. OPTICAL BEAM SCANNING ANTENNA 2D beam scanning antenna consists of a beam switching

circuit and a leaky waveguide. The optical signal is coupled to the leaky waveguide and its radiated beam direction is controlled by sweeping the wavelength. This two stage beam control

provides 2D beam scanning antenna. The beam switched waveguide is given by MZ phase shifter and silicon waveguide. The leaky waveguide is fabricated by silicon photonics. The waveguide layer with 210nm thickness is put on a SiO2 substrate and is covered by 2m thickness protection layer. To achieve high gain antenna, a waffled waveguide is used in this paper, because its antenna gain is 10dB higher than a grating waveguide with the same antenna length. The period of cavity array determines the beam tilt angle which is adjusted by wavelength sweep. Current tunable lasers have the potential of 100nm variable range in wavelength, which gives 10 beam scanning. To extend its range, 5 leaky waveguides with different tilt angles are fed through optical switches.

To demonstrate beam scanning of proposed waffled waveguide, a prototype antenna is fabricated by silicon photonics. Waveguide length is L=2000Λ and the number of cavity along y axis is N=20, where Λ is the period of the cavity. A tunable laser changes input optical wavelength from 1500 to 1600 nm and its radiation is detected through lens and multi-mode fiber terminated with optical power meter. Measured output beam patterns in different input wavelength, which shows 10 beam scanning is given by proposed waffled waveguide.

The link budget is calculated for short range systems, assuming the proposed leaky transmission antenna and a photo diode reception with sensitive size of 300m and dark current of 100pA [5]. The communication range of 8m is given by 50dBi antenna gain for 1mW input power and by 45dBi for 5mW, which is acceptable design parameters for short range communication systems.

ACKNOWLEDGEMENT A part of this work was supported by JSPS KAKENHI Grant

Number 25420359.

REFERENCES [1] J. Medbo, el. al., “Channel Modelling for the Fifth Generation Mobile

Communications”, EuCAP 2014. [2] P. F. McManamon, et al., ”A review of phased array steering for narrow-

band electro optical systems”, Proc. IEEE, vol.97, no. 6, pp.1078-1096, Jun. 2009.

[3] Karel Van Acoleyen, et al., “Off-chip beam steering with a one-dimensional optical phased array on silicon-on –insulator”, Opt. Lett., vol. 34, no. 9,pp.1477-1479,2009.

[4] Y. Morimoto and H. Arai, "Wavelength-Insensitive MZ type Switch for Dielectric Leaky Wave Antenna for Optical Transmission,” B-1-182, IEICE Spring Conf., Mar. 2014.

[5] www.kyosemi.co.jp/sensor/nir_photodiode/kpde030

Optical beam scanning antenna for ultra high speed short range communication system

Hiroyuki Arai Dept. of Electrical and Computer Engineering

Yokohama National University 79-5, Tokiwadai, Hodogaya-ku, Yokohama-shi, Japan

[email protected]

Abstract— This paper presents an optical beam scanning antenna for ultra-high speed short range communication system. The antenna consists of switched beam waveguide and leaky antenna for high gain beam scanning. The beam direction is changed by switching feed waveguide and the beam scanning is given by sweeping the wavelength from 1500 to 1600 nm. The communication range is also discussed by this high gain transmission antenna and low gain receiving antenna with photo diode.

I. INTRODUCTION For 5G and beyond wireless communication systems, data

transmission rate is expected to be more than 100GBps [1] or 1TBps. To achieve ultra-high speed wireless data transmission, frequency band is required more than a tens of THz, which makes us to explore optical short range communication systems with small spot coverage area. A massive MIMO concept is expected to enhance channel capacity in frequency bands higher than current cellular systems, however it is necessary to use high speed digital processing with a large number of antenna elements. High speed beam scanning, equivalent to the massive MIMO, is given by conventional frequency scanned array, which is an attractive antenna in optical frequency range. Fiber optics technologies have been developed well, then the next frequency band to explore for the future system is optical range. To overcome the difficulty of large propagation loss in this frequency range, narrow beam and high gain antennas are required to be developed for small cell base station antennas.

A combination of frequency sweep and phased array technique was proposed for optical beam scanning arrays [2], however it is expected to use for sensor applications with narrow scanning range. A thermal switched phase shifter for two-dimensional (2D) beam scanner was also proposed for display devices [3], which is not for the application in communication system. As another beam scanning array, this paper presents leaky waffled waveguides fabricated on silicon wafer excited by beam switching circuits to scan in two orthogonal planes. Its beam scanning is given by frequency sweep and beam switching circuit is given by Mach-Zehnder (MZ) type optical phase shifter [4]. The next section describes the proposed antenna geometry and also discusses commination range.

II. OPTICAL BEAM SCANNING ANTENNA 2D beam scanning antenna consists of a beam switching

circuit and a leaky waveguide. The optical signal is coupled to the leaky waveguide and its radiated beam direction is controlled by sweeping the wavelength. This two stage beam control

provides 2D beam scanning antenna. The beam switched waveguide is given by MZ phase shifter and silicon waveguide. The leaky waveguide is fabricated by silicon photonics. The waveguide layer with 210nm thickness is put on a SiO2 substrate and is covered by 2m thickness protection layer. To achieve high gain antenna, a waffled waveguide is used in this paper, because its antenna gain is 10dB higher than a grating waveguide with the same antenna length. The period of cavity array determines the beam tilt angle which is adjusted by wavelength sweep. Current tunable lasers have the potential of 100nm variable range in wavelength, which gives 10 beam scanning. To extend its range, 5 leaky waveguides with different tilt angles are fed through optical switches.

To demonstrate beam scanning of proposed waffled waveguide, a prototype antenna is fabricated by silicon photonics. Waveguide length is L=2000Λ and the number of cavity along y axis is N=20, where Λ is the period of the cavity. A tunable laser changes input optical wavelength from 1500 to 1600 nm and its radiation is detected through lens and multi-mode fiber terminated with optical power meter. Measured output beam patterns in different input wavelength, which shows 10 beam scanning is given by proposed waffled waveguide.

The link budget is calculated for short range systems, assuming the proposed leaky transmission antenna and a photo diode reception with sensitive size of 300m and dark current of 100pA [5]. The communication range of 8m is given by 50dBi antenna gain for 1mW input power and by 45dBi for 5mW, which is acceptable design parameters for short range communication systems.

ACKNOWLEDGEMENT A part of this work was supported by JSPS KAKENHI Grant

Number 25420359.

REFERENCES [1] J. Medbo, el. al., “Channel Modelling for the Fifth Generation Mobile

Communications”, EuCAP 2014. [2] P. F. McManamon, et al., ”A review of phased array steering for narrow-

band electro optical systems”, Proc. IEEE, vol.97, no. 6, pp.1078-1096, Jun. 2009.

[3] Karel Van Acoleyen, et al., “Off-chip beam steering with a one-dimensional optical phased array on silicon-on –insulator”, Opt. Lett., vol. 34, no. 9,pp.1477-1479,2009.

[4] Y. Morimoto and H. Arai, "Wavelength-Insensitive MZ type Switch for Dielectric Leaky Wave Antenna for Optical Transmission,” B-1-182, IEICE Spring Conf., Mar. 2014.

[5] www.kyosemi.co.jp/sensor/nir_photodiode/kpde030

(Invited paper)

45


IDC

Feed

An Electrically Small Isotropic Antenna Using Folded Split Ring Resonator

(Invited Paper)o Joon-Hong Kim, Sangwook Nam

Department of Electrical and Computer Engineering, INMC, Seoul National University, Seoul, Koreao [email protected], [email protected]

Ⅰ. IntroductionWith rapid growth of wireless communication

technologies, electrically small antennas have received

great attention due to their compact size in the systems.

Also isotropic radiation pattern which shows full spatial

coverage is very useful for some applications, such as radio

frequency identification (RFID) tags and wireless access

points (AP) due to stable link connection [1]. In this paper,

an electrically small isotropic antenna using three

dimensional (3-D) folded split ring resonators (FSRR) is

presented.

Ⅱ. Folded Split Ring Resonator AntennaThe designed FSRR antenna is shown in Fig. 1. To

miniaturize the electrical size of the antenna, its

structure is based on split ring resonators (SRR)

which show isotropic radiation characteristics.

Further miniaturization of the antenna is

implemented with interdigital capacitors (IDC) at the

end of its arms. However, the SRR antenna has poor

radiation characteristics. Thus the folded structure is

applied to the compact SRR structure to improve its

radiation characteristics [2].

Figure 1. Folded split ring resonator antenna

The simulated reflection coefficient and radiation

patterns of designed FSRR are presented in Fig. 2

and Fig. 3 respectively.

Figure 2. Simulated reflection coefficient of FSRR

Figure 3. Simulated radiation pattern of FSRR

Ⅲ. ConclusionThe compact isotropic antenna using SRR and folded

structure is presented. The electrical size of the designed

FSRR is ka = 0.54 with 1.7 % fractional bandwidth and

gain deviation is about 3.5 dBi.

Acknowledgements

This research was supported by a grant to Bio-Mimetic

Robot Research Center Funded by Defense Acquisition

Program Administration, and by Agency for Defense

Development (UD130070ID).

References

[1] Pan, Y.M, Leung, K.W,, Kai Lu “Compact Quasi-

Isotropic Dielectric Resonator Antenna With Small

Ground Plane," IEEE Trans. Antennas Propag., vol.

62, no. 2, Feb. 2014.

[2] R. E. Colin, Antennas and Radiowave Propagation,

MaGraw-Hill, 1985.

46


47

Centum Hotel (20,Centum 3-ro, U-dong, Haeundae-gu, Busan)

Welcome Reception2 (Rio Karaoke, 181-141 Millak-dong, Suyeong-gu, Busan)

Welcome Reception1 (Ahn-Chae, 26, Centum 3-ro, U-dong)

Banquet (Tiffany21 Cruise, 168 Marine city 1-ro, Haeundae, Busan)

MAP

Welcome Reception 1 will be held on 6:00PM 27th January, 2016, which is located at

HAEUNDAECENTUM

HOTELAhn-Chae, 2nd floor at CENTUM SQUARE Building

Ahn-Chae, 2nd floor at CENTUM SQUARE Building in the front of HAEUNDAE CENTUM HOTEL.


48

Banquet: Tiffany 21 Night Cruise Homepage: http://tiffany21.co.kr

Two Hours 7:00PM-9:00PM, Address 168 Marine City 1-ro, Haeundae-gu, BusanProf. Jaehoon Choi (Hanyang University, Korea)

Introducing Busan's Tiffany 21 Cruise Boat

The largest marine tourism city in Korea, Tiffany 21 Cruise is one of Busan's special marine attractions. Tiffany 21 blends

cruise excursions with a fine dining experience, offering a perfect venue for a variety of customer-tailored events.

Haeundae Beach has a beautiful 1.8 kilometer-long coastline. Passengers can take in the views of the numerous coastal

sites, including Nurimaru APEC House on Dongbaek Island, 49th Square in Suyoung-gu Namcheon-dong, and Centum

City in U-dong. With Gwangandaegyo Bridge, the nation's largest marine bridge, spanning Haeundae Beach, Tiffany 21

will give you an unforgettable cruise experience.

Surrounded by a wall of rocks, Igidae has some gorgeous scenery, with dozens of magnificent cliffs jutting out of the open

sea around it. Designated a Natural Monument No. 22, Oryukdo Island adds to Busan's coastal scenery. Also, Taejongdae

Park has huge waves crashing against the rocky shore, and has a beautiful, calm forest as well.

When you pass over Busandaegyo Bridge, which connects the mainland with Yeongdo island, you will be greeted with other

famous touristic attractions, such as Busan International Film Festival (PIFF) Square and Jagalchi Fish Market. Jagalchi

Fish Market is the largest of fish market in Korea, known for the loud noise made by middle-aged men and women

haggling for the best bargain of fresh sea food.

Banquet: We will take a Bus on 6:00PM 28th January, 2016 in the hotel lobby of HAEUNDAE CENTUM HOTEL, and

leaving at 6:20PM, go to board on 7:00PM Tiffany 21 Night Cruise, Two Hours 7:00PM-9:00PM. We will take a Bus at

9:00PM and arriving at HAEUNDAE CENTUM HOTEL around 9:10PM.

Things to enjoy on Tiffany 21

The cruise has established itself as a popular tourist attraction in Busan, with its breathtaking views of the Busan Sea.

Providing a variety of cruise experiences and a fine seafood buffet, as well as beverages, both alcoholic and non-

alcoholic. Watch the sunrise, moon, fireworks, and other various events, including family functions, weddings and

birthday parties while drifting along the beautiful coastal waters of Busan. It's especially great for groups like families

or domestic and foreign tourists, as it offers a truly unique experience of Busan.

Banquet


49

Transportation

Transportation from Gimhae International Airport to Hotel

Distance from Gimhae International Airport to Haeundae Centum Hotel: 26.87 km

Taxi

Takes approximately 45 minutes.

Fare: 20,000 - 30,000 KRW

Airport Limousine

Airport Limousine buses to BEXCO and Haeundae New Town departs every 20 minutes.

Get off at Haeundae Centum Hotel bus stop after approximately 50 minutes.

Fare : Adults 7,000 KRW / Children 4,500 KRW

First and last buses depart at 06:45 and 22:00, respectively.

The schedule is subject to change and dependent upon traffic conditions.

Bus

City bus 307

Gimhae International Airport → BEXCO

Fare: 1,200KRW / Approx. 1 hour 28 min.

Subway

Airport station (Busan-Gimhae line) → Sasang Station (Line 2) → Centum City Station (Exit No.3)

Fare: 2,000 KRW/ Approx. 55 min


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MEMO


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MEMO


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MEMO


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MEMO

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