che et al 2014 tie

164 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 1, JANUARY 2014

Operation of a Six-Phase Induction Machine UsingSeries-Connected Machine-Side Converters

Hang Seng Che, Emil Levi, Fellow, IEEE, Martin Jones, Mario J. Duran,Wooi-Ping Hew, Member, IEEE, and Nasrudin Abd. Rahim, Senior Member, IEEE

AbstractThis paper discusses the operation of a multiphasesystem, which is aimed at both variable-speed drive and generat-ing (e.g., wind energy) applications, using back-to-back converterstructure with dual three-phase machine-side converters. In thestudied topology, an asymmetrical six-phase induction machine iscontrolled using two three-phase two-level voltage source convert-ers connected in series to form a cascaded dc link. The suggestedconfiguration is analyzed, and a method for dc-link midpointvoltage balancing is developed. Voltage balancing is based onthe use of additional degrees of freedom that exist in multiphasemachines and represents entirely new utilization of these degrees.The validity of the topology and its control is verified by simulationand experimental results on a laboratory-scale prototype, thusproving that it is possible to achieve satisfactory dc-link voltagecontrol under various operating scenarios.

Index TermsAC machines, current control, generators, motordrives, multiphase machines, voltage control, wind energy.

I. INTRODUCTION

THE STANDARD solution for medium-voltage high-power variable-speed drives is nowadays based on theback-to-back converter topology. The converters are typicallythree-level, and a three-phase machine is used [1], [2]. Thesame configuration is also applicable in conjunction with windenergy generators, which is based on fully rated converters.Compared with conventional three-phase machines, multiphasemachines are credited with having lower torque ripple, bet-ter fault tolerance, and lower per-phase power rating require-ment [3]. These desirable features make multiphase machinesa promising candidate for high-power and/or high-reliabilityapplications. The majority of the literature deals with multi-phase motor drives. However, studies of multiphase generators,

Manuscript received August 24, 2012; revised November 7, 2012 andJanuary 18, 2013; accepted January 23, 2013. Date of publication February 22,2013; date of current version July 18, 2013.

H. S. Che is with the University of Malaya Power Energy DedicatedAdvanced Centre, University of Malaya, 59990 Kuala Lumpur, Malaysia, andalso with the School of Engineering, Technology and Maritime Operations,Liverpool John Moores University, Liverpool, L3 3AF, U.K. (e-mail: [email protected]).

E. Levi and M. Jones are with the School of Engineering, Technology andMaritime Operations, Liverpool John Moores University, Liverpool, L3 3AF,U.K. (e-mail: [email protected]; [email protected]).

M. J. Duran is with the Department of Electrical Engineering, University ofMalaga, 29071 Malaga, Spain (e-mail: [email protected]).

W.-P. Hew and N. A. Rahim are with the University of Malaya Power EnergyDedicated Advanced Centre, University of Malaya, 59990 Kuala Lumpur,Malaysia (e-mail: [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2013.2248338

Fig. 1. Six-phase generation system with (a) parallel machine-side converters,(b) series machine-side converters with dc-link midpoint connection, and(c) series machine-side converters without dc-link midpoint connection.

including five-phase [4], six-phase [5][10], nine-phase [11],twelve-phase [12][14], and eighteen-phase [15] systems havebeen also reported. Most of these works have been relatedto wind energy systems because the increasing power rating(currently up to 10 MW) and reliability requirements (partic-ularly in offshore farms) match the advantageous features ofmultiphase machinery.

Multiphase variable-speed drive systems with back-to-backconverter configuration typically utilize a multiphase two-levelor three-level voltage source converter (VSC) at the machineside [3]. Multiphase generator studies have concentrated on theuse of standard n-phase VSCs or diode rectifiers, depending onthe machine type, with n converter legs connected in parallelto the dc link. Such a topology, where two parallel three-phase VSCs drive a six-phase wind energy system, is shownin Fig. 1(a). This is, at the same time, the typical configurationof an asymmetrical six-phase machine when used in a variable-speed drive system.

In [5], a unique converter topology was introduced for anasymmetrical six-phase generator (a permanent magnet syn-chronous machine was considered). The topology uses two

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CHE et al.: OPERATION OF A SIX-PHASE INDUCTION MACHINE USING MACHINE-SIDE CONVERTERS 165

three-phase machine-side converters, which are connected inseries to form a cascaded dc link. A three-level neutral pointclamped (NPC) converter was used as the grid-side converterin a back-to-back manner, with a connection to the dc-linkmidpoint, as shown in Fig. 1(b). In this configuration, themachine-side converters can provide additional control to thedc-link midpoint voltage balancing and improve the transientperformance [16]. A similar concept, using a twelve-phasepermanent magnet machine and Vienna rectifiers, is presentedin [14]. The topology in Fig. 1(b) is equally applicable to thevariable-speed drives as well and can be used in conjunctionwith both permanent magnet synchronous and induction ma-chines. The series-connected machine-side topology means thatthe individual dc-link voltages [Vdc1 and Vdc2 in Fig. 1(b)] ofthe six-phase system need only be equal to 50% of the totalrequired dc voltage, achieving a higher total dc-link voltage(Vdc1 + Vdc2) for the same voltage rating of the converters.The dv/dt of the common-mode voltage (CMV), which isknown to be a main cause of leakage currents in high-powerapplications, is therefore halved as well. The total number ofthe semiconductor switches of the machine-side converter fora six-phase machine with two-level VSCs is equal to what isrequired for a three-level three-phase VSC. The elevated dc-linkvoltage in this series-connected topology reduces the currentrating and the cable size for the given power, hence giving apotential overall capital cost reduction.

Although the voltage of the converters is halved in theseries connection, it should be noted that the voltage stresswithin the machine can still reach a value up to the total dc-link voltage Vdc. Nevertheless, since not the whole machineis under this maximum voltage stress, the overall insulationrequirement is expected to be lower than that of a higher voltagemachine designed with phase voltage rating based on the fullVdc voltage. As a matter of fact, part of the voltage stress canbe reduced via a proper choice of the pulsewidth modulation(PWM) technique, as discussed later in Section III.

Although the topology in [5] is potentially interesting, thethree-wire connection shown in Fig. 1(b) can be uneconomicalif the dc link is long and the grid-side converter is locatedfar apart. While this may or may not be relevant for variable-speed drive applications, it would be relevant in offshore windfarms with a dc offshore grid and high-voltage dc (HVDC)transmission to the onshore grid-side converter [17]. Offshorewind farms are currently promoted by some country policies[18] and manufacturers (e.g., RePower [19]) due to better windresources and absence of the visual impact. Newly designedoffshore wind farms require higher powers and better reliability,which makes them suitable for the utilization of multiphasegenerators. If the wind farm distance to the shore is above acertain break-even distance (typically around 70 km [19]), theuse of HVDC transmission becomes more favorable, and thethree-wire topology in Fig. 1(b) is not adequate. For this reason,the topology in [5] has been modified in [20] to eliminate theneed for the dc-link midpoint connection to the grid side, asshown in Fig. 1(c). This provides a favorable arrangement forremote offshore wind generation with dc offshore grid.

Since the grid-side converter can no longer control the volt-age of the dc-link midpoint, the voltage drifting becomes a

problem for this topology unless the dc-link voltages can becontrolled from the machine side. Fortunately, the additionaldegrees of freedom of the six-phase machine allow the volt-age balancing using an additional controller [20]. This paperextends the initial discussion of the suggested topology andits control in [20] [see Fig. 1(c)] and provides simulation andexperimental verification of the voltage balancing controllerperformance. It should be noted that the additional degrees offreedom of a multiphase machine have been used in the past forvarious specific aims (e.g., torque enhancement with low-orderstator current harmonic injection, development of fault-tolerantcontrol algorithms for postfault operation, and independentcontrol of a multitude of series-connected multiphase machinesusing a single VSC [3]). However, there is no evidence that thecapacitor voltage balancing has ever been attempted before byutilizing these additional degrees of freedom.

The concept of cascading converters to achieve elevated dc-link voltage is of course not new, and it has been reported in sev-eral works [8], [14], [21][26]. However, such solutions usuallyrequire specially designed machines or customized converters.When diode-based rectifiers are used, the applicability is re-stricted to synchronous generators. The topology discussed hereuses standard three-phase VSCs, which provide advantages interms of economy and technology maturity. Moreover, the useof VSCs allows the topology to be used also with inductionmachines, in both motoring and generating modes.

This paper is organized as follows: Section II presents anoverall description of the system, including the topology, theinduction machine model, and the control structure. Section IIIanalyzes the merits and demerits of the topology and the dc-linkvoltage drifting issue. Detailed derivation of the dc-link voltagecontroller is given in Section IV. The theoretical development issupported using simulation and experimental results, which areprovided in Section V. Finally, concluding remarks are given inSection VI.

II. SYSTEM DESCRIPTION

A. System Topology

In this paper, the machine is an asymmetrical six-phase(two three-phase windings spatially shifted by 30) squirrel-cage induction machine with two isolated neutral points. It isdriven by two three-phase two-level VSCs, which are connectedin series to form a cascaded dc link. Since the focus of thediscussion is on the machine-side converters, a three-phase two-level VSC is considered as the grid-side converter in this paper,for the sake of simplicity. However, it should be emphasizedthat the grid-side converter is not restricted to the two-levelVSC and, as already noted, more advanced converters, such asmultilevel NPC converters, can be also used.

B. Induction Machine Model

Using the vector space decomposition (VSD) technique,the machine model can be decoupled into three orthogonalsubspaces, which are denoted as , xy, and zero-sequencesubspaces. For machines with distributed windings, only


components contribute to the useful electromechanical energyconversion, whereas xy and zero-sequence components onlyproduce losses. These xy current components represent theadditional degrees of freedom, previously referred to. A powerinvariant decoupling transformation is used to convert the phasevariables of the stator (a1, b1, c1, a2, b2, and c2) and rotorwindings into and xy variables [3], i.e.,

[T ] =13

xy

1 12 12

32

32 0

032

32

12

12 1

1 12 12 32

32 0

0 32

32

12

12 1

. (1)

Transformation (1) is the Clarkes matrix for an asymmetricalsix-phase system. Zero-sequence components are omitted fromthe consideration [and are therefore not included in (1)] sincethe machine has two isolated neutral points. Two pairs ofreal variables that result after application of (1) onto phasequantities ( and xy) can be combined into correspondingspace vectors [3].

A rotational transformation is applied next to transform the variables into a synchronously rotating reference frame(dq), which is suitable for vector control [3], i.e.,

[D] =

dqxy

cos s sin s sin s cos s

11

. (2)

The transformation of dq variables in (2) is identical as in thecase of a three-phase system [3]. The second pair of variables(xy) is not rotationally transformed since the equations forthese variables do not contain stator-to-rotor coupling (xyquantities do not contribute to the electromagnetic torque andhence electromechanical energy conversion). In (2), s is theangle of the rotational transformation for stator. Assumingthat the electrical angular speed of the machine is r andthe reference frame is rotating at an arbitrary speed (sothat s =

dt), the model of the induction machine can be

described using the following voltage and flux equations in thedq plane (indices s and r indicate stator and rotor quantities,respectively; motoring convention for the positive stator currentflow is used):

vds =Rsids + dds/dt qsvqs =Rsiqs + dqs/dt+ ds

0 =Rridr + ddr/dt ( r)qr0 =Rriqr + dqr/dt+ ( r)dr (3)

ds =(Lls + Lm)ids + Lmidr

qs =(Lls + Lm)iqs + Lmiqr

dr =(Llr + Lm)idr + Lmids

qr =(Llr + Lm)iqr + Lmiqs (4)

where Rs and Rr are the stator and rotor resistances, respec-tively; whereas Lls, Llr, and Lm are the stator and rotor leak-

age inductances and the magnetizing inductance, respectively.Additional stator equations, which describe the machine in thexy plane, are

vxs =Rsixs + dxs/dt

vys =Rsiys + dys/dt (5)

xs =Llsixs

ys =Llsiys. (6)

For a machine with p pole pairs, the electromagnetic torquesolely depends on the dq components and is given with

Te = pLm[idriqs idsiqr]. (7)Finally, the equation of rotor motion is

Te Tm = J dmdt

(8)

where m is the rotor mechanical speed, J is the inertia, andTm is the mechanical (prime mover or load) torque.

C. Control Structure

Based on the VSD model, the six-phase induction machineis controlled using indirect rotor flux oriented control (IRFOC).Rotor flux angle, which is required for the rotational transfor-mation (2), is calculated on the basis of the standard indirectorientation principles, using slip frequency sl, i.e.,

s =

(pm +

sl) dt (9)

sl =1

Tr

iqsids

. (10)

Due to the additional degrees of freedom, instead of usingjust two proportionalintegral (PI) controllers for the dq cur-rent control, two extra current controllers are required for thexy current control. The schematic of the machine-side currentcontrol is shown in Fig. 2. There are various control schemesthat can be used for the xy current control in multiphasemachines [27], [28]. Here, an antisynchronous reference frameis used for the xy PI current controllers since it allows aneasy implementation of the dc-link voltage balancing control.Superscript is added to the rotational transformation matrix[D] and xy currents in Fig. 2 to differentiate them from thosebased on (2). Detailed explanation of the dc-link voltage balanc-ing controller using PI current controllers in an antisynchronousreference frame is postponed for Section IV. The d-axis currentreference (ids) is set at a constant value to provide rated rotorflux, and the q-axis current reference (iqs) is provided by eithera maximum power point tracking (MPPT) controller, whichis based on the optimal torque control method [19], whenoperating as a generator or by a PI speed controller whenoperating in motoring mode.

Since the dc link practically decouples the grid-side converterfrom the machine-side converters and the total dc-link voltage is


Fig. 2. General structure of the machine-side controllers.

controlled by the grid-side converter, the grid-side converter canbe conveniently represented by a controllable current sourcein the simulation (and a constant voltage supply in the exper-iment). The current source (voltage supply) only operates tomaintain the overall dc-link voltage at a constant level and willtherefore not be discussed in detail here.

III. ANALYSIS OF THE SERIES-CONVERTER TOPOLOGY

A. Merits and Demerits of the TopologyThe topology in Fig. 1(c) combines an asymmetrical six-

phase induction machine with cascaded machine-side con-verters. The series connection of the converters elevates thedc-link voltage, thus reducing the current rating and losses forthe given dc-link voltage level. In generating mode, the highergenerating voltage also eases the step-up voltage process to thetransmission voltage level.

The cascaded dc links also allow supplying each set of three-phase windings with halved dc-link voltage [see Fig. 1(c)]. Thislowers the dc-link voltage (Vdc1 = Vdc1 = Vdc/2) and allowsthe use of switching devices with lower voltage ratings. Thereduced dc-link voltage also halves the dv/dt of the CMVfor each winding, which now has steps of Vdc1/3 = Vdc/6,compared with the value of Vdc/3 in three-phase machines, forthe given Vdc value.

As with other transformerless converter topologies [8], [23],one major challenge for this topology is the machines insu-lation requirement. In this topology, the upper and lower dc-link voltages (referenced to the dc-link midpoint) are Vdc/2and 0 for VSC1 and 0 and Vdc/2 for VSC2. Hence, whilethe voltage within each set of three-phase windings is restrictedto Vdc/2, the voltage stress between the two sets of windingscan reach Vdc.

Nevertheless, not all parts of the machine are subjected to thismaximum voltage stress. In particular, the voltage differencebetween neutral points of the two sets of windings variesdepending on the switching state, as summarized in Table I.Switching states are expressed in decimal form of the six-digit

TABLE INEUTRAL-TO-NEUTRAL VOLTAGE FOR DIFFERENT SWITCHING STATES

binary number Sa1 Sb1 Sc1 Sa2 Sb2 Sc2, where Si denotesthe switching condition of converter leg for phase i, with 0indicating lower switch turned on and 1 indicating upper switchturned on. Redundant states are given in bold.

As shown in Table I, the neutral-to-neutral voltage variesfrom 0 to as high as Vdc, depending on the switching state.By selecting a suitable PWM method, the neutral-to-neutralvoltage can be limited to lower values. PWM methods that useswitching state 56 (111000), as does the 24-sector space vectorPWM (SVPWM) [29], should be avoided, as they exert largevoltage stress Vdc on the windings. PWM using double zero-sequence injection [30] is used throughout this paper, in bothsimulations and experiments, since this method does not applyswitching state 56 (111000). As a matter of fact, only switchingstates that produce 0.333, 0.5, and 0.667 Vdc are used in thisPWM method, as shown later in the experimental results. Thepresence of redundant states suggests the possibility of furtherreducing the voltage stress using special SVPWM techniques,which restrict selection of certain switching states [31]. Thisis, however, beyond the scope of this paper and not discussedfurther.

Since not every part of the machine winding requires theinsulation to withstand maximum voltage stress of Vdc, it ispossible to relax the insulation requirements by proper ma-chine design. For instance, slot sharing between conductorsfrom different sets of windings should be avoided. Hence, asnoted, although the insulation requirement is high, it can becomparatively less than in a similar machine designed withphase voltage rating based on full Vdc voltage.

B. DC-Link Voltage DriftIn the considered topology, the dc-link midpoint at the ma-

chine side is not accessible by the grid-side converter any more,whereas the total dc-link voltage control is performed by thegrid-side converter. There is no guarantee whatsoever that thedc-link voltages [Vdc1 and Vdc2 in Fig. 1(c)] will always bebalanced. It is thus necessary to control the dc-link voltages at


Fig. 3. Simplified circuit diagram for a generation system with series-connected dc links.

the machine side using the machine-side converters, to avoiddc-link voltage drift. In order to analyze the dc-link voltagebalancing issue, the system is simplified by representing eachVSC as a controlled current source, as shown in Fig. 3.

By using Kirchhoffs current law for points W and Z, themachine-side converters currents can be written as

Idc1 = Idc3 Icap1 Idc2 = Idc3 Icap2. (11)

The currents Idc1 and Idc2 consist of two components, namely,a common component (Idc3) and a differential component(Icap1 and Icap2). At any time instant, the common currentcomponent will be drawn from both machine-side converters,whereas the instantaneous difference between the converterscurrents and the common current will be supplemented by eachof the converters capacitors. Ideally, the two sets of machinewindings are identical so the average converter current shouldbe the same despite the spatial difference. The average capacitorcurrents should thus be also the same.

The dc-link voltage balancing depends on the active powerbalancing between the two converters. The equations for theactive power of the machine-side converters are

P1 = Idc1Vdc1 P2 = Idc2Vdc2. (12)

During steady state, the average converters currents will beequal. Hence,

P1P2

=Vdc1Vdc2

. (13)

If the grid-side converter provides perfect control, the totaldc-link voltage will be maintained at a constant value of Vdc.Each individual dc-link voltage is expressed as a sum of its idealbalanced value (Vdc/2) and a deviation from the ideal value(Vdc1 and Vdc2). Since the sum of two dc-link voltagesis equal to Vdc, the voltage deviations must be equal but ofopposite signs, i.e., Vdc1 = Vdc2. Hence, the power (13)can be written as

P1P2

=(Vdc/2) + Vdc1(Vdc/2)Vdc1 . (14)

Rearranging (14), the voltage deviation can be expressed as afunction of the active powers, i.e.,

Vdc1 =

(P1 P2P1 + P2

)Vdc2

. (15)

From (15), it follows that any asymmetry in the system willcause dc-link voltage drifting unless some voltage balancingmechanism is included in the control scheme.

IV. DC-LINK VOLTAGE BALANCING CONTROL

A. DC-Link Voltage Balancing Using XY CurrentsBased on the analysis in Section III, it is obvious that

additional control needs to be provided by the machine-sideconverters to ensure that the dc-link voltages are alwaysbalanced.

This can be achieved using the additional degrees of freedomprovided by the xy currents, so that the torque and fluxproduction of the machine remains unaffected. Since the dc-link voltage unbalance is found to be a result of the activepower imbalance, it is necessary to first identify the relationbetween the xy currents and the active power differencebetween the two windings. With the VSD model, the quantitiesin the two pairs of windings are not related, whereas the powerof the two three-phase windings now need to be separatelycontrolled. In order to achieve separate power control with theVSD model, it is insightful to, at first, establish the relationshipbetween the dqxy components in the VSD model and thed1q1d2q2 components in the dual dq model [20], whichenables separate formulation of winding powers.

To start with, according to the stationary transformation ofthe dual dq model, the components of the two windingsare separately treated as 11 and 22 currents, which canbe given with

is1 =

2

3

(ia1 1

2ib1 1

2ic1

)

is2 =

2

3

(3

2ia2

3

2ib2

)

is1 =

2

3

(3

2ib1

3

2ic1

)

is2 =

2

3

(1

2ia2 +

1

2ib2 ic2

). (16)

Comparison of (16) with (1) shows that the following holdstrue:

is =

1

2(is1 + is2) is =

1

2(is1 + is2)

ixs =

1

2(is1 is2) iys =

1

2(is1 + is2). (17)

For control purposes, it is more useful to have the controlvariables in the dq synchronous reference frame, so that theyappear as dc quantities and can hence be easily dealt with


using PI controllers. For the dual dq model, currents in thesynchronously rotating frame are given as

ids1 = is1 cos s + is1 sin s

ids2 = is2 cos s + is2 sin s

iqs1 = is1 sin s + is1 cos siqs2 = is2 sin s + is2 cos s. (18)

For the VSD model, using the conventional rotational transfor-mation defined in (2), the following is obtained:

ids =1/2(ids1 + ids2)

iqs =1/2(iqs1 + iqs2)

ixs =1/2 [(ids1 ids2) cos s (iqs1 iqs2) sin s]

iys =1/2 [(ids1 + ids2) sin s + (iqs1 + iqs2) cos s] .

(19)

As can be seen from (19), the resulting xy components arenot dc quantities. Hence, an alternative transformation matrix isintroduced, i.e.,

[D] =

dqx

y

cos s sin s sin s cos s

cos s sin ssin s cos s

(20)

which rotates the xy components in the inverse (anti) syn-chronous direction. With this alternative rotational transforma-tion, a more suitable form of xy components (denoted byxy components) can be obtained, i.e.,

ids =1/2(ids1 + ids2)

iqs =1/2(iqs1 + iqs2)

ixs =1/2(ids1 ids2) =

1/2ids

iys =1/2(iqs2 iqs1) =

1/2iqs. (21)

Transformed xy components are now both dc signals andrepresent the difference between the dq components of thetwo windings. Controlling ixs to be positive will make ids1greater than ids2, whereas positive iys makes iqs1 smaller thaniqs2, and vice versa. Thus, the power drawn from the twowindings can be controlled by the proper injection of ixsiyscurrents. Moreover, since dq components are dc quantities,xy components will be also dc quantities, which allowsthe use of simple PI controllers. It is also worth noting that,from (21), injecting ixsiys changes the difference betweenids1iqs1 and ids2iqs2 but does not change the overall fluxand torque currents idsiqs. Hence, the injection of ixsiyscurrents will not affect the overall operation of the machine.

During generation, active power is injected into the dc link,and the torque current is negative (assuming positive rotationaldirection). When Vdc1 is positive, iqs should be positive toreduce the power injected by VSC1, and vice versa. Hence, adc-link voltage balancing controller can be constructed using a

Fig. 4. Structure of the (dotted box on the right) xy current controllersand the (dotted box on the left) dc-link voltage balancing controller.

PI controller with Vdc1 as input. Since iys and iqs haveopposite polarity, the output of the PI controller should beinverted (multiplied by 1), as shown in Fig. 4.

B. Voltage Balancing in Motoring Mode

The derivation of the dc-link voltage balancing controllerhas been based on the machine operation in the generatingmode. The applicability of the same controller structure in themotoring mode is therefore addressed here.

During motoring operation in positive rotational direction,the machine consumes active power from the dc link, and thetorque current is positive. When Vdc1 is positive, positiveiqs should be imposed so that VSC1 consumes more activepower than VSC2, thus reducing Vdc1. Since iqs again hasthe same polarity as Vdc1, the same controller structure canbe applied during motoring mode as well.

When the machine rotates in the negative direction in motor-ing mode, the sign of the torque current becomes negative. Inthis case, positive Vdc1 requires a negative iqs for voltagebalancing. For proper operation of the dc-link voltage balancingcontroller, the negative sign in Fig. 4 will have to be replaced bya positive sign. This consideration does not apply in generationsince the rotation of a generator is usually confined to a singledirection.

V. SIMULATION AND EXPERIMENTAL RESULTS

In order to verify the dc-link voltage balancing controlleroperation, experimental tests are conducted on a low-powerasymmetrical six-phase induction machine, which was obtainedby rewinding a 1.1-kW three-phase machine. The six-phasemachine is configured with two isolated neutral points. Eachof the two three-phase windings is connected to a custom-made multiphase two-level VSC, which is configured for three-phase operation. The dc links of the two VSCs are cascadedin series and connected to a dc power supply (Sorensen SGI600-25), which maintains the overall dc-link voltage at 300 V.A 7.5-kW dc machine is coupled to the six-phase machine andis controlled using ABB DCS800 in torque mode to provideloading of the six-phase machine.

The whole control algorithm for the six-phase machine isimplemented using a dSPACE DS1006 system. A switchingfrequency of 5 kHz and a current sampling frequency of 10 kHzare used. A dead time of 6 s is integrated in the VSChardware. Machine phase currents and dc-link voltages are


Fig. 5. Experimental setup for the series-converter topology in motoringmode.

TABLE IIEXPERIMENT AND SIMULATION PARAMETERS

measured (using the LEM sensors embedded in the VSCs)through a dSPACE DS2004 ADC module and displayed on anoscilloscope via the dSPACE DS2101 DAC module. Hence, thecurrent traces shown later are the filtered current waveforms,without the switching ripples. Overall, the configuration of theexperimental setup is shown in Fig. 5, and the detailed list ofmachine and control parameters is given in Table II.

Due to the lack of means for grid connection, experimentalresults apply only to operation in the motoring mode. How-ever, as maintained from the beginning and as discussed inSection IV-B, the capacitor voltage balancing controller is ofthe same structure in both motoring and generating applica-tions. All the parameters used for the simulation are identical tothose in the experimental setup (except for the machines inertiaand machines inherent asymmetries, which were obtained viatrial and error for simulation purposes).

In what follows, the experimental results are shown first,together with MATLAB/Simulink simulation results, for themotoring mode of operation. The study is then complementedwith simulation results for the generating mode of operation.

A. Experimental Verification in the Motoring ModeThe machine is controlled in motoring mode, using IRFOC

with closed-loop speed control. Only dq current controllersare initially used, without any xy current control (xy voltage

Fig. 6. Experimental results for no-load operation at 500 r/min, with only dqcurrent controllers. Channel 1: Vdc1 (100 V/div); Channel 2: Vdc2 (100 V/div);Channel 3: phase a1 current (1 A/div); Channel 4: phase a2 current (1 A/div);Horizontal: time (20 ms/div). (Note that markers for Channels 1 and 3 havebeen overlapped by markers for Channels 2 and 4, respectively.)

Fig. 7. Simulation results for no-load operation at 500 r/min, with only dqcurrent controllers.

references are set to zero). Fig. 6 shows the experimental resultswhen the machine operates at 500 r/min without load.

The dc-link voltages are not equal due to the inherent asym-metries that exist in the converters and the machine. VSC2is driven into saturation (overmodulation region) by the lowdc-link voltage and produces low order harmonics in phasevoltages, which cause flow of xy currents. The uncontrolledxy currents produce additional power losses and distort thecurrent waveform, causing the difference in the amplitudes ofthe currents in windings 1 and 2.

The same operating condition is simulated usingMATLAB/Simulink, and the results are shown in Fig. 7.The inherent asymmetry in the machine/converter is emulatedby adding additional resistance R = 2.8 in phases a1b1c1(the value was found via trial and error, so that a closeagreement with the experimental findings is obtained). Otherthan the value of R, all parameters used in the simulation areidentical to those used in the actual experiment. The simulatedcurrents closely resemble the experimental results, confirmingthe accuracy of the simulator.

The same test is repeated for the machine running at250 r/min, when a larger dc-link voltage drift is observed. Fig. 8shows the experimental results, where the phase currents aremore distorted than in the previous case because VSC2 is indeeper saturation due to the larger dc-link voltage imbalance.Again, simulated results in Fig. 9 show a good correlation withthe experimental findings.

Figs. 10 and 11 show the experimental and simulation results,respectively, when antisynchronous PI xy current controllersare activated with xy current references set to zero. Since


Fig. 8. Experimental results for no-load operation at 250 r/min, with onlydq current controllers. Channel 1: Vdc1 (100 V/div); Channel 2: Vdc2(100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4: phase a2 current(1 A/div); Horizontal: time (20 ms/div). (Markers for Channels 1 and 3 havebeen overlapped by markers for Channels 2 and 4, respectively.)

Fig. 9. Simulation results for no-load operation at 250 r/min, with only dqcurrent controllers.

Fig. 10. Experimental results for no-load operation at 500 r/min, with dq andxy current controllers activated. Channel 1: Vdc1 (100 V/div); Channel 2:Vdc2 (100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4: phase a2current (1 A/div); Horizontal: time (20 ms/div). (Markers for Channels 1 and 3have been overlapped by markers for Channels 2 and 4, respectively.)

Fig. 11. Simulation results for no-load operation at 500 r/min, with dq andxy current controllers activated.

one of the VSCs is now in saturation, the xy current con-trollers, which are designed to operate in the linear modulationregion, are unable to fully suppress the xy currents, and

Fig. 12. Experimental results for no-load operation at 500 r/min, showing theactivation of the dc-link voltage balancing controller at t = 1.0 s. Channel 1:Vdc1 (100 V/div); Channel 2: Vdc2 (100 V/div); Channel 3: phase a1current (1 A/div); Channel 4: phase a2 current (1 A/div); Horizontal: time(200 ms/div). (Markers for Channels 1 and 2 are overlapped.)

Fig. 13. Simulation results for no-load operation at 500 r/min, showing theactivation of the dc-link voltage balancing controller at t = 5.0 s.

Fig. 14. Experimental results for no-load operation at 500 r/min, with thedc-link voltage balancing controller activated. Channel 1: Vdc1 (100 V/div);Channel 2: Vdc2 (100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4:phase a2 current (1 A/div); Horizontal: time (20 ms/div). (Markers forChannels 1 and 3 have been overlapped by markers for Channels 2 and 4,respectively.)

only a slight improvement in terms of the current distortionis obtained. Dc-link voltages remain unbalanced in this casesince the active power consumed by the two VSCs is still notbalanced.

Figs. 12 and 13 show the activation of the dc-link voltagebalancing controller at t = 1.0 s (t = 5.0 s in Fig. 13), whenthe machine is running at 500 r/min without load. As predictedby the theoretical considerations, the dc-link voltages convergeand stay balanced after the activation of the controller. Figs. 14and 15 show that the currents for phases a1 and a2 becomemuch more balanced and are less distorted because the xycurrents are now under control and the dc-link voltages arecontrolled to achieve VSC operation in the linear region. Slightdistortion can be still observed due to the dead-time effect ofthe converter.


Fig. 15. Simulation results for no-load operation at 500 r/min, with the dc-linkvoltage balancing controller activated (Vdc1 and Vdc2 are overlapped).

Fig. 16. Experimental results for no-load operation at 250 r/min, showing theactivation of the dc-link voltage balancing controller at t = 1.0 s. Channel 1:Vdc1 (100 V/div); Channel 2: Vdc2 (100 V/div); Channel 3: phase a1current (1 A/div); Channel 4: phase a2 current (1 A/div); Horizontal: time(400 ms/div). (Markers for Channels 1 and 2 are overlapped.)

Fig. 17. Simulation results for no-load operation at 250 r/min, showing theactivation of the dc-link voltage balancing controller at t = 5.0 s.

The same procedure is repeated for operation at 250 r/minwithout load, as shown in Figs. 16 and 17. A slower conver-gence is observed since the same controller gains and limits areused, whereas the dc-link voltage difference is now larger.

Next, the performance of the dc-link voltage balancing con-troller is evaluated with the machine in variable-speed oper-ation. Fig. 18 shows the performance of the dc-link voltagebalancing controller when the speed reference increases from100 to 250 r/min at t = 1.2 s and then to 500 r/min at t = 2.2 s(t = 4.2 s and t = 5.2 s in the corresponding simulation studyin Fig. 19). It is evident that the dc-link voltages are kept atequal values during the whole transient operation.

The performance of the system under sudden load torquevariation is also evaluated in Figs. 20 and 21. A torque com-mand is first given to the DCS800, such that the dc machine

Fig. 18. Experimental results for speed variation from 100 to 250 to 500 r/minunder no-load conditions with the dc-link voltage balancing controller acti-vated. Channel 1: Vdc1 (100 V/div); Channel 2: Vdc2 (100 V/div); Channel 3:phase a1 current (1 A/div); Channel 4: speed (500 r/min/div); Horizontal: time(400 ms/div). (Markers for Channels 1 and 2 are overlapped.)

Fig. 19. Simulation results for speed variation from 100 to 250 to 500 r/min(Vdc1 and Vdc2 are overlapped).

Fig. 20. Experimental results for machine unloading at 250 r/min, with thedc-link voltage balancing controller activated. Channel 1: Vdc1 (100 V/div);Channel 2: Vdc2 (100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4:speed (500 r/min/div); Horizontal: time (400 ms/div). (Markers for Channels 1and 2 are overlapped.)

Fig. 21. Simulation results for machine unloading at 250 r/min, with the dc-link voltage balancing controller activated (Vdc1 and Vdc2 are overlapped).

provides a load torque to the six-phase machine. Using IRFOC,the six-phase machine is able to maintain the speed at250 r/min. At t = 1.6 s (t = 4.6 s in Fig. 21), the load torque


Fig. 22. Comparison of active powers, consumed by each VSC, (left bars)before and (right bars) after the activation of the dc-link voltage balancingcontroller, when the machine operates at 250 r/min with different load torque.

is removed by reducing the torque command to zero. A speedovershoot is observed together with the current amplitude re-duction, as a result of the load torque variation. Nevertheless,the dc-link voltages are kept at equal values throughout thisprocess, showing that the dc-link voltage balancing controlleris unaffected by the sudden load torque variation.

Based on (15), it is obvious that dc-link unbalance is aresult of power unbalance between the two windings. Fig. 22shows the comparison of the active power supplied to VSC1and VSC2 before and after the activation of the dc-link voltagebalancing controller. Results show that the power consumed byeach VSC (and hence each winding of the machine) is highlyunbalanced before compensation. The compensation restorespower balance between the two VSCs. This is accompanied bya slight reduction in total power, which is due to the reductionin loss producing harmonics associated with the saturationof one of the VSCs under unbalanced capacitor voltage. Theimbalance in power, apart from causing unbalanced capacitorvoltages, may also require derating of the machine to avoidoverloading of one of the windings/converters.

It is worth noting that the unbalance is, in this case, causedby inherent asymmetries in the machine windings/converters,which are manifested in the form of small differences in theper-phase resistances. As the load increases, the difference inRI2 power losses in these resistances becomes marginal; hence,power unbalance decreases with increasing load torque.

B. Experimental Results: Operation at LowerSwitching Frequency

The experimental results given so far were obtained for a5-kHz switching frequency. Since the topology is aimed athigh-power applications, experimental results for the operationof the system with a lower switching frequency of 1 kHzare presented next. Operating conditions are the same as forthe 5-kHz switching frequency, except that the PI controllergains have been reduced to ensure stable operation. Fig. 23shows the activation of the dc-link voltage balancing controller,and Fig. 24 shows the operation of the system under varyingspeed. While the dynamics of the system slightly differ, theresults show that the dc-link voltage controller can still bal-ance the dc-link voltage despite the reduction in the switchingfrequency.

Fig. 23. Experimental results for no-load operation at 250 r/min show-ing the activation of the dc-link voltage balancing controller at t = 1.0 s(switching frequency = 1 kHz). Channel 1: Vdc1 (100 V/div); Channel 2:Vdc2 (100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4: phasea2 current (1 A/div); Horizontal: time (400 ms/div). (Markers for Channels 1and 2 are overlapped.)

Fig. 24. Experimental results for speed variation from 100 to 250 to 500 r/minunder no-load conditions with the dc-link voltage balancing controller activated(switching frequency = 1 kHz). Channel 1: Vdc1 (100 V/div); Channel 2:Vdc2 (100 V/div); Channel 3: phase a1 current (1 A/div); Channel 4: speed(500 r/min/div); Horizontal: time (400 ms/div). (Markers for Channels 1 and 2are overlapped.)

Fig. 25. Experimental results for the machine running at 250 r/min withoutload with the dc-link voltage balancing controller activated. Channel 1: CMVfor winding 1 (100 V/div); Channel 2: phase a1 current (1 A/div).

C. Experimental Results: CMV andNeutral-to-Neutral Voltage

This subsection shows the experimental results for CMV andneutral-to-neutral voltage in the series-converter topology. Theresults are obtained for the motoring mode without load, withthe dc-link voltage balancing controller activated. Overall, thedc-link voltage is 300 V. Fig. 25 shows the CMV for winding 1.This voltage is measured as the neutral-point voltage of winding1 with respect to the dc-link midpoint. A zoomed-in view ofthe voltage waveform shows that the CMV changes in steps of50 V (Vdc/6), which agrees with the discussion in Section III.


Fig. 26. Experimental results for the machine running at 250 r/min withoutload with the dc-link voltage balancing controller activated. Channel 1: neutral-to-neutral voltage (100 V/div); Channel 2: phase a1 current (1 A/div).

Fig. 27. Wind speed variation used for simulating the system in generatingmode of operation.

Fig. 26 shows the neutral-to-neutral voltage of the machineunder the same operating condition. It is observed that theneutral-to-neutral voltage is around 0.333, 0.5, and 0.667 Vdc,since the PWM method used does not utilize states that giveneutral-to-neutral voltages of 0, 0.166, 0.833, and 1.0 Vdc (asstated in Section III). The dv/dt is also reduced since the stepsare Vdc/6.

D. Simulations in Generating ModeBy comparing the experimental and simulation results pre-

sented in Section V-A, it can be concluded that the MATLAB/Simulink simulator gives an accurate representation of theactual system. Here, investigations using the same simulator arepresented to verify the operation of the system in generatingmode. In the simulation, the total dc-link voltage is maintainedat 600 V, and the generator is subjected to a varying wind speedprofile, as shown in Fig. 27. All simulation results are shownfrom t = 2.0 s onward, when the machine has reached a steady-state operating point at rated wind speed.

An additional resistance R = 2.8 is added again in phasesa1b1c1 to emulate the winding asymmetry. Fig. 28 showsthe unbalanced dc-link voltage when the generator is con-trolled using only dq current controllers, with no dc-linkvoltage balancing control. The phase currents under this op-erating condition are unbalanced, with current amplitudes inwinding 1 lower than those in winding 2 (see Fig. 29). Thisis the result of the nonzero xy currents that, according to(21), lead to the current asymmetry. Since the voltage driftin this case is not severe enough to saturate the VSCs, thedistortions in phase currents are less significant. The remainingdistortion is caused by the uncontrolled xy currents and theuncompensated dead-time effect.

Fig. 28. Dc-link voltages under varying wind speed, with no dc-link voltagebalancing controller.

Fig. 29. Phase currents for generating operation without dc-link voltagebalancing controller.

Fig. 30. Dc-link voltages under varying wind speed, with dc-link voltagebalancing controller (Vdc1 and Vdc2 are overlapped).

Fig. 31. Phase currents for generating operation with dc-link voltage balanc-ing controller.

The same test is then repeated but with the dc-link voltagebalancing controller activated. Fig. 30 shows that the dc-linkvoltages are now balanced and are kept at an equal level.The phase currents in Fig. 31 still show different amplitudes,but now, the currents in winding 1 are higher than those inwinding 2. This is so because, in order to achieve dc-linkvoltage balancing, more active power needs to be generated in


Fig. 32. dq currents for generating operation with dc-link voltage balancingcontroller.

winding 1 to compensate for the larger copper loss due to theadditional resistance R. Although the dc-link voltage balancingcontroller injects nonzero steady-state xy currents, the amountof these circulating currents is lower than those in the previouscase (with uncontrolled xy currents), and this reduces thecurrent waveform distortion. The injection of these steady-statexy currents, which are caused by the generator asymmetries,produces additional stator winding losses. Fig. 32 shows thedq currents under varying wind speed conditions. The fluxcurrent is maintained at a constant level, whereas the torquecurrent is varied to follow the MPPT operation.

The flux and torque operation of the generator is practicallyunaffected by the dc-link voltage balancing controller. Thesimulation results for the generating mode show that, with thedc-link voltage balancing controller, the generator is able tooperate with proper flux and torque control.

VI. CONCLUSION

This paper has discussed the viability of an asymmetrical six-phase energy conversion system with cascaded machine-sideconverters and has presented a method for the voltage balancingof the dc-link midpoint. The topology and the concept areequally applicable to both variable-speed drive and generationapplications.

The series connection of the converters halves the individualdc-link voltages and the CMVs dv/dt. However, the voltageshifting between the two isolated neutral points can prevent theuse of low-voltage machines unless specific PWM techniquesare employed. The system is believed to be well suited toremote offshore wind farms with HVDC connection, where el-evation of the dc-link voltage and the use of only two cables forthe grid-side connection can reduce the overall infrastructurecost, but there is a potential problem with the drift of the dc-link midpoint voltage. This paper overcomes this limitation bydeveloping a dc-link voltage balancing controller that uses thexy currents to unbalance the winding currents in order tobalance the power sharing between the two sets of three-phasewindings. This represents an entirely new way of exploiting theexistence of additional degrees of freedom, which are availablein multiphase machines, which has never been reported before.Simulation and experimental results confirm that it is possibleto accurately control the dc-link midpoint voltage and to operatethe machine in variable-speed mode in both motoring andgeneration.

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Hang Seng Che received the B.Eng. degree in elec-trical engineering from the University of Malaya,Kuala Lumpur, Malaysia, in 2009. He is a recipientof the Kuok Foundation Postgraduate ScholarshipAward for his Ph.D. study and is currently workingtoward the Ph.D. degree under the auspices of a dualPh.D. program between the University of Malayaand Liverpool John Moores University, Liverpool,U.K.

His research interests include multiphase ma-chines and renewable energy.

Emil Levi (S89M92SM99F09) received theM.Sc. and Ph.D. degrees from the University ofBelgrade, Belgrade, Serbia, in 1986 and 1990,respectively.

From 1982 to 1992, he was with the Departmentof Electrical Engineering, University of Novi Sad,Novi Sad, Serbia. In May 1992, he joined LiverpoolJohn Moores University, Liverpool, U.K., where hehas been a Professor of electric machines and drivessince September 2000.

Dr. Levi serves as the Coeditor-in-Chief of theIEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, an Editor of the IEEETRANSACTIONS ON ENERGY CONVERSION, and the Editor-in-Chief of theIET Electric Power Applications. He was the recipient of the Cyril VeinottAward of the IEEE Power and Energy Society in 2009.

Martin Jones received the B.Eng. (first-class hon-ors) degree in electrical engineering and the Ph.D.degree from Liverpool John Moores University,Liverpool, U.K., in 2001 and 2005, respectively. Hewas a recipient of the Institution of Electrical Engi-neers Robinson Research Scholarship for his Ph.D.studies.

From September 2001 to Spring 2005, he was aResearch Student with Liverpool John Moores Uni-versity, where he is currently a Reader. His researchinterest is in the area of high-performance ac drives.

Mario J. Duran was born in Mlaga, Spain, in 1975.He received the M.Sc. and Ph.D. degrees in electricalengineering from the University of Mlaga, Mlaga,Spain, in 1999 and 2003, respectively.

He is currently an Associate Professor with theDepartment of Electrical Engineering, University ofMlaga. His research interests include modeling andcontrol of multiphase drives and renewable energiesconversion systems.

Wooi-Ping Hew (M06) received the B.Eng. degreeand the Masters degree in electrical engineeringfrom the University of Technology, Johor, Malaysia.He received the Ph.D. degree from the University ofMalaya, Kuala Lumpur, Malaysia, in 2000.

He is currently a Professor with the University ofMalaya Power Energy Dedicated Advanced Centre,University of Malaya. His research interests includeelectrical drives and electrical machine design.

Dr. Hew is a member of the Institution of Engi-neering and Technology, U.K. He is also a Chartered

Engineer in the U.K.

Nasrudin Abd. Rahim (M89SM08) receivedthe B.Sc.(Hons.) and M.Sc. degrees from TheUniversity of Strathclyde, Glasgow, U.K. He re-ceived the Ph.D. degree from Heriot-Watt University,Edinburgh, U.K., in 1995.

He is currently a Professor with the Universityof Malaya, Kuala Lumpur, Malaysia, where he isalso the Director of the Power Energy DedicatedAdvanced Centre.

Prof. Rahim is a Fellow of the Institution of Engi-neering and Technology, U.K., and the Academy of

Sciences Malaysia.

/ColorImageDict > /JPEG2000ColorACSImageDict > /JPEG2000ColorImageDict > /AntiAliasGrayImages false /CropGrayImages true /GrayImageMinResolution 300 /GrayImageMinResolutionPolicy /OK /DownsampleGrayImages true /GrayImageDownsampleType /Bicubic /GrayImageResolution 300 /GrayImageDepth -1 /GrayImageMinDownsampleDepth 2 /GrayImageDownsampleThreshold 1.50000 /EncodeGrayImages true /GrayImageFilter /DCTEncode /AutoFilterGrayImages false /GrayImageAutoFilterStrategy /JPEG /GrayACSImageDict > /GrayImageDict > /JPEG2000GrayACSImageDict > /JPEG2000GrayImageDict > /AntiAliasMonoImages false /CropMonoImages true /MonoImageMinResolution 1200 /MonoImageMinResolutionPolicy /OK /DownsampleMonoImages true /MonoImageDownsampleType /Bicubic /MonoImageResolution 600 /MonoImageDepth -1 /MonoImageDownsampleThreshold 1.50000 /EncodeMonoImages true /MonoImageFilter /CCITTFaxEncode /MonoImageDict > /AllowPSXObjects false /CheckCompliance [ /None ] /PDFX1aCheck false /PDFX3Check false /PDFXCompliantPDFOnly false /PDFXNoTrimBoxError true /PDFXTrimBoxToMediaBoxOffset [ 0.00000 0.00000 0.00000 0.00000 ] /PDFXSetBleedBoxToMediaBox true /PDFXBleedBoxToTrimBoxOffset [ 0.00000 0.00000 0.00000 0.00000 ] /PDFXOutputIntentProfile (None) /PDFXOutputConditionIdentifier () /PDFXOutputCondition () /PDFXRegistryName () /PDFXTrapped /False

/Description > /Namespace [ (Adobe) (Common) (1.0) ] /OtherNamespaces [ > /FormElements false /GenerateStructure false /IncludeBookmarks false /IncludeHyperlinks false /IncludeInteractive false /IncludeLayers false /IncludeProfiles false /MultimediaHandling /UseObjectSettings /Namespace [ (Adobe) (CreativeSuite) (2.0) ] /PDFXOutputIntentProfileSelector /DocumentCMYK /PreserveEditing true /UntaggedCMYKHandling /LeaveUntagged /UntaggedRGBHandling /UseDocumentProfile /UseDocumentBleed false >> ]>> setdistillerparams> setpagedevice

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