datasheet

LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGYAUGUST 1997 VOLUME VII NUMBER 3

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD and SwitcherCAD are trademarks of Linear TechnologyCorporation. Other product names may be trademarks of the companies that manufacture the products.

The LT1533 Heralds aNew Class of Low NoiseSwitching Regulators

by Jeff Witt

Every designer approaches switchingpower supplies with trepidation ordread—these circuits are infamousas generators of noise or electromag-netic interference (EMI). However,switching regulators are indispens-able when systems require small size,high efficiency, battery-poweredoperation or isolated power. Modernanalog systems combine theserequirements with demands for everincreasing performance; RF commu-nications devices are becoming moresensitive and analog-to-digitalconverters are pushing analog mea-surements to greater resolution andbandwidth. Such systems cannot tol-erate interference from any source.

Noise is Due toFast SwitchesIll effects due to ripple voltages andcurrents at the operating frequency ofa switching regulator can usually beeliminated by careful design. Low ESRcapacitors, closed inductors, LC fil-ters and proper trace routing to avoidground loops and IR drops will reducethis noise to acceptable levels.

However, switching regulator EMIproblems are most often the result ofhigh frequency noise generated byswitching high potentials or currents.In order to maintain high efficiency,these transitions are designed to occuras quickly as possible. These fastedges will couple capacitively or

inductively to nearby circuitry, andcan radiate energy to nearby or evenremotely placed circuits. The magni-tude of such stray coupling and itspotential for causing trouble are diffi-cult to predict. Once discovered,eliminating these effects requires alot of experimenting, numerous lay-out revisions, expensive shielding,black art and good luck.

Introducing the LT1533Low Noise SwitcherThe LT1533 is a new switching regu-lator that provides a solution to EMIproblems through two flexibleapproaches. First, the slew rates ofboth the current through the powerswitch and the voltage on it are easilyprogrammed with external resistors.Limiting these slew rates will removethe highest harmonics from theswitching waveforms. Second, theLT1533, with two 1A power switches,is designed to operate in push-pullcircuits. Such circuits, with their lowinput and output current ripple, areinherently quiet. The result is anintegrated switching regulator thatprovides very quiet output power andvery low emissions. Figure 1 gives anexample of what can be achieved. Thetop trace shows the output of a push-pull boost regulator generating 120mAat 12V from an input of 5V. This tracewas measured using a 10MΩ

continued on page 3

IN THIS ISSUE…COVER ARTICLEThe LT®1533 Heralds a New Classof Low Noise Switching Regulators..................................................1Jeff Witt

Issue Highlights ........................2

LTC in the News .........................2

DESIGN FEATURESThe LTC®1067 and LTC1067-50:Universal 4th Order Low Noise,Rail-to-Rail Switched CapacitorFilters ........................................6Doug La Porte

LTC1422: Simple and VersatileHot Swap™ Solution ................11David Soo

LTC1626 Low Voltage MonolithicStep-Down Converter Operatesfrom a Single Li-Ion Cell ..........15Tim Skovmand

LTC1340 Varactor Driver SavesPower in Cellular Phones .........18Dave Bell

DESIGN IDEASLT1635 1A Shunt Charger .......22Mitchell Lee

Negative-to-PositiveTelecommunication Supply .....23Kurk Mathews

Send Camera Power and Videoon the Same Coax Cable ..........24Frank Cox

MUX the LTC1419Without Software ....................25Kevin R. Hoskins

(More Design Ideas on pages 26–30;complete list on page 22)

DESIGN INFORMATIONUnderstanding and ApplyingVoltage References (Part Two)................................................31Mitchell Lee

New Device Cameos ..................37

Design Tools ............................39

Sales Offices ............................40

Linear Technology Magazine • August 19972

EDITOR’S PAGE

Issue HighlightsOur cover article for this issue intro-duces the LT1533, a new switchingregulator that provides a solution toEMI problems through two flexibleapproaches. First, the slew rates ofboth the current through the powerswitch and the voltage on it are easilyprogrammed with external resistors.Limiting these slew rates removes thehighest harmonics from the switch-ing waveforms. Second, the LT1533is designed to operate in push-pullcircuits. Such circuits, with their lowinput and output current ripple, areinherently quiet. The result is anintegrated switching regulator thatprovides very quiet output power andvery low emissions.

Another power control device, theLTC1626 low voltage monolithicstepdown converter, also debuts inthis issue. The LTC1626 is a mono-lithic, low voltage, step-down currentmode DC/DC converter with an inputsupply voltage range of 2.5V to 6V,making it ideal for single-cell Li-Ion or3- to 4-cell NiCd/NiMH applications.A built-in 0.32Ω (at VIN = 4.5V)P-channel switch allows up to 0.6A ofoutput current. The LTC1626 incor-porates automatic power saving BurstMode™ operation to reduce gate-charge losses when the load currentdrops below the level required forcontinuous operation. With no load,the converter draws only 160µA; inshutdown it draws a mere 1µA—mak-ing it ideal for current-sensitiveapplications.

Also introduced in this issue is aspecialized power device, the LTC1340varactor driver. This novel productneatly fills a “crack” that has ap-peared in the block diagram of portablecommunications equipment—theneed to provide 5V swing to the VCOcontrol input even though the rest ofthe product operates at 3V. TheLTC1340 is a low noise amplifier withan output that can swing to nearlytwice the supply voltage. A 4MHz on-chip charge-pump DC/DC converterproduces the doubled supply voltageto power the amplifier. The IC, to-

gether with three ceramic capacitors,consumes a mere 0.065 in2 of PCboard area.

On the filter front, we present theLTC1067 and LTC1067-50 univer-sal, 4th order switched capacitor filtersdesigned for rail-to-rail operation.Each part contains two identical, highaccuracy, very wide dynamic-range2nd order filter building blocks. Eachbuilding block, together with three tofive resistors, provides 2nd order fil-ter transfer functions, includinglowpass, bandpass, highpass, notchand allpass. These parts can be usedto easily design 4th order or dual 2ndorder filters. Both parts are doublesampled and can operate on suppliesfrom ±5V to 3V.

Also included in this issue is a newHot Swap™ controller, the LTC1422.Hot Swap controllers allow circuitboards to be safely plugged into a livebackplane. Like the LTC1421 HotSwap controller, introduced in theAugust 1996 issue of Linear Technol-ogy, the LTC1422 can handle supplyvoltages in the 3V–12V range. Thedifference between these two parts isin the number of supplies that can beindependently controlled: theLTC1421 is a 3-channel device,whereas the LTC1422 is a single-channel device.

This issue features an exception-ally large and varied selection ofDesign Ideas. There’s a solar-pow-ered battery charger, 3V to 1.xV, 12Vto 5V and –48V to 5V DC/DC convert-ers, a circuit that allows video fromand power to a surveillance camera toshare the same coax cable, a short-circuit protected isolated high-sideswitch, some novel applications forthe LTC1590 12-bit DAC and a methodfor MUXing the LTC1419 14-bit ADCwithout software.

This issue also includes the con-clusion of the in-depth examinationof voltage reference specifications, be-gun in the June 1997 issue of LinearTechnology. We conclude with a halfdozen New Device Cameos.

LTC in the News…Linear Technology Corporationannounced on July 22 that net salesfor its fiscal year ended June 29 were$379,251,000, a nominal increase overnet sales of the $377,771,000 for theprevious year. The Company alsoreported record net income o f$134,371,000 for the year, a nominalincrease over last year’s $133,964,000.Earnings per share decreased nomi-nally from $1.72 in FY’97, as theincrease in net income was offset by anincrease in shares outstanding.

According to Robert Swanson,president and CEO, “our businessaccelerated nicely in the second half ofthe year and we concluded the yearattaining record bookings, sales andnet profit. We are going into the newyear with good momentum. We areeven on Mars with several of our prod-ucts on both the Pathfinder spacecraftand the Sojourner robot.”

Net sales for the fourth quarterwere $104,075,000, a 16% increaseover the fourth quarter of the previousyear. Net income for the quarter was$37,402,000 or $0.47 per share, com-pared with $31,357,000 or $0.40 pershare, an increase of 19% over thefourth quarter of the previous year.Sequentially, the results for the fourthquarter were up 10% as compared tonet sales and net income reported forthe third quarter of $95,033,000 and$33,980,000 or $0.43 per share.

Linear Technology is increasing itsquarterly dividend from $0.05 to $0.06per share. Payment will be on August20, 1997 to shareholders of record onAugust 1, 1997.

These impressive financials are oneof the reasons that Linear Technologyhas again appeared in the “The Busi-ness Week Global 1000,” wherecompanies are ranked according to theirmarket value. LTC’s market value wasranked at 441, ahead of such firms asHarris, Southwest Airlines and Maxim.LTC has also been ranked in the “Top200,” a listing of the leading electroniccompanies worldwide, ranked by elec-tronics sales revenue, in the July issueof Electronic Business Today.

Chicago Corporation analyst DavidWu in New York said of LTC, “The globalmarkets are improving, orders are up,and the telecom industry, in whichLinear participates strongly is up. Inthe long run, Wu estimates “Linear’ssales can grow in the area of 25%.”

Linear Technology Magazine • August 1997 3

DESIGN FEATURES

oscilloscope probe with a six-inchground lead, demonstrating that thereis no significant inductively or ca-pacitively coupled noise. Probing theoutput of the LT1533 circuit with a50Ω low noise amplifier reveals thereal performance (second trace): peak-to-peak output ripple of the low noiseswitcher is only 150µV in a 10kHz to100MHz bandwidth.

A Closer Look at the LT1533The LT1533 is a fixed frequency cur-rent mode PWM switching regulator.Output voltage is regulated by con-trolling the peak switch current oneach cycle of the oscillator, resultingin good transient performance andrapid current limiting. The oscillator(see the block diagram in Figure 2)drives a toggle flip-flop, alternatelyenabling one of two 0.5Ω NPN powerswitches, QA and QB. The switchcurrent is monitored by a sense resis-tor at the emitter of the switch. Theoutput voltage (either positive or nega-tive) is compared with an accurateinternal 1.25V reference voltage byan error amplifier whose current out-put, along with loop compensationcomponents tied to the VC pin, deter-mine the peak switch current requiredfor regulation; a comparator turns offthe switch when this current level isreached.

The slew-control circuitry moni-tors the collector voltages and emittercurrents of the power switches andadjusts base drive to control both thevoltage and current slew rates. Thedesired rates are programmed by tyingthe RVSL and RCSL pins to ground withresistors between 4k and 68k, corre-sponding to slew rates from ~80V/µsto 5V/µs and 7A/µs to 0.4 A/µs. Thisallows the circuit designer to directlytrade off quiet, low EMI operationwith high efficiency: low slew ratesresult in slowly changing stray fields,which generate less interference, butincrease the conduction losses in theswitches.

The LT1533 oscillator presentsadditional opportunities for manag-ing EMI. Its wide frequency range(20kHz to 250kHz) allows the designerto avoid sensitive frequencies. Oper-ating frequency is set with a capacitor

–

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S

R

Q

FF

T

Q

QBk

FF

SLEW CONTROL

LDO REGULATOR

OSCILLATOR

ERROR AMP

COMPARATOR

NEGATIVE FEEDBACK AMP

OUTPUT DRIVERS

1.25V

QA

QB

RVSL

RCSL

COL BCOL APGNDVINSHDN

INTERNAL VCC

VC

NFB

FB

RT

CT

SYNC

DUTY

50k100k

GND

gm

+

–

1533_02.EPS

Figure 1. Output ripple of an LT1533switching regulator producing 120mAat 12V from a 5V input

LT1533, continued from page 1

Figure 2. LT1533 block diagram

5µs/DIV 1533_01.eps

200µV/DIV

20mV/DIV


DESIGN FEATURES

on the CT pin and a resistor of nomi-nally 17k on the RT pin. The LT1533can also be synchronized to an exter-nal clock, allowing accurate placementof both switching frequency andphase.

Push-Pull PWM Makes aQuiet Boost ConverterThe push-pull converter in Figure 3produces 200mA at 12V from an inputof 5V. The oscillator is set to 80kHz(note that the circuit operates at halfthis frequency) and the LT1533applies a pulse-width modulated 5Vto the primary side of the transformer.The rectified secondary voltage is fil-tered by L1 to generate 12V on C1. Inthis circuit, L1 is the primary energystorage device, so the transformer

can be made fairly small. Additionaloutput filtering is provided by L2 andC2.

This topology is inherently quiet.Current through L1 into the primaryoutput capacitor C1 is a continuoustriangle wave with little high frequencycontent, resulting in low conductedoutput noise. With an appropriatetransformer turns ratio, RMS inputcurrent is kept low, reducing thepotential for conducted noise on theinput.

It is advantageous to start with agood topology, but high frequencynoise will still get around via straycapacitance and mutual inductance;the best way to deal with this is toeliminate fast edges. Figure 4 showsseveral waveforms from the circuit as

18k 15k 2.49k

21.5k

0.015µF

LT1533

COL A

PGND

COL B

RCSL

RVSL

FBGND RT CT VC

5V

47µF 6.3V

25nH*

T1

4k–68k

4k–68k

1.2k

220pF

D1

D2

L1 300µH

L2 10µH

C1 22µF 20V

C2 22µF 20V

12V/200mA

*BEAD OR PCB TRACE T1 = COILTRONICS CTX02 13666-X1 L1 = COILTRONICS CTX300-2 L2 = COILTRONICS DT1608C-103 D1, D2 = MOTOROLA MBRS1100T3

VINNC

DUTY

SHDN

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1000pF

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10569

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it delivers 120mA of output current.The upper trace in each photo is thecurrent in switch QA as it turns off.Trace B is the output voltage probedwith a 10MΩ scope probe with a six-inch ground lead. The lower trace isthe output measured with a low noiseamplifier. In the left photo the switchslew rates are programmed to theirhighest values with 3.9k resistors onthe RCSL and RVSL pins. The fast switchtransients induce high frequencyripple on the output (the higher levelof noise on the middle trace is due tothe inductance of the scope probe’sground lead). By lowering the slewrates (RCSL = 24k and RVSL = 8.2k) thispotentially troublesome output rippleis eliminated, as shown in the rightphoto. The efficiency penalty is minor;the slower slew rates reduce efficiencyfrom 73% to 70%.

This combination of appropriatecircuit topology and controlled slewrates produces the exceptionally cleanoutput shown in Figure 1. This circuitis simply implemented with ordinaryPCB construction, and can be placedin close proximity to sensitive circuitswithout the need for expensive elec-trostatic or magnetic shielding.

DC Transformerwith Civilized EdgesGrounding the Duty pin of the LT1533disables the feedback loop and runseach switch at 50% duty cycle,allowing the LT1533’s use in DC trans-former circuits. Such circuits areuseful for generating bipolar or iso-lated supplies; Figure 5 shows an

Figure 3. 5V to 12V push-pull PWM converter

Figure 4. Lowering the slew rates of the power switches (Trace A) eliminates high frequency ripple at the output (Traces B and C).

0.2µs/DIV0.2µs/DIV 1533_04.eps

TRACE B 20mV/DIV

TRACE A 0.5A/DIV

TRACE C 500µV/DIV

TRACE B 20mV/DIV

TRACE A 0.5A/DIV

TRACE C 500µV/DIV


DESIGN FEATURES

example. The LT1533 switches 5Vacross a 3.3:1 transformer and a diodebridge rectifies the secondary sidevoltages to produce nominally 16Vbipolar outputs that are regulated to±12V. Short-circuit current limit atthe output is provided by the LT1533’sswitch current limit; the 1A switchlimit is transformed to 0.3A on thesecondary.

A common problem with isolated-output switchers is that fast edgescouple through stray capacitancebetween the primary and secondary

18k

LT1533

COL A

PGND

COL B RCSL

RVSL

FB

5V22µF 10V

25nH*

T1

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*BEAD OR PCB TRACE T1 = COILTRONICS CTX02 13716-X1

VINNC

DUTY

SYNC

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3.3

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1533_05.EPS

3300pF

12V 80mA

–12V 80mA

4 × 1N5819

LT1121-CS8

LT1175-CS8

22µF 35V

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150k

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windings of the transformer to createcommon mode noise on the outputs.Also, linear regulators are incapableof rejecting high frequency noise attheir inputs. Both problems are greatlyreduced by limiting the switch slewrates. Shielding between the wind-ings can be eliminated, reducingtransformer size and cost. LC filterson the isolated side are unnecessarywith the linear regulators rejectingripple at the operating frequency andthe controlled slew rates eliminatinghigh frequency ripple.

2200pF 0.01µF

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18k 10k 2.49k

1000pF

10569

LT1533

COL A

COL B

PGND

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RVSL

FB

C2 33µF 10V

L1 100µH

*BEAD OR PCB TRACE L1 = COILTRONICS CTX100-4 D1 = MOTOROLA MBRS120T3 C1 = AVX TPSD107M010R0100 C2 = AVX TPSC336M010R0375

VINNC

DUTY

SHDN

SYNC

1533_06.EPS

VOUT 5V/350mA

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3.3V to 5V Boost ConverterSimple switching topologies can alsobenefit from the LT1533’s low noisefeatures. In a boost regulator, forexample, the current into the outputcapacitor is a square wave, whichcontains the high frequency harmon-ics generated by a fast power switch.Even when the rectifying diode is off,fast voltage waveforms at the switchcouple through the Schottky diode’scapacitance. Fast switching can alsoexcite high frequency resonant circuitsformed by the diode’s capacitance andparasitic inductance due to boardtraces. All of these effects can bereduced by controlling the slew rate ofthe switch. Figure 6 shows the LT1533in a simple boost circuit generating5.0V from a 3.3V input, a typicalrequirement when interfacing 3.3Vlogic systems to 5V high performanceADCs. The collectors of the two powerswitches are tied together and alter-nately energize the boost inductor.Figure 7 shows several waveforms attwo different slew rate settings withthe circuit delivering 200mA of out-put current. Trace A is the switchvoltage, trace B is the current throughthe output capacitor and trace C isthe AC-coupled output voltage in a100MHz bandwidth. In the left photo,the slew rates are set to their maxi-mum values (RCSL = RVSL = 3.9kΩ).The rapidly switched current com-bined with the finite series inductanceof the output capacitor result in large

Figure 5. 5V to ±12V DC transformer

Figure 6. 3.3V to 5V boost converter continued on page 21


DESIGN FEATURES

The LTC1067 and LTC1067-50:Universal 4th Order Low Noise,Rail-to-Rail Switched Capacitor Filters

by Doug La PorteIntroductionModern products require ever smallerand more efficient designs. Much ofyesterday’s desktop equipment hasbeen replaced by small, often battery-operated, portable systems in today’sworld. Filter products must followthis trend and must address therequirements of today’s systemdesigner. These requirements includesmall package size, operation atreduced power supply voltages, lowpower consumption and large outputvoltage swings to accommodate highdynamic-range signals.

The LTC1068 family of universal,8th order switched capacitor filterswere the first products to answer thebell. They are half the size of theprevious generation of devices andtheir performance is superior in everyrespect. Next, the LTC1069 family ofpreconfigured, fully integrated,switched capacitor filters raised thebar in performance and ease of use.The LTC1069’s performance specifi-cations, coupled with standard SO-8packaging and zero external compo-nents, match the needs of modernsystems. Now, a brand new family ofparts has been added to the LinearTechnology filter portfolio: theLTC1067 and LTC1067-50.

Why Another Building Block?The LTC1068 family provides an ex-cellent solution for high performance8th order filters, but many systemsdo not need an 8th order filter; a 4thorder filter will do the job. Thesesystems will either use the LTC1068with extra unused sections or one ofthe older MF-10-type parts. The MF-10, in its many versions, was a fineproduct in its day, but its day haspassed. Modern systems must oper-ate on low power supply voltages. TheMF-10 parts are only good down to a

5V supply and most systems todayuse 3.3V supplies. To get acceptableperformance and high dynamic rangeon a 3.3V power supply, the filtermust have a low noise input struc-ture and provide a large output swing.If the output only swings to 800mVfrom the positive supply rail, you arethrowing away about 25% of theavailable dynamic range. For this rea-son, a rail-to-rail output stage isdesired. On the lower supply volt-ages, the part’s DC offset also becomesmore significant. The LTC1067 andLTC1067-50 were designed to fill theneeds of the 4th order filter market. Itis not just another version of the MF-10, but a completely new design. It ismade for modern systems; it operateson a 3V power supply, with low DCoffset, low input noise, rail-to-railoutput swing, low power consump-tion and small packaging. These newparts also deliver much tighter fre-quency accuracy than any previousproduct. This opens the door to reli-able, repeatable narrow-band filterdesigns.

LTC1067 andLTC1067-50 OverviewThe LTC1067 and the LTC1067-50are universal, 4th order switchedcapacitor filters with rail-to-railoperation. Each part contains twoidentical, high accuracy, very widedynamic-range 2nd order filter build-ing blocks. The block diagram forboth parts is identical (see Figure 1).Each building block, together withthree to five resistors, provides 2ndorder filter transfer functions, includ-ing lowpass, bandpass, highpass,notch and allpass. These parts can beused to easily design 4th order ordual 2nd order filters.

L inear Technology ’s newFilterCAD™ for Windows® filter designsoftware fully supports designs withthese parts. FilterCAD is a very pow-erful Windows-based filter designprogram, available free of charge fromLinear Technology. Consult LinearTechnology’s marketing or applica-tions staff for more information.

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CLK (16)

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SA (4)

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SB (13)

AGND (15)

15k

15k

1067_01.eps

Figure 1. LTC1067/LTC1067-50 block diagram

Windows is a registered trademark of Microsoft Corp.


DESIGN FEATURES

The center frequency of each 2ndorder section is tuned by an externalclock. The LTC1067 has a 100:1 clock-to-center frequency ratio. TheLTC1067-50’s clock-to-center fre-quency ratio is 50:1. Both of theseparts are double sampled. Doublesampling has the advantage of plac-ing all aliasing and imaging issues attwice the clock frequency. Ratios otherthan 100:1 or 50:1 are achieved byusing external resistors and choosingthe operating mode configuration. Formore details on the operating modes,consult the LTC1067/LTC1067-50data sheet.

The LTC1067 and LTC1067-50have the following salient features:

Rail-to-rail input and outputoperation

Operation from single 3V to ±5Vsupply

16-pin narrow SSOP or SO-16packages

>80dB dynamic range from a3.3V supply

Tight fO accuracy: ±0.2% typical,±0.55% maximum

Low DC offset: ≤ 5mV typical Low noise: < 70µVRMS (4th order

Butterworth LPF)

These parts also consume smallamounts of supply current. TheLTC1067 typically draws 2.0mA froma 3V supply, 2.55mA from a 4.75V

supply and 4.35mA from a ±5V sup-ply. The LTC1067-50 typicallyconsumes about one half of theLTC1067’s supply current: 1.0mAfrom a 3V supply, 1.45mA from a4.75V supply and 2.35mA from a ±5Vsupply. The LTC1067-50 is a lowerpower, slightly faster version of theLTC1067. However, the LTC1067 hasslightly lower noise, lower DC offsetand tighter fO tolerance than theLTC1067-50. These parts allow thedesigner to trade off one set of speci-fications for the other. The LTC1067is preferred, by virtue of its highersampling rate, for notch and highpassfilters, due to the broad-band natureof these filters.

Both parts are designed for single-or dual-supply-voltage applications.The parts have an integrated voltagedivider network; only a decouplingcapacitor is required to set a mid-supply potential for single-supplysystems. Figure 2 shows the powersupply connections for dual- andsingle-supply operation.

The LTC1067 and LTC1067-50 areideally suited for today’s systems.The ability to operate on low supplyvoltages, with low supply current,combined with low noise, low DC off-set and rail-to-rail output swing,makes these parts perfect for battery-operated systems that require highdynamic range. The small 16-pin nar-

row SSOP package is also in line withthe compact packaging requirements.These parts also feature tight fO accu-racy, allowing highly repeatable,manufacturable narrow-bandwidthbandpass and notch filters.

The LTC1067 and LTC1067-50 arealso mask programmable. This allowsthe designer to specify the filter’sresponse and have the part deliveredas a fully optimized, integrated solu-tion that has been tested to meet theexact system specifications. Mask-programmed parts have all externalresistors integrated, have the supplycurrent adjusted for the specificapplication and come in the smallSO-8 package. This results in aminimal-real-estate, zero-external-components design that is optimizedand tested to meet the design require-ments. For more information onmask-programmed parts, call LinearTechnology’s marketing department.

When designing filters, especiallyhigh Q filters, consult the “MaximumQ vs Center Frequency” curves andthe applications information in theLTC1067/LTC1067-50 data sheet.These curves serve as a guide to howfast you can push any design and stillmaintain a highly repeatable design.Using the FilterCAD design softwarewill automatically alert you to anydesign that is beyond the limits of thepart.

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FOR MODE 3, THE SA AND SB SUMMING NODE PINS ARE TIED TO THE AGND PIN

Figure 2a. Dual-supply ground plane connections Figure 2b. Single-supply ground plane connections


DESIGN FEATURES

Let the Games Begin:Some LTC1067 andLTC1067-50 Applications

High Dynamic-RangeButterworth Lowpass Filterwith Built-In Track-and-HoldChallenges Discrete DesignsFigure 3 shows an LTC1067 config-ured as a 5kHz Butterworth lowpassfilter. This circuit runs on a 3.3Vpower supply and uses an externallogic gate to stop the clock for track-and-hold operation. The transferfunction for this circuit, shown inFigure 4, is the classical Butterworthresponse. This circuit can be usedwith either the LTC1067 or theLTC1067-50. The broad-band noisefor the LTC1067 circuit is 45µVRMSand the DC offset is typically lessthan 10mV. For the LTC1067-50, thebroad-band noise is 55µVRMS and theDC offset is typically less than 15mV.

This circuit has tremendousdynamic range, even on low supplyvoltages. Figure 5 shows a plot of theLTC1067’s signal-to-noise plus totalharmonic distortion (SINAD) vs inputsignal level for a 1kHz input at threedifferent power supply voltages.SINAD is limited for small signals bythe noise floor of the LTC1067, formedium signals by the part’s linear-ity and for large signals by the outputsignal swing. The part’s low noiseinput stage and excellent linearityallow the SINAD to exceed 80dB forsignals as small as 700mVP-P, whilethe rail-to-rail output stage main-tains this level for input signals

approaching the supply rails. Previ-ous parts could not attain this highdynamic range due to higher inputnoise levels, poor linearity and lim-ited output-stage signal swing. Thelow noise and rail-to-rail output swingare especially crucial on the lower3.3V power supply, where every bit ofdetectable signal range is precious.Figure 6 shows the same plot for theLTC1067-50 circuit. The dynamicrange is not quite equal to that of theLTC1067, but is still very good. Recallthat, for the same clock frequency,the LTC1067-50 based filter hasdouble the bandwidth and half thesupply current of the LTC1067.

The LTC1067 and LTC1067-50 alsoperform a track-and-hold function.Stopping the clock holds the outputof the filter at its last value. TheLTC1067 is the best performing partin this area. The LTC1067’s hold stepis less than –100µV and the drooprate is less than –50µV/ms over thefull temperature range. These num-

bers compare very favorably with dedi-cated track-and-hold amplifiers.When the clock is restarted, the filterresumes normal operation within tenclock cycles and the output will thencorrectly reflect the input as soon asthe filter’s mathematical responseallows.

Elliptic Lowpass FilterThe LTC1067 family is capable ofmuch more challenging filters. Figure7 shows the schematic for a 25kHzelliptic lowpass filter using theLT1067-50 operating on a 5V supply.Maximum attenuation one octavefrom the –3dB corner is the designgoal for this filter. Figure 8 shows thefrequency response of the filter withthe –3dB cutoff at 25kHz and –48dBof attenuation at 50kHz. The broad-band noise of the filter is 85µVRMSand the DC offset is less than 15mVtypically.

Although Figure 7 shows the filterpowered by a single 5V supply, 3.3V

3.3V

0.1µF

1µF

VIN

R31, 49.9k

R41, 40.2k

R21, 40.2k

RL1, 59k

R42, 76.8k

R32, 40.2k

R22, 76.8kR11

52.3k

VOUT

CMOS LOGIC GATE

1067_03.EPS

V+

NC

V+

SA

LPA

BPA

HPA

INV A

CLK

AGND

V–

SB

LPB

BPB

HPB

INV B

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1067

500kHz

TRACK HOLD

1k 10k 100k

1067_04.EPS

–90.00

10.000

0

–10.00

–20.00

–30.00

–40.00

–50.00

–60.00

–70.00

–80.00

FREQUENCY (Hz)

GAIN

(dB)

0.1 1 10

1067_05.EPS

–100.00

–50.00

–55.00

–60.00

–65.00

–70.00

–75.00

–80.00

–85.00

–90.00

–95.00

INPUT VOLTAGE (Vp-p)

S/N

+THD

(dB)

VSUPPLY = 5V

VSUPPLY = ±5V

VSUPPLY = 3.3V

0.1 1 10–100.00

–50.00

–55.00

–60.00

–65.00

–70.00

–75.00

–80.00

–85.00

–90.00

–95.00

INPUT VOLTAGE (VP-P)

S/N

+THD

(dB)

1067_06.EPS

VSUPPLY = 3.3V

VSUPPLY = 5V

VSUPPLY = ±5V

Figure 3. High dynamic-range Butterworth LPF with track-and-hold control

Figure 4. Transfer function of the LTC10675kHz Butterworth LPF

Figure 5. Dynamic range of LTC1067Butterworth LPF

Figure 6. Dynamic range of LTC1067-50Butterworth LPF


DESIGN FEATURES

or ±5V supply operation is alsosupported. The maximum cutoff fre-quency is 15kHz for the 3.3V supplyand 35kHz for the ±5V supply. Thesame design and schematic used withan LTC1067 will achieve a somewhatlower noise, lower DC-offset filter.With the LTC1067, the broad-bandnoise is 70µVRMS and the DC offset istypically less than 10mV. The maxi-mum operating frequencies for theLTC1067 are one half of those for theLTC1067-50.

Narrow-Band Bandpass FilterDesign Extracts Small SignalsBuried in NoiseNarrow-band bandpass filters are dif-ficult to design but are easilyachievable with these parts. Mostapplications for these filters involveextracting a low level signal from anoisy environment. The noise may bethe standard broad-band, Gaussian-

type noise or it may consist of mul-tiple interfering signals. For example,the signal may be a low level tone or anarrow-bandwidth modulated signal,in a voice-band system. The presenceof the tone must be detected evenwhile the large voice signals arepresent. A narrow-band bandpass fil-ter will allow the tone to be separatedand detected even in this hostile envi-ronment. Numerous systems alsorequire a narrow bandpass filter to beswept across a band looking for thetones. Switched capacitor filters allowthe filter to be swept by simply chang-ing the clock frequency.

To achieve success in designingnarrow-band bandpass filters, youmust start with precision components.In an LC or RC design, you wouldhave to start with 0.1% resistors, 1%inductors and 1% capacitors to haveany hope of finishing with a success-ful, repeatable design in production.A competing solution, a digital filterimplementation, also requires preci-sion components. The full input signal(signal, noise and out-of-band inter-ference) must be correctly digitizedand then processed with a DSP deviceto finally determine the tone’s pres-ence. If an out-of-band interferingsignal is 20dB greater than the desiredtone, the ADC must have an extra20dB of dynamic range above thesignal’s requirement. To pull a small-signal tone from a large signalinterferer, you may need a 16-bit ADCto digitize the signal just to get 12-bitresolution of the tone after process-

ing. The added cost, power, boardspace and development time makethis approach unattractive.

A precision switched capacitor fil-ter provides a simple, small, low power,repeatable, inexpensive solution. Theolder MF-10-type parts do not havethe necessary fO accuracy to achievea reliable, repeatable design. Figure 9shows the schematic of a narrow-band bandpass filter centered at 5kHz.The design uses two identical cas-caded sections, each with a Q of 20.Multiply the individual Q of each sec-tion by 1.554 to calculate the total Qof a filter with two identical fO, identi-cal Q sections. This filter has a total Qof 31. For tunable filter applications,simply lowering the clock frequencylowers the center frequency of thefilter. Figure 10 shows the frequencyresponse of this filter. The broad-band noise of this filter is only90µVRMS. Highly selective bandpassfilters are possible due to theLTC1067’s excellent fO accuracy.

Higher Q, narrower bandwidth fil-ters are achievable with 0.1% resistors

V+

NC

V+

SA

LPA

BPA

HPA

INV A

CLK

AGND

V–

SB

LPB

BPB

HPB

INV B

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1067-50

–

+

5V

0.1µF

1µF

VIN

R61, 48.7k

R31, 49.9k

R21, 20k

R51 4.99k

RH1, 93.1k

RL1, 25.5k

R52 4.99k

1.25MHz

R62, 6.04k

R32, 21k

R22, 24.9k

RH2, 487k

RL2, 20k

RG, 21k

1/2 LT1498 VOUT

1067_07.EPS

R11 53.6k

1k 10k 100k 200k

1067_08.EPS

–90.00

10.000

0

–10.00

–20.00

–30.00

–40.00

–50.00

–60.00

–70.00

–80.00

FREQUENCY (Hz)

GAIN

(dB)

0.1µF

1067_09.eps

LTC1067

V+

NC

V+

SA

LPA

BPA

HPA/NA

INV A

CLK

AGND

V–

SB

LPB

BPB

HPB/NB

INV B

3.3V

R32, 200k

R22, 10k

R31, 200k

R21, 10kR11 200k

IN

R12, 200k

OUT

1µF

fCLK = 500kHz1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

FREQUENCY (kHz)4.0

GAIN

(dB)

6.0

1067_10.eps

4.5 5.0 5.5

0

–10

–20

–30

–40

Figure 7. 25kHz elliptic lowpass filter

Figure 8. Transfer function of LTC1067-5025kHz LPF

Figure 9. Low noise, low voltage narrow BPF

Figure 10. Frequency response of narrow BPF


DESIGN FEATURES

or matched resistor networks. AnLTC1067 mask-programmed part isideal for these ultranarrow filters.The well matched, on-chip resistors,coupled with specified test conditions,yield a fully functioning filter module,in an SO-8 package, without any ofthe hassles or cost of procuring preci-sion resistors or resistor networks.

Narrow-BandNotch Filter DesignReaches 80dB Notch DepthNarrow-band notch filters are espe-cially challenging designs. Therequirement for most notch filters isto remove a particular tone and notaffect any of the remaining signalbandwidth. This requires an infini-tesimally narrow filter that can onlybe approximated by a reasonablynarrow bandwidth. These types offilters, like the narrow-band bandpassdiscussed above, require precision fOaccuracy. Figure 11 shows the sche-matic of this type of filter. This filter isa 1.02kHz notch filter that is oftenused in telecommunication testsystems.

One of the challenges of designinga switched capacitor notch filterinvolves the broad-band nature of anotch filter. The broad-band noisecan be aliased down into the band ofinterest. Optimal high performancenotch filters should employ some formof noise-band limiting. To accomplishthe noise-band limiting, the design inFigure 11 places capacitors in paral-lel with the R2 resistors of each 2ndorder section. This forms a pole, set atfP = 1/(2 • π • R2 • C2), that will limitthe bandwidth. This pole frequencymust be low enough to have a band-limiting effect but must not be so lowas to affect the notch filter’s response.

0.1µF

1067_11.eps

LTC1067

V+ CLK5V

R52* 4.99k

R32 464k

R22 75k

R32 61.9k

R51* 4.99k

R61* 9.88k

R21 10k

R11 18.7k

VIN***

1µF

fCLK = 125kHz

R51, R61, R52, R62 ARE 0.1% TOLERANCE RESISTORS C21 AND C22 IMPROVE THE NOTCH DEPTH WHERE

NC

V+

SA

LPA

HPA/NA

BPA

INV A

1

2

3

4

5

7

6

8

16

15

14

13

12

10

11

9

AGND

V–

SB

LPB

BPB

HPB/NB

INV B

200Ω

R62* 10k

RH1 40.2k

C21** 300pF C22**

30pF

VOUT

* **

***

1 2π(R2X)(C2X)

(30)(fNOTCH) < < (75)(f NOTCH) WITHOUT

C21 AND C22 THE NOTCH DEPTH IS LIMITED TO –35dB

VIN ≤ 1.25VP-P

FREQUENCY (Hz)800

GAIN

(dB)

900 1000 1100

LT1067_12.eps

1200

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

Figure 11. Narrow-band notch filter

The pole should be greater than thirtytimes the notch frequency and lessthan seventy-five times the notch fre-quency for the best results. Figure 12shows the frequency response of thefilter. Note that the notch depth isgreater than –80dB. Without the useof the C21 and C22, the notch depthis only about –35dB.

ConclusionModern systems’ requirements aredifferent from those of the past. TheLTC1067 and LTC1067-50 are brandnew filter products optimized fortoday’s systems. They are small, highlyaccurate, low power, rail-to-rail, lownoise building blocks that have tre-mendous versatility.

Figure 12. Measured frequency response ofFigure 11’s narrow-band notch filter

for the latest information

on LTC products, visit

www.linear-tech.com


DESIGN FEATURES

LTC1422: Simple and VersatileHot Swap Solution by David Soo

IntroductionWhen a circuit board is plugged into alive backplane, the supply bypasscapacitors can draw large transientcurrents from the backplane powerbus as they charge. These inrushcurrents can destroy board traces or,at the least, disrupt the supply with alarge negative transient. There is aneed for a protection circuit to powerup the board in a controlled mannerand provide a low series-resistanceconnection during normal operation.

The LTC1422 allows a board to besafely inserted into and removed froma live backplane. The rate at whichthe supply ramps up at the board isprogrammable. In addition to hot swapprotection, it has a voltage monitorand reset output that can be used tosupervise the board’s circuitry. Thedevice also has overcurrent protec-tion, which disconnects the boardduring a short circuit. Most impor-

tantly, the LTC1422 combines thesefeatures in a compact 8-pin packagethat simplifies the design of any hot-swapped system.

Typical ApplicationLike the LTC1421 Hot Swap control-ler, introduced in the August, 1996issue of Linear Technology theLTC1422 can handle supply voltagesin the 3V–12V range. The differencebetween these two parts is in thenumber of supplies that can be inde-pendently controlled: the LTC1421 isa 3-channel device, whereas theLTC1422 is a single-channel device.

Figure 1 shows a typical applica-tion using the LTC1422. In order toprotect the board and capacitors, aseries-pass transistor, Q1, is placedbetween the connector and the ca-pacitors. The voltage at the gate of thepass transistor is raised at a con-

trolled rate. Because Q1 is an N-channel transistor, the supply on theboard follows the gate ramp minus athreshold voltage. The capacitor-charging current is limited to thecapacitor value times the ramp rate ofthe pass-transistor gate (I = C × dv/dt).

Due to special enhancements tothe LTC1422, the supply pin can with-stand any supply ramp-up rate.Following the connection of power tothe chip, there is a time delay beforethe GATE pin voltage starts toincrease. This allows time for the con-nector to seat after numerousbounces. This time delay is pro-grammed with the capacitor C2 onthe TIMER pin. The ON pin is nor-mally pulled high to allow the part toautomatically turn on the pass tran-sistor at power-up. If the ON pin isbrought low, the GATE pin will beheld at ground.

BACKPLANE

VCC

ON/RESET

CONN

ECTO

R 2

CONN

ECTO

R 1

GND

PLUG-IN CARD

1422_01.EPS

R1 0.005Ω

Q1 MTB56N06V

VCC 8

SENSE 7

GATE 6

R2 10Ω 5%

CHARGE PUMP

1.232V REFERENCE

GLITCH FILTER

LOGIC

C1 1.0µF

C4 2200µF

VOUT 5V/5A

12µA

µP

R3 6.81K

1%

R4 2.43K

1%

Q2

4

1

5

10µs FILTER

2.47 UVL

Cp1

Cp4

Cp2

REF

REF

LTC1422

2.0µA

Q1

TIMER 3

C2 0.33µF

ON 2

REF

GND

RESET

FB

Cp3

50mV

RESET

+–

+

–

+

–+

–

+

–

+

Figure 1. LTC1422 typical application


DESIGN FEATURES

ON

VCC

TIMER

2.45V

1.232V 1.232V

GATE

RESET

VOUT

1 2 3 4 5

1422_02.EPS

The timing for the board duringboard connection is shown in Figure2. When the connector makes initialcontact at time point 1, the ON pin ispulled high. If VCC is greater than the2.45V undervoltage lockout thresh-old, the TIMER pin starts charging C2with a 2µA current source. At timepoint 2, the supply opens. VCC nowdrops below the undervoltage thresh-old, signaling the LTC1422 todischarge the TIMER pin and wait forsupply connection. Once the connec-tor is seated, the TIMER pin completesa full timing cycle by charging to the1.232V threshold and the voltage atthe GATE pin begins to rise (timepoint 3). The rate at which the voltageat the gate of the pass transistor, Q1,increases is controlled by sourcing a10µA current source into C1. On-chip charge pumps provide gate driveof at least twice VCC to Q1.

Voltage MonitorThe LTC1422 uses a 1.232V refer-ence, a precision voltage comparatorand an external resistive divider tomonitor the output supply voltage.When the voltage at the FB pin risesabove its reset threshold voltage(1.232V), comparator Cp2’s outputgoes high and a timing cycle starts(Figure 2, time point 4). After a com-plete timing cycle, RESET is pulledhigh (time point 5). The 12µA pull-upcurrent source to VCC on RESET hasa series diode so the pin can be pulledabove VCC by an external pull-upresistor without forcing current backinto the supply.

70

60

50

40

30

20

100 40 80 120

FEEDBACK TRANSIENT VOLTAGE (mV)

GLIT

CH F

ILTE

R TI

ME

(µs)

1422_03.EPS

160 200 220

TA = 25°C

ON

30µs 40µs

15µs

TIMER

GATE

RESET

VOUT

1 2 3 4 5 6

1422_04.EPS

20µs

Figure 2. Typical insertion timing

Figure 4. Soft-reset timing

Figure 3. Glitch filter time vs feedbacktransient voltage


DESIGN FEATURES

Glitch FilterThe LTC1422 has a glitch filter toprevent RESET from generating asystem reset when there are transientson the FB pin. The filter is 20µs forlarge transients (greater than 150mV)and up to 80µs for small transients.The relationship between glitch-filtertime and the feedback transient volt-age is shown in Figure 3.

Soft ResetIn some cases, a system reset withouta power-down is desirable. The ONpin can signal the RESET pin to golow without turning off the power (asoft reset). This is accomplished byholding the ON pin low for only 15µsor less ( Figure 4, time point 1). Atabout 30µs from the falling edge ofthe ON pin (time point 2) the RESET

ON

VCC – VSENSE

VCC

TIMER

1.232V

GATE

RESET

VOUT

1 2

1.232V 1.232V 1.232V

1422_05.EPS

40µs

10µs

5V OUT

3.3V OUT

3.3V IN

5.0V IN

GND

RESET

ON

C1 0.33µF

16V

RESET

ON

TIMER

LTC1422

R2 0.01Ω 5%

R3 10Ω 5%

Q1 1/2 Si4936

D1 1N4148

Q2 1/2 Si4936

3.3V OUT

5.0V OUT

R7 10Ω 5%

C4 470µF 16V

C5 470µF 16V

R1 10k 5%

R4 2.74k 1%

TRIP POINT = 4.6V

R5 1k 1%

C2 0.022µF 25V

1422_06.EPS

C3 0.047µF 25V

R6, 1M 5%GND

1

2

3

4

8

7

6

5

VCC

SENSE

CURRENT LIMIT = 5A

GATE

FB

pin goes low and stays low for onetiming cycle.

If the ON pin is held low for longerthan 40µs, the GATE will turn off andthe RESET pin will eventually go low(time points 4, 5 and 6).

Hot Swapping Two SuppliesWith two external pass transistors,the LTC1422 can switch two sup-plies. In some cases, it is necessary tobring the dominant supply up firstduring power-up and ramp it downlast during the power-down phase.The circuit in Figure 6 shows how toprogram two different delays for thepass transistors. The 5V supply ispowered up first. R1 and C3 are usedto set the rise and fall delays on the 5Vsupply. Next, the 3.3V supply rampsup with a 20ms delay set by R6 andC2. On the falling edge, the 3.3Vsupply ramps down first because R6is bypassed by the diode, D1.

Electronic Circuit BreakerThe LTC1422 features an electroniccircuit-breaker function that protectsagainst short circuits or excessivecurrents from the supply. By placinga sense resistor between the supplyinput and SENSE pin, the circuit

Figure 5. Electronic circuit-breaker timing

Figure 6. Hot swapping 5V and 3.3V


DESIGN FEATURES

breaker will be tripped whenever thevoltage across the sense resistor isgreater than 50mV for more than10µs. When the circuit breaker trips,the GATE pin is immediately pulled toground and the external N-channelMOSFET is quickly turned off (Figure5, time point 1). When the ON pin iscycled off for longer than 40µs andthen cycled on (time point 2), thecircuit breaker is reset and anotherturn-on cycle is started. If the circuitbreaker feature is not required, theSENSE pin should be shorted to VCC.

Hot Swapping a 48V DC/DCModule with an Active-LowOn/Off Control SignalUsing a 5.1V Zener and a resistor, theLTC1422 can switch supplies muchgreater than its 12V VCC pin rating. Asshown in Figure 7, the pass transis-tor, Q1, is connected as a commonsource driver rather than as the source

follower used in previous applications.This allows the ground of the LTC1422to connect to the negative terminal ofthe 48V input. The clamp circuit con-sisting of R5 and D1 provides powerto the LTC1422.

The ON pin threshold is set at1.232V by the reference and com-parator Cp1 (Figure1). This allowsthe resistive divider of R1 and R2 inFigure 7 to monitor the input supply.When the supply is greater than 37Va timing cycle is generated and thepass transistor Q1 is turned on.

Because the pass transistor is in acommon-source configuration, caremust be taken to limit the inrushcurrent into capacitor C3. One way todo this is to precharge C3 using resis-tor R4. As the input rises, currentflows through R4 and charges capaci-tor C3. Once the input supply crosses37V, there is a timing cycle followedby the ramp-up of the GATE pin. By

R1 36k 5%

C3 100µF 100V

RESET1

VIN+ VOUT

+ 5V

VOUT–

SENSE+

SENSE–

1422-07.EPS

VIN–

AT&T JW050A1-E

50W

ON/OFF

ON

TIMER

GND

VCCQ2

MMBT5551LT1

OPTIONAL PRECHARGE RESISTOR

R4

510Ω, 5%

SENSE

LTC1422

GATE

FB

R2 1.2k 5%

CIRCUIT TURNS ON WHEN VIN > 37V CIRCUIT FOR ACTIVE LOW TURN-ON MODULES

C1 0.47µF 25V

C4 1µF 25V

R6 1M 5%

R7 270k 5%

C2 0.1µF 25V

D1 5.1V 1N751A

R3 10Ω 5%

R5 10k 5%

2

3

4

8

7

6

5

+ 48V

–

Q1 IRF530

this time, capacitor C3 is sufficientlycharged, thereby limiting the inrushcurrent. Another method for limitingthe inrush current is to slow down theramp-up rate of the GATE pin.

The reset comparator is used tomonitor the gate voltage. 290ms afterthe voltage at the GATE pin exceeds5.8V, the RESET pin will source 12µAinto Q2. The high voltage transistorQ2 is used to translate the RESETsignal to the module On/Off input.

ConclusionDesigning hot insertion systemsrequires a significant effort by anexperienced analog designer. One wayto reduce the design effort is to use aHot Swap building block, such as theLTC1422, which offers a charge-pumpgate driver, a millisecond timer andother specialized features. With theLTC1422, designers can easily createsafe and reliable hot-swappedsystems.

Figure 7. Hot swapping an AT&T 48V module


DESIGN FEATURES

LTC1626 Low Voltage MonolithicStep-Down Converter Operatesfrom a Single Li-Ion Cell by Tim Skovmand

IntroductionThe LTC1626 is a monolithic, lowvoltage, step-down current mode DC/DC converter with an input supplyvoltage range of 2.5V to 6V, making itideal for single-cell Li-Ion or 3- to 4-cell NiCd/NiMH applications. Abuilt-in 0.32Ω P-channel switch (VIN =4.5V) allows up to 0.6A of outputcurrent. The maximum peak inductorcurrent is externally programmableto minimize component size in lowercurrent applications.

The LTC1626 incorporates auto-matic power saving Burst Modeoperation to reduce gate-charge losseswhen the load current drops belowthe level required for continuousoperation. With no load, the converterdraws only 160µA; in shutdown itdraws a mere 1µA—making it idealfor current-sensitive applications.

Single-Cell Li-Ion OperationAs shown in Figure 1, a fully chargedsingle-cell Li-Ion battery begins thedischarge cycle at either 4.1V or 4.2V(depending upon the manufacturer’scharge voltage specifications). Duringthe bulk of the discharge time, the cellproduces between 3.5V and 4.0V.Finally, toward the end of discharge,the cell voltage drops fairly quickly

below 3V. It is recommended that thedischarge be terminated somewherebetween 2.2V and 2.8V (again,depending upon the manufacturer’sspecifications).

The LTC1626 is specificallydesigned to accommodate a single-cell Li-Ion discharge curve. Forexample, using the circuit shown inFigure 2, it is possible to produce astable 2.5V/0.25A regulated outputvoltage with as little as a 2.7V fromthe battery—thus obtaining the maxi-mum possible run time.

100% Duty Cyclein Dropout ModeAs the Li-Ion cell discharges, theLTC1626 smoothly shifts from a highefficiency switch-mode DC/DC regu-lator to a low dropout (100% dutycycle) linear regulator. In this mode,the voltage drop between the batteryinput and the regulator output isdetermined by the load current andthe series resistance of the internal P-channel power MOSFET, the currentsense resistor and the inductor resis-tance. When the battery voltage rises

again, the LTC1626 smoothly shiftsback to a high efficiency DC/DCconverter.

High Efficiency OperationUsing the circuit shown in Figure 3,efficiencies of greater than 90% aremaintained from 20mA to 250mA ofload current with a 3.5V input supplyvoltage, as shown in Figure 4.

Current Mode ArchitectureThe LTC1626 is a current mode DC/DC converter with Burst Mode opera-tion. This results in a power supplythat has very high efficiency over awide load-current range, fast tran-sient response and a very low dropoutcharacteristic.

An external, small valued senseresistor, RSENSE, provides current feed-back, which allows the LTC1626 tocontinuously control the inductorcurrent. When the load is heavy, theLTC1626 is in continuous operation.During the switch on-time, a built-incurrent comparator monitors the volt-age between the SENSE+ and SENSE–

pins connected across the external

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

0 1 2 3 4 5 6 7

DISCHARGE TIME (HOURS)

Li-Io

n CE

LL V

OLTA

GE (V

)

SGND

SHUTDOWN

1k

3900pF

1000pF

0.1µF

(VIN = 2.7V TO 4.5V)

VOUT

COUT†

100µF 10V

CIN††

47µF 16VPWR VIN VIN

SW

CT

ITH

SENSE+

SENSE–

PGND

LTC1626

D1 MBR0520LT1

L1* 22µH

RSENSE**

VFB

10k 1%

10k 1%

100pF

(2.5V/0.25A)0.1

+

+

Ω

CT 270pF

LBIN

LBOUT

SHDN

SINGLE Li-Ion CELL

+

*SUMIDA CDRH62-220 **IRC 1206-R100F †AVX TPSD107K010 ††AVX TPSD476K016

Figure 1. Typical single-cell Li-Ion dischargecurve Figure 2. Single-cell Li-Ion battery to 2.5V converter


DESIGN FEATURES

sense resistor in series with theinductor. When the voltage across theresistor reaches the comparator’sthreshold value, the internal P-chan-nel MOSFET is switched off. By usinga current mode architecture, theinductor current is predictable andwell controlled under all operatingconditions, making the selection ofthe inductor much easier.

Current mode control also givesthe LTC1626 excellent start-up andshort-circuit recovery characteristics.For example, when the output isshorted to ground, the off-time isextended to prevent inductor currentrun away. When the short is removed,the output capacitor begins to chargeand the off-time gradually decreases.The output returns smoothly to regu-lation without overshooting.

Operating FrequencyThe nominal off-time of the LTC1626is set by an external timing capacitorconnected between the CT pin andground. The operating frequency isthen determined by the off-time andthe difference between VIN and VOUT.

Figure 5 is a graph of normalizedoperating frequency versus powersupply voltage for the 2.5V regulatorshown in Figure 3. Note that the fre-quency is relatively constant whenthe supply voltage is between 3.6Vand 6V, but drops as the supply volt-age approaches the 2.5V regulatedoutput voltage.

The LTC1626 is capable of operat-ing at frequencies up to 700kHz, butthe highest efficiency is achieved

between 200kHz and 400kHz. As thefrequency increases, losses due tothe gate charge of the P-channel powerMOSFET increase. In space-sensitiveapplications, high frequency opera-tion allows the use of smallercomponents at the cost of a few effi-ciency points.

Low Voltage,Low RDS(ON) SwitchThe integrated P-channel switch inthe LTC1626 is designed to provideextremely low resistance at low sup-ply voltages. Figure 6 is a graph ofswitch resistance vs supply voltage.

Note that the RDS(ON) is typically0.32Ω at 4.5V and only rises toapproximately 0.40Ω at 3.0V. Thislow switch resistance ensures highefficiency switching as well as lowdropout DC characteristics at lowsupply voltages.

Low-Battery DetectionA built-in low-battery detector sensesthe input voltage through an externalresistive divider. The divided voltageconnects to the (–) input of a voltagecomparator (LBIN) and is compared tothe internal 1.25V reference voltage.

The LBOUT pin is an N-channelopen drain that goes low when thebattery voltage drops below thethreshold voltage. In shutdown, thecomparator is disabled and LBOUT isin the high impedance state. Figure 7is a schematic diagram detailing thelow-battery comparator connectionand operation.

SGND

SHUTDOWN

470Ω

3900pF270pF

1000pF

0.1µF

VIN

33µH VOUT

(2.7V to 6V)

COUT††

100µF 6.3V

CIN†††

47µF 16V

PWR VIN VIN

SW

CT

ITH

SENSE+

SENSE–

P GND

LTC1626 D1†

L* RSENSE**

VFB

10k 1%

10k 1%

100pF

2.5V/0.25A

0.1

+

+Ω

CT

*COILTRONICS CTX33-4 **IRC 1206-R100F †MBRS130LT ††AVX TPSC107M006R0150 †††AVX TPSD476K016

0.01 0.10 1.00

OUTPUT CURRENT (A)

EFFI

CIEN

CY (%

)

70

75

80

85

90

95

100

L1 = 33 µH V OUT = 2.5V R SENSE = 0.1 Ω C T = 270pF

VIN = 3.5V

INPUT VOLTAGE (V)

2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5

2.00

1.80

1.60

1.40

1.20

1.00

0.80

0.60

0.40

0.20

0

NORM

ALIZ

ED F

REQU

ENCY

L1 = 33 µH V OUT = 2.5V R SENSE = 0.1 Ω C T = 270pF

INPUT VOLTAGE (V)

1.00

0.90

0.80

0.70

0.60

0.50

0.40

0.30

0.20

0.10

0

R DS(

ON) (

Ω)

TJ = 25˚C

2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5

Typical Applications

3- or 4-Cell NiCd/NiMHDC/DC ConverterFigure 8 is a schematic diagram thatshows the LTC1626 being poweredfrom a 3- or 4-cell NiCd or NiMHbattery pack. (This circuit is also suit-able for operation from three or fouralkaline cells.) All the components

Figure 3. High efficiency 2.5V step-down converter

Figure 4. Efficiency vs output load current

Figure 5. Normalized operating frequency vssupply voltage

Figure 6. P-channel switch resistance vsinput voltage


DESIGN FEATURES

LTC1626

LBIN

VIN

R1 1%

R2 1%

–

+1.25V LBOUT

CFLTR 0.01µF

shown in this schematic are surfacemount and have been selected to mini-mize the board space and height. The

output voltage is set at 2.5V, but iseasily programmed to 3.3V for 4-cellapplications. Simply modify the twooutput ladder resistors, R1 and R2,from 10k each to 15k and 9.09k,respectively, as shown in Figure 8.

Single Li-Ion3.3V Buck/Boost ConverterThe circuit shown in Figure 9 pro-duces 3.3V from an input voltageranging from 2.5V to 4.5V. The twowindings of a common inductor coreare used to implement this circuit.Note that the current sense resistor is

connected to ground. The table inFigure 9 shows the output currentcapability as a function of batteryvoltage.

ConclusionThe LTC1626 is specifically designedto operate from a single-cell Li-Ionbattery. With its low dropout, highefficiency and micropower operatingmodes, it is ideal for battery operatedproducts and efficiency-sensitive de-vices such as cellular phones andhandheld industrial and medicalinstruments.

SGND

SHUTDOWN

1k

3900pF 270pF

1000pF

0.1µF

L1* 22µH VOUT

COUT†

100µF 10V

CIN††


SW

CT

ITH

SENSE+

SENSE–

PGND

LTC1626

D1 MBR0520LT1

VFB

R1 10k 1%

R2 10k 1%

100pF

2.5V/0.25A

RSENSE** 0.1Ω

+

+

CT

LBIN

LBOUT

SHDN

(VIN = 2.7V TO 6V)

3 OR 4 CELL

NiCD OR NiMH

+

FOR 3.3V: R1 = 15k 1% R2 = 9.09k 1%

*SUMIDA CDRH62-220 **IRC 1206-R100F †AVX TPSD107K010 ††AVX TPSD476K016

SGND

SHUTDOWN

1k

3300pF 75pF1000pF

0.1µF

(2.5V TO 4.2V)

VOUT

COUT †

100µF 10V

CIN††


SW

CT

ITH

SENSE+SENSE–

PGND

LTC1626D1

MBRS130LT1

L1B 33µH

RSENSE** 0.1Ω

VFB

15k 1%

9.09k 1%

100pF

3.3V

+

+

CT

LBIN

LBOUT

SHDN

+

L1A33µH

1 2

4

3

33µF 10V*

1

23

4TOP VIEW

L1B

L1B

L1A

L1A

VIN (V)

2.5 3.0 3.5 4.0 4.2

IOUT (mA)

200 350

0500* 0500* 0500*

MANUFACTURER COILTRONICS DALE

PART NO. CTX33-4 LPT4545-330LA

+Li-Ion

SINGLE CELL

* DESIGN LIMIT

*SANYO OS-CON CAPACITOR **IRC 1206-R100F †AVX TPSD107M010R0100†† AVX TPSE107M016R0100

Figure 7. Low-battery detector

Figure 8. 3- or 4-cell NiCd/NiMH to 2.5V converter

Figure 9. Single-cell Li-Ion to 3.3V buck/boost converter


DESIGN FEATURES

PHASE DETECTOR

FREQUENCY DIVIDER

LOOP COMP

REFERENCE FREQUENCY

LTC1340 VARACTOR

DRIVER VCO

VCCVCC

RF OUTPUT

DIVIDER PROGRAMMING

FROM µC

1340_01.EPS

LTC1340 Varactor Driver Saves Powerin Cellular Phones by Dave Bell

IntroductionDesigners of wireless communicationsequipment are constantly striving toincrease the operating time from abattery charge. For example, just afew years ago thirty to forty hoursstandby time was considered state-of-the-art for a Global System forMobile Communication (GSM) cellu-lar telephone. Today GSM phonesthat operate for more than one hun-dred hours in standby are available.One of the obvious tactics for achiev-

the frequency synthesizer may still bepowered from a low noise 5V powersupply. This penalizes the productwith high power consumption andgreater complexity.

Many component manufacturersnow offer frequency synthesizer ICsand voltage controlled oscillators(VCOs) that are specified for opera-tion down to 3V, but these still don’trepresent a total solution. The prob-lem lies with the tuning characteristicsof the varactor diode inside the VCO;even though the VCO itself may beoperable from a 3V supply, the con-trol voltage input (the cathode of avaractor diode within the VCO’s reso-nant LC tank) usually requires a swinggreater than 0V to 3V to span thenecessary frequency range. This isespecially true when considering theextra tuning range needed to accountfor production VCO tolerances. Whatis really needed is a simple method forboosting the control voltage appliedto the VCO’s varactor diode—andthat’s precisely what the LTC1340varactor driver is designed to do.

The LTC1340 is a low noise ampli-fier with an output that can swing tonearly twice the supply voltage. A4MHz on-chip charge-pump DC/DC

converter produces the doubled sup-ply voltage to power the amplifier.Size is extremely important in mod-ern cellular telephones, so theLTC1340 is packaged in the diminu-tive MSOP package. The IC, togetherwith three ceramic capacitors,consumes a mere 0.065 in2 of PCboard area.

FrequencySynthesizer BasicsFigure 1 depicts a basic phase lockedloop (PLL) frequency synthesizer blockdiagram. The output of the voltagecontrolled oscillator (VCO) is fed to aphase detector via a programmablefrequency divider. The phase detectorcompares the phase of this dividedsignal with a reference oscillator.During the period between rising edgesof the two signals, one of two currentsources is enabled at the output ofthe phase detector; a current issourced if the VCO phase is lagging,and a current is sinked if the VCOphase is leading. These current pulsesoccur once per cycle at the referenceoscillator frequency and are integratedon the loop compensation capacitor.The net effect is that the control volt-age applied to the VCO increases if

Figure 1. Frequency-synthesizer block diagram

Even in phones that operatealmost exclusively from 3V

supplies, the frequencysynthesizer may still be

powered from a low noise5V power supply. This

penalizes the product withhigh power consumptionand greater complexity.

ing such long standby time is reduc-tion of the operating voltage, sincereduced operating voltage usuallyproduces a substantial reduction inoperating power. Whereas most cel-lular phones used to operate from 5Vsupplies, newer models generally op-erate most circuitry in the range of 3Vto conserve power.

Good things in life usually havetheir trade-offs, and so does reducedoperating voltage. Most functionsinside a cellular phone can now beobtained with operation specifieddown to 3V. However, a few circuitsstill require higher operating volt-ages. The transmit power amplifiergenerally requires higher operatingvoltages to deliver the necessary out-put power—reducing the operatingvoltage here doesn’t reduce the totalpower needed. Another area that stillcauses problems is the frequency syn-thesizer. Even in phones that operatealmost exclusively from 3V supplies,


DESIGN FEATURES

LTC1340

PGND

CP AVCC

VCC

SHDN

DOUBLER CHARGE

PUMP WITH INTERNAL

FLYING CAPACITOR

IN

50Ω

PGND 1340_02.eps

62.3pF 1.15M

VS 0.62V

OUT ±20µA

AGND

CCP 0.1µF

(EXTERNAL)

0.1µF

COUT (EXTERNAL)

1.5M

47.9pF

–

+

–+

the phase is lagging and decreases ifthe phase is leading until the phaseerror approaches zero.

Loop compensation componentsare selected to optimize settling timefor the PLL, while keeping “referencespurs” at acceptable levels (referencespurs are unwanted sidebands in theVCO output spectrum at harmonicsof the reference frequency). The VCOfeedback divider ratio is controlled bya microcontroller to program thedesired output frequency to multiplesof the reference oscillator frequency.

As mentioned earlier, the phasedetector output comprises two cur-rent sources. Operating the phasedetector output voltage too close toeither ground or the positive supplyrail causes one of the two currentsources to saturate. Usually about400mV headroom is needed for thecurrent sources to perform withinspecifications. This means that theVCO control-voltage range is reducedeven further because of the head-room required at the phase detectoroutput, with this penalty increasingas a percentage of the total range assupply voltage is reduced. Forexample, with a 3V power supply theallowable control voltage is approxi-mately 0.4V to 2.6V. The LTC1340varactor driver is designed to beinserted between the phase detectorand the VCO in order to provide amuch larger voltage swing at the VCOwhile keeping the phase detector cur-rent sources in the linear region.

LTC1340 Circuit DescriptionThe LTC1340 is conceptually simple—it is a low noise amplifier combinedwith a charge-pump voltage doubleron the same IC (refer to Figure 2). Thisis possible because the intended load,a varactor diode, is reverse biasedand thus requires practically zero DCcurrent. All that is needed is suffi-cient output current to slew thevaractor diode’s junction capacitancein parallel with the amplifier’s ownoutput filter capacitor. The amplifieris capable of delivering approximately±20µA from its output, so the slewrate isn’t anything to brag about, butit’s sufficiently fast for GSM cellulartelephones that must be able to “fre-quency hop” in approximately 500µs.

A 2MHz on-chip oscillator drives abiphase charge-pump DC/DC con-verter, resulting in charge pulses beingdelivered at a 4MHz rate. This highfrequency operation has two benefits:it allows the use of small on-chipcharge-pump capacitors, and it makesthe resultant output ripple easy tofilter. The entire amplifier operatesfrom the AVCC pin, which is generallytied directly to the charge-pump out-put (CP). Although the amplifieroutput stage obviously needs thisboosted supply, the amplifier inputstage also benefits. Because of theboosted supply on the input stage,the amplifier input is rail-to-rail andoperates properly with a phasedetector signal anywhere between theinput rails.

Noise is the NemesisLow noise is without a doubt the mostcritical parameter of the varactordriver. Noise at the amplifier outputresults in unwanted frequency modu-lation of the VCO, which appears assidebands in the output spectrum.Most RF communications systemshave stringent limits on “out-of-band”energy, so this noise must be kept tovery low levels. Output noise for theLTC1340 is typically only 15µVRMS(25µVRMS maximum) over the 1kHz to100kHz bandwidth. This keeps theout-of-band energy about 15dB belowthe GSM cellular phone specificationlimits with a typical VCO.

Output noise comes primarily fromthree sources. The charge-pump DC/DC converter obviously producessome 4MHz ripple on the CP pin. Thiscan be reduced by a filter networkbetween the CP and AVCC pins, butthis measure is generally unneces-sary because of the low ripple presentat the charge-pump output, and theamplifier’s excellent high frequencypower supply rejection ratio (PSRR).A transconductance amplifier wasintentionally chosen because it has ahigh impedance current source out-put. This high impedance, coupledwith an output filter capacitor, resultsin excellent PSRR even at 4MHz. Infact, the 4MHz charge-pump ripplehas been reduced to the point whereit is undetectable in output spectrumplots in an actual frequency synthe-sizer system. The ground side of thefilter capacitor may be placed directly

Figure 2. LTC1340 block diagram


DESIGN FEATURES

adjacent to the VCO’s ground, allevi-ating additional noise that mightresult from high frequency grounddifferential between the LTC1340 andthe VCO.

The amplifier’s input stage alsocontributes to output noise, but thiseffect has been minimized by keepingthe input stage quiescent current ashigh as possible without exceedingthe desired power budget. The size ofthe input transistors was alsooptimized to put the 1/f noise cornerfrequency at approximately 10kHz—right where the PLL loop responsetypically begins to correct for phasenoise. The high impedance feedbackdivider also produces thermal noise,and this, surprisingly, has the largesteffect on output noise. The resistorvalues must be kept large to minimizeDC loading on the amplifier output,but large on-chip capacitors acrossthe resistors are used to keep noise incheck. Above 2kHz the feedbackimpedance is dominated by the ca-pacitors, minimizing the noisecontribution from the high valueresistors.

Other Design ConsiderationsAlthough MOSFET transistors have amuch higher 1/f corner frequencythan bipolar transistors, they werechosen for the input stage to keepbias current to a minimum. Inputbias current is guaranteed to be 1nAmaximum (25°C) in order to minimizereference spurs that can result frominput bias current. Because the phasedetector output impedance is highwhen both current sources aredisabled, even small amounts of biascurrent can result in voltage changeson the small loop compensation

capacitor between phase detectorsamples. These voltage changes looklike a low amplitude sawtoothwaveform and result in unwantedfrequency modulation of the VCO atthe reference frequency.

The transconductance amplifierarchitecture results in some addi-tional benefits. The current-sourceoutput topology of a transconductanceamplifier naturally provides nearlyrail-to-rail output swing, and this isessential for maximizing the varactordrive range. Because the outputcapacitor establishes the dominantpole for the amplifier’s feedback loop,the amplifier’s bandwidth and slewrate can be tailored by proper selec-tion of this capacitor. A 1nF outputcapacitor results in a bandwidth ofapproximately 125kHz. The minimumrecommended output capacitor forstability is 220pF, resulting in a band-width of approximately 560kHz. Bothbandwidth and slew rate are inverselyproportional to output capacitanceover a wide range of values. The rela-tionships among output capacitance,bandwidth and slew rate can be seenin Figure 3.

Voltage source VS in Figure 2 pro-duces intentional offset in theLTC1340’s transfer function. Refer-ring now to Figure 4, it can be seenthat this offset allows the amplifieroutput to go within 100mV of groundeven though its noninverting input isat approximately 400mV. Because theamplifier has a gain of approximately2.3, the amplifier output can be satu-rated high while the input is stillapproximately 400mV below the inputsupply (assuming a 3V nominal sup-ply voltage). This transfer functionprovides for rail-to-rail output swingwhile keeping the phase detector cur-rent sources in the linear region.

The whole point of reducedoperating voltages in portable com-munication equipment is powerreduction, so any additional currentconsumed by the LTC1340 itself mustbe kept small. Only 380µA total cur-rent is drawn from a 3V input supply,allowing significant reductions inoverall system power consumption.This power efficiency, coupled with

the small PCB area needed, make theLTC1340 very attractive for moderncellular telephones.

GSM Frequency SynthesizerEvaluation CircuitAn evaluation circuit has been devel-oped to demonstrate the performanceof the LTC1340 varactor driver in anactual frequency synthesizer appli-cation. This circuit replicates acomplete transmit frequency synthe-sizer for a GSM cellular telephone.Configuration switches allow pro-gramming the output to 890MHz,902.4MHz or 915MHz (bottom, middleand top of the GSM band).

The GSM cellular protocol sup-ports “frequency hopping”—changingthe frequency rapidly between timeslots. In the case of a GSM cellulartelephone, one synthesizer is oftenused for both transmit and receive ona time-division multiplexed basis. Thisrequires that the frequency synthe-sizer be able to change to a newfrequency and stabilize in just over500µs. The evaluation circuit may

INPUT VOLTAGE (V)0

OUTP

UT V

OLTA

GE (V

)

12

11

10

9

8

7

6

5

4

3

2

1

01 2 3 4

1340_04,eps

5 6

VCC = 2.7V

VCC = 5V

TA = 25°C COUT = 1nF IOUT = 0A VSHDN = VCC

VCC = 6V

1340_05.EPSSPAN = 1MHz RBW = 300Hz

10dB

/DIV

10

0

–10

–20

–30

–40

–50

–60

–70

–80

–90–5 –4 –3 –2 –1 0 1 2 3 4 5

COUT = 0pF

COUT = 220pF

COUT = 470pF

COUT = 1nF

Figure 3. LTC1340 small-signal response

Figure 4. LTC1340 DC transfer curve

Figure 5. Spectrum plot of frequencysynthesizer’s RF output


DESIGN FEATURES

also be programmed to hop betweenthe bottom and top frequencies onceevery millisecond, demonstrating itsability to hop and settle within 500µs.

Figure 5 shows a spectrum analyzerplot measured at the synthesizer’s RFoutput. The excellent noise perfor-mance of the LTC1340 can be seenfrom the roughly –75dBc measure-

voltage spikes on the output. Theright photo shows the same wave-forms with the slew rates lowered(RCSL = RVSL = 22k), eliminating thetroublesome transients. The penaltyis a drop in e f f ic iency f rom85% to 80%.

Conclusion: a Switcherfor Sensitive SystemsWith two 1A power switches, the abil-ity to control positive or negativeoutputs, and a wide input operating

range (2.7 to 30V), the LT1533 is ahighly flexible switching regulator.Thermal shutdown, in addition toswitch-current limit, provides circuitprotection. The LT1533 is packagedin the narrow 16-lead SO, and isavailable in commercial and indus-trial grades.

The LT1533 allows the circuitdesigner to add a switching regulatorto sensitive analog systems withoutfear of introducing uncontrollablenoise and interference. The program-

mable operating frequency and switchslew rates allow final tuning to occurin the circuit, when the system isrunning and interference problemsmay first become apparent. In addi-tion to providing a way to deal withunforeseen problems, this flexibilitymeans that sacrifices in efficiencywill be limited to those needed forproper system performance. TheLT1533 is the switching regulator ofchoice for high performance analogsystems.

Figure 7. Limiting switch slew rates (Traces A and B) lowers the high frequency content of the boost regulator’s output ripple (Trace C).

LT1533, continued from page 5

ment ±100kHz off carrier (each hori-zontal division is 100kHz). Small“reference spurs” can be observed at±200kHz off carrier due to thesynthesizer’s 200kHz reference fre-quency. The amplitude of thesereference spurs can be traded againstPLL settling time by optimization ofthe loop filter components.

SummaryThe LTC1340 varactor driver is thesmallest, most power-efficient solu-tion for delivering increased varactordrive to low voltage VCOs. This novelproduct neatly fills a “crack” that hasappeared in the block diagram ofportable communications equip-ment—the need to provide 5V swingto the VCO control input even thoughthe rest of the product operates at3V.

Authors can be contacted at (408) 432-1900



www.linear-tech.com

5µs/DIV5µs/DIV 1533_07.eps

TRACE B 0.5A/DIV

TRACE A 5V/DIV

TRACE C 20mV/DIV

TRACE B 0.5A/DIV

TRACE A 5V/DIV

TRACE C 20mV/DIV


DESIGN IDEAS

LT1635 1A Shunt Charger by Mitchell Lee

Most battery chargers comprisenothing more than a series-passregulator with current limit. In solar-powered systems, you can’t count onsufficient headroom to keep a seriesregulator alive, so a shunt method ispreferred. A simple shunt batterycharger is shown in Figure 1. It con-sists of an op amp driving a shunttransistor and ballast resistor, and isbuilt around an LT1635. This devicecontains both an op amp and a refer-ence, making it perfectly suited forregulator and charger applications.

Operation is straightforward: thebattery voltage is sensed by a feed-back divider composed of two 1Mresistors. The internal 200mV refer-ence is amplified to 7.05V andcompared against the feedback. RT1introduces a TC that accurately tracksthe battery’s correct charging voltage

over a wide temperature range.Because RT1 is designed to compensatefor changes in battery temperature, itshould be located close to the batteryand as far as possible from the shuntelements. When the battery chargesto 14.1V, the op amp output begins torise, turning on the Darlington shuntand resisting further increases in volt-age. Full panel power is divided equallybetween the transistor and 7.5Ωresistor when the battery is completelycharged. Don’t forget to provide ad-equate heat sinking and air flow forup to 15W dissipation.

The charger is designed to handle1A continuous, which is compatiblewith a “20W” panel. There is no needto disconnect or diode isolate thecharger during periods of darkness,because the standby current is only

230µA—less than 10% of the selfdischarge of even a small battery.

If a different or adjustable outputis desired, the feedback ratio can beeasily modified at the 1M divider.14.1V is a compromise between anaggressive charge voltage and a con-servative float voltage. Given the cyclicnature of insolation, allowing peri-odic charging at 14.1V is notdetrimental to Gelcell™ batteries. Thecircuit in Figure 1 will work withlarger or smaller batteries than thatshown. As a rule of thumb, the panelshould be sized from 1W per 10Ahbattery capacity (a float charge undergood conditions with a good battery)to 5W per 1Ah battery capacity (1 dayrecharge of a completely dischargedbattery under favorable conditions ofinsolation).Gelcell is a trademark of Johnson Controls, Inc.

–

+–

+

2A

12V, 5Ah Gelcell

7.5Ω/10W DALE HLM-10

TIP121

220Ω

100nF

OA 1/2 LT1635

REF 1/2 LT1635

1M

1M

64.5k

105Ω

2

3 7

6

48

1 7.05V

200mV

1A SOLAR ARRAY

2.43k RT1 7.5k

RT1 = THERM-O-DISC 1K752J

14.1V

Figure 1. 1A shunt battery charger (IDARK = 230µA; VFLOAT = 14.1V)

DESIGN IDEASLT1635 1A Shunt Charger .......22Mitchell Lee

Negative-to-PositiveTelecommunication Supply .....23Kurk Mathews

Send Camera Power and Videoon the Same Coax Cable ..........24Frank Cox

MUX the LTC1419Without Software ....................25Kevin R. Hoskins

Low Cost 3.3V to 1.xV 6 AmpPower Supply .......................... 26Sam Nork

Short-Circuit-Proof IsolatedHigh-Side Switch .....................27Mitchell Lee

The LTC1590 Dual 12-Bit DACis Extremely Versatile .............28Kevin R. Hoskins

12V Wall Cube to 5V/400mA DC/DCConverter is 85% Efficient .......30Steve Pietkiewicz



DESIGN IDEAS

Negative-to-PositiveTelecommunication Supply by Kurk Mathews

An increasing number of telecom-munication circuits require a positivesupply voltage derived from a –48Vinput. The traditional approach tonegative-to-positive conversion hasbeen to use a buck-boost converter(see Figure 1). Unfortunately, thistopology suffers drawbacks as thepower level and input-to-output volt-age difference increases.

A more appropriate solution for–48V to 5V conversion is shown inFigure 2. The LT1680 is used to imple-ment a forward converter with itsoutput referenced to the input com-mon. Compared to the buck-boostconverter, switch current is reducedby a factor of two and output capaci-tor ripple is reduced by a factor of five.

The LT1680 is referenced to –48Vand requires a 12V bias supply. The

12V is generated by using the RUN/SHDN and a bootstrap winding on theoutput inductor, L1. When input volt-age is first applied, R1 begins chargingC1. As C1 charges, Q1 is held on byR2, shorting R3. R4 and R5 form avoltage divider that holds the RUN/SHDN pin below its 1.25V thresholduntil the 12VIN pin reaches approxi-mately 14V. Once out of standby, Q1is turned off by Q2, reducing the runthreshold to approximately 9V andallowing C1 time to discharge slightlybefore the overwinding on L1 takesover. The only remaining issue is feed-back. Q3 translates the output voltageto a current, which flows to the VFBpin.

The LT1680’s unique differentialcurrent sense amplifier has an inputcommon mode range of –0.3V to 60V.

If VIN is expected to exceed 60V, thesense resistor could be relocated inthe main FET’s source and the inputcapacitors’ voltage increased. Becausethe forward converter is funda-mentally an isolated topology, anoptocoupler and reference could beadded to provide isolation betweenthe input and output of the supply.

+

+

–VIN

+VOUT

DI_1680_01.eps

9 10 13 15 1

7

8126543216

14

11

5VREF CT IAVG SS VC SGND PGND VREF

S+ S– GATE SYNC SL_ADJRUN/SHDN

12VINVFB

LT1680

+

++

+

+

+

T1

33Ω

33Ω

1.5nF

1.5nF

MBR2045CT

L1, OUTPUT 20µH

50Ω 1W

330µF 6.3V

SANYO OSCON

1k

300pF

4.22k

Q3 2N5401

IRF64010Ω

MBR0520LT1

220µF* 35V

220µF* 35V

220µF* 35V

220µF* 35V

24k

24k

1µF 63V

R1 24k

1.2k

0.1µF

0.1µF

0.22µF

1k

2.2nF

1nF

16k

20k1µF

R5 7.5k

R3 4.75k

Q2 2N3904

Q1 2N7000

20k

BAV21

R2 1M

C1 220µF 35V

0.1µF

R4 78.7k

L1, BIAS

–48V

INPUT COM

5V/6A

T1 = COILTRONICS VP5-1200, 1:1:1:1:1:1 (SIX WINDINGS, EACH 77 µH) L1 = PHILIPS EFD20-3F3-E63-S CORE SET (A l = 63nH/T2) OUTPUT 18T BIFILAR 22AWG, BIAS 54T BIFILAR 32 AWG * = SANYO CV-GX

0.015Ω 1W

Figure 2. –48V to 5V Telecommunication supply

Figure 1. Buck/boost converter


DESIGN IDEAS

–

+

–

+

–

+

++

+

+

CAMERAVIDEO OUT

12VR1 10k

R2 10k

C1 20µF

U4 LT1363

C2 4.7µF TANT

C3 1000µF

C4 1000µF

C6 1000µF

562Ω

R4 2k

R5 10k

C5 0.1µF

Q1 ZETEX ZTX749

OUT INU3

LT1086CT-12

R3 1k

U2 LT1363

C10 1000µF

0.1µF

R14 562Ω

R15 280Ω

R16 280Ω

C12 1000µF

C11 51pF

R17 75Ω

C13 100µF

R12 10k

R13 10k

24V 24V

TO MONITOR

3 7

6

42

3 7

6

42

U1 LT1206CT

R11, 100Ω

R10 0.1Ω

C9 20µF FILM

1

2 4

5

NC6

3

24V

4.7µF TANT R7

2.32k

0.1µF

4.7µF FILM

R9 500k

R8 11.5k

R6 75Ω

100' RG58U/U

C7 1000µF

24V

DI_VID_01.eps

+ +

+

+

+

20V DC

7

Send Camera Power and Videoon the Same Coax Cable by Frank Cox

Because remotely located videosurveillance cameras do not alwayshave a ready source of power, it isconvenient to run both the power andthe video signal through a single coaxcable. One way to do this is to use aninductor to present a high impedanceto the video and a low impedance tothe DC. The difficulty with this methodis that the frequency spectrum of amonochrome video signal extendsdown to at least 30Hz. The compositecolor video spectrum goes even lower,with components at 15Hz. This impliesa rather large inductor. For example,a 0.4H inductor has an impedance ofonly 75Ω at 30Hz, which is about theminimum necessary. Large inductorshave a large series resistance that

wastes power. More importantly, largeinductors can have a significantamount of parasitic capacitance andstand a good chance of going into selfresonance below the 4MHz videobandwidth and thus corrupting thesignal. The circuit shown in Figure 1takes a different approach to the prob-lem by using all active components.

The circuitry at the monitor end ofthe coax cable supplies all the powerto the system. U1, an LT1206 currentfeedback amplifier, forms a gyrator orsynthetic inductor. The gyrator iso-lates the low impedance power supplyfrom the cable by maintaining a rea-sonably high impedance over the videobandwidth while, at the same time,contributing only 0.1Ω of series resis-

tance. This op amp needs to haveenough bandwidth for video and suf-ficient output drive to supply 120mAto the camera. The selected part hasa guaranteed output current of 250mAand a 3dB bandwidth of 60MHz, mak-ing it a good fit. Because the videoneeds to be capacitively coupled, thereis no need for split supplies; hence asingle 24V supply is used. The 24Vsupply also gives some headroom forthe voltage drop in long cable runs.

The camera end has an LT1086fixed 12V regulator (U3) to supply12V to a black and white CCD videocamera. U4, an LT1363 op amp, sup-plies the drive for Q1, a fast, highcurrent transistor. Q1, in turn, modu-lates the video on the 20V DC. The

Figure 1. Circuit transmits video and 12V power on the same coax cable


DESIGN IDEAS

MUX the LTC1419 Without Software

The circuit shown in Figure 1 useshardware instead of software rou-tines to select multiplexer channelsin a data acquisition system. Thecircuit features the LTC1419 800ksps14-bit ADC. It receives and convertssignals from a 74HC4051 8-channelmultiplexer. Three of the four outputbits from an additional circuit, the74HC4520 dual 4-bit binary counter,are used to select a multiplexer chan-nel. A logic high power -on orprocessor-generated reset is appliedto the counter’s pin 7.

After the counter is cleared, themultiplexer’s channel selection inputis 000 and the input to channel 0 isapplied to the LTC1419’s S/H input.The channel-selection counter isclocked by the rising edge of the con-vert start (CONVST) signal thatinitiates a conversion. As eachCONVST pulse increments thecounter from 000 to 111, each multi-plexer channel is individually selectedand its input signal is applied to the

LTC1419. After each of the eight chan-nels has been selected, the counterrolls over to zero and the processrepeats. At any time, the input multi-plexer channel can be reset to 0 byapplying a logic-high pulse to pin 7 ofthe counter.

This data acquisition circuit has athroughput of 800ksps or 100ksps/channel. As shown in Figure 2, theSINAD is 76.6dB for a full-scale ±2.5V,1.19kHz sine wave input signal.

1

2

3

4

5

6

7

8

9

10

11

12

13

14

28

27

26

25

24

23

22

21

20

19

18

17

16

15

+AIN –AIN

VREF

COMP

AGND

D13 (MSB)

D12

D11

D10

D9

D8

D7

D6

DGND

AVDD

DVDD

VSS

BUSY

CS

CONVST

RD

SHDN

D0

D1

D2

D3

D4

D5

0.1µF

0.1µF

1µF

0.1µF

0.1µF

10µF

0.1µF0.1µF

10µF 10µF

14

13

12

11

10

1

2

7

15

16

2Q3

2Q2

2Q1

2Q0

2CE

1CLK

1CE

1CLEAR

2CLEAR

VCC

1Q0 1Q1 1Q2 1Q3

2CLK GND

0

1

2

3

4

5

6

7

13

14

15

12

1

5

2

4

AIN O

AIN 1

AIN 2

AIN 3

AIN 4

AIN 5

AIN 6

AIN 7

VCC

COM

INH

GND

VSS

16

3

6

8

7

A B C

9 8

3 4 5 6

74HC4051

74HC4520

LTC1419

5V

5V

–5V

DATA 0–13

CONVERT CONTROL

BUSY

5V –5V

CLEAR

COUNT

DI_MUX_01.EPS

+

++

–140

–120

–100

–80

–60

–40

–20

0

DI_MUX_02.EPS

INPUT FREQUENCY (kHz)

AMPL

ITUD

E (d

B)

0 10 20 30 40 50

fSAMPLE = 100ksps fIN = 1.19kHz VIN = ±2.5V

Figure 1. This simple stand-alone circuit requires no software to sequentially sample andconvert eight analog signal channels at 14-bit resolution and 100ksps/channel.

Figure 2. FFT of the MUXedLTC1419’s conversion of a full-scale1.19kHz sine wave

collector of Q1 is the input to the 12Vregulator. This point is AC groundbecause it is well bypassed as re-quired by U3. U1 is set up to deliver20V to the cable. Because the 12Vregulator in the camera end needs1.5V of dropout voltage, the balanceof 6.5V can be dropped in the series

resistance of the cable. The output ofthe LT1206 is set to 20V to give head-room between the supply and thevideo.

U2, another LT1363 video-speedop amp, receives video from the cable,supplies some frequency equalization

and drives the cable to the monitor.Equalization is used to compensatefor high frequency roll off in the cam-era cable. The components shown(R16, C11) gave acceptable mono-chrome video with 100 feet of RG58B/U cable.

by Kevin R. Hoskins


DESIGN IDEAS

Low Cost 3.3V to 1.xV 6 AmpPower Supply by Sam Nork

As voltage requirements for micro-processors drop, the need for highpower DC/DC conversion from a 3.xVsupply to a lower voltage keeps grow-ing. The LTC1430 is a very attractivechoice for such DC/DC applications,due to its low cost, high efficiency andhigh output power capability. How-ever, there are two problems: first,3.xV does not provide enough gatedrive to ensure low RDS(ON) using ex-ternal logic-level FETs; and second,the LTC1430 has a 4V minimum in-put requirement. These obstacles areboth overcome by using an LTC1517-5 regulated charge pump to generatethe input voltage for the LTC1430.

The circuit shown in Figure 1 usesthe LTC1430 to produce a synchro-nous 3.3V to 1.9V step-down DC/DCconverter. The circuit achieves 90.5%efficiency at 3 amps of output currentand has a 6 amp maximum outputcapability. (Refer to the LTC1430 datasheet for detailed description ofLTC1430-based designs). Power for

the LTC1430 is derived from the out-put of the LTC1517-5.

The LTC1517-5 is a switchedcapacitor charge pump available in atiny, 5-pin SOT-23 package. The partuses Burst Mode operation to generatea 5V output from a 2.7V to 5V input.It achieves regulation by sensing theoutput voltage via an internal refer-ence, comparator and resistor divider(see Figure 3). When the output hasdrooped below the lower trip point ofthe comparator, the charge pump isenabled, boosting VOUT back abovethe upper trip point. The LTC1517-5also contains thermal-shutdown andshort-circuit protection.

The regulated 5V supply powersthe internal circuitry of the LTC1430and ensures that the LTC1430 canprovide adequate gate drive to theexternal N-channel FETs. Withinsufficient gate drive, output powerand efficiency will be significantlyreduced due to high RDS(ON) of theFETs. In this circuit, typical supply

current drawn by the LTC1430 isbetween 25mA and 30mA, the vastmajority of which is needed to chargeand discharge the external FETs.Because the LTC1517-5 has a maxi-mum effective output impedance of50Ω, this current can be comfortablysupplied from a 3.3V input. If theinput voltage drops to 3V or lower, theLTC1517-5 output may also drop.However, with the FETs shown inFigure 1, the LTC1517-5 will providea 4.5V minimum supply to theLTC1430 at input voltages down to3V.

LTC1517-5

LTC1430CS

G1 PVCC1 PGND GND SENSE– FB SENSE+ SHDN

G2 PVCC2

VCC IFB

IMAX FREQSET

COMP SS

1 2 3 4 5 6 7 8

16 15 14 13 12 11 10 9

C3 0.22µF

5

1

2 3

4

C6 TO C9* 330µF 6.3V × 4

C15 10µF 10VC13

390pF

C14 0.012µFC12

0.1µF

C16 0.018µF

VOUT 1.9V 6A

C4 0.1µF

5V

C5 1µF

D1 BAT54

D2 MBRS120

R5 4.99K 1%

R6 10K 1%

L1 2.4µH, 8A SUMIDA

CDRH127-2R4

C11 1µF

Q1 Si4410

Q2 Si4410

R2 24k

R4 5.2k

C10 0.1µF

C1 Y5V CERAMIC

3.3µF

VIN 3.3V

C2 Y5V CERAMIC

10µF

+

C17 TO C21 * 330µF 6.3V × 5

*AVX TPS TANTALUM

1517 TA03

+

R1 100Ω

R3 1k

ON OFF +

Figure 1. 3.3V to 1.9V/6A power supply

LOAD CURRENT (A)

40

70

100

90

80

50

60EFFI

CIEN

CY (%

)

10

1517_02.EPS

0.1 1

VIN = 3.3V VOUT = 1.9V

Figure 2. Efficiency curve for Figure 1’s circuit


DESIGN IDEAS

Short-Circuit-Proof IsolatedHigh-Side Switch

Figure 1 shows a MOSFET switch,driven by the LTC1177-5 2.5kVRMSisolator. This device allows a logicsignal to control a power MOSFETand provides complete galvanic isola-tion. The device includes an internalcurrent limiting circuit, but at highervoltages limiting the current is justnot enough for effective protection ofthe MOSFET. Foldback (shown on theLTC1177 data sheet) helps, but thepart has trouble starting certain typesof loads when foldback current limit-ing is used. The circuit shown herelatches off in an overcurrent condi-tion and is restarted by cycling thelogic input.

Q1 and Q2 form an SCR with aholding current of less than 100nA. Ifthe load current exceeds approxi-mately 1A, the SCR fires, shorting theMOSFET gate to source. The LTC1177output current (about 7µA) is morethan adequate to hold the SCR on

indefinitely. The circuit resets whenthe logic input briefly cycles off.

Inductive loads present a specialproblem. If the load creeps up on theovercurrent threshold and fires theSCR, the load’s inductance will carrythe MOSFET source far below ground,which could destroy the MOSFET.Diode D1 clamps the gate at ground,

by Mitchell Lee

2.5kV ISOLATION BARRIER5V

LTC1177

VIN OUT

G1 SENSE G2

C1 10nF

Q2 2N3904

10MQ1 2N3906

10M

100Ω

20M

MTD3055EL

LOAD

0.5Ω 1W

D1 1N914

24V

OFF

ON

Figure 1. Short-circuit protected, isolated high-side switch

1517 BD

C1 0.22µF

VIN

CHARGE PUMP

LTC1517-5

VOUT

3.75M

1.25M

COUT

CIN

C1+C1–

800kHz OSC

THERMAL SHDN

1.25V REF

–

+

Figure 3. LTC1517-5 simplified block diagram

Pulling the SHDN pin on theLTC1430 low will shut down the powersupply. Q1 and Q2 will be forced offand the LTC1430 quiescent currentwill drop to 1µ A. Although theLTC1517-5 does not have a shutdownfeature, the no-load operating currentis an extremely low 6µA. This keepsthe overall shutdown current below10µA plus external FET leakage. (Forfurther reductions in shutdown cur-rent, an 8-pin LTC1522 may be usedin place of the LTC1517-5; theLTC1522 is the same as an LTC1517-5with shutdown.) The additionalLTC1517-5 circuitry will not take upmuch board space. The entire circuitconsumes only 0.045 in2.

turning the MOSFET back on, andsafely dissipates the stored magneticenergy in the MOSFET.

As shown the output rise time isabout 2ms, allowing the circuit tosuccessfully charge capacitors of upto 100µF. Increase C1 proportion-ately to handle higher value loadcapacitors.



DESIGN IDEAS

The LTC1590 Dual 12-Bit DACis Extremely Versatile by Kevin R. Hoskins

CMOS multiplying DACs make ver-satile building blocks that go beyondtheir basic function of converting digi-tal data into analog signals. ThisDesign Idea details some of the othercircuits that are possible when usingthe LTC1590 dual, serially interfaced12-bit DAC.

The circuit shown in Figure 1 usesthe LTC1590 to create a digitally con-trolled attenuator using DACA and aprogrammable gain amplifier (PGA)using DACB. The attenuator’s gain isset using the following equation:

VOUT = –VIN D 2n

where VOUT = output voltage VIN = input voltage n = DAC resolution in bitsD = value of code applied to DAC

The attenuator’s gain varies from4095/4096 to 1/4096. A code of 0can be used to completely attenuatethe input signal.

The PGA’s gain is set using thefollowing equation:

VOUT = –VIN 2n D

where VOUT = output voltageVIN = input voltage

n = DAC resolution in bitsD = value of code applied

to DACThe gain is adjustable from 4096/

4095 to 4096/1. A code of 0 is mean-ingless, since this results in infinitegain and the amplifier operates openloop. With either configuration, theattenuator’s and PGA’s gain are setwith 12 bits accuracy.

Further modification to the basicattenuator and PGA is shown in Fig-ure 2. In this circuit, DACA’sattenuator circuit is modified to givethe output amplifier a gain set by theratio of resistors R3 and R4. The

–

+

–

+

DATA IN

SERIAL CLOCK

CHIP SELECT/ DAC LOAD

DATA OUT

CLEAR

DIN

CLK

CS/LD

DOUT

CLR

13

14

11

4

15

24-B

IT S

HIFT

REG

ISTE

R AN

D LA

TCH

DACA

DACB

5V

7

10

16

0.1µF VIN A ±10V

VIN B ±10V

1

9 8

2

VREF B RFB B

VREF A RFB A 3

4

5

6

OUT1A

OUT2A

OUT2B

OUT1B

33pF

33pF

1/2 LT1358

1/2 LT1358

15V

–15V

0.01µF

0.01µF

VOUT A

VOUT B

VOUT = –VIN D 2n

VOUT = –VIN 2n D

LTC1590

DI1590_01.EPS

AGND

DGND

6

5

7

4

3

2 8

1

–

+

–

+

DATA IN

SERIAL CLOCK


DATA OUT

CLEAR

DIN

CLK

CS/LD

DOUT

CLR

13

14

11

4

15

24-B

IT S

HIFT

REG

ISTE

R AN

D LA

TCH

DACA

DACB

5V

7

10

16

0.1µF

VIN A ±10V

VIN B ±10V

1

9 8

2

VREF B RFB B

VREF A RFB A 3

4

5

6

OUT1A

OUT2A

OUT2B

OUT1B

33pF

33pF

1/2 LT1358

1/2 LT1358

15V

–15V

0.01µF

0.01µF

VOUT A

VOUT B

VOUT = –VIN 16D 2n

VOUT = –VIN 2n 16D

LTC1590

DI1590_02.EPS

R4 15k

15k

1k

R3 1k

15k1k

R2 15k

R1 1k

AGND

DGND

6

5

7

4

3

2 8

1

Figure 1. Driving DACA’s reference input (VREF) and tying the feedback resistor (RFB) to the opamp’s output creates a 12-bit accurate attenuator. Reversing the VREF and RFB connectionsconfigures DACB as a programmable-gain amplifier.

Figure 2. Modifying the basic attenuator and PGA creates gain for the attenuator (R3 and R4)and attenuation at the PGA’s input (R1 and R2).


DESIGN IDEAS

equation for this attenuator with out-put gain is

VOUT = –VIN 16D 2n

With the values shown, theattenuator’s gain has a range of–1/256 to –16. This range is easilymodified by changing the ratio of R3and R4. In the other half of the circuit,an attenuator has been added to theinput of DACB, configured as a PGA.The equation for this PGA with inputattenuation is

VOUT = –VIN 2n 16D

This sets the gain range from effec-tively –1/16 to –256. Again, this range

–

+

–

+

DATA IN

SERIAL CLOCK


DATA OUT

CLEAR

DIN

CLK

CS/LD

DOUT

CLR

13

14

11

4

15

24-B

IT S

HIFT

REG

ISTE

R AN

D LA

TCH

DACA

DACB

5V

7

10

16

0.1µF

VIN A

VIN B

1

9 8

2

VREF B RFB B

VREF A RFB A 3

4

5

6

OUT1A

OUT2A

OUT2B

OUT1B

U3A 1/2 LT1358

U3B 1/2 LT1358

15V

–15V –15V

0.01µF

15V

0.01µF

0.01µF

0.01µF

VOUT A

VOUT B

U1 LTC1590

DI1590_03.EPS

AGND

DGND

–

+U2B

1/2 LT1358

–

+U4B

1/2 LT1358

–

+U4A

1/2 LT1358

–15V

0.01µF

–

+U2A

1/2 LT1358

15V

0.01µF

10k

10k

10k

10k

10k

10k

RI

RI

CI

CI3

2 8

1

6

5

7

4

2

3

1

8

5

6

7

4

6

5

7

4

3

2 8

1

fC = D 2n+1 • π • RI • CI

can be modified by changing the ratioof R1 and R2.

The LTC1590 can also be used asthe control element that sets a low-pass filter’s cutoff frequency. This isshown in Figure 3. The DAC becomesan adjustable resistor that sets thetime constant of the integrator formedby U4 and CI. With the integratorenclosed within a feedback loop, alowpass filter is created.

The cutoff frequency range is afunction of the DAC’s resolution andthe digital data that sets the effectiveresistance. The effective resistance is

RREF = RI 2n D

Using this effective resistance, thecutoff frequency is

fC = D 2n+1 • π • RI • CI

The cutoff frequency range variesfrom 0.0000389/RC to 0.159/RC. Asan example, to set the minimum cut-off frequency to 10Hz, make RI =8.25k and CI = 470pF. At an inputcode of 1, the cutoff frequency is10Hz. The cutoff frequency increaseslinearly with increasing code,becoming 40.95kHz at a code of 4095.Generally, as the code changes by ±1bit, the cutoff frequency changes byan amount equal to the frequency atD = 1. In this example, the cutoff fre-quency changes in 10Hz steps.

Figure 3. This LTC1590-controlled dual single-pole lowpass filter uses RI and the DAC’s input code to create an effective resistance that sets theintegrator’s time constant and, therefore, the circuit’s cutoff frequency.


DESIGN IDEAS

12V Wall Cube to 5V/400mA DC/DCConverter is 85% Efficient

The ubiquitous 12V wall cube,power source of countless electronicproducts, generates an unregulatedDC voltage between 8V and 18V,depending on line voltage and load. Ifyou use a linear regulator to drop thevoltage to 5V, a 400mA load meansthe linear regulator must dissipate5W under worst-case conditions. Todeal with this heat, you must provideadequate heat sinking, increasingyour product’s size and weight. Addi-tionally, the heat is sometimesobjectionable to customers. Thesefactors can negate the cost advantageof a linear regulator. Figure 1’s cir-cuit, a negative buck converter,delivers 5V at loads up to 400mA from

a 7V–25V input with peak efficiencyof 85%, eliminating the need for aheat sink. Since the LT1307B (U1) isintended for use with a low inputvoltage, Q1 and Q2 are used to makea simple preregulator, providing 1.9Vfor U1’s VIN pin. The IC switches at600kHz, allowing a low cost 22µHinductor and 10µF ceramic outputcapacitor to be used. Q3 is needed tolevel shift the output voltage becauseU1’s feedback pin is referenced to thenegative input. Output ripple mea-sures 10mVP-P at a load of 400mA.The circuit’s efficiency is detailed inFigure 2, and response to a load stepfrom 150mA to 300mA is shown in

Figure 3. Input bypass capacitor C1sees worst-case RMS ripple currentequal to one-half the output currentand should have an ESR of less than0.5Ω. Take care during constructionto keep R1–R3 and Q3 close to U1’sFB pin and away from the SW pin toprevent unwanted coupling. Use aground plane and keep traces for thepower components short and direct.

Although it might seem unsettlingthat the negative side of the wall cubeis not grounded, remember that the9V wall cube floats. The circuit merelyregulates the negative side, ratherthan the more conventional positiveside.

SHDN VIN

SW

VC

FB

GND

U1 LT1307B

+

Q2 2N3904

10k

10k

30k

30k

Q1 2N3904 1µF

CERAMIC

C2 10µF

CERAMIC

1N5818

L1 22µH

R3 100k

Q3 2N3906

C1 33µF 25V

5V 400mA

L1 = SUMIDA CD54-220

12V UNREGULATED

SUPPLY

+

–

R2 42.2k, 1%

R1 12.1k 1%

DI_ADAP_01.EPS

1000pF

3 6

5

2

41

by Steve Pietkiewicz

5070

72

74

76

78

80

82

84

86

88

90

100 150 200LOAD CURRENT (mA)

EFFI

CIEN

CY (%

)

250 300 350 400

DI-ADAP_02.EPS

VIN = 8V

VIN = 12V

VIN = 18V

Figure 1. This negative buck converter delivers 5V at 400mA from a 7V–25V input.

Figure 2. Efficiency peaks at 85%; it is above80% over an input range of 8V–18V.

Figure 3. Load-step response; the load changes from150mA to 300mA.

VOUT 0.5V/DIV

50µs/DIVDI_ADAP_03.eps

IL1 0.5A/DIV

ILOAD 0.3A/0.15A


DESIGN INFORMATION

Board LeakageA new specter has entered the field ofreferences: board leakage caused bythe residues of water-soluble flux.The effect is not unlike that producedby the sticky juice extravasated froma ruptured electrolytic capacitor.Leakage from ground, supply railsand other circuit potentials into NC,trim and other sensitive pins throughconductive flux residues will causeoutput voltage shifts. Even if the leak-age paths do not shift the referenceout of spec, external leakage canmanifest itself as long-term outputvoltage drift, as the resistance of theflux residue changes with shifts inrelative humidity and the diffusion ofexternal contaminants. Water-solubleflux residues must be removed fromthe board and package surfaces, orcompletely avoided. In one case, theauthor observed an LT1009 shiftedout of spec by a gross leakage path ofapproximately 80kΩ between the trimpin and a nearby power supply trace.The leakage was traced to water-soluble flux.

Figure 8 shows how a good refer-ence can go bad with only a very smallleakage. A hypothetical industrial con-trol board contains an LT1027Aproducing 5V for various data acqui-sition circuits. A nearby trace carries24V. Just 147MΩ leakage into thenoise filtering pin (NR) causes a typi-cal device to shift +200ppm, and outof spec. Clearly, a 24V circuit trace

doesn’t belong anywhere near a 0.02%reference. This example is oversim-plified but clearly demonstrates thepotential for disaster.

A tightly packed circuit board mayleave no choice but to agglomerateincompatible traces. In this case, usea guard ring to eliminate referenceshift (see Figure 9). The output of thereference is divided down to 4.4V,equal to the potential on the NR pin,and used to bias a guard ring encir-cling the trace connecting NR to thenoise filter capacitor. This reducesthe effect of board leakage paths bymore than two orders of magnitude,shunting the errant leakage away fromthe guarded traces.

Trim-InducedTemperature DriftAbout half of LTC’s reference offer-ings include a pin for external(customer) trimming. Trimming maybe necessary to calibrate the system,but it can also adversely affect thetempco of the reference. For example,in the LT1019 bandgap reference,external trim resistors won’t matchthe tempco of the internal resistors.The mismatch causes a small(1ppm/°C) worst-case shift in theoutput voltage tempco, as explainedon the data sheet. The LT1021-5 andLT1236-5 standard trim circuit canbe modified, as shown in Figure 10, toprevent upsetting the references’

inherently low temperature coeffi-cients. Trimming the LT1027 has littleeffect on the output voltage tempco,and it needs no special consideration.Always check the reference data sheetfor specific recommendations.

Burn-InMost manufacturers of high-accuracysystems run their products through aburn-in procedure. Burn-in solvestwo problems at once: it relievesstresses built into the reference andcircuit board during assembly and itages the reference beyond the highestlong-term drift region, which occurswhen power is first applied to thepart. A typical burn-in procedure callsfor operating the board at 125°Cambient for 168 hours. If the mainconcern is stress relief, a shorter,unpowered burn-in cycle can be used.

Board StressBurn-in can help “relax” a stuffedboard, but additional mechanicalstress may be introduced when theboard is mounted into the product.Stress has a directly measurable effecton reference output. If the stresschanges over a period of time, it maymanifest itself as unacceptable long-term drift. Circuit boards are notperfectly elastic, so bending forcesmay cause permanent deformationand a permanent step-change in ref-erence output voltage. Devices in

Understanding and ApplyingVoltage References (Part Two)

by Mitchell Lee

LT1027 VOUTNR

24V

10V

5V

GND

VIN

REFACC_08.eps

1µF MYLAR

147MΩ

LT1027VOUTNR

24V

10V

5V

30k

220k

GND

VIN

REFACC_09.eps

GUARD RING

147MΩ

REFACC_10.eps

LT1236-5 OR

LT1021-5

R1 27k

1N4148GND

OUT VOUT

R2 50kTRIM

IN

Figure 8. Board leakage can wreak havoc witha precision reference. Here, a 147MΩ leakagepath to 24V pushes the 5V output out of spec.

Figure 9. Adding a guard ring protects againsterrant leakage paths.

Figure 10. The LT1021 or LT1236 output trimis made temperature insensitive by theaddition of a diode and a resistor.


DESIGN INFORMATION

metal (TO-5 and TO-46) packages arelargely immune to board stress, owingto the rigidity of the package and theflexibility of the leads. Plastic andsurface mount packages are anothermatter.

Board stress effects are easilyobserved by monitoring the output ofa reference while applying a bendingforce to the board. A controlledexperiment was performed to mea-sure the effect of board stress on anLT1460CS8-2.5 surface mount refer-ence. Devices were mounted in thecenter of 7" × 9" rectangular boards,as shown in Figure 11. The boardswere then deflected out-of-plane 18mils per inch, as shown in steps 1through 4. Figure 12 shows the neteffect on the output of one represen-tative sample measured over eightcycles of flexure.

The original board showed about60ppm peak-to-peak shift. The boardwas then slotted on a vertical mill,forming a 0.5" × 0.5" tab with thereference located in its center (alsoillustrated in Figure 11). The test con-tinued with the slotted configuration,and the output voltage variations werereduced to ±1 count (10µV) on themeter, or approximately 4ppm peak-

to-peak. This represents a tenfoldimprovement in stress-induced out-put voltage shift.

Several other techniques can beemployed to minimize this effect, with-out resorting to a milled board.Anything that can be done to restrictthe board from bending is helpful. Asmall, thick board is better than alarge, thin board. Stiffeners helpimmunize the board against flexure.Mount the circuit board with grom-mets, flexible standoffs or card-cagestyle so that minimal force is appliedto the mounting holes and board.

Part placement and orientation arejust as important. If a board issqueezed from opposite edges, thebending force tends to concentrate ina line down the center. Locate thereference away from the middle of theboard. Since the longer side of a boardis more flexible than the shorter, locatethe reference along the shorter edge.These recommendations are gener-alities; the placement, mountingmethod and orientation of other com-ponents and assemblies on the circuitboard will influence the mechanicalstrengths and weaknesses of the cir-cuit board.

Bench tests indicate that thestrongest axis for plastic packages isalong the shorter dimension of thebody of the plastic. Figure 13 showsthe correct orientation for surfacemount parts. Note that the part’slongest axis is placed perpendicularto that of the circuit board. The de-vices in Figure 13 are shown in thecenter of the board for illustrative

purposes only; comments aboutplacement still apply.

In spite of all precautions, extrane-ous effects may adversely affect thereference’s resistance to board stress.Watch out for adhesives and solderand flux debris under the package.These will create pressure points andinduce unpredictable stresses in thepackage. If a board has been sub-jected to a high bending force, some ofthe glass fibers and layers may breakor shear apart, permanently weaken-ing the board. Subsequent bendingforces will concentrate their stress atpoints thus weakened.

Figure 14 shows various schemesfor routing stress-relief slots on acircuit board, along with optimumpackage orientation. Note that thelongest axis of the reference is alignedwith the tab, not the shortest axis ofthe circuit board. This is in

REFACC_11.eps

9"

7"

1

SLOTTED AREA APPROXIMATELY 1/2 INCH BY 1/2 INCH

2

3

4

DEFLECTION NUMBER

0

80

40

160

120

OUTP

UT D

EVIA

TION

(ppm

)

40

REFACC_12.eps

0 10 3020

ORIGINAL CIRCUIT BOARD

SLOTTED CIRCUIT BOARD

4ppm = 10µV = METER RESOLUTION

REFACC_13.eps

LONGEST DIMENSION

LONGEST AXIS

S8

MSOP

REFACC_14.eps

(c)

(a)

(b)

(e)

(d)

Figure 11. Reference sensitivity to stress wasevaluated by assembling devices on a 7" × 9"circuit board and flexing, as shown in steps1–4.

Figure 12. Isolating stress by slotting thecircuit board reduces reference variations bymore than an order of magnitude.

Figure 13. Arranging the longest axes of theboard and package in perpendicularityminimizes stress-induced output changes.

Figure 14. Slotting the area around thereference can help isolate it from board stressif properly applied (see text).


DESIGN INFORMATION

TIME (MINUTES)

OUTP

UT V

OLTA

GE N

OISE

(20µ

V/DI

V)

12

REFACC_15.eps

0 6 8 1042

20µV

LT1021-7 (TO-5 PACKAGE) f = 0.01Hz TO 10Hz

FOAM CUP REMOVED

–

+

–

+

REFACC_16.eps

2XAAA ALKALINE CELLS

ZTX214C 1/2 LT1495

1/2 LT1495

R3 249k 0.1%

R4 24kΩ 5%

LT1634A-1.25 R2 1.00MΩ

0.1%

1µA COM 1.5V

R1 200kΩ 0.1%

R1, R2, R3 = MAR5 SERIES IRC (512) 992-7900

Figure 15. Air turbulence induces lowfrequency noise and compromises referenceaccuracy.

anticipation of flexing forces trans-mitted into the tab. The bestorientation for the tab is in line withthe longest axis of the board as in (b),(c) and (d). Bending forces along theweaker (longer) axis of the board couldbe coupled into (a) and (e). Note that

the ICs are aligned to resist thisforce. Use configuration (c) when thepart is located along the longer edgeof the board, and (d) when it is lo-cated along the shorter edge. Use (b)when the part is not located alongany edge.

Temperature-Induced NoiseEven though references operate onvery meager supply currents, dissi-pation in the reference is enough tocause small temperature gradients inthe package leads. Variations inthermal resistance, caused by unevenair flow, lead to differential lead

Figure 16. This pocket reference operates for five years on one set of AAA cells.

traP pagdnaBdeiruBreneZ ***edoM V52.1 V5.2 V5.4 V0.5 V0.7 V01

4001TL tnuhs 10.0 20.0

9001TL tnuhs 4.0

9101TL seires

5.1 5.1 5.1 5.1

A9101TL 3.1 3.1 3.1 3.1

1201TL seires 5.1 5.1 0.2

7201TL seires 8.2

9201TL tnuhs 6.0

1301TL seires 0.2

4301TL * tnuhs 20.0 20.0 *1.0

6321TL seires 5.1 5.1

0641TL seires 561.0 561.0 561.0

4361TL tnuhs

210.0

A4361TL 210.0

0001ZTL tnuhs **1.1

** reneZdeirubadnapagdnabahtobsedulcniecnereferlaud4301TLehT** Am5tnerrucreneZdednemmocer;tnerrucretaehedulcnitonseoD**

;tnerrucgnitarepomuminimsisecnerefertnuhsroftnerruC****** .tnerructnecseiuqmumixamsitisecnereferseiresrof

Table 3. Guaranteed supply current (mA) over temperature


DESIGN INFORMATION

traP pagdnaBdeiruBreneZ edoM V52.1 V5.2 V5.4 V0.5 V0.7 V01

4001TL tnuhs 23.0 18.0

9001TL tnuhs 2.0

9101TL seires 2.0 2.0 2.0 2.0

A9101TL seires 50.0 50.0 50.0 50.0

B1201TL

seires

0.1 7.0 5.0

C1201TL 50.0 50.0

D1201TL 0.1 7.0 5.0

A7201TL

seires

20.0

B7201TL 50.0

C7201TL 50.0

D7201TL 50.0

E7201TL 1.0

9201TL tnuhs

0.1

A9201TL 2.0

B1301TL

seires

50.0

C1301TL 1.0

D1301TL 2.0

4301TL * tnuhs 2.1 6.1 9.2–,3.4

A6321TL seires

5.0 5.0

C,B6321TL 1.0 1.0

A0641TL

seires

570.0 570.0 570.0

D,C,B0641TL 1.0 1.0 1.0

E0641TL 521.0 521.0 521.0

F0641TL 521.0 51.0 51.0

G0641TL 52.0 52.0 52.0

A,4361TL tnuhs 1.0

0001ZTL tnuhs 8.2–,2.4

ecnereferreneZdeirubadnapagdnabahtobsedulcniecnereferlaud4301TLehT*

Table 4. Guaranteed percentage initial accuracy at 25˚C


DESIGN INFORMATION

traP pagdnaBdeiruBreneZ edoM

erutarepmeTegnaR V52.1 V5.2 V5.4 V0.5 V0.7 V01

4001TL tnuhsC˚07–C˚0 **57 **57

C˚521–C˚55– **57 **57

9001TL tnuhsC˚07–C˚0 52

C˚521–C˚55– 53

9101TL

seires

C˚07–C˚0 02 02 02 02C˚58–C˚04– 02 02 02 02

C˚521–C˚55– 52 52 52 52

A9101TLC˚07–C˚0 5 5 5 5

C˚521–C˚55– 01 01 01 01

B1201TL

seires T NIM ≤TJ≤T XAM

5 5 5

C1201TL 02 02

D1201TL 02 02 02

A7201TL


2

B7201TL 2

C7201TL 3

D7201TL 5

E7201TL 5.7

9201TL tnuhs

C˚07–C˚0 43

C˚521–C˚55– 04

A9201TL T NIM ≤TJ≤T XAM 02

B1301TL


5

C1301TL 51

D1301TL 52

4301TL * tnuhs T NIM ≤TJ≤T XAM

04 04 )pyt(04

B4301TL 02 02 )pyt(04

A6321TL


5 5

B6321TL 01 01

C6321TL 51 51

B,A0641TL


01 01 01

C0641TL 51 51 51

E,D0641TL 02 02 02

G,F0641TL 52 52 52

4361TL tnuhs T NIM ≤TJ≤T XAM

02

A4361TL 01

0001ZTL tnuhs TJ C˚56= 1.0

* ecnereferreneZdeirubadnapagdnabahtobsedulcniecnereferlaud4301TLehT*nosirapmocfosesopruprofdetamitsesiocpmet;deetnaraugerutarepmetrevoegatlovtuptuoetulosbA**

Table 5. Temperature coefficient (ppm/˚C)


DESIGN INFORMATION

temperatures, thereby causing ther-moelectric voltage noise at the outputof the reference. Figure 15 dramati-cally demonstrates this effect. Thefirst half of the plot was made with anLT1021H-7 buried Zener reference,which was shielded from ambient airwith a small foam cup (Dart Con-tainer Corporation Stock No. 8J8 orsimilar). The cup was removed at sixminutes elapsed time for the secondhalf of the test. Ambient in both caseswas a lab bench-top with no excessiveturbulence from air conditioners,opening/closing doors, foot traffic or547 exhaust. Removing the foam cupincreased the output noise by almostan order of magnitude in the 0.01Hzto 10Hz band.

The kovar leads of the TO-5 work-ing against copper circuit traces arethe primary culprit. Copper leadframes used on DIP and surface mountpackages are not nearly as sensitiveto air turbulence because they areintrinsically matched. Still, externalcomponents create thermocouples oftheir own with potentials of 10 µV/°Cor more per junction. In a LT1021-7reference, this represents more than1ppm/°C shift from each thermoelec-tric generator. Temperature gradientsacross the circuit board and dissipa-tion within external components can

lead to the same kind of noise as wasshown in Figure 15.

Temperature gradients may arisefrom heat generators on the board.Position the reference and its associ-ated external components far fromheat sources and, if necessary, userouting techniques to create an iso-thermal island around the referencecircuitry. Minimize air movementeither by adding a small enclosurearound the reference circuitry, or byencapsulating the reference circuitryin self-expanding polyurethane foam.

Reference ApplicationsThe unique pocket reference shownin Figure 16 is a good match for a pairof AAA alkaline cells, because thecircuit draws less than 16µA supplycurrent. Two outputs are provided: abuffered, 1.5V voltage output, and aregulated 1µA current source. Thecurrent source compliance rangesfrom approximately 1V to –43V.

The reference is self-biased, com-pletely eliminating line regulation asa concern. Start-up is guaranteed bythe LT1495 op amp, whose outputsaturates at 11mV from the negativerail. Once powered, there is no reasonto turn the circuit off. One AAA alka-line contains 1200mAH capacity,enough to power the circuit through-out the 5-year shelf life of the battery.Voltage output accuracy is about0.17% and current output accuracyis about 1.2%. Trim R1 to calibratethe voltage (1kΩ per 0.1%), and R3 tocalibrate the output current (250Ωper 0.1%).

Low noise synthesizers need quietpower supplies for their VCOs andother critical circuitry. 3-terminalregulators exhibit far too much noisefor this application, calling insteadfor a regulator constructed from areference. A practical example isshown in Figure 17. Current throughthe LT1021-5 reference is used todrive the base of a PNP pass device,resulting in an available output cur-rent of at least 1A. In this example,the current is intentionally limited to200mA by the addition of emitterdegeneration and base clamping. Thelow noise of the reference is preserved,

giving a 100-fold improvement overthe noise of an equivalent 5V, 3-ter-minal regulator, not to mentionimproved initial accuracy and long-term stability. Typical output noise is7µ VP-P over a 10kHz bandwidth.

ConclusionWhen specifying a reference, keep inmind that initial accuracy, tempera-ture coefficient, and long-termstability all play a role in overall accu-racy of the finished product. By takingsome care in applying the reference,and by avoiding some key pitfalls, thereference’s inherent accuracy can bepreserved. Tables 3–5 compare qui-escent current, initial accuracy andtemperature coefficient for each ofLTC’s references.

For Further Reading“The Ultra-Zener—A PortableReplacement for the Weston Cell?” byPeter J. Spreadbury; IEEE Transac-t ions on Instrumentat ion andMeasurement, Vol. 40, No. 2, April1991, pages 343–346Application Note 42: Voltage Refer-ence Circuit Collection, by BrianHuffman; Linear Technology Corpo-ration, June 1991.

REFACC_17.eps

+

LT1021-5OUT

2.0Ω

10Ω 1/2W

5V/200mA OUTPUT

4.7Ω

ZBD949*

220Ω

RED LED**

ZETEX INC (516) 864-7630 GLOWS IN CURRENT LIMIT. DO NOT OMIT.

* **

9V-12V INPUT

10µF Ta

+47µF

IN

GND

Figure 17. Ultralow noise 5V, 200mA supplyoutput noise is 7µVRMS over a 10Hz to 10kHzbandwidth. Reference noise is guaranteed toless than 11µVRMS. Standard 3-terminalregulators have one hundred times the noiseand no guarantees.



www.linear-tech.com

Part one of this article appeared inthe June 1997 issue of LinearTechnology.


NEW DEVICE CAMEOS

The LT1497:Dual 50MHz, 125mACurrent Feedback AmplifierThe LT1497 dual 50MHz current feed-back amplifier is ideal for drivingtwisted pairs, buffering low imped-ance loads and driving multipleback-terminated cables. The devicefeatures high output drive and lowdistortion from a modest supply cur-rent. From a low 7mA maximumsupply current per amplifier, eachoutput drives a minimum of ±100mAto within 2V of each supply. Distor-tion at up to 1MHz is better than–70dBc driving ±40mA peak signalsinto a 135Ω twisted pair.

Operation is fully specified forsupplies from ±2.5V to ±15V. Theamplifiers have current and thermallimiting to provide protection againstfault conditions. The LT1497 bridgesthe performance between the 30mAoutput, 100MHz LT1229 and the250mA output, 60MHz LT1207 dualcurrent feedback amplifiers.

The LT1497 is available in a lowthermal resistance SO-16 packagefor operation with supplies up to ±15V.For operation at ±5V and below, thedevice is also available in a low ther-mal resistance SO-8 package.

The LT1634: Micropower,Precision Voltage ReferenceThe LT1634 is LTC’s newest micro-power, precision, shunt voltagereference. The LT1634 providesexcellent precision and drift perfor-mance at micropower bias currents.A low operating current of 10µA, guar-anteed temperature drift of 25ppm/˚C,tight initial voltage accuracy of 0.05%are some of the virtues of the LT1634.The LT1634 comes in four popularvoltages: 1.25V, 2.50V, 4.096V and5.0V.

The bandgap reference usestrimmed precision thin-film resistorsto achieve 0.05% initial voltage accu-racy. Advances in design, processingand packaging guarantee 10µA op-eration and low temperature-cycling

hysteresis. Improved curvature cor-rection techniques guarantee 25ppmmaximum temperature drift. Boardspace is minimized with surfacemount SO-8 and MSOP and through-hole TO-92 (Z) packages. To furtherreduce board space, the LT1634 re-quires no output compensationcapacitor: it is stable with any capaci-tive load over the full operating currentrange of 10µA to 20mA.

The LT1634 is a high performanceupgrade to the LM185/LM385,LT1004 and LT1034. The performanceof the LT1634 makes it the ideal choicefor applications where high precisionand minimum power consumptionare important. LT1634 is available inthe commercial grade.

The LTC1659: Ultrasmall,8-Lead MSOP Packaged, LowPower, 12-Bit MultiplyingVoltage-Output DACThe latest addition to LTC’s family of12-bit voltage-output DACs is themultiplying LTC1659. Packaged inan 8-lead MSOP, the LTC1659 oper-ates on a single supply of 2.7V to5.5V. The LTC1659 requires an ex-ternal reference voltage source. Thegain between the reference pin andthe output is VIN(CODE/4096). Forapplications that require rail-to-railoutput, connect VCC to the referencepin for an output swing from GND toVCC. The LTC1659 features a DOUTpin that allows data daisy-chaining,and a simple 3-line (CLK, DIN, CS/LD)serial interface, allowing easy con-nection to microcontrollers andmicroprocessors.

The LTC1448: Dual LowPower, 12-Bit Multiplying,Voltage-Output DACThe latest addition to LTC’s family ofdual 12-bit voltage-output DACs isthe multiplying LTC1448. Packagedin an SO-8, the LTC1448 operates ona single supply of 2.7V to 5.5V. TheLTC1448 requires an external refer-

ence voltage source. The gain be-tween the reference pin and the outputis VIN(CODE/4096). For applicationsthat require rail-to-rail output,connect VCC to the reference pin foran output swing from GND to VCC.The LTC1448 features a simple 3-line(CLK, DIN, CS/LD) serial interface,allowing easy connection to micro-controllers and microprocessors.

The LTC1412: Low Power,3Msps 12-Bit ADCAdding to LTC’s family of high speed,low power analog-to-digital convert-ers, the LTC1412 samples analoginput signals at up to 3Msps with 12-bit resolution and accuracy. TheLTC1412 operates on a ±5V supplyand dissipates just 150mW at fullconversion speed. The converter hasan analog input range of ±2.5V. It isdesigned for high speed, highresolution signal processing applica-tions, including telecom digital-datatransmission applications, widebandwidth multichannel data acqui-sition and baseband signal recoverythrough undersampling. Its simpleparallel interface and conversion startsignal input make it easy to use inDSP-based designs. The LTC1412includes an internal reference andconversion clock, microprocessor ormicrocontroller compatible 12-bitparallel interface and a fully differen-tial input that achieves better than70dB CMRR over a 0Hz to 3MHz band-width. Data is available immediately,without pipeline delay, at the conclu-sion of a conversion. The LTC1412 isavailable in a space-efficient 28-pinSO package.

The LTC1412’s unique S/H hastwo very beneficial features that ap-ply when sampling either single-endedor differential signals. The first is awide, 40MHz full-power input band-width. This wide input bandwidthallows the LTC1412 to undersamplesignals far above the converter’sNyquist frequency and preserve theirfidelity. The second is a common mode

New Device Cameos


NEW DEVICE CAMEOS

rejection of 60dB. When samplingdifferentially, the common moderejection is especially useful forsuppressing the perturbing effects ofcommon mode noise and groundloops.

The LTC1412 combines excellentdynamic performance with a highspeed conversion rate. The signal-to-noise + distortion (SINAD) is 72dBand the total harmonic distortion(THD) is –84dB when sampling a1.5MHz full-scale input signal at3Msps.

The LTC1414: Low Power,2.2Msps 14-Bit ADCFollowing on the success of theLTC1419, the world’s cleanest 14-bitADC, LTC has just released theLTC1414. This new high speed, lowpower analog-to-digital convertersamples analog input signals at up to

2.2Msps with 14-bit resolution andaccuracy.

The LTC1414 operates on a ±5Vsupply and dissipates just 150mW atfull conversion speed. The converterhas an analog input range of ±2.5V. Itis designed for high speed, highresolution applications, includingtelecom digital-data transmission,DSP-based signal processing,wide bandwidth multichannel dataacquisition and baseband signalrecovery through undersampling. TheLTC1414 includes an internalreference, a fast 14-bit parallel inter-face, and a fully differential inputS/H. Conversion results are availablewithout a pipeline delay. The LTC1414is available in a space-efficient 28-pin narrow SSOP package.

The LTC1414’s S/H has a wide,20MHz full power input bandwidth.This a l lows the LTC1414 to

undersample signals far above theconverter’s Nyquist frequency andpreserve their fidelity. Its input com-mon mode rejection is better than70dB over a 0Hz to 3MHz bandwidth.This is especially useful for sup-pressing the deleterious effects ofcommon mode noise and ground loopswhen sampling differentially. Thesefeatures of the LTC1414’s unique S/Happly when sampling either single-ended and differential signals.

The LTC1414’s high speed conver-sion rate does not sacrifice dynamicperformance. The signal-to-noise +distortion (SINAD) is 80.5dB and thespurious-free dynamic range is 95dBwith a 100kHz input signal. This spu-rious-free dynamic range is anincrease of 10dB over 12-bit devices.The SINAD is 78dB at fS = 2.2Mspsand fIN = 1.1MHz.



www.linear-tech.com


For further information on anyof the devices mentioned in thisissue of Linear Technology, usethe reader service card or callthe LTC literature servicenumber:

1-800-4-LINEAR

Ask for the pertinent data sheetsand Application Notes.


DESIGN TOOLS

Applications on DiskNoise Disk — This IBM-PC (or compatible) programallows the user to calculate circuit noise using LTC opamps, determine the best LTC op amp for a low noiseapplication, display the noise data for LTC op amps,calculate resistor noise and calculate noise using specsfor any op amp. Available at no charge

SPICE Macromodel Disk — This IBM-PC (or compat-ible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be usedwith any version of SPICE for general analog circuitsimulations. The diskette also contains working circuitexamples using the models and a demonstration copyof PSPICE™ by MicroSim. Available at no charge

SwitcherCAD™ — The SwitcherCAD program is a pow-erful PC software tool that aids in the design andoptimization of switching regulators. The program cancut days off the design cycle by selecting topologies,calculating operating points and specifying compo-nent values and manufacturer’s part numbers. 144page manual included. $20.00

SwitcherCAD supports the following parts: LT1070series: LT1070, LT1071, LT1072, LT1074 and LT1076.LT1082. LT1170 series: LT1170, LT1171, LT1172 andLT1176. It also supports: LT1268, LT1269 and LT1507.LT1270 series: LT1270 and LT1271. LT1371 series:LT1371, LT1372, LT1373, LT1375, LT1376 andLT1377.

Micropower SwitcherCAD™ — The MicropowerSCADprogram is a powerful tool for designing DC/DC con-verters based on Linear Technology’s micropowerswitching regulator ICs. Given basic design param-eters, MicropowerSCAD selects a circuit topology andoffers you a selection of appropriate Linear Technologyswitching regulator ICs. MicropowerSCAD also per-forms circuit simulations to select the other componentswhich surround the DC/DC converter. In the case of abattery supply, MicropowerSCAD can perform a bat-tery life simulation. 44 page manual included.

$20.00

MicropowerSCAD supports the following LTC micro-power DC/DC converters: LT1073, LT1107, LT1108,LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,LT1300, LT1301 and LT1303.

Technical Books1990 Linear Databook, Vol I —This 1440 page collec-tion of data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conver-sion and interface products (bipolar and CMOS), inboth commercial and military grades. The catalogfeatures well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 pagesupplement to the 1990 Linear Databook is a collectionof all products introduced in 1991 and 1992. Thecatalog contains full data sheets for over 140 devices.The 1992 Linear Databook, Vol II is a companion to the1990 Linear Databook, which should not be discarded.

$10.00

1994 Linear Databook, Vol III —This 1826 pagesupplement to the 1990 and 1992 Linear Databooks isa collection of all products introduced since 1992. Atotal of 152 product data sheets are included withupdated selection guides. The 1994 Linear DatabookVol III is a companion to the 1990 and 1992 LinearDatabooks, which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 pagesupplement to the 1990, 1992 and 1994 Linear Da-tabooks is a collection of all products introduced since1994. A total of 80 product data sheets are includedwith updated selection guides. The 1995 Linear Data-book Vol IV is a companion to the 1990, 1992 and 1994Linear Databooks, which should not be discarded.

$10.00

1996 Linear Databook, Vol V —This 1152 page supple-ment to the 1990, 1992, 1994 and 1995 LinearDatabooks is a collection of all products introducedsince 1995. A total of 65 product data sheets areincluded with updated selection guides. The 1996Linear Databook Vol V is a companion to the 1990,1992, 1994 and 1995 Linear Databooks, which shouldnot be discarded. $10.00

1990 Linear Applications Handbook, Volume I —928 pages full of application ideas covered in depth by40 Application Notes and 33 Design Notes. This cata-log covers a broad range of “real world” linear circuitry.In addition to detailed, systems-oriented circuits, thishandbook contains broad tutorial content togetherwith liberal use of schematics and scope photography.A special feature in this edition includes a 22-pagesection on SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —Continues the stream of “real world” linear circuitryinitiated by the 1990 Handbook. Similar in scope to the1990 edition, the new book covers Application Notes40 through 54 and Design Notes 33 through 69.References and articles from non-LTC publicationsthat we have found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —This 976 page handbook maintains the practical outlookand tutorial nature of previous efforts, while broaden-ing topic selection. This new book includes ApplicationNotes 55 through 69 and Design Notes 70 through144. Subjects include switching regulators, measure-ment and control circuits, filters, video designs,interface, data converters, power products, batterychargers and CCFL inverters. An extensive subjectindex references circuits in LTC data sheets, designnotes, application notes and Linear Technology maga-zines. $20.00

Interface Product Handbook — This 424 page hand-book features LTC’s complete line of line driver andreceiver products for RS232, RS485, RS423, RS422,V.35 and AppleTalk® applications. Linear’s particularexpertise in this area involves low power consumption,high numbers of drivers and receivers in one package,mixed RS232 and RS485 devices, 10kV ESD protec-tion of RS232 devices and surface mount packages.

Available at no charge

Power Solutions Brochure — This 84 page collectionof circuits contains real-life solutions for commonpower supply design problems. There are over 88circuits, including descriptions, graphs and perfor-mance specifications. Topics covered include batterychargers, PCMCIA power management, microproces-sor power supplies, portable equipment power supplies,micropower DC/DC, step-up and step-down switchingregulators, off-line switching regulators, linear regula-tors and switched capacitor conversion.


High Speed Amplifier Solutions Brochure —This 72 page collection of circuits contains real-lifesolutions for problems that require high speedamplifiers. There are 82 circuits including descrip-tions, graphs and performance specifications. Topicscovered include basic amplifiers, video-related appli-cations circuits, instrumentation, DAC and photodiodeamplifiers, filters, variable gain, oscillators and currentsources and other unusual application circuits.


Data Conversion Solutions Brochure — This 52 pagecollection of data conversion circuits, products andselection guides serves as excellent reference for thedata acquisition system designer. Over 60 productsare showcased, solving problems in low power, smallsize and high performance data conversion applica-tions—with performance graphs and specifications.Topics covered include ADCs, DACs, voltage refer-ences and analog multiplexers. A complete glossarydefines data conversion specifications; a list of se-lected application and design notes is also included.


Telecommunications Solutions Brochure — This 72page collection of circuits, new products and selectionguides covers a wide variety of products targeted forthe telecommunications industry. Circuits solving reallife problems are shown for central office switching,cellular phone, base station and other telecom applica-tions. New products introduced include high speedamplifiers, A/D converters, power products, interfacetransceivers and filters. Reference material includes atelecommunications glossary, serial interface stan-dards, protocol information and a complete list of keyapplication notes and design notes.


continued on page 40

DESIGN TOOLS

Acrobat is a trademark of Adobe Systems, Inc. AppleTalkis a registered trademark of Apple Computer, Inc. PSPICEis a trademark of MicroSim Corp.

Information furnished by Linear Technology Corporationis believed to be accurate and reliable. However, LinearTechnology makes no representation that the circuitsdescribed herein will not infringe on existing patent rights.

Linear Technology Magazine • August 1997© 1997 Linear Technology Corporation/Printed in U.S.A./

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7417(408) 432-1900 FAX (408) 434-0507www.linear-tech.comFor Literature Only: 1-800-4-LINEAR

InternationalSales OfficesFRANCELinear Technology S.A.R.L.Immeuble “Le Quartz”58 Chemin de la Justice92290 Chatenay MalabryFrancePhone: 33-1-41079555FAX: 33-1-46314613

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JAPANLinear Technology KK5F NAO Bldg.1-14 Shin-Ogawa-cho Shinjuku-kuTokyo, 162JapanPhone: 81-3-3267-7891FAX: 81-3-3267-8510

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SINGAPORELinear Technology Pte. Ltd.507 Yishun Industrial Park ASingapore 2776Phone: 65-753-2692FAX: 65-754-4113

SWEDENLinear Technology ABSollentunavägen 63S-191 40 SollentunaSwedenPhone: 46-8-623-1600FAX: 46-8-623-1650

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UNITED KINGDOMLinear Technology (UK) Ltd.The Coliseum, Riverside WayCamberley, Surrey GU15 3YLUnited KingdomPhone: 44-1276-677676FAX: 44-1276-64851

World HeadquartersLinear Technology Corporation1630 McCarthy BoulevardMilpitas, CA 95035-7417Phone: (408) 432-1900FAX: (408) 434-0507

U.S. AreaSales OfficesNORTHEAST REGIONLinear Technology Corporation3220 Tillman Drive, Suite 120Bensalem, PA 19020Phone: (215) 638-9667FAX: (215) 638-9764

Linear Technology Corporation266 Lowell St., Suite B-8Wilmington, MA 01887Phone: (508) 658-3881FAX: (508) 658-2701

NORTHWEST REGIONLinear Technology Corporation1900 McCarthy Blvd., Suite 205Milpitas, CA 95035Phone: (408) 428-2050FAX: (408) 432-6331

SOUTHEAST REGIONLinear Technology Corporation17000 Dallas ParkwaySuite 219Dallas, TX 75248Phone: (214) 733-3071FAX: (214) 380-5138

Linear Technology Corporation5510 Six Forks RoadSuite 102Raleigh, NC 27609Phone: (919) 870-5106FAX: (919) 870-8831

CENTRAL REGIONLinear Technology CorporationChesapeake Square229 Mitchell Court, Suite A-25Addison, IL 60101Phone: (708) 620-6910FAX: (708) 620-6977

SOUTHWEST REGIONLinear Technology Corporation21243 Ventura Blvd.Suite 227Woodland Hills, CA 91364Phone: (818) 703-0835FAX: (818) 703-0517

Linear Technology Corporation15375 Barranca ParkwaySuite A-211Irvine, CA 92718Phone: (714) 453-4650FAX: (714) 453-4765

CD-ROMLinearView — LinearView™ CD-ROM version 2.0 isLinear Technology’s latest interactive CD-ROM. It al-lows you to instantly access thousands of pages ofproduct and applications information, covering LinearTechnology’s complete line of high performance ana-log products, with easy-to-use search tools.

The LinearView CD-ROM includes the complete prod-uct specifications from Linear Technology’s Databooklibrary (Volumes I–V) and the complete ApplicationsHandbook collection (Volumes I–III). Our extensivecollection of Design Notes and the complete collectionof Linear Technology magazine are also included.

A powerful search engine built into the LinearView CD-ROM enables you to select parts by various criteria,such as device parameters, keywords or part numbers.All product categories are represented: data conver-sion, references, amplifiers, power products, filtersand interface circuits. Up-to-date versions of LinearTechnology’s software design tools, SwitcherCAD,Micropower SwitcherCAD, FilterCAD, Noise Disk andSpice Macromodel library, are also included. Every-thing you need to know about Linear Technology’sproducts and applications is readily accessible viaLinearView. LinearView 2.0 runs under Windows® 3.1,Windows 95 and Macintosh® System 7.0 or later.

Available at no charge.

World Wide Web SiteLinear Technology Corporation’s customers can nowquickly and conveniently find and retrieve the latesttechnical information covering the Company’s prod-ucts on LTC’s new internet web site. Located atwww.linear-tech.com, this site allows anyone withinternet access and a web browser to search throughall of LTC’s technical publications, including data sheets,application notes, design notes, Linear Technologymagazine issues and other LTC publications, to findinformation on LTC parts and applications circuits.Other areas within the site include help, news andinformation about Linear Technology and its salesoffices.

Other web sites usually require the visitor to downloadlarge document files to see if they contain the desiredinformation. This is cumbersome and inconvenient. Tosave you time and ensure that you receive the correctinformation the first time, the first page of each datasheet, application note and Linear Technology maga-zine is recreated in a fast, download-friendly format.This allows you to determine whether the document iswhat you need, before downloading the entire file.

The site is searchable by criteria such as part numbers,functions, topics and applications. The search is per-formed on a user-defined combination of data sheets,application notes, design notes and Linear Technologymagazine articles. Any data sheet, application note,design note or magazine article can be downloaded orfaxed back. (Files are downloaded in Adobe Acrobat™PDF format; you will need a copy of Acrobat Reader toview or print them. The site includes a link from whichyou can download this program.)

DESIGN TOOLS, continued from page 39

Acrobat is a trademark of Adobe Systems, Inc.; Windows isa registered trademark of Microsoft Corp.; Macintosh is aregistered trademark of Apple Computer, Inc.

40K

datasheet

Documents

portable communications equipmentthe

cin 47f 16v

2n 1 ri ci

updated selection guides

cout 100f 10v

baseband signal recovery

ltc1340 varactor driver

unwanted frequency modulation