deep-ocean tests of an acoustic modem insensitive to multipath distortion

8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion

1/8

Deep-Ocean

Tests

of

an

Acoustic

Modem

Insensitive to Mult1path Distortion

Winfield Hill, Gerald Chaplin, David Nergaard

Sea

Data,

Inc., A Pacer Systems Company

1 Bridge

Street, Newton, Massachusetts,

02158

concept,

design

and ocean

testing of a new low-power

modem

is

presented. The telemetry

system

em

a novel chirp frequency sweep and has

other

fea

allow

operation

in the presence of multi

path

inter

The chirp system uses fsk data modulation and

a carrier sweep starting at 9 or 31kHz, depend

upon the model, to obtain the benefits of frequency

without requiring a frequency synthesizer, multi

or

a

FFT

analyzer.

Intended

for

retrofittable

to

instruments,

the

new

system

is designed for use in

deep

ocean

and

the continental

shelf over dis tances

to

were performed in

about

4000

meters of

using

the

low frequency version. Additional shalIow

are

planned,

including a typical harbor.

acoustic

telemetry

diversity modem

Introduction

multipath

chirp

of data is a commonly desired capa

is not commonly available in undersea oceano

instruments. Although convenience and peace-of

occaisional motivations for these desires, strong

have been

made

for

the

value of this capability(1).

include use in

real-time

operational systems, multi

deployments

(where

it s

impractical

to

wait until the

for the data),

performance

monitoring, repair flexibil

and

expendible instrumentation.

We

report

here on

the

and

initial ocean tests of

a new chirp acoustic teleme

method, which

has

simplicity and reliability properties

for fitting acoustic telemetry data links to existing

designs.

Background

- Ocean Acoustic Telemetry

travelling a substantial distance in

the

suffers from severe amplitude fluctuations

and

phase

Acoustic temporal incoherence may by caused

by multiple

sound

pathways,

bottom

and surface scattering

and moving inhomogeneities in

the

ocean(2). However the

repeated observation

of

such

degradation

has obscured the

fact that sound transmission quality over direct vertical or

slanted pathways (other than in a sound channel) may be

quite

good(3).

Kearney

and Laufer(4)

demonstrated

this

point

while de

livering a paper at Oceans '84, by playing a

cassette

tape

recording

of

voice and music transmitted from 1500 me

ters

depth

to

a

shipboard

recorder; my

memory

is

that

the

primary degradation was due to the use of a very poor cas

sette recorder. Designers of acoustic high-resolution pos

tioning systems have long

taken

advantage of good direct

transmission paths by detecting

the

arrival of

short acou;tic

pulses using narrow-band Q

>

30) filters(5). When used

at

low frequencies (10kHz), these positioning systems re

quire several milliseconds

of

phase coherence in

the

leading

edge

of the

pulse.

Short-range «

400m) acoustic

telemetry

systems have been constructed(6,7) using simple frequency

shift keying (fsk) modulation in the expectation of a reliable

acoustic path, with some success. One system transmits at

the

very slow

rate

of 1 bit-per-second(8) to achieve

up to

a

1000m range.

2.1

Multipath. Despite the good quality of a direct path

or

reliable-acoustic-path signal transmission channel, most

practical underwater acoustic systems must contend with

strong undesired signals

scattered

from the surface or the

bottom. This is especially true when one of the acoustic

transducers is near

the

ocean surface. Superimposing the

surface-scattered signals

upon the direct-path

signal causes

fading and phase instabilities, possibly including

complete

cancellation of the

desired signal for a few milliseconds from

destructive

interference. Therefore, a pulsed one- or two

frequency signal, which begins with

good receive quality,

deteriorates as multi

path

interference arrives.

As

an

example of simple two-frequency fsk

telemetry

per

formance

when

surface and bottom scatter have a strong

influence, consider the experiences

of

Ryerson

at

Sandia

Labs(6). Transmission from a 10m subsurface buoy

with

a

slant

range of 180 to 280 meters to a surface buoy was

desired. Water depth was 200 meters. Optimum perfor

mance was obtained only after a variety of system-tuning

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2/8

were made.

Operating

frequencies were selected

near

50kHz)

to

reduce

transducer

backside

and

side-lobe

esponse

and to attenuate

long, multiple-reflection paths.

n mid-experiment, the

receive

transducer depth

was in

reased by one meter. Also, lower error rates were achieved

with

a -12dB power change (0.6

watts

instead of the design

level of 10

watts).

An 85 to 90% success

rate

was achieved.

I we

-

h

•

1

2.2 Surface Multipath.

A common surface-path

sit

uation is illustrated

in figure

1.

The

offending surface

scattered

backside arrival) signals clearly travel a longer

path

than the direct

path

signal and therefore take a longer

time to

arrive.

The

earliest-arriving

scattered

signals take

an

extra

delay

time

Td)

to

travel

an extra path

delay

P

d

)

as follows:

1)

d ..; h d

P

d

= B h

2

[ h - d) tanA - dtanBj2

os cos A

2a)

where d is

the

receive

transducer depth,

h

is the transmitter

depth,

A

is

the transmitting

slant

angle, B

is the scattered

signal

receive

angle

both

angles are measured from

the

vertical) and e is the speed of sound in seawater,

about

1.5m/ms. The surface

watch-circle radius

we)

is

related

to the slant angle

by

we = h -

d)

tan A.

If

the

watch circle

radius is known instead

of

the slant

angle

A, equation

(2a)

can be written:

P

d

= _ d _ Jh2 we

- dtan B)2

- J h -

d)2

wc

2

cosB

2b)

When

the

transmitter is

straight

below

A =

0 and

we

=

0), equations (2)

above

simplify to:

P

d

= _ d _

.

Ih2 dtanBp - h - d)

2c)

cos

BY

The

first delayed

surface-scatter multipath

arrival occurs

at Td

=

2d/e, when

the

arrival angle B

=

0 (surface angle

= 90°), followed by

more

sound arriving for B

>

O

The

first arrival delay

is about

27ms for a receive hydrophone

depth

of 20m.

For

slanted sound paths A >

0) equation (2a) shows

that

the surface-scattered

first-arrival delay time is slightly faster

than

for the direct overhead case; the shortest

path

occurs

for equal angles of incidence and

scatter

at

the

surface. As

an example,

for a transmitter in 3000m of water,

to

a

60m

deep hydrophone

at

a 2000m watch circle

distance, A = 34°

so the surface

incident

angle (given by

90°-A) is

about

55°.

Sound

scattered

at 55°

from

the surface

B

= 35°) will ar

rive with a 67ms delay compared

to

80ms for

the

straight

below case). Straight-line sound travel

has

been assumed

throughout,

even

though

for a

slanted

direct

path

sound

travel

is actually slightly

curved, due

to

refraction by

the

sound-speed

depth

profile;

This

does

not

affect

our

conclu

sions.

After the first

multipath

arrival, sound travelling longer

paths continues to arrive for a substantial period of

time;

this additional sound

constitutes most

of the

multipath

in

terference. Some of

the sound has

travelled very complex

pathways, involving

volume scatter

as well.

Although the

multipath

signals suffer surface-scattering losses( 6)

of

10

to

20dB, the beneficial effects of these losses are reduced by

,he

large

area of the

surface.

The

desirable losses are fur

ther

reduced during high sea-state conditions, when acous

tic surface

scattering

increases (e.g. see

the

backscattering

curves in ref 15 p. 264). However, for frequencies

above

20kHz (e.g.

33kHz), wind

velocities above 10

to

15

m/s

may actually cause reduced surface-scatter sound due

to

sound

attenuation

by small-bubble p o p u l t i o ~ in

the top

5 meters of

the

ocean( 7).

2.3 Fighting Multipath.

Several

methods

have

been

sug

gested

to reduce

signal

degradation

by

multipath

interfer

ence. One

is

to

use a

transducer with

high

back rejection

(or use a baffle).

In

the

3000m

example

above,

the

first

offending sound arrived at angle of 145.

0

from

the

trans

ducer forward direction (given by 180°

- A,

assuming the

transducer is pointed

down). A second

method

is

to

cre

ate a highly-directive receive transducer array(18). These

approaches increase

the

cost of

the

system, are painful

to

implement at low frequencies

and

have

limited

utility for a

variety of reasons. Furthermore, in shallow

water, directive

sensors may not be very helpful.

276

Acoustic transmission in shallow

water s

much more dif

ficult

than

in deep water, since it suffers from the exis

tance of many strong sound pathways to the destination,


3/8

numbers of surface

and bottom

reflec

Computer modelling(T,18) indicates

that

for 10kHz

in 200m deep water,

the Direct-to-Multipath

Ratio

(DMR)

may be

as

poor 88

6

dB at

ranges

of

than lOOOm. Actual measurements in

the

ocean may

poorer DMR. Higher-frequency transmissions will

sea-water absorption attenuation for

longer

multiple-bounce

pathways,

but

less

than 5dB of

is

calculated

at

50kHz,

due to this

effect.

sound transmission

methods

have been

to solve

the multipath

problem. Systems

many frequency channels(l1,12,IS) have been

proposed,

up to

32 frequencies(H), so

that the system can

switch

a new frequency before the multipath interference ar

In a common approach, the frequencies in use

are

every 50 to lOOms, allowing

the

multipath

energy

decay on the old channel. Since

the

decay

time

allowed

can be reused is proportional to the num

of available frequency channels, this may well be a true

more is

better .

Of course

the telemetry system

more

complex,

but

the improved results

that

be

obtained

in all environments are very

attractive.

f O ~

lot-to ~ I I I

1-

2

A

New Chirp Telemetry

Method

new Sea

Data chirp

acoustic

telemetry system

is

based

a variation

of

the

frequency-diversity idea: use

an

in

number of frequencies. This is achieved by sweep

the telemetry carrier

frequency while applying fsk data

(fig 2).

The

transmitted

signal

P t)

is a single

starting at II, and

changing

at

a

smooth rate

/dt, plus fsk frequency shifts with amplitude 12:

P t)

=

cos[w t)t]

3)

w t)

=

211 11

t t

12

M

t»)

4)

dl

=

s

-

dt

Ts

5)

where Ts is

the duration

of

the

sweep and 11

and

Is

are

the starting and

ending sweep frequency

and M

t)

=

0

or

I according

to the

data

bits. The

modulation ampli

tude, 12

is chosen large enough, e.g.

>

150Hz,

to

eliminate

doppler-shift spreading problems, which will be less than

40Hz (O.33Hz/kt per kHz). .

If the

receive frequency

is

accurately swept

to match the

transmitter,

a small receive

bandwidth (constrained by the

data

rate and

the

fsk 0,1 frequency shift) can be used,

just

as in a conventional fsk system. A small bandwidth will im

prove

the

signal-to-noise

ratio

(SNR)

not

only by rejecting

ambient noise

but

also by rejecting

the

(delayed)

multipath

energy from

the

old-channel frequencies.

n

a chirp

telemetry

system, the effective frequency-diversity

channel usage time Tu)

can

be equal to

the

time required

for

the carrier

frequency sweep to change by

more than the

receive

bandwidth

BW),

as follows:

Ts

Tu BW

3 - I

6)

The

usage time

can

be easily

set at

under 50ms (e.g. BW

= ~ O O H z

sweep 4000Hz in 650ms), allowing excellent re

jection of multipath signals.

As

an

added benefit,

the

new

chirp telemetry approach

can

be inexpensive, compared to other frequency-diversity

methods,

since multiple frequencies are not required (i.e.

no synthesizer)

and

receive decoding

can be

simplified (i.e

o multiple

filters

or FFT

analyzer). To

understand our

approach and

the

role

of

all

the

elements in

the

sweep wave

form

of

figure 2, we'll

start

by considering how

the

receiver

works (see figure 3).

277

3.1

Signal

Description.

Because a

telemetry

receiver

contains many circuit elements

that

consume electrical power,

it's desirable to switch

the

power to a

portion of these

cir

cuits.

n the

receiver design above, a

number

of

components

are continuously

powered

in order

to

detect the

arrival

of

an alert

signal.

These

are

the

preamp, AI,

a

bandpass

fil

ter, BP, the 10 detector and a power-control

circuit

(for

our

experiment,

10

=

9.0kHz). The

bandpass

filter is designed

to pass signals over

the

entire 10 to

f3 range

of

the

system

(a

double-

or

triple-tuned filter)

and to

reject

intense

low

frequency noise from shipping, etc

(extra LF

cutoffs).

The

fa energy detector operates on a principle similar to that

used by

many

high-resolution acoustic positioning

systems

(5):

an amplitude

limiter (to

establish constant power),

a

sharp fo

filter

and

a

comparator with

a

time constant,

work

together

to determine if the fo energy present is above

the

background noise adjacent to fo by a

threshold

amount.

When

the

10 alert-tone energy

is detected, the remainder

of the

receiver, including

the

microprocessor,

is turned

on.

After a short time,

to, the

transmitter shifts its frequency to

(for

our test

11

=

10 600Hz),

creating

the data-trigger


4/8

tone. This trigger tone start pulse is detected by the

11 energy detector (similar to

the

1 energy detector)

and

is

used to start

a sweep

generator

and voltage-controlled

oscillator (VeO). The resulting frequency ramp is designed

to

precisely

track the transmitter's

sweep

with

a fixed offset,

hr where frr

>

13 - 10)/2 to avoid images.

This

ramp

frequency

is

the local oscillator (LO) input to a mixer, and

has a frequency, lLO , similar to equations (3) to (5)

except

as follows:

dl

fLO = f t) =

I i dt

t IIr

7)

The resulting

intermediate

frequency (IF) output from the

mixer after IF-stage

filtering is:

v t)

=

sin[27rI t)t] n t)

8)

f t) = fIr 12M t)

9)

where n t) is the received noise, with a noise bandwidth

given by the IF-stage bandpass. This signal is limited and

applied to a frequency

discriminator

to

track 12 and

deter

mine whether M

=

0 or

1.

A

data

precursor time delay, t

in fig 2, allows the circuits

to

settle before data discrimina

tion must start. Also a warmup time, tw = to less the

1

detect

time,

is available

forthe

crystal

in

the

receiver's

microprocessor

to

start, etc.

The

frequency discriminator in fig 3

is

a phase-locked loop

(PLL) circuit, which forms a tracking filter

to further nar

row the noise

bandwidth

of

the

receiver.

The

input

stage

of the PLL

is

a

limiter that

responds to the

strongest

sig

nal within

the

IF bandpass and acts

to

reject any weaker

signals,

thereby

further rejecting (quieting)

unwanted

mul

tipath signals . A full-wave mixer phase-detector circuit

(exclusive-OR) and the PLL loop filter act in a

+5

BP

\

FO

DETECTOR

POWER

CONTROL

SWEEP

GEN

+5

=

SWITCHEC

POWER

3

manner

to

maintain the v o

output

frequency

near

wet)

in formula (9). Further filtering of the

input

to

the v o

-

a varying dc voltage

vet) ex wet)

- along

with

ac-coupling

and clamping, yields the original data-stream signal, M t).

The

PLL loop filter and low-pass filtering of M t)

set

the

noise bandwidth,

BW,

of the

telemetry

receiver. The re

ceiver

should be

able

to operate

with very low SNRs, al

though

the

data error

rate

may not then be zew.

4.

Transmitter

The transmitter (figure 4) helps illustrate the

simplicity

of the chirp telemetry scheme. A few low-data-rate con

trollable

outputs

from the instrument's microprocessor

are

sufficient

to

operate the transmitter. These outputs include

the sweep

generator power

control,

an

enable for

the output

driver as soon as

the

v o

is stable,

a

start

pulse (SP) shift

ing

the

frequency for a data trigger, a sweep

enable

(SE)

and

the

data bit (DB)

modulation

signal.

Another

line

sets

the sweep rate (SR) to allow

optimizating the

system for

deepsea or

shallow-water

use.

BATTERY

LI

SYSTEM

,p

SP SWEEP VB0

I . : . . . . = ~ = ~ = M = O = D G UE L = A ~ = O = R = - _ - = : ? r n c ~ _ - , rm

R

ENABLE TI

4

278


5/8

At low frequencies, e.g. 10kHz, obtaining a transmit op

erating range of

5kHz

is

a challenge, due

to the

narrow

band nature

of a

tuned

acoustic

transducer. In

figure 4,

the

reactive component of

transducer

Xl is removed us

ing tuning coil L1, with a series resistor R1 to increase the

frequency range.

In the

9

to

14kHz experiment

to

be de

scribed, a modified ITC

type

3013

transducer

(which nor

mally

has

transmit-voltage-response

peaks at

9 and 14kHz)

was used with a 22mHy choke

and

a 50 ohm

damping

resis

tor.

A very

acceptable

calculated

network

output

flatness

(+ 140±2dB/V) was obtained over a 7.5 to 14kHz range,

and verified

with

pulsed measurements in

the

local

YMCA

swimming pool. If necessary, a more complex network could

be devised. When operating

the

system with 5kHz sweeps

at 33kHz, using a custom-designed

transducer,

a

damping

resistor

is

less important.

4.1 Power.

In

the

deep ocean

test, the output

stage con

sisted of a pair of VMOS transis tors driving a

center-tapped

transformer

with

a regulated 12V input.

This

provided

about 20 watts of power into the transducer network and

yielded a

modest

calculated source level of + 179dB re l/LPa

at

1m, confirmed

in

the

pool test.

The current drain

from

the instrument battery was less than 2A during transmis

sion, a very acceptable level for any instrum.nt with several

,tacks of alkaline

batteries.

Although lower power levels may be used in practise, our

thought

was

to

get good quality eata on

the experiment

OAT

tape and

subsequently degrade it

with

noise when we

tested transmit codes and receiver designs in the lab. How

~ v r the

higher-power energy usage

is not unattractive:

At

300 baud, less than 0.1 Joules per bit is required, including

alert tone,

etc. Since a single stack of alkaline D-cells con

tains about 0.5MJ of

energy,

it could

power

about

100,000

transmissions of 50-bit data blocks.

SR

IK

R4

1

R5

o R6 R7 2V

2 K 165K 1

R3

R8

4.99K

rl

1

RO

RI

R2

4.2

Design Simplicity. Because I always miss

the

ab

sence of electronic-circuit schematics at IEEE conferences,

I ll be sure to include one here. Figure 5 shows details of

the transmitter

sweep

generator and

serves

to further

illus

trate

the

simplicity of our new approach, while giving me

a chance to dispel any concerns over drifts, tuning, etc.

The most

important

component is the voltage-controlled

oscillator (VCO) chip U4,

an

Analog Devices AD537, which

operates

at

twice

the transmitter output

frequency.

This

VCO chip creates a very stable frequency and has low

power-supply and temperature drift coefficients (0.01 /volt

and 0.03 /10

degrees C).

When

used with

stable

compo

nents (capacitor

Cl and

resistor RIO are low-tc compo

nents), the

AD537 may allow a circuit with lifetime factory

calibration. The

VCO follows

the

formula I = Vs/[1O(R9

+

RlO)Cl].

Here R9 sets

the

exact coefficient for

the

VCO

frequency-programming voltage, V

3

which comes from am

plifier A3 (LMIO, chosen for ImA sink capability when

V

ou

=

0.2V

at

the end

of

the

sweep).

This

amplifier s

summing junction allows the telemetry system

operating

parameters to

be exactly ratiometrically determined by pre

cision resistors RO, Rl

and

R2 according

to the

following

formula:

(10)

where

Fo

=

I/RO

sets

the 1

alert frequency,

1

= I /Rl

sets

the

11 - 1 data trigger frequency shift,

2

= I/R2

sets

the h fsk

modulation

level and

s

=

1/R3

sets the

sweep

rate

(and hence Is). Amplifier

Al

(OP-20, chosen

for low offset voltage) creates a reference I-volt above

the

amplifier-reference signal (also 1 volt), so

that

k

= R8.

In

the

experiment, an electronic switch selected two values of

R2

to

allow two fsk

modulation

levels.

Amplifier A2 (OP-90, chosen for low input

current

and off

set voltage) is a ramp, which operates (when switch SE

opens) with

an integration

constant =(R4+R5)C2.

The

integrator uses voltage source trim R6

to

allow two cali

brated sweep rates according to the resistor ratio

R4/R5

CI

2f

5

1 E t

VR

279


6/8

IIwitch

SR. The

sweep

generator operates on

5.0 volts

supply,

a.nd

the entire circuit requires only two

sim

calibration points, yet

we're able

to

get our infinite

channels.

. Signal Propagat ion

Loss,

Noise, S N R

expected

signal-to-ambient-noise

ratio

(SNR)

can be

.

alculated

(in

dB) by subtracting the speading

and

attenu-

losses and the background noise level from the- trans

source level:

SNR =

SL -

20Iog(r)

-

r

_

NSL

- lOlog BW) 11)

1000

SL

is the

source level (dB

re

1tLPa

at

1

yard),

cor

for

the transducer

directivity

index, r

is the range

-

not

km), a is

the

seawater attenuation coefficient

than 1dB/km

for frequencies below 15kHz), NSL is

the

spectral

level

(dB re

1tLPa/v1fz ),

and BW

the receiver

bandwidth

(Hz).

The

equation assumes a

preamp

and

does not include the

improve

a

directive

receive

transducer

will

provide

in

rejecting

ambient

noise, which could exceed

seawater

attenuation

is due to

magnesium-sulfate ionic

with

an

absorption

coefficient of

about 0.7 and

dB/kyd

(at

10°C

and zero depth)

for 12

arrd

35Hz, re

(see ref. 15,

page

109

and

ref. 22).

Over

the

of 8

to

50kHz,

the absortion

coefficient increases

by

square of the

frequency, decreases

about

7% for

each

of depth

and

increases

about

2% for each °C of tem

decrease. The latter two effects

tend to

cancel

other

out in

the top half

of

the

deep ocean. Applying

formulas

to

expected ocean conditions yields

the

values

which

can be integrated

over

the

sound propaga

to determine the absoption

loss for various

applications.

Depth (m)

Temp OC)

Attenuation (dB/kyd)

@10kHz

@33kHz

0 20

0.65

4.S

3000

4

O.SO

3,3

6000

4

0.38

2.4

using

the system in

deep

water, with

a Skm

path,

we

calculate

a

16dB expected SNR

for (poor)

20m/s

wind

as follows: Given

the TVR of

the

transducer

t

+141

dB

per

volt, and considering

a

l.SdB

loss for

the

tuning

resistor,

we can calculate an output

acoustic

of +178dB, for 20 watts

(this

was confirmed in

pool

test).

We lose -74dB from Skm spreading and -

from attenuation

(at

10kHz). The resulting calculated

280

signal strength

of

+99dB

re 1J LPa near

the surface,

is about

16dB

louder than the wind

noise for

20m/s

(NSL

= +S8dB

at 10kHz), assuming a 300Hz receiver

bandwidth (+25dB)

and

an isotropic receive

transducer

(Dr

=

OdB).

5.1

Shallow

water .

Using 33kHz in shallow

water

at

lOoC, 3km of range will result in

about -22dB

of absorption

loss,

assuming the actual (scattered and

reflected) path

is

30% longer

than

the

range. Since

the sound is

in a channel

the spreading

loss

may be

less

than the

-69dB value

r o ~

formula

(11),

say

+lOdB

for

30m water depth

(2).

The

fi-

nal system SNR is similar to the case above since NSL is a

bit lower at 33kHz. Because 3km of range in shallow water

will be subjected

to

severe

multipath

interference,

the

sweep

rate

may increased and

the

data

rate

may

be

decreased to

combat this. Also,

the

telemetry system's processor

can

easily allow using slower

data

rates, with 25ms dead

peri

ods in between

each

bit,

to

allow

the

immediate

multipath

energy to

decay.

6.

Sea Trial using

a

DAT Recorder

The

experiment

was performed on 16

to 17

June

1988

dur

ing cruise

OC200

of

the

WHOI

vessel R.V.

Oceanus, t

a

site approximately

400 miles

east of Cape Hatteras, just

north

of the Gulf Stream, in 3775m of water.

The

undersea

transmitter

for

the

experiment

operated

over a range of 9

to

14.5kHz and was installed in a Sea

Data

model 1665 In

verted Echo Sounder (rES), deployed on

the bottom. The

receive hydrophone was

the standard

EG G

acoustic

re

lease deck-set sensor (an

ITC

3013

transducer),

suspended

over

the

side of

the

ship

about 18m

below

the

surface. A

custom-built preamp

with a 5kHz 2nd-order

bandpass

filter

was used with

the

hydrophone.

To

test the

new telemetry system, we elected

to

transmit

test

signals from

the

ocean

bottom to

various lab-based re

ceiver circuits

via

a

shipboard

precision audio recorder. In

this way we

could

perform receiver tests in the lab with var

ious noise levels and different types of interference, using a

TEST

;

~

TR NSMIT

;; 7 } > I p 1


7/8

large variety

of transmission types

as

they

were

actually

received in

the

ocean. We used a small

portable

16-bit dig

ital audio

tape

(DAT) recorder (Technics model

SV-MDl,

complete

wit.h a

manual entirely

in

Japanese). Thus

we

were able

to obtain

perfect (90dB

dynamic range,

flat

to

18kHz, 0.01% time

stability)

digital analog recordings

of

the received hydrophone signals.

During the experiment the transmitter

variables were cy

cled

through

a

variety

of combinations

using a

parameter

table

in

the

microprocessor's

program.

The parameters

in

cluded: data

rates

(100

to

360

baud),

modulation

index

(300 and 500Hz), chirp sweep rate (10

and

40kHz/s), chan

nel decay quiet times (62ms to 14s), transmit

duration

(1

to

10

bytes/record and

3

to

80 records)

and the transmitted

data patterns.

During

the experiment the

receive variables included

slant

range

and

Dolphin

activity

level.

Winds of

lOkts and a

steady

rainfall

both occurred at

various

times

during the

experiment. The experiment

was

performed with

the

low

frequency version (to = 9kHz), since all

the

available com

ponents

(IES

transmit

stage, receive hydrophone

and

DAT

recorder) weighed

against

the

33kHz version.

Initial oscilloscope examination of

the

DAT tapes shows 3

to

lOdB of

fading after the

sweep

was under

way

(due to

multipath?), -6

to -lOdB of

delayed (obvious) multi

path

interference

and

a + 10 to +20dB

S i ~

(4kHz noise band

width), depending upon surface conditions.

The concept of

using a DAT recording

to

provide receiver

test signals has proven

to be

very useful.

At

this

writing,

excellent

performance has

been

obtained playing back the

tapes

into

our prototype

receiver. In

this

fashion we will

easily be able

to

optimize

the

performance of

the

receiver

design

with tests using bench instruments, e.g. the SNR

can

be

degraded

with

noise

generators.

Already, we were able

to

painlessly

test the

improvement that a CMOS-switch

ana

log mixer provided over a

limiter/XOR-gate

mixer. Further

DAT

recorder ocean experiments are planned

in

shallow

water.

6.1 Dolphins. We experienced considerable interference

from dolphins, who were curious

about the ship and

enjoyed

playing with

the hydrophone.

A few dolphins used

their

variable-rate

pulse

sonar to locate and

"ping"

the

trans

ducer;

at

closest

approach they

increased

the

ping

rate to

buzz. Like

our hydrophone,

the

dolphins

could

hear

the

transmitter on

the ocean bottom.

Amazingly, they did a

good

job of

mimicking

the

9 to 14kHz sweep signal

of the

telemetry But we

haven't yet

decoded

their

transmissions

(Does anyone know, do they use ASCII code? And if so,

is it

Is

b first?).

t

was necessary

to

move

the ship

several

times,

and to turn

off

the

fantail lights. This may

be an

argument in favor of

higher

frequencies, such as

our

33kHz

version.

281

1

System Considerations

The

receiver and

transmitter of

the

acoustic

modem

each

occupy one

card,

as does

the

processor.

The

33kHz trans

ducer is very small, l.6-in (4cm) in diameter,

and

is con

structed with an O-ring

groove

and 3/4-16 stud with

em

bedded

wires,

to

allow it

to be

screwed directly

into an

endcap. Thus, the system can

easily

be added to many

existing designs. A standalone version mounted in a small

housing with

a

battery

is

planned

as well.

The

final

telemetry

system software will employ a data

transmission

protocol suited for systems applications, and

error checking features. A unique code can be sent from

each

transmitter

for identification. Controlled

redundancy

can greatly

reduce

the

error rate: block error-correction

codes

such

as

the

Reed-Solomon code(21) can allow for cor

rection (after reception) of up

to

15 errors within a ISS-bit

block while achieving

an

80% code

rate

(125 data

bits).

Although both

receiver

and transmitter cards

will often

be

located at both ends of

a

system,

creating a full

underwater

MODEM, telemetry

systems

can be substantially

simplified

if one-way data transmission is used. f stable timebase os

cillators are

employed(19), offset time-slot channels

may be

established

so that many

undersea instruments can trans

mit

to

a

central

receiver(20) on a single frequency, without

requiring

a

command

for

the

transmission. Furthermore,

studies

have shown(7) that a one way acoustic data

trans

mission

system

can

be optimal,

e.g. "Analysis of

the

ADTL

data indicated that command and retry

provided only

min

imal improvement in

the amount of

data passed

without

errors.

8. Conclusion

t is

our expectation

that considerable

improvement

over

other

traditional

methods

will

be

experienced with

our

new

swept-frequency telemetry, at a reduced cost.

t

is our hope

that our

work will help lead

to

a

greater and

happier

use

of

acoustic telemetry in

the

ocean.

9. Acknowlegments

One

of

us (GC) wrote some of the transmitter

software

and

singlehandedly

( )

performed

the

undersea experiment,

during the

wee hours when

the rest

of

the ship

was asleep,

while another (DN) modified

the

IES undersea transmitter

and constructed preamps and

prototype

receivers

to ana

lyze

the

DAT

tapes.

Special

thanks are due

to Kevin Boyce

for creating a

major portion

of

the

original IES micropro

cessor code,

to Dan

Frye

and others at WHO

I for

their

suggestions

and

review of

telemetry system

goals

and to

Prof. Randy Watts at URI

for his

encouragement.


8/8

References

Parker,

B.B. (1985) Real-Time Oceanographic Model Systems: Present

and

Oceans

'85,

Proc.

IEEE-MTS

Conf.,

pp W.-214.

Urich,

ILJ.

(1982) Sound

Propagation

in

the

Sea, Peninsula Publishing, Los

chapter 10-12.

Coffey,

D.M.

and

PaquetteiI985): Aaura

deep-ocean tests of an acoustic modem insensitive to multipath distortion

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