Design and Control of a Bi-directional Resonant DC-DC Converter for Automotive Engine/Battery

Hybrid Power Generators

Junsung Park, Minho Kwon and Sewan Choi, IEEE Senior Member Department of Electrical and Information Engineering Seoul National University of Science and Technology

E-mail: schoi@seoultech.ac.kr

Abstract— In this paper a bidirectional DC-DC converter is proposed for automotive engine/battery hybrid power generators. The two-stage bidirectional converter employing a fixed-frequency series loaded resonant converter(SRC) is designed to be capable of operating under zero-current-switching(ZCS) turn on and turn off regardless of voltage and load variation, and hence its magnetic components and EMI filters can be optimized. Also, a new autonomous and seamless bidirectional voltage control method that combines two individual controllers for low voltage side control and high voltage side control by introducing a variable current limiter is proposed to provide uninterrupted power to critical AC loads and reduce the size of the DC bus capacitor. Experimental results from a 5 kW prototype are provided to validate the proposed concept.

Keywords— hybrid power generator, bidirectional DC-DC converter, series loaded resonant converter, zero cur-rent switching, seamless transition

I. INTRODUCTION Standby or emergency generators are often used as backup

power supplies for buildings, industrial facilities, and power plants in the event of a loss of utility power [1]. In addition, remote power generation for military, industrial, and personal use requires a reliable, compact, and lightweight power generation system. The diesel generation system has been used as backup power supplies or remote power generators[2]. Since the engine generator may not be able to respond to sudden load changes, energy storage devices should be used along with the engine generator to level out the erratic changes in power balance between the generation and load consumption[3][4]. Energy storage device is used along with a bidirectional DC-DC converter(BDC) in order to match the voltage level and/or achieve efficient charging and discharging operation[2].

Fig. 1 shows an automotive engine/battery hybrid power generation system. The BDC is located between the high voltage DC bus and the low voltage battery which is also connected to DC loads such as anti-lock brakes, electric power steering, heated seats, electronic ignition and HVAC in the vehicle. The DC-AC inverter converts the DC power to AC power to supply the critical AC load in the vehicle such as

broadcasting equipment of outside broadcast van and communications equipment of tactical vehicle. The AC-DC converter converts the AC power from the engine generator to the DC power, regulating the high voltage DC bus[5]. If the engine generator is capable of supplying total demanded power of AC and DC loads, the AC-DC converter will be able to regulate the high voltage DC bus, and the BDC will deliver the power from the engine generator to the low voltage side. If the engine generator is shut down or total demanded power of the AC and DC loads is greater than the maximum power of the engine generator, high side bus voltage will drop off to a voltage depending on the capacitances of the DC bus capacitor. Then, the BDC is required to take over the regulation duty of the high voltage DC bus by changing over from VL control (battery charging) to VH control (battery discharging) so that it should be able to deliver power from the battery to the AC load. Therefore, in order to provide uninterrupted power to the critical AC loads and reduce the size and cost of the DC bus capacitor, the transition from VL control to VH control of the BDC should be seamless and as short as possible. This is a crucial performance of the BDC, especially, in the automotive application where electrolytic capacitors cannot be used due to limited lifespan and bulky nature[6]-[9]. So far, bidirectional voltage control methods with seamless mode transition of the BDC have not been discussed.

The BDC should provide galvanic isolation and high step up/down voltage conversion ratio in the application where the low voltage battery is used. Typical topology candidates with

Fig. 1. Automotive engine/battery hybrid power generation system

740978-1-4799-0482-2/13/$31.00 ©2013 IEEE

these requirements include half-bridge, full-bridge and push-pull PWM converters [10][11], dual active bridge(DAB) converters [12][13], and resonant converters [14]-[17]. The half-bridge, full-bridge and push-pull PWM converters usually necessitate passive or active clamping on the low voltage side with inductors to clamp the surge voltage generated by the leakage inductance of the transformer. The active clamping technique makes the converter not only clamp the surge voltage, but achieve zero-voltage-switching (ZVS) turn on of all switches. A drawback of the active clamped PWM converter is high switch turn off losses[18]. The DAB has a modular and symmetric structure and can achieve ZVS turn on without auxiliary components. However, the DAB has limited ZVS range and high circulating currents for applications requiring wide voltage variation. The ripple current of the DAB converters is high and especially problematic in the low voltage application[19]. The bidirectional resonant DC-DC converter generally sees different resonant tanks in forward and reverse modes, respectively, resulting in different voltage gains. This often makes it difficult to satisfy the required voltage gain for both modes of operation[14][15]. The dual bridge SRC converter has large current in the resonant tank compared to the DAB converter[16]. A minimum current operation for the dual bridge SRC converter has been proposed with a complicated switching method[17]. The bidirectional CLLC resonant converter in [14] requires four resonant components to be designed and therefore has a challenging issue for high volume manufacturing associated with resonant component tolerances.

In this paper, a two-stage BDC is proposed for automotive engine/battery hybrid power generators. The proposed two-stage BDC consists of a non-isolated converter and a fixed frequency SRC. The SRC is designed be capable of operating under ZCS turn on and turn off regardless of voltage and load variation in both forward and reverse operation. A method of adjusting dead time of the SRC will be presented to minimize the switch turn on losses associated with energy stored in MOSFET’s output capacitances during the ZCS turn on process. Also, a new autonomous and seamless bidirectional voltage control strategy is proposed to provide uninterrupted power to the critical AC loads and reduce the size of the DC bus capacitor.

II. PROPOSED BIDIRECTIONAL DC-DC CONVERTER The proposed BDC consists of two power conversion

stages: a non-isolated converter and a fixed frequency SRC, as shown in Fig. 2. Since the SRC is operated at fixed frequency and fixed duty all components can be designed with minimum voltage and current rating. The non-isolated converter is operated to regulate either high side voltage VH or low side voltage VL according to demanded load power and availability of the engine generator. Figs. 3 and 4 show key waveforms and operation states of the proposed SRC, respectively. Mode I

begins with Lr-Cr resonance when switches SH1 and SL2 are turned on at t0. The angular resonant frequency of the resonant circuit can be expressed as,

12r rr r

fL C

ω π= =⋅

(1)

where resonant inductance and resonant capacitance can be determined respectively by,

2

2m ks

r kpm ks

L n LL LL n L

⋅= +

+ (2)

1 23r r rC C C= + . (3)

It is seen from Figs. 3 and 4 that low side current iL (=iSL2) at Mode I(t0-t1) becomes purely sinusoidal if the on-time duty cycle is selected such that DTs = 0.5fr . Then it can be expressed as,

,( ) sin2L DC

L r

Ii t t

πω= (4)

Fig. 2. Proposed two-stage bidirectional DC-DC converter

SH1, SL2

iLr

vLr

vCr2

iSH1

vSH1

iSL2

vSL2

t0 t1 t2

iLm

0.5Ts0

VSH,on

ILm,pk

DdTs

Ts

iSH2

vSH2

SH2, SL1

vSL1

iSL1

t3 t4

VSL,off

Near ZCS turn-offNear ZCS turn-on

ZCS turn-offZCS turn-on

Near ZCS turn-offZCS turn-on

ZCS turn-offZCS turn-on

vCr1

VSL,on

IL,pk

DTs= 2fr

1

2 Vi

Vi

VSL,ON

VSH,onISH,off = ILm,pk

Fig. 3. Key waveforms of the proposed SRC

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Also, voltage across Lm can be expressed as,

( ) ( )LLm L ks

div t n V Ldt

= − + (5)

Therefore, from eqns. (4) and (5) the magnetizing current at Mode I (t0-t1) can be expressed using iLm(t0) = - iLm(t1) by,

,( ) sin2 2

ks L DCL LLm r

r m m m

n L In V nVi t t tL L L

ππ ωω

⎛ ⎞= − + ⎜ ⎟

⎝ ⎠ (6)

The resonant current can then be obtained using eqns. (4) and (6) by,

, ,( ) sin2 2 2

ks L DC L DCL LLr r

r m m m

n L I In V nVi t t tL L L n

π ππ ωω

⎛ ⎞= − + −⎜ ⎟

⎝ ⎠. (7)

Neglecting voltage oscillation after turning on of SL2, voltage across low side switch SL1 at Mode I (t0-t1) is expressed as,

1( ) 2 LSL L ks

div t V Ldt

= + (8)

The turn off voltage of low side switch can be obtained by,

,, 2

2r ks L DC

SL off L

L IV V

πω= − (9)

It should be noted that VSL,off should be greater than zero for the proposed operation. Therefore, from eqns. (4) and (9) the secondary side leakage inductance should be limited such as,

,

4 Lks

r L DC

VLIπω

< (10)

Switch SH1 is turned off at t1, and turn off current of the high side switch, ISH,off, becomes equal to peak magnetizing current ILm,pk. Since Lm is made very large in the proposed SRC, ILm,pk is very small, resulting in negligible switch turn off losses.

During Mode II, the output capacitors of SH1 and SH2 are charged and discharged, respectively by ILm,pk, as shown in Fig. 3. The charging and discharging operation may not be completed at the end of Mode II if ILm,pk is not sufficiently large, which may lead to a non-zero turn on v oltage of high and low side switches. The turn on voltages of the high and low side switches can be determined respectively by,

2,

, 2 2 2( )2 2

Lm pk d sSH on SH i

OSSp OSSs

n I D TV v t V

n C C= = −

+ (11)

,,, 1 2 2

4( )

2 2 2Lm pk d sL r ks L DC

SL on SLOSSp OSSs

nI D TV L IV v t

n C Cπω−

= = −+

(12)

Note that SH2 and SL1 are turned on with ZCS, but there exists turn on losses of high and low side switches associated with energy stored in MOSFET’s output capacitances as follows[20][21],

2, ( ) ,0.5SH loss on OSSp SH on sP C V f= . (13)

2, ( ) ,0.5SL loss on OSSs SL on sP C V f= . (14)

iL

iLr

iLm

IL,DC

(t0~t1)Mode I

(t1~t2)Mode II

COSSp

Cr2

SH1

SH2

Lks

SL1

SL2

n : 1 : 1

Vi Lm

Cr1

CL

Lks

TransformerLkp

COSSp

COSSs

COSSs

COSSp

Cr2

SH1

SH2

Lks

SL1

SL2

VL

n : 1 : 1

Vi

Cr1

CL

Lks

Lkp

COSSp

COSSs

COSSs

VL

Lm

Transformer

a

b

a

b

iL

iLr

iLm

IL,DC

Fig. 4. Key waveforms of the proposed SRC

fs1 (kHz)50 60 70 80 90 100 11030 40

Po=1 kWPo=3 kWPo=5 kW

Gai

n

20

11.0

10.5

10.0

9.5

9.0

12.5

12.0

11.51.67 X 10-4

10.0005

10.0000

9.9995

9.9990

10.0010

fr= 2D 1 fs1

4 0 4 5 50 55 6 0

Operating frequency

(a) (b) Fig. 5. Voltage gain curve (a) charging mode (b) discharging mode (Po=5 kW, Vi=280 V, VL=28 V, fs1=48 kHz, n=5, Lr=5.8 uH, Cr=1.88 uF)

t

iSH

Near ZCS turn-on Near ZCS turn-off

vSH

Load increase

tZCS turn-on

iSL vSL

ZCS turn-off

Load increase

(a) (b)

Fig. 6. Switch voltage and current waveforms of the proposed SRC (a) high side switch (b) low side switch

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However, the turn on losses of the switches may be considerable in the high voltage application. The turn on loss PSL,loss(on) of the low side switch is negligible since VSL,on is small in this low voltage application, and the turn-on loss PSH,loss(on) of the high side switch is also small due to the two-stage configuration. In the proposed SRC, PSH,loss(on) can further be reduced by increasing DdTs and, in turn, decreasing VSL,on, as shown in Fig. 3. However, increasing DdTs may cause increased current ratings and undesired resonance, and hence it should properly be chosen. Therefore, it is noted that both turn off and turn on switching losses of the proposed SRC is made negligible in this application.

In the conventional frequency-controlled SRC, in general, resonant inductance Lr should be made large to reduce the switching frequency range. In the proposed SRC, on the contrary, Lr is chosen to be small since the SRC is not used for regulation, resulting in very small gain variation according to load variation in both charging and discharging modes, as shown in Fig. 5. Furthermore, small Lr leads to less sensitive to the resonant component tolerances, eliminating the voltage regulation issues and saturation problem of magnetic devices that was introduced in the conventional frequency-controlled SRC[22][23]. This also allows Lr to be easily embedded in the transformer. Also, the proposed SRC is able to achieve ZCS turn on and turn off of the switch without regard to voltage or load variation, as shown in Fig.6, by choosing the resonant frequency fr as follows,

1 11 1

2 1 2r s sd

f f fD D

= =− (15)

III. PROPOSED CONTROL STRATEGY The high side DC bus is regulated to either 400V by the

AC-DC converter or 380V by the BDC, respectively, according to condition of VH. The conventional control of the BDC is in general realized with two individual controllers of VL control for battery charging and VH control for battery discharging, and therefore may not be able to avoid large transient during the transition from VL control to VH control of the BDC.

In this paper a new autonomous and seamless bidirectional voltage control strategy, as shown in Fig. 7, is proposed to provide uninterrupted power to the critical AC loads and reduce the size of the DC bus capacitor. The two outer loop

voltage controllers for VL control and VH control are combined by VCL whose output I *

LB is automatically selected to be either ILB,H, the output of the high side voltage controller, or ILB+, the positive limit of VCL which varies with the output of the low side voltage controller. This makes it possible to share inner loop current controller, resulting in autonomous and seamless transition from VL control(charging mode) to VH control(discharging mode), and vice versa. The peak values of the positive and negative limit, ILB-,pk and ILB+,pk, of the VCL are determined by,

, ,i

LB pk LB pki

PI I

V+ −= = (16)

According to C-rate of the battery used, ILB+,pk may be chosen smaller than eqn. (16). ILB+ varies with magnitude of VL while ILB- is always fixed at ILB-,pk. The anti-windup is used to

Anti-windup

ILB

GL(s)

Controller

GH(s)

Controller

GI(s)

ControllerPWM Generator

Variable current limiter

Anti-windup

Anti-windup

Fig. 7. Control block diagram of the proposed battery charger

Mode III Mode I Mode II

10K

5K

0K

400395390385380375

292827262524

40

20

0

-20

Pow

er [W

]Vo

ltage

[V]

Volta

ge [V

]C

urre

nt [A

]

Fig. 8. Simulation waveform of the proposed bidirectional voltage control for seamless transition from VL control to VH control

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prevent the saturation of the controllers. For the sake of simplicity it is assumed that the DC-load is constant and all the power losses of the AC-DC converter, the DC-AC inverter and the BDC in Fig. 1are neglected.

A. Transition from VL control to VH control Figs. 8 shows PSIM simulation waveforms for illustration

of the operating principle of the proposed bidirectional control strategy for transition from VL control to VH control.

Mode I : Assume that the battery has already been fully charged. The engine generator is supplying the AC and DC loads during this mode. VH is regulated to 400V by the AC-DC converter, and the reference voltage V *

H of the BDC is set at 380V. Since the high side voltage controller GH(s) is saturated the reference current I *

LB of the BDC is determined by ILB+ which is the same as IL,DC.

Mode II : This begins when the AC load increases and the sum of the AC and DC loads is greater than PG,max, the maximum power that can be produced by the generator. Then, the AC-DC converter is not able to regulate the DC bus, and VH drops off from 400V, which makes I *

LB be changed to ILB,H, as shown in Fig. 8. I *

LB decreases, changes its sign and continuously increases. This means that the BDC starts to discharge the battery and regulate VH to 380V. As the battery voltage decreases, ILB+ which is the output of the low side voltage controller GL(s) increases up to ILB+,pk . This is the end of the mode.

Mode III : I * LB is fixed at constant value since the AC load is

constant. The BDC keeps discharging the battery and regulating VH to 380V. ILB+ is kept at ILB+,pk.

B. Transition from VH control to VL control Figs. 9 shows PSIM simulation waveforms for illustration

of the operating principle of the proposed bidirectional control strategy for transition from VH control to VL control.

Mode I : This mode is identical to Mode III of Section III-A. The BDC is discharging the battery and regulating VH to 380V. I *

LB is determined by ILB,H, and ILB+ is the same as ILB+,pk.

Mode II : This mode begins when the AC load decreases and the sum of the AC and DC loads becomes smaller than PG,max. This makes the AC-DC converter be capable of regulating VH, recovering it back to 400V. Therefore, I *

LB(=ILB,H) decreases and changes its sign, meaning that the BDC is able to regulate VL to 28V, and continuously increases until it reaches to ILB+,pk. Now, I *

LB is determined by ILB+(=ILB+,pk) since the high side voltage controller GH(s) is saturated. Then, the BDC starts to charge the battery with constant current of IB,CC which is determined by,

, , ,B CC LB pk Load DCI I I+= − (17)

Mode III : When the battery voltage VL gets close to V * L the

reference current I * LB which is determined by ILB+ starts to

decrease. During this mode the BDC charges the battery with constant voltage of V *

L .

IV. EXPERIMENTAL RESULTS A 5 kW prototype of the proposed BDC has been built

under the following system parameters.

• Po = 5 kW • VH 340~440 V • VL = 24~32 V • NP : NS = 5 : 1

• fs1 = 48 kHz • fs2 = 20 kHz • Cf = 110 μF • CH = 45 μF

• Ci = 100 μF • CL = 380 μF • LB = 1 mH • Lf = 0.42 μH

• Lr = 5.8 μH • DdTs = 600 ns • Cr1 (=Cr2)=0.94 μF Figs. 10 and 11 show key experimental waveforms of the

charging and discharging modes at full load, respectively. As we can see from Figs. 10(b), (c) and Figs. 11(b), (c) all switches of the SRC are being turned on and off with ZCS in both charging and discharging modes. In fact, all switches of the SRC are always turned on and off with ZCS without regard to voltage and load variations.

Figs. 12 shows the experimental waveforms of the mode transition. A 24 V/100 Ah lead acid battery was used at the low voltage side. Fig. 15 shows that the BDC is regulating VL to charge the battery, and VH is regulated to 400 V by the AC-DC converter. When the engine generator is shut down, VH drops but is recovered to 380 V since the BDC changes over to VH control to discharge the battery.

The efficiency of the proposed BDC including gate drive and control circuit losses is measured by YOKOGAWA WT3000 and shown in Fig. 13. The maximum efficiencies are 95.13 % at 1.3 kW in charging mode and 95.08 % at 1.5 kW in

Mode III Mode I Mode II

CC CV

IB,CC

10K

5K

0K

400395390385380375

292827262524

40

20

0

-20

Pow

er [W

]Vo

ltage

[V]

Volta

ge [V

]C

urre

nt [A

]

Fig. 9. Simulation waveform of the proposed bidirectional voltage control for seamless transition from VH control to VL control

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discharging mode, respectively. Fig. 14 shows the photograph of the proposed BDC prototype.

V. CONCLUSIONS This paper proposes a bidirectional DC-DC converter for

automotive engine/battery hybrid power generators. The features of the proposed BDC are as follows.

• The proposed topology preserves the advantages of the two-stage DC-DC converter: 1) The switching method is simple in that voltage regulation and mode transition are carried out only by the non-isolated converter; 2) All components’ ratings of the isolated converter are optimized.

• Small Lr can be used since the proposed SRC is not used for regulation, which leads to the following

advantages: 1) The SRC has very small gain variation according to load variation, and therefore the proposed BDC can be designed for wider voltage range; 2) The SRC is less sensitive to the resonant component tolerances, and therefore suitable for high volume

(a) (b) (c)

Fig. 10. Experimental waveforms of the charging mode (a) inductor current ILB, switch voltages VSB,1 and VSB,2 of the non-isolated converter (b) primary current Ipri, high side switch voltages VSH,1 and VSH,2 of the SRC (c) primary current Ipri, low side switch voltages VSL,1 and VSL,2 of the SRC

Ipri [20A/div]

VSL,1 [50V/div]VSL,2 [50V/div]

[5μs/div]

ZCS turn on & turn off

(a) (b) (c)

Fig. 11. Experimental waveforms of the discharging mode (a) inductor current ILB, switch voltages VSB,1 and VSB,2 of the non-isolated converter (b) primary current Ipri, high side switch voltages VSH,1 and VSH,2 of the SRC (c) primary current Ipri, low side switch voltages VSL,1 and VSL,2 of the SRC

Fig. 12. Experimental waveforms of transition from VL control for charging to VH control for discharging

Eff

icie

nty

(%)

Fig. 13. Measured efficiencies of the proposed BDC including gate drive and control circuit losses

Fig. 14. Photograph of the proposed BDC prototype

745

manufacturing; 3) Small Lr can be easily embedded in the transformer.

• The proposed SRC is capable of achieving ZCS turn on and turn off regardless of voltage and load variation. A method of adjusting dead time of the SRC has been presented to minimize the switch turn on losses associated with energy stored in MOSFET’s output capacitances during the ZCS turn on process.

• An autonomous and seamless bidirectional voltage control method with a variable current limiter has been proposed to provide uninterrupted power to critical AC loads and reduce the size of the DC bus capacitor.

Experimental results from a 5kW prototype were provided to validate the proposed concept. The maximum efficiencies including gate drive and control circuit losses are 95.13 % at 1.3 kW in charging mode and 95.08 % at 1.5 kW in discharging mode, respectively.

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