Microelectronics Journal 42 (2011) 1252–1256

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Microelectronics Journal

0026-26

doi:10.1

n Corr

E-m

journal homepage: www.elsevier.com/locate/mejo

Design of fractal-based CMOS bandpass filter for WirelessHD system

Wei-Yu Chen a,n, Shoou-Jinn Chang a, Min-Hang Weng b, Cheng-Yuan Hung c

a Institute of Microelectronics and Department of Electrical Engineering, Advanced Optoelectronic Technology Center, Center for Micro/Nano Science and Technology, National

Cheng Kung University, Taiwanb Medical devices and opto-electronics equipment department Metal Industries Research & Development Center, Taiwanc Department of Electronics Engineering and Computer Sciences Tung-Fang Institute of Technology, Taiwan

a r t i c l e i n f o

Article history:

Received 12 February 2011

Received in revised form

3 August 2011

Accepted 9 August 2011Available online 9 September 2011

Keywords:

Fractal

Dual-mode

mm-wave

CMOS

WirelessHD

92/$ - see front matter & 2011 Elsevier Ltd. A

016/j.mejo.2011.08.004

esponding author.

ail address: mark00.chen@gmail.com (W.-Y. C

a b s t r a c t

We proposed a fractal-based dual-mode bandpass filter (BPF) using a standard CMOS process for

application of 60 GHz WirelessHD system. We first investigated the effect of coupling feedlines of I/O

ports set at different layer of M3 and M4 layer on the transmission loss of the resonator, and verified

the nature coupling of fractal-based dual-mode filter. Experimental result shows that the designed filter

with a fractional-bandwidth (FBW) of 23%, an insertion loss about 7 dB and return loss larger than

10 dB. Additionally, two transmission zeros are appeared at the passband edges, thus much improve

the selectivity of the proposed CMOS BPF. The result indicates that fractal-based structure is feasible

and can meet the requirement in the mm-wave application.

& 2011 Elsevier Ltd. All rights reserved.

1. Introduction

The wireless transmission with high capacity and high datarates always plays a desired goal to meet in communicationsystem. Currently, the blooming WirelessHD system servers toorganize an industry-led standardization effort to define a next-generation wireless digital interface specification for consumerelectronics and PC products. Specifically, the WirelessHD specifi-cation is based on the unlicensed 60 GHz band with the 7 GHz ofcontinuous bandwidth and a theoretical data rates as high as25 Gbps. It enables wireless connectivity for uncompressedstreaming high definition audio, video and data between sourcedevices and high-definition displays [1]. Moreover, great effortshave been made to integrate the front-end components on thelow-resistivity silicon substrate using stand CMOS processrecently [2–5]. The CMOS technology offers the potential for thefront-end component to realize the RF system-on-chip (SoC).

As is well known, the bandpass filter (BPF) is the key componentin the front-end circuit of wireless communication system; however,there are few investigations concerning about the WirelessHDapplication and implemented using standard CMOS process at thesame time. Besides, the dual-mode filter is attractive since itsadvantages of narrow bandwidth, two symmetric transmission zeros,and high-selectivity characteristics in filter frequency response [6,7].

For the consideration of large substrate loss of the low-resistivitysilicon up to mm-wave frequencies, the resonator with high quality

ll rights reserved.

hen).

factor shall be used. Among the various types of resonator, dual-mode resonator is found to have high quality factor. Moreover, thedescription of geometry of Sierpinski based fractal, designed guide-line and the discussions of high quality and miniaturization havebeen investigated [8,9]. In Refs. [8], it is the first study of theSierpinski square resonator. The property of resonant behavior andthe external quality factor were well analyzed and discussed. In Ref.[9], the dual-mode bandpass filter was realized and implemented onthe ultra-thin liquid crystal polymer substrate using four Sierpinskisquares with third order. However, it is a challenge to design anmm-wave bandpass filter using standard CMOS process whenconsidering the large substrate loss of the low-resistivity silicon upto mm-wave frequencies.

In this paper, we developed a fractal-based dual-mode band-pass filter using a standard CMOS process for the WirelessHDsystem, as shown in Fig. 1. First, the design of input/output (I/O)ports is analyzed. We investigated the effect of coupling feedlinesof I/O ports set at different layer of M3 and M4 layer on thetransmission loss of the resonator. Second, we discussed thatnature coupling of dual-mode filter based on the Sierpinskistructure. Finally, the designed BPF was fabricated and measuredto experimentally verify the simulated results of our design.

2. Fractal-based dual-mode CMOS BPF

2.1. Design of the feedlines for the fractal-based BPF

In this design, the Sierpinski based dual-mode resonator isadopted for the reasons of the high quality factor and

Fig. 1. (a) Circuit layout and (b) cross section of layer configuration in standard

CMOS process. W1¼15 mm, W2¼10 mm, L1¼480 mm, L2¼440 mm M1¼0.665 mm,

M2¼0.64 mm, M3¼0.64 mm, M4¼0.925 mm, IMD1¼1.665 mm, IMD2¼1.64 mm,

IMD3¼1.64 mm. (Via: Via hole, M1: Metal-1, M2: Metal-2, M3: Metal-3, M4:

Metal-4, IMD1: Metal-1 and thin oxide-1, IMD2: Metal-2 and thin oxide-2, IMD3:

Metal-3 and thin oxide-3.)

Fig. 2. Layout of the resonator structure with the feedlines of I/O port located at

(a) M3 layer, (b) M4 layer and (c) transmission loss of the resonator structure with

two different coupling schemes.

W.-Y. Chen et al. / Microelectronics Journal 42 (2011) 1252–1256 1253

miniaturization. In our previous study, the resonant frequenciesof the Sierpinski square are lower and the external qualityfactors are higher as the higher iteration is performed. Up to thefourth order Sierpinski square, the external quality factorswill achieve the saturation value about 210. Therefore, in thisstudy, the fourth order Sierpinski square are chosen for the designof dual-mode filter [8]. In order to meet the requirement of theWirelessHD system on the silicon substrate, we chose thefourth order of Sierpinski square geometry as the main resonatorstructure, as shown in Fig. 1. Moreover, since the fractal-basedgeometry is mainly generated according to the iterativemethod. When the iterative order is larger than seven, theminimum line-width is smaller than 0.25 mm. Namely, the line-width in the structure will limit the iterative order of theSierpinski square when using the stand 0.35 mm CMOS process.The I/O ports with the coupling feedlines are designed toprovide the energy transfer to the resonator structure. Since thecenter frequency of the Sierpinski dual-mode filter is designed at60 GHz, the side length of 480 mm is derived using Eq. (1) [9]and the slightly discrepancy is well optimized by the EM simula-tion [10]

L1 ¼150� R

fD �ffiffiffiffierp ðUnit:mmÞ: ð1Þ

where the resonant frequency ratio, R, is defined as fN/f0, fN is theresonant frequency of the nth iteration order resonator and f0 isthe resonant frequency of the patch resonator. fD is the designedfrequency (fD¼60 GHz, R¼0.38).

To provide the higher coupling energy in the proposed dual-mode filter, we adopted a novel scheme. Namely, the enhancedfeedlines are set at M3 layer and the I/O ports are set at M4 layerfor the consideration of on-wafer measurement and the lowerseries resistance due to the thickness of the metal layer. The I/Oports set at M4 layer is connected to M3 layer using the via holes,as shown in Fig. 1(b). To further know the reason why wedesigned the enhanced feedlines set at M3 layer, the transmissionloss of the resonator using the proposed new scheme andconventional scheme of feedlines are simulated and compared.Layout of the resonator structure with the feedlines of I/O portlocated at M3 layer and M4 layer is shown in Fig. 2(a) and (b),respectively. Note that, the feedlines shown in Fig. 2(b) is setaside the resonator with a gap of 5 mm and a length of 480 mm.The transmission loss of the resonator structure with two differ-ent coupling schemes is shown in Fig. 2(c). It is found themagnitude of S21 is about �2.9 dB for the novel scheme using

the feedlines set at M3 layer, on the other hand, the magnitude ofS21 is about �41.3 dB for the conventional scheme using thefeedlines set at M4 layer. It indicates that the feedlines set at M3layer provides a strong coupling strength than that set at M4layer, since the feedlines set at M3 layer provides a highercapacitive factor.

2.2. Coupling characteristics

To form the filter response, the I/O ports are located ortho-gonally to the fractal based dual-mode resonator and the pertur-bation element shall be added in the corner of the resonator [6].Moreover, it is known for that a BPF based on a dual-moderesonator can achieve elliptic response or the response with tworeal-axis transmission zeros [7]. The appearance and location ofthe transmission zeros depends on the cutting or filling ofperturbation element. To identify the nature of the couplingbetween degenerate modes of the proposed fourth order Sier-pinski square resonator, we use square cells with an area of225 mm2 for the basic cell of perturbation element for the cuttingor filling, as shown in Fig. 3. A full-wave electromagnetic (EM)simulator [10] is used to characterize the coupling between thedegenerate modes. Fig. 4 shows the current distribution of thefourth order Sierpinski square resonator without and with filledor cut perturbation elements. The current distributions on fourcorners are identical at the resonance frequency, as shown inFig. 4(a). Therefore, the degenerate resonance modes in the fourthorder Sierpinski square resonator do not couple with each other.When seven basic cells are filled at the corner B (or D) on thesymmetry axis, as shown in Fig. 4(b), this type of the perturbationwould result in an increase in the inductance per unit lengthwithin the resonator and consequently an inductive effect existson the dual-mode Sierpinski square resonator. Additionally, theBPF shows an elliptic response, as confirmed by our simulationsshown in Fig. 5. On the other hand, when 13 basic cells are cut at

Fig. 3. Conception of the cell division.

Fig. 4. Simulated current distribution pattern for the fourth order Sierpinski square resonator at the resonant frequency for a single mode. (a) Without perturbation

element, (b) with 7 cells of perturbation element filled at the inner corner and (c) with 13 cells of perturbation element cut at the inner corner.

Fig. 5. Responses of fourth order Sierpinski square filter with elliptic response

(Fig. 4(b)) and a response with two real-axis transmission zeros (Fig. 4(c)).

W.-Y. Chen et al. / Microelectronics Journal 42 (2011) 1252–12561254

the corner B (or D), as shown in Fig. 4(c), this kind of theperturbation would increase the capacitance per unit length ofthe resonator due to the maximum current distribution on the

corner B (and C) on the symmetry axis. At this time, the BPFshows a response with two real-axis transmission zeros, as shownin Fig. 5. Therefore, to achieve a better passband selectivity, thecells filled at the inner corner are used as the perturbationelements for this design, as shown in Fig. 1(a).

To further investigate the perturbation’s size on the couplingproperty, the simulated split resonance frequencies of thesedegenerate modes of the Sierpinski dual-mode resonatorand the corresponding coupling coefficient as functions of per-turbation’s size are shown in Fig. 6. It is found that the splitbetween the mode frequencies decreases when the size of theperturbation element filled or cut at the inner corner is lessthan 18 cells (4050 mm2) and 16 cells (3600 mm2), respectively,while it increases when the size further is larger than 18 cells(4050 mm2) filled or 16 cells (3600 mm2) cut at the innercorner. The corresponding coupling coefficient (K) can becalculated as (1)

K ¼f 22�f 2

1

f 22�f 2

1

ð2Þ

where f2 and f1 are the resonant frequency of the mode I andmode II with respect to the degenerate modes, respectively.

Fig. 6. Simulated two resonant frequencies (mode I and mode II) and coupling

coefficient as functions of different size of the perturbation element.

Fig. 7. The simulated and measured responses of the proposed CMOS BPF

(inserted picture is the die photo).

W.-Y. Chen et al. / Microelectronics Journal 42 (2011) 1252–1256 1255

To meet an optimized filter responses and a better coupling oftwo generated modes, the perturbation element is filled by 25cells and the coupling coefficient is about 0.126.

3. Implement and measurement results

After careful layout, the proposed dual-mode CMOS BPF,utilizing the fourth order Sierpinski resonator discussed above,was implemented using standard 0.35 mm CMOS process (D35)having a thickness about 4 mm. It is assumed that the dielectricconstants of each poly layers within 0.35 mm process are thesame, thus the equivalent dielectric constant can be obtained as3.9 according to the formulation [11]

ere ¼XN

n ¼ 1

tn

ern

!�1 XN

n ¼ 1

tn

!ð3Þ

where tn is the height of the nth substrate layer and ern is thedielectric constant of the nth substrate layer. The physical size ofthe fabricated CMOS BPF is compact and only 480 mm�480 mm,i.e., approximately 0.192lg by 0.192lg, where lg is the guidedwavelength at the center frequency of 60 GHz. The fabricatedCMOS filter was measured on the probe station by an HP8510CNetwork Analyzer after the standard on wafer calibration ofShort-Open-Load-Thru (SOLT). Fig. 7 shows the simulated andmeasured results where the inserted picture is the die photo inmeasurement. To meet the requirement of measurement, one ofthe I/O ports is bended. The fabricated CMOS BPF has a measuredcenter frequency of 60 GHz, a fractional-bandwidth (FBW) of 23%,an insertion loss less than 7 dB and a return loss larger than 10 dB.It is known that the circuit quality factor, Q0, is defined as

1

Q0¼

1

Qcþ

1

Qd¼

l0ðacþadÞ

pffiffiffiffiffiffierep ð4Þ

where ac and ad are the attenuation coefficient corresponding tothe conductor and dielectric losses, respectively, and Qc and Qd,are the quality factors corresponding to conductor and dielectriclosses, respectively. In our previous study, the external qualityfactors of the fourth order Sierpinski square resonator realized onthe low loss dielectric substrate achieve the saturation valueabout 210. Since the low-resistivity silicon has a very largesubstrate loss up to mm-wave frequencies, the external qualityfactors of the proposed filter using fourth order Sierpinski squareresonator is 4. However, we still verified that the measuredpassband perform of our design using standard 0.35 mm CMOSprocess is in the acceptable range. The measured insertion lossabout 7 dB is due to the close proximity of the ground plane to the

signal line, which significantly degrades the inductive quality ofthe circuit. It can be improved using the advanced standard CMOSprocess with the thicker thickness i.e. the 0.18 mm or 0.13 mmprocess. Moreover, using the cells filled at the inner corner as theperturbation element, two transmission zeros appeared at thepassband edges are clearly observed, as discussed in Fig. 4(b),thus much improving the selectivity of the proposed CMOS BPF.

4. Conclusions

We have designed and implemented a fractal-based CMOS BPFfor the WirelessHD system. Sierpinski based fractal is used as themain dual-mode resonator. Using a new scheme of the feedlines,the coupling strength of the I/O ports can be enhanced. The naturecoupling of the dual-mode filter is also verified and discussed. Thephysical size of the fabricated CMOS BPF is compact and only480 mm�480 mm, i.e., approximately 0.192lg by 0.192lg, wherelg is the guided wavelength at the center frequency of 60 GHz.The measured results of the fabricated CMOS BPF at 60 GHz havea fractional-bandwidth (FBW) of 23%, a return loss larger than10 dB and an insertion loss about 7 dB, which is due to the closeproximity of the ground plane to the signal line, significantlydegrading the inductive quality of the circuit. Moreover, twotransmission zeros appeared at the passband edges are clearlyobserved, thus much improving the selectivity of the proposedCMOS BPF. The proposed filter provides a feasible result to applythe fractal-based structure in the mm-wave application.

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