development of a low voltage static synchronous compensator statcom for educational … ·...

Development of a Low Voltage Static

Synchronous Compensator STATCOM

for Educational Use

2016

Thesis Report submitted to the school of Engineering and Information Technology, Murdoch

University, in partial completion of a Bachelor of Engineering.

Author: Jordan Goodchild

Thesis Supervisor: Mr Craig Carter

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Abstract

Electrical power quality is essential in the effective transmission and distribution of power to

consumers. With the increase in non-linear loads it is becoming more prevalent to maintain the

quality of power supplied. Voltage sagging or swelling and power factor are two of the major

power quality issues. By introducing a static synchronous compensator (STATCOM) into a system

these power-quality issues can be significantly reduced and maintained effectively.

This project developed and tested the idea of constructing a low voltage static synchronous

compensator for educational use, in which students could safely experiment with the concept

and tools of power quality control. The end design goal was to construct a device capable of

displaying the concepts of a STATCOM by producing a low voltage Alternating Current (AC)

waveform to supply the given loads. Features of this project were to include a real-time display of

system parameters and a switch with the ability to change between voltage control and power

factor control.

The project incorporates the use of various components to achieve the desired STATCOM.

Integrated into this system is an Arduino Uno microcontroller. This controller produces the

triangular carrier waveform through direct digital synthesis. Additionally, it is responsible for

the measurement and display of system data. The control signals produced by the

comparison of the filtered Direct Digital Synthesis (DDS) triangular waveform with a

sinusoidal control signal are fed into an Infineon Dual Half-Bridge. Following this the output

of the bridge is fed through a low pass filter to produce the required output sinusoid for the

STATCOM.

Through several experiments the output waveform of the STATCOM was monitored and

recorded. One of the parameters monitored for the STATCOM output was the total harmonic

distortion of the output waves. At low amplitude modulation values the output waveform

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resembles a sinusoidal output however as expected at higher values the output waveforms

begin to distort and resemble a square wave.

The major recommended future works are to design a control algorithms capable of

performing voltage control and power factor control to display the concepts of power quality

control. In addition to this a method of switching between the two control modes will require

installation.

The project is unfortunately not considered to be at a completed, ready to implement stage.

However, sufficient work has been undertaken and completed for another willing student to

finalise the project.

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Table of Contents

ABSTRACT ................................................................................................................................................... II

TABLE OF CONTENTS ................................................................................................................................. IV

LIST OF FIGURES ........................................................................................................................................ VI

LIST OF TABLES ....................................................................................................................................... VIII

GLOSSARY OF ACRONYMS AND ABBREVIATIONS ..................................................................................... IX

ACKNOWLEDGEMENTS .............................................................................................................................. X

1. INTRODUCTION .................................................................................................................................. 1

1.1. COMMERCIALLY AVAILABLE SOLUTIONS ..................................................................................................... 2

2. LITERATURE REVIEW........................................................................................................................... 3

2.1. DC – AC CONVERTERS ........................................................................................................................... 3

2.1.1. HALF BRIDGE INVERTERS ............................................................................................................... 4

2.1.2. FULL BRIDGE INVERTERS ................................................................................................................ 6

2.2. PWM SWITCHING SCHEME .................................................................................................................... 8

2.2.1. PWM WITH BIPOLAR VOLTAGE SWITCHING .................................................................................... 14

2.2.2. PWM WITH UNIPOLAR VOLTAGE SWITCHING ................................................................................. 15

2.3. FLEXIBLE ALTERNATING CURRENT TRANSMISSION SYSTEMS ......................................................................... 17

2.3.1. STATIC SHUNT COMPENSATORS .................................................................................................... 18 2.3.1.1. STATIC VAR COMPENSATORS .................................................................................................................... 19 2.3.1.2. STATCOM ......................................................................................................................................... 20

2.4. ARDUINO PLATFORM ........................................................................................................................... 22

2.5. DIRECT DIGITAL SYNTHESIS ................................................................................................................... 25

3. PROJECT DESIGN .............................................................................................................................. 30

3.1. POWER SUPPLY .................................................................................................................................. 30

3.1.1. DC POWER SUPPLY .................................................................................................................... 30

3.1.2. AC POWER SUPPLY .................................................................................................................... 32

3.2. LINE AND LOAD BANK .......................................................................................................................... 33

3.2.1. LOAD DESIGN ............................................................................................................................ 33

3.2.2. LINE IMPEDANCE DESIGN ............................................................................................................. 41

3.3. LIQUID CRYSTAL DISPLAY INTERFACE ....................................................................................................... 42

3.3.1. 1602 LCD SHIELD PARALLEL (4 BIT) INTERFACE .............................................................................. 43

3.3.2. 2004 LCD I2C INTERFACE ........................................................................................................... 45

3.4. MEASUREMENT .................................................................................................................................. 48

3.4.1. ZERO CROSSOVER DETECTION ....................................................................................................... 48

3.4.2. CURRENT MEASUREMENT ............................................................................................................ 51

3.4.3. VOLTAGE MEASUREMENT ............................................................................................................ 56

3.5. PWM SWITCHING SCHEME .................................................................................................................. 60

3.5.1. TRIANGULAR CARRIER AND SINUSOIDAL CONTROL WAVEFORMS ......................................................... 61

3.5.2. PWM SWITCHING METHOD ........................................................................................................ 65

3.6. STATCOM ....................................................................................................................................... 68

3.6.1. FULL BRIDGE VSI ....................................................................................................................... 68

3.6.2. STATCOM FILTER ..................................................................................................................... 72

3.6.3. DC CAPACITOR .......................................................................................................................... 73

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4. RESULTS ........................................................................................................................................... 76

4.1. MEASUREMENT AND CONTROL WAVEFORMS ........................................................................................... 76

4.1.1. VOLTAGE SCALING ...................................................................................................................... 76

4.1.2. CURRENT SCALING ...................................................................................................................... 78

4.1.3. TRIANGULAR WAVE .................................................................................................................... 80

4.2. STATCOM ....................................................................................................................................... 82

4.2.1. OUTPUT WAVEFORM .................................................................................................................. 82

4.2.2. BASIC STATCOM IMPLEMENTATION ............................................................................................. 86

5. FUTURE WORKS ................................................................................................................................ 88

5.1. 2.5 VOLT BIAS ................................................................................................................................... 88

5.2. MICROCONTROLLER ALTERNATE ............................................................................................................ 89

5.3. VSI ALTERNATE .................................................................................................................................. 89

5.4. ADDITIONAL FEATURES ........................................................................................................................ 90

5.5. MANUFACTURED PCB ......................................................................................................................... 90

5.6. DC-DC BOOST CONVERTER .................................................................................................................. 91

5.7. CONTROL ALGORITHM ......................................................................................................................... 91

6. CONCLUSION .................................................................................................................................... 93

APPENDICES .............................................................................................................................................. 95

A. SOFTWARE ....................................................................................................................................... 96

B. FULL CIRCUIT DRAWINGS ............................................................................................................... 103

C. RELEVANT DATASHEET INFORMATION ........................................................................................... 108

LM319 DATASHEET INFORMATION ................................................................................................................... 108

D. SCALING RATIO TABLES .................................................................................................................. 109

VOLTAGE SCALING RATIO TABLES ...................................................................................................................... 109

CURRENT SCALING RATIO TABLES ...................................................................................................................... 110

E. STATCOM OUTPUT WAVEFORMS ................................................................................................... 111

WORKS CITED ......................................................................................................................................... 122

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List of Figures

FIGURE 2-1. HALF BRIDGE INVERTER (LEFT). OUTPUT WAVEFORM (RIGHT). [5] .................................................................. 5

FIGURE 2-2. BASIC H-BRIDGE INVERTER TOPOLOGY (LEFT), OUTPUT WAVEFORM (RIGHT) [8] ............................................. 6

FIGURE 2-3. BASIC H-BRIDGE SWITCHING SCHEME [9] ................................................................................................. 7

FIGURE 2-4. PWM SWITCHING SCHEME CARRIER (BLUE) AND CONTROL (RED) WAVEFORMS [9] ......................................... 9

FIGURE 2-5. VOLTAGE CONTROL BY VARYING MA [4] .................................................................................................. 10

FIGURE 2-6. OVER-MODULATION, BORDERLINE SQUARE-WAVE OPERATION [9] .............................................................. 11

FIGURE 2-7. CONTROL PULSE GENERATION EXAMPLE ................................................................................................. 12

FIGURE 2-8. PWM SWITCHING SCHEME OUTPUT WAVEFORM; UNFILTERED (BLUE), FILTERED (RED) [9] ............................ 13

FIGURE 2-9. BIPOLAR PWM SWITCHING SCHEME [11] .............................................................................................. 15

FIGURE 2-10. UNIPOLAR PWM SWITCHING SCHEME [11] .......................................................................................... 16

FIGURE 2-11. BASIC FULL BRIDGE INVERTER ............................................................................................................. 16

FIGURE 2-12. STATIC SHUNT COMPENSATOR DIAGRAMS [13] ..................................................................................... 18

FIGURE 2-13. STATIC SERIES COMPENSATOR EXAMPLES [14] ....................................................................................... 19

FIGURE 2-14. STATCOM SCHEMATIC EXAMPLE [12] ................................................................................................ 21

FIGURE 2-15. ARDUINO UNO/GENUINO MICROCONTROLLER [25] ............................................................................... 23

FIGURE 2-16. ATMEGA 328P PDIP CHIP PIN LAYOUT [26] ......................................................................................... 23

FIGURE 2-17. ARDUINO MEGA/GENUINO MEGA MICROCONTROLLER [27] .................................................................... 24

FIGURE 2-18. AD9833 PIN LAYOUT [31] ................................................................................................................ 25

FIGURE 2-19. FUNDAMENTAL ARCHITECTURE OF A DDS GENERATOR [33] ..................................................................... 26

FIGURE 2-20. FREQUENCY TUNEABLE DDS GENERATOR [32] ...................................................................................... 27

FIGURE 2-21. PHASE WHEEL SHOWING PHASE ACCUMULATOR REGISTER M = 1 [24] ...................................................... 28

FIGURE 2-22. LOOKUP TABLE OF ONE TRIANGULAR WAVEFORM PERIOD ....................................................................... 29

FIGURE 3-1. AC (BOTTOM) AND DC (TOP) POWER SUPPLY BOX..................................................................................... 31

FIGURE 3-2. LOAD BANK ....................................................................................................................................... 33

FIGURE 3-3 3.0MH AIR CORE INDUCTOR ................................................................................................................. 42

FIGURE 3-4. 16 PIN INTERFACE EXAMPLE .................................................................................................................. 43

FIGURE 3-5 1602 LCD ARDUINO SHIELD WITH TACTILE SWITCHES [43] ........................................................................ 43

FIGURE 3-6 EXAMPLE OF AN I2C-BUS CONFIGURATION [44] ....................................................................................... 46

FIGURE 3-7. 2004 LCD DISPLAY [45] ...................................................................................................................... 46

FIGURE 3-8. I2C DRIVER [46] ................................................................................................................................. 46

FIGURE 3-9. 2004 LCD DISPLAY ............................................................................................................................. 48

FIGURE 3-10. MODIFIED RAMP GENERATOR [47] ...................................................................................................... 49

FIGURE 3-11. HIGH SPEED COMPARATOR AND EXCLUSIVE-OR GATE ZERO CROSS OVER [47] ............................................ 49

FIGURE 3-12. DS3486 VS. MC3486 DESIGN .......................................................................................................... 50

FIGURE 3-13. DS3486 ZERO CROSSING DETECTOR [47] ............................................................................................ 50

FIGURE 3-14. BASIC EXAMPLE OF A CURRENT TRANSFORMER (LEFT) [49], CIRCUIT SYMBOL (RIGHT) [48] ............................ 52

FIGURE 3-15. CURRENT TRANSFORMER PRIMARY WINDING INCREASING EXAMPLE [48] ................................................... 52

FIGURE 3-16. ALTTEC 5A 1000:1 L01-6210 THROUGH HOLE CURRENT TRANSFORMER [50] .......................................... 53

FIGURE 3-17. CURRENT SENSOR INTERFACING WITH AN ARDUINO ANALOG INPUT [51] .................................................... 55

FIGURE 3-18. PROJECT CIRCUIT - CURRENT MEASUREMENT ........................................................................................ 55

FIGURE 3-19. BRIDGE RECTIFICATION CIRCUIT WITH OUTPUT WAVEFORM [52] .............................................................. 56

FIGURE 3-20. VOLTAGE MEASUREMENT WITH TRANSFORMER [54] .............................................................................. 57

FIGURE 3-21. VOLTAGE MEASUREMENT WAVEFORMS [54] ........................................................................................ 57

FIGURE 3-22. 3VA 12+12 VAC TRANSFORMER [55] ................................................................................................ 58

FIGURE 3-23. SINUSOID MIDPOINT OBSERVATIONS .................................................................................................... 59

FIGURE 3-24. SINUSOID MIDPOINT OBSERVATION (AFTER POTENTIOMETER ADDED)........................................................ 59

FIGURE 3-25. MARTIN NAWRATH 12.5KHZ CHEBYSHEF LOW PASS FILTER [30] .............................................................. 61

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FIGURE 3-26. MODIFIRED CHEBYSHEF LOW PASS FILTER ............................................................................................. 62

FIGURE 3-27. TRIANGULAR WAVEFORM AT VARIOUS FREQUENCIES, 1500HZ (LEFT), 5000HZ (RIGHT) ............................... 63

FIGURE 3-28. MODIFIED 12.5KHZ CHEBYSHEF LOW PASS FILTER ................................................................................. 64

FIGURE 3-29. LM319 DIL PACKAGE PIN LAYOUT [56] ............................................................................................... 65

FIGURE 3-30. SINUSOIDAL CONTROL SIGNAL OVERTOP OF TRIANGULAR CARRIER WAVEFORM ........................................... 66

FIGURE 3-31. OSCILLOSCOPE SNAPSHOT OF SINUSOIDAL SIGNAL (TOP) AND PWM COMPARATOR OUTPUT (BOTTOM) ........... 66

FIGURE 3-32. INVERSED PWM SWITCHING PULSES.................................................................................................... 67

FIGURE 3-33. VARYING CARRIER MAGNITUDE EXAMPLE ............................................................................................. 68

FIGURE 3-34. INITIAL H-BRIDGE DESIGN CIRCUIT [57] ................................................................................................ 69

FIGURE 3-35. INFINEION BTN8982TA MOTOR DRIVER SHIELD [58] ............................................................................ 70

FIGURE 3-36. APPLICATION CIRCUIT FOR A BI-DIRECTIONAL MOTOR CONTROL WITH BTN8982TA [58] ............................... 71

FIGURE 3-37. LCL FILTER LAYOUT [61] .................................................................................................................... 72

FIGURE 3-38 RG CHASSIS MOUNT ELECTROLYTIC CAPACITOR. [65] .............................................................................. 75

FIGURE 4-1. ATTENUATED AND BIASED MEASUREMENT WAVEFORM ............................................................................. 77

FIGURE 4-2. SPREAD OF VOLTAGE MEASUREMENTS ................................................................................................... 77

FIGURE 4-3. SPREAD OF CURRENT MEASUREMENTS ................................................................................................... 79

FIGURE 4-4. PLOT OF TRIANGULAR AMPLITUDE VERSUS VALA VARIATIONS WITH LIMITS .................................................... 81

FIGURE 4-5. 1.5KHZ TRIANGULAR CARRIER WAVEFORM WITH VALA OF 15 .................................................................... 82

FIGURE 4-6. BASELINE HARMONIC CONTENT TDS1001B SNAPSHOT. ........................................................................... 83

FIGURE 4-7. BASELINE HARMONIC CONTENT EXCLUDING FIRST ORDER .......................................................................... 84

FIGURE 4-8. STATCOM OUTPUT WAVEFORM VALA OF 20 ........................................................................................ 85

FIGURE 4-9. STATCOM OUTPUT WITH LINE VOLTAGE ............................................................................................... 87

FIGURE 4-10. STATCOM OUTPUT WITH LINE VOLTAGE CORRECTED PHASE ................................................................... 87

FIGURE 5-1. FINE-TUNED VOLTAGE DIVIDER [68] ...................................................................................................... 89

FIGURE 5-2. XC4514 DC-DC CONVERTER [70] ........................................................................................................ 91

FIGURE 6-1. OVERALL PROJECT ENCLOSURES, POWER SUPPLIES (LEFT), MEASUREMENT (CENTRE) AND LOADS (RIGHT).......... 93

FIGURE C-1. LM319 METAL CAN PACKAGE [56] .................................................................................................... 108

FIGURE C-2. LM319 TYPICAL APPLICATIONS [56] ................................................................................................... 108

FIGURE E-1. VALA OF 100, OUTPUT WAVEFORM (TOP), FFT WAVEFORM (CENTRE), OUTPUT HARMONICS (BOTTOM) ....... 112

FIGURE E-2. VALA OF 90, OUTPUT WAVEFORM (TOP), FFT WAVEFORM (CENTRE), OUTPUT HARMONICS (BOTTOM) ......... 113








FIGURE E-10. VALA OF 10, OUTPUT WAVEFORM (TOP), FFT WAVEFORM (CENTRE), OUTPUT HARMONICS (BOTTOM) ....... 121

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List of Tables

TABLE 2-1. HALF BRIDGE SWITCHING SCHEME ............................................................................................................ 5

TABLE 2-2. H-BRIDGE SWITCHING POSITIONS ............................................................................................................. 8

TABLE 2-3. COMPARATOR SWITCHING SCHEME ......................................................................................................... 11

TABLE 2-4. UNIPOLAR PWM COMPARATOR CONTROL SIGNALS ................................................................................... 15

TABLE 2-5. UNIPOLAR PWM VSI OUTPUT CONTROL SIGNALS ..................................................................................... 17

TABLE 2-6. ARDUINO UNO SPECIFICATIONS [29] ....................................................................................................... 24

TABLE 3-1. RESISTIVE LOAD CALCULATIONS. ............................................................................................................. 34

TABLE 3-2. RECALCULATED RESISTIVE LOADS ............................................................................................................. 36

TABLE 3-3. REACTIVE LOAD RESULTS ....................................................................................................................... 37

TABLE 3-4. CAPACITOR VALUE COMPARISON ............................................................................................................ 38

TABLE 3-5. INDUCTANCE CALCULATION RESULTS ........................................................................................................ 40

TABLE 3-6. INDUCTOR VALUE COMPARISON .............................................................................................................. 41

TABLE 3-7. 1602 LCD PIN CONNECTIONS [43].......................................................................................................... 44

TABLE 3-8. DC CAPACITOR SPECIFICATIONS [65] ....................................................................................................... 75

TABLE 4-1. PEAK-TO-PEAK VOLTAGE VALUES OF VARYING VALA TRIANGULAR WAVES ..................................................... 80

TABLE 4-2. STATCOM OUTPUT HARMONIC CONTENT ............................................................................................... 85

TABLE 5-1. FUTURE WORKS ................................................................................................................................... 88

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Glossary of Acronyms and Abbreviations

AC Alternating Current

CSI Current Source Inverter

DC Direct Current

DDS Direct Digital Synthesis

FACTS Flexible Alternating Current Transmission Systems

FFT Fast Fourier Transform

I2C Inter-Integrated Circuit

I/O Input / Output

IC Integrated Circuit

IGBT Insulated-Gate Bipolar Transistor

LCD Liquid Crystal Display

LC Inductive-Capacitive

LCL Inductive-Capacitive-Inductive

LPF Low Pass Filter

MOSFET Metal Oxide Semiconductor Field-Effect Transistor

MSC Mechanically Switched Capacitor

MSR Mechanically Switched Reactor

NCO Numerically Controlled Oscillator

PCB Printed Circuit Board

PROM Programmable Read Only Memory

PWM Pulse Width Modulation

RC Resistive-Capacitive

RL Resistive-Inductive

RMS Root Mean Square

STATCOM Static Synchronous Compensator

SVC Static Var Compensator

TCR Thyristor Controlled Reactors

THD Total Harmonic Distortion

TSC Thyristor Switched Capacitors

TSR Thyristor Switched Reactors

UPS Uninterruptible Power Supply

VA Volt-Amps

Var Volt-Amps Reactive

VAC Volts AC

VDC Volts DC

VSC Voltage Source Converter

VSI Voltage Source Inverter

VSD Variable Speed Drive

W Watts

1602 16 Columns by 2 Rows

2004 20 Columns by 4 Rows

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Acknowledgements

I would like to extend my most sincere acknowledgement and thanks toward Craig Carter for his

assistance, guidance and patience throughout the duration of this project. In addition, I would like

to thank the staff within the School of Engineering and Information Technology at Murdoch

University who assisted and facilitated my journey through this undergraduate degree.

Furthermore, I would like to acknowledge my friends and family, in particular I would like to give

my most heartfelt thank you to my Mother Bev, Father Martin, Brother Cameron and partner

Caitlyn Morrison for their untiring assistance and support for whom without this support my

degree would not have been possible. I would like to acknowledge my fellow engineering

students for offering assistance when needed and making the project room of Murdoch

University an enjoyable box with one window, particularly, Adam Gioffri, Jade Sciberras and

Michael Colson.

- Jordan Goodchild

January 2016

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1. Introduction

Within a transmission and distribution system there are countless factors affecting the quality

of the power transmitted. These factors include variation from non-linear loads, losses within

the transmission lines and harmonics introduced into a system. Electrical power quality is an

essential component for effective transmission and distribution of power to users.

The resulting effects these aforementioned factors have on power quality invoke voltage

sagging or swelling, reduced power factor, introduced harmonic noise and overheating [1]

within inductive motors. A solution comes from introducing a static synchronous

compensator (STATCOM) into a system with which these power quality issues can be

effectively managed and reduced.

Whilst STATCOMs are not directly introduced in any engineering units at Murdoch University,

the basic concepts surrounding a STATCOM are discussed in various levels of detail. Within

ENG349, ENG243 and ENG348 the concepts of voltage source inverters, power factor

correction and various voltage networks are introduced. The STATCOM will enable students

to visually see the combination of multiple engineering processes acting together in unison to

produce a device capable of various tasks.

Although this device would typically be aimed towards third year engineering students who

have supervised access to systems with voltages around 440 VAC, this device is designed to

operate at a far lower voltage around 12 VRMS.

The project focuses toward the design and construction of a low voltage static synchronous

compensator with the end users being third year engineering students. The purpose is enable

student’s practical exposure to a unique power quality controller. The specifications for this

project are for the STATCOM control to be switchable between voltage or power factor

2

control. In addition, the system data should be displayed at a reasonable rate for students to

visualise the changes the STATCOM has on the system.

The overall project goal is to produce a working prototype of the STATCOM that can be

implemented into a classroom environment.

1.1. Commercially Available Solutions

One prominent commercially available product is produced by Festo Didactic Inc. [2] which

purchased the company formerly known as LabVolt. The product available is a combination of

components that represent a STATCOM. These components consist of various LabVolt racks

most notably the Data Acquisition and Control Interface, Model 9063 and the IGBT

Chopper/Inverter, Model 8837-B.

These components can range upward of $300 AUD each. Although the university already has

some LabVolt equipment the additional cost to purchase these modules may outweigh their

value making this a non-viable solution.

3

2. Literature Review

The purpose of this literature review is to introduce the fundamental concepts involved

behind the design and implementation of a static synchronous compensator (STATCOM). The

intention of this information is to give a reader (who is assumed to have a basic knowledge of

electrical power) the ability to understand the design stages of this project. However, it

should not be regarded as an all-encompassing reference for the entire topic.

2.1. DC – AC Converters

The core objective of a static power converter is to generate an Alternating Current (AC)

sinusoidal output waveform by means of a Direct Current (DC) power supply. The applications

of this waveform generation extends into and beyond the functions of variable speed drives

(VSDs), uninterruptible power supplies (UPS), Static Var Compensators (SVCs), Flexible

Alternating Current Transmission Systems (FACTS), and voltage compensators. Typically, for

sinusoidal AC outputs control should be applicable to the magnitude, frequency, and phase of

the waveform. [3]

In order to transform a DC voltage to a sinusoidal AC voltage similar to that observable at the

mains outlet of a household a converter is required. These converters are referred to as

inverters and as the name suggests inverts a DC voltage into an AC voltage. Whilst this

chapter will focus on the sinusoidal style inverter there are numerous options commercially

available including square wave inverters and modified sine-wave inverters.

For this project the decision was to focus on a switch-mode DC-AC converter. The input to the

inverter will be assumed to be a DC voltage source with a DC Capacitor in parallel [4]. This

style of inverter is denoted as a Voltage Source Inverter (VSI). These VSIs can be further

divided into the following three categories:

4

1. Pulse-Width-Modulated (PWM) Inverters.

2. Square-wave inverters.

3. Single-phase inverters with voltage cancellation. [4]

For this report and the concepts discussed in this chapter the third category has been

ignored. These devices have been ignored as the voltage cancellation technique is only

applicable to single phase. If this project is scaled up to a three phase system, these devices

would become redundant. Additionally, the added complexity of introducing active common

mode voltage cancellation into the project would be an undesirable for any advantages that

may result.

Additionally, a voltage source inverter can be achieved through many configurations.

Typically, they are designed around full bridge inverters. Within this chapter a basic

introduction to half bridge and full bridge inverters will be discussed.

2.1.1. Half Bridge Inverters

In order to better understand the functionality of a full bridge inverter an investigation into

half bridge inverters is useful. A half bridge inverter is a simple device which consists of two

switching mechanisms. Typically, a semiconductor device acts as the switches connected in a

cascade arrangement along with two series capacitors. Figure 2-1 (left) below shows the

5

topology of a half bridge inverter.

Figure 2-1. half bridge inverter (Left). Output waveform (Right). [5]

Capacitors C1 and C2 are equal value capacitors which are series connected across the DC

source. Each capacitor will represent half of the input source voltage ½ VDC. By design,

sufficiently large capacitance should be used to ensure the potential midpoint of the

capacitors remains constant with respect to the negative bus [4].

By means of applying a square wave switching scheme to switches X1 and X2, Table 2-1 details

the resulting square-wave output waveform, an example of the output waveform can be seen

above in Figure 2-1 on the right. There is a possibility of four switching states for this

topology. One of these four states is undesirable and will result in a short circuit of the DC

input voltage. Whilst included in Table 2-1 this state should be avoided entirely.

Table 2-1. Half Bridge Switching Scheme

Output Voltage Switch X1 Switch X2

+ ½ VDC On Off

- ½ VDC Off On

0 Off Off

Short Circuit On On

This switching style is the most basic version applicable. By switching X1 and X2 on and off

complementary to each other at a rate of half the desired output period the fundamental

frequency can be achieved. The resulting duty cycle for this approach will be D=0.5.

6

Furthermore, it can be shown from substituting D=0.5 into Equation 2-1 below that the DC

component of the waveform is zero.

𝑉𝑎𝑣𝑔 = (2𝐷 − 1)𝑉𝑖𝑛 Equation 2-1 [6]

It is possible to apply more complex switching schemes and filtering to a half-bridge inverter

to achieve a more sinusoidal output only at a fixed frequency though. Nonetheless, for the

purpose of this report and project these switching schemes will not be discussed for a half-

bridge inverter. A complex switching scheme can be applied when using cascaded half-

bridges, [7] is an example of this application.

2.1.2. Full Bridge Inverters

A full bridge inverter is often referred to as an H-Bridge Inverter. This referral comes from the

physical resemblance to the capital letter H that the components and load form. Figure 2-2

shows this configuration with an example of the output waveform.

Figure 2-2. Basic H-Bridge Inverter Topology (Left), Output Waveform (Right) [8]

A full bridge inverter consists of two legs. Each leg of the inverter contains two cascaded

switching devices typically, MOSFETs or IGBTs with respective anti-parallel diodes [4]. A

7

distinct advantage of the full bridge inverter is that it has the capability to produce a

maximum output voltage twice that of a half bridge inverter with the same DC input voltage.

A variety of switching schemes can be applied to a full bridge inverter to achieve a desired

output waveform. The most basic method for switching an H-Bridge is to switch opposing

switches (X1 and X4) complementary to the other opposing switches (X2 and X3). Figure 2-3 is

an example of this basic switching scheme. As previously mentioned this basic switching

scheme is referred to as a square-wave switching scheme, in which each leg of the inverter is

active for one half-cycle (180°) of the desired output frequency. [4]

Figure 2-3. Basic H-Bridge Switching Scheme [9]

Similar to a half bridge inverter there are four basic modes (+V, -V, 0 and Short Circuit). As

previously mentioned if two switches in the same leg are closed simultaneously a short circuit

is created. This operation should be avoided entirely. Practically, a period known as the

blanking time should be put in place between switch transitions to avoid this short circuit

8

condition. For the purpose of this report covering ideal scenarios the blanking time can be

ignored [10].

From this, Table 2-2 details the switch positions and corresponding output voltages for the

basic switching scheme.

Table 2-2. H-Bridge Switching Positions

Output Voltage Switch 1 Switch 2 Switch 3 Switch 4

VDC On Off Off On

- VDC Off On On Off

0 Off Off Off Off

0 On Off On Off

0 Off On Off On

Short Circuit On On Off Off

Short Circuit On On On On

Short Circuit Off Off On On

The project focus will be driven toward the usage of a full bridge inverter acting as the

STATCOM VSI. In order to achieve a sinusoidal output voltage waveform, a more complex

method of control will be applied to the full bridge switches. Chapters 2.2 will detail the two

styles of Pulse Width Modulation (PWM) switching schemes; Bipolar and Unipolar

respectively.

2.2. PWM Switching Scheme

To produce a sinusoidal output voltage waveform with control of the magnitude and

frequency a PWM switching scheme is used. The output frequency control results from

comparing a sinusoidal waveform with a triangular waveform. Figure 2-4 illustrates this

waveform comparison. The triangular waveform 𝑣𝑡𝑟𝑖 (blue) shown is referred to as the carrier

frequency with a switching frequency 𝑓𝑠. This switching frequency is the frequency at which

the bridge will operate. The other waveform is the fundamental frequency that the inverter

9

output voltage will represent. The control signal 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙 (red) is used to modulate the switch

duty ratio and has a frequency 𝑓1. [4]

Figure 2-4. PWM Switching Scheme Carrier (Blue) and Control (Red) waveforms [9]

Generally, the amplitude of the carrier 𝑣𝑡𝑟𝑖 remains constant with the control signal’s

magnitude 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙 variable. These two variables are used to control the amplitude of the

output waveform through Equation 2-2 which represents the amplitude modulation.

𝑚𝑎 =𝑐𝑜𝑛𝑡𝑟𝑜𝑙

𝑡𝑟𝑖

Equation 2-2 [4]

Notably, the amplitude modulation ratio can take the form of any value. However, for ease

of understanding the effect of this ratio it is broken into three sections linear, over-

modulation and square-wave respectively represented as:

𝑚𝑎 ≤ 1.0

1.0 < 𝑚𝑎 < 3.24

𝑚𝑎 ≥ 3.24

The modulation ratio will determine the output voltage magnitude for the case of the

𝑚𝑎 ≤ 1.0 the magnitude increase will operate in a linear arrangement. Conversely, for

over-modulation there is a non-linear increase in the output voltage. If the modulation

Voltage Amplitude

(V)

Time (ms)

10

increases above a value of 3.24, the output waveform will become a square-wave of

relatively fixed magnitude. Figure 2-5 details a graph of these three sections. Figure 2-6

depicts an example of the waveform comparison for over-modulation. This example is

bordering on square-wave operation with 𝑚𝑎 = 3.0.

Figure 2-5. Voltage Control by Varying mA [4]

11

Figure 2-6. Over-modulation, Borderline Square-Wave Operation [9]

Whilst it is desirable to increase the output voltage using over-modulation there are

drawbacks. Most significantly, as previously mentioned, the amplitude does not increase

linearly. Additionally, over-modulation will result in an increase in the harmonics in the

output waveform. Equation 2-3 underneath, adapted for full bridge inverters, shows the

peak output voltage with respect to the amplitude modulation ratio when operating in the

linear range.

𝑜𝑢𝑡 = 𝑑𝑚𝑎 Equation 2-3 [4]

For a full bridge inverter the switches are controlled based on the comparison of 𝑣𝑡𝑟𝑖(𝑡)

and 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙(𝑡) . This control is performed by a series of pulses which are generated

dependent upon the voltage level of the control waveform in comparison to the voltage

level of the carrier wave. Table 2-3 shows the comparator output for the comparison levels.

Table 2-3. Comparator Switching Scheme

Comparison Comparator Output

𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 > 𝒗𝒕𝒓𝒊 High

𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 ≤ 𝒗𝒕𝒓𝒊 Low

Voltage Amplitude

(V)

Time (ms)

12

Figure 2-7 details an example of how these pulses are generated based on the comparison

of the two waveforms. The spacing of the control pulses is dependent upon the frequency

modulation, which defines the ratio between the carrier frequency 𝑓𝑠 and the control signal

frequency 𝑓1. The formula for frequency modulation shown below in Error! Reference

source not found..

𝑚𝑓 =𝑓𝑠

𝑓1 Equation 2-4 [4]

Figure 2-7. Control Pulse Generation Example

Figure 2-8 details an example of the output voltage waveform from the comparison

between 𝑣𝑡𝑟𝑖(𝑡) and 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙(𝑡) at an amplitude modulation of 0.5 and a frequency

modulation of 30.

Voltage Amplitude

(V)

Time (ms)

13

Figure 2-8. PWM Switching Scheme Output Waveform; Unfiltered (Blue), Filtered (Red) [9]

It is desirable to use as high a frequency modulation ratio as possible. A major drawback to

increasing the switching frequency is that the switching losses in inverter switches increase

proportionally to the switching frequency [4]. These losses are attributed to the turn-on

and turn-off losses with every switching cycle. As the frequency increases there are a

greater number of these on-off events and thus greater losses. Typically, in applications

using a PWM switching scheme the switching frequencies are chosen to be below six kHz or

above 20 kHz. From Mohan; Power Electronics: Converters, Applications and Designs

Chapter 8 the value of 𝑚𝑓 has been divided into two categories; 𝑠𝑚𝑎𝑙𝑙 𝑚𝑓 ∶ 𝑚𝑓 ≤ 21 and

𝑙𝑎𝑟𝑔𝑒 𝑚𝑓 ∶ 𝑚𝑓 > 21. [4] Whilst these 𝑚𝑓 ranges are an arbitrary assumption this report

will continue to use these assumptions.

For small 𝑚𝑓 values the triangular carrier and the sinusoidal control signal should be

synchronised. This synchronisation can be seen above in Figure 2-7 in which the zero point

of both waveforms align. This synchronisation requires that 𝑚𝑓 be an integer. This integer

𝑚𝑓 will reduce the undesirable sub-harmonics. Additionally, 𝑚𝑓 should be an odd integer

except in the case of single-phase inverters using Unipolar PWM switching [4].

Voltage Amplitude

(V)

Time (ms)

14

Conversely for large values of 𝑚𝑓 values can be asynchronous. The reason behind this is

that at larger values of 𝑚𝑓 the harmonics present are significantly smaller. Although

asynchronous PWM will introduce larger harmonic currents into a system which is greatly

unwelcome. Therefore, even at larger 𝑚𝑓 values synchronous PWM should be the option

considered.

2.2.1. PWM with Bipolar Voltage Switching

Given the output waveform switches between a positive and negative DC voltage this

switching scheme is called Bipolar PWM. Bipolar PWM switching was illustrated within

section 2.2, however an additional and brief explanation will be detailed within this section.

The basic method employs an individual triangle carrier waveform compared with a

sinusoidal control signal. For this approach, the opposing upper and lower switches of each

inverter leg are switched as pairs.

Figure 2-9 details all signals associated with Bipolar PWM. The first waveforms detail the

input waveforms into the comparator. The second and third waveforms detail the control

signals sent to the opposing gates. Finally, the bottom illustration shows the total output

waveform both filtered (red) and unfiltered (blue). This simple PWM switch scheme

requires minimal components to perform. Essentially this method can be achieved with a

comparator and NOT gate.

15

Figure 2-9. Bipolar PWM Switching Scheme [11]

2.2.2. PWM with Unipolar Voltage Switching

In this type of PWM scheme when switching occurs the output voltage changes between

zero and +𝑉𝐷𝐶 or between zero and −𝑉𝐷𝐶 [4]. This is how the PWM scheme is given the

name unipolar switching.

Unipolar PWM switching differs from Bipolar PWM switching in that the triangular carrier

waveform is compared with two sinusoidal control signals. These control signals are 180°

out of phase from one another fundamentally represented as 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙 (red) and −𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙

(green). This is depicted in the top illustration of Figure 2-10.

Table 2-4 shows the comparator output signals for the comparison of the carrier with

𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙 or −𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙.

Table 2-4. Unipolar PWM Comparator Control Signals

Comparison Comparator Output

𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 𝒐𝒓 − 𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 > 𝒗𝒕𝒓𝒊

𝐻𝑖𝑔ℎ

𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 𝒐𝒓 − 𝒗𝒄𝒐𝒏𝒕𝒓𝒐𝒍 ≤ 𝒗𝒕𝒓𝒊

𝐿𝑜𝑤

Voltage Amplitude

(V)

Time (ms)

16

The opposing control signals for the inverter are shown within the second and third

diagram of Figure 2-10. The overall output waveform both filtered (red) and unfiltered

(blue) shown above in the last illustration of Figure 2-10 differs significantly from that of

the bipolar PWM output.

Figure 2-10. Unipolar PWM Switching Scheme [11]

Figure 2-11 detailing a basic full bridge inverter will demonstrate how the output voltage

waveform is generated.

Figure 2-11. Basic Full Bridge Inverter

Table 2-5 details how, when opposing switches S1 and S4 or S2 and S3 are high, the

output voltage appears as VDC or –VDC respectively. However, when parallel switching

devices S1 and S3 or S2 and S4 are simultaneously high or low the output waveform

appears as zero volts.

17

Table 2-5. Unipolar PWM VSI Output Control Signals

Bridge Switch Positions Output Voltage

1 2 3 4

High Low Low High 𝑉𝐷𝐶

Low High High Low −𝑉𝐷𝐶

High Low High Low 0

Low High Low High 0

A significant advantage of unipolar PWM compared with bipolar PWM switching is that

the switching frequency is essentially doubled. This advantage will reduce the noise and

harmonics present in the filtered waveform.

For this project, a modified bipolar switching scheme has been utilised on the full bridge

inverter. Whilst the unipolar PWM switching scheme is a more ostentatious solution, the

requirement to produce an inverted sine wave would require more complex components

and circuits. These would consist of an operational amplifier and a split rail power supply

to power this amplifier circuit. This increase in equipment outweighs the added

advantage of a better output waveform an improved STATCOM filter can achieve similar

results.

2.3. Flexible Alternating Current Transmission Systems

Flexible Alternating Current Transmission Systems have become essential to transmission

and distribution networks to assist in alleviating the reduced quality power supply from

the resultant increase of demand on the lines. FACTS technology allows for a new

spectrum of methods for controlling power and increasing the present capabilities of

existing lines.

18

2.3.1. Static Shunt Compensators

Static Shunt Compensators are typically known as Static Var Compensators (SVCs) or

Static Synchronous Compensators (STATCOMs). An SVC is an umbrella reference for

numerous devices such as:

Thyristor Controlled Reactors (TCRs)

Thyristor Switched Reactors (TSRs)

Thyristor Switched Capacitors (TSCs)

Mechanically Switched Capacitors/Reactors (MSCs or MSRs).

Examples of these devices can be seen below in Figure 2-12. The purpose of these shunt

compensation devices are to influence the electrical qualities of the transmission line to

increase the steady-state transmittable power and/or to control the voltage profile along

the line. [12]

Figure 2-12. Static Shunt Compensator Diagrams [13]

In addition to Static Shunt Compensators there are also Static Series Compensators. These

devices are run parallel to a series reactive component as shown below in Figure 2-13. For

this report Static Series Compensator devices won’t be discussed or considered.

19

Figure 2-13. Static Series Compensator Examples [14]

For the purpose of this report the overall SVC operational characteristics will briefly be

discussed within this chapter however TCRs, TSRs and TSCs will not be discussed in any

further detail. For a more comprehensive detail into these devices the following references

should be consulted [12], [15], [16], [17].

2.3.1.1. Static Var Compensators

Throughout the duration of any given day, the electrical loads present on a transmission

line will either absorb and generate reactive power. Given this variation of loads the

reactive power balance on a grid will vary considerably. The end result from this variation

can take the form of intolerable voltage amplitude variations, a voltage depression or even

a voltage collapse. [18]

The IEEE definition of a Static Var Compensator is as follows:

A shunt-connected static var generator or absorber whose output is adjusted to exchange

Source

Source Load

Load

20

capacitive or inductive current so as to maintain or control specific parameters of the

electrical power system (typically bus voltage). [12]

Dynamically an SVC has the capability to control voltage swings, provide support for

reactive power demand and reduce power oscillations thereby improving the performance

and quality of the transmission and distribution network supply. These control strategies

are achieved through the SVC’s capacity to vary the value of the shunt compensation

applied to the transmission line as well as affect the power flow of the system.

The placement of an SVC will vary the effective conditions on the transmission line. There

are several options to be considered when designing an SVC. Generally, the point of

connection for the SVC is the point at which the voltage is to be controlled. If the SVC is

situated at the middle of the transmission line the voltage at this point can be controlled

such that it has the same value as the end line voltages [12]. The advantage of this

placement is that the power transmission across the system is increased. Conversely, the

SVC can be located parallel to the load at the end of the line. This placement will enable the

device to stabilise the voltage caused by variations in the load, line outages or variations

from generators. Furthermore, incorporating multiple SVCs into a system will allow the

network to increase transfer capability and reduce losses while maintaining a smooth

voltage profile under different network conditions [18].

2.3.1.2. STATCOM

A Static Synchronous Compensator is defined by the IEEE as:

A static synchronous generator operated as a shunt-connected static var compensator

whose capacitive or inductive output current can be controlled independent of the AC

system voltage. [12]

21

Furthering this definition, a STATCOM has the capacity to act as a controlled reactive-

power source delivering the network voltage support by means of generating or absorbing

reactive power. Figure 2-14 shows a Voltage Source Inverter-based configuration for a

STATCOM.

Figure 2-14. STATCOM Schematic Example [12]

The control strategy for a STATCOM is dependent upon the amplitude of the output voltage

in relation to the network voltage amplitude. Moreover, the amplitude control can be

divided into three (3) sections using the format from Figure 2-14 these sections consist of:

1. VSTATCOM < VLINE

2. VSTATCOM > VLINE

3. VSTATCOM = VLINE

For the first case VSTATCOM < VLINE, where the amplitude of the AC system line voltage VLINE is

greater than the STATCOM voltage VSTATCOM, a lagging current results and the STATCOM is

seen as an inductor. In this case, reactive power is absorbed [12]. Following on from this, in

case two VSTATCOM > VLINE where the amplitude of the line voltage is below that of the

amplitude of the STATCOM voltage the resulting current is leading. In the event of a leading

22

current the system sees the STATCOM as a capacitor and in-turn generates reactive power.

For the third instance when the amplitudes of both voltages are equal the system exchanges

no power as ideally no current will flow.

In reality a VSI is not a lossless converter; the switching elements within the VSI will have a

small amount of resistance. In the case of this project, the MOSFETS used will have on and off

resistances, typically of the order of mΩ. In conjunction other internal and external losses will

occur within the converter. A solution is to have the output voltage of the converter lag the

AC system by a slight amount in order to absorb a small amount of active power to balance

the losses.

2.4. Arduino Platform

Arduino was originally developed and co-founded in Italy by Massimo Banzi at the Ivrea

Interaction Design Institute [19] in 2005. The purpose was to encouraging students to begin

developing skills in electronics and programming. Arduino is an open-source programming

platform designed around the idea of implementing easy to use hardware and software. The

Arduino programming language is based around Wiring [20] an open-source programming

language which was developed by Hernando Barragán in 2003 [21]. Additionally, the software

known as the Arduino IDE (currently 1.0.5) is based around a system called Processing [22].

Processing was co-developed in 2001 by Ben Fry and Casey Reas [23].

An example of the physical Arduino platform is the Arduino Uno shown below in Figure 2-15.

This is an all-inclusive printed circuited board (PCB) with a microcontroller housed around I/O

connectors and other complementary circuitry. Typically, these microcontrollers are based on

23

the Atmel AVR microcontroller an example of this is that of the 8 bit ATmega328P (Figure

2-16) with 32KB of on board program (Flash) memory [24].

Figure 2-15. Arduino Uno/Genuino Microcontroller [25]

Figure 2-16. Atmega 328P PDIP Chip Pin Layout [26]

A vast range of Arduino boards are commercially available. These boards range in a variety of

sizes and prices such as the Arduino Nano, Arduino Uno, Arduino Mega and Arduino Due.

Initially the project design was to use either the Arduino Uno or the Arduino Mega (Figure

2-17). These microcontrollers were considered over others in the Arduino range because of

their ability to connect to external adapters known as “shields”. These shields are directly

24

pluggable and stackable adapters built for singular purposes such as a motor controller shield

or an RGB lighting shield.

Figure 2-17. Arduino Mega/Genuino Mega Microcontroller [27]

After considerations, the Arduino Uno was chosen for the project. Whilst the Arduino Mega

notably has more Digital I/O pins than the Arduino Uno and thus capability the size and

cost of the microcontroller board for this project was unnecessary. The Arduino Uno was at

an affordable cost of £20.00 + VAT (approximately $40 AUD [28]) as of 28th of December

2015. Additionally, the Arduino Uno microcontroller was sufficiently capable for the

project. Table 2-6 details the specifications of the Arduino Uno.

Table 2-6. Arduino Uno Specifications [29]

Microcontroller ATmega328P

Operating Voltage 5 Volts

Input Voltage (recommended) 7-12 Volts

Input Voltage (limit) 6-20 Volts

Digital I/O pins 14 (of which 6 are PWM output)

Analog Input Pins 6

DC current per I/O pin 20 mA

Clock Speed 16MHz

The Arduino microcontroller makes up the basis of the measurement instrument as well as

producing the controllable triangular waveform used to produce the PWM outputs. The

triangular waves will be discussed within Chapter 3.5.1. The triangular waves are based on

25

the open-source code produced by Martin Nawrath’s Direct Digital Synthesis generator [30]

discussed in Chapter 2.5.

2.5. Direct Digital Synthesis

Direct digital synthesis (DDS) is a technique of generating an analogue waveform, typically, a

sine wave, by generating a time-varying signal in digital form and then performing a digital-to-

analog conversion [31], generally through a low-pass filter.

Currently there are a wide variety of commercially available chipsets that are capable of

performing DDS waveform generation. Figure 2-18 the Analog Devices AD9833 is an example

of this chipset. DDS devices like the AD9833 are programmed through a high speed serial

peripheral-interface (SPI; Pins 6, 7 and 8) and need only an external clock to generate simple

sine waves. [31]

Figure 2-18. AD9833 Pin Layout [31]

Other DDS devices commercially available are capable of producing waveforms of

frequencies from less than 1 Hz through to 400MHz. The advantages of these chipsets are

their reliability, size and cost. Despite these attractive advantages this project will utilise

the internal clock of the Arduino Uno microcontroller and code to produce the DDS output

waveform. As previously mentioned this method is based on the open-source code

available from Martin Nawrath’s DDS generator at [30].

MCLK

26

Figure 2-19 details basic architecture of a DDS generator. An adequate amount of digital

phase-to-amplitude data that corresponds to one period of an arbitrary waveform is stored

into programmable read-only-memory (PROM). This PROM acts as the LOOKUP TABLE. A

regular clock signal drives the address counter which in-turn steps through and accesses

each of the PROM’s memory locations. The contents are then presented to a high-speed

Digital to Analog Converter (DAC) [32]. In the case of this project the DAC and the Low-Pass

Filter (LPF) are one and the same.

Figure 2-19. Fundamental Architecture of a DDS Generator [33]

The output performance of this basic DDS generator is acceptable for basic single

frequency systems, as this method lacks the flexibility of easily adjusting the output

frequency. The output frequency can be altered by changing the frequency of the reference

clock or reprogramming the PROM. Whilst these options are achievable, they do not allow

for high-speed output frequency changes.

A solution to this inflexibility is to modify the DDS generator architecture. Figure 2-20

details this adapted architecture.

27

Figure 2-20. Frequency Tuneable DDS Generator [32]

The main components of the above frequency tuneable DDS generator are the Phase

Accumulator, the DAC and the phase-to-amplitude converter (typically, this is a lookup

table as previously discussed). By introducing a phase accumulator function, this

architecture resembles a numerically controlled oscillator (NCO) which is the heart of a

flexible DDS system [32].

The output frequency fOUT is dependent upon two variables: the reference-clock frequency

and the binary number programmed into the frequency register (tuning word M) [31]. The

main input into the phase accumulator is the tuning word. If a look-up table is used the

phase accumulator computes a phase (angle) address for the look-up table which outputs

the digital value of amplitude corresponding to the sine of that phase angle to the DAC

[31]. The D/A converter and LPF are one and the same in this case converts the digital

pulses from the phase-to-amplitude converter into an analog voltage output waveform. For

the Arduino this waveform will have an ideal value ranging from 0-5 Volts.

In order to produce a constant frequency output waveform the phase increment which is

determined by the turning word is summed to the phase accumulator each clock cycle. If

the phase increment is large the phase accumulator will step through the look-up table

quickly and thus generate a higher frequency waveform. Conversely, if the phase increment

is small the phase accumulator will take many more steps accordingly generating a slower

8 BITS

DIGITAL TO ANALOG CONVERTER

28

waveform [31]. Fundamentally this can be observed in both Figure 2-21 and Equation 2-5.

For a larger tuning word M the output frequency will be correspondingly larger. As well in

Figure 2-21 the larger M value rotates through the phase wheel faster thus producing a

higher frequency waveform.

𝐹𝑂𝑈𝑇 =

𝑀 × 𝑓𝑐

2𝑛 Equation 2-5. [31]

Figure 2-21. Phase Wheel showing Phase Accumulator Register M = 1 [24]

For Martin Nawrath’s code the tuning word is defined by:

const double refclk=31372.549;

dfreq = 5000.0; // initial output frequency = 5000.0 Hz

tword_m = pow(2,32)*dfreq/refclk; // calulate DDS new tuning word

In this case Equation 2-5 is re-arranged to solve for the tuning word M. In addition to this

the code also supplies a sine wave lookup table, although this has been modified to

produce a triangular output waveform for this project the operation is the same. This code

can be seen below.

29

// table of 256 triangle values stored in flash memory

PROGMEM prog_uchar tri256[] = 0x2,0x4,0x6,0x8,0xa,0xc,0xe,0x10,0x12,0x14,0x16,0x18,0x1a,0x1c,0x1e,0x20,

0x22,0x24,0x26,0x28,0x2a,0x2c,0x2e,0x30,0x32,0x34,0x36,0x38,0x3a,0x3c,0x3e,0x40,



0x81,0x83,0x85,0x87,0x89,0x8b,0x8d,0x8f,0x91,0x93,0x95,0x97,0x99,0x9b,0x9d,0x9f,

0xa1,0xa3,0xa5,0xa7,0xa9,0xab,0xad,0xaf,0xb1,0xb3,0xb5,0xb7,0xb9,0xbb,0xbd,0xbf,

0xc1,0xc3,0xc5,0xc7,0xc9,0xcb,0xcd,0xcf,0xd1,0xd3,0xd5,0xd7,0xd9,0xdb,0xdd,0xdf,

0xe1,0xe3,0xe5,0xe7,0xe9,0xeb,0xed,0xef,0xf1,0xf3,0xf5,0xf7,0xf9,0xfb,0xfd,0xff,

0xfd,0xfb,0xf9,0xf7,0xf5,0xf3,0xf1,0xef,0xed,0xeb,0xe9,0xe7,0xe5,0xe3,0xe1,0xdf,

0xdd,0xdb,0xd9,0xd7,0xd5,0xd3,0xd1,0xcf,0xcd,0xcb,0xc9,0xc7,0xc5,0xc3,0xc1,0xbf,

0xbd,0xbb,0xb9,0xb7,0xb5,0xb3,0xb1,0xaf,0xad,0xab,0xa9,0xa7,0xa5,0xa3,0xa1,0x9f,

0x9d,0x9b,0x99,0x97,0x95,0x93,0x91,0x8f,0x8d,0x8b,0x89,0x87,0x85,0x83,0x81,0x80,

0x7e,0x7c,0x7a,0x78,0x76,0x74,0x72,0x70,0x6e,0x6c,0x6a,0x68,0x66,0x64,0x62,0x60,



0x1e,0x1c,0x1a,0x18,0x16,0x14,0x12,0x10,0xe,0xc,0xa,0x8,0x6,0x4,0x2,0x0,

; // end of lookup table

Figure 2-22. Lookup Table of One Triangular Waveform Period

30

3. Project Design

This chapter of the report will detail how the project was designed. The design approach will

incorporate the above fundamentals discussed in Chapter 2. The objective of this section is to

enable a reader continuing or recreating this project research to clearly follow the previously

completed sections.

3.1. Power Supply

This section of Chapter 0 will detail the decision behind how the power supplies were sized

and chosen appropriately. For a full set of wiring diagrams for both power supply

connections, Full Circuit Drawings should be referred to particularly DWG 000-001.

3.1.1. DC Power Supply

The purpose of the DC supply for this project is to provide a dedicated supply for the Arduino

microcontroller, Integrated Circuits (ICs) and supply power to the cooling fans. From the

Arduino website [34] the Arduino Uno has a recommended supply voltage of 7-12 Volts DC.

By design, the current consumption of the Arduino board and cooling fans played a factor in

the decision of the DC supply. The Arduino consumes around 50mA with each I/O pin adding

up to 20mA when connected to a load that requires this level of current such as a light

emitting diode. From this information, the approximate maximum current that will be drawn

from the Arduino Uno is 500mA. Adding in the DC fans consumption and the consumption of

any ICs used in the project the overall current is approximately 800mA.

Utilising this specification, numerous options were explored for the DC supply. Initially, a DC

plug pack was considered. Whilst using a DC plug pack rated at one Ampere is a viable option

the need to have loose plug packs and power boards transported with the project constantly

31

became tedious. Additionally, plug packs can open the project up to failure in the event a

larger DC voltage supply or AC voltage supply was connected instead of the intended supply.

From this the option was to allow the AC supply (discussed in Section 3.1.2) provide the DC

power through rectification and a voltage regulator. However, this was quickly rejected as it

was not an effective solution and reduced the current supply capabilities of the AC portion of

the project.

Overall, the DC supply chosen was the M8730 25W 12 VDC switch-mode power supply sourced

from Altronics. Figure 3-1 shows the DC supply. This supply will provide around 2.08 Amps of

current. Whilst this is well above the required amount this will allow for future additions as

well as account for discrepancies and losses. In order to provide a level of safety for the

project an external 1.5A quick blow fuse was added to prevent any accidental short circuits or

larger current draws from causing damage to the equipment or persons.

Figure 3-1. AC (bottom) and DC (top) power supply box

32

3.1.2. AC Power Supply

Initially the project was designed around the use of a 24 VAC supply, however upon

investigation it was decided that the voltage of the system was to be reduced from 24 to 12

Volts RMS. This decision was attributed to the need to allow for a safety margin for the

components. By sizing the AC power supply to this lower level, the end user can have the

system account for voltage increases from either from the mains supply or from any increase

in voltage via the STATCOM.

In previous deliverables for this thesis project it had been discussed that the power supply

could consist of one of various options ranging from an E-I core plug pack through to an

amplifier fed from a function generator. However, after exploring multiple options it was

decided that the most realistic and viable solution was to utilise a M5012A 12 + 12 50VA

Toroidal Transformer sourced from Altronics, which can be seen in the above Figure 3-1.

The advantages of using the toroidal transformer over an E-I core plug pack with external

connecters are the reduced electromagnetic noise, which is more heavily induced by an E-I

core transformer as well as the disadvantage of having multiple cabled plug packs.

The chosen transformer has the capability of the dual secondary windings being configured in

either parallel or series to provide different voltage and current levels. By connecting the

secondaries in parallel the transformer is capable of supplying 12 VAC at 4.1 A. This

configuration has been chosen to allow a sizeable overhead for the project.

The 12 VAC will be supplied through an external quick blow fuse rated at 2 Amps. The fuse is

designated at this level based on calculations of approximately 1.3 Amps drawn by the lowest

impedance load whilst still allowing for an overhead for any unforeseen extra current draw.

33

3.2. Line and Load Bank

This section of the Chapter 0 details the design behind the load bank and line impedances. A

detailed drawing showing the connections for the Load Impedances can be found in Full

Circuit Drawings labelled DWG 200-001.

3.2.1. Load Design

The load bank (Figure 3-2) for this project will consist of a variable switch box housing 6

resistive loads, 6 inductive loads and 6 capacitive loads. Initially the design concept was to

install 12 loads for each variant. After investigation it was found that the cost of these loads

was too high with respect to user functionality and practicality.

Figure 3-2. Load Bank

The overall setup will result in the loads being parallel Resistive-Inductive (RL) or parallel

Resistive-Capacitive (RC). The design behind this load arrangement is to produce varying

leading or lagging loads at the end of the low voltage line and to see the response of the

STATCOM in either controlling the voltage at the load or correcting the load’s power factor.

34

For simplicity of starting the calculations the loads were assumed to range from 1 – 12 W and

±1 – 12 VARs. By assuming these start values and a nominal voltage of 12VAC calculations for

the load current and load impedances could be carried out.

Using these assumed values the following equations for apparent power(S), real power (P)

and reactive power (Q) will produce the values shown for the resistive loads in Table 3-1

below.

𝑉𝑅𝑀𝑆 = 𝐼𝑅𝑀𝑆𝑅 = 𝐼𝑍 Equation 3-1 [35]

𝑆 = 𝑃 + 𝑗𝑄 Equation 3-2 [35]

𝑆 = 𝑉𝐼 Equation 3-3 [35]

𝑃 = 𝑉𝐼𝑐𝑜𝑠∅ Equation 3-4 [35]

Assuming cos∅=1 representing a purely resistive load.

𝑃 = 𝑉𝐼 =𝑉2

𝑅 Equation 3-5 [35]

Rearranging to solve for R.

𝑅 =𝑉2

𝑃 Equation 3-6 [35]

Table 3-1. Resistive Load Calculations.

Real Power (W)

Calculated Resistance

(Ω)

1 144

2 72

3 48

4 36

5 28.8

6 24

7 20.57

8 18

9 16

10 14.4

11 13.09

12 12

35

From the calculated resistances and research on what sizes are readily available it was

decided that the highlighted six would be chosen as the resistive loads. From Table 3-1 the

lowest resistance is 12Ω. Using Equation 3-5 and the unloaded line voltage of 13.45 VAC the

calculated power dissipated within the resistor provides a rough estimation to the design of

the components. The resultant equation yields 15.08W. Commercially and readily available

resistors can be found to range in 10, 15, 25 and 50W sizes. Whilst 10W resistors would

work for the higher resistive loads uniformity is a factor in the design, therefore the 10W

size will be disregarded. This logic can also be applied to the 15W resistors. Although the

15W resistors would be viable they do not leave much flexibility for an increase in the

system current and therefore power dissipation capabilities. The resistors have been

selected to be 25W resistors as they give adequate flexibility for an increase in current as

well as satisfying the uniform design approach for the loads.

Table 3-2 details the resistances chosen from commercially available values. The design

approach was to use as few resistors as possible to achieve the desired values. The

exceptions are evident with two series resistors being used for the 3 and 5 Watt loads. In

addition, it should be noted that the 1 and 7 Watt load resistors are not exact to the

calculated resistances. These differences have been deemed negligible in the overall design

with the recalculated powers show in the last column.

It was suggested that the use of parallel resistor combinations could be used to closer

achieve the values for the 1 and 7 Watt loads. The example given was the 150Ω resistor in

parallel with two (2) series 1.8kΩ resistors resulting in 144Ω. Despite the fact that this is an

elegant and simple solution the resulting combination requires more resistors increasing

component footprint and cost.

36

Table 3-2. Recalculated Resistive Loads

Real Power (W)

Calculated Resistance (Ω)

Chosen Resistor(Ω) Resistor Tolerance

Recalculated Real Power

(W) Overall 1 2

1 144 150 ±5% 0.96

3 48 48 47 1 ±5% 3

5 28.8 28.8 27 1.8 ±5% 5

7 20.57 20 ±5% 7.2

9 16 16 ±5% 9

12 12 12 ±5% 12

Following these calculations the values for the inductors and capacitors were calculated

based on the similarly assumed principles of ±1-12 VARs and 12 VAC nominal. The equations

below produce the values seen in Table 3-3.

𝑉 = 𝐼𝑋 Equation 3-7 [35]

𝑄 = 𝑉𝐼𝑠𝑖𝑛∅ Equation 3-8 [35]

Assuming sin∅=1 for a purely reactive load.

𝑄 = 𝑉𝐼 =𝑉2

𝑋 Equation 3-9 [35]

Rearranging to solve for X.

𝑋 =

𝑉2

𝑄 Equation 3-10

Solving for the reactance in this case leads to two further equations based on whether the

load is designed to be capacitive or inductive. These equations are:

𝑋𝐶 =

1

𝜔𝐶=

1

2𝜋𝑓𝐶 Equation 3-11 [35]

𝐶 =

1

2𝜋𝑓𝑋𝐶 Equation 3-12

𝑋𝐿 = 𝜔𝐿 = 2𝜋𝑓𝐿 Equation 3-13 [35]

𝐿 =

𝑋𝐿

2𝜋𝑓 Equation 3-14

37

A further assumption for these equations is that the frequency is 50Hz.

Table 3-3. Reactive Load Results

Reactive Power (VAr)

Reactance (Ω)

Inductance (mH)

Capacitance (µF)

1 144 458.37 22.1

2 72 229.18 44.21

3 48 152.79 66.31

4 36 114.59 88.42

5 28.8 91.67 110.52

6 24 76.39 132.63

7 20.57 65.48 154.73

8 18 57.3 176.84

9 16 50.93 198.94

10 14.4 45.84 221.05

11 13.09 41.67 243.15

12 12 38.19 265.26

As a generalisation to achieve the desired design of 6 inductive loads and 6 capacitive

loads, the inductors have been selected as the odd VAr values and the capacitors as the

even VAr values. The calculated values shown above in Table 3-3 for both the inductors and

capacitors are a non-standard size set and will require some form of customisation.

For the design of this project the capacitors would need to be suitable higher voltage

bipolar capacitors. From inquiries into readily available bipolar capacitors it was found that

by utilising parallel combinations the values desired could be closely achieved. Once the

capacitor load bank was constructed the overall capacitance of each parallel combination

was measured. Table 3-4 details the comparison between the calculations, ideal and

measured values. As well it shows the parallel combinations of the capacitor values

designed to achieve the calculated values.

38

Table 3-4. Capacitor Value Comparison

Original Capacitance

(µF)

Purchased Capacitance

(µF)

Measure Capacitance

(µF)

Parallel Combination

44.21 44.00 41.80 22//22

88.42 88.00 83.90 22//22//22//22

132.63 132.00 126.20 100//22//10

176.84 176.80 170.90 100//47//22//6.8//1

221.05 221.00 204.00 220//1

265.26 265.0 246.00 220//22//22//1

One of the most notable variances in the above Table 3-4 is the measured capacitance.

Generally, capacitors are designed to have a slightly higher capacitance than the value stated

on the external packaging. This suggests that the micron Q1150A digital capacitance and

inductance meter [36] used to measure these results is uncalibrated or is faulty. Excluding

the final two capacitor combinations the difference in reactance is minimal which will in-turn

result in minimal difference of the original reactive VArs. Despite the discrepancy these

capacitor values will be utilised for the project as the consequence is negligible. Whilst it is

undesirable to have these variances the overall objective of the STATCOM is to correct the

loads power factor or line voltage. These values will not hinder this operation of the project.

Initial investigation into sourcing pre-made inductors was unsuccessful. The required

inductors would need to be custom made. This customisation was the approach undertaken

for the project. Unfortunately, the original inductors that were constructed were wrapped

around small iron powder cores. These cores were not designed to operate at the low 50 Hz

voltages and quickly became saturated when installed in the system causing the AC fuse on

the secondary side to blow.

After consultation with Craig Carter the approach chosen was to use a large toroidal

transformer core to prevent the inductors from saturating. The approach and calculations

used for the original inductors for wrapping copper wire around the toroidal cores was

employed.

39

Knowing that a toroid inductor has an inductance given by the following equation:

𝐿 =𝜇𝑁2𝐴

2𝜋𝑟 Equation 3-15 [37]

Where:

N = number of turns

A = cross sectional area of the core (m2)

µ = relative permeability of the core

r = toroid radius (m)

L = Inductance in Henry’s (H)

Re-arranging to solve for the number of turns.

𝑁 = √2𝜋𝑟𝐿

𝜇𝐴 Equation 3-16

𝜇 = 𝜇𝑅𝜇𝑂 Equation 3-17 [38]

𝜇𝑂 = 1.257 × 10−6

From the specifications available on [39] the following variables are known:

𝐴 = 3.182 × 10−3𝑚2

𝑟𝑎𝑣𝑔 = 37.2 × 10−3𝑚

In order to determine the µ for calculation purposes the process was to wrap a wire around

the toroid and measure the various inductances. By applying this method to several

different turns’ ratios 𝜇𝐴

2𝜋𝑟 could be calculated for each case and averaged. From this

average value in conjunction with the above values for area and radius it was possible to

solve for the permeability of the core. The resultant permeability yields a value that would

40

be indicative of a ferrite core [40]. This is a reasonable assumption as most cores for

transformers tend to consist of an iron ferrite core.

𝜇 = 3.99413 × 10−4

By using these known variables the number of turns required to produce each inductor can

be solved. Table 3-5 displays the recalculated inductances based on the number of integer

turns. The design for this new inductor is to exploit the style of a tapped transformer

utilising one copper wire wrapped round the core and tapped off at each desired point.

Table 3-5. Inductance Calculation Results

Original Inductance (mH)

Number of Turns

Re-Calculated Inductance (mH)

Difference (mH)

458.37 290 457.29 1.08

152.79 168 153.47 -0.68

91.67 130 91.89 -0.22

65.48 110 65.79 -0.31

50.93 97 51.16 -0.23

41.67 88 42.11 -0.44 .

The inductance was measured during and after construction of the toroid. Once the

inductors were fully constructed the inductance was recorded from measurements using a

micron Q1150A digital capacitance and inductance meter [36]. As noted above in the

discussion relating to the capacitor values the meter may not have been calibrated. This

uncalibrated meter is evident from Error! Reference source not found. Table 3-6 from the

measurement of the actual inductance values.

This inductor was hand wound and will have imperfections. Once the copper wire had been

taped down for insulation, the defects could have been accentuated by overlapping wire

causing shortened or reduced turns ratios. However, as mentioned previously, the

difference in the values is un-desirable but will not affect the operation of the STATCOM

function.

41

Table 3-6. Inductor Value Comparison

Original Inductance (mH)

Re-Calculated Inductance (mH)

Actual Inductance (mH)

458.37 457.29 400.0

152.79 153.47 129.8

91.67 91.89 76.5

65.48 65.79 58.2

50.93 51.16 45.2

41.67 42.11 34.1

3.2.2. Line Impedance Design

Once the power supply had been designed the unloaded voltage was measured to be around

13.45 VAC. From the design brief the line voltage set point was to be a 12 VAC line feeding the

load. Thus the voltage drop across the line is required to be approximately 1.45 Volts.

Unfortunately, multiple factors will affect the supply voltage. These factors consist of but are

not limited to varying load impedance and fluctuating voltage present at the mains.

A function of the STATCOM is to control the voltage level of the line. Therefore, as long as the

voltage has the ability to range above and below the desired 12 VAC set point it can be used to

demonstrate how the STATCOM controls the line voltage.

By arbitrary design, the line resistance will need to be a small value. Initially, the project

utilised an on hand resistor (1.5Ω ±5% 50W) to act as the line resistance. Through

rudimentarily determining the line reactance based on the lines X/R ratio equalling one the

line would have an inductance of 4.77 mH.

Initial designs assumed the line inductance to be a toroidal inductor. Inopportunely, as

mentioned above in Section 3.2.1 the core originally chosen was not suitable for 50 Hz

operation. After sourcing a local electronics store the resulting find was an Air Core 3.0mH

inductor [41] shown in Figure 3-3. This inductor will eliminate the requirement of the 1.5Ω

50W resistor as the DC copper resistance of the inductor is 1.36Ω. Additionally being air cored

42

the inductor is highly unlikely to saturate. From specifications the power handling capability is

100 W RMS. Re-calculating from these new values, the X/R ratio is now 0.693. This X/R ratio is

an acceptable value for the line and takes a similar X/R ratio to that of underground power

conductors.

Figure 3-3 3.0mH Air Core Inductor

Using the inductor’s nominal values and the impedance values found in Section 3.2.1 the

lowest overall impedance for the system can be calculated. Knowing the lowest impedance

load will be 7.99∠48.24°Ω and the line impedance is 1.65∠34.72 °Ω the lowest overall

impedance will result in 9.60∠45.94°Ω.

From Equation 3-1 the current through the system with a line voltage of 13.45 VAC gives 1.4∠-

45.94° Amps. From this the largest voltage drop across the line impedance is 2.31∠-11.22°

volts. This level of voltage drop is acceptable for the small-scale system.

3.3. Liquid Crystal Display Interface

There are several styles and sizes of Liquid Crystal Displays (LCDs) for interfacing with an

Arduino platform. Generally, the Hitachi HD44780 driver [42] commonly identified by the 16-

pin interface is the basis for interfacing with the Arduino platform. Figure 3-4 gives an

43

example of this 16-pin interface with the connectors shown along the top of the green LCD

board.

Figure 3-4. 16 pin interface example

3.3.1. 1602 LCD Shield Parallel (4 Bit) Interface

The initial approach for the project was to utilise a display system students would be familiar

with from ENG109. This familiarity is the 1602 LCD screen with tactile switches. Figure 3-5

displays the 1602 LCD shield with 16 Characters per line with two lines available. The display

strategy for this LCD was to incorporate the functionality of the tactile switches. Assigning

each switch a set parameter such as voltage, current, power factor or frequency. The student

could then toggle through the information to be displayed.

Figure 3-5 1602 LCD Arduino Shield with Tactile Switches [43]

44

Integrating the 1602 LCD with the Arduino is simple with the LCD mounted onto an Arduino

Uno compatible shield. The LCD shield utilises a 4-bit parallel interface with information

transferred through four digital pins. In an attempt to reduce the number of pins consumed

the tactile switches output a set voltage level between 0-5 VDC. Through this approach the five

switches coalesce to an analog input. This is evident within the example code [43] supplied.

void loop()

int x;

x = analogRead (0);

lcd.setCursor(10,1);

if (x < 60)

lcd.print ("Right ");

else if (x < 200)

lcd.print ("Up ");

else if (x < 400)

lcd.print ("Down ");

else if (x < 600)

lcd.print ("Left ");

else if (x < 800)

lcd.print ("Select");

By reading the 10-bit analog input the voltage observed would correspond to a particular

switch. To allow for variations in the voltage the divisions are approximately 200 points. This

division level represents a voltage difference of approximately one volt. Table 3-7 details the

pin connections for the 1602 LCD shield.

Table 3-7. 1602 LCD pin connections [43]

Pin Function

Analog 0 Buttons (select, up, right, down and left)

Digital 4 DB4

Digital 5 DB5

Digital 6 DB6

Digital 7 DB7

Digital 8 RS (Data or Signal Display Selection)

Digital 9 Enable

45

The advantage of an LCD shield compared with a stand-alone LCD was an attractive solution

and initial experiments with the 1602 keypad LCD were successful. However, the end

determination was that it would not effectively allow a student to visualise the changes to the

system rapidly enough. In addition to this, concerns became prominent when introducing the

MOSFET H-bridge shield into the project. Unfortunately, the component height of the 1000µF

blue capacitor shown in Figure 3-35 meant that the shields could not be directly stacked on

top of one another. Furthermore, a violation between pins occurred such that both the LCD

and H-Bridge require the use of analog input A0. A solution was to employ two Arduino

Uno’s transferring the data by serial, I2C communications (Figure 3-6) or doubling up on

cabling. After deliberation, the final solution came from the purchase of a 2004 I2C interfaced

LCD discussed in the section below.

3.3.2. 2004 LCD I2C Interface

Interfacing with the 2004 I2C LCD (shown below in Figure 3-7) consumes fewer pins than the

standard parallel 4-bit interface discussed above in Section 3.3.1.

Developed by NXP Semiconductors, formerly known as Phillips Semiconductors, the I2C

communications protocol uses only “two wires, serial data (SDA) and serial clock (SCL), to

carry information between the devices connected to the bus. Each device is recognized by a

unique address (whether it is a microcontroller, LCD driver, memory or keyboard interface)

and can operate as either a transmitter or receiver, depending on the function of the device.

An LCD driver may be only a receiver, whereas a memory can both receive and transmit data.”

[44]

46

Figure 3-6 Example of an I2C-Bus Configuration [44]

Figure 3-7. 2004 LCD display [45]

In order to communicate to the LCD via the I2C bus a new library was required. This library

is available online at Wikispaces Blue I2C LCD [46]. Also available at this source is a setup

guide describing how to install the updated library. The website details a few examples

dependent upon the particular style of the I2C bus driver installed on the LCD. For this

particular project, the driver installed is similar to the LCM1602 IIC A0 A1 A2 [46] with the

actual driver board shown in Figure 3-8.

Figure 3-8. I2C driver [46]

47

A series of set up commands are required within the code in order to obtain successful

communication with the LCD screen. The code snippet shown below details these

initialisation codes.

#include <Wire.h>

#include <LiquidCrystal_I2C.h>

LiquidCrystal_I2C lcd(0x27, 2, 1, 0, 4, 5, 6, 7, 3, POSITIVE);

// Set the LCD I2C address

lcd.begin(20,4);

// initialize the lcd for 20 chars 4 lines and turn on backlight

With successful communication established, the code snippet shown below details how the

project communicates the required information to the display. Using the simple set of

commands ‘setCursor’ and ‘print’ the display can communicate the system information to

the user. As shown below within the Power Factor portion of the code the level of precision

displayed is alterable by including the number of decimal places to present. Opposing the

1602 method of display there are no switches to change between displayed values. The

objective of the 2004 LCD is to display all information on the screen at once as shown in

Figure 3-9.

void LCDdisplay() // LCD screen display

// RMS Voltage

lcd.setCursor(1,0);

lcd.print("V: ");

lcd.setCursor(4,0);

lcd.print(Vrms);

// Power Factor

lcd.setCursor(0,3);

lcd.print("PF: ");

lcd.setCursor(4,3);

lcd.print(power_factor,4);

// end of LCD display function

48

Figure 3-9. 2004 LCD display

3.4. Measurement

This sections of the report will detail what procedure was taken to measure the system

values with considerations given to the voltage, current and zero crossing of the voltage

waveform.

3.4.1. Zero Crossover Detection

Zero crossover is the point at which a waveform passes from positive to negative voltage.

This point is where ideally no voltage is present. Zero crossing detection is an important

system for measurement techniques. Whilst it appears to be a simple task at higher

frequencies, zero crossing detection can become quite complex particularly if high accuracy

is required. Zero crossover detection is used in a variety of applications. Common

applications are for synchronisation of AC waveforms, determining phase angle between two

signals and in light dimming. The purpose behind using zero cross over detection to switch

voltages is that random switching along a waveform can introduce large amounts of

harmonic interference. This interference can become quite dangerous if not attenuated

before entering a system.

After researching techniques for zero crossing the most useful source came from Elliot

Sounds Productions at [47]. Techniques such as using a modified ramp generator shown in

49

Figure 3-10 and using high speed comparators coupled with Exclusive-Or gates shown in

Figure 3-11 were considered.

Figure 3-10. Modified Ramp Generator [47]

The major downfall of the basic 50Hz zero crossover detector (Figure 3-10) is that the pulse

width of each trigger is typically around 600µS. Assuming that the width is even this would

result in 300µS on either side of the actual zero cross over. This is not accurate enough for

the application.

Figure 3-11. High Speed Comparator and Exclusive-Or Gate Zero Cross Over [47]

Whilst the high speed comparator option overcame the pulse width issue significantly

other issues arose when enquiring into the available chipsets. One of these issues was the

need to have a split rail voltage source, which would require additional equipment to run

this one component. Continuing research on Elliot Sounds blog located the final solution.

The use of a Quad RS-422, RS-423 Line Receiver (Figure 3-13) with a propagation delay of

50

19nS and a required supply voltage of only +5V became an elegant solution to both

problems.

For the zero cross over detection the design was to use the DS3486 Quad RS-422, RS-423

Line Receiver. However, the package’s small surface mount design made it difficult and

cumbersome to work with. Through research on the datasheet the product is also

produced in a 16 pin DIL package, the MC3486. Figure 3-12 shows the comparison of both

the DS3486 and the MC3486 side by side to demonstrate the differences. This component

solves the issue of size whilst still offering the ±25V max input voltage and retaining the

same pin layout.

Figure 3-12. DS3486 Vs. MC3486 Design

Figure 3-13. DS3486 Zero Crossing Detector [47]

SMD DS3486

DIL MC3486

51

Initial designs were to fully utilise the circuit shown in Figure 3-13 taken from [47]. In this

circuit a second IC a HC7402 Not-OR (NOR) gate outputs a small pulse displaying the zero

cross over point on a waveform. Initially it was thought that it would be possible to simply

time the difference between the edges of these pulses for both the voltage waveform and

current waveform. However, after building and testing the circuits the current waveform did

not represent a clean sinusoidal wave when an inductive load was introduced. The resulting

output waveform pulses from the MC3486 were odd shaped and misaligned.

After deliberation the solution was to ignore the current waveform and the HC7402 chip for

the zero cross over detection. Instead the project would rely on a series of equations within

the code to compute the power factor for the system. The purpose of the MC3486 is to now

produce a square wave pulse when a negative edge is detected on the AC mains sinusoid.

This square wave pulse is responsible for timing and synchronisation between the STATCOM

output waveform and the AC mains.

3.4.2. Current Measurement

In order for the user to observe the current waveform for measurement, calculations and

display purposes a Current Transformer (CT) is required. CTs are designed to reduce a high

current down to a useable level for accurate instruments. By isolating the current flowing in

the AC transmission line the current can be safely monitored. The principal of operation of a

current transformer is no different from that of an ordinary transformer. [48] The main

difference between Current Transformers and Voltage Transformers (VTs) are the inverse

ratios between primary and secondary windings. CTs consist of only one or very few turns as

the primary. [48] The secondary windings conversely, consist of a high number of turns

encased around a low-loss magnetic material. Figure 3-14 shows a basic example of this

principle, alongside is the circuit symbol usually used to represent a CT in a drawing.

52

Figure 3-14. Basic Example of a Current Transformer (left) [49], Circuit symbol (right) [48]

Commonly a CT will be identified by the current that the primary side of the CT is capable of

handling in conjunction with a turns ratio such as 1000:1. From Figure 3-15 it can be shown

that, by increasing the number of turns on the primary wire, the turns ratio can be reduced

to make the secondary more sensitive to changes in the current. Additionally, this decrease

in turns ratio allows a higher value current transformer to provide the maximum output

current for an ammeter when used on smaller primary current lines. [48]

Figure 3-15. Current Transformer Primary Winding Increasing Example [48]

One loop Two loops Three loops

Primary

Secondary

300A/5A 150A/5A 100A/5A

53

Knowing that the maximum current present on the primary line is approximately 1.5 A, the

design strategy is to acquire a CT capable of handling a current greater than 1.5 A on the

primary side whilst still allowing a safety margin. After Investigation the typical range for

Current Transformers is 1A, 5A, 10A, 15A, 20A and onward. By intention, the transformer has

been selected as the L01-6210 Alttec 5A 1000:1 Through Hole Current Transformer [50]

shown underneath in Figure 3-16. In order to measure the line current effectively the

employment of the principle shown in Figure 3-15 will be incorporated into the design. By

wrapping the primary wire around the transformer twice the effective current seen doubles.

Figure 3-16. Alttec 5A 1000:1 L01-6210 Through Hole Current Transformer [50]

Whilst the only use for the current waveform is to determine the system current and power

factor the burden resistor should be appropriately calculated to ensure correct operation.

Making use of the following formula (Equation 3-18) the burden resistor can be calculated

for measurement with the Arduino.

𝑅𝑏𝑢𝑟𝑑𝑒𝑛 =

𝐴𝑅𝐸𝐹 × 𝐶𝑇 𝑇𝑈𝑅𝑁𝑆

2√2 × 𝐼𝑃,𝑀𝑎𝑥

Equation 3-18 [51]

54

Where:

AREF is the analog reference voltage of the controller

- 5 Volts.

CTTURNS is the turns ratio of the transformer

- 1000:2 → 500:1

IP,Max is the maximum primary current

- Max Current 2 Amps (fuse rating)

Substituting this information into Equation 3-18 the Burden resistor results in:

𝑅𝑏𝑢𝑟𝑑𝑒𝑛 = 5 × 500

2√2 × 2

𝑅𝑏𝑢𝑟𝑑𝑒𝑛 = 2500

4√2

𝑅𝑏𝑢𝑟𝑑𝑒𝑛 = 441.94 Ω

441Ω is not a common value. The options available are 430Ω and 470Ω. Using components of

a ±1% tolerance the selection for the burden resistor will be 470 Ω. Rearranging Equation

3-18 to solve for IP,Max yields 1.88 A, whilst this maximum current is lower than the previously

prescribed maximum, this will have no effect on the overall operation of the CT.

At this point the current transformer cannot be connected to the Arduino analog input

though. The output of the CT will vary from a positive to negative with respect to ground.

However, the Arduino analog inputs require a positive voltage [51]. By applying a voltage to

one lead of the CT secondary, the output with respect to ground will now swing between this

voltage point. The Arduino analog inputs have the ability to measure between 0-5 V on a 10-

bit scale. By applying the mid-point voltage of 2.5 V to one lead of the CT secondary, the

signal would vary around this point. Provided the output voltage was never greater than 5V

55

peak-peak, the input voltage would always remain positive. Figure 3-17 shown below details

the circuit required to achieve this 2.5 V bias.

Figure 3-17. Current Sensor Interfacing with an Arduino Analog Input [51]

For this project, however a 100kΩ trim pot has been used in place of the R1 and R2 voltage

divider. Figure 3-18 shows the project circuit. This allows the voltage bias to be adjusted to

achieve the desired 2.5 V more precisely. While a smaller value trim pot in series with a

resistor on each end would be more sensitive to adjustments, the 100kΩ was an on-hand

component. Capacitor C1 provides a path for the alternating current to bypass the resistor

[51]. The value assigned to this capacitor is a 2200µF electrolytic. A small RC LPF filter with a

cut off frequency of 339 Hz has also been included into the circuit to attenuate any noise

input from the CT.

Figure 3-18. Project Circuit - Current Measurement

2200µF 10V

Electrolytic

0.047µF

56

3.4.3. Voltage Measurement

The voltage measurement for this project will serve multiple uses. Theses uses will be

measurement, calculation, voltage level comparison and display. In order for these tasks to

be undertaken the voltage must first be reduced down from the 12 VAC line. The biggest

requirement for this reduction is that of the analog input into the Arduino Uno, as previously

mentioned the Arduino is only capable of accepting a range of 0-5 V.

Several techniques were considered when initially designing this project in order to measure

the AC voltage. At the initial stages of this project the idea was to use a method of AC – DC

rectification and voltage division. By applying the AC voltage to a bridge rectifier with parallel

smoothing capacitors of a sufficiently large capacitance the output would resemble a DC

voltage with minimal ripple. Figure 3-19 beneath shows an example of this full wave

rectification included is the output waveform detailing the ripple voltage in the DC output.

Attaching a voltage divider across the output terminals the voltage could be reduced to a

level the Arduino could manage.

Figure 3-19. Bridge Rectification Circuit with Output Waveform [52]

After progressing the project further, it was discovered that this rectification voltage

measurement method was no longer applicable as the zero crossing measurement for

calculating power factor had become more complex. The solution came from research into an

Arduino controlled home energy monitor [53]. This monitor utilised an AC plug pack to isolate

57

and reduce the voltage of a mains supply. Following this transformer, a voltage divider is

inserted across the output terminals to further attenuate the voltage level. In order for the

voltage to be measure with the Arduino a second voltage divider applying a DC bias of 2.5V to

the secondary “neutral” side of the transformer is required. Figure 3-20 shows the circuit

layout for this method as well Figure 3-21 details their corresponding voltage waveforms

primary (green), secondary (red) and the measurement waveform (blue).

Figure 3-20. Voltage Measurement with Transformer [54]

Figure 3-21. Voltage Measurement Waveforms [54]

2.5V

58

Initially, this method was an effective solution allowing the waveform to be measured and

the voltage levels calculated within the Arduino. Issues began to arise when the STATCOM

was connected to the line. Unknowingly a ground violation had been created. This ground

violation was caused by the improper connection of the 2.5 VDC bias and the neutral line.

Unfortunately, being unknowledgeable on ground violations the transformer shown above in

Error! Reference source not found. was disregarded as the power supply transformer had

already isolated the mains. This was not the case as the STATCOM was feeding onto the

secondary side of this 240 VAC mains isolating transformer. The solution emanated from

simply including another low power isolation transformer to isolate the two circuits (Figure

3-22).

Figure 3-22. 3VA 12+12 VAC Transformer [55]

Furthermore, after implementing this solution the biasing voltage divider was changed to a

variable resistance 100kΩ potentiometer. This allowed the bias to be more accurately

adjusted. For finer tuned adjustments the potentiometer could be incorporated with fixed

resistances. The decision behind changing the voltage divider came from observing the

triangular waveform in comparison to the sinusoidal signal. The observation was that the

midpoint of the sinusoid was not 2.5 V instead it was around 2.23 V. This variation began

causing issues with the PWM switching as the lower peaks of the sinusoid and triangular

59

waveform were almost touching. This is observable in Figure 3-23 where the highlighted area

shows the two waveforms almost aligning at the low peaks.

Figure 3-23. Sinusoid Midpoint Observations

After implementing the changes, the difference was immediately observable. Figure 3-24

below shows how this midpoint increase better aligns the two waveforms.

Figure 3-24. Sinusoid Midpoint Observation (After Potentiometer Added)

With the changes to the sinusoidal waveform the code needed a small adaptation for the scaling

factor. The voltage waveform shown above is taken into the analog input and converted into a

number between 0 – 1023 for a 10-bit system. This number is then manipulated back into a

voltage value. This voltage value in conjunction with the value obtained from the current are

60

combined through various formula to calculate the powers and power factor. The code snippets

below details a small portion of how this operation is performed. For the full code Appendix A.

Software should be consulted.

// Voltage Variables

double sum_of_voltage,

square_voltage,

Vrms,

instant_voltage,

voltage_scale,

voltage_offset;

int raw_voltage_data,

sum_of_voltage = 0;

square_voltage = 0;

voltage_offset = 2.5;

voltage_scale = 19.8;

// Raw input data

raw_voltage_data = analogRead(A3); // raw analog voltage data sample

// Voltage, current and power conversion calculations

instant_voltage = ((raw_voltage_data * (5.0/1023)) - voltage_offset)

* voltage_scale;

// Convert the analog reading to an AC voltage

instant_power = instant_voltage * instant_current;

// Calculation for instant real power

// RMS Voltage Calculations

square_voltage = double(instant_voltage * instant_voltage);

sum_of_voltage = sum_of_voltage + square_voltage;

3.5. PWM Switching Scheme

As previously discussed in Chapter 2.2 the subsequent chapters associated the PWM

switching scheme for this project will encompass a form of bipolar switching. Generally, in

bipolar switching the sinusoidal control signal and carrier waveform traverse across a positive

and negative axis, such that they have positive and negative peaks.

Additionally, mentioned in Chapter 2.5 the triangular waveform used as the carrier signal for

the bipolar PWM switching will be generated through a DDS method from an Arduino Uno.

One of the issues associated with this method is that the output voltage of the Arduino is only

61

capable of positive voltages (0 – 5 Volts), for this reason the sinusoidal control signal will need

to be within this range also.

3.5.1. Triangular Carrier and Sinusoidal Control Waveforms

In the design of this project the triangular carrier waveform will need to have a sufficiently

high frequency such that the mf ratio for PWM switching is high. One of the major drawbacks

of the Arduino DDS method is that the frequency cannot be higher than around 10kHz. In

addition to this, increasing the frequency of the carrier results in larger and more significant

losses through the filter.

After testing several reconstruction filters for the output of the triangular carrier it was

determined that the most effective was a slightly modified version supplied by Martin

Nawrath at [30]. Figure 3-25 shows the original Martin Nawrath Filter.

Figure 3-25. Martin Nawrath 12.5kHz Chebyshef Low Pass Filter [30]

Originally, the filter was tested as a simple low pass 1kHz RC filter, but whilst it sufficiently

filtered the waveform into a clean triangle the losses were too large for to be used. Following

this a second order 2.3 kHz LC filter was implemented. This filter greatly reduced the losses

present. The down fall of this LC filter was that the output had become skewed slightly for

some reason. This unknown skew was also present within the reference source: Michael

Chapman’s Thesis [24] where he detailed the use of an LC filter on the DDS output. The

62

resultant filter was chosen to be a modified version of the 12.5kHz Chebyshef low pass filter

used in Martin Nawrath’s project [30]. The transfer function in the s domain is shown below

in Equation 3-19.

𝐻(𝑠) =1

(1.32 × 10−24)𝑠5 + (1.04 × 10−19)𝑠4 + (8.77 × 10−15)𝑠3 + (6.91 × 10−10)𝑠2 + (1.27 × 10−5)𝑠 + 1 Equation 3-19

Figure 3-26. Modifired Chebyshef Low Pass Filter

Figure 3-26 details the modifications made to the filter. These modifications consist of

removing the end two parallel resistances: the fixed 270Ω and 100kΩ potentiometer. The

removal of these components was decided upon when testing the original filter. The output

waveform suffered significant losses and after removing these two resistors the filtered

waveform became drastically less affected. Unfortunately, there was no investigation into

changing the value of the initial series 270Ω resistor, however it is expected that, by reducing

this value lower attenuation will result.

Moreover, as previously mentioned in Chapter 2.2 the reduction in magnitude can also be

attributed to the increase in output frequency. This was observed when increasing the carrier

frequency from 1.5kHz to 5 kHz shown below in Figure 3-27.

63

Figure 3-27. Triangular Waveform at various frequencies, 1500Hz (left), 5000Hz (Right)

At 1.5kHz the peak to peak voltage was 4.08 V whilst at 5kHz the peak to peak voltage was

reduced to below 3.12 V. This reduction was unacceptable for the project and the end

decision became to use a 1.5kHz carrier waveform. This lower carrier waveform frequency

still results in a frequency modulation ratio of 30 which is more than enough to achieve the

desired results for the project.

Additionally, in order to ensure the output voltage remained at a constant midpoint of 2.5

Volts for the PWM comparator a decoupling capacitor and voltage divider was installed at the

end of the filter. Figure 3-28 details the final filter circuit used for the project. DWG 100-001

located in Full Circuit Drawings should be consulted to see the overall connections of the filter

within the entire circuit.

64

Figure 3-28. Modified 12.5kHz Chebyshef Low Pass Filter

Given that the triangular waveform will have a voltage range from 0-5 V the sinusoidal control

signal will need to follow suit. A solution to this problem is to utilise the voltage measurement

sinusoid discussed above in Chapter 3.4.3. Knowing that the waveform is already within the

desired range for its use on the analog inputs resolves this issue. One downfall of this is that

the sinusoid magnitude will vary slightly with changes in the loads but this is not significant

enough to warrant creating a separate solution.

Typically, the control sinusoid is the variable magnitude waveform within a PWM switching

scheme. Conversely, for this project it has been inverted such that the triangular carrier will

act as the variable magnitude waveform. This is done to allow the system to control the

magnitude of the carrier through a control algorithm which affects the output magnitude.

Shown below is a snippet of the code that affects the magnitude of the output waveform.

byte valA;

const byte valB = 100;

// defines intial values for system operation

valA = 100;

// triangle waveform magnitude variables (100 = 100% (max))

OCR2A=(valA*(pgm_read_byte_near(tri256 + (uint8_t)(icnt)))/valB);

// read value from ROM sine table and send to PWM DAC out on pin 11

By attenuating the values in the lookup table by an integer value up to 100 the overall output

waveform can be controlled and reduced in the sense of controlling the output as a

65

percentage between 0-100%. Currently no control algorithm has been produced to

manipulate this variable and control the STATCOM output.

3.5.2. PWM Switching Method

As mentioned above the PWM switching scheme for this project will consist of a modified

bipolar switching scheme. The PWM pulses are produced through the comparison of the

sinusoid and triangular waveform as detailed in Chapter 2.2. In order to produce these pulses

an LM319 high speed comparator (Figure 3-29) has been implemented into the system. This

comparator chip can operate from a single +5 Volt supply making it ideal for the current

application.

Figure 3-29. LM319 DIL Package Pin Layout [56]

From the Typical Applications section of the LM319 Datasheet [56], in particular the window

detector application, it can be seen that in order to produce the required TTL output style

pulses a pull up resistor connected across the 5 Volt supply and the output pin is required.

Additionally, in order for this chip to operate on a single 5 Volt supply the V- pin (6 on the DIL

package) requires a ground connection along with GND 1 and 2 Pins (3 and 8 respectively).

One issue discovered with the datasheet is that the pin connections for the typical

66

applications refers to the Metal Can Package of the LM319. A simple cross reference solves

this issue. Relevant Datasheet Information subsection LM319 Datasheet Information will

detail the Metal Can Package layout in Figure C-1. In addition, Figure C-2 will show the circuit

diagrams for the typical applications.

By applying the sinusoidal control signal to pin 4 (input 1+) and the triangular carrier wave to

pin 5 (input 1-) shown in Figure 3-30, the resulting output on pin 12 (output 1) will provide

the PWM switching signal shown in Figure 3-31.

Figure 3-30. Sinusoidal Control Signal Overtop of Triangular Carrier Waveform

Figure 3-31. Oscilloscope Snapshot of Sinusoidal Signal (top) and PWM Comparator Output (bottom)

67

In order to control the H-Bridge, the PWM pulses and the inverse of these pulses are sent to

the control pins of the H-Bridge. To produce the inverse of these pulses a simple NOT gate is

applied. An example of these inverse switching pulses can be seen in Figure 3-32. Although

practically there is an inconsequential delay with the NOT gate and the signals have the

potential to be high simultaneously. It is not significant enough to cause any issues with the

H-Bridge. The reason for this not effecting the H-Bridge will be discussed below in Chapter

3.6.1.

Figure 3-32. Inversed PWM Switching Pulses

Varying the magnitude of the carrier wave the two waveforms peaks can be brought closer

together or further away. The effect of this variation in magnitude is that the STATCOM

output magnitude can be inversely controlled. As previously discussed, if the amplitude

modulation ratio is 1.0, the STATCOM output magnitude will be exactly that of the DC

source voltage. A control algorithm can be designed to incorporate this information in

order to obtain the required output voltage magnitude of the STATCOM. Figure 3-33 shows

an example of the variation in the triangular carrier waveform in relation to the sinusoidal

control signal.

68

Figure 3-33. Varying Carrier Magnitude Example

3.6. STATCOM

This section of the report will detail the design procedure behind the construction of the

STATCOM components. Several allowances have been made for the sizing of the

components to ensure adequate levels of voltage and current for operation of the STATCOM.

3.6.1. Full Bridge VSI

As previously discussed the Voltage Source Inverter is one of the main components of the

STATCOM. The purpose of this device within the system is to provide the control device for

which the DC capacitor will supply or absorb reactive power.

The original design of the H-Bridge that was developed utilising discrete components

including two P-Channel and two N-Channel MOSFETS was unsuccessful. Given the low

number of components and vast resources available it was thought that constructing an H-

Bridge from these components would be the simplest option. Unfortunately, upon

completion of the build the H-Bridge was non-functional. Initial thoughts were that a faulty

component was the reason. After replacing the component and re-testing the circuit similar

results were obtained in which only half of the bridge was operating. After a few tests an

69

accidental short circuit damaged the PCB and remaining components. The circuit that was

followed for this construction is shown below in Figure 3-34.

Figure 3-34. Initial H-Bridge Design Circuit [57]

Following this, the option was to redesign the H-Bridge circuit or investigate an off-the-shelf

option. The end decision was to source an off-the-shelf option this would enable

replacement of the part to be a lot easier should anything be damaged. After investigation

into what products were commercially available an Arduino shield produced by Infineon

shown below in Figure 3-35 was found. The core component of this shield is two half bridge

ICs which are the Novalith IC™ BTN8982TA.

70

Figure 3-35. Infineion BTN8982TA Motor Driver Shield [58]

The BTN8982TA is an integrated high current half-bridge [59] typically used for motor drive

applications. It is part of the NovalithIC™ family containing one P-Channel high-side MOSFET

and one N-Channel low-side MOSFET with an integrated driver IC in one package. Interfacing

to a microcontroller is made easy by the integrated driver IC which features logic level inputs

and protection against over temperature, under voltage, overcurrent and short circuit. [60]

As mentioned above these IC chips are capable of being driven from logic level input voltages.

This is a major advantage enabling direct operation from the Arduino microcontroller. In

addition to this the BTN8982TA Chip encompasses very low path resistances:

Path resistance of max. 20.4 mΩ @ 150 °C (typ. 10.0 mΩ @ 25 °C)

High Side: max. 10.5 mΩ @ 150 °C (typ. 5.3 mΩ @ 25 °C)

Low Side: max. 9.9 mΩ @ 150 °C (typ. 4.7 mΩ @ 25 °C) [60]

This is a significant advantage as the lower the path resistance, the lower the losses within

the VSI. Furthermore, this shield is capable of handling high PWM frequencies up to around

71

30kHz. This is well within the range required for the PWM switching scheme discussed above

in Chapter 3.5.

A typical application of this MOSFET dual half bridge shield is shown in Figure 3-36. This

layout has been applied to the STATCOM in which the motor is replaced with an LCL filter to

be discussed in section 3.6.2. From this circuit diagram it can be seen that each bridge has

two inputs labelled IN and INH. The IN labelled input is responsible for the PWM input control

signals whilst the INH inputs are designed to put the bridge into a sleep mode if no high signal

is applied.

Figure 3-36. Application circuit for a bi-directional motor control with BTN8982TA [58]

Additionally, as previously mentioned, the PWM signals both being high simultaneously

would normally cause a short circuit. However, with this design comprising itself of two half

bridges there is not opportunity for a short circuit to occur. The reasoning for this is that

when a low signal is sent to one IN input pin the low-side MOSFET is active, conversely a high

signal will activate the high-side. If both bridges receive a high signal, then both bridges will

have the high-side MOSFET active and no current will flow.

72

3.6.2. STATCOM Filter The voltage output of the VSI STATCOM will need to adequately resemble a 50Hz sinusoidal

waveform. This can be achieved through the PWM switching in conjunction with a low pass

filter designed to block the switching frequency of the PWM. The PWM signal will consist of

the desired low frequency 50Hz waveform along with the high frequency switching noise. By

utilising an LCL low pass filter shown in Figure 3-37 the signal can be filtered without large

resistive losses in comparison to an RC low pass filter.

Initially for this project it was thought a simple LC filter would suffice, however after

observing a waveform on the oscilloscope it could be seen that the PWM switching noise was

present on the negative half of the waveform. Adding in the additional inductance resulted in

the waveform being filtered in a far more adequate way.

Figure 3-37. LCL Filter Layout [61]

Utilising readily available components a filter design was established based on a series of

equations and rules of thumb for determining LCL filters. These equations are as follows:

𝜔𝑟𝑒𝑠 = √𝐿1+𝐿2

𝐿1𝐿2𝐶1 Equation 3-20 [62]

10𝑓𝑔 < 𝑓𝑟𝑒𝑠 < 0.5𝑓𝑠𝑤 Equation 3-21 [62]

Using Equation 3-20 and the following values the filter was designed to satisfy Equation 3-21.

𝑓𝑔 = 50𝐻𝑧

73

𝑓𝑠𝑤 = 1500𝐻𝑧

𝐿1 = 𝐿2 = 470𝜇𝐻

𝐶1 = 220𝜇𝐹

𝜔𝑟𝑒𝑠 = √470𝜇 + 470𝜇

470𝜇 × 470𝜇 × 220𝜇

𝜔𝑟𝑒𝑠 = 4397.99 𝑟𝑎𝑑/𝑠

𝑓𝑟𝑒𝑠 = 699.96 𝐻𝑧

3.6.3. DC Capacitor

One of the main components of a STATCOM is the DC capacitor. This DC capacitor is

responsible for the absorbing and supplying the reactive power. For this project, the method

for designing the DC capacitor corresponds to methods used in industry. According to the

following equations:

𝐶𝐷𝐶 =

√2𝐼(1 − 𝑠𝑖𝑛𝛼)

2𝜔𝜀𝑉𝐷𝐶 Equation 10. [63]

𝑉𝐷𝐶 =

√3𝑉𝑠

𝜋 Equation 10.1. [64]

𝑉𝑠 = √2𝑉𝑝𝑒𝑎𝑘 Equation 10.2. [63]

𝜔 = 2𝜋𝑓 Equation 11. [35]

Where:

α is the value of the switching angle for the H-bridge.

- For the largest capacitance α=0⁰

ε is the voltage ripple factor of dc voltage, whose value may vary from 5 - 20% for a

practical application.

- Assumed to be 10%.

- The smaller the ε value, the larger the capacitance and thus the smaller the

voltage ripple.

74

I is the rated current of the H-Bridge.

- Whilst the bridge has a current handling capability of 35 Amps, the rated

current of the system is 2 Amps.

VS is the peak phase voltage of the 12VAC supply voltage.

VIN is the supply voltage.

- VIN has a nominal voltage of 12Vac.

Substituting in these assumed parameters yields the following values:

𝑉𝐷𝐶 =√3 × √2 × 18

𝜋

𝑉𝐷𝐶 = 14.03𝑉

𝐶𝐷𝐶 =√2 × 2(1 − sin(0))

2 × 2 × 50 × 𝜋 × 0.1 × 14.03

𝐶𝐷𝐶 =2√2

280.6 × 𝜋

𝐶𝐷𝐶 = 0.003209 𝐹

𝐶𝐷𝐶 = 3209 𝜇𝐹

From these calculations, the required capacitance is 3209µF. This is unfortunately not a

standard value obtainable; the suggestion is to choose the next available standard value for

the capacitor.

After some research through a JayCar catalogue, the decision was to incorporate a 4000µF

high capacitance, high voltage capacitor. Although the value for the capacitance has

increased, there will be no negative effect on the STATCOM. Conversely, the increase will

allow for a greater reactive power capability. Incorporated into the selection decision for

this capacitor were the ripple current and the physical mounting structure. Having a ripple

current twice that of the system requirement should eliminate the chance of damage from

an overcurrent in the capacitor. Additionally, taking into consideration the physical

75

mounting of the component allowed for ease of installation with minimal tool work

required.

Figure 3-38 shows a representation of the capacitor selected. See Table 3-8 below for the

specifications for the elected capacitor.

Figure 3-38 RG Chassis Mount Electrolytic Capacitor. [65]

Table 3-8. DC Capacitor Specifications [65]

Property Value

Capacitance: 4000 µF

Voltage: 75V

Ripple Current: 4.6A

Physical Size: 51mm X 35mm

Rated Temperature: 85®C

76

4. Results

This section of the report will detail the obtained results from testing the individual circuits

detailed in the design phase of this project report.

4.1. Measurement and Control Waveforms

The measurement and control waveforms will be discussed within this section; in particular

how the scaling was achieved. In addition, the variation in triangular waveform will be

discussed.

4.1.1. Voltage Scaling

In order for the Arduino Uno to display the line voltage on the LCD a scaling factor is required

to transform the value seen at the ADC into a voltage representative of the line voltage. To

determine a value that would sufficiently scale all measurements a repeatable test was

conducted. The test consisted of measuring the RMS line voltage using a UNI-T UT803 Digital

Multimeter, and measuring the Peak-to-Peak value of the attenuated and biased voltage from

the measurement transformer discussed in chapter 3.4.3 on a TDS1001B Tektronix digital

oscilloscope. By measuring the Peak-to-Peak value of the attenuated waveform shown below

in Figure 4-1 an RMS value can be calculated from Equation 4-1.

𝑉𝑅𝑀𝑆 =

𝑉𝑃−𝑃

2√2 Equation 4-1. [66]

77

Figure 4-1. Attenuated and Biased Measurement Waveform

By repeating this test for each possible load a series of data points were obtained to

generate the graph (Figure 4-2) shown below. The full table of results can be found in Scaling

Ratio Tables.

Figure 4-2. Spread of Voltage Measurements

0.55

0.6

0.65

0.7

0.75

11.5 12 12.5 13 13.5 14 14.5

Me

asu

red

Val

ue

RMS Line Voltage

Spread of Voltage Measurement

Capacitor

Inductor

78

The graph shows the spread of RMS line voltage versus the Measured RMS voltage at the

Arduino for both inductive and capacitive loads. This graph shows a relatively linear trend

which is a good indication of the reliability of the voltage measurements.

In order to obtain the best fit scaling factor the calculated RMS values were multiplied by a

starting value of 20. From here, the absolute value of the difference between the RMS line

voltage and this scaled line voltage was calculated. Summing all of these variances together

the aim was to utilise the ‘Goal Seek’ capability on Microsoft Excel to calculate the best fit

scaling factor to reduce this value as close to zero as possible.

The results gave a final scaling factor of 19.8 with an average variation in the voltage value of

0.0961 V. This small variation will translate to the voltage displayed on the LCD being

sufficiently accurate for the purpose of this project.

4.1.2. Current Scaling

Similarly to the voltage, a scaling factor is required to display the RMS line current value. By

repeating the same method as described above, the voltage waveform across the 470Ω

burden resistor can be observed and transformed into a value representative of the primary

RMS line current. Converting the Peak-to-Peak voltage waveform into an RMS current

requires Equation 4-1 and Equation 4-3.

Tabulating and plotting the obtained data Figure 4-3 shows the spread of the RMS line

current versus the calculated RMS line current. From this graph it can be seen that the

𝐼𝑃 = 𝐼𝑆

𝑁𝑆

𝑁𝑃 Equation 4-2 [48]

𝐼𝑃 =

𝑉𝑅𝑀𝑆

𝑅𝑏

𝑁𝑆

𝑁𝑃 Equation 4-3

79

overall shape is slightly exponential however it is quite distributed. This indicates that the

accuracy of these measurements is not sufficient to achieve power factor control as the

results would be too skewed. If the STATCOM were required to control the power factor it

would be advised that an alternate CT or approach be adopted for measuring the current in

order to produce more accurate readings.

Figure 4-3. Spread of Current Measurements

Although these readings are considered inaccurate a scaling factor for display purposes is still

required. Applying the same technique of using the ‘Goal Seek’ functionality of Microsoft

Excel discussed previously. The results return a scaling factor of 0.95 with an average variance

of 0.129 A. This is a large variance in terms of the current sizes for this project thus supporting

the inaccurate results notion.

0

0.2

0.4

0.6

0.8

1

1.2

1.4

0 0.2 0.4 0.6 0.8 1 1.2 1.4

Me

asu

re C

urr

en

t

RMS Line Current

Spread of Current Measurement

Inductor

Capacitor

80

4.1.3. Triangular Wave

As previously discussed, the triangular waveform will be utilised as a means of controlling the

STATCOM output voltage. This control comes from the ability to alter a variable within the

code referred to as ValA. By adjusting this value in the range of 0 – 100 the amplitude

modulation value and in-turn the PWM switching scheme can be altered to increase or

decrease the STATCOM output magnitude.

Repeating the process used to determine the Peak-to-Peak voltage of the Scaling Ratio

waveforms the triangular waveform Peak-to-Peak voltages for varying ValA values can be

obtained. Table 4-1 tabulates these results, with Figure 4-4 plotting the values and applying a

linear trend line.

Table 4-1. Peak-to-Peak Voltage Values of Varying ValA Triangular Waves

ValA Pk-Pk

100 3.72

90 3.4

80 3.04

70 2.76

60 2.28

50 1.96

45 1.72

40 1.6

30 1.24

20 0.8

15 0.68

10 0.48

5 0.28

0 0

From Chapter 2.2 the limit of the amplitude modulation before producing a square wave

output is ma = 3.24. By measuring the Peak-to-Peak value of the largest sinusoid control

signal, the lowest possible Peak-to-Peak amplitude of the triangular carrier can be

determined from re-arranging Equation 2-2. For a sinusoid with a Peak-to-Peak of 2.12 Volts,

81

the smallest possible Peak-to-Peak amplitude for the Carrier is 654 millivolts shown below in

Figure 4-4 as the green line.

Figure 4-4. Plot of Triangular Amplitude versus ValA Variations with limits

Utilising the linear equation y = 0.0383x the minimum value of ValA to coincide with ma = 3.24

can be determined. The resultant value is 17.08. Knowing ValA can only be of integer values

within the range of 0 – 100 a minimum and maximum limit can be set for ValA of 18 ≤ ValA ≤

100. The vertical red line above in Figure 4-4 depicts this lower integer limit of ValA. These

limits in conjunction with the above equation y = 0.0383x will set up the basis for the control

algorithm for controlling the output voltage of the STATCOM.

One of the issues facing the triangular waveform is noise, as the value of ValA is reduced the

noise becomes increasingly prevalent. Within Figure 4-5 it can be seen that the noise

dominates sections of the waveform. This noise could become a severe issue when low ValA

triangular waves are compared with the sinusoidal control signal resulting in inaccurate

pulses or pulse widths. Although this waveform is passed through a low pass filter a solution

y = 0.0383xR² = 0.9965

0

0.5

1

1.5

2

2.5

3

3.5

4

0 20 40 60 80 100

Tria

ngu

lar

Wav

e A

mp

litu

de

(P

k-P

k)

ValA Value

Triangular Amplitude Vs. ValA Variable

82

could come from investigating a more effective filter with shielding or isolation through an

opto-isolater that is capable of transffering analog signals.

Figure 4-5. 1.5kHz Triangular Carrier Waveform with ValA of 15

4.2. STATCOM

This section will detail the results of the STATCOM output waveforms including the output

waveform quality and a basic implementation of the STATCOM.

4.2.1. Output Waveform

As mentioned above by varying the ValA variable the output magnitude of the STATCOM can

be altered. To test the output waveforms of the STATCOM a 12 VDC source is been applied to

the VSI with the measurements taken from the output of the low pass filter while the

STATCOM is disconnected from the line.

In order to evaluate the output waveform of the STATCOM a measure of the waveform

quality is required. To achieve this the Total Harmonic Distortion (THD) percentage is

calculated from the Fast Fourier Transforms (FFT) performed on the TDS1001B oscilloscope.

The analysis was performed using the mains frequency in order to obtain a base line FFT. This

83

base line produced a THD+N% of 1.41%. To determine this value Equation 4-4 and Equation

4-5 were used to transform the dB scale into powers and then converted to a THD%.

Figure 4-6 shows the oscilloscope snapshot detailing the baseline harmonic content. From

this it can be seen that the first order 50Hz content dominates the other orders as expected.

Figure 4-6. Baseline Harmonic Content TDS1001B Snapshot.

By using Equation 4-4 and excluding the first order content the graph (Figure 4-7) below

shows the remaining harmonic content present on the line. The even order harmonics are

negligible, however the 7th and 9th order harmonics are significant in comparison to the other

odd harmonics. This level of sub harmonics is possibly introduced from external noise or

other external sources.

𝑃𝑛(𝑊) = 0.001 × 10

(𝑎𝑛10

)

Equation 4-4 [67] where 𝑎𝑛 is the dB value

𝑇𝐻𝐷 + 𝑁(%) = 100 × √𝑃2 + ⋯ + 𝑃𝑛

𝑃1 Equation 4-5 [67]

84

Figure 4-7. Baseline Harmonic Content excluding First Order

By repeating the oscilloscope FFT measurement process for each value of ValA listed below in

Table 4-2 the THD+N% can be calculated. From this table it is shown that as the value of ValA

is decreased the harmonic content increases, particularly for lower values of ValA where the

output waveform begins to resemble a square wave. This square wave output is evident from

Figure 4-8. Additionally, the THD+N% for ValA at 20 is 37.75% which is close to the THD+N%

of an actual square wave. The reduction in Peak-to-Peak voltage at this level could be from

the introduction of the noise within the triangular waveform as previously discussed.

Knowing the amplitude of the sinusoidal control signal (1.96 V) the ma values can be

calculated to show a relationship between the ValA and the size of the output waveform.

Although these ma values will change depending on the size of the sinusoid control signal

these results can be used as an example of approximately how the ValA will affect the

STATCOM output and in-turn the line voltage.

2nd 3rd 4th 5th 6th 7th 8th 9th 10th

Order of Harmonic 0.46 5.02 0.07 2.19 0.02 16.64 0.09 11.51 0.18

0.00

5.00

10.00

15.00

20.00

Po

we

r (µ

W)

Baseline Harmonic Content

85

Table 4-2. STATCOM Output Harmonic Content

ValA Value Voltage (Pk-Pk) THD+N% ma

100 10.6 4.41 0.527

90 11.8 6.32 0.576

80 13.6 5.69 0.644

70 15.4 5.13 0.710

60 18.8 5.97 0.859

50 23.4 9.41 1.000

40 27.0 18.71 1.225

30 27.0 23.56 1.581

20 24.0 37.75 2.45

10 34.8 53.31 4.083

Figure 4-8. STATCOM Output Waveform ValA of 20

The harmonic content is high at certain values of ValA and could in-turn cause the line to be

saturated with these harmonics resulting in heating of the inductors or an increase in current.

To reduce the harmonic content present in the STATCOM output a solution could come from

investigating a way to increase the carrier frequency without reducing the amplitude of the

waveform. Although there are several methods of producing a triangular waveform, a

method allowing the Arduino to control the amplitude would be ideal such as a voltage

controlled amplifier.

86

Other solutions could stem from increasing the order of the output filter to better attenuate

the noise. As well, it is known that the mf value should be an odd integer; however, this is

unlikely to produce much change in the harmonic content as the sinusoidal control signal

frequency is constantly fluctuating, as it is generated from the mains frequency. This

fluctuation will cause the mf value to be neither an integer nor an even or odd set of the

control signal and will introduce an increase in sub-harmonics for the output waveforms.

Despite these harmonics these output waveforms were utilised to develop the prototype of

this project.

STATCOM Output Waveforms will detail each of the output waveforms with their

corresponding FFT analysis.

4.2.2. Basic STATCOM Implementation

At this point, the STATCOM and the line have not been directly connected. After viewing the

waveforms together there is a significant time difference between peaks shown below in

Figure 4-9. This time difference of 2.00 milliseconds correlates to a 36˚ phase difference; this

will introduce real power transfer between the systems.

87

Figure 4-9. STATCOM output with Line Voltage

A solution to resolving this issue came from increasing the capacitance of the blocking

capacitor in the DC biasing circuit. The previously implemented capacitor was too low and

caused issues with the circuit. Previously a 1µF bipolar capacitor it has now been changed to a

2200 µF electrolytic low ESR capacitor. Figure 4-10 shows the waveforms to be relatively well

aligned. Although there may be a slight difference on the µS order this should allow the two

systems to be paralleled together to acquire results and implement a control algorithm.

Figure 4-10. STATCOM Output with Line Voltage Corrected Phase

88

5. Future Works

Unfortunately, after a multitude of unforeseen or unknowledgeable setbacks, the project is

not 100% completed. There are still several tasks to be finished in order for this to be

considered a finalised product that future students can utilise. This chapter will describe the

tasks that will require attention with a priority level assigned to assist a future student

working on this project. Table 5-1 briefly identifies the remaining tasks with the designated

priority code, this code will be a colour based system with:

Red Urgent

Orange Moderate

Green Low

Table 5-1. Future Works

Task Number Task to be Completed Priority Level

1 Fine tune 2.5 VDC bias

2 Investigate alternate microcontroller

3 Investigate alternate VSI

4 Investigate additional features

5 PCB all in one board

6 Implement a DC-DC boost converter

7 Write, code and test control algorithm

The requirements of each future work task are described below in the order in which they are

listed 1 through to 7.

5.1. 2.5 Volt Bias

As previously discussed in Chapter 3.4 the 2.5 VDC bias used to bring the voltage and current

waveforms out of the negative voltage plane could be better designed and allow for a more

fine-tuned approach. This is a simple task to achieve in which a screw driver adjustable trim-

pot is incorporated with a couple of series fixed value resistors. Figure 5-1 details the layout.

89

The output voltage B can now be finely tuned around the 2.5 VDC range when R1 and R2 are

appropriately sized generally with a 5 VDC source R1 = R2 as well R1 and R2 >> 1000Ω.

Figure 5-1. Fine-Tuned Voltage Divider [68]

5.2. Microcontroller Alternate

After completing a vast amount of this project it was realised that, although the Arduino Uno

is capable of completing the desired task, there is potential for improvement in this section of

the project. An example of this could be to investigate the Arduino Nano. This device could

potentially be capable of performing the same tasks as the Uno whilst consuming less

physical space. Other alternate options should be considered as well. Arduino was chosen as

it was a familiar base for students although this could be changed to suit the design more.

5.3. VSI Alternate

Whilst the Infineon BTN8982TA Shield is capable and well suited to the task of acting as the

VSI there are alternate options available for the project that will once again reduce the overall

physical footprint of the design. These devices could be researched and put forth as part of a

project plan when completing a future thesis on this work.

90

5.4. Additional Features

Investigating additional features would be an effective means of allowing students more

interaction with the equipment, such as introducing more adjustable controls. These features

could range from adjusting the voltage set point between some levels with the use of a

potentiometer or via digital means. In addition, it may be worthwhile introducing the ability

to change line parameters, allowing students to vary the voltage drop across differing line

types. These could be designed to be representative of real-world line types or varied X/R

ratios.

One feature that needs to be included is the ability to control the fans via a temperature

sensor and relay. The load box can become hot when the lower resistance loads are left

active for a period of time, the fans will significantly reduce the heat, however if not activated

the box will continue to heat until damage is caused.

5.5. Manufactured PCB

Currently the circuits are soldered onto small cut sections of prototyping board. This should

only be a temporary solution and should be rectified with the production of a manufactured

PCB. There are several online sources which will take a circuit diagram and convert this to a

PCB layout. An example of this is Eagle 7.3.0 downloadable at [69]. Although the Light Edition

program of this is limited to a small board size it could be possible to produce two or three

separate boards for the task. An advantage of Eagle is that it can be linked with Element14 an

online parts supplier which will produce the board, although at some cost. There are cheaper

alternatives such as sourcing the board production from China but these tend to have a

requirement of producing a large number of identical boards. This could be advantageous if

producing a final product in which several of these devices are going to be constructed.

91

5.6. DC-DC Boost Converter

The purpose of this DC – DC boost converter is to provide a higher DC source across the DC

capacitor for the STATCOM. Unfortunately, a mislabelled product the XC4514 (Figure 5-2) [70]

from JayCar Electronics meant that the converter originally acquired was actually a DC-DC

Stepdown Converter. Whilst this component is still installed within the system, it has been set

to simply allow the full 12 VDC to pass through to the capacitor.

Figure 5-2. XC4514 DC-DC Converter [70]

5.7. Control Algorithm

The control algorithm is one of the most important features to be completed. This control

strategy is responsible for ensuring the voltage input into the system is at the correct level for

the desired effect. As previously discussed in Chapter 2.3.1.2 when the voltage output from

the STATCOM is below the line voltage the STATCOM is seen as a capacitor and generates

reactive power. In opposition to this when the voltage of the STATCOM is above the line

voltage the STATCOM is seen as an inductor and absorbs reactive power.

In conjunction with the control algorithm a switch allowing the user to change between

power factor control and voltage control should be implemented as part of the system. This

92

would allow for a more versatile and configurable project. Each of these modes would require

a separate control algorithm.

An investigation into these control algorithms would be required. This would involve testing

various scenarios and obtaining an understanding as to what effects the various voltages

levels have on the system and line voltage. Following this, implementing a controller into this

device could be in the form of a PI controller, which would satisfactorily control the voltage

on the line, or if appropriate a PID controller or something far more complex could be

implemented. Typically, this could be done by a student undertaking Industrial Computer

Systems Engineering alongside Instrumentation Control Engineering at Murdoch University.

93

6. Conclusion

Although the project is incomplete at this stage, the low voltage static synchronous

compensator developed over the duration of the project is essentially a working prototype

show below in Figure 6-1. The construction of the STATCOM satisfies the majority of the

initial design objectives with only a few minor adjustments required.

Figure 6-1. Overall Project Enclosures, Power Supplies (Left), Measurement (Centre) and Loads (Right)

The end design utilises a BTN8982TA dual half bridge DC motor driver shield produced by

Infineon Technologies to create the voltage source inverter required to produce the DC – AC

conversion. In conjunction, an LM319 high speed comparator has been utilised to produce

the required control signals for feeding the above VSI. These signals fed to the comparator

are generated using two methods. The triangular carrier is produced through direct digital

synthesis from an Arduino Uno through a third order low pass reconstruction filter, whereas

the sinusoidal control signal is generated from the AC mains voltage through a series

attenuation and biasing.

Although the STATCOM is not fully operational the output waveforms from the VSI appear to

be of a reasonable quality for the fundamentals of the project with the THD% ranging from

4.14% to 37.75% for useable values of ValA.

94

With the phase difference resolved the lines are capable of being connected in parallel. An

investigation into connecting the systems through small resistive values initially should be

undertaken.

The project requires additional work – as previously stated the main objective for completion

would be to write the control algorithms into the code. In addition to this refinement on

filters and adjustment potentiometers would further improve the functionality of this device.

From a construction point of view, the project is considered to have been an overall success,

as the majority of the build portion of the project was finalised. Unfortunately, due to the

numerous setbacks and issues however the project is considered incomplete and not ready

for implementation in a classroom environment. It is hoped that this project will be furthered

and improved upon to allow students more access to equipment to better understand the

electrical engineering concepts and practicality of the degree.

95

Appendices

See following page.

1

A. Software

The following Arduino sketch was written in Arduino IDE 1.0.6 for use with an Arduino Uno. A soft

copy of this file is included in

2

Software on the compact disk that is associated with this project as V5_0_STATCOM_ARDUINO CONTROL CODE

/* this code has been developed by Jordan Goodchild and adapted through the use of many sources, some sources are

linked within the body of the code. Notable sources are Martin Nawrath’s DDS Sine wave Generator code that has been

adapted to output a triangle waveform with other alterations included. */

#include <Wire.h>

#include <LiquidCrystal_I2C.h>

#include "avr/pgmspace.h"

// table of 256 triangle values, one traingular period stored in flash memory

PROGMEM prog_uchar tri256[] =

0x2,0x4,0x6,0x8,0xa,0xc,0xe,0x10,0x12,0x14,0x16,0x18,0x1a,0x1c,0x1e,0x20,




0x81,0x83,0x85,0x87,0x89,0x8b,0x8d,0x8f,0x91,0x93,0x95,0x97,0x99,0x9b,0x9d,0x9f,

0xa1,0xa3,0xa5,0xa7,0xa9,0xab,0xad,0xaf,0xb1,0xb3,0xb5,0xb7,0xb9,0xbb,0xbd,0xbf,

0xc1,0xc3,0xc5,0xc7,0xc9,0xcb,0xcd,0xcf,0xd1,0xd3,0xd5,0xd7,0xd9,0xdb,0xdd,0xdf,

0xe1,0xe3,0xe5,0xe7,0xe9,0xeb,0xed,0xef,0xf1,0xf3,0xf5,0xf7,0xf9,0xfb,0xfd,0xff,

0xfd,0xfb,0xf9,0xf7,0xf5,0xf3,0xf1,0xef,0xed,0xeb,0xe9,0xe7,0xe5,0xe3,0xe1,0xdf,

0xdd,0xdb,0xd9,0xd7,0xd5,0xd3,0xd1,0xcf,0xcd,0xcb,0xc9,0xc7,0xc5,0xc3,0xc1,0xbf,

0xbd,0xbb,0xb9,0xb7,0xb5,0xb3,0xb1,0xaf,0xad,0xab,0xa9,0xa7,0xa5,0xa3,0xa1,0x9f,

0x9d,0x9b,0x99,0x97,0x95,0x93,0x91,0x8f,0x8d,0x8b,0x89,0x87,0x85,0x83,0x81,0x80,




0x1e,0x1c,0x1a,0x18,0x16,0x14,0x12,0x10,0xe,0xc,0xa,0x8,0x6,0x4,0x2,0x0,

; // end of lookup table

//Define Variable Names

#define cbi(sfr, bit) (_SFR_BYTE(sfr) &= ~_BV(bit))

// these are from Martin Nawraths code and define whether a bit is set or not.

#define sbi(sfr, bit) (_SFR_BYTE(sfr) |= _BV(bit))

#define DCCapRelay 7 // DC relay pin

#define INH_1 12

// Infinieon DC Motor Control Shield with BTN8982TA half bridge sleep/wake pin control left half bridge

#define INH_2 13

3

// Infinieon DC Motor Control Shield with BTN8982TA half bridge sleep/wake pin control right half bridge

// LCD Initialisation

LiquidCrystal_I2C lcd(0x27, 2, 1, 0, 4, 5, 6, 7, 3, POSITIVE);

// Set the LCD I2C address https://arduino-info.wikispaces.com/LCD-Blue-I2C

// Initialise Variables

// Voltage Variables

double sum_of_voltage,

square_voltage,

Vrms,

instant_voltage,

voltage_scale,

voltage_offset;

int raw_voltage_data,

number_of_samples,

n;

// Current Variables

double sum_of_current,

square_current,

Irms,

instant_current,

current_scale,

current_offset;

int raw_current_data;

// Power Variables

double instant_power,

sum_inst_power,

real_power,

apparent_power,

power_factor;

// Timer and Interrupt Variables

volatile double dfreq;

const double refclk=31372.549;

byte valA;

const byte valB = 100;

// variables used inside interrupt service declared as voilatile http://gammon.com.au/interrupts

volatile byte icnt; // var inside interrupt

byte icnt1; // var inside interrupt

4

volatile byte c4ms; // counter incremented all 4ms

volatile unsigned long phaccu; // phase accumulator

volatile unsigned long tword_m; // dds tuning word m

int startbridge; // triggers bridge to turn on

void setup() // setup code here:

startbridge = 0;

lcd.begin(20,4); // initialize the lcd for 20 chars 4 lines and turn on backlight

// pinmode define

pinMode(DCCapRelay, OUTPUT); // Charge DC capacitor through relay

pinMode(11, OUTPUT);

// Defines the output pin for the triangle waveform OCR2A will be outputting on this pin

pinMode(INH_1,OUTPUT); // Define INH_1 pin as an output for H-Bridge

pinMode(INH_2,OUTPUT); // Define INH_2 pin as an output for H-Bridge

// Charge capacitor upon start-up

// digitalWrite(DCCapRelay, HIGH); // sets the Relay on

// delay(1000); // waits for a second to charge capacitor.

// digitalWrite(DCCapRelay, LOW); // sets the Relay off

// timer initialisiation code

Setup_timer2(); // call timer2 setup

risefalledge(); // set up interrupt for changing state of 50Hz sinewave triggering.

// disable interrupts to avoid timing distortion

cbi (TIMSK0,TOIE0); // disable Timer0 !!! delay() is now not available

sbi (TIMSK2,TOIE2); // enable Timer2 Interrupt

// defines intial values for system operation

valA = 100; // triangle waveform magnitude variables (100 = 100% (max))

dfreq = 5000.0; // initial output frequency = 5000.0 Hz

tword_m = pow(2,32)*dfreq/refclk; // calulate DDS new tuning word

// initialise pins

digitalWrite(INH_1,0); // initialise D12 as low

digitalWrite(INH_2,0); // initialise D13 as low

// end of set up

void loop()

while(1)

5

if (c4ms > 250) // timer wait for a full second

c4ms=0; // reset 4ms counter

// end of if statement

// reset variable values after for loop calculations

sum_of_voltage = 0;

square_voltage = 0;

sum_of_current = 0;

square_current = 0;

sum_inst_power = 0;

number_of_samples = 3000;

voltage_offset = 2.5;

current_offset = 2.5;

voltage_scale = 19.8;

current_scale = 0.95;

for (n=0; n<number_of_samples; n++)

// Raw input data

raw_voltage_data = analogRead(A3); // raw analog voltage data sample

raw_current_data = analogRead(A2); // raw analog current data sample

// Voltage, current and power conversion calculations

instant_voltage = ((raw_voltage_data * (5.0/1023)) - voltage_offset) * voltage_scale;

// Convert the analog reading to an AC voltage

instant_current = ((raw_current_data * (5.0/1023)) - current_offset) * current_scale;

// Convert the analog reading to an AC current

instant_power = instant_voltage * instant_current; // Calculation for instant real power

// RMS Voltage Calculations

square_voltage = double(instant_voltage * instant_voltage);

sum_of_voltage = sum_of_voltage + square_voltage;

// RMS Current Calculations

square_current = double(instant_current * instant_current);

sum_of_current = sum_of_current + square_current;

// Real Power Calculations

sum_inst_power = sum_inst_power + instant_power;

// end of for loop

// system calculations

6

Vrms = sqrt(sum_of_voltage/number_of_samples); // Vrms calculation

Irms = sqrt(sum_of_current/number_of_samples); // Irms calculation

real_power = abs(sum_inst_power / number_of_samples);

apparent_power = Vrms * Irms;

power_factor = real_power / apparent_power;

LCDdisplay(); // call LCD display function to display data onto LCD screen

// end of while loop

// end of void loop coding

void Setup_timer2() // Timer2 Setup https://www.arduino.cc/en/Tutorial/SecretsOfArduinoPWM

TCCR2A = _BV(COM2A1) | _BV(COM2B1) | _BV(WGM20);

// Output A and B set to clear (10) ** PWM set to PWM to 255 (001)

TCCR2B = _BV(CS20); //prescaler set to 1

// end of timer2 setup function

ISR(TIMER2_OVF_vect) // Interrupt Service Routine Timer2 Overflow

phaccu = phaccu + tword_m; // soft DDS, phase accu with 32 bits

icnt = phaccu >> 24; // use upper 8 bits for phase accu as frequency information

OCR2A=(valA*(pgm_read_byte_near(tri256 + (uint8_t)(icnt)))/valB);

// read value from ROM sine table and send to PWM DAC out on pin 11

if(icnt1++ == 125) // 4ms clock generator 125 loops at 32us

// increment variable c4ms every 4 milliseconds

c4ms++;

icnt1=0;

// end of if statement

// end of ISR function

void LCDdisplay() // LCD screen display

// RMS Voltage

lcd.setCursor(1,0);

lcd.print("V: ");

lcd.setCursor(4,0);

lcd.print(Vrms);

// RMS Current

lcd.setCursor(1,1);

7

lcd.print("I: ");

lcd.setCursor(4,1);

lcd.print(Irms,3);

// Real Power

lcd.setCursor(1,2);

lcd.print("W: ");

lcd.setCursor(4,2);

lcd.print(real_power);

// Apparent Power


lcd.print("VA: ");


lcd.print(apparent_power);

// Power Factor

lcd.setCursor(0,3);

lcd.print("PF: ");

lcd.setCursor(4,3);

lcd.print(power_factor,4);

// end of LCD display function

void risefalledge() //https://www.arduino.cc/en/Reference/AttachInterrupt

attachInterrupt(0, action, CHANGE); // attach interrupt handler

void action ()

phaccu = 0; // reset phase accumulator at zero crossing trigger

TCNT2 = 0; // reset timer 2 counter at zero crossing trigger

startbridge++ ;

// increase startbridge to trigger after a set number of edges detected (50 edges = 25 periods at 50 Hz thus 500ms)

if ( startbridge == 50)

digitalWrite(INH_1,HIGH); // initialise D12 as high

digitalWrite(INH_2,HIGH); // initialise D13 as high

8

B. Full Circuit Drawings

Within this section is the circuit diagrams detailing the overall connections of the system.

Currently four (4) drawings exist:

DWG 000-001: AC AND DC POWER SUPPLY BOX WIRING DIAGRAM

DWG 100-001: MEASUREMENT AND CONTROL BOX WIRING

DWG 100-002: ARDUINO UNO WIRING DIAGRAM

DWG 200-001: LOAD BOX WIRING DIAGRAM

13

C. Relevant Datasheet Information

LM319 Datasheet Information

Figure C-1. LM319 Metal Can Package [56]

Figure C-2. LM319 Typical Applications [56]

14

D. Scaling Ratio Tables

Voltage Scaling Ratio Tables

R/C Vline Vmeas Ratio Vline Vmeas Ratio Vline Vmeas Ratio

41.8uF 83.9uF 126.2uF

12Ω 11.72 0.587 19.972 11.84 0.594 19.937 11.86 0.601 19.735

16Ω 12.11 0.608 19.917 12.22 0.615 19.867 12.33 0.615 20.046

20Ω 12.42 0.622 19.963 12.54 0.629 19.929 12.68 0.636 19.928

28.8Ω 12.79 0.636 20.101 12.9 0.650 19.833 13.06 0.658 19.863

48Ω 13.12 0.650 20.171 13.27 0.665 19.967 13.43 0.672 19.996

150Ω 13.52 0.679 19.920 13.67 0.693 19.730 13.84 0.707 19.576

R/C Vline Vmeas Ratio Vline Vmeas Ratio Vline Vmeas Ratio

170.9uF 204uF 246uF

12Ω 12.04 0.608 19.802 12.1 0.615 19.672 12.18 0.622 19.577

16Ω 12.44 0.622 19.995 12.51 0.629 19.881 12.6 0.629 20.024

20Ω 12.8 0.643 19.895 12.86 0.650 19.771 12.96 0.658 19.711

28.8Ω 13.2 0.665 19.862 13.26 0.665 19.952 13.38 0.679 19.714

48Ω 13.57 0.686 19.787 13.66 0.693 19.715 13.78 0.721 19.109

150Ω 13.98 0.721 19.386 14.08 0.735 19.149 14.22 0.749 18.975

R/L Vline Vmeas Scaling Vline Vmeas Scaling Vline Vmeas Scaling

34.1mH 45.2mH 58.2mH

12Ω 11.56 0.580 19.940 11.57 0.580 19.957 11.58 0.580 19.974

16Ω 11.89 0.601 19.785 11.91 0.601 19.819 11.94 0.601 19.869

20Ω 12.19 0.615 19.818 12.2 0.615 19.834 12.21 0.615 19.851

28.8Ω 12.47 0.629 19.818 12.49 0.629 19.850 12.51 0.629 19.881

48Ω 12.81 0.650 19.694 12.84 0.658 19.528 12.86 0.658 19.559

150Ω 13.24 0.672 19.713 13.27 0.672 19.757 13.28 0.672 19.772

R/L Vline Vmeas Scaling Vline Vmeas Scaling Vline Vmeas Scaling

76.5mH 129.8mH 400mH

12Ω 11.59 0.580 19.992 11.6 0.580 20.009 11.6 0.580 20.009

16Ω 11.96 0.608 19.670 11.97 0.608 19.687 11.98 0.608 19.703

20Ω 12.22 0.615 19.867 12.23 0.615 19.883 12.24 0.615 19.900

28.8Ω 12.53 0.629 19.913 12.57 0.636 19.755 12.58 0.636 19.771

48Ω 12.88 0.658 19.589 12.9 0.658 19.619 12.92 0.658 19.650

150Ω 13.29 0.672 19.787 13.3 0.672 19.802 13.33 0.679 19.640

15

Current Scaling Ratio Tables

R/C Iline Imeas Ratio Iline Imeas Ratio Iline Imeas Ratio

41.8uF 83.9uF 126.2uF

12Ω 0.92 0.835 1.102 0.97 0.903 1.075 1.03 0.993 1.037

16Ω 0.76 0.699 1.087 0.81 0.805 1.006 0.89 0.910 0.978

20Ω 0.62 0.624 0.993 0.68 0.745 0.913 0.77 0.857 0.898

28.8Ω 0.46 0.549 0.838 0.54 0.699 0.772 0.67 0.850 0.788

48Ω 0.32 0.534 0.599 0.44 0.707 0.622 0.56 0.865 0.647

150Ω 0.2 0.579 0.345 0.34 0.737 0.461 0.54 0.888 0.608

R/C Iline Imeas Ratio Iline Imeas Ratio Iline Imeas Ratio

170.9uF 204uF 246uF

12Ω 1.13 1.121 1.008 1.23 1.218 1.009 1.34 1.346 0.995

16Ω 1.01 1.061 0.952 1.11 1.166 0.952 1.25 1.316 0.950

20Ω 0.91 1.015 0.896 1.03 1.143 0.901 1.18 1.294 0.912

28.8Ω 0.83 1.023 0.811 0.95 1.151 0.826 1.13 1.316 0.859

48Ω 0.76 1.030 0.738 0.91 1.166 0.781 1.1 1.346 0.817

150Ω 0.73 1.053 0.693 0.89 1.203 0.740 1.08 1.384 0.780

R/L Iline Imeas Ratio Iline Imeas Ratio Iline Imeas Ratio

34.1mH 45.2mH 58.2mH

12Ω 0.91 0.767 1.186 0.91 0.767 1.186 0.91 0.767 1.186

16Ω 0.74 0.624 1.185 0.74 0.624 1.185 0.74 0.624 1.185

20Ω 0.6 0.504 1.191 0.6 0.504 1.191 0.59 0.504 1.171

28.8Ω 0.44 0.414 1.064 0.44 0.414 1.064 0.43 0.414 1.039

48Ω 0.29 0.354 0.820 0.28 0.354 0.792 0.27 0.354 0.764

150Ω 0.11 0.346 0.318 0.11 0.346 0.318 0.1 0.346 0.289

R/L Iline Imeas Ratio Iline Imeas Ratio Iline Imeas Ratio

76.5mH 129.8mH 400mH

12Ω 0.9 0.760 1.185 0.9 0.760 1.185 0.89 0.760 1.172

16Ω 0.73 0.624 1.169 0.73 0.624 1.169 0.72 0.624 1.153

20Ω 0.59 0.504 1.171 0.58 0.504 1.151 0.58 0.504 1.151

28.8Ω 0.43 0.414 1.039 0.42 0.414 1.015 0.42 0.414 1.015

48Ω 0.27 0.354 0.764 0.27 0.354 0.764 0.26 0.354 0.736

150Ω 0.1 0.346 0.289 0.09 0.346 0.260 0.09 0.346 0.260

16

E. STATCOM Output Waveforms

This appendix presents output waveforms from the STACOM with varying amplitude modulation

values. The graph in each case represents the amplitude of the harmonics for each of the

harmonic orders from the 2nd order harmonics through to the 10th order harmonics. In addition,

the magnitude of each harmonic order is presented as a 2 d.p. value.

See following page.

17

Figure E-1. ValA of 100, Output Waveform (Top), FFT Waveform (Centre), Output Harmonics (Bottom)


Order of Harmonics 0.35 18.24 0.96 5.02 0.17 1.82 0.02 0.24 0.09

0.00

2.00

4.00

6.00

8.00

10.00

12.00

14.00

16.00

18.00

20.00

Po

wer

(µ

W)

STATCOM Output Harmonics

18




0.00

5.00

10.00

15.00

20.00

25.00

30.00

35.00

40.00

Po

wer

(µ

W)

STATCOM Ouput Harmonics

19




0.00

5.00

10.00

15.00

20.00

25.00

30.00

Po

wer

(µ

W)


20




0.00

5.00

10.00

15.00

20.00

25.00

30.00

35.00

40.00

45.00

Po

wer

(µ

W)


21




0.00

10.00

20.00

30.00

40.00

50.00

60.00

70.00

80.00

Po

wer

(µ

W)


22




0.00

50.00

100.00

150.00

200.00

250.00

300.00

Po

wer

(µ

W)


23




0.00

500.00

1000.00

1500.00

2000.00

2500.00

Po

wer

(µ

W)

STATCOM OutputHarmonics

24



Order of Harmonics 95.73 4,581.74 31.70 502.38 87.30 458.17 18.24 50.24 2.64

0.00

500.00

1,000.00

1,500.00

2,000.00

2,500.00

3,000.00

3,500.00

4,000.00

4,500.00

5,000.00

Po

wer

(µ

W)


25




0.00

2000.00

4000.00

6000.00

8000.00

10000.00

12000.00

Po

wer

(µ

W)


26




0.00

2000.00

4000.00

6000.00

8000.00

10000.00

12000.00

14000.00

16000.00

Po

wer

(µ

W)


27

Works Cited

[1] S. K. O. Deeksha Bansal, “Mitigation of Voltage Sag and Swell using D-STATCOM to

improve Power Quality,” IJISET - International Journal of Innovative Science, Engineering &

Technology, June 2014. [Online]. Available: http://ijiset.com/v1s4/IJISET_V1_I4_07.pdf.

[Accessed 05 March 2016].

[2] Festo Didactic Inc., “Festo Didactic,” 2015. [Online]. Available: https://www.labvolt.com/.

[Accessed 05 January 2016].

[3] A. M. B. S. Sunil Panda, “Control of Voltage Source Inverters using PWM/SVPWM,” May

2009. [Online]. Available:

http://ethesis.nitrkl.ac.in/1133/1/Control_of_Voltage_Source_Inverters_using_PWM.pdf.

[Accessed 20 December 2015].

[4] T. M. U. W. P. R. Ned Mohan, “Power Electronics: Converters, Applications and Design,” in

Chapter 8: SWITCH-MODE dc-ac INVERTERS dc - SINUSOIDAL ac, New Jersey, John Wiley &

Sons, Inc., 2003.

[5] C. H. Mei, “Single Phase Half Bridge Inverter,” 2012. [Online]. Available:

http://www.powere.dynamictopway.com/i6.htm. [Accessed 20 December 2015].

[6] M. E. E. Edison R. C. da Silva, “Fundamentals of Power Electronics,” 2013. [Online].

Available:

https://www.google.com.au/url?sa=t&rct=j&q=&esrc=s&source=web&cd=1&ved=0ahUKE

wjs3dD8vJfLAhUJOJoKHVNqD24QFggbMAA&url=http%3A%2F%2Fwww.springer.com%2Fc

da%2Fcontent%2Fdocument%2Fcda_downloaddocument%2F9781447151036-

c2.pdf%3FSGWID%3D0-0-45-1404710-p175009438&. [Accessed 20 December 2015].

[7] C. I. ODEH, “Cascaded half-full-bridge PWM multilevel inverter configuration,” Department

of Electrical Engineering, University of Nigeria, Nsukka, 2014.

[8] J. D. Reeder, “Controlling The Real World With Computers,” learn-c.com, 2011. [Online].

Available: http://www.learn-c.com/experiment7.htm. [Accessed 21 December 2015].

[9] Christoph, “SINGLE PHASE INVERTER TUTORIAL,” 25 July 2013. [Online]. Available:

https://barela.wordpress.com/2013/07/25/single-phase-inverter-tutorial/. [Accessed 21

December 2015].

[10] T. M. U. W. P. R. Ned Mohan, “Power Electronics Converters, Applications and Design,” in

Chapter 7: full bridge dc-dc converter, New Jersey, John Wiley & Sons Inc., 2003, p. 189.

[11] H. S. W. Anuja Namboodiri, “Unipolar and Bipolar PWM Inverter,” Vishwakarma Govt.

Engineering College, Chandkheda, December 2014. [Online]. Available:

http://www.ijirst.org/articles/IJIRSTV1I7111.pdf. [Accessed 22 December 2015].

28

[12] G. Glanzmann, “Flexible Alternating Current Transmission Systems,” 14 January 2005.

[Online]. Available:

https://www.eeh.ee.ethz.ch/uploads/tx_ethpublications/Glanzmann_FACTS_internal.pdf.


[13] Hermoine12, “Examples of FACTS for shunt compensation (schematic),” Wikipedia, 1

August 2007. [Online]. Available:

https://en.wikipedia.org/wiki/File:FACTS_shunt_compensation1.PNG. [Accessed 28

December 2015].

[14] Hermoine12, “Examples of FACTS for series compensation (schematic),” Wikipedia, 1

August 2007. [Online]. Available:

https://en.wikipedia.org/wiki/File:FACTS_series_compensation1.PNG. [Accessed 28

December 2015].

[15] L. G. Narain G. Hingorani, Understanding FACTS: Concepts and Technology of Flexible AC

Transmission Systems, New York: IEEE Press, 2000.

[16] R. M. a. R. Varma., Thyristor-based facts controllers for electrical transmission, Piscataway:

IEEE Press, 2002.

[17] Y.-H. Song., Flexible ac transmission systems (FACTS)., London: The Institution of Electrical

Engineers, 1999.

[18] ABB Australia, “Static Var Compensators SVCs,” 2015. [Online]. Available:

http://www.abbaustralia.com.au/cawp/gad02181/c1256d71001e0037c1256bd60043b75c

.aspx. [Accessed 28 December 2015].

[19] Arduino, “Introduction,” 2015. [Online]. Available:

https://www.arduino.cc/en/Guide/Introduction. [Accessed 28 December 2015].

[20] H. Barragán, “Wiring,” 2015. [Online]. Available: http://wiring.org.co/. [Accessed 28

December 2015].

[21] H. Barragán, “Wiring Copyright,” 2015. [Online]. Available:

http://wiring.org.co/copyright.html. [Accessed 28 December 2015].

[22] C. R. Ben Fry, “Processing,” 2015. [Online]. Available: https://processing.org/. [Accessed 28

December 2015].

[23] C. R. Ben Fry, “Processing Copyright,” 2015. [Online]. Available:

https://processing.org/copyright.html. [Accessed 28 December 2015].

[24] M. Chapman, “Development of a Low Voltage Three Phase Power Supply for Educational

Use,” Murdoch University, Murdoch, 2013.

[25] Arduino, “Arduin Uno/Genuino,” 2015. [Online]. Available:

https://store.arduino.cc/product/GBX00066. [Accessed 28 December 2015].

29

[26] Atmel Corporation, “ AVR® 8-Bit Microcontroller,” 2009. [Online]. Available:

http://www.atmel.com/Images/doc8161.pdf. [Accessed 28 December 2015].

[27] Arduino, “Genuino Mega2560,” 2015. [Online]. Available:

https://store.arduino.cc/product/GBX00067. [Accessed 28 December 2015].

[28] XE, “XE Currency Converter,” 28 December 2015. [Online]. Available:

http://www.xe.com/currencyconverter/convert/?Amount=20&From=GBP&To=AUD.


[29] Arduino, “Arduino Uno,” 2015. [Online]. Available:

https://www.arduino.cc/en/Main/ArduinoBoardUno. [Accessed 01 July 2015].

[30] M. Nawrath, “Arduino DDS Sinewave Generator,” Academy of Media Arts Cologne, 2011.

[Online]. Available: http://interface.khm.de/index.php/lab/interfaces-advanced/arduino-

dds-sinewave-generator/. [Accessed 23 September 2015].

[31] C. S. Eva Murphy, “All About Direct Digital Synthesis,” Analog Devices, August 2004.

[Online]. Available: http://www.analog.com/library/analogDialogue/archives/38-

08/dds.pdf. [Accessed 28 December 2015].

[32] Analog Devices, “A Technical Tutorial on Direct Digitial Synthesis,” 1999. [Online].

Available: http://www.ieee.li/pdf/essay/dds.pdf. [Accessed 28 December 2015].

[33] Analog Devices, “Fundamentals of Direct Digital Synthesis (DDS),” 2009. [Online].

Available: http://www.analog.com/media/en/training-seminars/tutorials/MT-085.pdf.


[34] Arduino, “Arduino Board Uno,” 2015. [Online]. Available:

https://www.arduino.cc/en/Main/ArduinoBoardUno. [Accessed 14 December 2015].

[35] A. R. Hambley, Electrial Engineering Principles and Applications, New Jersey: Pearson,

2011.

[36] Micron, “Q1150A DIGITAL LC METER,” 25 March 2015. [Online]. Available:

http://www.altronics.com.au/p/q1150a-digital-inductance-capacitance-meter/. [Accessed

22 September 2015].

[37] C. R. Nave, “Approximate Inductance of a Toroid,” 2014. [Online]. Available:

http://hyperphysics.phy-astr.gsu.edu/hbase/magnetic/indtor.html. [Accessed 23

September 2015].

[38] Wikimedia, “Permeability_(electromagnetism),” 28 August 2015. [Online]. Available:

https://en.wikipedia.org/wiki/Permeability_(electromagnetism). [Accessed 23 September

2015].

30

[39] Altronics Distributors Pty Ltd., “M5345A 45 + 45 160VA Toroidal Transformer,” 2015.

[Online]. Available: http://www.altronics.com.au/p/m5345a-powertran-45-45-160va-

toroidal-transformer/. [Accessed 17 December 2015].

[40] Wikimedia Foundation, Inc., “Permeability (electromagnetism),” 4 December 2015.

[Online]. Available: https://en.wikipedia.org/wiki/Permeability_(electromagnetism).


[41] Jaycar Electroncis, “3mH Crossover Inductor,” 2015. [Online]. Available:

https://www.jaycar.com/productView.asp?ID=LF1332&CATID=85&form=CAT2. [Accessed

19 December 2015].

[42] Arduino, “TUTORIALS > Examples from Libraries > LiquidCrystal > HelloWorld,” 17 August

2015. [Online]. Available: https://www.arduino.cc/en/Tutorial/HelloWorld. [Accessed 18

December 2015].

[43] HobbyTronics Ltd, “Arduino LCD Keypad Shield,” 2015. [Online]. Available:

http://www.hobbytronics.co.uk/arduino-lcd-keypad-shield. [Accessed 23 September

2015].

[44] NXP Semiconductors , “UM10204 I2C-bus specification and user manual,” 4 April 2014 .

[Online]. Available: http://www.nxp.com/documents/user_manual/UM10204.pdf.


[45] Pictronics, “HD44780 LCD Alphanumeric,” 2010. [Online]. Available:

http://www.pictronics.com.br/artigos-tecnicos/43-eletronica-e-automacao/74-display-lcd-

hd44780-lm016l.html. [Accessed 18 December 2015].

[46] Tangient LLC, “LCD-Blue-I2C wikipsaces,” 2015. [Online]. Available: https://arduino-

info.wikispaces.com/LCD-Blue-I2C. [Accessed 18 December 2015].

[47] R. Elliott, “Elliot Sounds Production,” 08 January 2011. [Online]. Available:

http://sound.westhost.com/appnotes/an005.htm. [Accessed 02 July 2015].

[48] W. Storr, “Current Transformer Basics,” Basic Electronics Tutorials, 17 December 2015.

[Online]. Available: http://www.electronics-tutorials.ws/transformer/current-

transformer.html. [Accessed 18 December 2015].

[49] S. Zeichnung, “Current Transformer,” 15 April 2009. [Online]. Available:

https://upload.wikimedia.org/wikipedia/commons/archive/3/3b/20090415151550%21Str

omwandler_Zeichnung.svg. [Accessed 14 January 2016].

[50] RS Components Australia, “Alttec L01-6210 5A Through Hole Current Transformer,

1000:1,” Alttec, 2015. [Online]. Available: http://au.rs-online.com/web/p/current-

transformers/8329781/. [Accessed 18 December 2015].

31

[51] Open Energy Monitor, “CT sensors - Interfacing with an Arduino,” 2012. [Online].

Available: http://openenergymonitor.org/emon/buildingblocks/ct-sensors-interface.

[Accessed 23 September 2015].

[52] W. Storr, “The Full Wave Rectifier,” Electronics Tutorials, August 2014. [Online]. Available:

http://www.electronics-tutorials.ws/diode/diode_6.html. [Accessed 30 December 2015].

[53] OpenEnergyMonitor, “Open Energy Monitor Introduction,” 2015. [Online]. Available:

http://openenergymonitor.org/emon/. [Accessed 30 December 2015].

[54] OpenEnergyMonitor, “Measuring AC Voltage with an AC to AC power adapter,” 2015.

[Online]. Available: http://openenergymonitor.org/emon/buildingblocks/measuring-

voltage-with-an-acac-power-adapter. [Accessed 30 December 2015].

[55] Altronics, “M7024A 3VA 12+12V PCB Transformer,” PowerTran, 2015. [Online]. Available:

http://www.altronics.com.au/p/m7024a-powertran-3va-12+12v-pcb-transformer/.


[56] National Semiconductor Corporation, “LM119/LM219/LM319 High Speed Dual

Comparator,” August 2000. [Online]. Available: http://www.ti.com/lit/ds/symlink/lm319-

n.pdf. [Accessed 29 December 2015].

[57] Talkingelectroncis.com, “The H-Bridge Design 6,” 01 February 2012. [Online]. Available:

http://www.talkingelectronics.com/projects/H-Bridge/H-Bridge-1.html. [Accessed 02 July

2015].

[58] Infineon, “Motor Control Shield With BTN8982TA for Arduino,” March 2013. [Online].

Available: http://www.infineon.com/dgdl/Infineon-

Motor_Control_Shield_with_BTN8982TA_for_Arduino-UM-v00_09-

EN.pdf?fileId=5546d4624ca27d02014cb20b89867eed&ack=t. [Accessed 23 September

2015].

[59] Infineon Technologies AG , “BTN8982TA High Current PN Half Bridge Infineon,” Infineon

Technologies , 17 May 2013. [Online]. Available: http://www.infineon.com/dgdl/Infineon-

BTN8982TA-DS-v01_00-EN.pdf?fileId=db3a30433fa9412f013fbe32289b7c17. [Accessed 02

March 2016].

[60] Infineon Technologies AG, “BTN8982TA,” Infineon, 2015. [Online]. Available:

http://www.infineon.com/cms/en/product/power/motor-control-and-gate-driver-

ics/intelligent-motor-control-ics/single-half-bridge-

driver/BTN8982TA/productType.html?productType=db3a30443ef951e3013f0f6c88742068

. [Accessed 30 December 2015].

[61] A. AKA, “designing LC filter for linearizing PWM output,”

http://electronics.stackexchange.com/, 9 April 2014. [Online]. Available:

http://electronics.stackexchange.com/questions/105931/designing-lc-filter-for-linearizing-

pwm-output. [Accessed 30 December 2015].

32

[62] M. S. A. A.-D. S. M. M. A. Reznik, “LCL Filter Design and Performance Analysis for Grid

Interconnected Systems,” 2015. [Online]. Available:

http://inside.mines.edu/~mSimoes/documents/paper_54.pdf. [Accessed 30 December

2015].

[63] D. J. Kumar, “Size Selection Of Energy Storing Elements For A Cascade Multilevel Inverter

STATCOM,” October 2012. [Online]. Available: http://www.ijert.org/view-pdf/1327/size-

selection-of-energy-storing-elements-for-a-cascade-multilevel-inverter-statcom. [Accessed

23 September 2015].

[64] M. A. HANNAN, “Effect of DC Capacitor Size on D-STATCOM Voltage Regulation

Performance Evaluation,” 2012. [Online]. Available:

http://pe.org.pl/articles/2012/12a/50.pdf. [Accessed 23 September 2015].

[65] Jaycar Electronics, “4000uF 75V RG Chassis Mount Electrolytic Capacitor RU6708,” 2015.

[Online]. Available: http://www.jaycar.com.au/Passive-

Components/Capacitors/Electrolytic/4000uF-75V-RG-Chassis-Mount-Electrolytic-

Capacitor/p/RU6708. [Accessed 18 December 2015].

[66] Learning about Electronics, “Peak-to-peak Voltage (VPP) Calculator,” 2015. [Online].

Available: http://www.learningaboutelectronics.com/Articles/Peak-to-peak-voltage-

calculator.php#peak. [Accessed 08 January 2016].

[67] Microsemi Corporation, “Basic Total Harmonic Distortion (THD) Measurement,” January

2015. [Online]. Available: http://www.microsemi.com/document-

portal/doc_view/134813-an30-basic-total-harmonic-distortion-thd-measurement.


[68] L. Bumm, “Lab 2: Voltage Dividers,” 2015. [Online]. Available:

http://www.nhn.ou.edu/~bumm/ELAB/Labs/lab02_VD_v1_0_0.html. [Accessed 02

January 2016].

[69] CadSoft Eagle, “Eagle 7.3.0,” 2011. [Online]. Available:

http://www.cadsoftusa.com/download-eagle/. [Accessed 02 January 2016].

[70] JayCar Electronics, “XC4514 Arduino Compatible DC - DC Booster Module,” 2015. [Online].

Available: http://www.jaycar.com.au/Kits%2C-Science-%26-Learning/Science-Lab-

Equipment/Instruments/Arduino-Compatible-DC---DC-Booster-Module/p/XC4514.


[71] Toronto Surplus and Scientific Inc., “LabVolt EMS 8251,” 2015. [Online]. Available:

http://www.torontosurplus.com/test-equipment/educational-training/lab-volt-labvolt-

8251-capacitor-start-motor-resistance-split-phase-motor-ems-electromechanical-training-

system.html. [Accessed 02 January 2016].

[72] Festo Didactic Inc,, “Data Acquisition and Control Interface (without Function Set) 9063,”

2015. [Online]. Available:

33

https://www.labvolt.com/solutions/6_electricity_and_new_energy/50-9063-

00_data_acquisition_and_control_interface_without_function_set. [Accessed 05 January

2016].

[73] Festo Didactic Inc., “IGBT Chopper/Inverter 8837-B,” 2015. [Online]. Available:

https://www.labvolt.com/solutions/6_electricity_and_new_energy/50-8837-

B0_igbt_chopper_inverter. [Accessed 05 January 2016].

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