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Abstract—This project aimed to design a wireless powering

circuit to power and control a left ventricular assist device (LVAD) that was both efficient and functional. The approach taken to achieve this was to first simulate and calculate proper design characteristics. This was followed by implementation into physical manifestations that were analyzed to verify effectiveness and efficiency. The third step, although not fully completed, was to create a single PCB that included all the various components. Innovations from previous attempts at this project were successful implementation of a feedback circuit, wireless measurement of power and wireless control of speed.

Index Terms — Amplifier, Critical Coupling, Feedback, Left Ventricular Assist Device (LVAD), Motor controller, Radiofrequency (RF) Link, Rectifier, Receiver, Relax Oscillator, Resonant Frequency, Transmitter, Zero Cross Detector

I. INTRODUCTION

IRELESSLY-POWERED left ventricular assist devices aim to both provide non-tethered power to the implant and

also externally control implant flow output in the circulation. The main reason to switch to a wireless power and communication system versus a wired system is the fact that a wired system makes the patient more prone to infection and the surgery more invasive. The wired system has a driveline coming out of the abdomen wall, essentially an open wound, that is then connected to the power and control system. A wireless system negates the need for the driveline and as such the complications for the surgery are reduced and the patient is less likely to suffer from infections [1]. The procedure to implant the wireless system is also less invasive as compared to the wired system.

In this paper, we discuss the development of a wireless powering circuit that both transmits power to the Jarvik 2000 Child LVAD but also provides control of the device speed via wireless communication. A block-diagram of the entire system developed in this paper is shown in Figure 1. The fundamental principle that allows for wireless power transmission is mutually-coupled inductors, called a transmitter and receiver, tuned to operate at a resonant frequency. Moreover, various circuits are attached to both transmitter and receiver coils to generate, amplify and interpret the transmitted power signal. Attached to the transmitting coil is a feedback circuit which enables for the efficient power transfer across tissue despite

changes in alignment between the transmitting and receiving coils. In addition, an oscillator block generates an initial signal through its relax oscillator. A Class E amplifier, used to amplify the signal to provide adequate power, is also connected on the transmitter side. In terms of the receiving circuitry, a rectifier is used to convert the transmitted AC magnetic flux to a DC voltage. The DC voltage is then used to power a motor controller which supplies power to the LVAD. Lastly, a wireless communication circuit enables wireless control of the motor controller through a pulse width modulator (PWM) as well as data acquisition from the LVAD to measure output power. In the following sections, each block within the entire system will be explained.

Fig. 1. Block diagram of wireless LVAD system

II. FEEDBACK CIRCUIT

The feedback circuit is comprised of four main stages: the zero cross detector, relaxation oscillator, PWM and pulse comparator. A block-diagram of the feedback circuitry is shown in Figure 2. Each stage of the feedback circuit functions to adjust the input signal to the amplifier when the transmitting and receiving coils are misaligned relative to each other. The feedback circuit works in direct collaboration with the amplifier, providing its input and receiving its output. The basic design of our feedback circuit used is based from research conducted in the field of transcutaneous power transmission [2], [3].

Nolan R. Bumstead, Timothy A. Forster, Mubashir Mahmood, Paula Delos Santos, Sebastian

D. Antoon, Daniel L. Diaz

Dr. John Valdovinos

California State University, Northridge

November 9, 2018

Design of a Fully Implantable Wireless LVAD System

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Fig. 2. Block diagram for the feedback circuit The process to design and build the feedback circuit started

with PSPICE simulations to ensure that both the correct values and the correct connections are made. An image of the feedback circuitry schematic is shown in Figure 3 below.

Fig. 3. Feedback and amplifier PSPICE schematic Following the completion of the simulations, each

component was built separately on a breadboard. The full circuit was not built on the breadboard due to the contributions of internal capacitances within the breadboard to the output signals at each stage. With our designed plans finalized, the circuit was then assembled step by step on a perforated board. An image of the final circuitry on a perforated board is shown in Figure 4.

Fig. 4. Feedback and amplifier on perforated board

The zero cross detector compares the voltage at the input terminals and returns either the positive supply voltage or zero. With the inverting terminal tied to ground, the comparator senses whether the input to the non-inverting terminal is on a positive or negative swing and outputs positive supply voltage or zero, respectively. This generates a square wave output, switching from high supply voltage to zero whenever the input to the non-inverting terminal crosses zero. Due to the 3.5 V supply voltage of the comparators, the square wave generated by the zero cross detector ideally ranges from 0 V to 3.5 V with a 50% duty cycle. We were able to verify the correct function of our zero cross detector circuit by analyzing the output signal of CH 2 in Figure 5. Due to the peak-peak voltage of 3.16 V and a minimum voltage of 0 V, signified by the marker in the left margin, the circuit showed correct results. The drop in peak-peak voltage from the theoretical value of 3.5 V to the experimental value of 3.16 V is attributed to lowering the supply voltage of the comparator to ensure a clean signal.

Fig. 5. CH 1 input sine wave, CH 2 output square wave, of the zero cross detector. Refer to Appendix A for PSPICE results

Analogous to the zero cross detector, the relax oscillator

functions by comparing the voltage at the input terminals. When the input to the non-inverting terminal is greater, the capacitor, C4, connected to the inverting terminal will charge. On the contrary, once the voltage on the inverting terminal exceeds the voltage on the non-inverting terminal the capacitor discharges. This charging and discharging of the capacitor creates an output signal that ranges from 600 mV to 1.25 V, which is offset due to the 0 V to 3.5 V input square wave. Once power is supplied to the entire circuit, a key responsibility of the relax oscillator in the feedback circuit is to generate a periodic signal at the resonant frequency of the transmitting and receiving coils. This is accomplished through the aforementioned capacitor and leads to the rest of the circuit functioning correctly. Observing Figure 6, we verified the relax oscillator was functioning properly by the output signal produced, measured on CH 2. Ranging from 600 mV to 1.23 V, as shown by the CH 2 marker in the left margin, the output follows the charging and discharging pattern expected. Moreover, the experimental results in Figure 6 resemble the simulation results in Figure AD of the Appendix A.

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Fig. 6. Ch 1 input square wave, CH 2 output charging/discharging capacitor wave of the relax oscillator. Refer to Appendix A for PSPICE results

The pulse width modulator compares the output of the relax

oscillator with a DC voltage generated from three series connected resistors. By varying the DC voltage to the inverting terminal of OP3, the duty cycle of the output signal can be changed. We chose a duty cycle value of 45% as an initial starting point but moved to a value of 41%. This lowering of the duty cycle was implemented because of its positive contribution to the voltage gain of the amplifier. The 41% duty cycle value was accomplished by applying a DC voltage of 1.05 V, which intersects the output of the relax oscillator at 41% of its downward swing. By analyzing Figure 7, we observed that the circuit was functioning correctly by the CH 2 output square wave. In comparison to the input signal in CH 1, the output signal shows a 41% duty cycle which is reduced from the original 50% duty cycle of the input.

Fig. 7. CH 1 output square wave from zero cross detector, CH 2 output square wave with 41% duty cycle from the pulse width modulator. Refer to Appendix

A for PSPICE results

The pulse comparator compares the output signal from the pulse width modulator (Figure 7 CH 2) with the initial input signal from the zero cross detector (Figure 5 CH 1). When the 600 mVpp, 0-offset input sine wave is compared to the 41% duty cycle square wave from 0 V to 3.5 V, a resultant square wave is produced and then fed to the amplifier. We verified the proper function of the pulse comparator circuit by analyzing the

CH 2 output shown in Figure 8. Although the duty cycle is higher than our simulation results found in Figure AH, the resultant square wave produced the expected voltage gain of the amplifier. Ideally the peak-peak voltage of 2.12 V would have been closer to 3 V, but propagated losses in the circuit deteriorated the signal.

Fig. 8. CH 1 input sine wave from zero cross detector, CH 2 output square wave from pulse comparator. Refer to Appendix A for PSPICE results

III. AMPLIFIER

Similar to our choice of feedback network, the amplifier selected was originally designed in previous research [3]. The class E amplifier was used because of its high switching speeds and its ability to operate at high frequencies. Adjustments were made to the design due to the transmitting and receiving coils’ inherit inductances which required specific capacitor values.

The main purpose of the amplifier is to convert a DC supply voltage to an amplified AC signal that powers the transmitting coil. Once the amplifier was constructed, the output from the pulse comparator was connected to the input of the amplifier. The input signal had a peak-peak voltage of 2.12 V as seen in Figure 8. That voltage was then amplified to a peak-peak voltage of 37.6 V as seen in Figure 9. The transmitter and receiver were both tuned to the same resonant frequency of 760 kHz. The choke inductor, L1 from Figure 3, value used for the amplifier was 33 uH. The final values used to obtain that high of a voltage were 10 nF and 16 nF for the series and parallel capacitors, respectively. The output voltage was measured across the transmitting inductor, L2 in Figure 3 (L2 is also referred to as LRx in section IV).

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Fig. 9. CH 1 transmitting coil output, CH 2 receiving coil input

During the testing process of the amplifier and feedback circuits, we found that we were able to create the proper outputs of each step. However, we could not correctly connect the output signal of the feedback circuit to the input of the amplifier. This was due to the inability to overcome the gate to source voltage of the Power MOSFET which was tested to be 10 V peak-peak. This prohibited the direct powering of the amplifier with the feedback circuit. Instead we supplied a 10 V signal from the function generator in order to achieve our fully functioning system as seen in Figure 8. In future design iterations, a non-inverting amplifier should be used to bridge these two circuits.

IV. RF LINKING COILS

A key stage in ensuring wireless power transmission is properly designing and tuning the transmitting and receiving coils, also known as radiofrequency (RF) linking coils. The RF link consists of a transmitting coil that is connected to the amplifier, that would be external to the body, and a receiving coil that is connected to the rectifier and would be implanted subcutaneously. In order to maximize the signal from the transmitter to the receiver, the coils must be operated at their resonant frequency [2].

Fig. 10. The images above show the construction of the 25 mm diameter, hand-wound transmitter coil using Litz wire and the 20 mm diameter receiving coil that was wet etched from copper clad Pyralux laminate

As seen in Figure 10, the transmitting coil was hand-wound out of 26 AWG, nylon insulated Litz wire. This wire is notable for its multistrand composition that reduces losses at higher operating AC frequencies, including those due to the skin effect, proximity effect and eddy current losses [4]. The 17-turn coil was made to a diameter of 25 mm and set in epoxy to give the wire coil structured firmness.

The receiving coil was etched from a Pyralux copper clad laminate. Etching from a Pyralux laminate enabled the coil to be thin, flexible, and relatively small. These features are critical when designing an RF receiving coil that is to be implanted subcutaneously. Using a design from previous senior design groups, a 20 mm diameter coil with 1 mm wide traces and 1mm spacing between traces, was laser printed onto the copper surface, shown in Figure BA of Appendix B. Wet etching using a ferric chloride solution (FeCl3) was necessary to remove unwanted copper from the Pyralux copper clad laminate sheet. This process involved submerging the cut-out coil in the solution for up to 10 minutes. The solution was heated to 50 and a stirring mechanism was utilized to ensure the reaction was maintained. The FeCl3 solution removed all copper from the Pyralux that was not covered in ink. Once the ink and remaining residue were cleaned with acetone, the fresh copper receiving coil was all that was left on the sheet, as seen in Figure 10. 30 AWG, flexile silicone wire was soldered to each end of the trace to complete the receiving coil circuit.

Using an RLC meter, the inductances of each coil were measured. The 17-turn transmitter, LTx, measured 3 µH. The 3-turn receiver, LRx, measured 0.4 µH. Since the operating frequency was known, finding resonance was reduced to solving for the capacitances that would enable coupling [2]. Taking equation (1) with 𝑓equal to the operating frequency of 760 kHz, ω equal to the angular frequency, L being the inductance of the coil and C the capacitance, it is possible to derive the capacitance for the transmitter, CTx (2), as well as for the receiver, CRx (3), that are required to achieve critical coupling.

2𝜋𝑓 = 𝜔 =√

(1)

𝑐 = (2)

𝑐 = (3)

The transmitting and receiving coils were tested to determine

the signal transfer across both inductors at various frequencies. A testing rig that aligns both coils was designed in Solidworks and 3D printed using the Formlabs Form 2 printer as seen in Figures BB and BC of Appendix B. The final design consists of a stand with a 20 mm diameter recess of 0.2 mm for the receiving coil, a stand with a 25 mm diameter recess of 2 mm for the transmitting coil, and a base to make sure the stands would be properly aligned for testing to ensure critical coupling. The assembly is shown in Figure 11.

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Fig. 11. 3D printed transmitter and receiving coil testing stands and base aligner

Using a network analyzer, the transmission coefficient for the

coupled transmitter and receiver system was determined for a range of frequencies, displayed in Figure BD of Appendix B. From this test, we verified that the coils resonated at 760 kHz, since this is the frequency at which maximum transmission occurred. This meant that despite increasing distances between the transmitter and receiver coils, the magnitude of the voltage transfer function was maximized and allowed for power transfer up to 10 mm, including through synthetic skin that was fabricated from previous research. The transmission coefficient for various separation distances between transmitting and receiving coils was measured. In addition, the transmission coefficient for the coils with a material mimicking skin was also measured. The results are shown in Figure 12. With synthetic skin which had an average thickness of 4 mm, the magnitude of the transmission coefficient was 0.13, nearly the same result as the 4 mm magnitude across air.

Fig. 12. The above graph shows the voltage transfer magnitude at the resonant frequency of 760kHz

V. RECTIFIER

The rectifier converts the transmitted AC power signal from the receiving coil into the DC signal needed to power the motor controller. The rectifier used is a full wave rectifier. This circuit utilizes four diodes in a bridge configuration that take the positive and negative cycles of the AC wave and invert the negative cycle, outputting two positive cycles with double the frequency. A capacitor across the DC terminals smooths and

provides a constant DC signal that goes to a load [5]. The rectifier used is the MB16S with four Schottky diodes built in a surface mount package [6]. Schottky diodes are utilized due to the 760 kHz input signal from the receiving coil.

The signal from the receiving coil is connected to the AC pins on the rectifier. The DC signal is then output from the positive and negative terminals on the rectifier. Equation (4) is used to calculate a smoothing capacitor where ILOAD is the load current in A, 𝑓 is the frequency in kHz, and VRIPPLE is the ripple voltage in mV [5]. Capacitors from 10 µF to 220 µF were used.

𝑉 = 𝑓

(4)

Simulation of the rectifier was performed on PSPICE prior

to construction, as shown in Figure 13. The simulation was tested with an AC signal of 20 V amplitude at 60 Hz and 760 kHz with a 1 kΩ resistor and a 220 µF capacitor.

Fig. 13. PSPICE schematic of the rectifier circuit

The simulation confirms the correct configuration for a bridge rectifier, as only the positive half cycles are output, as shown in Figure CA of Appendix C. For comparison, the circuit was tested at 760 kHz with the output displayed in Figure CB of Appendix C. A 220 µF capacitor was put in parallel with the load resistor to smooth the alternative signal to a DC signal. The output can be seen in Figure CC of Appendix C with a constant DC signal around 19 V.

Prior to connecting to the motor controller, the rectifier was first bench tested and then soldered to a perforated board. The results from the perforated board circuit are shown in Figure 14, with the rectifier outlined.

0.2090.187

0.140.103 0.074 0.055

0

0.1

0.2

0.3

0 2 4 6 8 10 12

Mag

nitu

de

Distance Apart (mm)

Magnitude of Voltage Transfer vs Sepraration of Coils

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Fig. 14. Rectifier and Motor Controller circuit on perforated board The input to the circuit is a 20 VPP, 1 kHz signal as shown in

Figure CD of Appendix C. The full wave rectified output with a 1 kΩ load resistor is shown in Figure CE of Appendix C. This signal only contains the positive peaks of the input wave with double the frequency, indicating that the chip is operating properly. To measure a DC output, a 220 µF capacitor was connected in parallel. The measured output is shown in Figure CF of Appendix C, with a DC value of around 8 V. The ripple of the signal is measured under 100 mV. The bench results match those of the simulation. The amplitude of the bench test was 10 V, whereas the simulation was 20 V. The rectified signals in the simulation and bench test are just shy of those values due to the voltage drop from the diodes.

The rectifier was also tested when connected to the receiving coil and measured with the motor controller acting as the load. The signal from the coil was measured at 759 kHz and 9.6 Vpp, with an amplitude of 4.8 V. The rectified voltage measured is 4.32 V. The measurements are shown below in Figure 15.

Fig. 15. Rectifier output when connected to receiving coil The rectifier chosen must be able to withstand a specific peak

inverse voltage (PIV). The PIV is equal to twice the peak voltage. In our case, for a peak-peak voltage of 20 V, this would be 40 V. The two rectifiers close to our specifications were the MB14S with a 40 V PIV and the MB16S with a 60 V PIV. The MB16S was chosen to provide safety if a higher peak-peak voltage was input. The maximum RMS voltage is also taken

into consideration. From equation (5), a VRMS for a peak voltage, Vp, of 10 V would be 7.07 V. The MB16S is capable of handling a VRMS of 42 V. [7]

𝑉 = 0.707 ∗ 𝑉 (5)

VI. MOTOR CONTROLLER

The motor controller used in the circuit is the DRV11873PWPR chip. This is a 16 pin IC motor controller that provides a three-phase wye output. The soldered motor controller can be seen in Figure 14. A supply voltage input ranging from 5 V to 20 V is required to power the motor controller. This signal is provided from the output of the rectifier previously mentioned. The controller also requires a PWM square wave input ranging from 0 V to 5 V with a frequency range of 7 kHz to 100 kHz. Increasing or decreasing the DC input voltage or the PWM duty cycle will increase or decrease the RPM of the motor, respectively. Figure CG of Appendix C shows the schematic for the motor controller [8].

The output of the DRV11873PWPR is a three-phase motor in a wye configuration, with three phases and a neutral, or common line. However, the motor for the Jarvik 2000 Child LVAD is a three-phase delta connection without a neutral line. In order to connect the Jarvik to the motor controller, a wye to delta conversion must be done. The schematic for the conversion is shown in Figure CH of Appendix C [9]. The three-phase outputs, W, U and V, each have a 1 kΩ resistor connected from the motor controller. The other end of the resistors connect together in a wye configuration. The COM pin is then connected to the ends of the resistors. The outputs of the three phases are also tapped and connected to the Jarvik motor, providing a delta input. The resistor values can be adjusted depending on the motor used.

The evaluation model of the motor controller was first tested with a BA-BL 1230 Brushless Motor. The motor controller on the perforated board will be the next phase of testing. The initial motor startup was achieved with a PWM signal of 25 kHz ranging from 0 V to 5 V, a duty cycle of 20% and a DC input of 12 V. The input voltage and duty cycle could then be changed to decrease or increase the motor speed. The purpose of the test was to prove that the motor controller can increase or decrease the speed of the motor with changes in either supply voltage or PWM duty cycle. Results were collected and plotted using MATLAB and can be seen in Figure 16. The figure displays that an increase in duty cycle increases the current draw from the power supply. Ultimately, this increases the motor speed. A fixed running voltage of 6.4 V allowed for a PWM signal range from 20% to almost 90% before shut down. An increased fixed voltage of 7.5 V and 8.5 V resulted in a shut off at a duty cycle of 40% and 50%, respectively.

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Fig. 16. Increase in Motor Speed with Changing Duty Cycle

Once the motor controller was successfully tested, the Jarvik

pump was connected and tested. The Jarvik was connected to the motor controller with a basin of water to pump through the system. The TS420 transit-time perivascular flowmeter was connected with the ME 10PXL sensor to read the flow at a rate of a liter per minute. The Jarvik pump operated successfully while varying both the duty cycle and input voltage to increase or decrease the flow rate. Data and measurements were taken and plotted using MATLAB. Figure CI in Appendix C shows the results for a change in flow rate due to a change in duty cycle with a fixed input voltage of 15 V. The plot resembles a linear graph where an increase in duty cycle from 20% to 60% increases the flow rate from around 2 L/min to 5 L/min. Another test was performed for a fixed duty cycle of 50%. Flow rate was then measured for a change in input voltage. The results are seen in Figure CJ in Appendix C. The results show an almost linear graph, with an increase in input voltage from 5 V to 18 V corresponding to an increase in flow rate from about 1.25 L/min to 5 L/min. The results conclude that the motor controller was successful in varying the speed of the Jarvik and therefore the flow rate by varying the input voltage and duty cycle of the PWM.

VII. WIRELESS PWM CONTROL

Along with the requirement of this LVAD system to be completely untethered to an external system, a wireless controller for the speed of the pump is required. To begin addressing the requirements of a wireless controller for the speed of the pump, two things are necessary; first, a microcontroller with pulse-width modulation capability and secondly, a Wi-Fi shield that can be attached to the microcontroller to give it Wi-Fi capabilities. To address both necessities, an ESP8266 microcontroller is used. The ESP8266 has relatively low power requirements, and is Wi-Fi enabled with a large array of peripheral circuitry built in.

For the purposes of the wireless PWM control, all that will be mentioned is that the ESP8266 has a 10-bit resolution pulse width modulation circuit. To control this PWM circuit wirelessly, the ESP8266 must receive an integer value between 0 and 1024, from a smartphone or computer device, and use this integer to change the duty cycle of the pulse width modulator. The reason the integer must be between 0 and 1024 is due to the 10-bit resolution of the circuitry: 10 bits provide a 1-1024

range. A value of 0 shuts down the PWM as intended by the manufacturer.

To receive data using the user datagram protocol, a Wi-Fi network was created by the microcontroller, then the code as seen in Figure DA was implemented. This code allows the ESP8266 to monitor a specified port, wait for a packet to come in, and convert the packet’s contents into an integer data type. Once the packet’s contents are received in the form of an integer, the output of the function “getCam” can be used to control the duty cycle of the PWM circuitry by calling the function “analogWrite” which accepts both the pin number and the duty cycle as parameters. By using the function “analogWrite” with the pin parameter as 16 and the duty cycle parameter as the output of the “getCam” function, we achieve wireless pulse width modulator control (16 was chosen arbitrarily). The reason this is important is the motor controller, or the device that controls the speed of the motor, can be controlled via the duty cycle of a square wave.

In actual implementation, the code and PWM circuitry functioned almost exactly as intended. The only discrepancy is the fact that the pulse width modulator does not output a square wave with the exact frequency that is detailed in the code. However, the difference is insignificant, as can be seen in Figure 17.

Fig. 17. CD PWM Sq. Wave with Intended frequency of 25kHz

Using a network traffic analyzer (Wireshark), a packet from a smartphone can be seen broadcasted on port 32001 with the contents being a character array containing the value 310, shown in Figure DB of Appendix D. This value of 310 is then turned into an integer and used as the duty cycle parameter for calling the “analogWrite” function. The decimal value of 310 divided by 1024 is calculated by inputting the integers into float type variables and performing simple division, then rounding down to the lowest hundredth’s place number. It is obvious that this does not meet IEEE standard 754 on floating point numbers and rounding in general, however supreme accuracy was not needed for this project’s purposes. The outcome of the algorithm outlined above is then entered into a character array and broadcasted on the same port (32001) as can be seen in the blue highlighted text “DutCyc=0.30” in Figure DC. This is for error detection and redundancy purposes, as the user knows what duty cycle they are inputting, it is simply for the best that

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the device repeats what the user has said, or in this case broadcasted, to it.

VIII. WIRELESS DATA ACQUISITION

In hopes of allowing the user a better sense of how much power the motor controller is supplying to the motor; a data acquisition circuit is required. Beyond this requirement, the LVAD system must still be wireless in terms of its physical detachment from anything outside of the body. To know the power being transferred to the motor, many measurements could be taken that lead to the same calculation, so the idea of finding two voltage values was redundantly chosen. Knowing the voltage before and after a series resistor between the receiving coil and the motor controller would allow a calculation of current through the resistor, and a rough estimate of how much power the motor controller would be receiving. There may be other circuitry in line with the series resistor, so it is important to note that this series resistor would have to be the closest series resistor to the motor controller, to achieve the greatest accuracy of power delivered to it.

The general circuit path between the receiving coils and the motor controller will be referred to from here on as “the line”. The voltages expected on the line range anywhere from 0-20 volts, and the operating range of the ADC on the microcontroller is 0-3.3 V. To bridge the gap between 20 V and 3.3 V, a 1 by 20 resistive divider is achieved by using the standard resistor values 220 kΩ, 10 kΩ, and 1.5 kΩ. The 220 kΩ, 10 kΩ, and 1.5 kΩ resistors are all placed in series with respect to each other, and in parallel with respect to the line. The voltage across the summation of the 10 kΩ and 1.5 kΩ resistors is probed, and that voltage is effectively a 1/20 multiplication of the voltage on the line. To achieve a wireless data acquisition circuit, while still making good use of the parts needed for the Wireless PWM circuitry, two ESP8266 microcontrollers are needed. This is because the ESP8266 contains one ADC which has only one channel, and the code as seen in Figure DA would allow the wireless communication between two ESP8266 devices. This would allow one device to know both voltage measurements, and from this point the device would be able to calculate the power being delivered to the motor. To read the value of the Analog to Digital Converter, a simple call to the function “AnalogRead”, with the parameter A0, is made. From this point, the data on the second ESP8266 is broadcasted to the first via the function “SendIt” (Appendix D Figure DD), and the first ESP8266 broadcasts the corresponding data for the user. At the time of creation of this report, the full circuitry has not been developed yet, and the exact location on the line that will be probed is not known yet. The circuitry has however been tested, and the corresponding data can be seen in Figure DE, with ADC1 belonging to the ESP8266 that controls the PWM circuitry, and ADC2 belonging to the second ESP8266.

IX. PCB DESIGN

Creating a printed circuit board (PCB) is the final aspect of this project. The PCB design was created on Autodesk’s Eagle software version 9.2.2. Interconnecting the components to each other electronically would minimize the number of components and connections of the final design. Datasheets were collected

per components such as the amplifier, feedback, rectifier, motor driver, PWM, and data acquisition. Datasheets of all components were added to the custom-made library named “LVAD”. Each part was named based on the PSPICE schematic. Initially, the group decided to make two separate PCBs; one for the feedback/amplifier and the other for the rectifier and PWM. Labeling the parts was important to the group to avoid any confusion when interconnecting the components together. The parts were verified by the group members involved in the design phase such as quantity, part name, package type, and datasheets.

As a PCB creator, datasheets were utilized to design each part. It provided the mechanical layout and the part’s schematic. Some components were available on the Eagle library by default while the other parts were made from scratch based on the mechanical layout and/or downloaded from a user-friendly website named “SnapEda.”

To build a device on Eagle, footprint and symbols must be created first. Footprints were based on the respective mechanical layout from the datasheet while the symbol was picked from the default library on Eagle. Once footprints and symbols were developed, the device could now be built and named the same as the ones placed on the component’s PSPICE schematic. When constructing the device, the pins from the footprint must be connected to the pins on the symbols. The connections were verified by looking back at the datasheets.

Moreover, the devices were then connected to one another. The connections were from the PSPICE schematic. All devices and components were built for the production of two PCBs. However, due to complications with some of the parts of the project, the group decided to only develop one PCB, which is the feedback and amplifier of the wireless LVAD. See image below.

Fig.18. Eagle diagram of feedback circuit (an enlarged image can be viewed

in Figure EA of Appendix E)

Lastly, a library was successfully developed on Eagle that consists of footprints, symbols, and devices that were utilized in the implemented design. This library can be modified and utilized for future research that may include manufacturing and marketing of a wireless LVAD for the biomedical field.

X. CONCLUSION The design of our wireless powering circuit that both

transmits power to the Jarvik 2000 Child LVAD and wirelessly controls speed was largely a success. The transmitter side of the design effectively amplified a signal to be transmitted across

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the inducting coils to power the receiving side of the circuit. Moreover, the receiving side of the circuit successfully rectified the incoming signal and powered the motor controller. Along with the PWM, the motor controller was able to successfully power and control the Jarvik 2000 Child LVAD. The pump speed could be increased or decreased by modifications to either the rectified input voltage or the PWM duty cycle, translating to a specific flow rate that could be measured. Wireless implementations of speed control and data acquisition were also effectively utilized, providing an understanding of how well the overall circuit was functioning.

A shortcoming of our design was the inability to successfully connect the feedback circuit to the amplifier. Although successful readings were taken throughout the feedback circuit, ensuring its proper implementation, the MOSFET switch of the amplifier could not be biased with the voltage generated by the feedback circuit. This resulted in the amplifier having to be driven by a function generator when testing the system. If the feedback circuit was able to bias the MOSFET it was shown the circuit could sustain a signal and function autonomously.

REFERENCES [1] A. Kilic, M. Acker and P. Atluri, "Dealing with surgical left ventricular

assist device complications", Jtd.amegroups.com, 2018. [Online]. Available: http://jtd.amegroups.com/article/view/5703/5955. [Accessed: 05- Nov- 2018]

[2] M. Baker and R. Sarpeshkar, "Feedback Analysis and Design of RF Power Links for Low-Power Bionic Systems”, IEEE Transactions on Biomedical Circuits and Systems vol. 1, no. 1, 2013.

[3] P. Troyk and M. Schwan, “Closed-Loop Class E Transcutaneous Power and Data Link for MicroImplants”, IEEE Transactions on Biomedical Engineering vol. 39, no. 6, 1992.

[4] "Litz Wire Benefits and Applications | New England Wire Technologies", Newenglandwire.com, 2018. [Online]. Available: https://www.newenglandwire.com/litz-wire-benefits-and-applications/. [Accessed: 05- Nov- 2018]

[5] Electronics-tutorials.ws “Full Wave Rectifier” [online] Available at: https://www.electronics-tutorials.ws/diode/diode_6.html [Accessed 5 Nov. 2018]

[6] “Micro Commercial Components, ‘1 Amp Surface Mount Schottky Bridge Rectifier 20 to 100 Volts” MB12S thru MB110S datasheet, 13 Jan. 2015.

[7] “RMS Voltage Calculator” [online] Available at: https://www.allaboutcircuits.com/tools/rms-voltage-calculator/ [Accessed 5 Nov. 2018].

[8] Texas Instruments, “DRV11873 12-V, 3-Phase, Sensorless BLDC Motor Driver” DRV11873 datasheet, Nov. 2012 [Revised Nov. 2014]

[9] Texas Instruments, ‘DRV11873/10873 Connection for Delta Wiring Motor (3-wire)” DRV11873/10873 datasheet

APPENDIX A (FEEDBACK/AMPLIFIER)

Fig. AA. PSPICE schematic of the zero cross detector

Fig. AB. PSPICE simulation results of the zero cross detector

Fig. AC. PSPICE schematic for the relax oscillator

Fig. AD. Relax Oscillator PSPICE simulation results

Fig. AE. FPSPICE schematic of the pulse width modulator

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Fig. AF. Pulse width modulator PSPICE simulation results

Fig. AG. PSPICE schematic of the pulse comparator

Fig. AH. Pulse comparator PSPICE simulation results

Fig. AI. Power MOSFET PSPICE simulation results

Fig. AJ. Amplifier output PSPICE simulation results

Fig. AK. Perforated board design of feedback and amplifier circuits

APPENDIX B (RF COILS)

Fig. BA. Laser printed RF coils on paper and Pyralux copper clad laminate

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Fig. BB. Newly 3D printed transmitter and receiver coil stands

Fig. BC. Newly 3D printed transmitter and receiver coil stands

Fig. BD. Using a network analyzer to verify magnitude of voltage transfer

APPENDIX C (RECTIFIER/MOTOR CONTROLLER)

Fig. CA. 60 Hz rectifier output without capacitor PSPICE simulation results

Fig. CB. 760 kHz rectifier output without capacitor PSPICE simulation results

Fig. CC. 760 kHz rectifier output with 220 uF capacitor PSPICE simulation

results

Fig. CD. Test input signal, 20 VPP, 1 kHz

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Fig. CE. Full wave rectified signal

Fig. CF. DC output

Fig. CG. Motor Controller schematic

Fig. CH. Wye to delta schematic

Fig. CI. Flow rate vs. duty cycle

Fig. CJ. Flow rate vs. input voltage

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APPENDIX D (PWM/DATA ACQUISITION)

Fig. DA. Function “getCam” for receiving data

Fig. DB. Broadcasted packet from smartphone containing data: “310”

Fig. DC. Broadcasted packet from ESP8266 reporting data highlighted in blue

Fig. DD. Function “sendIt” for broadcasting data

Fig. DE. Measured and Reported Voltages for ADC 1 & 2

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