EVALUATION OF HIGH POWER IGBTs FOR TRACTION APPLICATIONS

M. A. Hollander, G. E. Zetterberg

ABB Daimler Benz Transportation (Sweden) Ltd, Sweden

Trondheim

Abstract. This paper describes a test circuit for high power IGBTs, aimed for traction applications. A typical application for the IGBTs is Metro cars with 750 V or 1500 V DC supply. The test circuit is used for application oriented type-tests of IGBT modules. Also the calculation program TULIP, which calculates power losses and junction temperatures of semiconductors in a three-phase inverter is described.

Keywords. Traction, Inverter, IGBT, Module

To ensure good quality of traction converters using IGBTs, a test equipment for high power IGBT-modules with maximum ratings 3.3 kV and 2 kA has been designed, built and tested.

Two electrical circuits are used for the type-testing of IGBTs: the Phase-Leg and the Full-Bridge. In both cases two IGBT-modules forming a single-phase switching leg are tested simultaneously; one acting as an IGBT-transistor, the other as an inverse diode. In the Phase-Leg the switching behaviour of the IGBTs can be studied by single switchings and the switching losses can be measured. In the Full-Bridge the IGBTs are stressed thermally. The power losses can be measured

. by comparing measured temperatures with the ones of an earlier made calibration with pure DC-current. Furthermore an application test can be made with the Full-Bridge, in which the IGBTs are exposed to the same conditions as in the real application. The test equipment has been tested successfully and measurements have been made on many different types of IGBT-modules, the largest up to now is with ratings 3.3 kV and 1.2 kA.

The evaluation also includes the calculation program TULIP which calculates the power losses and junction temperature of semiconductor switches used in a three phase inverter. Comparison can be made between semiconductor switches from different suppliers, different driving conditions, different cooling conditions, different operation points and so on.

Temperature- and load-cycling tests are other important issues for traction applications. However, this is not considered here because this asks for a different circuit.

IGBT TEST CIRCUIT

The main requirement of the test circuit is to type test IGBT-modules with maximum ratings of 3.3 kV and 2 kA. In a type-test, one or a few samples are tested and identified in order to be able to compare the properties of that type with another type. One key parameter to be measured is the switching energy.

Control panel Temperature

control

Measuring equipment

380V 3-

50Hz

I

Protection system

Figure 1. Principle circuit diagram of the test circuit

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Control electronics

-'Auxiliary

: phase

I I I I I

The mechanical construction of the test circuit is designed to test IGBTs in different kinds and sizes of housing. Besides the IGBT itself, surrounding components like clamping circuits and driver circuits can be tested with the test circuit. This requirement was met by building the test circuit in a modular way.

The scope of the test circuit described in this paper is shown in the principle circuit diagram in Figure 1. The power supply transforms and rectifies the three-phase voltage from the mains and feeds the DC-link. With the control panel the power supply is switched on and off and the DC-link voltage can be controlled between 0 and 2000 V .

In case of component or system failure, the protection system switches off the power supply and discharges the DC-link capacitors.

Two phases are connected to the DC-link: one Test phase with two IGBTs under test and an Auxiliary phase which is part of the test equipment. With switch S 1 in Figure 1, the selection between the Phase-Leg and the Full-Bridge is made. In the Phase-Leg, only the Test phase is active and single switchings can be made; in the Full-Bridge both phases are active and the IGBTs in the Test phase are stressed thermally.

The control electronics generate suitable gate pulses for the tests in the Phase-Leg and the Full-Bridge.

The temperature control has a dual function . In the Phase-Leg the IGBTs are warmed up to the desired temperature. In the Full-Bridge the IGBTs have to be cooled and the power losses are measured with the help of this system.

The measuring equipment measures the voltage across and the current through the IGBTs.

PHASE-LEG

The switching waveforms are studied by single switchings in the Phase-Leg (Figure 2). In this Phase­Leg, a step down chopper with inductive load, the dynamic parameters of IGBTs can be measured. For converter design, the switching energy is an important parameter to be measured. The switching energy should be known, in order to be able to predict the power loss of the IGBTs in the converter and to dimension the cooling system for the IGBTs. Furthermore, switching speed is important to know when designing low­inductive intermediate links, gate-driver circuits and clamping circuits.

In Figure 2 it can be seen that a clamping-capacitor is used close to the IGBTs in order to minimise the equivalent stray inductance.

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DC·link capacitor

4411f

4.7mF

Figure 2. Arrangement for measuring switching energy

The switching energy is measured in the well-known way [1] by measuring the voltage across, and the current through, the IGBT. The time integral of the product of this voltage and current is the switching energy. An example of the switching waveforms and switching energy for a 3.3 kV / 1.2 kA device is shown in Figure 3.

1200 A

I

Uee \ie

r....

- lk n

I /

vertical : Uee 300 V/div. ie 200 Aldiv. p olf lOOOkW/div. Eon 500 mJ/div.

I I 1650V-

I

o Y , OA

OkW

1':::clL= 1670mJj-OmJ

horizontal: time 2 jlS/div.

Figure 3 Tum-off at Ude = 1650 Y, Ie = 1200 A, Tj = 115°C

The switching energy is measured at different currents, temperatures, DC-link voltages and driving conditions. Figure 4 shows the turn-off energy at Ddc = 1650 V as a function of collector current and junction temperature.

1800

1600

1400 _ 1200

! 1000 I:: 800 0 ~ 600

400

200

0

0

Turn-ofT energy at Udc = 1650 V

300 600

Ie (A)

900 1200

-0-115 "C

~75 "C

--tr- 25 "C

Figure 4 Tum-off energy of an 3.3 kY / 1.2 kA IGBT

Figure 5 shows the voltage across and the current through the IGBT, the switching power and the switching energy at transistor turn-on at Ude = 1650 V and 1.: = 1200 A.

I I ~v 1\

1\ \ "- 1200 A

I--ic \uc< 11 'h. OV, OA

A J f- OkW

/ J It = 880 mJ t--

OmJ vertical : horizontal:

Uee 300 V/div, time ) ~/div .

ie 400 Ndiv. pon 1000 kW/div , Eon 500 mJ/div,

Figure 5 Tum-on at Ude = 1650 V, Ie = 1200 A, Tj = 115°C

Figure 6 shows the turn-on energy at Udc = 1650 V as a function of collector current and junction temperature,

Tum-on energy at Ude = 1650 V

=f 700 -o-115 ' C ::; 600 -<>- 75 ' C .§. 500 c

400 -tr- 25 ' C

" ~ 300 200

100 0

0 300 600 900 1200 Ie (A)

Figure 6 Tum-on energy of an 3,3 kV / 1,2 kA IGBT

The diode reverse recovery energy is measured in a similar way to the described measurement of the switching energy of the transistor.

FULL-BRIDGE

The knowledge about the losses is vital in converter design, It is desirable to verify the switching energy measurement by a measurement of the total power losses, This can normally be done in a prototype converter, but it is of course desired to do this testing before a prototype is built.

With the test circuit described in this paper it is possible to do a type-test in which the IGBT-module is tested under the same conditions as in the application i,e, at the right DC-link voltage, current, cos <p (only the

1.414

fundamental frequency is considered), type of modulation, modulation index and temperature, This kind of test should be seen as an "overall type-test" on the IGBT­modules before they are put into a prototype inverter.

A problem is that both an IGBT-transistor and an inverse diode are present in the same housing, both warming up the module, Because of this, AC-current is applied to the two IGBT-modules in one phase of a full-bridge .

To obtain the right cos <p , and thus the right stress for the IGBT-transistor and the diode, the right ratio of active and reactive power to the load is needed , This can be done by having a partly inductive and partly resistive load. The problem with a (partly) resistive load is that much power is dissipated which asks for a huge power supply. The goal is to obtain the right power factor in the IGBTs under test without unnecessary consumption of active power.

A full-bridge with inductive load is used, Figure 7.

Phase A Phase B

Figure 7 The Full-Bridge for AC-current

By applying a modulation in phase A which will make ua= Da sin(wt) and a modulation in phase B which will make Ub= Db sin(wt + a) a voltage Uload is set across the load inductor Lload ' Figure 8 shows the situation.

Figure 8 Vector diagram of voltages and current in Figure 9.

The angle <p is found between Ua and iload' The IGBTs under test (AI and A2) are stressed in the same way as they will be in the application. It is possible to obtain

the desired <p by varying a; however the amplitude of

the load current Iload is set by Uload, wand Lload ' The load current iload can be varied by varying Db, the amplitude of Ub ' Changing Ub (by increasing the modulation index) changes the angle <p too . The system is therefore difficult to control.

For a given application (ua, cos cp, iload) an EPROM (Erasable Programmable Read Only Memory) with the modulations for both phases is programmed. Using one EPROM has the advantage that the fundamental frequency of Ua and Ub are exactly equal.

As an example a test with a 3.3 kY / 1.2 kA IGBT­module with the following operation data is explained.

The given parameters for the phase under test (phase A)

are:

Udc = 1650 Y Iload = 480 A cos cp = 0.85 fo = 50 Hz SIN-PWM rnA = 0.7 fswitch.A = 1000 Hz

(DC-link voltage) (RMS load current)

(between Ua and iload) (fundamental frequency) (sinusoidal PWM) (modulation index phase A) (switching frequency phase A)

Rth.h.a = 0.035 K!W (thermal resistance of heatsink) T ambient = 30°C (ambient temperature)

The given parameters of the test circuit are:

Lload = 0.75 mH (load inductance) Rload = 8 ron (resistance of load inductor)

The adjustable parameters of the auxiliary phase (phase B, which is part of the test circuit):

a = 10.30 (phase angle between Ua and Ub)

mB = 0.91 (modulation index phase B) fswitch.B = 1000 Hz (switching frequency phase B)

These adjustable parameters are calculated from the given parameters of the phase under test and test circuit. In fact the principle can be considered in the following way:

The DC-link voltage and the pulse pattern to the gates of the phase under test (phase A) are given. Because of this, the phase voltage Ua is given. The amplitude and phase angle (cp) of iload is determined by the way phase B is modulated. Phase B can be regarded as a current generator which is controlled by a and mB.

It should be noted that even the switching frequency of phase B can be adjusted in order to change the higher frequency harmonics of the load current. Here, only the fundamental frequency is considered, which is a good approximation as can be seen in Figure 9.

Figure 9 shows the load current iload and the voltage Uce between the collector and the emitter of IGBT module A2 (in Figure 7).

1 .415

'"'"' ~

! ~ I ~ i / J )

II \., II \., OA

J \. J \. J r--r fh'"' f'""'"

I In

I

OV

vertical: honzontal: i10ad 300 Aldiv. time 5 ms/div. Uce 300 V/div.

Figure 9 Udc = 1650 V, Iload = 480 A, f""itch,A = 1000 Hz.

In the frequency test in the Full-Bridge, the heats ink and the ambient temperature is measured . Before this test is made, the thermal resistance R Ib.h.a of the heatsink is determined with a well-defined constant power dissipation of the IGBT-module. In this calibration, the difference between the heatsink and ambient temperature, ~Th.a is measured at the expected power dissipation for the test in the Full-Bridge. At the time the test is made in the Full-Bridge the temperature

differe nce ~Th.a is measured again and the power dissipation Ploss is determined with :

~oss (1)

In this w'ay the power dissipation of the IGBT-module is determined for the given operation point and given gate­drive conditions.

CALCULA nON OF LOSSES WITH "TULIP"

In order to easily calculate the power losses and the junction temperature of the IGBTs in a three phase inverter from the measurements made in the Phase-Leg, a calculation program called TULIP was developed.

TULIP was designed to compare the power losses and junction temperatures between IGBTs from different suppliers, different cooling conditions, different gate­drive conditions, different operation points and so on.

Only the three semiconductor phases of the inverter are taken into consideration. Snubber circuits, output filters and transformers (if any) are not included in the calculations. Furthermore, higher harmonics are not taken into consideration, TULIP calculates for the stationary (steady-state) case only.

The program consists of a program manager and a library with data on the different IGBTs. The following data is in this library for each type of IGBT: supplier, type, ratings, thermal resistances, driving conditions, on-state voltage of the IGBT and the diode and the switching energy of the IGBT and the diode. The on­state voltage and switching energy is described as a second order polynomial of the current at three different junction temperatures. The program manager uses these data to calculate the power losses of the IGBT and the diode and from this, with the thermal resistances, the junction temperature of the IGBT and the diode.

To calculate the power losses, only one phase (phase A in Figure 7 and 8) of a three phase inverter is considered because of symmetry. Again because of symmetry the losses in switch A 1 and A2 are equal.

To calculate the IGBT conducting losses, the forward voltage drop across the IGBT is assumed to be described by a second order polynomial as a function of (time dependent) current.

(2)

where the constants ace> bce and Cce are determined from measurements.

The average conducting losses, PIGBT, ct>nd,a"e

described by: 1 T

PICHT,cm,d,,,, , = T f aai(t) + b e/ (t) + cj'(t) d(t) o

T

= aI ICBT , ... + bIicBT.mIJ + d· f i;GRT (t) d(t) o

where,

1 IT . 3 () () 3 .J2 (4 3n ) - llCBT t d t = l loud"", - - + m-cosqJ To' 2n 3 8

are

(3)

(4)

(5)

(6)

Now, the IGBT conduction loss is written as a function of known parameters. Please note that this is valid for only one given junction temperature. In order to take the temperature into consideration, this calculation is made with the data at three different junction temperatures. Because the junction temperature is dependent on the losses and the losses are dependent on the junction temperature, TULIP calculates the junction temperature in an iterative way.

The sum of the turn-on and turn-off switching energy of the IGBT is assumed to be described by a second order polynomial as a function of (time dependent) current.

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(7)

where the constants aM' bslV and esw are determined from measurements in the Phase-Leg.

Every time the IGBT switches, the energy described by equation 7 is dissipated. The switching losses should therefore be described by a sequence. The average switching power losses, PIGBT,swirch ,av, is

1/2fm"IChfo

~gbl ,swilch,ave = 10 I, ESWilch[tn] 1.=0

(8)

where:

10 fswitCh

is the frequency of the fundamental wave

is the switching frequency of the IGBT

E SWilch [t n ] is the switching energy which is lost at the

discrete times t n

Please observe that the IGBT switches only during the positive half period of the fundamental wave and so does the diode during the negative half period. Because it is assumed that no higher harmonics are present on the load current llo.d, the series in equation 8 can written as an integral. The average switching losses of the IGBT, PIGBT,S\rirch,a,' .. is given by:

P"" 'lrh.tn., = !"'leh 21. J (a,., + b",.1sin(ax)+ c".12sin2(ax)) d(wt) o

(9)

The total power dissipation in the IGBT-transistor is :

P,CBT ,IOIUJ = ~GBT.cofld . (J\'t + P'CBT .. ndrc:h .(J\'t. (10)

The conducting and reverse recovery losses of the diode are calculated in a similar way as for the IGBT­transistor. The heatsink and junction temperature of the IGBT-transistor and the diode are determined for the stationary state from the calculated power losses with the thermal model in Figure 10.

~ R.. •.• ~

~ ~ ' il. U ] <'

Figure 10 Thermal model for temperature calculation

The thermal resistance Rth.h.• is determined by own measurements, the other thermal resistances in Figure 10 are delivered by the semiconductor supplier.

The junction temperature of the IGBT, 0.lGBT, is:

~.ICBT = R1h.j-cPrCBT.lolal + (R1h •C_ h + R 1h .h- o ) PrCBT.IOIol

+ (R1h •C _ h + R 1h .h_a )~iod,. IOIUI (11 )

In Figure 11 a typical output from the program TULIP is shown. In this figure it can for example easily be seen that the diode losses of supplier Y and Z are almost equal, while the junction temperatures of the diodes are very different. In the same figure it can be seen that this is due to a higher heatsink temperature in case of supplier Z, caused by the high switching losses of the IGBT-transistor of supplier Z. Furthermore the thermal resistance between junction and case Rth.j -c of supplier Z is higher than the one of supplier Y.

TULIP has proved to be a very powerful tool for easy and fast evaluation of the power losses and junction temperatures of IGBT-modules. Despite the relatively

easy equations and all the assumptions made, the calculations are quite accurate. As an example it can be mentioned during the test in the Full-Bridge the measured heatsink temperature was 83°C for supplier Y, while the TULIP-calculation said 85°C.

REFERENCE

1. Petterteig, A.; Rogne, T.: IGBT-turn-off losses - in hard switching and with a capacitive snubber, EPE Conference '91,1991, pp 0.203 - 0.208

ADDRESS OF THE AUTHORS

Adtranz Sweden AB, Dept. DPAK, S-721 73 Vastwls, Sweden. Tel.: +4621322000, Fax.: +46 21 135132

TULIP 1.1 fiJ

Comparison 3.3 kV /1.2 kA IGBT-module running mode

Comparison Transistor losses

1600 TJ= 128 'C

1-100 Tj = lOS 'C

469 Tj = 103 ' C

200

~urrliCf X SUppilcr Y supplier Z

Comparison Diode losses

700

bOO

c p.

Tj= 97 'C 453 462

• Pcond.d1.

~200 t-

100

supplier X supplic:rY suppliaZ

sheetn:une supplier X supplier Y supplier Z RIh.j-c@Tr 0.0120 0.0083 0.0100 K/W Rlh. j<@Di 0.02-10 0.0166 0.01 80 K /W

Rlh. c-h 0.0060 0.0060 0.0060 K t W Thc.atsink 82 85 101 'C

Status: Operation point: operation point ©

component data ©

results ®

Figure II Typical output data of the program TULIP

Vdc = 1650 V; Iphase = 480 A; POUI = 749 kVA;

cos II' = 0.85; m = 0.70; VOUI = 901 V; f swilch = 1000 Hz;

T ambienl = 30 ' C; Rlh.h-a = 0.0350 K!W

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