Marques, F. Medina, and M. Sorolla, Super compact split-ring reso-nators CPW band-pass filters, 2004 IEEE MTT-S Int MicrowaveSymp, Forth Worth, TX, 2004, pp. 1483–1486.

7. F. Falcone, T. Lopetegi, J.D. Baena, R. Marques, F. Martın, and M.Sorolla, Effective negative-� stop-band microstrip lines based on com-plementary split ring resonators, IEEE Microwave Wireless ComponLett 14 (2004), 280–282.

8. F. Falcone, T. Lopetegi, M.A.G. Laso, J.D. Baena, J. Bonache, R.Marques, F. Martın, and M. Sorolla, Babinet principle applied to thedesign of metasurfaces and metamaterials, Phys Rev Lett 93 (2004),197401–197404.

9. C. Nguyen, Development of new miniaturized bandpass filters havingultrawide bandwidth, Electron Lett 30 (1994), 767–768.

10. Y.-S. Lin, W.-C. Ku, C.-H. Wang, and C.H. Chen, Wideband coplanarwaveguide bandpass filters with good stopband rejection, IEEE Mi-crowave Wireless Compon Lett 14 (2004), 422–424.

11. J.-S. Hong and M.J. Lancaster, Microstrip filters for RF/Microwaveapplications, Wiley, New York, 2001.

© 2005 Wiley Periodicals, Inc.

FEASIBILITY OF A LOW-COST HYBRIDTUNABLE PHASE SHIFTER BASED ONNONLINEAR TRANSMISSION LINES

Akil Jrad,1 Raffi Bourtoutian,1 Philippe Ferrari,2 andAmer El Helwani11 LPA, Laboratoire de Physique AppliqueeLebanese UniversityFaculty Of Sciences IIITripoli, Lebanon2 IMEP, Institut de MicroelectroniqueElectromagnetisme et Photonique, ENSERG23 av. des Martyrs BP 25738016 Grenoble Cedex 1, France

Received 22 January 2005

ABSTRACT: This paper presents the realization of a hybrid tunablephase shifter. The principle advantage of the hybrid approach is the lowcost of the varactor diodes. The phase shifter employs a variable-veloc-ity transmission line obtained by periodically loading a coplanarwaveguide (CPW) with the varactors. The designed phase shifter is ableto produce a variable phase shift from 0° to 180° at 3.8 GHz with amaximum insertion loss of 4 dB. The return loss is better than �10 dBover all the phase states. © 2005 Wiley Periodicals, Inc. MicrowaveOpt Technol Lett 46: 286–289, 2005; Published online in Wiley Inter-Science (www.interscience.wiley.com). DOI 10.1002/mop.20967

Key words: tunable phase shifter; transmission line; coplanarwaveguide; hybrid technology

1. INTRODUCTION

Tunable phase shifters are very useful in many high-frequencyapplications: couplers, sweeping antennas, reflectometers, and soforth. Our purpose is to design hybrid voltage-variable phase-shifters. Varactor-loaded transmission lines have been used inseveral applications, using both nonlinear [1–4] and linear (small-signal) [5–7] effects of the diodes. In some recent demonstrations[8, 9], diode-loaded transmission lines have been used as delaylines in antenna applications. In [10], a monolithic phase shifterwas designed with a continuous phase shift from 0° to 360° at 20GHz, a minimum insertion loss of 4.2 dB and a return loss betterthan �12 dB over all the phase states.

In this paper, after providing a brief background theory and thedesign-procedure description, we first design a hybrid phase shifter

optimized to work at 3.8 GHz. The phase shifter uses a coplanar-waveguide (CPW) transmission line periodically loaded by varac-tor diodes mounted in area between the central strip and the groundplane (two diodes for each section). The principle advantage of thisapproach is the low cost of the silicon diodes that have been used.Our phase shifter is able to produce a variable phase shift from 0°to 180° at 3.8 GHz with a maximum insertion loss of 4 dB. Thereturn loss is better than �10 dB over all the phase states. Then wedemonstrate that a 0° to 360° phase shifter can be designed ifdiodes having a more abrupt doping profile are used.

2. BACKGROUND THEORY AND DESIGN

The design of a hybrid nonlinear transmission line is slightlydifferent than the design of a monolithic one. In the hybrid case,the varactor specifications cannot be optimized; they are imposedby the manufacturer components.

2.1. Background TheoryThe equivalent electrical circuit of the high impedance transmis-sion line loaded by varactors is given in Figure 1.

C(V) and Rs are the voltage variable capacitance and the seriesresistance of the reverse biased diodes. Lsect is the length of anelementary section. The phase shifter is comprised of n sections ofsuch elementary section. C(V) is given by the following expres-sion:

C�V� �Cj0

�1 �V

�V���M ,

where Cj0 is the zero biased capacitance, V� is the barrier poten-tial of the diode, and M is the grading coefficient.

If a lumped model is used, the characteristic impedance ZL ofa nonlinear transmission line is expressed as

Figure 1 Transmission-line Model

Figure 2 Minimization of the reflection-coefficient method

286 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 3, August 5 2005

ZL � � L0

C0 �C�V�

Lsect

,

where L0 and C0 are the inductance and the capacitance per unitlength of the high impedance transmission line, respectively, L0 �(Z0/v0); C0 � (1/Z0v0), with Z0 and v0 the characteristic imped-ance and phase velocity, respectively, v0 � (c/��eff), with c thelight velocity, and �eff the relative permittivity.

2.2. DesignTo completely design the phase shifter, we must calculate thefollowing:

● number of sections n,● length of each section Lsect, and● characteristic impedance of the high impedance line Z0.

We use the return loss as a figure of merit to optimize the phaseshifter. The goal was to obtain a minimum reflection coefficient,given by

� �ZL � Zc

ZL � Zc.

Z0 was varied between 70� and 120�, leading to realizabletransmission lines, and nmax was fixed to 20 sections in order tominimize the cost of the circuit.

The complete diode model given by the manufacturer has beenused.

The reflection coefficient minimization method is shown inFigure 2. Vmin � 0 and Vmax leads to

Zmin � � L0

C0 �Cmax

Lsect

and Zmin � � L0

C0 �Cmax

Lsect

.

We can consider that the device has been correctly optimized whenthe same absolute reflection coefficient is obtained for Vmin andVmax, leading to d1 � d2 in Figure 2.

Our experience in working with commercial Schottky diodesindicates that often specifications given by the diode’s manufac-turer are not correct in high frequencies. So specific measurementswere carried out to extract diode’s realistic model.

The optimal design using the extracted model leaded to a 0° to180° phase shifter working at 3.8 GHz with n � 20 sections, Z0 �115�.

2.3. Circuit FabricationDiodes HSMS-286K-BLK (with Sot-323 package) from Agilentwere used. According to the manufacturer’s statements, theSchottky diodes have a series resistance (Rs) of 6�, a reversebreakdown voltage Vbr of �7 V, a grading coefficient M of 0.5,and a zero bias capacitance Cj0 of 180 fF. A high-impedance CPW

Figure 3 (a) Layout and (b) photograph of the fabricated phase-shiftercircuit

Figure 4 Return loss vs. frequency for the two extreme bias voltages 0V and �7 V

Figure 5 Insertion loss versus frequency for the two extreme biasvoltages 0 V and �7 V

Figure 6 Maximal differential phase shift vs. frequency, correspondingto �7-V bias voltages

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 3, August 5 2005 287

line was fabricated on FR4 (�r � 4, h � 1600 �m), with W �0.76 mm, S � 1.5 mm, and Lsect � 7.4 mm.

To preserve symmetry, two varactors were connected in paral-lel from the CPW central conductor to both ground planes. Figure3 shows the fabricated circuit.

3. RESULTS

Microwave measurements were made on a Wiltron 360 networkanalyzer that was calibrated using the TRL method. The circuit iswell matched to 50� and the return loss is better than �10 dB overall phase states, as shown in Figure 4.

The maximum insertion loss at 3.8 GHz occurs at zero bias andis limited to 4 dB (Fig. 5).

Figure 6 shows that the phase shifter is able to produce avariable phase shift from 0° to 180° at 3.8 GHz.

These results permitted us to calculate the real range of varia-tion of the Schottky diodes’ capacitance. Although according tothe manufacturer it was supposed to be 360 to 104 fF, respectively,for Vbias � 0 V and Vbias � �7 V, the simulations led to a tuningrange of 205 to 140 fF (width). These are poor characteristics forachieving wide phase tuning.

4. EXPECTED FUTURE RESULTS

In this section, we present the simulation results obtained usingAgilent 5314 beam-led Schottky diodes at 10 GHz. These diodesare supposed to have a more abrupt profile than the diodes used inthe fabrication of our phase shifter, as well as limited values of theequivalent capacitance of the diode, which permits us to designphase shifters that work at higher frequencies. The capacitancetuning range given by the manufacturer is 130 to 50 fF.

Using these varactor diodes, we designed a distributed phaseshifter able to produce a continuously variable 0° to 360° phaseshift at 10 GHz with a maximum insertion loss of �2 dB andreturn loss better than �14 dB over all the phase states.

The number of section was fixed to 20. Z0 was fixed at 130�to optimize the phase shift. For the same substrate used as above,Lsec � 1 mm optimized the return loss to be better than �14 overall the phase states (Fig. 7).

The maximum insertion loss at 10 GHz occurs at zero bias andis limited to 2 dB due to the limited value of the series resistanceof the diode (Rs � 4�), as shown in Figure 8.

Figure 9 shows the phase shift for three bias voltages corre-sponding to the reference, the average, and the maximal bias. Thephase shifter is able to produce a variable phase shift from 0° to360° at 10 GHz.

5. CONCLUSION

The design of a hybrid microwave tunable phase shifter has beenproposed and demonstrated. The principle advantage of this hybridapproach is the low cost of the diodes used in our research and thesimplicity of fabrication.

The phase shifter that we designed is able to produce a variablephase shift from 0° to 180° at 3.8 GHz with a maximum insertionloss of 4 dB, although the diodes characteristics were not optimum.The return loss is better than �10 dB over all phase states.

These results can be improved if we use better quality diodes(with wider tuning range). We showed that a 0° to 360° hybridphase shifter working at 10 GHz can be realized.

REFERENCES

1. M.J.W. Rodwell, M. Kamegawa, R. Yu, M. Case, E. Carman, and K.S.Giboney, GaAs nonlinear transmission lines for picosecond pulsegeneration and millimeter-wave sampling, IEEE Trans MicrowaveTheory Tech 39 (1991), 1194–1204.

2. M.J.W. Rodwell, S.T. Allen, R. Yu, M. Case, U. Bhattacharya, M.Reddy, E. Carman, M. Kamegawa, Y. Konishi, J. Pusl, R. Pullela, andJ. Esch, Active and nonlinear wave propagation devices in ultrafastelectronics and optoelectronics, Proc IEEE 82 (1994), 1035–1059.

3. R. Yu, M. Reddy, J. Pusl, S.T. Allen, M. Case, and M.J.W. Rodwell,Millimeter-wave on-wafer waveform and network measurements us-ing active probes, IEEE Trans Microwave Theory Tech 43 (1995),721–729.

4. U. Bhattacharya, S.T. Allen, and M.J.W. Rodwell, DC 725-GHzsampling circuits and subpicosecond nonlinear transmission lines us-ing elevated coplanar waveguide, IEEE Microwave Guided Wave Lett5 (1995), 50–52.

5. D. Jager and W. Rabus, Bias dependent phase delay of Schottky-contact microstrip lines, Electronics Lett 9 (1973), 201–203.

6. H. Hasegawa and H. Okizaki, MIS and Schottky slow-wave coplanarstriplines on GaAs Substrates, Proc 12th Euro Microwave Conf, 1982,pp. 328–333.

7. Y. Fukuoka and T. Itoh, Design consideration of uniform and periodiccoplanar Schottky variable shifter; in Proceeding 13th Euro MicrowaveConf, 1983, pp. 278–281.

8. W.M. Zhang, R.P. Hsia, C. Liang, G. Song, C.W. Domier, and N.C.

Figure 7 Return loss for different bias voltages

Figure 8 Insertion loss for different bias voltages

Figure 9 Phase shift for different bias voltages

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Luhman, Jr., Novel Low-Loss Delay Line for Broadband PhasedAntenna Array Applications, IEEE Microwave Guided Wave Lett 6(1996), 395–397.

9. R.P. Hsia, W.M. Zhang, C.W. Domier, and N.C. Luhman, Jr., A hybridnonlinear delay line-based broad-band phased antenna array system,IEEE Microwave Guided Wave Lett 8 (1998), 182–184.

10. A.S. Nagra and R.A. York, Distributed Analog Phase Shifters withLow Insertion Loss, IEEE Trans Microwave Theory Tech 47 (1999),1705–1711.

11. W. Heinrich, Quasi-TEM Description of MMIC Coplanar Lines In-cluding Conductor Loss Effects, IEEE Trans Microwave Theory Tech41 (1993), 45–52.

© 2005 Wiley Periodicals, Inc.

PBG-ASSISTED SHARED-APERTUREDUAL-BAND APERTURE-COUPLEDPATCH ANTENNA FOR SATELLITECOMMUNICATION

Nemai C. Karmakar,1 Md. N. Mollah,2 Shantanu K Padhi,1 andJeffrey S. Fu2

1 Department of Electrical and Computer Systems EngineeringMonash University Clayton CampusVIC., Australia 38002 Communication Research LabSchool of Electrical and Electronic EngineeringNanyang Technological University, SingaporeSingapore 639798

Received 8 February 2005

ABSTRACT: Design and development of a photonic bandgap (PBG)-assisted shared-aperture dual-band orthogonal aperture-fed rectangularmicrostrip patch antenna element, which is suitable for a portable verysmall aperture terminal (VSAT), are presented in this paper. The dual-band dual-polarized antenna element achieves 21% input impedancebandwidth at the S- and C-bands. A comparison of the antenna with andwithout 2D PBG grids shows that the inclusion of PBG structures(PBGSs) improves the antenna performances. © 2005 Wiley Periodicals,Inc. Microwave Opt Technol Lett 46: 289–292, 2005; Published onlinein Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.20968

Key words: VSAT; return loss; bandwidth; PBG; ACPA

1. INTRODUCTION

Recently, VSATs have drawn the attention of antenna researchersdue to their worldwide use in different applications such as dis-tance education, rural telephony, telemedicine, DBS TV, private/corporate networks, and broadband internet. At present, more than500,000 VSATs are in operation worldwide and it is expected thatthis number will be doubled within the next three years [1]. Themain constraint of VSAT systems is their antennas, which areparabolic reflectors and hence they are bulky, create visual pollu-tion, and are expensive. In this context, various efforts have beenmade to replace the reflector dishes with suitably high-gain low-profile planar-antenna arrays, which will be robust, portable, andless expensive. For such arrays, microstrip patch antennas are themost suitable candidates due to their light weight, low profile, lowcost, ease of fabrication, and integration with other electroniccomponents. They are compatible with hybrid microwave inte-grated circuit (HMIC) and monolithic microwave integrated circuit(MMIC) design. Despite all these dimensional advantages, theysuffer from narrow bandwidth and low gain. In the case of the

practical implementation of microstrip patch antennas in VSATsystems, the crucial issues are the gain (on the order of 30 dBi),large bandwidth (up to 20%) to cover both transmit and receivefrequencies [2], high isolation (on the order of 30 dB betweentransmitting and receiving ports), linear orthogonal polarization,and low side-lobe levels (on the order of �30 dB). Presently,research is continuing on the printed microstrip patch antennas inorder to satisfy all these requirements. S. D. Targonski et al. [3]and D. M. Pozar et al. [4] have reported an L/X-band dual-polarized shared-aperture array for space-borne synthetic apertureradar (SAR) applications. The design yields 3.2% bandwidth at theL-band and 15.5% bandwidth at the X-band. Y. Bao et al. [5] havereported a receiving dual-polarized directive antenna array at theX-band. Their design, a 6 6 element planar subarray, yields3.5% bandwidth at 9.7 GHz and an average gain of 25 dBi. Adual-band inflatable 6 6 element L-band SAR antenna array [6]yields 13% bandwidth and 16 dBi gain at 1.25 GHz. F. Rostan etal. [7] reported an 8 8 element dual-polarized single-bandantenna array at 5.3 GHz. The array is an ensemble of aperture-coupled patch antennas (ACPAs) on a 0.5-mm-thick RT/Duroid5880 and yields a 2.8% �15-dB return-loss bandwidth. C.Mangenot et al. [8] reported an S- and X-band dual-polarizedsubarray for SAR. This antenna yields a 16.6% bandwidth at theS-band and 4.2% bandwidth at the X-band. So far, all the previousdesigns are for either single-frequency operations or dual-fre-quency operations for a single narrowband system. Very recently,we designed a compact and broadband (20% BW at the S- andC-bands) shared-aperture dual-band dual-polarized aperture-cou-pled patch-antenna (ACPA) panel [9], which will cater not only fordual-system services (both Intelsat and ST1 services) but also forportable VSAT terminals.

Figure 1 shows such a VSAT system, where the antennaremains under the suitcase lid and is able to maintain the alignmentwith the satellite by adjusting the lid angle. In this work, weinclude dual orthogonal 2D periodic PBG patterns on the antennaground plane in order to enhance the antenna performance. PBGinclusion in the antennas enhances the gain and bandwidth,smoothens the radiation patterns, and yields polarization puritiesby suppressing the surface wave in the dielectric slabs [10, 11].

Recently, PBG structures (PBGSs) have been widely used forimproving the performances of different microwave componentsand devices [12–16]. Y. L. Kuo et al. [17] used PBG structures ina dual-band rectangular-patch antenna. A single probe feed at acertain point on the rectangular patch excites dual resonances attwo orthogonal TM01 and TM10 modes. In such a case, wideband

Figure 1 Proposed portable VSAT terminal with panel antenna

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 3, August 5 2005 289