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Bi-directional Power Converter for

Flywheel Energy Storage Systems

A Thesis

Submitted for the Degree of

Master of Sciencein the Faculty of Engineering

By

S R Gurumurthy

Department of Electrical EngineeringIndian Institute of Science

Bangalore - 560 012

India

Jan 2006



i

To

my mother

and

to the memory of

my father



Acknowledgements

I am grateful to Prof. V. Ramanarayanan and Shri. M. R. Srikanthan, Project Manager

(M), RMP for accepting me as a student. I thank them for guiding me in the project and

providing me all the facilities for experimental work. I express my heartfelt thanks to Prof.

V. Ramanarayanan for giving me an insight into power electronics during my course work

as well as for his continuous support and guidance throughout the project work.

I am extremely grateful to Shri. T. K. Bera, Project Director, RMP, for his guidance and

for giving me an opportunity to undertake the postgraduate studies at Indian Institute of

Science, Bangalore. I also thank Shri. S.Sarkar, Project Manager (Process), RMP, and Shri.

H. A. Balasubramanya for their continuous support throughout my project work.

I thank Prof. V. T. Ranganathan and Prof. G. Naranyanan for their advice and suggestions.

I thank Bhabha Atomic Research Centre, Mysore for all the support, facilities and oppor-

tunities provided to me. I owe my gratitude to IISc administration for providing excellent

hostel and mess facilities during my stay in the Institute campus. I am grateful to Shri.

D. M. Channe Gowda and his team in the Electrical Engineering Department office for the

smooth conduct of administrative activities.

I want to specially thank my colleagues in the power electronics laboratory Venugopal,Kaushik, Kannan, Lakshmi, Debmalya Banerjee, Amit, Mirzaei, Kamalesh, Chandrashekhar,

Milind and Vishal for their help and support during the project work and technical docu-

mentation of the project. My technical discussions with them have helped me learn a lot,

which I am sure, will be useful throughout my life.

I specially thank Kaushik for his help and support during my course work. I owe a lot to

him for his help during the difficult phase of my stay. I will always cherish his friendship. I

cannot forget the helps rendered by Venugopal during experimentation and documentation

ii



Acknowledgements iii

work of the project. I am very much thankful to him.

Discussions with Dr. R. Anbarasu, A. Nandakumar and J. Nataraj has helped me a lot in

conceptualizing the ideas. I am ever indebted to them. I specially thank Satheesh kumar

who was ever ready for helping me in fabrication, testing of hard ware and documentation

of the work. This work would not have been possible without the help and cooperation of

my wife Seetha and son Viveka. I am very much thankful to them.

I would like to attribute all of my success, my achievements to my father and mother. My

father remain an incessant source of inspiration and support all through my life. Finally

I would like to thank the Almighty for all that I have got in my life and for creating the

opportunities for me to pursue the work of my interest.



Abstract

In many power processing applications such as traction, elevators, cranes etc, it is common-place to encounter loss of stored energy. The main reason is that, the power converters are

not capable of returning the stored energy during transients. In applications where frequent

transients are involved, this results in substantial loss of energy. Bi-directional converters in

such applications can lead to higher operating efficiency. In a typical traction application,

stored energy while running can be restored during deceleration. This process saves the

energy and improves the efficiency. Such applications need a bi-directional interfacing con-

verter. The bi-directional converter facilitates the energy flow, to and from the device. Basic

requirements of this interface are, simple structure, ease of control and energy efficiency.Such an application is the target of the development work reported in this thesis.

The aim of this work is to develop a bi-directional power converter/controller to facilitate

the energy storage to and from the storage device. The storage device employed in this

application is a flywheel. The bi-directional power converter (BDC) drives the brushless DC

(BLDC) machine coupled to the flywheel. The total system is a Flywheel Energy Storage

(FES) system. The analysis, design, fabrication and evaluation of such a system is covered

in this thesis.

In the FES system considered, there is a flywheel, coupled to the rotor of an electrical ma-chine. This machine uses the power from a dc bus to accelerate the flywheel (in charging

mode) and keep it running. The same machine discharges the flywheel (in deceleration mode)

to provide power back to the dc bus. The machine acts as a motor during charging and as a

generator during discharging. To store the energy in the flywheel, it is run up to the rated

speed of the motor (using input dc power). Under this condition, the flywheel stores the en-

ergy. When the input power fails, the flywheel continues to run due to its inertia, driving the

generator. If an electrical load is connected across the generator terminals, it draws current

iv



Abstract v

and utilizes the energy stored in the flywheel. In this process, the flywheel discharges its

energy and decelerates. The terminal voltage of the machine exponentially drops during de-

celeration. It is desired that power harvested from the flywheel during the deceleration is at

constant voltage. The machine therefore, is interfaced to the dc bus through a bi-directional

power converter. The bi-directional power converter serves two purposes. It accelerates the

motor while charging and discharges the flywheel while decelerating. Further, the control

in the converter can be exercised to obtain constant dc output voltage for a wider range of

flywheel speed. The amount of power transferred to the load and its duration is a function

of the running speed, overall efficiency and the control strategies adopted.

The primary aim of this thesis is to design and fabricate the bi-directional power con-

verter/controller, identify its operating modes, study the effect of various system parameters

on the performance; and evaluate the system.

To start with, the overall system design is considered. The selection of prime mover and bi-

directional power converter are discussed. Basic design of the BLDC machine is presented.

The design is verified through Finite Element Method (FEM) of analysis and validated

through experimental results. A six-switch voltage source inverter topology is selected as

bi-directional power converter. Operating modes of the power converter are identified as con-

trolled current acceleration (CCA) mode during charging and constant voltage deceleration(CVD) mode during discharging. Transfer function and steady state equations are derived

for both CCA and CVD modes. Controller design for both the modes are proposed. Design

considerations and selection of various circuit elements are explained. Sources of various

losses in the system are discussed in detail. Method of apportioning the losses are given.

The effect of various system parameters on its performance is explained. Design recommen-

dations to move towards lower losses are given.

The controller and Human Machine Interface (HMI) are implemented using a Motorola,

Digital Signal Processor 56F805. All subsystems are fabricated, integrated and tested. Ex-perimental results are presented. Energy transfer, to and from the flywheel is demonstrated.



Contents

Acknowledgements ii

Abstract iv

List of Tables x

List of Figures xi

Nomenclature xiv

1 Introduction 11.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2 Two quadrant BLDC drive . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.2.1 BLDC machine . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.2.1.1 Source of Loss and its reduction . . . . . . . . . . . . . . . . 3

1.2.1.2 Machine Type . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.2.1.3 Construction . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.2.1.4 Sizing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

1.2.2 Bi-directional Power Converter . . . . . . . . . . . . . . . . . . . . . 51.2.2.1 Power Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . 5

1.2.2.2 Sizing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

1.2.3 Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

1.3 Overall system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

1.3.1 Principle of working of the system . . . . . . . . . . . . . . . . . . . . 9

1.4 Scope of the Thesis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

1.5 Organization of the thesis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

vi



Contents vii

2 Selection, Design and Analysis of Brushless DC Machine 13

2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.2 Selection of the type of machine . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.3 Basic design of the BLDC machine . . . . . . . . . . . . . . . . . . . . . . . 14

2.4 Construction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.5 Equation for the induced voltage . . . . . . . . . . . . . . . . . . . . . . . . 17

2.5.1 Induced voltage equation . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.5.2 Computation of phase and line voltages . . . . . . . . . . . . . . . . . 20

2.6 Armature leakage inductance and coil resistance . . . . . . . . . . . . . . . . 20

2.7 Model of the machine . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 222.8 FE Method of Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

2.9 Experimental results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

2.10 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

3 Bi-directional Converter 27

3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

3.2 Power converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

3.3 Transfer function of BDC in CCA mode . . . . . . . . . . . . . . . . . . . . 30

3.3.1 Simplified Equivalent circuit . . . . . . . . . . . . . . . . . . . . . . . 30

3.3.2 Small signal modeling . . . . . . . . . . . . . . . . . . . . . . . . . . 31

3.4 Transfer function of BDC in CVD mode . . . . . . . . . . . . . . . . . . . . 31

3.4.1 Simplified equivalent circuit . . . . . . . . . . . . . . . . . . . . . . . 34

3.4.2 Small signal modeling . . . . . . . . . . . . . . . . . . . . . . . . . . 34

3.5 Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

3.6 Switching between controllers . . . . . . . . . . . . . . . . . . . . . . . . . . 36

3.7 Selection of circuit elements . . . . . . . . . . . . . . . . . . . . . . . . . . . 363.7.1 Series Inductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

3.7.2 DC bus capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

3.7.3 Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

3.7.4 Switching frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

3.8 Effects of system parameters on the performance . . . . . . . . . . . . . . . . 38

3.8.1 Source resistance of the machine . . . . . . . . . . . . . . . . . . . . . 38

3.8.2 Weight of the flywheel . . . . . . . . . . . . . . . . . . . . . . . . . . 40



viii Contents

3.8.3 Diameter of the flywheel . . . . . . . . . . . . . . . . . . . . . . . . . 41

3.9 Effect of Backup time on the system performance . . . . . . . . . . . . . . . 41

3.10 Simulation of the system in CVD mode . . . . . . . . . . . . . . . . . . . . . 42

3.11 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

4 Digital Implementation and Performace Evaluation of the System 45

4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

4.2 Pulse generation in CCA (motor) mode . . . . . . . . . . . . . . . . . . . . . 45

4.3 Pulse generation in CVD (generator) mode . . . . . . . . . . . . . . . . . . . 48

4.4 Software implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

4.4.1 Human Machine Interface (HMI) . . . . . . . . . . . . . . . . . . . . 48

4.4.1.1 Keypad . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

4.4.1.2 Display . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

4.4.2 Timer interrupt service routine . . . . . . . . . . . . . . . . . . . . . 51

4.4.3 Main program . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

4.5 Hardware implementation of the system . . . . . . . . . . . . . . . . . . . . 52

4.5.1 Controller hardware . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

4.5.2 Power converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

4.6 Testing and performance analysis of the system . . . . . . . . . . . . . . . . 55

4.6.1 Apportioning of various losses . . . . . . . . . . . . . . . . . . . . . . 55

4.6.1.1 No load test . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

4.6.1.2 Retardation test with flywheel . . . . . . . . . . . . . . . . . 55

4.6.1.3 Copper losses in the armature winding . . . . . . . . . . . . 57

4.6.1.4 Switching and conduction losses in the converter . . . . . . 58

4.6.2 Power backup time test . . . . . . . . . . . . . . . . . . . . . . . . . . 59

4.6.3 Source resistance effect . . . . . . . . . . . . . . . . . . . . . . . . . . 60

4.6.4 Current dependent eddy current loss in the core . . . . . . . . . . . . 63

4.6.5 Comparison of various loss components . . . . . . . . . . . . . . . . . 64

4.6.6 Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

4.6.7 Harvestable Energy . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

4.6.8 Current waveforms at various speeds . . . . . . . . . . . . . . . . . . 66

4.7 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68



Contents ix

5 Conclusions 69

5.1 The present work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

5.2 Guidelines emerging from the work . . . . . . . . . . . . . . . . . . . . . . . 71

5.3 Spin off technology from the present system . . . . . . . . . . . . . . . . . . 71

5.4 Applications of the system . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

A Specifications of IGBT and Capacitor 72

A.1 IGBT Module . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

A.2 Capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

B Specifications of Digital Signal Processor DSP56F805 73B.1 Digital Signal Processing Core . . . . . . . . . . . . . . . . . . . . . . . . . . 73

B.2 Memory . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

B.3 Peripheral Circuits for DSP56F805 . . . . . . . . . . . . . . . . . . . . . . . 74

C Block Diagram of Controller 76

D Specifications of Hall effect position Sensor 77

E Publication 78

F Photographs of the test setup 79

G Further improvements in the system 83

G.1 Method used in apportioning various losses . . . . . . . . . . . . . . . . . . . 83

G.2 Relation between the speed and the loss: . . . . . . . . . . . . . . . . . . . . 83

G.3 Interpretation of the equation : . . . . . . . . . . . . . . . . . . . . . . . . . 85

G.4 Loss reduction techniques: . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

G.5 conclusions: . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

References 90



List of Tables

2.1 Comparison of results at 10000 RPM . . . . . . . . . . . . . . . . . . . . . . 24

4.1 Retardation test data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

4.2 Max voltage gain test data . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

4.3 Comparison of various losses . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

4.4 Test results at various load conditions . . . . . . . . . . . . . . . . . . . . . . 65

G.1 Total loss at various mass as a function of rotor speed . . . . . . . . . . . . . 84

G.2 Various losses with out flywheel . . . . . . . . . . . . . . . . . . . . . . . . . 86

G.3 Various losses in watts with a flywheel of 11 Kg . . . . . . . . . . . . . . . . 86



G.6 Loss reduction techniques . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

G.7 Comparison of various losses . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

G.8 Comparison of various systems . . . . . . . . . . . . . . . . . . . . . . . . . . 89

x



List of Figures

1.1 Block schematic of two quadrant BLDC drive . . . . . . . . . . . . . . . . . 2

1.2 Bi-directional Power Converter circuit . . . . . . . . . . . . . . . . . . . . . . 6

1.3 Current vs Time characteristics . . . . . . . . . . . . . . . . . . . . . . . . . 6

1.4 Voltage vs Time characteristics . . . . . . . . . . . . . . . . . . . . . . . . . 8

1.5 Typical flywheel energy storage system . . . . . . . . . . . . . . . . . . . . . 8

1.6 Speed vs Time characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . 9

1.7 Voltage vs Time characteristics . . . . . . . . . . . . . . . . . . . . . . . . . 10

2.1 Cross sectional view of basic PM BLDC machine . . . . . . . . . . . . . . . 14

2.2 Cross sectional view of PM BLDC machine . . . . . . . . . . . . . . . . . . . 16

2.3 Placement of magnets in the interior of rotor . . . . . . . . . . . . . . . . . . 16

2.4 Magnetic field produced in the air gap . . . . . . . . . . . . . . . . . . . . . 17

2.5 Aig gap flux density vs mechanical angle . . . . . . . . . . . . . . . . . . . . 18

2.6 Stator winding connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.7 Induced voltage as a function of time . . . . . . . . . . . . . . . . . . . . . . 19

2.8 Lumped parameter equivalent circuit for the calculation of leakage inductance. 21

2.9 Magnetic equivalent circuit of the armature leakage inductance. . . . . . . . 21

2.10 Equivalent circuit of the machine . . . . . . . . . . . . . . . . . . . . . . . . 22

2.11 FEM generated air gap flux density plot . . . . . . . . . . . . . . . . . . . . 23

2.12 FEM generated induced voltage waveform plot at 10000 RPM . . . . . . . . 24

2.13 Induced voltage vs speed characteristics . . . . . . . . . . . . . . . . . . . . . 24

2.14 Measured induced voltage waveform of the machine at 10,000 RPM . . . . . 25

3.1 Circuit diagram of the bi-directional power converter . . . . . . . . . . . . . 27

3.2 Converter output voltage waveforms in CCA mode . . . . . . . . . . . . . . 28

xi



xii List of Figures

3.3 Current path during different intervals of time in the CCA mode . . . . . . . 29

3.4 Equivalent circuit of converter in CCA mode . . . . . . . . . . . . . . . . . . 30

3.5 Simplified equivalent circuit of converter in CCA mode . . . . . . . . . . . . 30

3.6 Induced voltage waveforms of the machine . . . . . . . . . . . . . . . . . . . 32

3.7 Current path during different intervals of time in the CVD mode . . . . . . . 33

3.8 Equivalent circuit of converter in CVD mode . . . . . . . . . . . . . . . . . . 34

3.9 Block diagram of the controller in CCA mode . . . . . . . . . . . . . . . . . 36

3.10 Block diagram of the controller in CVD mode . . . . . . . . . . . . . . . . . 36

3.11 Schematic representation of various losses in the system . . . . . . . . . . . . 39

3.12 Representation of voltage dependent and current dependent losses . . . . . . 393.13 Effect of source resistance on the converter performance . . . . . . . . . . . . 40

3.14 Energy efficiency vs backup time . . . . . . . . . . . . . . . . . . . . . . . . 41

3.15 Energy harvested vs backup time . . . . . . . . . . . . . . . . . . . . . . . . 42

3.16 Plot of speed vs time in CVD mode . . . . . . . . . . . . . . . . . . . . . . . 43

3.17 Plot of output voltage vs time in CVD mode . . . . . . . . . . . . . . . . . . 43

4.1 Placement of the position sensors and their output signals . . . . . . . . . . 46

4.2 Flow chart of the position sensor interrupt . . . . . . . . . . . . . . . . . . . 47

4.3 Sensorwise IGBT gate trigger logic diagram . . . . . . . . . . . . . . . . . . 47

4.4 Power and control schematic in CCA mode . . . . . . . . . . . . . . . . . . . 48

4.5 Power and control schematic in CVD mode . . . . . . . . . . . . . . . . . . . 49

4.6 Flow chart of the key pad interface routine . . . . . . . . . . . . . . . . . . . 50

4.7 Flow chart of the display routine . . . . . . . . . . . . . . . . . . . . . . . . 51

4.8 Flow chart of the timer interrupt service routine . . . . . . . . . . . . . . . . 52

4.9 Flow chart of the main program . . . . . . . . . . . . . . . . . . . . . . . . . 53

4.10 Block diagram of the test setup . . . . . . . . . . . . . . . . . . . . . . . . . 544.11 Various losses and their sources in the system . . . . . . . . . . . . . . . . . 55

4.12 Iron losses in the machine in no load test . . . . . . . . . . . . . . . . . . . . 56

4.13 Speed vs time in the retardation test with flywheel . . . . . . . . . . . . . . 57

4.14 Losses in the machine with and with out flywheel . . . . . . . . . . . . . . . 58

4.15 Equivalent circuit of boost converter for copper loss calculation . . . . . . . . 58

4.16 dc bus voltage control with P o = 818 watts . . . . . . . . . . . . . . . . . . . 60

4.17 dc bus voltage control with P o = 450 watts . . . . . . . . . . . . . . . . . . . 61



List of Figures xiii

4.18 Voltage gain as a function of duty cycle with α = 0.024 . . . . . . . . . . . . 62

4.19 Source resistance as a function of duty cycle . . . . . . . . . . . . . . . . . . 63

4.20 Overall efficiency as a function of back up time . . . . . . . . . . . . . . . . 65

4.21 Energy harvested as a function of back up time . . . . . . . . . . . . . . . . 66

4.22 Armature current waveform in CCA mode at speed = 2200 RPM . . . . . . 66

4.23 Armature current waveform in CCA mode at speed = 9640 RPM) . . . . . . 67

4.24 Armature current waveform in CVD mode . . . . . . . . . . . . . . . . . . . 67

C.1 Block Schematic of DSP Board . . . . . . . . . . . . . . . . . . . . . . . . . 76

F.1 Bi-directional Power Converter . . . . . . . . . . . . . . . . . . . . . . . . . 80F.2 Brushless DC machine coupled to Flywheel . . . . . . . . . . . . . . . . . . . 81

F.3 Test set up of Flywheel Energy Storage System . . . . . . . . . . . . . . . . 82

G.1 Various losses and their sources in the system . . . . . . . . . . . . . . . . . 84

G.2 Losses in the machine with and with out flywheel . . . . . . . . . . . . . . . 85



Nomenclature

Symbols : Definitions

α : Ratio Rs

RL

B : Airgap flux density in Tesla

C : Filter capacitor connected across DC bus

D : Duty cycle of switching waveform

Dmax : Steady state maximum duty cycle

Dm : Diameter of the machine

Dmin : Steady state minimum duty cycle

ˆd : Perturbation in duty cycleeb : Instantaneous back EMF induced in armature coils per phase in Volt

E b : Peak value of back EMF induced in steady the state

η : Efficiency

f s : Switching frequency in Hz

H c : Coercivity of the permanent magnet

I a : Armature current in steady the state

ia : Instantaneous armature current

ˆia : Perturbation in armature currentI aRMS : RMS value of armature current

I ∗ : DC current reference

I dc : DC bus current in Ampere

I fb : DC bus current feedback

I Lmax : Peak value of armature current in Amp.

J : Rotational moment of inertia of the flywheel

K i : Current feedback constant

xiv



Nomenclature xv

K v : Voltage feedback constant

La : Armature leakage inductance per phaseL : Active length of the machine

Lch : External series Inductance per phase

lg : Length of air gap in meters

Lt : Total circuit Inductance (Lt = La + Ls)

N : Number of conductors per slot

ω : Rotor mechanical speed in rad/sec

P ei : Current dependent eddy current loss in the core in watts

P eφ : Flux dependent eddy current loss in the core in wattsP h : Hysteresis loss in the core in watts

Φa : Flux produced by armature current

Φf : Flux produced by field poles

P m : Mechanical losses in watts

p : Number of poles

Ra : Winding resistance in ohms

RL : Load resistor connected across DC bus

Rs : Equivalent source resistance as seen by boost converterR : Stator bore diameter

S : Number of slots per phase per pole

S r : Rotor speed in RPM

T bu : Backup time in seconds

tf : Fall time of the IGBT

T g : Generated torque in the machine

T l : Load torque

T PT : Pole transition timetr : Rise time of the IGBT

T s : Period of switching waveform in sec

V B : Instantaneous voltage applied across B-phase armature winding

V dc : DC bus voltage in Volts

V dc : Perturbation in DC bus Voltage

V ∗ : DC voltage reference



xvi Nomenclature

V fb : DC bus voltage feed back

V line : RMS value of line voltage in VoltV ph : RMS value of phase voltage in Volt

V R : Instantaneous voltage applied across R-phase armature winding

V Y : Instantaneous voltage applied across Y-phase armature winding

w : Width of magnet in meters

Abbreviations

BDC : Bi-directional Converter

BLDC : Brushless DC

CCA : Controlled Current AccelerationCCM : Continuous Current Mode

CVD : Constant Voltage Deceleration

FES : Flywheel Energy Storage

PMSM : Permanent Magnet Synchronous Machine



Chapter 1

Introduction

1.1 Introduction

In many drives applications such as traction, elevators, cranes etc, it is commonplace to

encounter loss of stored energy. The main reason is that, the power converters are not capable

of returning the stored energy during transients. In application where frequent transientsare involved, this results in substantial loss of energy. Bi-directional converters in such

applications can lead to higher operating efficiency. In a typical traction application, stored

energy while running can be restored during deceleration. This process saves the energy and

improves the efficiency. Such applications need a bi-directional interfacing converter. The bi-

directional converter facilitates the energy flow, to and from the device. Basic requirements

of this bi-directional converter are, simple structure, ease of control and energy efficiency.

An attempt is made to develop one such interfacing converter.

The topic addressed in this thesis is a bi-directional power converter driving a two quadrantbrushless DC (BLDC) machine. Such a system is simple in structure; easy to control and well

suited for several low cost applications such as electric traction, material handling equipments

etc. The storage device employed in this application is a flywheel. The bi-directional power

converter (BDC) drives the brushless DC (BLDC) machine coupled to the flywheel. The

total system is a Flywheel Energy Storage (FES) system. This thesis covers the analysis,

design, fabrication and evaluation of such a system applied to FES. Key features of the

system are demonstrated and the contributions are outlined.

1



2 Chapter 1. Introduction

Ifb

BDC

Vfb

Vdc

Controller

Controller

(a)

(b)

ω

T

ω

T

Motor

Generator

BDC

vdc

−

+

−

+

Figure 1.1: Block schematic of two quadrant BLDC drive



1.2. Two quadrant BLDC drive 3

1.2 Two quadrant BLDC drive

A typical two quadrant drive is shown fig 1.1. The power flow direction in motor mode is

shown in fig 1.1(a). This mode of operation is identified as controlled current acceleration

(CCA) mode. Once acceleration is complete, the motor will be running at constant speed

drawing the loss power from the dc source. The power flow direction in generator mode is

shown in fig 1.1(b). This mode of operation is identified as constant voltage deceleration

(CVD) mode. In this mode, the energy in the flywheel is transfered to the dc bus at constant

voltage.

This system can be divided into following subsystems.

• BLDC machine.

• Bi-directional power converter (BDC).

• Controller.

1.2.1 BLDC machine

In the FES system, the electrical machine accelerates the flywheel during charging and

discharge the flywheel during deceleration. The following are the key features of the machine.

• Capability to be operated at speeds of the order of 10,000 RPM or above.

• Capability to be operated as a motor or a generator during charging or discharging of

the flywheel, respectively.

• High efficiency.

1.2.1.1 Source of Loss and its reduction

In an energy storage/retrieval application, the energy loss is a major concern. The losses in

the machine are listed as,

• Iron losses in the armature magnetic material.

• Copper losses in the armature conductors.

• Friction and windage losses in the rotational system.




The iron loss is a function of property of the core material, its quantity, operating frequency

and flux density. In general, two pole machines are preferred for the high speed motoring

operation since the value of the operating frequency is lower. This in turn, results in lower

iron loss. Iron loss in the machine is related to the specific magnetic loading in the air-gap

(B) [5]. Copper loss in the machine is related to specific electric loading in the armature

periphery (J A/m) [5]. Low electric and magnetic loading will make the losses low; the

machine will however, be bigger.

1.2.1.2 Machine Type

It is preferred to have a commutator-less machine, as it eliminates frequent maintenance

problems, reduce the EMI and increase the efficiency. Permanent Magnet Synchronous

Machines (PMSM) or Brushless DC (BLDC) machines can be used because they can be

operated as generator or motor conveniently. PM machines use magnets to produce air-gap

magnetic flux instead of field coils. This configuration eliminates rotor copper loss as well

as the need for maintenance of the field exciting circuit. This has been made possible by the

easy availability of high performance permanent magnets with high coercivity and residual

magnetism, such as Samarium cobalt and Neodyum-Iron-Boron (NdFeB) magnets. It can

be shown that a permanent magnet excitation circuit does not use any copper and saves 30

percent of the iron used in comparison with an electromagnetic excitation.

1.2.1.3 Construction

The permanent magnet machines consist of a three phase stator windings similar to that of

induction machine and a rotor with permanent magnets. The machine characteristics depend

on the magnets used and the way they are located in the rotor. The permanent magnets

(PM) are either mounted on the surface or buried in the interior of the rotor. Accordingly

they are called as Surface Mounted PM machines or Interior PM machines. PM machines

can be broadly classified into two categories [6].

• Sinusoidal waveform machines: These machines have a uniformly rotating stator field

as in induction machines. The stator winding is sinusoidally distributed or the mag-

nets are shaped to get sinusoidal induced voltage waveforms. Hence sinusoidal stator

currents are needed to produce ripple free torque.




• Trapezoidal waveform machines: These are known as brushless DC or electronically

commutated DC machines. Induced voltage is trapezoidal in its shape. The con-

centrated windings on the stator are the reason for the trapezoidal-shaped back emf

waveform. The armature current is switched in discrete steps. The control of such mo-

tors is very simple. Only six discrete rotor positions per electrical revolution are needed

in a three-phase machine to synchronize the phase currents with phase back emfs for

effective torque production. A set of hall effect sensors are mounted on the armature

to provide rotor position information. This eliminates the need for high-resolution

encoder or position sensor required for the PMSM [6]. The back emf waveforms are

fixed with respect to rotor position. Square wave phase currents are supplied such that

they are synchronized with the back emf peak of the respective phase. The controller

achieves this using the rotor position feedback information. The motor basically oper-

ates like a DC motor, with such a controller configuration, from a control point of view;

hence the motor is designated as a brushless DC motor. There are several advantages

of using PM for providing excitation in AC machines. Permanent magnets provide loss

free excitation in a compact way without complications of connections to the external

stationary electric circuits. These types of machines become very attractive option due

to their high torque densities, high power density, excellent performance and with lowrotor losses [6].

1.2.1.4 Sizing

The machine has to deliver rated power only for a short time - during acceleration while

charging and during break time while discharging. In the motoring mode the machine is

required to provide only the losses under steady state. In other words, the machine is used

for intermittent operation only. A low loss short time rated PM machine is the best choice.

1.2.2 Bi-directional Power Converter

1.2.2.1 Power Circuit

The bi-directional power converter is of the voltage source bridge topology. Such a converter

can transfer power from a constant voltage dc source to an ac load, or from an ac source

to a dc voltage bus. The power converter is made up of solid-state devices. This controls

the flow of bulk power from source to motor terminals or from generator terminals to the




dc power bus. The schematic of the converter is shown in the fig 1.2. In drive mode, the

converter operates as a standard voltage source inverter with six switches. In the generating

mode, the same converter operates as 3-phase boost converter pumping energy from each of

the phases to the dc bus.

ea

eb

ec

La

La

La

LoadC

+dc bus

−dc bus

T1 T3 T5

T2T6T4

Figure 1.2: Bi-directional Power Converter circuit

Current

Time

Load current

Armature current

Figure 1.3: Current vs Time characteristics




1.2.2.2 Sizing

In drive mode (CCA), the power converter supplies the required power to the motor, to

accelerate the flywheel and to keep it running. With the closed loop current controller, the

current during the acceleration is controlled. In the generating (CVD) mode, bi-directional

power converter supplies the rated power at constant voltage to the load connected across

the dc bus. The characteristics of the bi-directional converter in CVD mode is shown in the

fig 1.3. It may be noticed that, the machine (armature) current increases (to deliver constant

power) as the flywheel is slowing down. Accordingly the current rating of the converter is

decided based on the output voltage and power requirement, at lowest operating speed of

the flywheel.

1.2.3 Controller

The controller block is shown in the fig 1.1. This is based on Digital Signal Processor. The

controller senses the input commands and feedback signals. These signals are processed to

generate the switching pulses for the power converter semiconductor switches.

The motor is supplied with controlled current at the desired frequency through the bridge

circuit. This ac current decides the accelerating torque of the motor. This is the operating

mode during charging of the flywheel. This may be termed as the current (ac) controlled

acceleration (CCA) mode.

In the discharge duration, the same converter transfers the energy from the flywheel through

the machine to the dc bus. In this mode, the machine voltage varies widely since the

flywheel is decelerating and loosing the energy. However, the converter is capable of operating

over a wide range of machine voltage, pumping power to dc bus at constant voltage. This

mode is called the constant voltage (dc) deceleration (CVD) mode. The current versustime characteristics of the boost converter is shown in the fig 1.3. The voltage versus time

characteristics of the boost converter is shown in fig 1.4. It may be seen from the figures 1.3

and 1.4, that with the boost converter it is possible to draw power at constant voltage right

down to 50 percent of the speed. This corresponds to a harvest of 75 percent of the stored

energy.




Tbu

Energy harvestingtime

Input Voltage

Output voltage

Voltage

Time

Figure 1.4: Voltage vs Time characteristics

Vfb

Bearing (B)Ifb

DSP 56F805

Motorola make

Controller Converter(D)

(C)

(E)

In Out

(A)

SignalsPositionRotor

dc power

Two−Quadrant

drive motor

Flywheel

Bi−directional

Figure 1.5: Typical flywheel energy storage system

1.3 Overall system

FES system has the following subsystems.

• Flywheel that stores the energy (A).

• Bearings that support the rotor and the flywheel (B).

• Two quadrant drive motor (C).

• Bi-directional power converter (D).

• DSP based electronic controller (E).



1.3. Overall system 9

1.3.1 Principle of working of the system

The flywheel is coupled to the rotor of the electrical machine. This machine is used to accel-

erate the flywheel (charging mode). The same machine discharges the flywheel (deceleration

mode) to provide power to the dc bus. The machine acts as a motor during charging and as

a generator during discharging. The machine is therefore driven by a bi-directional power

converter as shown in the fig 1.5. To store the energy in the flywheel it is run up to the

rated speed of the motor. Under this condition, the flywheel stores the energy. When the

input power fails, the flywheel continues to run due to its inertia driving the generator. If an electrical load is connected across the generator terminals, it draws current and utilizes

this energy. In this process, the flywheel discharges its energy and decelerates. Speed-time

and voltage-time characteristics are shown in the fig 1.6 and fig 1.7.

0

2000

4000

6000

8000

10000

0 20 40 60 80 100

S p e e d i n R P

M

Time in seconds

Figure 1.6: Speed vs Time characteristics

The amount of power transferred to the load and its duration is a function of the running

speed, overall efficiency and the control strategies adopted. From fig 1.7 it is seen that

the terminal voltage of the machine exponentially drops during deceleration. It is desired

that power harvested from the flywheel during the deceleration is at constant voltage. The




0

50

100

150

200

250

300

0 20 40 60 80 100

G e n e r a t o r v o l t a g e i n v o l t s

Time in seconds

Figure 1.7: Voltage vs Time characteristics

bi-directional power converter serves two purposes. It accelerate the motor while charging

and discharges the flywheel during deceleration. Further, the control in the converter can

be exercised to obtain constant dc output voltage for a wider range of flywheel speed. More

about the control methodologies are explained later in the chapter 3.

1.4 Scope of the Thesis

This thesis covers the Analysis, Design, Fabrication and Evaluation of a two quadrant bldc

drive. This system consists of a bidirectional converter/controller, a brushless dc machine

and a flywheel. The scope of the work is partitioned as follows,

• Overall system design

• Basic design and performance analysis of the brushless dc machine.

• Dynamic modeling and development of control strategy for the power converter to

operate as bldc drive in motoring mode (Current controlled acceleration) & as boost

converter in generating (constant voltage deceleration) mode.

• Verification of the design through simulation



1.5. Organization of the thesis 11

• Development of a suitable control platform (DSP) and the associated software to realize

the control strategy.

• Development of suitable Human Machine Interface (HMI).

• Performance evaluation of the full system.

1.5 Organization of the thesis

The thesis is organized as follows:

Chapter 2: Chapter 2 covers the design and performance of the bldc machine. Basic de-

sign of a permanent magnet bldc machine under idealized condition is presented. The bldc

machine constructed employs standard sizes of rotor/stator frame/stampings etc, that are

readily available. The Finite Element Method of analysis is carried out to verify the basic

design. The analysis results are validated through the speed vs emf characteristics of the

machine running as a generator.

Chapter 3: Chapter 3 presents the bi-directional power converter. The operating modes

are identified. The circuit topologies of the converter and defining equations of the drive in

acceleration and deceleration modes are presented. The control strategies under both modes

of operation are also outlined. Design considerations like device rating, critical values of in-

ductor and dc link capacitor, Switching frequency are high lighted. Effects of non-idealities

in the machine parameters and other circuit components on the performance of the BDC

are also discussed. Simulink modeling of the overall system in CVD mode is also presented

in this chapter.

Chapter 4: The central processing unit of the controller employed is a Digital Signal

Processor (DSP). The hardware realization of the power converter and the DSP controlleris presented in this chapter. The software issues covering flow charts, priority allocation

and scheduling of events etc, are presented. Experimental results covering the charging and

discharging mode of the FES system are presented to validate the design.

Chapter 5: Chapter 5 gives contributions and the conclusions made from this work. Fea-

tures of this system and important findings of this thesis are presented. Design guidelines

emerging from this work, the spin off technology from the present system and other appli-

cations of the system are presented.



Chapter 2

Selection, Design and Analysis of

Brushless DC Machine

2.1 Introduction

This chapter covers the selection, basic design, construction, analysis and testing of the

brushless dc machine. Equation for induced voltage in the machine is obtained analytically.

Analytical computation of induced voltages are verified by the FEM analysis and validated

through the experimental results.

2.2 Selection of the type of machine

The prime mover for the flywheel energy storage system is an electrical machine. The

machine accelerates the flywheel to charge the same and discharge the flywheel during the

deceleration.

Following are the key features of the machine.

• Capability to be operated at a speed of the order of 10,000 rpm.

• Capability to be operated as a generator or a motor during charging and discharging

mode respectively.

• High efficiency.

• No brushes.

13



14 Chapter 2. Selection, Design and Analysis of Brushless DC Machine

Permanent Magnets

Radius

L e n g t h

Rotor

Armature conductors

Figure 2.1: Cross sectional view of basic PM BLDC machine

Permanent magnet synchronous machine (PMSM) or Brushless DC (BLDC) machine can be

used for the above features. This is due their capability to be operated as a generator or a

motor conveniently. BLDC machine is selected for this application on account of simplicity

of control.

2.3 Basic design of the BLDC machine

With reference to fig 2.1, basic design of the machine has been done by using idealized

permanent magnet BLDC machine equations [1]. Design specifications of the machine are

as follows:

dc bus voltage (V dc) = 300V.

Out put power (P o) = 1000W.Specific electric loading (J) = 12000A/m.

Specific magnetic loading (B) = 0.2 T.

Efficiency (η) = 0.75.

Rotor speed (S r) = 10,000 RPM.

It may be noted that the flux density taken is 0.2 T. The operating frequency will be 333

Hz at 10,000 RPM. To reduce the iron losses in the magnetic material of the machine, the

air gap flux density is kept substantially low. Idealized design equation relating the machine



2.4. Construction 15

output power to the mechanical dimensions is given by [1] [2],

P o = π2

BJ Dm

2

LS rη60

(2.1)

Substituting the design parameters in the equation 2.1, we get,

• Dimensions:

Dm2L = 338 e -6 m3; If L and Dm are same in magnitude, then the we have,

L = 70 mm, Dm = 70 mm.

Nearest dimensions of the standard available frame is Dm = 86 mm and L = 75 mm.

• Coil current:

I aRMS =

2

3P o

V dcη = 3.64A. (2.2)

• No of slots and poles:

No of slots and poles are selected as 36 and 4 respectively.

• Turns per coil:

Voltage per turn = Φtotal

T PT , Where T PT is the pole transition time.

e1 =2BL

πDm

4 (4S r)

60 = 1.35V. (2.3)

Total no of turns = V dc

e1 = 222 Turns.

Turns per coil = Total no of turns / (2 * No of slots per phase/2) = 18.5 Turns.

A suitable magnet of dimensions 12 mm x 18 mm is selected for the permanent magnets.

With the above design data, the machine is fabricated. The design validation is done through

back emf test as well as the FEM analysis of the magnetic circuit of the machine.

2.4 Construction

The winding distribution is uniform throughout the periphery of the stator as shown in the

fig 2.2. Permanent magnets are buried in the interior of the rotor as shown in the fig 2.3.




α( )t

L

28

29

10

11

20 19

35 1363

B

R

Y

γ

R2

B

Y

Figure 2.2: Cross sectional view of PM BLDC machine

These magnets are used for producing air-gap magnetic flux. The magnetic field produced

in the air gap is shown in the fig 2.4. A set of hall sensors are mounted on the stator,

facing the rotor magnets. These sensors are placed 120o (Electrical) apart to give the rotor

position information. The machines with this type of construction are known as brushless

dc or electronically commutated dc machines.

Rotor shaft

Magnets

0o

90o

o

270o

180

Figure 2.3: Placement of magnets in the interior of rotor



2.5. Equation for the induced voltage 17

Figure 2.4: Magnetic field produced in the air gap

2.5 Equation for the induced voltage

The arrangement of the magnets in the rotor and their magnetization directions are shown

in the fig 2.4. The flux distribution in the air gap is rectangular in shape. This is shown in

the fig 2.5. Following assumptions are made in order to derive the expression for the induced

voltage in terms of electrical, magnetic and mechanical quantities.

• No saturation in the active magnetic material circuit of the core.

• No eddy current and hysteresis losses.

• Air gap is uniform

• Magnets have infinite resistivity.

• Permeability of the magnet is equal to that of air.




Flux density

0 90o 180o 270oθm

B

−B

Figure 2.5: Aig gap flux density vs mechanical angle

Vphase

Vline

R Y B

Figure 2.6: Stator winding connection

2.5.1 Induced voltage equation

The flux linked by each coil as a function of mechanical angle is shown in the fig 2.7(a). The

induced voltage across a conductor moving in the magnetic field is given by,

e = BLv (2.4)

Where ‘B’ is the flux density in Tesla, ‘L’ is the length of the conductors in meters and ‘v’

is the relative velocity of the conductor w.r.t the flux.

Using this relation one can find the total emf induced in one coil (there are N conductors in

one coil). It is given by,

eb = 2BLNR ω (2.5)



2.5. Equation for the induced voltage 19

φa

(a)

(b)

(c)

(d)

(e) V

3V

e2

e3

V

−V

θ

θm

θm

θm

m

0o 60o40o20o 80o 100o 120o 140o 160o 180o

60o

e1

θm

Flux

−3V

−V

Figure 2.7: Induced voltage as a function of time

The shape of the induced voltage is rectangular as shown in fig 2.7(b). Induced voltages in

the conductors in all other slots are also identical in their shapes and magnitudes. But they

are shifted in phase depending on their position in the stator slot. As all the coils are in

series, the total induced voltage across the coil is the algebraic sum of the induced voltage

of each coil as shown in fig 2.7(e). Peak of the line voltage is given by the equation 2.7.

Total induced emf per phase per pole is of trapezoidal wave shape. The peak value is

given by,

V = 6BLNR ω (2.6)

For a ‘p’ pole machine, this value is multiplied by p

2. The machine under consideration is

a four pole machine; the phase voltage is two times V. The line voltage as referred in the

fig 2.6, is given by equation 2.7.

V line = 2E b = 24B pLNR ω (2.7)




The plot of the induced voltage waveforms is given in the fig 2.7.

2.5.2 Computation of phase and line voltages

Machine data:

ω = 1047.2rad/s (Corresponding to a speed of 10,000 rpm)

B = 0.18 T

L = 0.075 m

N = 19

R = 0.043 m

Air gap length = 1.1 mm.

Substituting the above data of the machine in the equations 2.7, we get,

V ph = 138.6 V

V line = 277.2V

The line to line induced voltage can also be expressed as a function of speed in RPM. This is

done by substituting the actual dimensions in the above equation. If S r is the motor speed,

the equation 2.7 becomes,

V line = 2E b = 0.0277S r (2.8)

2.6 Armature leakage inductance and coil resistance

Following assumptions are made to find an expression for leakage inductance.

• There are no slots in the stator and the air gap is uniform throughout the periphery.

• The flux produced by one armature coil is not linked by the other armature coil.

• The relative permeability of magnets used is equal to 1.0.

Inductance of the coil is given by,

La = N 2

R (2.9)

Where ‘N’ is the no of turns and the ‘R’ is the reluctance of the magnetic path. The path

of the flux produced by an armature conductor and the equivalent magnetic circuit is shown

in fig 2.8 and fig 2.9 respectively. It may be noted that, the ‘R’ is the series combination of



2.6. Armature leakage inductance and coil resistance 21

w

Air gaps length

MagnetPermanant

Armature coil l

l

g

g

Rotor core

Stator core

Figure 2.8: Lumped parameter equivalent circuit for the calculation of leakage inductance.

Rm

R

aNI

aφ

g1

Rg2

Figure 2.9: Magnetic equivalent circuit of the armature leakage inductance.

Rg1 (reluctance of air gap1), Rm (reluctance of magnet) and Rg2 (reluctance of air gap2).

Therefore,

R = Rg + Rm + Rg (2.10)

R =

4πDmLµo

(lg + w + lg) (2.11)




b2E

Vline

+

_

(2.096mH) (0.752 Ω)

1.94mH 0.776 Ω

Figure 2.10: Equivalent circuit of the machine

Using this relation we get,

La = 2

3N 2πDm

4 Lµo

(2lg + w)

= 0.97mH (2.12)

Where lg (= 1.1 mm) is the air gap length and w (= 12 mm) is the width of the magnet.

The winding resistance is obtained from the conductors cross section (ac = 0.519mm2), mean

length of a turn (L = 0.30m), number of turns (T = 19) and resistivity (ρ = 0.0177Ωm/mm2)

of the copper as given below:

Ra = 2 ρLT

ac = 0.388Ω. (2.13)

The measured values are as follows:

La = 1.048mH.

Ra = 0.376Ω.

The equivalent circuit of the machine is shown in fig 2.10.

2.7 Model of the machine

The model of the machine is as shown in the fig 2.10. The values shown inside the parenthesisare the measured values of inductance and resistance.

2.8 FE Method of Analysis

The machine is analyzed using the FEM software package ‘Magnet’. Mechanical model of

the machine (shape and actual dimensions) and the properties of the materials used in the



2.9. Experimental results 23

construction of the machine are fed and the analysis is carried out. Following are the machine

parameters given as input.

Figure 2.11: FEM generated air gap flux density plot

Stamping material : Rote alloy

No of stator slots: 36Core length : 75 mm

Stator inside dia : 86 mm

Turns per coil : 19

Hc in kA/m : 723

Radial air gap length : 1.1 mm

Plots of flux density vs mechanical angle and induced voltage vs time are obtained and given

in the fig 2.11 and fig 2.12.

2.9 Experimental results

A flywheel is coupled to the machine. Machine was run up to a speed of 10,000 rpm and

dc voltage applied was 315 volts. Then the supply has been cut off and the machine was

allowed to decelerate freely. The motor speed and the corresponding induced voltages are

recorded. The induced voltage versus the generator speed is plotted as shown in the fig 2.13.

The induced voltage waveform recorded is shown in fig 2.14.




Figure 2.12: FEM generated induced voltage waveform plot at 10000 RPM

0

50

100

150

200

250

300

350

400

0 2000 4000 6000 8000 10000

I n d u c e d v o l t a g e i n v o l t s

Speed in RMP

Figure 2.13: Induced voltage vs speed characteristics

The value of induced emf was computed using analytical and FE methods. The induced

emf is measured by experimental method also. The results are shown in the table 2.1.

Method Induced Voltage Wave shape Speed

Analytical Method 138.6 Trapezoidal 10,000rpm

FE Method 135 Trapezoidal 10,000 rpm

Experimental 120 Trapezoidal 10,000 rpm

Table 2.1: Comparison of results at 10000 RPM



2.10. Conclusions 25

Figure 2.14: Measured induced voltage waveform of the machine at 10,000 RPM

(Time: 1ms/div, Voltage: 50V/div)

2.10 Conclusions

In this chapter the idealized BLDC machine performance equations were given. An ideal-

ized basic design was carried out to get the key geometrical measures of the machine. The

same was modified to take into account the standardized frame sizes available. The ma-chine was fabricated and the design was validated through back emf measurement and the

equivalent circuit measurement. This equivalent circuit is used in the next chapter to design

the bi-directional power converter to drive the machine. A suitable digital controller for the

converter also presented in the chapter-4.



Chapter 3

Bi-directional Converter

3.1 Introduction

The bi-directional power converter (BDC) facilitates running up the flywheel in the presence

of power supply and recovering the stored energy from flywheel when power fails. This

chapter covers the BDC. The operating modes are identified; control strategies are developed.

The system design and the performance evaluation through simulation is presented in this

chapter.

3.2 Power converter

Power converter circuit diagram and the output waveforms are shown in the fig 3.1 and

fig 3.2 respectively.

VR

Ra

La

VY

VB

Ra Ra

La

La

eb

eb

eb

/2dc

V /2dc 1

4

3

6

5

2

Load

+dc bus

−dc bus

−

+

−

+

Figure 3.1: Circuit diagram of the bi-directional power converter

27



28 Chapter 3. Bi-directional Converter

VR

VY

VB

36003000240018001200600

−Vdc

2

dcV

2

00 t

t

t

Figure 3.2: Converter output voltage waveforms in CCA mode

The voltage is applied across the windings by turning ‘ON’ the appropriate combination of switches in the converter. Two switches will be ‘ON’ at any time; each device conducts for a

duration of 120o. Commutation will take place every 60o. One can divide the whole cycle into

six intervals of 60o each. The equivalent circuit for the intervals 0−60o, 60o−120o, 180o−240o

and 240o − 300o are shown in the fig 3.3. The path of the current is shown in bold line.

From the figure it can be seen that, the current from dc bus flows through two switches and

two armature coils at any instant. The equivalent circuit is as shown in fig 3.4.

The dynamic equations of the system can be written as,

2La

diadt

= V dcu(t) − 2iaRa − 2eb (3.1)

J dω

dt = T g − Bω − T l (3.2)

Where u(t) = 1 during +ve half cycle and u(t) = -1 during - ve half cycle.



3.2. Power converter 29

VR

Ra

La

VY

VB

Ra

Ra

La

La

VR

Ra

La

VY VB

Ra

Ra

La

La

VR

Ra

La

VY VB

Ra

Ra

La

La

VR

Ra

La

VY

VB

Ra

Ra

La

La

eb

eb

eb

eb

eb

eb

eb

60000 −

1200

600 −

3000

2400−

2400

1800

Vdc /2

Vdc /2

Vdc /2

Vdc /2

Vdc /2

Vdc /2

Vdc /2

Vdc /2

−

+

eb

Ia

Ia

Ia

Ia

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

+dc bus

−dc bus

+dc bus

−dc bus

+dc bus

+dc bus

−dc bus

−dc bus

Interval1

Interval2

Interval3

Interval4

−

eb

eb

+e

b

eb

−

+

−

+

−

+

−

+

−

+

−

+

−

+

−

+

Figure 3.3: Current path during different intervals of time in the CCA mode




Vdc

2

Vdc

2

VR

Ra

La

VY

Ra

La

RL

eb

eb

+

−

6

31

4

+dc bus

− dc bus

aI

−

+

−

+

Figure 3.4: Equivalent circuit of converter in CCA mode

3.3 Transfer function of BDC in CCA mode

3.3.1 Simplified Equivalent circuit

During the charging of the flywheel, the armature current of motor is controlled. The

armature current of the motor can be controlled by adjusting the duty cycle of the applied

voltage. Fig 3.4 can be re-written as shown in fig 3.5

2e b

2Ra2La

Vdc

OFF

ON

Figure 3.5: Simplified equivalent circuit of converter in CCA mode

The dynamic equations are given by, (For period DT s; Switch is ON)

2La

diadt

= V dc − 2iaRa − 2eb (3.3)



3.4. Transfer function of BDC in CVD mode 31

(For period (1 − D)T s; Switch is OFF)

2La diadt

= −(2iaRa + 2eb) (3.4)

3.3.2 Small signal modeling

Averaged equation is given by,

2La

diadt

= V dcD − 2iaRa − 2eb (3.5)

Assume that the current controller response time is much smaller compared to the mechanical

time constant. Consider the perturbations ia = I a + ia, d = D + d and eb = E b (Mechanical

time constant is high; Speed change takes place slowly). The small signal model is as follows,

2La

diadt

= V dc d − 2iaRa (3.6)

Current control transfer function of the system is,

ia(s)

d(s)=

V dc2(Ra + sLa)

(3.7)

3.4 Transfer function of BDC in CVD mode

When the flywheel is decelerating, the machine works as a generator. The bottom switches

of BDC are continuously gated with pulses with switching frequency f s; top switches are

kept OFF. The control objective is to keep the dc bus voltage constant. For this purpose,

it is necessary to obtain the dc bus voltage control transfer function. Induced voltages of

machine is as shown in fig 3.6.

The current path in the converter during the positive half cycle of R-phase (30o − 150o) is

shown in fig 3.7.

From the fig 3.7, it may be observed that,




o30 o900o 150o 210o 270o 330o

t

t

t

VR

VY

VB

b

−Eb

E+

Figure 3.6: Induced voltage waveforms of the machine

• When active switch is ON,

The generator terminals are short circuited through one active ‘ON’ switch, one diode

and two series inductors (2La). The current path is shown in fig 3.7. The machine

current increases. Part of the energy in the flywheel is now transfered to the machine

inductance.

• When active switch is OFF

The generator terminals are connected to the dc bus through two diodes and two

inductors (armature leakage) in series. The current path is shown in fig 3.7. The

inductor current is now pumped into the dc bus and to the load. Eventually the

inductor current drops down.



3.4. Transfer function of BDC in CVD mode 33

VR

Ra

VY

VB

Ra

Ra

VR

Ra

VY VB

Ra

Ra

VR

Ra

La

VY VB

Ra

Ra

La

La

VR

Ra

La

VY

VB

Ra

Ra

La

La

Activeswitch

"OFF"

Activeswitch"ON"

switch

"OFF"

Active

Active

"ON"switch

Ia

Ia

eb

eb

eb

eb

eb

eb

eb

eb

eb

eb

eb

+

−

+

−

Ia

Ia

+

−

La

La

La

La L

a L

a

300

Interval 900−150

0

eb

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

1

4

3

6

5

2

Load

+dc bus

+dc bus

+dc bus

+dc bus

−dc bus

−dc bus

−dc bus

−dc bus

Interval

2C

2C

2C

2C

2C

2C

2C

2C

−+

−900

Ia

Ia

Figure 3.7: Current path during different intervals of time in the CVD mode




RL

Vdc

2La

2Ra

CON

OFF

2eb−

+

Figure 3.8: Equivalent circuit of converter in CVD mode

3.4.1 Simplified equivalent circuitDuring the deceleration of the flywheel, the power is transfered to the dc bus at constant

voltage. This is achieved by adjusting the duty cycle of the active switches. The equivalent

circuit is shown in fig 3.8

The dynamic equations are given by,

• Period: DT s (Switch is ON)

− C dV dc

dt =

V dcRL

(3.8)

2La

diadt

= 2eb − 2iaRa (3.9)

• Period: (1 − D)T s (Switch is OFF)

C dV dc

dt = ia −

V dcRL

(3.10)

2La

dia

dt = 2eb−

V dc−

2iaRa (3.11)

3.4.2 Small signal modeling

Averaged equations are given by,

C dV dc

dt = ia(1 − D) −

V dcRL

(3.12)

2La

diadt

= 2eb − V dc(1 − D) − 2iaRa (3.13)



3.5. Controller 35

We may consider that the speed (and back emf) changes are slow on account of high me-

chanical time constant. With small perturbations ia = I a + ia, d = D + d, eb = E b and

vdc = V dc + V dc to the system, the perturbed equations are,

C d V dc

dt = ia(1 − D) − I a d −

V dcRL

(3.14)

2La

diadt

= V dc d − V dc(1 − D) − 2iaRa (3.15)

Transfer function of the system becomes,

V dc(s)

d(s)

= 2E b

(1 − D)2 ×

(1 −2sLa

RL(1 − D)2)

1 + 2La

RL(1 − D)2s + 2LaC

(1 − D)2s2

(3.16)

The transfer function of the system is of second order. Its corner frequency is dependent

on the duty ratio also. At minimum (0.1) and maximum (0.75) duty ratios, these corner

frequencies (ωn) are at 97 rad/s and 350 rad/s. The system also has a right half plane zero

with a corner frequency 2812 rad/s. The response time of the controller of 0.5 second (which

is 2 percent of the power supply backup time) is considered. With a controller band width

of around 10 rad/s (corresponding to a response time of 0.5 sec), dynamics due to the poles

and zeros of the transfer function can be neglected. Then, it is possible to approximate thetransfer function, simply as a gain.

V dc(s)

d(s)=

2E b(1 − D)2

(3.17)

3.5 Controller

A PI controller is employed for maintaining the current in the motoring (CCA) mode and

voltage in the generating (CVD) mode. The block diagram of the controller in CCA mode

and CVD mode is shown in the fig 3.9 and fig 3.10 respectively. The current controller

will facilitate charging the flywheel while operating in CCA mode. The voltage controller

enable the system to deliver power at constant voltage in CVD mode. The current control

bandwidth (in CCA mode) of 100 rad/s with unity gain and the voltage control bandwidth

(in CVD mode) of 10 rad/s with unity gain are achieved. Protection features like dc over

voltage, dc over current, IGBT pulse inhibition in fault conditions are also provided.




Vdc

2(Ra+ sLa)

I1/K

ioCurrent

PI+−

I fb

PWM

^

I*d^ ^

Figure 3.9: Block diagram of the controller in CCA mode

v1/K

b2E V ^dc Voltage

PI+−

V fb

PWM 2(1 − D)

^

d^ V*

Figure 3.10: Block diagram of the controller in CVD mode

3.6 Switching between controllers

The software along with the power condition monitoring circuits, switch the controller from

CCA mode to CVD mode or vice-versa depending on the input power supply condition.

During normal operation the converter is operated as a brushless dc motor drive with current

limit. When the supply fails the converter is operated in boost mode maintaining the dc

bus voltage constant. Switching from CCA mode to CVD mode and vice-versa is done

automatically, depending on the input power supply condition.

3.7 Selection of circuit elements

3.7.1 Series Inductor

• Motoring (CCA) mode

Consider that the armature current reaches the rated current within 10 percent of the

conduction time. The circuit equation for establishing the current is given by,

2Ldiadt

= V dc (3.18)



3.7. Selection of circuit elements 37

Substituting following data

f = 333Hz at 10,000 rpm; T s = 3ms; Conduction time = 1.0ms.

I a = 3.3A

V dc = 300V

We get, L < 4.55mH.

• Generating (CVD) mode

The condition for the CCM of the boost converter is given by [3],

L > RLDminT s(1 − Dmin)2

2 (3.19)


T s = 300µs

Dmin = 0.1

RL = 240 Ω Corresponding to minimum load of P o = 375 watts

We get, L > 2.9mH.

The machine armature inductance (La) is 1.0mH/phase; inductance across the two ter-minals of the star connected machine is 2.0 mH. The desired external series inductance

(considering both CCA and CVD mode of operation) is,

0.9mH ≤ Ls ≤ 2.55mH .

An external inductance of 2.5mH/phase has been provided.

3.7.2 DC bus capacitor

The dc bus capacitor is selected based on the voltage ripple considerations [3].

C > I dcDmaxT s

∆V (3.20)

The capacitor ripple voltage component on account of ESR (Equivalent Series Resistance)

is also to be kept within 1.0 percent of V dc. The ESR of the capacitor,

ES R ≤∆V

I = 1Ω (3.21)

Selected capacitor value, C = 1650µF with an ESR of 0.1Ω (at 20oC , 100 Hz).




3.7.3 Switches

The current rating of the switches is selected based on the instantaneous maximum value.Maximum current through the switch (which is same as that of inductor) is given by the

relation [3],

I Lmax = 2E b

1

RL(1 − Dmax)2 +

DmaxT sLt

(3.22)

Where, Lt = 2(La + Ls)


T s = 300µs

Dmax = 0.7

2E b = 180 volts (Lowest input voltage)

RL = 180Ω corresponding to P o = 450 watts

We get I max = 11.0 Amps. Device selected is Semikron make, model: SK30 GD 123 (Data

sheet given in Appendix-A)

3.7.4 Switching frequency

Electrical time constant of the machine is La/Ra. With L = 7.096 mH and R = 2.35 Ω,

this value is 3.02 mSec. The switching period is required to be 10 times less ( ≤ 300µsec).Switching frequency selected is 3.3 kHz (T s = 300µsec).

3.8 Effects of system parameters on the performance

3.8.1 Source resistance of the machine

Energy conversion process can be shown as given in the fig 3.11. Losses in the machine can

be represented as shown in the fig 3.12. The resistance connected in parallel to the source

represents the flux dependent losses. They are, hysterisis loss and flux dependent eddy

current loss of the machine. The series component of the source resistance is a combination

of the armature winding resistance and the current dependent eddy current loss component

of the machine. The series resistance can be calculated as follows [4],

Rs =

P ei

I dc1 − D

2 + 2Ra (3.23)



3.8. Effects of system parameters on the performance 39

= Vdc Idc Tbackup

(ω2

max− ω2min)= 1/2 J

Available Energy

Flywheel

LossesCopper

LossesIron Power

ConverterLosses

Bearing friction& drag

losses in Flywheel

Bearing friction& drag

losses in Machine

Energy available

at Load

BDCand

Machine

Figure 3.11: Schematic representation of various losses in the system

2Eb Rh Re1

Re2

Rh

Re1

Re2

− Flux dependent eddy current loss component

− Hysteris loss component

BDC

− Current dependent eddy current loss and Copper loss component

Figure 3.12: Representation of voltage dependent and current dependent losses




0

1

2

3

4

5

0 0.2 0.4 0.6 0.8 1

V o l t a g e g a i n = 2 E b

/ V d c

Duty ratio

Max voltage gain

M

Figure 3.13: Effect of source resistance on the converter performance

On account of resistance Rs, the voltage conversion ratio during discharging

V dc2E b

gets

degraded. The voltage gain can be found out by the following relation [4]:

V dc =

E b

(1 − D)

1

1 +

Rs

RL

(1 − D)2

(3.24)

The effect of source resistance on the voltage gain is shown in the fig 3.13 [4]. As Rs

increases the maximum voltage gain reduces. Better voltage gain, higher efficiency and

increased backup time are achieved by keeping Rs as low as possible.

3.8.2 Weight of the flywheel

The inertia increases linearly with mass of the flywheel [7]. Stored energy will increase by

the same amount. An increase in the mass of the flywheel however, increases the frictional

loss also and is not desirable.



3.9. Effect of Backup time on the system performance 41

3.8.3 Diameter of the flywheel

The inertia increases with increasing the diameter of the flywheel in square law [7]. Thereforeincreasing the diameter keeping mass constant will result in increase in the backup time.

3.9 Effect of Backup time on the system performance

• Efficiency :

Overall efficiency of the system can be calculated by the relation,

η = P oT backup

0.5J (ω2

max − ω2

min)

= 1 − P lossT backup

0.5J (ω2

max − ω2

min)

(3.25)

From the above relation it is evident that the overall efficiency reduces as the back

up time increases (for a given ωmax and ωmin). The energy efficiency as a function of

backup time is given in the fig 3.14. It may be noted that energy efficiency is maximum

at lowest backup time. The bearing friction and iron losses are depending on speed

(voltage) and independent of load current. Armature copper loss depends on the load

current. If more power (more current at constant voltage) is drawn in less time, the

energy lost due to copper loss will increase; energy lost due to other losses mentionedabove will remain same. The copper loss is much less in comparison with other losses.

Overall energy lost reduces and the efficiency increases. Therefore efficiency is higher

at high power and low back up time.

0

0.2

0.4

0.6

0.8

1

0 100

E f f i c i e n c y

Backup Time in seconds

2.72A

1.80A1.50A

1.12A

19 3841 53

Figure 3.14: Energy efficiency vs backup time




• Energy output : The energy output as a function of back up time is shown in fig 3.15.

In the case of low backup time (higher power), the ratio Rs

RL

increases. This is on

account of increase in current dependent eddy current losses. This limits the maximum

duty cycle (in other words maximum voltage gain) operation. Thereby, limiting the

minimum speed down to which output voltage is maintained constant. This in turn,

reduces the amount of energy that can be extracted from the spinning flywheel. In the

case of high backup time (low output power), the energy lost due to iron and bearing

friction losses increases. This case also reduces the amount of energy that is extracted.

The energy extracted is maximum at particular backup time (in other words load).

This is evident from the fig 3.15. Maximum output energy can be extracted from a

given stored energy in the flywheel, if the system is operated at optimum backup time.

0

5000

10000

15000

20000

25000

0 10 20 30 40 50 60 70 80

E n e r g y i n J o u l e s

Time in seconds

Figure 3.15: Energy harvested vs backup time

3.10 Simulation of the system in CVD mode

This model is simulated in MATLAB/SIMULINK to obtain the dynamic performance of the

overall system. Following system parameters are used to simulate the system,

Moment of Inertia of the flywheel : 0.075 kgm2

Armature leakage inductance : 0.97mH / phase

Frictional torque : 2 Nm

Filter capacitance: 1650 µF and Load resistance: 100 Ω

Voltage constant of the machine : 0.2 V/rad



3.10. Simulation of the system in CVD mode 43

The results obtained are shown in the fig 3.16 and fig 3.17. The flywheel is accelerated to

a constant speed of 1046 rad/s. The input power is switched off when the time is 5 sec as

shown in fig 3.16. The flywheel starts decelerating towards zero speed as shown in fig 3.16.

The dc bus voltage is maintained constant at 300V by the BDC in boost mode (CVD) up

0 5 10 15 20 25 30 350

2000

4000

6000

8000

10000

12000

Time in seconds

G e n e r a t o r s p e e d i n R P M

Figure 3.16: Plot of speed vs time in CVD mode

to the instant of 27 sec. Therefore, the total back up time is 22 seconds (27 - 5) as shown infig 3.17.

0 5 10 15 20 25 30 350

50

100

150

200

250

300

350

Time in seconds

d c

b u s

v o l t a g e i n

v o l t

Figure 3.17: Plot of output voltage vs time in CVD mode




3.11 Conclusions

• Speed (and the back emf) changes are slow on account of mechanical time constant.

Accordingly we got a single pole transfer function in the acceleration (CCA) mode and

simple gain in the decelerating (CVD) mode. PI controllers with a bandwidth of 100

rads/sec and 10 rads/sec are employed for the CCA mode and CVD mode respectively.

• The circuit inductor should be low enough to allow the rise of armature current to

reach its rated current within 10 percent of conduction time in the accelerating (CCA)

mode. This inductor should be high enough to maintain the constant current in decel-

erating (CVD) mode. Accordingly an external inductance is added to satisfy both theconditions.

• Low mass, higher diameter flywheel is preferred to higher mass low diameter flywheel.

This gives better efficiency of the system for a given energy storage.

• An increase in equivalent source resistance increases the losses and also put a limit on

the voltage gain. This also reduces the amount of energy that is extracted. Current de-

pendent eddy current loss contributes to this resistance. It has to be kept as minimum

as possible. This can be done by the usage of thinner lamination for the machine.

• Overall energy efficiency is an inverse function of backup time. Energy delivered has a

maxima as function of power and time. This is identified and demonstrated.

Actual implementation of the system and the performance results obtained are presented in

the chapter-4.



Chapter 4

Digital Implementation and

Performace Evaluation of the System

4.1 Introduction

This chapter covers the hardware implementation, Human Machine Interface and testing

of the system. A power converter based on three phase full-bridge voltage source topology

is used. A Motorola make, Digital Signal Processor (DSP) 56F805 is used for controlling

the converter. Algorithm for the generation of switching pulses using the rotor position

information is presented; flow chart of the software is given; results of various tests conductedare presented in this chapter.

4.2 Pulse generation in CCA (motor) mode

There are three hall effect sensors for sensing rotor position. Placement of these sensors

is shown the fig 4.1(a). It may be noted that the angular distance between any of the

two adjacent sensors is 120o (electrical). Output signals of these sensors with reference to

induced voltage is shown in the fig 4.1(b). A detailed specifications of these sensors is givenin the Appendix-D. Output signals of these sensors are shaped in the signal conditioning

board to make them compatible to the DSP port. These signals are connected to port-B

of the DSP. Port-B is programmed in edge triggered interrupt mode. Highest priority is

given to this interrupt. The program flow of this interrupt service routine (ISR) is shown

in fig 4.2. This interrupt service routine generates six train of pulses (corresponding to six

IGBT switches) at the output of PWMA module as per the logic given in the fig 4.3. Out

of these six, three pulse trains (G1, G3 and G5) are used directly as the triggering signals to

45



46 Chapter 4. Digital Implementation and Performace Evaluation of the System

30 60 90 120 15000

180 210 240 270 300 330 3600 0 0 0 0 0 0 0 0 0 0 0ae

be

θe

θe

θe

θe

θe

θe

Ha

Hb

Hc

1200

Hc

aH0120

S e n s o r

P o

s i t i o n

S i g n a l s

I n d u c e d V o l t a g e

Hb

(b)

(a)

R

Y

R

B

Y

B

ec

Figure 4.1: Placement of the position sensors and their output signals

the top side power switches (T1, T3 and T5 ; E = 0). Next three train of pulses (G2, G4 and

G6) are ANDed with high frequency PWM pulses and used for driving bottom side power

switches (T2, T4, T6). This is shown in the fig 4.4. This high frequency PWM pulses are

generated by the current controller. The current controller design is given in chapter-3. Thedc bus current is taken as the feedback for the current controller. The output of the current

controller is used to track the dc current feedback. The output of the current controller is

compared with the high frequency (f s) triangular carrier to generate the PWM at desired

duty cycle pulse train. Note that, once the machine accelerates to full speed the current

drawn by the machine will drop below the set value. The machine, then will be floating on

the dc bus at full speed (approx. 10,000 rpm).

Power circuit diagram and control schematic of the converter in CCA (motor) mode is given



4.2. Pulse generation in CCA (motor) mode 47

Is it H c

bIs itH

Is it H aIs it H a

Is it H b

Is it H c T3=OFFT5=ON

T1=OFF

T3=ON

T1=ONT5=OFF

Position

ISR

N

N

N

N

N

N Return

Y

Y

YY

Y

Y

T6=OFFT2=ON

T4=OFF

T6=ON

T2=OFFT4=ON

Edge

Figure 4.2: Flow chart of the position sensor interrupt

H

H

H

H

H

H

T1

T2

T3

T4

T5

T6

T5

T6

T1

T2

T3

T4

a

c

b

a

c

b

Signal ON OFF

Figure 4.3: Sensorwise IGBT gate trigger logic diagram




GateDriveCKT

GateDriveCKT

If

G4

Position

G2G6G4

G1G3G5

_+

ea

eb

ec

DSP Core

G3G1

G6 G2

G5T1

T6

T3 T5

T2

Signals

La

La

La

+15V

Iref

E = 0.

T4

POS ISR

Port B

−

+

Figure 4.4: Power and control schematic in CCA mode

in the fig 4.4.

4.3 Pulse generation in CVD (generator) mode

Power circuit diagram and the control schematic of the converter in CVD (generator) mode

is given in the fig 4.5. The dc bus voltage is taken as the feedback for the voltage controller.

The output of the voltage controller is used to track the dc bus voltage feedback. The voltage

controller design is given in chapter-3. The output of the voltage controller is compared with

the high frequency (f s) triangular carrier to generate drive for bottom switches (T2, T4, T6)

at desired duty cycle as shown in the fig 4.5. The topside switches of the bridge are kept off

permanently in this mode (E = 1).

4.4 Software implementation

4.4.1 Human Machine Interface (HMI)

HMI is a digital interface between the controller and the operator. It takes the input com-

mand and data through the keys, and passes it on to the controller. The processor uses these



4.4. Software implementation 49

GateDriveCKT

GateDriveCKT

+

_

LR

G4

G2G6G4

G1G3G5

G3G1

G6 G2

G5

C

E = 1.

Vref

T1 T3 T5

T2T6T4

+ dc bus

− dc bus

ea

eb

ec

La

La

La

DSP Core

Figure 4.5: Power and control schematic in CVD mode

data to control the machine as desired by the operator. HMI also takes the messages and

data from the controller and display on the LCD panel. This can be read by the operator

to know various system parameters.

4.4.1.1 Keypad

This program is executed whenever ‘KeyPadFlag’ is set. It checks whether one or more

keys are pressed. If so, the processor checks whether it is a command or data and takes the

appropriate action. The flow chart of the keypad interface program is shown in the fig 4.6.

4.4.1.2 Display

This program is executed whenever the ‘DisplayFlag’ is set. Display interface routine sends

the messages to LCD display unit. This LCD diplay unit consists of 4 rows of 20 characters

each. The message is serially transferred, character by character to the LCD unit. The flow

chart of of this routine is shown in the fig 4.7.




ModeData

Y

Y

NKey Pressed

Stop

Start

Mode

Parameter

Key pad routine

Data = Data −1 Pointer = Pointer −1

Mode = ! Mode

Return

Enter

Y

N

Out put the messageDisable the pulsesSwitch OFF input

Start soft start timer

Switch ON inputEnable the pulses

Y

N

N

N

Y

ParameterData Is MODE

Data = Data + 1 Pointer= Pointer + 1

Data = Temp Buffer

N

Y Y

N

Decreament

Increament

Figure 4.6: Flow chart of the key pad interface routine



4.4. Software implementation 51

Y

N

put the char inTx buff

Return

sentcharacters are

All N

Y Any

message tobe sent

Display routine

Figure 4.7: Flow chart of the display routine

4.4.2 Timer interrupt service routine

Timer is programmed to interrupt the processor at regular intervals of time. The timer in-

terrupt service routine sets flags like DisplayFlag, KeyPadFlag, SoftStartFlag etc at different

intervals of time. The flow chart of the timer ISR is given in the fig 4.8. These flags are used

by different tasks to start the execution. This scheduling of various tasks is given in the flow

chart of main program (next subsection).

4.4.3 Main program

This is the main program. This program monitors and controls different tasks of the to-

tal system. This program calls various functions like StartADC, ReadADC, PIController,

ScanKeyPad, Display, ActOnAlarm etc at predetermined time intervals allowing them to

carry out their tasks. Processor also takes appropriate actions during interrupts like ‘rotor

position sensor interrupt’, ‘timer interrupt’ etc. The flow chart of the main program is shown

in the fig 4.9.




N

N

Y

TrafCharFlag = 1

ADC Flag = 1

Timer ISR

DisplayFlag = 1

N

Y

N

N

Y

Y

Return

Is it 10ms

UpdateFlag =1

KeypadFlag = 1

Is it 50ms

Is it 100msIs it 600us

Figure 4.8: Flow chart of the timer interrupt service routine

4.5 Hardware implementation of the system

4.5.1 Controller hardware

The entire control algorithm is realized on a digital controller platform. The main proces-

sor is a special purpose DSP controller (Motorola make, 56F805) with a set of peripherals

tailored for motor drive applications. It is a 16-bit, fixed point processor and is operated at

an internal clock of 80MHz. A 12-bit, 8-channel (multiplexed) unipolar Analog to Digital

Converter (ADC) enable sampling of analog signals. Besides, there are digital I/O ports

for handling digital variables. Another on-chip module is two sets of carrier based PWM

switching schemes. This processor also has a serial communication port for communicatingwith the user interface units. Programmable internal timers clock a set of counters, which

can be routed to software interrupts. These can be used for start of sampling of analog

signals, scanning the keys, refreshing the display etc. A signal processing card is also a part

of the platform. This scales and conditions the input analog and digital signals to make it

compatible to DSP ports. This also amplify the output signals to suit the interfacing cir-

cuitry external to the processor (DSP). All control functions and HMI interface are carried

out using this processor.



4.5. Hardware implementation of the system 53

Mode

CallV PI

Call

I PI

Read ADC

YN

Stop keypressed

Receive key siganl

&take actionKeypad flag

Switch OFF input contactor

Disable pulses

to PWM value register

Load the PI output

display

Refresh the

Update theset value

Alarm flag

N

Y

Display flag

Update flag

N

Y

Y

Y

N

Generator Motor

Start ADC

ADC Flag

If

Switch ON input contactor, Start softstart timer

Make PI output = 0, Enable pulses

Y

NStart key

pressed

START

Refresh display to give start message

InterruptsEnable the

N

N

message to display

Trip and send

Initialize the ports

Y

Figure 4.9: Flow chart of the main program




GateDriveCKT

GateDriveCKT

Vf If

BLDC

Machine

ChargeR

G2

G4

G1G3G5

A D C

P o r t B

G6

D1 D3 D5 G3G1

G6 G2

G5

Position

Signal

P W M ’ A ’ P o r t

T1

T6

T3 T5

T2

+15v

E=0

D4 D6 D2

C RL

Signal processingcircuitry

G4 T4

m o n i t o r i n g s i g n a l

I n p u t p o w e r c o n d i t i o n

Raw ACInput

HMI

−dc bus

+dc bus

Power switching module

DSP core

Figure 4.10: Block diagram of the test setup

4.5.2 Power converter

The power converter is built using the IGBT as the power device. Complete block diagram

of the system is shown in the fig 4.10. The system design is validated on an experimental

setup. The rating of the converter are,

• Input voltage: 230volts, 50Hz, line-to-line.

• Output voltage: 300 V DC

• Output power: 1.0 kW.

• Switching frequency: 3.3kHz.

• Line to line inductance (including machine leakage): 7.096mH.

• DC bus capacitance: 1650 µF , 400 V



4.6. Testing and performance analysis of the system 55

Windage losses

LoadFlywheel Machine BDC

Hysterisis + Eddy Current+ Copper losses

Conduction and

Switching lossesBearing Friction losses

Figure 4.11: Various losses and their sources in the system

4.6 Testing and performance analysis of the system

4.6.1 Apportioning of various losses

Between the flywheel (which stores the energy) and load (which consumes the energy) there

are different devices like, bearing, electrical machine, bi-directional power converter as shown

in the fig G.1. A portion of the energy which is extracted from the flywheel is dissipated as

loss in these devices. It is necessary to find out these losses. Following tests are conducted

to find and separate out the various losses.

4.6.1.1 No load test

The flywheel is decoupled from the motor shaft. The motor is made to run at different

speeds up to a speed of 10,000 RPM. Machine draws power only to meet the losses. The

weight of the rotor is very small compared to that of flywheel. The mechanical losses in this

condition is assumed to be negligible. All the power drawn is to meet the iron loss of the

machine. Power consumed by the machine at various speeds ( V dcI dc) is recorded and plotted

as shown in the fig 4.12. This test enables us to find the iron loss in the machine which is

predominantly speed dependent.

4.6.1.2 Retardation test with flywheel

In order to evaluate the mechanical losses in the machine, the classical retardation test is

done. The machine along with the flywheel is made to run up to a speed of 10,000 RPM.

Power is cut off and the machine is allowed to decelerate. This data is given in the table 4.1.




0

100

200

300

0 2000 4000 6000 8000 10000 12000

I r o n l o s s i n w a t t s

Speed in RPM

No load test

(without flywheel)

Figure 4.12: Iron losses in the machine in no load test

The same data is plotted as shown in the fig 4.13. The stored energy in the flywheel is

Time 0 30 60 90 120 150 180 210 240 270 300

in secs

Speed 9670 8200 6700 5550 4600 3700 2900 2200 1600 1100 650in RPM

Table 4.1: Retardation test data

consumed as the machine decelerates. The losses in this test condition include the bearing

friction losses, drag and the iron losses in the machine. The power loss at each speed can be

calculated by the following relationship,

P (ω) = J ωdω

dt (4.1)

The power lost at various speeds is computed from the retardation test data. The same

is plotted as a function of speed as shown in the fig G.2. For the sake of comparison, the

iron loss calculated from the no load test are also plotted in fig G.2. The point on curve

‘retardation test with flywheel’ gives the total loss and the point on curve ‘No load test

without flywheel’ gives only iron loss of the machine. The difference between these two gives

the mechanical losses (bearing friction, drag) at that speed.




0

2000

4000

6000

8000

10000

0 50 100 150 200 250 300

R o t o r s p e e d i n R P M

Time in seconds

Figure 4.13: Speed vs time in the retardation test with flywheel

4.6.1.3 Copper losses in the armature winding

In CVD mode, the power converter works in boost mode. The converter input is connected

to machine terminals. The machine is connected in star. Therefore, the converter sees two

sets of armature windings and series chokes. The equivalent circuit of converter connected

to the machine is shown in fig 4.15. Copper losses in the winding can be calculated by,

P cu = I 2aRs (4.2)

Where Rs = 2(Ra + Rch). In boost mode I a and I dc are related by,

I a = I dc1 − D

(4.3)

V dc and 2E b are related by,V dc

2E b= 1

1 − D (4.4)

Earlier, in chapter-2 (equation 2.8), the back emf of the machine has been shown to be,

2E b = 0.0277S r (4.5)

Where S r is the rotor speed in RPM.

Combining equations 4.3, 4.4 and 4.5 we get,

I a = I dcV dc

(0.0277S r) (4.6)




0

100

200

300

400

500

0 2000 4000 6000 8000 10000 12000

0

100

200

300

400

500

I r o n a n d m e c h a n i c a l l o s s i n w a t t s


Speed in RPM

Retardation test with

flywheel(No load)

No load test

without flywheel

Figure 4.14: Losses in the machine with and with out flywheel

Eb

RL Vdc

La Lch2( + ) Ra Rch2( + )

Ia

Idc

2

C

Figure 4.15: Equivalent circuit of boost converter for copper loss calculation

Therefore, the copper loss is given by,

P cu = 2

I dcV dc

(0.0277S r)

2

[(Ra + Rch)] (4.7)

These losses are presented later in table G.7.

4.6.1.4 Switching and conduction losses in the converter

Switching and conduction losses in the power converter are computed as follows:

• Switching losses: If I ON is the on state current and V OFF is off state voltage of the

switch, tr and f s are the rise time and the switching frequency respectively of the




switch, then the switching loss in a switch driving an inductive load is given by [3],

P sw = 0.5I ON V OFF trf s (4.8)

While BDC is operating in CVD mode, I ON is I a and V OFF is V dc. Substituting for I a

from equation 4.6, we get for each switch,

P sw = 0.5 I dcV dc

(0.0277S r)V dctrf s + 0.5

I dcV dc(0.0277S r)

V dctf f s (4.9)

Switching loss in the diodes is very small and neglected.

• Conduction loss: The current flowing across the switch is I a. Assuming the switch

drop to be V d, the conduction loss in the switch (P c1) can be computed by,

P c1 = I aV d (4.10)

Substituting for I a from equation 4.6, we get loss in each device.

P c1 = I dcV dc

(0.0277S r)V d (4.11)

There are three devices and each conducting for a duration of 120 o in a cycle. The

total loss of all devices is given by,

P c = 3

1

3

I dcV dc(0.0277S r)

V d

(4.12)

These losses are presented later in table G.7.

4.6.2 Power backup time test

This test is conducted to find out following parameters.

• Time duration for which the system supplies the power to the load (in the absence of

input power) at desired (constant) voltage.

• The minimum speed (or induced voltage) up to which the system maintains the output

dc bus voltage.




The input supply is switched off and the flywheel is allowed to decelerate. The time up to

which the dc bus voltage is maintained constant is recorded. This test is conducted with

a load current of 2.72A, at a dc bus voltage of 300V. The power delevered to the load is

818W. Fig 4.16 shows the plot of variations of output voltage with time. It may be noted

that, even though the generator voltage reduces with time (as speed is reduced), the dc bus

voltage is maintained constant for a duration of 19 sec. It is also observed that the minimum

speed (or the voltage) up to which system maintains the output dc bus voltage is 6894 RPM.

(corresponding induced voltage is 185 volts). It is to be noted that when the power drawn

0

100

200

300

400

0 10 20 30 40 50 60 70

V o l t a g e i n v o l t s

Time in seconds

Controlled

operation

Uncontrolled

operation

185V

19 Sec

Generator Voltage (2Eb)

dc bus Voltage (Vdc)

Figure 4.16: dc bus voltage control with P o = 818 watts

is less, the duration of power delivery at constant voltage is longer. Fig 4.17 shows a similar

result at a load of 450W. At this reduced power delivery, the break time is seen to be 41 sec.

4.6.3 Source resistance effect

The voltage gain of the system can be calculated by the following relationship [3].

V dcE b

=

1

(1 − D)

1

1 +

Rs

RL

(1 − D)2

(4.13)




0

100

200

300

400

0 10 20 30 40 50 60 70

V o l t a g e i n v o l t s

Time in seconds

Controlled

operation

Uncontrolled

operation

127V

41 Sec

Generator Voltage (2Eb)

dc bus Voltage (Vdc)

Figure 4.17: dc bus voltage control with P o = 450 watts

It may be noted that, Rs is the source resistance which is the series combination of armature

winding resistance, Ra of the machine and the resistance of the series choke Rch. Ra, Rch

and RL in the equivalent circuit is shown in the fig 4.15. Voltage is applied between the

input terminals of the star connected machine. The equivalent source resistance is (due to

two windings coming in series),

Rs = 2(Ra + Rch) (4.14)

In the experimental setup, the values of Ra = 0.38Ω, Rch = 0.8Ω and RL = 100Ω. The

ratio α = Rs

RL

is 0.024. The plot of voltage gain as a function of duty ratio for the value of

α = 0.024 is shown in fig 4.18. It can be seen from this plot that the voltage gain increases as

the duty ratio is increased. After a certain value of duty cycle, it starts decreasing. Prefered

operating duty ratio of the boost converter is from zero to ‘M’ as shown in the fig 4.18. The

peak of this graph is the maximum voltage gain one can get from the boost converter for agiven α. This peak value is a function of α. The ‘α’, will limit the maximum operating duty

cycle. It may be observed from fig 4.18 that the maximum voltage gain that can be obtained

from the setup (with α = 0.024) is 3.23 and the corresponding Dmax = 0.85. This means that

if dc bus voltage to be maintained is 300 volts, then the input voltage can go down to 93 volts

(and corresponding speed is 3900 rpm). It may be observed from the ‘Power backup time

test’ described in the previous section that the minimum voltage down to which converter

maintains the output dc bus voltage at 300V is only 185V. This means that the maximum




0

0.5

1

1.5

2

2.5

3

3.5

0 0.2 0.4 0.6 0.8 1

V o l t a g e g a i n V d c

/ 2 E b

Duty cycle

M

Figure 4.18: Voltage gain as a function of duty cycle with α = 0.024

voltage gain obtained is only 1.6. This is in contradiction to the estimated maximum gain of

3.23. The experiment was carried out at several other load settings to rule out experimental

error. The maximum voltage gain obtained at these loads are given in the table 4.2. It is

Output power Minimum speed Max Voltage Estimated max

in Watts in RPM gain voltage gain

818 6894 1.6 3.23

450 4691 2.36 4.33

338 4291 2.52 4.97

Table 4.2: Max voltage gain test data

observed that the voltage gain is more at lower armature currents and less at higher armature

currents. It is clear that the equivalent series resistance seen by the circuit is more than the dc

resistance measured. It is also seen that, this equivalent resistance is a function of armature

current. To confirm this, a test was conducted to compute Rs at different duty cycles. In this

test, the converter output voltage (dc bus voltage), input voltage (generator voltage) and




duty cycle are recorded. The source resistance Rs is computed from the following relation,

V dcE b

=

1

(1 − D)

1

1 +

Rs

RL

(1 − D)2

(4.15)

The ratio of source resistance to load resistance as a function of duty cycle is plotted as

shown in the fig 4.19. It is clear from the fig 4.19 that the source resistance is more at higher

0

0.05

0.1

0.15

0.2

0.25

0.2 0.25 0.3 0.35 0.4 0.45 0.5 0.55 0.6 S o u r c e R e s i s t a n c e / L o a d R e s i s t a n c e

Duty ratio

RL = 180

RL = 100

Ratio = 0.013

Ratio = 0.024RL = 100

RL = 180

Figure 4.19: Source resistance as a function of duty cycle

armature current (low load resistance) and is less at lower armature current. This increase

in Rs may be due to the loss that is taking place in the core, reflecting as series resistance

(since it is current dependent). It is proposed that, this is due to the the current dependent

losses in the core. This may be on account of the the eddy current losses due to the leakage

flux around the teeth of the core. This flux is produced by the current flowing through the

armature conductors. Therefore these losses are armature current dependent. This loss in

the core is reflected as a series resistance.

4.6.4 Current dependent eddy current loss in the core

Duty cycle and the output (dc bus) voltages are recorded at various induced voltages for a

given load resistance. The source resistance Rs is calculated by using the equation 4.15. If




Rs is the equivalent source resistance as seen by the converter, Ra is the armature winding

resistance and Rch is the resistance of the series choke, then the current dependent eddy

current loss is given by,

P ei = I 2a [Rs − 2(Ra + Rch)] (4.16)

Substituting for I a from equation 4.6 we get,

P ei =

I dcV dc

(0.0277S r)

2

[Rs − 2(Ra + Rch)] (4.17)

This loss also presented later in table G.7.

4.6.5 Comparison of various loss components

Various losses are computed/measured at an output power of 450 Watts. This is shown in

the table G.7. It may be noted that the copper loss P copper shown is the total of loss caused

by the armature dc resistance (P cu) and the current dependent eddy current loss (P ei). It is

Speed P mech P iron P copper P c P sw

in RPM in watts in watts in watts in watts in watts

9835 233 163 48 2.5 1

(6.7 + 41.4)

4691 66 50 115 5.25 2

(30 + 85)

Table 4.3: Comparison of various losses

evident from these tests that the highest contribution to the loss is from the bearing friction

and drag (Mechanical losses). Second highest contribution is from the iron losses in the core

of the machine. Next contribution is from the the copper losses of the machine. This copper

loss is the sum of the current dependent eddy current loss in the core and the power loss

in the dc resistance of copper wire used. The switching and conduction losses in the power

converter are very low and negligible.

4.6.6 Efficiency

The above exercises of apportioning the losses and quantifying the same will help us in

understanding the operating efficiency of the FES. The total energy that can be harvested




depends on the losses in the system. Tests are conducted with various loads (different backup

time) and the energy efficiency is calculated at each load. The results are plotted as shown

in fig 4.20. It is found that the energy efficiency is an inverse function of time. The bearing

friction and iron losses are dependent on the operating speed (voltage). Amount of energy

lost will increase if backup time is increased and efficiency decreases. This is explained in

detail in the section 3.9 of chapter-3.

0

0.2

0.4

0.6

0.8

1

0 20 40 60 80 100

E f f i c i e n c y

Time in seconds

Figure 4.20: Overall efficiency as a function of back up time

4.6.7 Harvestable Energy

The aim of this test is to find out the amount of energy that can be harvested at different

loads. Tests were conducted with various loads connected across the dc bus. The results are

given in the table 4.4. It is evident from these tests that the energy harvested has a maxima

Output power Backup time Energy harvested Efficiency

in Watts in secs in Joules

818 19 16359 0.78

450 41 18953 0.62

338 53 17937 0.56

Table 4.4: Test results at various load conditions




0

5000

10000

15000

20000

25000

0 10 20 30 40 50 60 70 80

H a r v e s t a b l e E n e r g y i n J o u l e s

Time in seconds

Maximum

energy

harvested

38 SecEnergy loss = 9973J

Figure 4.21: Energy harvested as a function of back up time

as a function of power and time. This is represented graphically as shown in fig 4.21. In this

set up it has occurred at 542 Watts, 38 sec. The reason for drop in harvested energy both

in lower backup time and higher backup time is explained in the section 3.9 of chapter-3.

4.6.8 Current waveforms at various speeds

Sample current waveforms of the machine are recorded in CCA mode and CVD mode. They

are shown in the fig 4.22 to fig 4.24. It may be observed that the current waveform shown

Figure 4.22: Armature current waveform in CCA mode at speed = 2200 RPM

(CH1(Top): Time:1ms/div, Current:1A/div,CH2(Bottom): Time:1ms/div, Voltage:2V/div)




Figure 4.23: Armature current waveform in CCA mode at speed = 9640 RPM)

(Time: 1ms/div, Current:1A/div)

Figure 4.24: Armature current waveform in CVD mode

(Time: 2ms/div, Current: 1A/div)




in fig 4.23 has positive slope as well as negative slope in the middle of the waveform. This

is due to the advancement of the triggering pulses given to the switches.

4.7 Conclusions

Various losses in the machine are separated out. It has been found that the major contribu-

tion to the losses is from mechanical losses, iron loss and the copper losses. The total energy

that can be harvested depends on the losses in the system. Current dependent eddy current

loss contribute for increase of source resistance of the converter. This will put a constraint

on the maximum voltage gain. This in turn, limit the total energy that can be extracted

for a given top speed. Usage of low loss core material like Nickel-iron or cobalt-iron for the

machine and ferrite for chokes will improve the overall efficiency of the system. This will

also help in reducing the source resistance of the converter. From the experiments it is found

that the energy extracted has a maxima as a function of time. For a given system there is

clearly a peak energy output, where the energy extracted will be maximum.



Chapter 5

Conclusions

In many power drive applications such as traction, elevators, cranes etc, it is commonplace to

encounter loss of stored energy. The main reason is that, the power converters are not capable

of returning the stored energy during transients. In application where frequent transients

are involved, this results in substantial loss of energy. Bi-directional converters in such

applications can lead to higher operating efficiency. In a typical traction application, stored

energy while running can be restored during deceleration. This process saves the energy

and improves the efficiency. Such applications need a bi-directional interfacing converter.

The bi-directional converter facilitates the energy flow, to and from the device. The desired

features of such a system are,

• Good energy efficiency.

• Simple control.

• Reliability.

• Low cost.

• Small size.

The aim of this work is to develop a bi-directional power converter/controller to facilitate

the energy storage, to and from the storage device. The storage system employed in this

application consists of a BLDC machine and a flywheel; together they serve as a flywheel

energy storage system. The analysis, design, fabrication and evaluation of such a system has

been covered in this thesis. The full system has been evaluated and design guidelines are

obtained.

69



70 Chapter 5. Conclusions

5.1 The present work

Chapter 1 focused on the essential basic specifications, guiding the selection of subsystems.

These systems are, bi-directional power converter, controller, BLDC machine and the fly-

wheel. The bi-directional power converter selected is of six - switch voltage source bridge

topology. The BLDC machine is selected for this application is due to its high power density,

low rotor losses and simplicity of control. The operating modes during charging and dis-

charging of the flywheel are identified. The control platform selected is a DSP of Motorola

make, 56F805. A flywheel of suitable dimensions and capable of storing the energy required

to supply 1.0kW load for a duration of 20 sec has been selected.

Chapter 2 is on the prime-mover required to drive the flywheel. Basic design of BLDC

machine has been carried out. The design is verified using a FE method of analysis using

‘Magnet’ software. The machine is fabricated using standard available frame of nearest

dimensions. The machine is tested up to a speed of 10,000 rpm. Design of the machine is

validated through the experimental results.

Chapter 3 is on the bi-directional power converter and its control. A 6-switch IGBT

bridge of voltage source topology is selected. The operating modes are identified as CCA

and CVD modes. Equivalent circuit and the transfer function in both the modes are obtained.

Suitable controllers for both the modes of operations are designed. The system is numerically

simulated to check the performance.

Chapter 4 presents the digital realization of the system. The controller and HMI are

implemented using a digital signal processor of Motorola make, 56F805. Seamless changeover

from CCA mode to CVD mode and vice versa in the controller is implemented. The power

converter and controller are fabricated; bi-directional converter/controller is integrated with

BLDC machine and the flywheel. The complete system is tested to evaluate the performance.

The results are analyzed to obtain design guidelines for such systems.

Features of the system are,

• Speed of operation is limited to 10,000 due to drag and safety issues.

• Control is simple

Important findings of this thesis are,

• Low efficiency on account of high iron losses and bearing losses.



5.2. Guidelines emerging from the work 71

• Current dependent eddy current loss contribute to the increase of source resistance of

the converter. This puts a constraint on the operating maximum voltage gain. This

in turn, limit the total energy that can be harvested for a given top speed.

5.2 Guidelines emerging from the work

In the present system, the source resistance is the result of the core losses in the machine.

This is required to be reduced to improve the efficiency. The current dependent losses has

to be reduced for harvesting maximum energy stored in the flywheel. These are achieved by

the usage of,

• Vacuum enclosure for rotating parts.

• Two pole machine. For two pole machine, the operating frequency is 167Hz. This, in

turn, results in lower iron loss.

• Low specific loss core materials like, Ni-Iron, Co-Iron etc.

• Low loss, contact-less bearing like magnetic bearing.

Study of these solutions have been done and low cost implementation is given in Appendix-G

5.3 Spin off technology from the present system

This system can be tailored to store and extract energy from super capacitor as the storage

device. The same power circuit can be used as multi-phase chopper in the super capacitor

energy storage application.

5.4 Applications of the system

The FES system which is developed, can be used as an UPS where short support time is

required. This will be useful for the installations which are backed up with diesel genera-

tors. This bi-directional converter along with the BLDC machine can be directly used for

hybrid vehicles and material handling equipments to improve their performance in terms of

efficiency, control and reduction of environmental pollution. With some modifications in the

control circuit and software, the same bi-directional converter can be used for energy storage

applications using the ultra capacitors.



Appendix A

Specifications of IGBT and Capacitor

A.1 IGBT Module

Parameter : Value

Make : Semikron

Model : Semitop; 6-devices pack

PartNumber : SK 30 GD 123

V CES : 1200 V

V GES : +/- 20 V

V CESat : 3.1 V

I C : 22A at 80o

C / 33A at 25o

C tdon : 65ns

tr : 100ns

tdoff : 430ns

tf : 35ns

A.2 Capacitor

Parameter : Value

V alue : 3300 µF

Surge : 440 V

ES R : 49 mΩ at 20oC at 100Hz

Impedance : 36 mΩ at 20oC at 10kHz

I Ripple : 12.7 A at 85oC at 100Hz

: 16.5 A at 85oC at 10kHz

72



Appendix B

Specifications of Digital Signal

Processor DSP56F805

B.1 Digital Signal Processing Core

• 16-bit DSP engine with dual Harvard architecture

• 40 Million Instructions Per Second (MIPS) at 80 MHz core frequency

• Single-cycle 16-bit parallel Multiplier-Accumulator (MAC)

• Two 36-bit accumulators, including extension bits

• 16-bit bidirectional barrel shifter

• Hardware DO and REP loops

• Three internal address buses and one external address bus

• Four internal data buses and one external data bus

• Instruction set supports both DSP and controller functions

• Controller style addressing modes and instructions for compact code

• Efficient C compiler and local variable support

• Software subroutine and interrupt stack with depth limited only by memory

• JTAG/OnCE debug programming interface

73



74 Appendix B. Specifications of Digital Signal Processor DSP56F805

B.2 Memory

• Harvard architecture permits as many as three simultaneous accesses to program and

data memory

• 32K, 16 bit words of Program Flash

• 512, 16-bit words of Program RAM

• 2K, 16-bit words of Data RAM

• 4K, 16-bit words of Data Flash

• 2K, 16-bit words of BootFlash

• Off-chip memory expansion capabilities programmable for 0, 4, 8, or 12 wait states

• 64K, 16 - bits of data memory

• 64K, 16 bits of program memory

B.3 Peripheral Circuits for DSP56F805• Two Pulse Width Modulator modules (PWMA and PWMB) each with six PWM

outputs, three Current Sense inputs, and four Fault inputs, fault tolerant design with

dead-time insertion; supports both center and edge aligned modes

• 12-bit Analog-to-Digital Converters (ADC) which support two simultaneous conver-

sions with two 4-multiplexed inputs

• Two Quadrature Decoders

• Two General Purpose Quad Timers

• CAN 2.0 Module

• Two Serialm Communication Interfaces (SCI0 and SCI1)

• Serial Peripheral Interface (SPI)

• 14 dedicated General Purpose I/O (GPIO) pins, 18 multiplexed GPIO pins



B.3. Peripheral Circuits for DSP56F805 75

• Computer Operating Properly (COP) watchdog timer

• Two dedicated external interrupt pins

• External reset pin for hardware reset

• JTAG/On-Chip Emulation (OnCE) module for debugging

• Software-programmable, Phase Lock Loop-based frequency synthesizer for the DSP

core clock

• Fabricated in high-density CMOS with 5V tolerant, TTL-compatible digital inputs

• Uses a single 3.3V power supply

• On-chip regulators for digital and analog circuitry to lower cost and reduce noise

• Wait and Stop modes available



Appendix C

Block Diagram of Controller

SP1

SCI#0

SCI#1

CAN

TIMERGPIO

PWM#1

PWM#2

A/D

3.3V,GNDXTAL

JTAG/OnCEJTAGConnector

Low Freq

Crystal

PrimaryUNI−1

SecondaryUNI−3

Power Supply

+3.3V,+5V

RS232Interface

10Bit D/A4 Channel

D Sub 9−Pin

CAN Interface

Debug LED’s

PWM LED’s

Over V Sense

Over I Sense

ZC Detector

Expantion

Pheripheral

Connector

Memory ExpConnector

16KX16 Bit

Data Memory

D Sub

25−Pin

DSP56F805

Signal

Processecing

Circuitry

& Interface

RESET

MODE

16KX16 BitMemoryProgram

MODE/IRQLogic

RESET Logic

Parallel JTAGInterface

Add,Data

Control

Figure C.1: Block Schematic of DSP Board

76



Appendix D

Specifications of Hall effect position

Sensor

Make : Honeywell

Part Number : SS413A

Type : Bi-polar

Supply Voltage : 3.8 to 30 V

Supply Current : 10mA

Output Type : Sink

Output Voltage : 40 VOutput Current : 20mA

tr : 0.05 µs typ

tf : 0.15 µs typ

77



Appendix E

Publication

1. S.R. Gurumurthy, V. Ramanarayanan, M.R. Srikanthan ‘Design and Evaluation of DSP

controlled BLDC drive for Flywheel energy storage system’ presented in National Power

Electronics Conference, NPEC 2005, Indian Institute of Technology, Kharagpur, India, Dec

22 - 24, 2005.

78



Appendix F

Photographs of the test setup

79



80 Appendix F. Photographs of the test setup

Figure F.1: Bi-directional Power Converter



81

Figure F.2: Brushless DC machine coupled to Flywheel



82 Appendix F. Photographs of the test setup

Figure F.3: Test set up of Flywheel Energy Storage System



Appendix G

Further improvements in the system

G.1 Method used in apportioning various losses

Various losses take place in the system are shown schematically in the fig G.1. It is possible to

compute the sources of losses and their contributions to the total loss. With this information

one can adopt different techniques that can be adopted to reduce the losses. Following tests

were conducted to find out the sources of losses.

• No load test (without flywheel)

• Retardation test (with flywheel)

No load test is conducted without flywheel and hence the mechanical losses are negligible.

Therefore, with this no load test data the iron loss of the machine is computed. Retardation

test with flywheels of different mass were carried out. The losses in the system are computed

and tabulated using data obtained from retardation test are given in the table G.1. This

gives the sum total of the mechanical losses along with the iron losses (electrical losses).

These losses are plotted as a function of speed as shown in the fig G.2.

G.2 Relation between the speed and the loss:

The relation between the loss as a function of speed is found out using curve fitting. If P Loss

is the total loss in the system and ”N” is the operating speed in RPM of the machine, then

the equation obtained are given as below:

• Without flywheel

P Loss = 0.000001N 2 + 0.0045N (G.1)

83



84 Appendix G. Further improvements in the system

Windage losses

LoadFlywheel Machine BDC

Hysterisis + Eddy Current+ Copper losses

Conduction and

Switching lossesBearing Friction losses

Figure G.1: Various losses and their sources in the system

Speed Power loss (watts) Power loss (watts) Power loss (watts) Power loss (watts)in RPM without flywheel with 11Kg flywheel with 15Kg flywheel with 21Kg flywheel

1880 13 31 46 56

3595 35 88 110 128

5395 60 145 195 250

7200 96 206 320 400

8950 138 294 441 550

9892 165 350 - -

Table G.1: Total loss at various mass as a function of rotor speed

• With flywheel of 11 Kg

P Loss = 0.000002N 2 + 0.0163N (G.2)


P Loss = 0.000003N 2 + 0.0191N (G.3)


P Loss = 0.000004N 2 + 0.0266N (G.4)

It is found that the right hand side of the equation contains two terms, one is proportional

to the speed and the other is proportional to the square of speed. This is in expected lines.

From the theory, it can be shown the term which is proportional to speed is corresponding



G.3. Interpretation of the equation : 85

0

100

200

300

400

500

600

0 2000 4000 6000 8000 10000 12000

0

100

200

300

400

500

600

I r o n a n d m e c h a n i c a l l o s s i n w a t t s


Speed in RPM

21 Kg

15 Kg

11 Kg

No load test

without flywheel

Figure G.2: Losses in the machine with and with out flywheel

to loss due to the bearing friction and hysteresis in the core of the machine; the term which

is proportional to the square of the speed is corresponding to loss due to the air drag and

eddy current in the core of the machine.

G.3 Interpretation of the equation :

• Without flywheel (no load test):

When flywheel is not coupled to the rotor shaft, the loss due to the bearing friction as

well as the drag can be neglected. This is on account of low mass and surface area of

the rotor. Therefore, the loss computed from this test can be entirely due to the iron

loss of the machine.

• With flywheel:

When flywheel is coupled to the rotor shaft, loss computed is the sum total of all the

losses in the machine. They are, bearing friction, drag, hysteresis and eddy current

loss. Subtracting the square component of ”no load test” from square component of

loss obtained from this test gives the drag loss in the system. Similarly, subtracting the

linear component of ”no load test” from the linear component of loss obtained from

this test gives the bearing friction loss of the machine.



88 Appendix G. Further improvements in the system

G.4 Loss reduction techniques:

Drag loss can be reduced by providing the vacuum enclosure for the rotaing parts of the

system. Both eddy current and hysteresis loss can be reduced by the using a two pole machine

(half the supply frequency) as well as by using low specific loss core material. Bearing friction

loss can be reduced by using a active magnetic bearings. This is a complicated technology

and the system becomes expensive. Using a two pole machine will be a cheaper option

compared to costly low specific loss core material. Usage of two pole machine will reduce

the flux dependent as well as armature current dependent eddy current loss to the extent

of 75 percent (compared to 4 - pole machine). Vacuum enclosure will be a cheaper optioncompared to magnetic bearings. Keeping the rotating parts in a enclosure with vacuum

of 0.1 mb, the loss due to drag will get reduce to the extent of 75 percent (compared to

atmosphere). Therefore, implementing the system with vacuum enclosure and two pole

machine will reduce the overall loss by 50 percent. Extra-polation of the results obtained

Speed P d P bf P h P e P cu P conv P Total

in RPM in watts in watts in watts in watts in watts in watts in watts

9835 96 116 44 97 48 3.5 4054691 22 55 21 22 115 7.25 243

Table G.7: Comparison of various losses

from the experiments conducted will validate this point. From the results shown in the

chapter-4 it is found that the energy havested is maximum at a backup time of 41 sec.

Hence various losses are computed at a back up time of 41 seconds. The maximum and

minimum speed considered are 9835 RPM and 4691 RPM. The experimental set up consistsof the flywheel of J = 0.075Kgm2 running in air (atmospheric pressure) coupled to a 4-pole

BLDC machine. Break up of various losses for the existing set up are given in the table G.7.

If a 2-pole machine is used, then the flux dependent as well as armature current dependent

eddy current loss will become 1

4th and the hysteresis loss will become (

1

2)half. The losses are

computed accordingly. If the rotating parts are kept in a vacuum enclosure (with a pressure

of 0.1mb) the drag losses becomes 1

4th. With the reduction in the system losses, the overall

efficiency of the system will improve. Overall efficiency of the system is computed using the



G.5. conclusions: 89

Power loss Power loss Average Overall System

at 9835 rpm at 4691 rpm power loss efficiency conditions405 243 325 0.60 4-pole machine

Running in air

278 151 215 0.72 2-pole machine

Running in air

206 134 170 0.78 2-pole machine

Running in vacuum of 0.1mb

Table G.8: Comparison of various systems

relation,

η =

P oT backup

0.5J (ω2

max − ω2

min)

= 1 −

P lossT backup

0.5J (ω2

max − ω2

min)

(G.5)

Summary of these loss reduction and the effect on the overall efficiency is given in the

table G.8.

G.5 conclusions:

Low cost and easy solutions for improving the efficiency of the system are vacuum enclosure

for rotaitng parts and usage of two-pole machines. With these techniques the efficiency will

be as high as 80 percent.



References

[1] Brushless Permanent-magnet Motor Design - Duane C.Hanselman University of Maine,

Orono, Maine; McGraw-Hill, Inc (1994).

[2] Electrical Engineering Design Manual - S.Parker Smith and M.G.Say; Second Edition

1950; Chapman and Hall Ltd (1950).

[3] Power Electronics Circuits - Issa Batarseh, University of Central Florida; John Wiley

and Sons, Inc (2004).

[4] Fundamentals of Power Electronics - Robert Errickson, Second Edition; Kluwer

Acadamic Publishers.

[5] Electric Drive suitable for Flywheel Energy Storage System - R.Anbarasu; Ph.D Thesis,

IIT, Delhi (1987).

[6] Electric and Hybrid Vehicles, Design Fundamentals - Iqbal Husain; CRC Press (2003).

[7] The key factors in the design and construction of advanced flywheel system and their

application to improve telecommunication power back up - Roger E Horner Interna-

guru murthy

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