homodyne olt-onu design for access optical networks

Homodyne OLT-ONU design for access opticalnetworks

Josep M. FabregaAdvisor: Josep Prat

Department of Signal Theory and CommunicationsUniversitat Politecnica de Catalunya - BarcelonaTech

[email protected]

May 17, 2010

J. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 1 / 59

Outline

Introduction

State of the art

Lock-In amplifier OPLL architecture

Advances in phase diversity architecturesKarhunen-Loeve series expansion phase estimationTime switched phase diversity

Simplified schemeDirect drive time switchingFuzzy data estimation

ONU and OLT architectures

Case studies

Conclusions


Introduction

I Novel multimedia applicationsI Voice over IPI Video on demandI HDTV

I User bit rate demand expected to be increasingI Nielsen law: bandwidth per user increments in a 50 % per yearI In 2020 each user would demand an average bandwidth of 1 Gb/s


FTTH roadmap and tendencies in PONs

Actual tendencies:I PON standardization bodies pushing towards high capacity

systems by increasing the aggregate bit rate (10 Gb/s)I ONU operates at a very high bit rate in the opto-electronics

transceivers just to use a small fraction of it (≈ 3 %)

High power consumption!New philosophy proposed, exploiting the pure WDM dimensionJ. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 4 / 59

Towards this new philosopy

Advantages DrawbacksIM-DD

I Simplicity I sensitivityI optical filters selectivity

Coherent

I Use of advanced modulation formatsI Electrical filtering for channel selectionI Detection amplitude, phase and polarizationI Linear transformation optical⇒ electricalI Increase of sensitivity

I Image frequency (heterodyne)I Phase noise (Homodyne)


Thesis objectives

I Identify current coherent systems architectures

I Propose advanced architectures to fit with the specifications of passiveoptical networks.

I Evaluate some of the advanced techniques by means of simulations andexperiments:

I Optical Phase-Locked LoopsI Lock-In amplified PLL

I Phase diversity receivers with zero intermediate frequencyI Karhunen-Loeve phase estimationI Time-switched phase diversity

I Implement a fully working transceiver prototype

I Research the published work on advanced access network architecturesand propose network scenarios to achieve ultra dense WDM operability

I Experimentally demonstrate the integration of the transceiver prototypein the more promising network schemes


Outline

Introduction

State of the art





Case studies

Conclusions


Reminding about homodyne receivers

eS (t) =√

PS exp(

j(ω0t + φS (t)

))ES (t) =

√PSejφS (t)

eLO (t) =√

PLO exp(

j(ω0t + φLO (t)

))ELO (t) =

√PLOejφLO (t)

I1(t) = <

∣∣∣∣∣√

1

2

(ES (t) + ELO (t)

)∣∣∣∣∣2

=<2

(PS + PLO ) + <√

PSPLO cos(φS (t)− φLO (t)

)

I2(t) = <

∣∣∣∣∣√

1

2

(− ES (t) + ELO (t)

)∣∣∣∣∣2

=<2

(PS + PLO )− <√

PSPLO cos(φS (t)− φLO (t)

)

Ip(t) = I1(t)− I2(t)

= 2<√

PSPLO cos(φS(t)− φLO(t)

)J. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 8 / 59

SNR and BER for BPSK and DPSK signals

I The bit error probability Pe for BPSK can be calculated as [1]:

Pe =12

erfc

(√SNR

2

)I For a coherent DPSK the bit error probability Pe is found [1]:

Pe =12

exp

(−SNR

2

)

Pe = 10−9 Pe = 10−3

BPSK 15.6 dB 9.8 dBDPSK 16 dB 10.9 dB

I 0.4 dB difference at Pe = 10−9

I 0.9 dB difference at Pe = 10−3


Phase errors in BPSK and DPSK

BPSK:

Pe =1

2

∫ π−π

p(φe)erfc

(√SNR

2cos(φe)

)dφe

=1

2√

2πσ2φe

∫ π−π

e−

φ2e

2σ2φe erfc

(√SNR

2cos(φe)

)dφe

being φe the phase error

DPSK:

Pe =1

2

∫ π−π

p(θ) exp

(−SNR

2cos2(θ)

)dθ

=1

2√

2πσ2θ

∫ π−π

e− θ2

2σ2θ exp

(−SNR

2cos2(θ)

)dθ

θ = φe(t0)− φe(t0 − Tb)

At SNR =∞ Pe is limited only by the phase error


BER limits and floorBER-floor is found to be [2]:

Pe =1√

2πσ2φe

∫cosφe<0

e−

φ2e

2σ2φe dφe =

2√2πσ2

φe

∫ +∞

π2

e−

φ2e

2σ2φe dφe

BER Standard deviation for 1 dB penalty BER-floor equivalent10−9 11◦ 2.31 · 10−16

10−3 19◦ 2.04 · 10−6

4.86 · 10−6 14.9◦ 10−9

2.54 · 10−1 28◦ 10−3


How to minimize this phase error?

I OPLLs: Several architectures proposedI Decision driven [3]I Costas [4]I Balanced [5]I Subcarrier modulated [6]

I Phase diversity with zero IF receiverI Analog mutiple differential detection [7]I Digital phase estimation:

I Wiener filter [8, 9]I Regenerative frequency dividers [10]I Viterbi & Viterbi [11, 12]

But many of them are not the cheap solutions we search for!

New solutions have to be proposed


How to minimize this phase error?

I OPLLs: Several architectures proposedI Decision driven [3]I Costas [4]I Balanced [5]I Subcarrier modulated [6]

I Phase diversity with zero IF receiverI Analog mutiple differential detection [7]I Digital phase estimation:

I Wiener filter [8, 9]I Regenerative frequency dividers [10]I Viterbi & Viterbi [11, 12]

But many of them are not the cheap solutions we search for!New solutions have to be proposed


Outline

Introduction

State of the art





Case studies

Conclusions


Principle of operation

I Low-cost PLL based on balanced OPLL [5]

I Search of maximum eye-opening using a lock-in amplifier

I Sinusoidally dither the phase of the local laser by a small amount

I Dithering makes possible the measurement of the phase error

I Amplitude modulated phase error after photodetection


Loop analysis and linearization

eS (t) =√

PS exp(j(ωt + φS (t)))

ES (t) =√

PS exp(jφS (t))

φS (t) = φD(t) + φNS (t)

eLO (t) =√

PLO exp(j(ωt + φLO (t)))

ELO (t) =√

PLO exp(jφLO (t))

φLO (t) = φC (t) + φNLO (t) +AKVCO

ωcsin(ωc t)

φLO (t) = φTLO (t) +AKVCO

ωcsin(ωc t)

V1(t) =1

2<RL

(PS + PLO − 2

√PSPLO cos(φS (t)− φLO (t))

)V2(t) =

1

2<RL

(PS + PLO + 2

√PSPLO cos(φS (t)− φLO (t))

)V3(t) = V2(t)− V1(t) = 2<RL

√PSPLO cos

(φS − φTLO (t) +

AKVCO

ωcsin(ωc t)

)= 2<RL

√PSPLO · η

Amplitude reduction factor: η = cos(φS − φTLO (t) +

AKVCOωc

sin(ωc t))



Amplitude reduction η can be expanded using trigonometric identities

η = cos(φS(t)− φTLO(t)

)cos

(AKVCO

ωcsin(ωc t)

)−

− sin(φS(t)− φTLO(t)

)sin(AKVCO

ωcsin(ωc t)

)AKVCOωc

is the dithering amplitude and can be enough small

η = cos(φS(t)− φTLO(t)

)(1−

A2K 2VCO

4ω2c

)+ (1)

+AKVCO

ωcsin(φS(t)− φTLO(t)

)sin(ωc t) + (2)

+A2K 2

VCO

4ω2c

cos(φS(t)− φTLO(t)

)cos(2ωc t) (3)



Spectral distribution of η

-

6

ωωc 2ωc

1 2 3

(2) =AKVCO


)sin(ωc t)

V4(t) ≈ 2<RL√

PSPLOAKVCO


)sin(ωc t)



V4(t) = 2<RL

√PSPLO

AKVCO

ωcsin(φS (t)−φTLO (t)

)sin(ωc t)

V5(t) = A sin(ωc t)

V6(t) =<RL

√PSPLOA2KVCO

ωcsin(φS (t)− φLO (t)

)V7(t) = 2<RL

√PSPLO · γ ∗ hf2(t) ∗ f (t) + A cos(ωc t)

γ = η ∗ hf1(t) =AKVCO

ωcsin(φS (t)− φTLO (t)

)sin(ωc t)

φLO (t) =

∫2KVCO<RL

√PSPLO · γ ∗ hf2(t) ∗ f (t)dt +

AKVCO

ωcsin(ωc t) + φNLO (t)

dφC (t)dt = K sin(φS(t)− φLO(t)) ∗ f (t) K =

A2<RL

√PSPLOK 2

VCOωc

Dithering amplitude adds a small controlable penalty

σ2d =

A2K 2VCO

2ω2c⇒ σ =

√σ2

AN + σ2PN + σ2

d =

√√√√ ωn

2SNRBe

(ξ +

1

4ξ2

)+

2π∆ν

2ξωn+

A2K 2VCO

2ω2c


Simulations

The Lock-In amplified OPLL performances were evaluated by means ofcomputer simulations, aiming to analyze:

I Phase noise cancellation

(Matlab/Simulink)

I Step response performances

(Matlab/Simulink)

I Comparison with other loops

(VPI TransmissionMaker)

The system was designed to operate at:

I Phase error BW of 200 MHz (possible highspeed phase variations [13])

I Dithering frequency of 700 MHz

I Dithering amplitude around 50 mrad (2.86◦ ⇒ σd = 2◦)

I Filters used based on Butterworth approximations (2.5 ns average delay)

I The filter f (t) was a PI control, leading to a second order PLL

Parameters to optimize: Damping factor, natural frequency and loop delay


Simulations

The Lock-In amplified OPLL performances were evaluated by means ofcomputer simulations, aiming to analyze:

I Phase noise cancellation (Matlab/Simulink)

I Step response performances (Matlab/Simulink)

I Comparison with other loops (VPI TransmissionMaker)

The system was designed to operate at:

I Phase error BW of 200 MHz (possible highspeed phase variations [13])

I Dithering frequency of 700 MHz

I Dithering amplitude around 50 mrad (2.86◦ ⇒ σd = 2◦)

I Filters used based on Butterworth approximations (2.5 ns average delay)

I The filter f (t) was a PI control, leading to a second order PLL

Parameters to optimize: Damping factor, natural frequency and loop delay


Phase noise simulations

Linewidth 1 ns delay 5 ns delay 10 ns delay1 MHz 11.74◦ 15.05◦ 18.23◦

2 MHz 17.93◦ 23.35◦ 28.24◦

3 MHz 23.98◦ 28.34◦ 35.08◦

4 MHz 28.29◦ 34.69◦ 40.35◦

5 MHz 32.23◦ 36.42◦ 53.82◦

I 10−9 1 dB penalty near 1 MHz and 1 ns loop delay

I 10−9 floor achieved with 1 MHz loop delay near 4.5 ns

I 1 dB penalty of a 10−3 BER for 2 MHz with 1 ns loopdelay, and 1MHz with 10 ns


Time response simulations


Comparison with other loops

I Compared to other oPLL architectures:I Costas [4]I Balanced [5]

I Subcarrier modulated [6]I Damping factor set to 9 to assure overdampingI 10 ns loop delay (eq. 20 cm of fiber)I Table shows 1 dB penalty BER tolerances

Linewidth tolerance BER 10−9 Linewidth tolerance BER 10−3 Pull in range Hold in rangeBalanced 420 kHz 1.2 MHz 19 MHz 1.28 GHzCostas 1.15 MHz 2.65 MHz 72 MHz 2.55 GHzSCM 1.35 MHz 2.75 MHz 176 MHz 7.68 GHz

Lock-In amplified 675 kHz 3.1 MHz 20 MHz 896 MHz

Lock-In amplifier OPLL is a competitive low-cost solution!


Comparison with other loops

I Compared to other oPLL architectures:I Costas [4]I Balanced [5]

I Subcarrier modulated [6]I Damping factor set to 9 to assure overdampingI 10 ns loop delay (eq. 20 cm of fiber)I Table shows 1 dB penalty BER tolerances

Linewidth tolerance BER 10−9 Linewidth tolerance BER 10−3 Pull in range Hold in rangeBalanced 420 kHz 1.2 MHz 19 MHz 1.28 GHzCostas 1.15 MHz 2.65 MHz 72 MHz 2.55 GHzSCM 1.35 MHz 2.75 MHz 176 MHz 7.68 GHz

Lock-In amplified 675 kHz 3.1 MHz 20 MHz 896 MHz

Lock-In amplifier OPLL is a competitive low-cost solution!J. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 22 / 59

Experiments and discussion

I LO standard DFB laser (833 kHz)@ 1544.07 nmI Locking observed by tuning one of the lasers until

20 MHz differenceI Hold in range was found to be 868.24 MHzI Laser output was fusion spliced to the photodetector

input to obtain 10 ns delayI Phase error standard deviation measured to be of

11.49◦ using the procedure described in [13] for a200 MHz of integration bandwidth

Lock-In OPLL demonstrated to be competitive at low cost, but it is mainly limited by the loop delay


Outline

Introduction

State of the art





Case studies

Conclusions


Karhunen-Loeve series expansion phase estimation

I Phase estimation based on Karhunen-Loeve expansionI For this case it is equivalent to Fourier seriesI Approximation to the phase noise spectrum

w(t) =∞∑

n=1

cnϕn(t)

ϕn(t) =

√2

Tsin(ωn t)

ωn =

√2π∆ν

λn=

(2n + 1)π

2T

cn =

√2

T

∫ T

0w(t) sin(ωn t)

λn = E{c2n} =

8T 2∆ν

(2n + 1)2π

w(t) = c>ϕ =(

c1 c2 c3 . . . cn)

ϕ1ϕ2ϕ3...ϕn

I Process block by block, highly correlatedI In a block, phase noise will have a variance of 2π∆νTI Coeffs are randomly initialized and calculated by LMSI Eigenfunctions are stored in a DSP memoryI Truncate the series at M = 5


Algorithm performances and discussion

I BPSK data stream simulated running at 10 Gb/sI Linewidths ranging from 100 kHz to 10 GHz

I 220 = 1048576 symbols and 16 samples per symbolI blocks of 16 samples (one symbol)I Compared to Wiener filter with a lag of 10 symbols [8]

I 1 dB penalty for 10−9 BER (11◦)

I Wiener filter: 0.6 %⇒ KL expansion: 3.9 %

I 1 dB penalty for 10−3 BER (19◦)

I Wiener filter: 1.9 %⇒ KL expansion: 11 %

I BER-floor at 10−9 (14.9◦)

I Wiener filter: 1.3 %⇒ KL expansion: 6.6 %

I BER-floor at 10−3 (28◦)

I Wiener filter: 5 %⇒ KL expansion: 25 %


Time switched phase diversity

I Phase diversity by switching from I to Q component at each bitI Less components duplicity than standard phase diversity (lower cost)I 3 dB penalty respect to an ideal systemI Several approaches presented


Simplified scheme and phase noise analysis

Vout =4<2R2

L PSPLO

2d(t)

[cos(φe(t)− φe(t − Tb)

)−

− cos(φe(t) + φe(t − Tb)

)+

+ cos(φe(t − Tb/2)− φe(t − 3Tb/2)

)+

+ cos(φe(t − Tb/2) + φe(t − 3Tb/2)

)]

Vout =C2

2d(t)

[1− sin2

(∆φ1(t) + ∆φ2(t)

2

)− sin2

(∆φ2(t) + ∆φ3(t)

2

)+

+1

2cos(2φe(t)

)[2 cos

(∆φ1(t) + ∆φ2(t)

)sin2

(∆φ1(t) + ∆φ3(t)

2

)+ sin

(∆φ1(t) + ∆φ2(t)

)sin(

∆φ1(t) + ∆φ3(t))]

+

+1

2sin(2φe(t)

)[2 sin

(∆φ1(t) + ∆φ2(t)

)sin2

(∆φ1(t) + ∆φ3(t)

2

)− cos

(∆φ1(t) + ∆φ2(t)

)sin(

∆φ1(t) + ∆φ3(t))]]

∆φ1(t) = φe(t)− φe

(t −

Tb

2

)∆φ2(t) = φe

(t −

Tb

2

)− φe(t − Tb) ∆φ3(t) = φe(t − Tb)− φe

(t −

3Tb

2

)I ∆φ1(t), ∆φ2(t) and ∆φ3(t); are independent identically distributed Gaussian random processes with zero meanI Terms containing φe(t), tend to 0 as they are multiplied by sine terms near 0I The only remaining terms will be near 1I More detais are given in the thesis


Experiments and discussion

I Total linewidth from 350 kHz to 30 MHzI Bitrate 1 Gb/sI TX laser output power 9 dBmI MZM modulator biased at nullI 10−3 BER-floor found at 18 MHz linewidthI -38.7 dBm sensitivity obtained at 10−9 BER

I -44.3 dBm sensitivity obtained at 10−3 BERI BER-floor dotted line is a numerical model assuming

∆φ3(t) = 0 (more details are provided by the thesis)

Results published at ECOC 2006 [14]→ tolerance record at lower cost!


Frequency drift

Effects of freq. difference (λTX − λLO)

Vout ≈C2

2d(t)2 cos

(2πfdTb

)

Pe =12

exp

(−SNR · cos2

(2πfdTb

)2

)

I 10−9 1 dB penalty point at 7.4 %

I 10−3 1 dB penalty point at 7.5 %

I Experimental operating limit at60 MHz, leading to 10−3 BER


Channel spacing I

I Evaluate adjacent channel interferenceI Photodetected spectrum doubled because of dithering

eS (t) =N∑

i=−N

√PS exp(j(ωt + 2πDit + φSi (t)))

eLO (t) =√

PLO exp(j(ωt +π

2p(t)))

Ip(t) =N∑

i=−N

2<√

PSi PLO cos(φSi (t)+2πiDt−

π

2p(t))

+n(t)

First simple model assuming gaussian statistics

penalty(dB) = −10 · log(

1−SNR

SIR

)

Gp(f ) =N∑

i=−N

4<2PS0PLOT

4

[sinc2

(πT (f − iD)

2

)]+ N(f )

SIR ≈

∫ +∞−∞ |H(f )|2

[sinc2(πTf

2)]

df∫ +∞−∞ |H(f )|2

[sinc2(πT (f−D)

2)]

df

Model limited by the fast varying phase of interference


Proposed mathematical model for channel spacing

θ(t) = φS1(t) + 2πDt +π

2p(t)

X ∝ d(

1 +1√SIR

cos(θ))

Pe(θ) =12

exp(− SNR

2

(1 +

1√SIR

cos(θ)

)2)

Pe =1

4π

∫ +π

−πexp

(− SNR

2

(1 +

1√SIR

cos(θ)

)2)

d θ


Experiments, results and discussion

I Experimental prototype assembledI Three external cavity tuneable lasers were usedI Total laser linewidth was 300 kHzI MZM modulator biased at nullI 27 km standard G-652 fiber spool

I Parameter measured: 10−9 BER sensitivity penaltyI Channel spacing ranged from 1 GHz to 6 GHzI Bit rate was 1 Gb/s

I 1 dB penalty found at around 3 GHz for both, Gaussianmodel and the accurate approach

I For the ideal system the 1 dB penalty channel spacing isof only 1.25 GHz.

I Experimental results:I 3 GHz spacing found for a 1 dB penalty

I 3 dB point found between 1.5 GHz and 2 GHz

Experimental results match theoretical results3 GHz channel spacing feasible at 1 dB penalty


Direct drive time switching I

Ipd (t) = 2<√

PSPLO cos(φS (t)− φLO (t) + φd (t) +

π

4+

AcγKLO

2πfcsin(2πfc t)

)

Im(t) = 2<2PSPLOd(t)[

cos(θ1(t)

)−

∞∑n=−∞

Jn(β) sin(θ2(t)

)]

β =AcγKLO

πfcθ1(t) = φe(t)− φe(t − Tb)

θ2(t) = φe(t) + φe(t − Tb) + 2πnfc t

In case γ is near√

2, β takes a value that makes J0(β) very low


Direct drive time switching II

Id (t) = Im(t) + Im

(t −

Tb

2

)= 2<2PSPLOd(t)

[cos(θ1(t)

)+ cos

(θ1

(t −

Tb

2

))− 2J2(β) sin

(θ2(t)

)− 2J2(β) sin

(θ2

(t −

Tb

2

))]= 2<2PSPLOd(t)

[cos(∆φ1 + ∆φ2) + cos(∆φ2 + ∆φ3) +

+2J2(β) sin(

2φe(t)± 2π2fc t −3∆φ1 + 2∆φ2 + ∆φ3

2

)cos(

∆φ1 + ∆φ2

2

)]

∆φ1(t) = φe(t)− φe

(t −

Tb

2

)∆φ2(t) = φe

(t −

Tb

2

)− φe(t − Tb) ∆φ3(t) = φe(t − Tb)− φe

(t −

3Tb

2

)J. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 35 / 59

Direct drive time switching III

I When phase noise becomes important, the cos(∆φ2 + ∆φ3) and cos(∆φ1 + ∆φ2) arguments growI The ideal receiver bandwidth of such a diversity receiver is 1.5 times data rateI Noise terms contributed by the Bessel function are over a carrier running at a frequency two times the data rate

I The phase noise will vary more slowly than data rate

Id (t) ≈ <2PSPLOd(t)[

cos(∆φ1 + ∆φ2) + cos(∆φ2 + ∆φ3)]

Sensitivity example detecting on Im(t) (no delay & add = NDAD) and on Id (t) (DAD)

I When phase noise can be neglected performing decision on Im(t) is slightly better than deciding on Id (t)I Low linewidth can be cancelled by the differential detection itself and the interference of the Bessel function terms


Simulations

I DPSK configuration with Monte-Carlo BER estimationI Linewidth per bit rate tolerance was evaluated at 1 Gb/sI Linewidths from 10 MHz to 100 MHz with infinite SNRI The maximum tolerated linewidth at 10−3 BER is 2%

bitrate for γ = 1I When γ ≈

√2 the maximum tolerated linewidth is

3.4% bitrate

I Sensitivity penalty was obtainedI Laser linewidth was disabled

I At 10−9 BER, a penalty is observed when replacing

square signal by the sinusoid:I 0.5 dB when γ =

√2

I 0.7 dB when 1 for a 10−9 BERI Almost no penalty is observed at 10−3 BER

Simulations confirm the theoretical behaviorAn optimum is obtained at γ =

√2


Experiments

I Experimental prototype developed and testedI bitrate was 1 Gb/sI Total linewidth was 300 kHz (2x150 kHz)

I γoptimum found near√

2I Frequency drift tolerance 1 dB penalty at 75 MHz

(matching the theoretical results)

I Potential candidate to a true low-cost PONtransceiver when laser is directly driven


Experiments

I Experimental prototype developed and testedI bitrate was 1 Gb/sI Total linewidth was 300 kHz (2x150 kHz)

I γoptimum found near√

2I Frequency drift tolerance 1 dB penalty at 75 MHz

(matching the theoretical results)I Potential candidate to a true low-cost PON

transceiver when laser is directly driven


Fuzzy data estimation I

I Heuristic method to estimate dataI Differential detection needed to separate I & Q

I Universe of discourse classified into 5 membershipfunctions

I The membership of the digitzed signal after the ADC isdetermined by means of a look-up table

I/Q Neg. Mod. Null Mod. Pos.Neg. Pos.

Negative Zero Zero Zero Zero -Mod. Neg. Zero Zero Zero - One

Null Zero Zero - One OneMod. Pos. Zero - One One OnePositive - One One One One


Fuzzy data estimation II

I After extensive numerical simulations, the optimum method isfound to be the so called Method of Maximum (MoM)

I Monte-Carlo bit error counting performances evaluationI Linewidth from 22.5 kHz to 100 MHzI Bit rate was 1 Gb/sI Improvement from 2.8% bit rate to 3.5% bit rate @ 10−3 BER-floorI This is a 26 % improvement of the bit rate per linewidth product


Outline

Introduction

State of the art





Case studies

Conclusions


Summary of the performances

Phase noise

Technique Linewidth Penalty Required key Complexitytolerance component

Decision-drive loop 5 MHz 0 dB 90◦ hybrid HighCostas loop 4.9 MHz 0 dB 90◦ hybrid Medium/High

Subcarrier loop 5.1 MHz 0 dB 90◦ hybrid HighBalanced loop 2.4 MHz 2 dB Optical coupler LowLock-In loop 6.4 MHz 1 dB Optical coupler Low

Full phase diversity 5% bitrate 0 dB 90◦ hybrid MediumTime-switch (Scrambler) 1.8% bitrate 4 dB Phase modulator Medium

Time-switch (Direct drive) 3.4% bitrate 4 dB High-chirp laser Low

Penalty is respect to an ideal system whereas tolerance is for a BER-floor of 10−3

Three main approaches: Optical phase locked loop, full phase diversity and time-switched diversity

Polarization mismatch

Local control Polarization diversity Polarization switchingPenalty 0 dB 0 dB 3 dB

Key component Polarization actuator Pol. beam splitter Pol. scrambler/switchResponse time 1 ms – 1 s < 10 µs < 10 µs

Complexity High Med./high Low (if placed at CO)

Several transceiver architectures are going to be discussed in the following slides:I The targeted modulation format is DPSK/BPSK, although other multilevel modulations can be usedI Channel selection is performed by tuning the local laser to the right wavelength and filtered by the electrical filters


Transceiver architectures proposedA

B

C

D

E

F

G

H

I

I Architectures A, C, E and G are intended for BPSK modulationI DPSK modulation format has to be used in architectures B, D, F, and II Polarization is managed at OLT for architectures A, B, E, F, G, H and II In transceivers C and D a PBS used for achieving polarization diversity

I For digital approaches (A, C, E, G), Inside digital I and Q post-processing several basic operations are performed:I phase estimationI frequency estimation and controlI data estimation

I polarization switching combinationJ. M. Fabrega (UPC-BarcelonaTech) Homodyne OLT-ONU design May 2010 43 / 59

Transceiver comparison

Arch. Phase handling Polarization handling Processing Sens. penalty Linewidth tolerance CostA 90◦ hybrid Switch at CO Digital 3 dB 5 MHz Med./HighB 90◦ hybrid Switch at CO Analog 4 dB 5 MHz Med./HighC 90◦ hybrid PBS Digital 0 dB 5 MHz Very highD 90◦ hybrid PBS Analog 1 dB 5 MHz Very highE Switch (Scr.) Switch at CO Digital 6 dB 1.8 MHz MediumF Switch (Scr.) Switch at CO Analog 7 dB 1.8 MHz MediumG Switch (Dir.) Switch at CO Digital 6 dB 3.4 MHz LowH Switch (Dir.) Switch at CO Analog 7 dB 3.4 MHz LowI OPLL Switch at CO Analog 4 dB 675 kHz Low

I Bitrate 1 Gb/s

I Architecture with 90◦ hybrids a PBS and DSP (C):I No additional penalty with respect to an ideal systemI Achieves high linewidth tolerance

I It is costly because it implies the duplication of many components needed

I Direct-drive time-switching (G):I High penalty (6 dB)I High linewidth toleranceI Reduced complexity and cost

I Requires a fully engineered laser capable to be phase modulated

I OPLL approach (I):I Low complexityI Lower linewidth tolerance (525 kHz)I Delay associated to the optical path length

I Local laser should be embedded with the optical reception front-end


Outline

Introduction

State of the art





Case studies

Conclusions


Case studies

Two cases of future deployment were tested in the laboratory:

I Subband WDM tree PON, featuring wavelength grooming [15]

I Ring-tree ultra-dense WDM-PON, with transparent remote nodes [16]

Both networks are based on the ultra-dense WDM concept, aiming to giveservice to a high number of users (around 1000), at very high speed (1 Gb/s)


Subband WDM tree PON

I 4 GHz channel spacing and 1 Gb/s data rateI 32 channels accommodated in an ITU-T G.694.1

100-GHz D-WDM channelI Serve 40 x 32 = 1280 users⇒ more than 1 Tb/sI 25 km fiber spool simulated the access trunk fiberI Losses at the AWG were measured to be 6.47 dBI 1:32 power splitter, adding 16 dB lossesI Total network losses were measured to be 27.67 dBI 9 dBm optical output power at CO

I −38.7 dBm of sensitivity (BER=10−9)I Power budget calculated to be 47.7 dB


Ring-tree ultra-dense WDM PON

I Totally passive and transparentI Simple and resilient architecture


Experimental evaluation

I Network designed to offer connectivity to 1024 users

I 4-node configuration

I 8 secondary trees with 1:128 splitting factor

I 4 GHz channel spacing

I Three different cases were investigated: RN1, RN2 and RN4


Results and discussion

BER = 10−9 BER = 10−3

RN1 RN2 RN4 RN1 RN2 RN4Sensitivity −43 dBm −41.3 dBm - −49.1 dBm −49.3 dBm −49.1 dBm

Link Losses 39.4 dB 41 dB 44.2 dB 39.4 dB 41 dB 44.2 dBPower Budget 42.9 dB 41.2 dB - 49 dB 49.2 dB 49 dB

I In normal operation (BER = 10−9), the maximum power budget reached is 41.2 dB arriving to RN2I In resilient mode FEC codes are used to overcome the possible fiber cutI Using FEC codes BER = 10−3 is operable and RN4 can be reached, featuring a power budget of 49 dB


Outline

Introduction

State of the art





Case studies

Conclusions


Conclusions I

I The overall goals of this study were homodyne OLT and ONUdesigns for upgrading the current standard PONs

I Several coherent detection techniques have been proposed,improving the performances of the receivers that shape thecurrent state of the art:

I A novel OPLL has been analyzed and prototypedI Karhunen-Loeve series expansion phase estimationI Fuzzy logic data estimationI Time-switched phase diversity

I A very simple and robust architecture has been prototypedfeaturing time-switching phase diversity

I Cost effective although some optical elements should be integratedI A similar structure has been recently integrated in InP substrate [17]I Tolerating linewidths up to 1.8% of the bitrate (BER-floor 10−3)I 3 GHz channel spacing for 1 dB penalty at 10−9 BER


Conclusions II

I Several transceiver architectures have been proposed anddiscussed

I Trade-off between performances and cost is difficult to overcomeI Time-switched diversity transceiver has been implemented

I Upgrading of PON architectures has been discussed forimplementing full ultra-dense WDM networks

I Laboratory testbeds have been developed for two network conceptsI The proposed topologies have been demonstrated to be feasible,

achieving transmission of up to 1 Gb/s in links higher than 25 km


Future lines

Some improvements to be achieved, resulting in a step forward:I Compact coherent transceiver

I State of polarization mismatch between local oscillator andreceived signal

I Careful design of the modulation formats to be usedI Full bidirectionality over a single fiber

I Rayleigh backscatteringI Light reflections

I Spectrum managementI Spectral efficiency maximizationI Wavelength monitoring, control and stabilization


Thank you!!Time for questions...


Bibliography I

[1] K.-P. Ho.Phase-Modulated Optical Communication Systems.Springer-Verlag, 2005.

[2] K. Kikuchi and et al.Degradation of bit-error rate in coherent optical communications due to spectral spread of the transmitter and the localoscillator.IEEE / OSA Journal of Lightwave Technology, 2(6), December 1984.

[3] L. G. Kazovsky.Decision-driven phase-locked loop for optical homodyne receivers: performance analysis and laser linewidth requirements.IEEE / OSA Journal of Lightwave Technology, 3:1238, 1985.

[4] S. Norimatsu, K. Iwashita, and K. Sato.Psk optical homodyne detection using external cavity laser diodes in costas loop.IEEE Photonics Technology Letters, 2(5), 1990.

[5] L. G. Kazovsky.Balanced phase-locked loops for optical homodyne receivers: performance analysis, design considerations, and laserlinewidth requirements.IEEE / OSA Journal of Lightwave Technology, 4:182, 1986.

[6] S. Camatel, V. Ferrero, and P. Poggiolini.2-psk homodyne receiver based on a decision driven architecture and a sub-carrier optical pll.In Proceedings of the Conference on Optical Fiber Communication and the National Fiber Optic Engineers Conference,2006 (OFC/NFOEC 2006), Anaheim (CA), March 2006.

[7] Y. H. Cheng, T. Okoshi, and O. Ishida.Performance analysis and experiment of a homodyne receiver insensitive to both polarization and phase fluctuations.IEEE / OSA Journal of Lightwave Technology, 7:368–374, 1989.


Bibliography II

[8] M. G. Taylor.Phase estimation methods for optical coherent detection using digital signal processing.IEEE / OSA Journal of Lightwave Technology, 27:901–913, 2009.

[9] M. G. Taylor.Accurate digital phase estimation process for coherent detection using a parallel digital processor.In Proceedings of 31th European Conference on Optical Communications (ECOC 2005), Glasgow (Scotland), September2005.

[10] R. Noe.Phase noise-tolerant synchronous qpsk/bpsk baseband-type intradyne receiver concept with feedforward carrier recovery.IEEE / OSA Journal of Lightwave Technology, 23(2), 2005.

[11] D. Van den Borne, C. R. S. Fludger, T. Duthel, T. Wuth, E. D. Schmidt, C. Schulien, E. Gottwald, G. D. Khoe, andH. de Waardt.Carrier phase estimation for coherent equalization of 43-gb/s polmuxnrz-dqpsk transmission with 10.7-gb/s nrz neighbours.In Proceedings of 33th European Conference on Optical Communications (ECOC 2007), Berlin (Germany), September2007.

[12] M. Seimetz.Laser linewidth limitations for optical systems with high-order modulation employing feed forward digital carrier phaseestimation.In Proceedings of the Conference on Optical Fiber Communication and the National Fiber Optic Engineers Conference,2008 (OFC/NFOEC 2008), San Diego (CA), March 2008.

[13] L. G. Kazovsky.A 1320- nm experimental optical phase-locked loop : Performance investigation and psk homodyne experiments at140mb/s and 2gb/s.IEEE / OSA Journal of Lightwave Technology, 8(9), 1990.


Bibliography III

[14] J. M. Fabrega and J. Prat.Homodyne receiver implementation with diversity switching and analogue processing.In Proceedings of 32th European Conference on Optical Communications (ECOC 2006), Cannes (France), September2006.

[15] C. Bock, J. M. Fabrega, and J. Prat.Ultra-dense wdm pon based on homodyne detection and local oscillator reuse for upstream transmission.In Proceedings of 32th European Conference on Optical Communications (ECOC 2006), Cannes (France), September2006.

[16] J. M. Fabrega and J. Prat.Ultra-dense, transparent and resilient ring-tree access network using coupler-based remote nodes and homodynetransceivers.In Proceedings of International Conference on Transparent Optical Networks ICTON’09, Ponta Delgada (Azores, Portugal),July 2009.

[17] A. Ramaswamy, L. A. Johansson, J. Klamkin, C. Sheldon, H. F. Chou, M. J. Rodwel, L. A. Coldren, and J. E. Bowers.Coherent receiver based on a broadband phase-lock loop.In Proceedings of the Conference on Optical Fiber Communication and the National Fiber Optic Engineers Conference,2007 (OFC/NFOEC 2007), Anaheim (CA), March 2007.


homodyne olt-onu design for access optical networks

Technology

phase diversityi

homodyne receiversest

gbsi onu

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advanced architectures

advanced techniques

rate demand

network scenarios