Implementation of the Ultra-Low Power Load-Independent LC VCO

Denys Martynenko, Gunter Fischer, Oleksiy Klymenko IHP, Frankfurt (Oder), Germany

e-mail: martynenko@ihp-microelectronics.com

Abstract — In this paper the influence of the external load on the key parameters of the commonly used cross-coupled LC VCO architecture is investigated. It is shown, that even high ohmic external load decreases the impedance of the resonator, limits the frequency tuning range, degraded the phase noise and increases the power dissipation of the structure. Additionally, based on the carried out analyses, the novel load-independent LC VCO architecture is implemented and characterized. In the presented VCO topology, the resonator is isolated from the external load, by connecting the VCO outputs in the common collector configuration. As a result, the oscillator with the power consumption of 540 μW from the 1.8 supply voltage is achieved. The measured frequency tuning range of the VCO lies between 6.30 GHz and 8.59 GHz and the phase noise is better than -100 dBc/MHz at the frequency offset of 1 MHz within entire tuning range.

Keywords-component; ultra-low power LC VCO; BiCMOS VCO; UWB; IEEE 802.15.4a

I. INTRODUCTION LC VCOs are commonly used in the RF integrated circuit

due to their easy implementation and good phase noise characteristics [1-5]. Typically in order to achieve a low power oscillator with a good phase noise performance, the resonator of the oscillator is designed for the highest possible quality factor in combination with the wide tunable varactors. However, the low power or ultra-low power VCO design for tunable frequency range of wider than 25% requires special design solutions, typically, due to the overloading the resonator with the high capacitive load or the limitation of the tunable capacitive range of the single varactor.

In the literature, several realizations of wideband low power VCO structures are described. In [1], the authors use two wideband VCOs in order to provide the required 40% tuning range. Although the concept is attractive because of the low power consumption and the simple implementation, it suffers from the enlargement of area occupation. In [2] a switchable inductor in addition to a varactor is used in order to increase the tuning range. As a result, high phase noise or high power dissipation is obtained. A more complex concept is presented in [3]. The authors describe the wideband VCO with switchable active core and varactor size. The complex digital control and VCO core implementation result in a slightly increased area occupation.

- G

Cvar

Gtank

Lind Rs

Fig. 1. The simplifies circuit diagram of the –Gm LC VCO. Nevertheless, the scalable power consumption with the frequency and wideband tuning range make this architecture attractive for low power wideband applications. Other, widely used realizations of the wideband VCOs are presented in [4]-[6]. The frequency tuning range is achieved by a fixed value of the inductor and a bank of the switchable capacitors or switchable varactors. This realization is very convenient for wideband VCO design, because it is simple, robust, and does not increase the area occupation. However, the utilization of a big amount of switchable capacitors may have a negative effect on the power consumption or phase noise due to the overloading the LC tank.

Consequently, the most challenging issue of wideband low power VCO design is to support the stable oscillation with the low power dissipation within the wide tuning range, independently of the strong variation of the internal resonator capacitance and the resonator quality factor. Besides the strong variations of resonator parameters which are caused by the internal VCO components, an external load is also critical and should be taken into account. It introduces not only an extra capacitance but also degrades the impedance of the resonator.

This paper analyses the influence of the external load on the power dissipation, quality factor, phase noise and the tuning range of the typical cross-coupled VCO architecture. In addition, the novel design of an ultra-low power LC VCO with isolated resonator is proposed. The proposed topology overcomes the negative effects introduced by the external load and, as a result, the VCO with the power dissipation of 540 µW from the 1.8 V supply voltage and measured tunable frequency range of 31% is achieved. The measured phase noise is better than -100 dBc/Hz at 1MHz offset frequency within whole tuning range.

978-1-4673-3119-7/12/$31.00 ©2012 IEEE 27

II. PROBLEM STATMENT The widely used LC VCO architecture (simplified circuit

diagram is shown in Fig.1) which is based on the cross coupled transistor pair and the parallel-tuned resonator is taken for investigation. The high quality factor of the resonator and large value of the inductance provide the conditions for the low power, low phase noise operation. The voltage variable capacitance supports the wide frequency tuning range of the VCO. However, the negative influence of the external load can strongly limit the performance of the VCO as it will be shown further in the paper.

A. Power dissipation The power dissipation of the unloaded LC VCO is defined

by the losses of the resonator components (inductor and varactor) and the value of the negative impedance which is generated by the cross-coupled transistor pair for determined tail current. With respect to the energy conservation for parallel-tuned resonator, the maximum energy stored in the capacitor has to be equal to the maximum energy stored in the inductor, if the losses of the resonator are compensated. Thus, the losses of the LCR tank (Fig. 2a) can be expressed with the equations [7]:

222

222 121

21

peakind

LpeakVVloss V

LR

VCRP S

s ωω == ,

where fπω 2= , SS VL RR , are the series resistances of the

inductor and varactor, respectively. peakV is the peak voltage

across the resonator, indL and VC are the values of the inductance and varactor capacitance, respectively. Let us rewrite the equations for the power loss of the resonator using the quality factors of the passive components:

SL

indind R

LQ ω=

VLV CR

QS

ω1=

,

where Qind and RLs are the quality factor and serial resistance of the inductor, QV and RVs are the quality factor and serial resistance of the varactor, respectively. Now, the power loss of the resonator can be represented as:

22

22

1211

11

21

11211

21

peakindind

Vpeak

indind

Vind

peakindind

peakind

Ls

indloss

VQL

CVQ

LCL

VQL

VLR

LP

==

===ωωω

,

Lind Rs Rs Lind

CV Rv

CTR CTR

CVRv

Lind Lind

RvP RvP

CV

CTR CTR

CV

RindPRindP

(a) (b)

Fig. 2. Simplified representation of the serial (a) and parallel (b) resonator.

Lind Lind

RvP RvP

CV

CTR CTR

CV

RindPRindP

Cext Cext

Rext Rext

Fig. 3. Simplified representation parallel resonator including the parasitic components of the external load.

In the same way:

22 121

21

peakVind

VpeakVVsVloss V

QLCVCRCP == ωω ,

From the equation it is seen, that the power losses of the resonator are reversely proportional to the inductance and to the quality factor of the resonator and directly proportional to the capacitance and to the voltage swing. The ratio C/L depends on the required VCO frequency tuning range and the voltage swing across the resonator is important in term of the phase noise. Therefore, the most effective way to decrease the power dissipation is to increase the R/L [7].

The short analysis of the power dissipation which is shown above includes just the internal components of the resonator. However, the resonator of the VCO is usually connected to an external load (buffers, dividers etc) and any extra resistance connected in parallel to the parallel-tuned resonator causes the drop of the unloaded Q-factor to a smaller value. Thus, external load, although having a high impedance, still reduces the impedance of the resonator and hence the Q-factor. Fig. 3 shows the equivalent circuit for the parallel-tuned LC resonator loaded by the external load with the impedance of Rext. Now, the unloaded quality factors of the passive components of the resonator could be expressed in term of the dynamic resistances RLp and RVp [8]:

ωind

Lind L

RQ P

U=

28

VVV CRQPU

ω=,

where PU Lind RQ , are the unloaded quality factor and the

dynamic resistance of the inductor and PU VV RQ , are unloaded

quality factor and the dynamic resistance of the varactor. Thus, Rext connected in parallel to

PLR and PVR decreases the Q-

factors of the passive components to:

ext

L

ind

ind

extL

extL

ind

extLind

RR

QL

RRRR

LRR

QP

ULP

P

P

L

+=

+==

1ωω,

ext

L

VVextVV

RR

QCRRQ

P

U

PU

+==

1ω ,

where LindQ and

LVQ are the loaded quality factor of the inductor and varactor, respectively. Now, let us write the equation for the power losses of the loaded LC resonator using the equation for the loaded Q-factor of the passive components:

21

21

peakind

ext

L

ind

Vloss V

QRR

LCP

U

P+=

21

21

peakV

ext

L

ind

Vloss V

QRR

LCP

U

P+=

Thus, the external load increases the losses of the resonator on

ext

orVL

R

Rpp )(1+ and causes additional increasing of the power

dissipation. Therefore, minimizing the influence of the external load on the quality factor of the resonator components is of great importance for the VCO design for ultra-low power applications.

B. Frequency tuning range The tunable frequency range of the unloaded LC VCO can

be calculated using the equations:

min. oscillation frequency:

( )parVind CCLf

+=

max2

1min π

max. oscillation frequency:

( )parVind CCL

f+

=min

21

min π

where CVmax and CVmin are the maximum and minimum capacitances of the varactor and Cpar is the internal parasitic capacitances of the VCO (such as capacitances of the transistor cross-coupled pair, inductor and etc.). However, any extra capacitance (including the capacitance of the external load (Fig. 3)) connected in parallel to the parallel-tuned resonator causes the tuning range reduction and shifts the VCO oscillation to the lower frequencies. That can be expressed with the equations: min. oscillation frequency:

( )extparVind CCCLf

++=

max2

1min π

max. oscillation frequency:

( )extparVind CCCLf

++=

min2

1max π

where Cext is the parasitic capacitance caused by the external load. Since, the extra capacitance has to be compensated in order to provide the oscillation in the required frequency range, then the inductance of the resonator or the tuning capacitance of the varactor has to be decreased. Any of those changes decreases the tuning range of the VCO. However, the inductance reduction increases the power dissipation and degrades the phase noise of the VCO. Therefore, to adjust the varactor value is more convenient.

In addition, the value of the parasitic capacitance of the external load depends on the application (number of required buffers after the VCO, etc.). Therefore, the C/L ratio of the resonator has to be adjusted for different external loads individually. This basically means that the VCO requires redesign each time when the external load is changed. Consequently, the parasitic capacitance of the external load introduces significant challenges in design and affects not just the frequency tuning range, but also the power dissipation of the VCO.

C. Phase noise The phase noise spectral density of the oscillator in the

2

1f

region can be predicted using the Lesson’s phase noise

model [9]:

( )⎥⎥⎦

⎤

⎢⎢⎣

⎡⎟⎟⎠

⎞⎜⎜⎝

⎛Δ

+=Δ2

0

212

ωωωUS QP

FkTL

where Ps is the signal power across the resonator and QU is the unloaded quality factor of the resonator. The quality factor of the unloaded parallel LCR resonator can expressed as:

ind

VVindU L

CRRQPP

= ,

29

if PP Vindres RRR = , than the loaded quality factor of the

resonator LQ is:

ext

res

U

ind

VextresL

RR

QLCRRQ

+==

1,

Further, the signal power across the resonator could be expressed with the equation:

extrestailS RRIP 2= ,

where Itail is the current through the resonator and Rres is the tank resonance impedance. Accordingly to the Lesson’s equation, the VCO phase noise is improved if the quality factor and the resonance impedance of the LC tank are maximized. However, with respect to the previous analyses, the external load of the VCO degrades the quality factor of the resonator. As it can be seen from the equations, it is not possible to compensate the degradation of the resonator impedance with the additional power dissipation. Consequently, an external load has unconvertible effect on the VCO phase noise. The Lesson’s equation which includes the quality factor of a loaded VCO resonator is shown below:

( ) ( )( )

⎥⎥⎥⎥⎥

⎦

⎤

⎢⎢⎢⎢⎢

⎣

⎡

⎟⎟⎟⎟⎟

⎠

⎞

⎜⎜⎜⎜⎜

⎝

⎛

Δ

⎟⎟⎠

⎞⎜⎜⎝

⎛+

++=Δ

2

0

2 2

112

ω

ωω

U

ext

res

extrestail

extres

QRR

RRIRRFkTL ,

Consequently, the external load negatively affects the key parameters of the VCO and cannot be completely compensated by the additional power dissipation or any other way.

III. PROPOSED VCO TOPOLOGY. IMPLEMENTATION AND MEASUREMENT RESULTS

Based on the considerations which are presented above, the LC VCO architecture (Fig.4a) has been developed. In order to avoid the negative effects which are introduced by the external load, the outputs of the oscillators are connected at the emitters of the transistors M1 and M2. Thus, the LC resonator is isolated from the external effects. As a result, the VCO carrier frequency and frequency tuning range are insensitive to the external parasitic capacitances. Additionally, the impedance of the resonator is free from the degradation caused by the parasitic impedances of the external load. Similar to the conventional structure, the oscillation frequencies of the proposed VCO is mostly defined by the values of the inductor (Lind) and the voltage variable capacitance (Cv) of the parallel-tuned resonator. The PMOS current source determines the current of the VCO core. The cross-coupled transistor pair introduces negative impedance for compensating the losses of the resonator and supporting the stable oscillation. Besides the isolation of the resonator, the common collector configuration of the oscillator outputs allows driving low impedance loads.

The resonator of the VCO consists of the inductor and variable capacitor. The inductor is designed with the thickest metal layer for the high quality factor. The inductance value is

equal to 2nH and the quality factor is around 30 at the frequency range between 6.0 GHz and 8.5 GHz (simulated). Large inductance value supports the start up oscillation condition and high voltage swing across the resonator with the low power dissipation. The varactor varies its capacitance in the range between 450 fF and 1400 fF (simulated).

Vdd

OUT- OUT+

M1

R

M2

R

Bias

L L

Varactors

Cpar Cpar

Lind Lind

RvP RvP

CV

CTR CTR

CV

RindPRindP

Cext Cext

Rext Rext

ISOLATION

(a) (b) Fig. 4. (a) The simplified schematic of the proposed LC VCO; (b) the equivalent circuit of the LC VCO loaded by the external load.

Fig. 5. The measured frequency tuning range of the LC VCO.

Fig. 6. The measured phase noise of the LC VCO

30

Voltagecontrolledoscillator

Fig. 7. The micrograph of the IC. In order to better observe the advantages of the proposed

architecture the equivalent circuit of the proposed VCO is shown in Fig. 4b. The equivalent circuit includes the inductor and varactor, their dynamic resistances, the negative impedance provided by the cross-coupled pair and the parasitic elements caused by the external and internal loads. The absence of the external parasitic capacitance allows increasing the VCO tuning range from usual 20% - 25% until 31 % without incorporating the digital switching and without increasing of the power consumption. The measured VCO tuning range (Fig. 5) covers the frequencies between 6.30 GHz and 8.59 GHz. The power consumption of the VCO is equal to 540 µW from the 1.8 V supply voltage. The measured phase noise of the VCO (Fig. 6) is better than -100 dBc/MHz at 1 MHz offset frequency within the tuning range. The micrograph of the die is shown in Fig. 7. The performance of the VCO is summarized in Table 1.

TABLE I. THE SUMMARIZED PERFORMANCE OF THE VCO Technology 0.25 µm BiCMOS Frequency tuning range 6.30 GHz – 8.59 GHz (31%)Tuning voltage 0…3 VPower dissipation 540 µW from 1.8 V supplyPhase noise - 100 dBc/MHz at 1MHz offset Figure of Merit - 180.73

IV. CONCLUTIONS This paper presents how the external load degrades the key

parameters of the commonly used LC VCO topology. We

show, that even high ohmic external load decreases the impedance of the resonator, limits the frequency tuning range, makes worse the phase noise and increases the power dissipation of the structure. Based on this knowledge, the novel LC VCO architecture with isolated resonator has been developed and implemented in the 0.25 µm BiCMOS process offered by IHP [11]. The isolated LC tank allows to avoid the negative effects of the external load and to obtain the ultra-low power, wideband VCO topology with acceptable phase noise performance.

REFERENCES [1] F. Herzel, S. Osmany, K. Hu, K. Schmalz and others, “An

integrated 8-12 GHz fractional-N frequency synthesizer in SiGe BiCMOS for satellite communications,” Analog Integrated Circuits and Signal Processing, Vol. 65, No. 1, October 2010, pp. 21-32(12).

[2] F. Herzel, H. Erzgraber and P. Weger, “Integrated CMOS wideband oscillator for RF applications,” Electronics Letters, Vol. 37, No. 6, March 2001, pp. 330-331.

[3] D. Hauspie, E. Park and J. Craninckx, “Wideband VCO with simultaneous switching of frequency band, active core, and varactor size,” JSSC, Vol. 42, No. 7, July 2007, pp. 1472-1480.

[4] C. Stagni, A. Italia, and G. Palmisano, “Wideband CMOS LC VCOs for IEEE 802.15.4a applications,” EuMA, Oct. 2008, pp. 246-249.

[5] Y. The, M. Annamalai Arasu, Y. Gao, Y. Zheng, “A 0.18-µm CMOS 3.2-10 GHz qaudrature VCO for IEEE 802.15.4a UWB transceivers,” APMC, Dec. 2009, pp. 245-248.

[6] Z. Zhujin, L. Ning, L. Wei, R. Junyan, “A power-optimized CMOS quadrature VCO with wide-tuning range for UWB Receivers,” ISCAS, Oct. 2007, pp. 437-440.

[7] J. Maget, “Varactors and inductors for integrated RF circuits in standard MOS technologies,” PhD thesis, Oct. 2002, p. 29.

[8] P. Tobin, “PSpice for circuit theory and electronic devices,” Series ISSN: 1932-3174 electronic, 2007, pp. 84-87.

[9] D. Cordeau, J.-M. Paillot, “Minimum phase noise of an LC oscillator: Determination of the optimal operating point of the active part,” AEU, International Journal of Electronics and Communications, Feb. 2009.

[10] Foundry Service of IHP, Available: http://www.ihp-microelectronics.com

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