191978-1-4799-5296-0/14/$31.00 © 2014 IEEE

PROC. 29th INTERNATIONAL CONFERENCE ON MICROELECTRONICS (MIEL 2014), BELGRADE, SERBIA, 12-14 MAY, 2014

Analog Front End Stage of a Fiber Optic Magnetic Field Point Scanner

S. J. Petričević, P. Mihailović, M. Barjaktarović, J. Radunović

Abstract – Fiber optic Faraday magnetic field scanners are sophisticated sensing devices whose important part is electronic signal processing block. This paper presents some aspects of a design of an AFE (Fig. 2) for FOFMS that meets these requirements. Electronic section containing the AFE has been constructed and implemented into electronic processing section containing analog to digital conversion (ADC). This unit has been integrated into a sensor and used to collect and process data in order to asses the sensor performance figures..

I. INTRODUCTION

Faraday magnetic field sensing has been under

continuous investigation for several decades, with various applications and research directions in mind [1]-[7]. Advantages of these devices over other techniques used to sense magnetic fields include complete dielectric isolation, no field perturbation, enormous bandwidth and passive sensing head construction. Techniques and devices implemented in fiber optic communications have become readily available on the market making construction of the fiber optic Faraday magnetic field scanner (FOFMS) an attractive option.

Particular area of interesting applications for point type scanner include: magnetic flux leakage detection, power systems equipment monitoring and environment monitoring. Magnetic flux leakage detector operates with induction levels of up to tens of mT. Power systems monitoring involves fiber optic measuring of current via sensing its induced magnetic field. Health concerns have led to regulations regarding recommended maximum induction values near substations, ranging from 6 µT at 3 kHz up to 40 mT at low frequencies. Therefore potential applications for Faraday magnetic field scanner require a device capable of 100:1 signal to noise ratio, measurement range from 100 µT to 100 mT and 3 kHz frequency bandwidth.

Light from the source (usually a LED or a laser diode) is coupled with a transmitting fiber that carries the light to the sensor head. Polarization state of light is modulated by magnetic induction at the head, then converter to intensity change and carried to electronic processing block by two receiving fibers. These are converted by photo detectors (photodiodes or phototransistors) and converted to voltage for further processing.

This paper is concerned with a design of an electronic section that performs such function (also know as analog front end or AFE for short) tasked with providing adequate conversion of light to voltage. It must be designed to

measure low level modulation of an optical carrier imposed by the magnetic field with sufficient resolution, yet be capable of adapting to variations in the light carrier intensity due to other sources (aging, temperature, mechanical coupling etc.). Further, it must comply with conventional requirements for an analog amplifier stage such as offset, bandwidth, stability and others. These requirements are tightly coupled and mandate careful design approach.

Fiber optic sensing generally requires that some kind of normalization be applied in signal processing in order to eliminate effects arising from light source intensity changes, fiber bending, temperature, aging and other causes with similar spectrum to the sensed quantity. Our design is based on Δ/Σ normalization that can be expressed as

1 1 2

1 2

1 sin2

U Ur const BU U

(1)

Quantity r represents numerical value linearly related

to magnetic field induction B and is obtained from two voltages (U1 and U2). These represent intensities of light from two receiving fibers impinging on photodetectors (AFE output).

II. THE ELECTRONIC PROCESSING BLOCK

Electronic processing section (seen in fig. 1) of a

FOFMS determines to great extent the performance of a device. Analog front end (AFE) consists of transimpedance stages and part of power supply system and influences signal-to-noise ration of a sensor together with analog to digital converter

Fig. 1. Electronic processing section in a fiber optic magnetic field scanner.

AFE

LED

ADC

MCU

USB

POWER

PCUSB

TX FIBER

RX FIBER 2

RX FIBER 1

192

AFE output (voltages U1 and U2) are fed to 16 bit analog to digital converter inside a microcontroller unit (SiLabs C8051F064) that samples the voltages and creates data stream transmitted to a PC computer (via USB) for processing (Eq. 1). MCU also controls light source (LED). Power to the entire block is supplied from the same USB link by a multi stage switcher that creates clean dual analog supply (± 5V) for AFE and supply for other blocks.

III. ANALOG FRONT END

Two crucial aspects of AFE design need to be taken

into account: signal-to-noise ration and dynamic range. Light modulation that takes place in the sensing head is small, about 0.15% per mT of magnetic field induction [8]. If 100 points are required in the full range (about 10 μT resolution), SNR must be at least 1:70000. Transimpedance stages must produce voltages lower than ADC maximum input range (2.5 V) so voltage resolution translates to 35 μV per 1 LSB of ADC. AFE noise must remain below this level.

Fiber motion, light source aging, mechanical coupling effects and other causes cause wide range light intensity variations at the head input and at the photodetectors. This can cause the power at photodetectors to vary anywhere from few μW up to hundreds of μW. As mentioned earlier, voltage at ADC inputs (AFE outputs) must remain within specified range. Therefore transimpedance stage must have adjustable gain to adapt to such a condition of wide dynamic range.

Accuracy could also be a substantial problem. A look at Eq. 1 shows operations of subtraction, addition and division followed by inverse trigonometric function. Changes in values for voltages U1 and U2 are small (as explained due to low modulation) and if these contain errors from say op amp offsets, results calculated can be quite incorrect. Another source of error is the photodiode dark current that causes a DC voltage at the op amp output dependant on the transimpedance gain. These two error sources can be compensated for by op amp offset compensation techniques.

Of less concern is the bandwidth requirement. As stated in introduction, bandwidth of 3 kHz is required and can be accomplished with relative easy by using large bandwidth op amp. This is then compensated with a capacitor calculated for required bandwidth at maximum transimpedance gain (maximum transimpedance resistance).

Solution for AFE is seen in fig. 2. Wide dynamic range problem is solved by using a digital potentiometer for transimpedance resistance (Analog Devices AD5242). These dual units feature 1 MΩ resistance thus allowing for 2 μA of minimum photocurrents from detectors. They are controlled using I2C interface (SCL and SDA lines) allowing for several chips on the same line for reasons of expansion. Op amp of choice is Analog Devices AD8034 (dual JFET input high bandwidth) with compensating

capacitors (CTA and CTB) dimensioned for 3 kHz bandwidth. Offset and dark current compensation is accomplished by applying voltages VOA and VOB with additional damping networks.

CTA

47pF

VTIAA

3

21

84

TIAAAD8034

+5

C

B

E12

SHIE

LD

0

FTA

SFH250V SFH350V

VPDA

CTIA1100n

GND

GND GND

CTB

47pF

GND

VTIAB

5

67

TIAB

AD8034

+5

SCL

C

B

E12

SHIE

LD0

FTB

SFH250V SFH350V

SDA

VPDB

+5

RTIAA

100

RTIAB

100

-5

VBIAS

VBIAS

SELB

SELA

3

21

411

OPA

LM6134BIN

COP1100n

CTIA2100n

GND

GND

10

98

OPC

LM6134BIN

RC1

4K7

5

67

OPB

LM6134BIN

VBIAS

VC

+5

-30COP2100n

GND

U?

SYM 1 OF 1

?

?

AD7398BR

VC

C16

CA1 10

VOUTD14

REFLO 2

CA0 11

VOUTC13 VOUTB5 VOUTA4

GN

D1

CA2 7

VREFD 15

SDA 9SCL 8

VREFC 12VREFB 6VREFA 3

DAC

LTC2609

SCLSDA

GND

+5

+5

GND

GND

VOA

1.25VVAVBVCVD

RA2

4K7

RA1

4K7

VB

1.25V

CA2

100n

RB2

4K7

CB2

100n

RB1

4K7

VOA

X012

X114

X215

X311

Y01

Y15

Y22

Y34

INH6

A10

B9

VEE7

X 13

Y 3

VC

C16

GN

D8

MUX

MC74HC4052-5

+5

GND

SELASELB

VINA

VINB

COA10u

ROA1

100K

GND

VOB

COB10u

ROB1

100K

GND

VOB

CDAC

100n

GND

CDPOT100n

GND

ROA2100K

ROB2100K

RC2

68K

GND

12

1314

OPD

LM6134BIN

RD1

4K7

VCAL

VD

RD2

15K

GND

+

CC2

100uF

1.25V

VA

DTIABTMMBAT42

GND

DTIAATMMBAT42

GND

DTIAA-

TMMBAT42

GND

DTIAB-

TMMBAT42

GND

VINB

VINA

VPDA

VPDB

VCAL

VCAL

CD2100n

CMUX2100n

GND

CMUX1100n

GND

GND

GND

GND

A12 B1 4

W1

3

A216 B2 14

W2

15

O1

1

O2

13

AD

09

AD

110

SCL

7

SDA

8

SHD

WN

6

VSS 12

DG

ND

11

VDD5

DPOTAD52421M

RCAL1M

Fig. 2. Analog front end schematics.

Since analog voltages are required a digital to analog

converter (DAC, model LTC2609 from Linear Technologies) is required. A four channel unit (voltage VA, VB, VC and VD) used can also supply voltage for photo detector bias (VBIAS) and calibration (VCAL). Additional quad op amp (LM6134) and associated network provide for single to dual supply voltage conversion and scaling. Photodetectors FTA and FTB can be either photo diodes (model SFH250V from Siemens) or phototransistors (SFH350V) suitable for fiber coupling. Choice depends on the amount of optical power available at the receiving fiber. Photodiodes provide superior temperature and bandwidth performance.

Since channel resistance of AD5242 also exhibit error and channel differences, these must be taken into account by calibration. This must be done before sensing photocurrents and is accomplished by multiplexing the same calibration current from 1 MΩ RCAL into VINA or VINB transimpedance stage input. Channels are calibrated in turn and the calibration current can be adjusted by DAC to accommodate for dynamic range. Once calibration is done, multiplexer switches photo detector currents (VPDA and VPDB) to transimpedance inputs and the measurement can begin. The ADC is capable of clibration correction of offsets and gains for its channels and this option is used to compensate for operational amplifier offsets and resistance differences in digital potentiometer channels.

Multiplexer characteristics do not affect the performance since multiplexer channel resistance mismatches and values are much smaller then calibration resistance and detector output impedances.

Detector bias can be set to reduce their inherent pn junction capacitances but was found by test to be redundant.

The rest of the circuit involves some protection diodes and decoupling networks that do not affect its function.

193

IV. EXPERIMENTAL RESULTS AND DISCUSSION

Experiments with the FOFMS have been carried out in

laboratory conditions in order to test the performance of the scanner. Noise testing was accomplished by placing the scanners head inside Helmholtz coils with controlled current and then acquiring magnetic induction values for a long time. Details of the setup are discussed in [8].

0 20 40 60 80 100 120 140 160 180 2002.077

2.078

2.079

2.080

2.081

2.082

2.083

2.084

2.085

2.086

B [m

T]

t (s) Fig. 3. Scanner noise.

Result can be seen in fig. 3 demonstrating that the sensor’s noise level is about 8 μT (peak to peak). This noise level contains contribution noises form LED noise, photodetectro noise, AFE noise and ADC nois (e.g. system noise).

Fig. 4 demonstrates a scan of magnetic induction inside Helmholtz coils in a plane parallel with coils but placed in the middle between them.

Fig. 4. Magnetic field scan obtained by FOFMS

The field is quite uniform in the coil center, but rises as the sensing head approaches the windings, as can be expected. Excellent SNR and lineariry can be inferred by simetrical shape and smooth surface.

Amplitude response of the electronic block is seen in fig. 5 obtained by sweeping the excitation frequency.

0 1k 2k 3k 4k 5k 6k 7k-5

-4

-3

-2

-1

0

1

B(f)/

B(50

Hz)

[dB]

f[Hz] Fig. 5. Normalized amplitude response of the FOFMS.

V. Conclusion

Dedicated electronic processing block has been

constructed for a fiber optic magnetic field Faraday effect scanner. Attention in the paper has been placed on analog front end stage whose design is of crucial importance for the performance of the device. A solution adaptable to various conditions has been proposed and tested in laboratory conditions.

ACKNOWLEDGEMENT

This work was supported by the Ministry of

Education and Science of the Republic of Serbia, project III 45003.

REFERENCES

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