implementation of a high efficiency, high lifetime and low cost converter for an autonomous...

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Implementation of a High Efficiency, High Lifetime,

and Low Cost Converter for an Autonomous

Photovoltaic Water Pumping System João Victor Mapurunga Caracas, Student Member, IEEE, Guilherme de Carvalho Freitas, Student Member, IEEE, Luis Felipe

Moreira Teixeira, Student Member, IEEE, Luiz Antonio de Souza Ribeiro, Member, IEEE

Abstract— This paper proposes a new converter for photovoltaic

water pumping or treatment systems without the use of chemical

storage elements, such as batteries. The converter is designed to

drive a three-phase induction motor directly from photovoltaic

(PV) energy. The use of three-phase induction motor presents a

better solution to the commercial DC motor water pumping system.

The development is oriented to achieve a more efficient, reliable,

maintenance free and cheaper solution than the standard ones, that

uses DC motors or low voltage synchronous motors. The developed

system is based on a current-fed multi-resonant converter also

known as Resonant Two Inductor Boost Converter (TIBC) and a

Full-Bridge Three-phase Voltage Source Inverter (VSI). The classic

topology of the TIBC has features like high voltage gain and low

input current ripple. In this work, it is further improved with the

use of a non-isolated recovery snubber along with a hysteresis

controller and the use of a constant duty cycle control to improve

its efficiency. Experimental results show a peak efficiency of 91 %

at rated power of 210 W, for the DC/DC converter plus the three-

phase VSI, and 93.64 % just for the DC/DC converter. The system

is expected to have high lifetime, due to the inexistence of

electrolytic capacitors, and the total cost of the converter is below

0.43 U$/Wp. As a result, the system is a promising solution to be

used in isolated locations and to deliver water to poor communities.

Index terms– Solar power generation; Photovoltaic power systems;

DC-DC power conversion; AC motor drives; DC-AC power

conversion.

I. INTRODUCTION

Currently, over 900 million people in various countries

do not have drinkable water available for consumption. Of this

total, a large amount is isolated, located on rural areas where the

only water supply comes from the rain or distant rivers [1], this is

also a very common situation in the north part of Brazil, where

this work was developed. In such places, the unavailability of

electric power rules out the pumping and water treatment through

conventional systems. One of the most efficient and promising

way to solve this problem is the use of systems supplied by PV

solar energy. This kind of energy source is becoming cheaper and

has already been put to work for several years without the need

of maintenance. Such systems are not new, and are already used

for more than three decades [1]. Nevertheless, until recently, the

majority of the available commercial converters in Brazil are

based on an intermediate storage system, performed with the use

of lead-acid batteries, and DC motors to drive the water pump [3].

More sophisticated systems have already been developed with the

use of low voltage synchronous motor [4], but these, although

presenting higher efficiency, are too expensive to be used in poor

communities that need these systems.

The batteries allow the motor and pump system to

always operate at its rated power even in temporary conditions of

low solar radiation. This facilitates the coupling of the electric

dynamics of the solar panel and the motor used for pumping [5].

Generally, the batteries used in this type of system have a low life

span, only two years on average [5], which is extremely low

compared to the useful life of 20 years of a PV module. Also, they

make the cost of installation and maintenance of such systems

substantially high. Furthermore, the lack of batteries replacement

is responsible for the failure of such systems in isolated areas.

The majority of commercial systems use low-voltage

DC motors, thus avoiding a boost stage between the PV module

and the motor [7]. Unfortunately, DC motors have lower

efficiency and higher maintenance cost compared to induction

motors and are not suitable for applications in isolated areas,

where there is no specialized personnel for operating and

maintaining these motors. Another problem is that low voltage

DC motors are not ordinary items in the local markets. Because

of the aforementioned problems this work adopted the use of a

three-phase induction motor, due to its greater robustness, lower

cost, higher efficiency, availability in local markets and lower

maintenance cost compared to other types of motors.

The design of a motor drive system powered directly

from a photovoltaic source demands creative solutions to face the

challenge of operating under variable power restrictions and still

maximize the energy produced by the module and the amount of

water pumped [8]. These requirements demand the use of a

converter with the following features: high efficiency – due to the

low energy available; low cost – to enable its deployment where

it is most needed; autonomous operation – no specific training

needed to operate the system; robustness –minimum amount of

maintenance possible; and high life span – comparable to the

usable life of 20 years of a PV panel.

This paper proposes a new DC/DC converter and

control suitable for photovoltaic water pumping and treatment

that fulfill most of the aforementioned features. The paper is

organized as follows: in section II it is described the proposed

system; in section III the DC/DC converter itself is presented and

analyzed; in section IV the control strategy is explained, and in

section V the experimental results are presented.

II. PROPOSED CONVERTER

To ensure low cost and accessibility of the proposed

system, it was designed to use a single PV module. The system

should be able to drive low power water pumps, in the range of

1/3 HP, more than enough to supply water for a family. Fig. 1

presents an overview of the proposed system. The energy

produced by the panel is fed to the motor through a converter with

two power stages: a DC/DC TIBC stage to boost the voltage of

the panels and a DC/AC three-phase inverter to convert the DC

voltage to three-phase AC voltage. The inverter is based on a

classic topology (three legs, two switches per leg) and uses a

sinusoidal pulse width modulation (SPWM) strategy with 1/6

optimal third harmonic voltage injection as proposed in [9]. The

use of this PWM strategy is to improve the output voltage level

as compared to sinusoidal PWM modulation. This is a usual

topology and further analyses on this topology are not necessary.

For the prototype used to verify the proposed system, a careful

selection of the VSI components is more than enough to

guarantee the efficiency and cost requirements.



3Φ Induction

Motor

Centrifugal

Water Pump

PVThree

phase

inverter

DC/DC CONVERTER

Modified Two Inductor boost converter3Φ INVERTER

HF TransformerHigh Voltage

DC Bus

Voltage

doubler

rectifier

Non-dissipative

Snubber

Current

Fed

inverter

Resonant

tank

Figure 1. Simplified block diagram of the proposed system.

The required DC/DC converter for this kind of system

needs to have large voltage conversion ratio, because of the low

voltage characteristic of the PV panels and small input current

ripple so it do not causes oscillation over the maximum power

point (MPP) of the PV module [10]-[12], thus ensuring the

maximum utilization of the available energy.

The commonly used isolated voltage-fed converters

normally have a high input current ripple, which forces the

converter to have large input filter capacitors. These are normally

electrolytic, which are known to have a very small lifetime and

thus affect the overall life span and mean time before failure

(MTBF) of the converter. Furthermore, the inherent step-down

characteristic of the voltage-fed converters, the large transformer

turn ratio needed to boost the output voltage, the high output

diode voltage stress, and the need of an LC output filter [13]

makes voltage-fed converters not the best choice for this

application.

When compared to the voltage-fed topologies, current-

fed converters have some advantages. Usually they have an

inductor at the input so the system can be sized to have input

current ripple as low as needed, thus eliminating the need of the

input capacitor at the panel voltage. Current-fed converters are

normally derived from the boost converter, having an inherent

high step-up voltage ratio, which helps to reduce the needed

transformer turn ratio. The classical topologies of this kind are

the current-fed push-pull converter [14],[15], the current-fed full-

bridge [16] and the dual half-bridge converter [17]. Although the

current fed topologies have all the above mentioned advantages

they still have problems with high voltage spikes created due to

the leakage inductance of the transformers, and high voltage

stress on the rectifying diodes [18].

One of the solutions to the current-fed pulse width

modulation converters it is the use of resonant topologies able to

utilize the component parasitic characteristics, as the leakage

inductance and winding capacitance of transformers, in a

productive way to achieve zero current switching (ZCS) or zero

voltage switching (ZVS) condition to the active switches and

rectifying diodes [19].

In this paper the use of a modified two inductor boost

converter (TIBC) for the first stage DC/DC converter is proposed,

due to its very small number of components, simplicity, high

efficiency, easy transformer flux balance [20]-[21], and common

ground gate driving for both switches. These features make it the

ideal choice for achieving the system’s necessary characteristics.

Besides the high DC voltage gain of the TIBC, it also compares

favorably with other current-fed converters concerning switch

voltage stress, conduction losses and transformer utilization [22]-

[23]. In addition, the input current is distributed through the two

boost inductors having its current ripple amplitude halved at

twice the PWM frequency. This last feature minimizes the

oscillations at the PV module operation point and makes it easier

to achieve the maximum power point (MPP).

In its classical implementation the TIBC is a hard

switched overlapped pulse modulated converter, this way at least

one of the switches is always closed, creating a conduction path

for the input inductors current. Nevertheless, the TIBC can be

modified to a multi-resonant converter by adding a capacitor at

the transformer’s secondary winding [24]-[25]. A multi-resonant

tank is formed by the magnetizing inductance of the transformer,

its leakage inductance and the added capacitor, as show in Fig.

2(a). The intrinsic winding capacitance of the transformer is

included in the resonant capacitor.

By adding this capacitor and using the parasitic

components of the transformer to create the resonant tank it is

possible to achieve (ZCS) condition for the input switches and

output rectifying diodes, this enables the converter to operate at

high frequencies with greater efficiency.

Q1 D1 Q2 D2

Np : NsPV

Resonant

Tank

Voltage

doubler

rectifier

Li1 Li2

LrLm Cr

Ds1

Ds2

Do1 Co1

Do2 Co2

Cs

(a)

(b)

Snubber(c)

Vin

Vout

VG1 VG2

A

B

IT

Iin

GND

Figure 2. Modified TIBC topology: (a) resonant tank; (b) voltage doubler

rectifier; (c) snubber.

With the use of a voltage doubler rectifier at secondary

side of the transformer, Fig. 2(b), it is possible to reduce the

transformer turns ratio, the necessary ferrite core, and the voltage

stress on the MOSFETs to half of the original ones. As a result,

the transformer is cheaper; the MOSFETs are cheaper, and the

number of diodes in the secondary side is halved. Also the output

DC bus capacitor can be integrated with the capacitors of the

rectifier, especially because the second stage, three-phase VSI,

has almost DC input current, exempting the bus capacitor from

any AC decoupling current.

Classically the TIBC have a minimum operation load to

maintain an established output voltage. Below a certain load level

the energy transferred to the output capacitor is not completely

transferred to the load and causes an increasing in the output

voltage. This happens because the inductors are charged even if

there is no output current. As a result, this converter has a

drawback when used in motor drive systems. Since the motor is

a variable load and it has large time constants, it will demand low

power at some operation points, i.e. at low speed and start-up/stop

transients. As a solution, a non-dissipative regenerative snubber

circuit is presented.

The regenerative snubber is formed by two diodes and

a capacitor connecting the input side directly to the output side of

the converter, as shown in Fig. 2(c). This makes it a non-isolated

converter, which have no undesirable effect in the photovoltaic

motor driver applications. The voltage over the MOSFETs is

applied to a capacitor connected to the circuit ground and the



voltage of this capacitor is coupled in series with the output of the

rectifier. This modification allows part of the energy to be

transferred from the input directly to the output, through the

snubber, without going through the transformer, reducing its size

and improving even more the efficiency of the converter.

It is also proposed a modification in the control strategy

of the proposed converter, when compared to the classical TIBC

control. Section IV shows that for the application of a

photovoltaic water pumping system, the correct design of

converter voltage gain makes the converter able to operate with a

constant gain, using a fixed duty cycle modulation for the primary

switches. To solve the minimum load condition it is proposed the

use of a hysteresis controller for the DC output voltage of the first

stage. When operating below the minimum required load, an

uncontrolled increase of output voltage will be seen, once this

voltage reaches the upper threshold, both primary switches are

turned off. This would be impossible in all the previously

reported implementations of the TIBC. However, with the added

snubber, even with both switches turned off there is still a path

for the input inductors’ current. Their energy is directly

transferred to the snubber capacitor Cs. This capacitor must be

sized to have a minimum voltage increase during this hysteresis

action. As this capacitor is in series with the output rectifier, the

same voltage increase will be noted in the output voltage. After

the switches are turned off, and the input inductors’ energy is

transferred to the snubber, the output voltage will start to

decrease, reaching the lower hysteresis threshold and restarting

the PWM operation with the fixed duty cycle.

III. OPERATION PRINCIPLE

To simplify the analysis of the proposed converter the

following assumptions need to be true during a switching

interval: the input inductors Li1 and Li2 are sufficiently large so

that their current is almost constant; the capacitors Co1, Co2 and

Cs are large enough to maintain a constant voltage; the output

capacitors Co1 and Co2 are much larger than Cr, to clamp the

resonant voltage.

In hard switched operation of the TIBC, the two primary

switches Q1 and Q2 operates at an overlapped duty cycle

switching scheme to guarantee a conduction path for the primary

inductors current. When both Q1 and Q2 are turned on, Li1 and Li2

are charged by the input energy. When Q1 (Q2) is opened the

energy stored in Li1 (Li2) is transferred to Co1 (Co2) through the

transformer and the rectifier diode Do1 (Do2).

Once the multi-resonant tank is introduced, two

different resonant processes occur: 1) when both switches are

closed the leakage inductance Lr participates along with

capacitance Cr in the resonance at the primary current switching

and current polarity inversion, allowing ZCS operation for the

primary switches; 2) during the conduction time interval

(between t4 and t5 on Fig. 3), when at least one of the switches is

open, Lr is associated in series with Li1 or Li2, not participating on

the transformer’s secondary current resonance, formed only by

Lm and Cr. The key waveforms for a switching period of the TIBC

are presented in Fig. 3. In this figure VgQ1 and VgQ2 are the gate

signals of the switches Q1 and Q2, respectively; VdsQ2 is the drain

to source voltage of MOSFET Q1; IQ2 is the current of MOSFET

Q2; VT is the voltage at the primary of the transformer; IT is the

current at the primary of the transformer; ILi1 and ILi2 are the

current of inductors Li1 e Li2, respectively; Iin is the input current

of the converter and also the current supplied by the PV panel.

t1 t2 t3 t4 t5 t6

VgQ1

VgQ2

VdsQ2

IQ2

VT

IT

ILi1,

ILi2

Iin

VDs2

IDs2

ICs

VCr

VDo2

IDo2

Figure 3. Key waveforms of the TIBC during a switching period.

At time t1, the rectifying diode Do1 is already conducting

and the voltage on resonant capacitor Cr is clamped at +Vout /2. At

this instant, the switch Q1 is activated by VgQ1. As the switch is

turned on, its voltage drops to zero, and the snubber diode Ds1 is

forced to stop conducting. From t1 to t2, Cr transfer its energy to

the leakage inductance Lr, beginning the primary switch’s

resonant process and forcing the current IQ2 on the switch Q2 to

decrease.

At the time t2, the rectifying diode Do1 stops conducting

and Cr continues to resonate with the magnetizing inductance Lm.

From t2 to t3, the primary switch’s resonance (Q2) continues to

force its current to decrease until it reverses its polarity. When IQ2

is negative, the switch can be turned off. This happens at instant

t3 when VgQ2 forced to zero.

At the time t3 the voltage VdsQ2 starts to increase and Q2

is completely blocked, the snubber diode Ds2 begins to conduct,

transferring energy directly to the snubber capacitor Cs. Between

t3 and t4 Cr and Lm continue to resonate, decreasing the voltage on

the doubler rectifier’s input and on VCr. At instant t4, the voltage



across Cr reaches -Vout /2 and the rectifying diode Do2 starts to

conduct, clamping VCr in -Vout /2.

From t4 to t5, the capacitor Co1 is charged, and the

current of Do2 starts to decrease. At the instant t5, Q2 is turned on,

initiating the resonant process on Q1. As Q2 is activated, Ds2 is

forced to stop conduction. At the instant t6, the current in Do2

reaches zero, and Do2 stops conducting, reinitiating the resonance

between Cr and Lm. From this moment until the end of the

switching period, the process repeats symmetrically as explained

for the other input switch.

An extended description of multi-resonant TIBC

without the snubber is presented and analyzed in [24], resulting

in a detailed mathematical modeling for both resonant processes

during its operation. However, the analysis in [24] is based on

several complex mathematical models and, consequently, the

presented design method shows several dependent variables,

which translates in a design methodology difficult to be

implemented.

In this paper it is applied a simplified methodology

based on the effect of each resonant process, the resonant

frequencies and the switching frequency. Spice simulations and a

prototype are used to show that despite the simplicity of the

design methodology, the correct operation of the converter is

guaranteed, specially the soft switching of the primary switches

for the whole operating load range.

Although the resonant process affects the output

voltage, depending of the resonant tank component values and the

load, this can be neglected because of its small influence and

complex effect. Thus, neglecting the resonant effect over the

output voltage, including the voltage doubler rectifier and the

snubber connecting the primary and secondary side of the

converter, the static voltage gain (𝐾𝑣) of the converter is defined

as:

𝑉𝑜𝑢𝑡

𝑉𝑖𝑛= 𝐾𝑣 =

1

1 − 𝐷(2

𝑁𝑠

𝑁𝑝+ 1) (1)

Where D represents the duty cycle of each switch and

must be higher than 50% to guarantee the necessary overlapping

for the correct operation. Ns / Np represent the transformer turns

ratio.

According to [24], to minimize the influence of the load

on the resonant process on the primary current commutation

interval, the switching frequency (Fsw) should be higher than the

resonant frequency (Frs) of Lm and Cr by a value of at least 1.1.

Thus:

𝐹𝑟𝑠 =1

2𝜋√𝐿𝑚𝐶𝑟

≤𝐹𝑠𝑤

1.1 (2)

During the primary current commutation interval, when

both switches Q1 and Q2 are turned on, inductor Lr participates on

the resonance in parallel with Lm and Cr, thus the resonant

frequency for this interval is defined as:

𝐹𝑟𝑝 =1

2𝜋√𝐿𝑚𝐿𝑟

𝐿𝑚 + 𝐿𝑟𝐶𝑟

(3)

Considering that Lm represents the magnetizing

inductance and Lr the leakage inductance, then Lm is much larger

then Lr, thus (3) can be simplified to:

𝐹𝑟𝑝 =1

2𝜋√𝐿𝑟𝐶𝑟

(4)

The duration of the commutation interval is equal to the

overlapping time (𝑇𝑜𝑣) of the pulse driving signals, and can be

calculated as:

𝑇𝑜𝑣 =(𝐷 − 0,5)

𝐹𝑠𝑤 (5)

The whole resonant process of the primary switches has

the duration of half a resonant cycle, as showed in Fig. 3. To

guarantee the ZCS condition for the entire load range, the

following conditions must be satisfied:

𝑇𝑜𝑣 =𝜋

𝜔𝑟 (6)

𝐹𝑟𝑝 =𝐹𝑠𝑤

2𝐷 + 1 (7)

Another important constraint is the energy accumulated

in Cr at the beginning of the primary switching resonant process.

This energy needs to be completely transferred to the leakage

inductance Lr during this process. From this condition, the

following equation is derived:

𝐿𝑟 ≤𝑉𝑜

2

2𝐼𝑖𝑛2 𝐶𝑟 (8)

IV. CONTROL OF THE SYSTEM

There are three main aspects in the proposed converter’s

control: 1) during normal operation a fixed duty cycle is used to

control the TIBC MOSFETs, thus generating an unregulated high

bus voltage for the inverter; 2) a MPPT algorithm is used along

with a PI controller to set the speed of the motor and achieve the

energy balance of the system at the maximum power point of the

PV module; 3) a hysteresis controller is used during the no load

conditions and start-up of the system. Each of these aspects is

described in following sections.

A. Fixed duty cycle control

One of the most important control aspects of this system

is the fact that it is possible to use an unregulated DC output

voltage and fixed duty cycle for the first stage DC/DC converter.

As a resonant converter, there are definite time intervals in the

switching period for the resonance process occurs. By altering the

duty cycle or the switching period to control the output voltage,

the converter may no longer operate at ZCS condition. So, the

fixed duty cycle is used to overcome these design problems and

ensure that the converter is going to operate in ZCS condition

despite the input voltage or output load.

The duty cycle was chosen to guarantee that the amount

of transferred energy occurs during most part of the switching

interval. So, it is possible to transfer the same amount of energy

with a smaller RMS current. Therefore, the losses in the input

inductors (Li1, and Li2), in the MOSFETs (Q1, and Q2), and in the

transformer are smaller. As a result, the efficiency of the

converter improves.

The operation with fixed duty cycle makes the converter

work with a constant voltage gain 𝐾𝑣, almost independent of the

input voltage. With the correct design of 𝐾𝑣 the system will

always be able to transfer energy from the PV module to the



motor. Assuming that the converter is always operating at the

maximum power point of the solar panel, the output DC/DC

converter voltage (DC bus voltage) will be:

𝑉𝐵𝑈𝑆 = 𝑉𝑀𝑃𝑃𝐾𝑣

(8)

𝑉𝑀𝑃𝑃 being the maximum power point voltage of the

solar panel. Fig. 4 shows the I-V characteristic curves for a typical

solar panel. It is shown that the voltage at the maximum power

point (𝑉𝑀𝑃𝑃) has only small variations for different radiation

levels. The different 𝑉𝑀𝑃𝑃 points for various radiation levels are

represented by the black dashed line.

VMPP voltages

Voltage (V)

Current (A)

VMPP,max

VMPP,min

β

Figure 4. Photovoltaic I-V curves. The dashed line shows the VMPP points.

On the other hand, it is important to analyze the

minimum DC voltage on the inverter DC bus necessary to drive

the motor at a specific power level. For this application the V/Hz

controller was used to maintain approximately constant the

pump’s motor flux. By doing this, the controller maintains the

capability of the motor to generate nominal torque at any speed

below its rated value. Neglecting the effect of rotor slip in

induction machines, and considering that the centrifugal water

pump has its torque proportional to the square of the motor speed

and that the frequency (𝑓) in the V/Hz control is proportional to

the voltage (𝑉), the motor output power (𝑃) can be expressed as

a cubic function of the motor voltage:

𝑃 ∝ 𝑇𝑓

(9)

𝑇 ∝ 𝑓2

(10)

𝑓 ∝ 𝑉

(11)

Considering that nominal voltage is necessary to

achieve nominal power, Fig. 5 shows a comparison between the

output voltage of the DC/DC converter (dashed line) and the

minimum DC bus voltage required to operate the motor (solid

line – cubic function) in the full power range. This minimum DC

bus voltage was calculated by considering that the inverter is

operating at the maximum voltage with a modulation index of 1

(no overmodulation is allowed). The correct design of 𝐾𝑣

guarantees that the output voltage of the first stage will always be

greater than the minimum voltage necessary to drive the motor.

Using the SPWM strategy with third harmonic voltage

injection allows the inverter to generate a maximum line voltage

equal to the bus voltage. For a three-phase pump with 𝑉𝑟𝑚𝑠 as

nominal line voltage, the gain 𝐾𝑣 can be calculated by:

𝐾𝑣 >

𝑉𝑟𝑚𝑠 × √2

𝑉𝑀𝑃𝑃,𝑚𝑎𝑥

(12)

PR

Converter output voltage

for a constant gain Kv

Voltage (V)

VMPP,max × Kv

Inverter minimum

DC voltage

VMPP,min × Kv

Power (W)

β

Figure 5. DC bus value and minimum voltage needed for pump operation in

constant V/Hz.

B. MPPT control

The Maximum Power Point Tracking (MPPT) is a

strategy used to ensure that the operating point of the system is

kept at the MPP of the PV panel [28]. The widely used hill-

climbing algorithm was applied due to its simple implementation

and fast dynamic response.

This MPPT technique is based on the shape of the power

curve of the PV panel. This curve can be divided in two sides, to

the left and to the right of the MPP. By analyzing the power and

voltage variation, one can deduce in which side of the curve the

PV panel is currently operating and adjust the voltage reference

to get closer to the desired point. The voltage reference is used on

a PI controller to increase or reduce the motor speed and

consequently adjust the bus and panel voltage by changing its

operating point.

C. Hysteresis control

The main drawback of the classical TIBC is its inability

to operate with no load or even in low load conditions. The TIBC

input inductors are charged even if there is no output current, and

the energy of the inductor is lately transferred to the output

capacitor raising its voltage indefinitely until its breakdown.

Classically the input MOSFET cannot be turned off

because there is not an alternative path for the inductor current.

However, with the addition of the proposed snubber the TIBC

switches can be turned off. Thus, a hysteresis controller can be

set up based on the DC bus voltage level. Every time a maximum

voltage limit is reached, indicating a low load condition, this

mode of operation begins. In this case, the switches are turned off

until the DC bus voltage returns to a normal predefined level. As

a result, the switching losses are reduced during this period of

time.

V. SYSTEM DESIGN

The hardware was designed to achieve a final system

with low cost, thus an accessible and market friendly product.

The motor used in the centrifugal pump is a 0.2Hp (mechanical

power) three-phase induction motor. Considering the nominal

power and efficiency of the motor, a 210W PV panel was chosen

[29]. The main characteristics of the used panel and motor are

shown in Table I.

TABLE I. PANEL AND MOTOR PARAMETERS



Parameters Values

PV Model KD210GX

PV Power 210 W

PV Open circuit voltage (𝑉𝑜𝑐) 29.9 V

PV Short circuit current (𝐼𝑠𝑐) 6.98 A

PV Maximum MPP voltage

(𝑉𝑀𝑃𝑃,𝑚𝑎𝑥) 26.6 V

Motor nominal power 0.2 HP

Motor nominal voltage (𝑉𝑟𝑚𝑠) 220 V 3 𝜙

Motor nominal frequêncy 60 Hz

The main parameters for the converter are summarized

in Table II. The maximum allowed input current ripple was

defined to guarantee a small oscillation around the maximum

power point (MPP). The bus voltage was defined using the

minimum necessary voltage for the chosen inverter topology and

PWM strategy, as shown in (13).

𝑉𝑏𝑢𝑠 > 𝑉𝑟𝑚𝑠 × √2 = 350 𝑉 (13)

The TIBC switching frequency must be a tradeoff

between switching losses and the size of the transformer and

inductors. A frequency of 100 kHz was used based on simulation

and previous experimental knowledge.

The 𝐾𝑣 gain necessary for the converter can be

calculated using (12):

𝐾𝑣 >𝑉𝑟𝑚𝑠 × √2

𝑉𝑀𝑝𝑝,𝑚𝑎𝑥= 11.69 (14)

To guarantee the overlapping conduction of the

MOSFETs while still maintaining the largest possible duty cycle

for the constant duty cycle operating control, D was chosen to be

53 % based on the minimum required overlapping and

commutation times of the chosen drivers and MOSETs. Using (1)

the minimum ratio 𝑁𝑠

𝑁𝑝 can be determined by (15).

𝑁𝑠

𝑁𝑝>

𝐾 × (1 − 𝐷) − 1

2= 2.25 (15)

TABLE II. CONVERTER DESIGN SPECIFICATIONS

Parameters Values

Converter Input Current Ripple 5 %

Nominal Bus Voltage 350 V

TIBC Switching Frequency 100 kHz

Inverter Switching Frequency 7.7 kHz

Constant voltage gain K 11.69

Transformer turns ratio 𝑁𝑠 𝑁𝑝⁄ 2.25

The high efficiency, low cost and robustness of the

system depends directly on the choice of components. To obtain

the expected level of efficiency each component was carefully

selected. The TIBC and inverter’s MOSFETs were chosen to

minimize the conduction losses. Extra low equivalent series

resistance (ESR) MOSFETs were used. Normally the ESR is

inversely proportional to the gate charge and output capacitance

of the MOSFET. These features have large impacts on the

switching losses, but this effect is reduced by the fact that the

DC/DC converter is resonant and has reduced switching losses.

All the diodes used in the voltage doubler rectifier and in the

snubber of the TIBC were chosen as superfast diodes. Therefore,

not only the conduction losses but also the ringing caused by the

diodes reverse recovery time are reduced. Table III shows the

main components used in the design.

TABLE III. COMPONENTS USED IN THE PROTOTYPE.

Parameters Values

TIBC MOSFET Q1, Q2 IPB039N10N3G

Diodes Do1, Do2 ER3G-TPMSCT

Diodes Ds1, Ds2 MURD620

Inverter MOSFETs IXFH60N50

Li1, Li2 100uH

Inductor Core NEE-40/17/12

Inductor turns 10

Transformer Core NEER-40/22/13

Np 12

Ns 31

Cr 3 x 2nF Mica

Co1, Co2, Cs 1.5F polypropylene

Considering the fact that the system is isolated and only

fed by a single photovoltaic module, the energy used by the

control components must be derived from the module’s power. A

high efficiency integrated switching power supply is used to

perform this task. It was observed that a gain of 1% in the system

efficiency is obtained using this as opposed to a linear regulator.

To reduce power consumption and cost the whole

control system was implemented on a single low power digital

signal controller, the DSPIC30F3010. Fig. 5 presents an

overview of the control system. Based on the measured PV panel

voltage (VPV) and current (IPV), the MPPT estimates a frequency

reference (fOP) to drive the motor, which indirectly serves to

regulate the PV voltage by modifying the amount of power

transferred to the motor. A Volt-Hertz controller calculates the

output voltage based on the operating frequency. Then, the output

voltage is used to calculate the modulation index (MPV) for the

switching pattern.

Hill-

Climbing

Algorithm

VPV

IPV

Volt-

Hertz

SPWM

with Third

Harmonic

Injection

PI

Controller

MPPT

DC/DC Converter

Control

Hysteresis

Controller

Q1_PWM

Q2_PWMVBUS

S1

S2

S3

S4

S5

S6

Inverter

Control

VREF

fOP MPV

Figure 6. Block diagram of the control system.

VI. SIMULATION AND EXPERIMETAL RESULTS

The proposed converter was simulated on Pspice Orcad

v16.5. Fig. 7 shows the schematics used for the first stage TIBC.



All parasitic series resistances were included in the transformer

and capacitors. The control of the primary MOSFETs was

simulated using a fixed pulse modulation and a voltage controlled

source to implement the hysteresis control within the limits of

380V ± 10V.

Fig. 8 shows the overlapped pulses used to control Q1

and Q2, the current in both input inductors, and the current in the

PV module. It is observed that each one of the inductors has a

current ripple at the converter switching frequency and out of

phase with each other, however both currents are supplied by the

PV module and when they are analyzed together (IPV) it is seen a

reduction in the ripple amplitude to half of the original ones.

Fig. 9 shows the gate to source voltage, the drain to

source voltage and current for one of the primary MOSFETs. It

is observed that there is no voltage spikes or increased voltage

stress over the switches. In addition, the figure shows that both

turn-on and turn-off occurs at almost zero current switching.

Fig. 10 shows voltage and current on the output

rectifying diode Do1. It is shown that not only the primary

MOSFETs are operated under ZCS condition but also the

rectifying diodes. The operation under ZCS condition allows the

use of fast recovery diodes instead of the expensive silicon

carbide ones, thus reducing the total cost of the system.

8

4

12

0

20

Volt

s (V

)A

mper

es (

A)

(a)

(b)

3.980 3.984 3.988 3.992 3.996

Time (ms)

4.0

PWM VgQ1 PWM VgQ2

IPV

ILi1 ILi2

Figure 8. Simulation results of the input of the DC/DC converter. (a) input

switches’ driving signals; (b) inductors’ input currents ILi1 and ILi2, and total input

current (IPV).

0

-40

80

0

-10

20

Vo

lts

(V)

Am

per

es (

A)

Vo

lts

(V)

(a)

(b)

3.980 3.984 3.988 3.992 3.996

Time (ms)

4.0

PWM VgQ2IQ2

VdsQ2

ZCS of

switches

Figure 9. Verification of the ZCS condition on the input switch Q2. (a) VgQ2

driving signal and IQ2 current; (b) VdsQ2 drain-source voltage.

0

200

400

0

-2.0

2.0

-200

Vo

lts

(V)

Am

per

es (

A)

Vo

lts

(V)

(a)

(b)

3.980 3.984 3.988 3.992 3.996

Time (ms)

4.0

VDo1

VdsQ2

IDo1

ZCS of output

diodes

Figure 10. Verification of the ZCS condition on the rectifying diode Do1: (a)

diode Do1 current and forward voltage; (b) diode Do1 reverse voltage.

Fig. 11 shows a picture of the developed prototype. It

was tested using real water pumping system and a PV module. A

complete autonomous operation was obtained. This includes the

system automatic start-up and shutdown according to the

radiation level.

Q1 Q2

RI4

10m

RI3

10m

TR1

COUPLING=0.999999

L4

96u

L3

96u

Ds2

Ds1

Cs4.5u

RM3

9.7m

LD2

1.3u

Do1

MUR860

Do2

MUR860

Co11.5u

RM4

49m

R7500m

C175.8n

Co21.5u

R15100m

R14100m

R1220m

C191u

R13

750

V6

26.6

Rg2

12

Rg1

12

Lm3

437u

Lm4

3098u

U6

VC_SWITCH

U7

VC_SWITCH

THERSHOLD =380ON_RESIST=0.1OFF_RESIST=100KHYSTERESIS=10

V4

VPULSE

V1=0V2=15TD=0TR=20nTF=20nPW=5.2uPER=10u

V5

VPULSE

V1=0V2=15TD=5uTR=20nTF=20nPW=5.2uPER=10u

THERSHOLD =380ON_RESIST=0.1OFF_RESIST=100KHYSTERESIS=10

+

-

Figure 7. Circuit used in the simulation.



Figure 11. Developed (a) prototype and (b) test bench composed of: PV panel;

water pump; electronic converter; water tank; PV voltage, PV current and water

flow sensors.

A programmable source was used to emulate the solar

radiation during the period of a day. The converter was connected

to this source to evaluate its overall performance. Fig. 12 presents

the obtained curves. The system is supposed to operate on the

maximum power point curves at all times, meaning it needs to

stay as close as possible to the green curves. These green curves

are the MPPT voltage, current and power levels produced by the

programmable source. The system was able to operate near the

ideal voltage (𝑉𝑀𝑃𝑃) and current (𝐼𝑀𝑃𝑃), during the programed

time, thus achieving operation close to the maximum power

point. It is also shown the generated power of the programmable

source and the corresponding water flow for the same period of

time.

The MPPT routine analysis was also performed using a

real PV module. A real time analysis of the operation point was

done, in other words, the capability of keeping a steady position

on the MPP of the PV panel. Fig. 13 demonstrates a real time I-

V curve, and the operating point of the system is emphasized by

the darker spot. The system starts its operation at the open circuit

voltage (VOC) and them stabilizes on the maximum power point.

As the solar radiation varies, the maximum power points move.

The system was able to stable track these points. The black

dashed line shows all the points during a day.

VPV

VMPP

VOC

IPV

IMPP

ISC

150

100

50

0

0.4

0.2

0 1 2 3 4 5 6 7 8

0

200

0 1 2 3 4 5 6 7 8

10

5

0

0

10

20

30

40

Time (h)

Time (h)

PPV

PMPP

PV Voltages

PV Currents

Pump Water Flow

PV Power

Volt

s (V

)A

mper

es (

A)

Watt

s (W

)F

low

(l/

s)

Figure 12. Voltage and current curves during PV emulation test. Red curves

show open circuit voltage and short circuit current. Green curves show ideal

voltage and current at the maximum power point. Blue curves show the system

voltage and current during the test.

Operation on VMPP

for different radiation

levels over a day

Voltage (V)

Current (A)

β

Real-time

MPP operation

System

Start-up

VOC

Figure 13. Real time MPP operation at a real PV module.

Fig. 14(a) shows the current and voltage waveforms in

one of the MOSFETs during a switching interval, with the

converter operating at full-load. It is observed that ZVS is

achieved for both turn-on and turn-off events. Fig. 14(b) shows

the TIBC output voltage (green line), the MOSFET Q1 drain to

source voltage (pink line) and the MOSFET Q1 Current (orange

line) for operation with no output load (the AC motor was

removed). It is noted the stable operation of the hysteresis

controller for the DC output voltage. In this test, the hysteresis

voltage band was set to 380V ± 7.5V. A small part of the figure

was amplified (zoom picture) to analyze the output voltage

variation. It is observed that when the system reaches 372.5V, at

time Th1, the converter start to operate and the output voltage

increases at a constant slope. Once it reaches 387.5, at time Th2,

the PWM signals are turned off. However, the output voltage still

increases with a different slope between times Th2 and Th3. This



is caused by the energy transferred from the input inductors L1

and L2 through the snubber to the capacitor Cs. Once all the

energy is transferred the PWM signals stay OFF until the output

voltage decreases reaching again 372.5V. At this voltage level,

the system is restarted.

ZCS Turn off

MOSFET Q1 CurrentVDS

ZCS Turn on

Zoom

Th1 Th2 Th3

(a)

(b)

Figure 14. Voltage and current curves in the mosfet for two different operation

conditions: (a) Drain to source voltage and current showing ZCS condition for

a MOSFET: (b) Hysteresis control of the output voltage.

Because of the high frequency nature of the output

voltage supplied to the motor by this converter, it is not trivial to

measure its efficiency, at least without very expensive equipment.

Considering that in an induction motor only the fundamental

voltage and current components are converted into mechanical

power, we based our measurements only in the fundamental

output power. A first order filter was used to obtain the

fundamental component of the voltage and a digital oscilloscope

was used to obtain the output power of the converter. Fig. 15

presents the measured efficiency considering the input power

versus the fundamental power supplied to the motor. A maximum

efficiency of 93.64% was obtained for the TIBC, first stage

DC/DC converter, and 91% for the complete system. These

curves were obtained during a real operation with a PV panel,

driving the water pump with varying solar radiation. The

efficiencies take into account all the energy used for the control

electronics and drivers and are valid for a commercial prototype.

0.5

0.55

0.6

0.65

0.7

0.75

0.8

0.85

0.9

0.95

1

0 0.2 0.4 0.6 0.8 1

Effi

cie

ncy

(%

)

Input Power (pu)

TIBC and VSI efficiency x Power input

TIBC+VSI

TIBC

Input Power (pu)

Eff

icie

ncy

Figure 15. TIBC and VSI measured efficiency considering only fundamental

output power.

Nowadays, there are reports of more efficient

converters. During the design of the proposed converter, some

compromises were made to achieve high efficiency and low cost.

The semiconductor components were preferably cheaper than

efficient, since they are the most expensive components of the

converter. Also, the magnetic components were handmade,

though in the cost calculation commercial prices were used.

Fig. 16 presents the distribution of costs of the entire

system and of the prototype. From Fig. 16(a) it can be seen that

the converter takes only a small share (approximately 10%) of the

overall system cost. Fig. 16(b) shows the price distribution of the

converter itself. The overall price of the prototype considering

large scale production was U$ 90,90. This cost can be further

divided into U$52,28 for the VSI and U$38,62 for the TIBC. The

control system and housing were divided equally between VSI

and TIBC.

Motor+Pump

$280,00

(33.70 %)

PV Panel

$460,00

(55.36 %)

Converter

$90.90

(10.94 %)

Miscelaneous 16.52 %

Controller 4.00 %

Semiconductors 32.84 %

Regulators 10.59 %

PCB

5.5 %

Capacitors 16.03 %

Drivers IC 6.64 %

Magnetic 7.87 %

(a)

(b)

Figure 16. Price distribution of the (a) system, and (b) converter.

VII. CONCLUSION

In this paper, a converter for photovoltaic water

pumping and treatment systems without the use of storage

elements was presented. The converter was designed to drive a

three-phase induction motor directly from PV solar energy, and

was conceived to be a commercially viable solution having low

cost, high efficiency, and robustness. The paper presented the

system block diagram, control algorithm, and design. The

experimental results suggest that the proposed solution could be

a viable option after more reliability tests are performed to

guarantee its robustness.

ACKNOWLEDGMENT

The authors would like to thank the financial support

and motivation provided by the Federal University of Maranhão

(UFMA), Eletrobrás, CP Eletrônica, Elétrica Visão, Texas

Instruments, and IEEE-PELS.

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João Victor Mapurunga Caracas received the B.S. degree and the M.S. degree

in electrical engineering from the Federal University of Maranhao, Brazil, in

2010 and 2013, respectively. Currently he is working toward his Ph.D. degree at

the same University. He has experience in Electrical Engineering with emphasis

in Power Electronics and Renewable Energy Sources, with focus in

grid connected systems and control.

Guilherme de Carvalho Farias is currently finishing his Bachelor degree in

Electrical Engineering. His main research interests include microinverters for

photovoltaic applications and control of switched mode converters. He is

currently working on grid-tied microinverters at the Instituto de Energia Elétrica

(IEE).

Luis Felipe Moreira Teixeira is currently finishing his Bachelor degree in

Electrical Engineering. His main research interests include control of switched

mode converters, and on grid-tied microinverters.

Luiz Antonio de Souza Ribeiro received the M.Sc. and Ph.D. degrees from the

Federal University of Paraiba, Campina Grande, Brazil, in 1995 and 1998,

respectively. From 2004 to 2006, he was a Visiting Researcher at the University

of Wisconsin, Madison, working in sensorless control of permanent magnet

synchronous machines. Since 2008, he has been an Associate Professor at the

Federal University of Maranhão, São Luís, MA, Brazil. His main areas of interest

include electric machine and drives, power electronics, and power systems using

renewable energy sources.

implementation of a high efficiency, high lifetime and low cost converter for an autonomous...

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