Abstract

In this work, the design and experimental results of com-pact low-loss filters, based on the extension of the clas-sical coaxial waveguide resonator to Substrate IntegratedWaveguide technology, are successfully demonstrated.The design, fabrication and measurement of a conti-nuously tunable cavity resonator and a two-pole filter atS-band in single-layer SIW technology are presented.These structures keep the low-cost fabrication scheme ofsingle-layer PCB processing, while requiring less than halfthe area compared to a conventional SIW design.

Keywords: Tunable filters, varactor tuning diodes, com-bline resonator, compact filter design, microwave filter,continuous tuning, Substrate Integrated Waveguide(SIW).

1. Introduction

The concept of Substrate Integrated Waveguide (SIW)has been developed to synthesize waveguide cavitieswith planar structures into a single dielectric material,combining integrated high Q-factor filters with low costfabrication techniques. A SIW component consists on asynthetic waveguide formed by the top and the bottommetal plates of a dielectric slab and is bounded transver-sely by two sidewalls of cylindrical metallic via-holes. Therows of vias connect top and bottom metal layers for-ming the cavity walls. Input and output coupling is achie-ved using tapered microstrip transmission lines orcoplanar current probes. Adjusting the geometrical di-mensions of the taper or probes, the input/output cou-pling can be controlled.

The SIW components can be fabricated on printed circuitboards (PCBs) or low-temperature co-fired ceramics(LTCCs) technologies. They enable a significant reductionof size, weight and cost compared with metallic wave-guide designs. As the SIWs are based in the TEn0 modes,the characteristic of conventional waveguides are preser-ved and the propagation of energy of these modes is subs-tantially limited in the substrate. Thus, they have a higherQ-factor and lower loss than other planar guided-wavestructures, for instance microstrip lines and coplanar wa-veguides (CPW). In fact, conventional planar resonatorsprovide a moderate unloaded Q-factor, typically less thanQu

grown on electronically tunable SIW cavity resonators asan enabling technology for the design of very low-loss tu-nable filters [7, 8] that can be an alternative to filter banksin several applications.

In this paper we present a novel structure for implemen-ting high-Q frequency agile resonators based on com-bline SIW cavities loaded with GaAs varactor tuningdiodes. Following this concept, the design of compactand continuously tunable filters based on a combline to-pology is presented. The resonators and filters are fabri-cated in a low-cost microwave substrate using aconventional single-side PCB fabrication process and off-the-shelf GaAs varactors.

Some previous research has been very recently done inthe study of reconfigurable SIW structures [9-11], con-cerning mainly the design of discretely tunable filters. Afirst attempt of using perturbing conductive elements fortuning the resonant frequency of a SIW cavity was pro-posed in [9], although no experimental device was thenreported. More recently, a PIN diode-based discretely tu-nable SIW resonator consisting on switchable perturbingelements has been proposed [10]. Finally, a different ap-proach for center frequency tuning of multi-layer filtershas made use of relatively large piezoelectric actuatorsdisks with remarkable results, however at the price of amore sophisticated design [11].

In this paper we propose a continuously tunable SIW filterbased on a combline topology. Then, an S-band comblinefilter is successfully designed, fabricated and measured. Ex-perimental and simulation results are in good agreement,showing the potential advantages of this structure in termsof size and design flexibility, without increasing insertionlosses compared to its conventional SIW counterpart. Themain advantages of our structure are the following: con-tinuous tuning range, more compact design, negligiblepower consumption and low-cost fabrication process. Mo-reover, a model for the proposed resonator is considered,and a systematic procedure for the design of these minia-turized coupled resonator bandpass filters is presented.

2. Combline Tunable Resonator in SIWTechnology

2.1 Resonator StructureThe structure of the tunable combline SIW resonator isshown in Fig. 1. It consists on a square cavity (size a)where a conducting post has been inserted at the centerof the resonator using a plated via hole. A metal patchconnected to that post and separated from the SIW top

plate by a small gap is created on the top side of the ca-vity. So, the inner via is short-circuited at the bottom me-tallic plan and open ended at the top. A loadingcapacitance C0 is therefore established between the postand the top metal layer of the SIW through the fringingfields across the annular gap.

Thus, the proposed resonator can be seen as a square-circular coaxial waveguide cavity, and can be modeled asa piece of TEM mode transmission line short-circuited atone end and capacitively terminated at the other. Ascheme of the equivalent circuit is shown in Fig. 2.

Due to the previous model, the resonant frequency of theSIW combline resonator can be expressed as a functionof the substrate thickness l, the annular gap capacitanceC0 and the characteristic impedance of the rectangular-circular coaxial waveguide Z0.

In order to validate the proposed concept, we have stu-died the tuning range and unloaded quality factor of atunable combline SIW resonator. Firstly, a device about 4GHz has been designed using full-wave 3D simulation ofthe structure using HFSS. The dimensions of the designedresonator can be seen in Fig. 3. It is worth pointing outthat the excitation of the resonator is performed throughmagnetic coupling through CPW probes.

As shown, the resonant frequency fR is not only a func-tion of the width and length of the cavity, but also a func-

70 ISSN 1889-8297/Waves - 2012 - year 4

Figure 1. Drawing of the resonator, showing the con-ducting post and the floating and metal patches.

Figure 2. Equivalent circuit of the combline SIW resonatorloaded at the capacitive end with a tuning varactor diode.

The availability of high-Q tunable resonators is of pri-mary concern in the design of low-loss reconfigurable fil-ters which are called to be key components in the futuremicrowave communication subsystems.

tion of size a of the metallic patch, the isolating gap andthe diameter of the central conducting post. These pro-perties of the combline resonator can be used to reducethe resonant frequency, without a significant decreasingin the Q factor of the resonator.

Fig. 4(a) and 4(b) show the dependence of fR and unloa-ded Q0 factor on the dimensions of the central patch for

the proposed combline resonator on �r = 3.55 substrate.It is seen that a high percentage change of the fR can beobtained changing the length a between 2 and 3 mm,with low degradation of Q0 factor.

2.2 Tuning ElementThe resonant frequency of the filter can be controlled bychanging the loading capacitance C0 of the combline re-sonator as shown in Fig. 5. Thus, GaAs varactor diodeshave been inserted connected to the metal patch for mo-dulating the capacitance of the resonator.

A hyper-abrupt junction GaAs varactor has been used inthis work as tuning element. Particularly, we have chosenthe MA46H200 from MA-COM. It exhibits a capacitancevariation from 0.25 to 1 pF for a reverse bias voltage bet-ween 22 and 0 V respectively. The nominal Q−factor at50 MHz is 3000 giving an estimated series resistance of2.2 Ω.

An important advantage of using tuning varactors ins-tead of switching elements (i.e. PIN diodes) is the negli-gible power consumption of the device while operatingin reverse bias conditions. Moreover, the need of only onevaractor per cavity significantly reduces the complexity ofthe bias network.

2.3 Continuously Tunable Resonator The loading capacitance C0, that would initially includethe capacitance between the metal patch and the SIWtop plate, can be controlled by inserting a varactor con-necting both sides. In order to simplify the biasing andoptimizing the varactor RF performance, a floating metalisland (it can be seen in Fig. 5) is created between themetal patch and the SIW top plate.

The cathode of the varactor is then connected to this flo-

ating pad while a series capacitor is inserted between the

former and the top metal side of the cavity for providing

a low impedance path for the microwave signal. The se-

ries DC blocking capacitor has been chosen of 1.5 pF in

order to slightly reduce the capacitive effect of the varac-

tor and improve a little bit the unloaded Q0 of the reso-

71Waves - 2012 - year 4/ISSN 1889-8297

Figure 5. Drawing of the cross-section of the comblineresonator. The varactor is connected between the islandand the metal patch.

Figure 3. Layout of the designed resonator. Via holeand conductive post diameters are 0.5 and 0.3 mm res-pectively. Metal patch gap is 0.1 mm.

Figure 4. Change in resonant frequency fR (left) andQ0 (right) as a function of the size a of metallic patch.

nator. We have chosen to keep the varactor bias circuitas simple as possible, so a thin wire has been connectedfrom the floating metal island to a bias pad. Thus, a con-venient bias voltage can be applied between the floatingpad and the top metal of the SIW cavity by using this thinwire. DC bias and ground connections have been isolatedby using high value resistors (i.e. R>100 kΩ), specifically1 MΩ resistors have been used.

Then, internal ports have been created at the placeswhere the varactor and the fixed capacitance will be in-serted in order to combine 3D EM simulations with cir-cuital simulations using the equivalent models of thelumped components. The equivalent circuit model of thetunable combline resonator is shown in Fig. 6, includingparasitic components for the varactor diode and the DCblocking capacitance.

The resonator has been strongly under-coupled (i.e. the S21parameter has been kept below -20 dB) in order to enableus the accurate measurement of the unloaded Q factorfrom the transmission response of the device [12]. Simula-tions for different values of varactor capacitance have beencarried out. The simulated results are shown in Fig. 7.

The estimated unloaded Q0 ranges from 180 to 70 for acapacitance variation between 0.25 pF and 1.25 pF. Forthe same capacitance range, the resonance frequencychanges from 3.3 GHz to 2.8 GHz. Thus, an estimatedtuning range about 15% can be extracted from the ob-tained results. The nominal parasitic due to the varactorpackaging and the series capacitor have been includedin the simulations.

3. Filter Design in SIW Technology

3.1 Combline FilterFor high-Q filter, it is critical to achieve center frequencytuning without degrading the Q factor of the filter. Sincetunable lumped components are generally low Q at mi-crowave frequency, they will decrease the overall qualityfactor. The proposed tunable structure allows the possi-bility of employing low-Q varactors or high-Q MEMS va-ractors to achieve center frequency control, with slightlydegradation of the overall Q, as shown in the previouschapter.

The structure of a 2−pole combline filter in SIW techno-logy can be seen in Fig. 8. The filter consists of two tu-nable SIW resonators, coupled by irises. In fact, narrown-order bandpass filters can be implemented by insertinginductive windows coupling adjacent resonators.

Each resonator consists on a square/circular SIW cavity ofside a where a plated via hole has been inserted at thecenter of the resonator, where the electromagnetic fieldof the fundamental mode is maximum. This inner via isshort-circuited at the bottom side and connected to ametal patch electrically isolated from the top side througha small gap.

A capacitive effect is therefore established between thismetal patch and the top plane of the SIW. This capaci-tance value depends on the perimeter of the metal patch,the spacing of the gap and the thickness of the metalli-zation layer.

72 ISSN 1889-8297/Waves - 2012 - year 4

Figure 6. The equivalent circuit model of the tunableresonator used in circuit simulations.

Figure 7. Simulated results of the under-coupled tunableSIW resonator for different values of varactor capacitance.

Figure 8. Scheme of the 2−pole combline filter in SIWtechnology.

For the input/output couplings to the resonators, wehave maintained the same structure of previous comblineresonator. These are performed by means of coplanarwaveguide CPW probes as shown in Fig.8. Therefore, theexternal coupling can be therefore controlled by the ge-ometrical dimensions of the probes. The inter-resonatorscoupling is controlled by means of irises created on theadjacent walls. Since the diameter of the rows of viaholes is considered fixed, the coupling level can be ad-justed changing the size of the iris and the distance bet-ween iris via holes.

As shown, the proposed tunable SIW resonator can beseen as a TEM square-circular coaxial transmission line oflength h (i.e. the substrate height) short-circuited at oneend and capacitively terminated at the other. The suscep-tance � (�) of the coaxial resonator can be expressed as

[1]

Where, C0 is the total capacitance of the central metalpatch, Y0 the characteristic admittance of the TEM modeshort-circuited transmission line formed by the inner (cir-cular) and outer (rectangular) conductor, h the substratethickness and the phase constant of the coaxial line atfrequency � is

[2]

Where, �r is the dielectric permittivity of the substrate,while C0 is the speed of light in vacuum. Now, the reso-nant frequency of the SIW cavity can be computed fromthe equation � (�)=0, that is

[3]

The inductive contribution comes from the TEM modeshort-circuited resonator, which can be seen as a shortpiece of coaxial transmission line of length and charac-teristic admittance embedded into the dielectric materialof permittivity �r. For a circular inner conductor of dia-meter d and a square contour of side a, the characteristicadmittance can be well approximated when a>>d by [13]

[4]

Finally, from [14], the capacitance established betweenthe metal disk of radius and the SIW top metal can becomputed. It is worth mentioning, the thickness of themetallic layers is not considered in this expression.

3.2 Filter DesignThe synthesis and dimensioning of the tunable filter is per-formed by the classical approach for directly coupled reso-nator filters. Firstly, the low-pass prototype parameters g0 ,g1, … , gn+1 are computed from the desired response. From

these values, the external quality factor Qext and inter-re-sonator coupling coefficients Kij are obtained from thewell-known expressions [12]. Secondly, a SIW combline re-sonator at the center frequency of the filter is designed ta-king into account the dielectric constant �r and thesubstrate thickness h. The side of the square SIW cavity andthe diameter of the inner via will determine the Y0 value ofthe coaxial resonator. Additionally, the dimension of themetal patch and the isolating gap width control the loadingcapacitance C0.

The prototype of the combline bandpass filter can be mo-

deled using shunt resonators and frequency-invariant ad-mittance inverters, as shown in Fig. 9. Given a filterresponse with centre frequency and bandwidth, beingand the upper and lower cutoff frequencies, the loadingcapacitance and the admittance of the coaxial resonatorcan be obtained by mapping the resonator susceptanceand the admittance inverters of the bandpass filter withthose of the low-pass prototype. The former conditioncan be expressed as

[5]

Where b is the resonator slope parameter, ��and�c = 1 rad/s the angular frequency and cut-off frequencyof the low-pass prototype respectively. The level of b willimpact the unloaded Q factor of the resonator, and mustbe a trade-off between low losses, compactness and fe-asibility of the synthesized values of C0 and Y0.

Now, substituting (1) into (4) for the corresponding pass-band edge frequencies ��=± 1 �={��, �L}we canobtain

[6]

And

73Waves - 2012 - year 4/ISSN 1889-8297

Figure 9. Equivalent prototype of an Nth-order comblineresonator bandpass filter with frequency invariant admit-tance inverters.

� (�)=�C0-Y0 cot(�h)

� =� ��r

C0

�C0=Y0 cot h� ��r

C0

Y0= In ad��r

601.079

-1

� (�)=��b �

�c

�0

C0= b�

�0

cot�+cotL��cotL-�Lcot�

The proposed resonator can be seen as a square-circularcoaxial waveguide cavity, and can be modeled as a pieceof TEM mode transmission line short-circuited at one endand capacitively terminated at the other.

[7]

Finally, the admittance inverters can be computed fromthe low-pass prototype coefficients using the well-known expressions

[8]

For i = 1 to (n – 1), where YA is the admittance of theinput/output access ports.

Following this approach, a 2-pole Chebyshev filter cen-tered at 3.1 GHz with 5% fractional bandwidth will besynthesized. Once the SIW cavity resonator has been de-signed, the values of Qext and Kij are computed as a func-tion of the input/output CPW probe dimensions and thewidth of the coupling irises using 3D full-wave simula-tions with HFSS software, as shown in Fig. 10.

In order to match the desired response of the filter, theresonators are tuned from the reflection group delay res-

ponse of short-circuited cavities. For adjusting the centrefrequency, the dimensions of the central square post orthe size of the cavities are modified. Moreover, by chan-ging the diameter of the inner vias the central frequencycan be coarsely adjusted.

A final optimization of the filter dimensions is then per-formed for the minimum capacitance value of the tuningelement, combining 3D full-wave results with circuital si-mulations using the equivalent models of the lumpedcomponents. As previous exposed, in the 3D model ofthe circuit, internal ports have been created at the placeswhere the varactors 46H200 (from MA-COM) and thefixed capacitance will be inserted.

Since all the parameters are now determined, the simu-lated response of the designed filter can be obtainedusing ANSOFT Designer and is shown is shown in Fig. 11.The minimum capacitance of the varactor (0.25 pF) hasbeen chosen to run this simulation. The centre frequencyof the complete bandpass filter is 2.88 GHz and the in-sertion loss is estimated less than 1.5 dB.

4 Experimental Results

4.1 Fabrication and Measurement of Tunable SIWResonatorA prototype of the resonator has been fabricated using Ro-gers RT/Duroid 4003C laminate (�r = 3.55 and tan�= 2.7 · 10−3). A picture of the fabricated device includingthe biasing wires is shown in Fig. 12. The size of the reso-nator is 27 × 27 mm2, showing an important size reductioncompared to its equivalent conventional SIW counterpartdue to the capacitive loading of the metal patch.

The measured S21 parameters for the tunable comblineSIW resonator are shown in Fig. 13. A tuning range ofalmost 20% is observed when changing the reverse biasvoltage of the varactor from 2 to 22 V.

74 ISSN 1889-8297/Waves - 2012 - year 4

Y0= b�

�0

��+�L��cotL-�Lcot�

J0,1 =�

b ��0

YA

g0 g1Jn,n+1 =

�

b ��0

YA

gn gn+1

Ji,i+1 =�

1gi g i+1

b ��0

Figure 10. Simulated results of the group delay (left) asa function of dimensions of the coupling iris, and intensityof the electromagnetic field (right).

Figure 11. Simulated results of the design filter (thick)and ideal response (dashed) for a minimum varactor ca-pacitance of 0.25 pF.

The resonant frequency and the unloaded Q of the reso-nator as a function of the reverse bias voltage are shownin Fig. 14. As can be seen, the tunable Q of the resonatorranges from 40 to 150, being above 100 for the first200 MHz of frequency shift. The power consumption ofthe device is negligible along the whole tuning voltages(i.e. typically less than 1 �W).

Although, the percentage value of tuning range is similarto estimated value, the range of frequency has moved tolower frequencies, since it ranges from 3.1 GHz to2.6 GHz. The differences with the simulated results areattributed to an underestimation of the capacitance bet-ween the metal patch and the floating island where the

DC bias is applied. This effect slightly increases the totalloading capacitance, decreasing both the resonant fre-quency of the structure and the achievable tuning rangeof the resonator. Fabrication and permittivity tolerancescontribute also to the small shift to lower frequencies ofthe response. The small increase of losses between expe-rimental and simulated results is attributed to the moun-ting and soldering of the components.

4.2 Fabrication and Measurement of Tunable SIW 2-pole FilterTo validate our filter topology, a 2−pole bandpassChebyshev filter was designed, fabricated and measuredin 1.524 mm-thick Rogers RO4003C substrate. The fixedcombline filter is first designed with equiripple fractionalbandwidth of 5%, center frequency of 3.1 GHz while therequested in-band return losses are −20 dB. Simulationresults of the tunable filter for the whole varactor capa-citance range can be seen in Fig. 15.

The prototype was fabricated using standard single-layerPCB processing technology. The layout and dimensionsof the fabricated filter is depicted in Fig. 16. Filter size are55×27.5 mm2, while the equivalent conventional SIW fil-ter would require about twice the area for square cavitiesusing the dominant TE101 mode.

75Waves - 2012 - year 4/ISSN 1889-8297

Figure 12. Photography of the fabricated device undermeasurement.

Figure 14. Resonance frequency (right) and unloadedQ (left) of the tunable resonator as a function of the re-verse voltage applied to the varactor.

Figure 13. Simulated results of the under-coupled tunableSIW resonator for different values of varactor capacitance.

The input/output CPW lines present a trace width of1.3 mm and a ground plane spacing of 0.15 mm, so thecharacteristic impedance is closed to 50 Ω. The diameter ofthe vias for the SIW cavities and irises is 600 �m, and the

spacing between the contour vias is 1.25 mm. The centralvia hole presents a diameter of 300 �m and the metal patchisolating gap is 100 �m. Photography of the fabricated filterand the measured S-parameters are shown in Fig. 17.

Measured results agree quite well with the simulations.The device centre frequency can be tuned between 2.64and 2.88 GHz for a bias voltage ranging from 3.5 – 22 V.The insertion losses vary from 1.27 dB to 3.63 dB acrossthe whole tuning range. Moreover, a variation of the3−dB fractional bandwidth between 4 %−7 % has beenobserved due to the short length of the coaxial resonator.The tuning range of the proposed approach is mainly li-mited by the moderate Q-factor of the tuning elementsand the frequency dependence of the input/output andinter-resonators couplings. However, the more compactdesign would enable us to integrate several switchablereconfigurable filters in order to cover a broader fre-quency band. Moreover, the use of higher Q tuning ele-ments (e.g. MEMS switches and varactors) will allow anincrease of the tuning range without degrading the EMperformance of the device.

The presence of the varactor introduces non-linearity ef-fects that limit the power handling of the filter. Resultsof intermodulation distortion and linearity of the fabrica-ted filter can be seen in Fig. 18. It is worth mentioning

76 ISSN 1889-8297/Waves - 2012 - year 4

Main advantages of combline SIW tunable filter are thefollowing: continuous tuning range, negligible powerconsumption, more compact design and low-cost fabri-cation process.

Figure 15. Simulation results of the fabricated filter for avaractor capacitance ranging from 0.25 pF (f0 = 2.88 GHz)to 1 pF (f0 = 2.64 GHz).

Figure 17. (Left): Photography of the tunable filter during the reconfiguration time measurement. (Right): MeasuredS-parameters of the fabricated tunable 2-pole filter.

Figure 16. (Left): Photography of the tunable filter during the 1-dB compression point measurement. (Right): Layoutand dimensions of the fabricated tunable filter.

that these effects have been measured for the worst-caseof a high varactor capacitance.

Thus, a two-tone third-order intermodulation interceptpoint (IIP3) of +24 dBm has been measured for a bias vol-tage Vb = 5 V and a two-tone separation f = 50 kHz.However, the 1−dB compression point of the filter at thesame bias voltage is only +10 dBm.

The reconfiguration time of the filter has been measuredusing a crystal power detector at the filter output and a4 dBm tone centered at 2.85 GHz at the input. Then, thebias voltage has been changed between 20 and 14 Vwith a 100 �s-period square signal. The measurementsetup is shown in Fig. 17. The low-to-high and high-to-low measured reconfiguration times of the filter are 120and 720 ns for a centre frequency shift about 40 MHz.

5 Conclusions

A tunable combline SIW resonator has been proposed inthis paper. The capacitively loaded end of the resonatorhas been used in order to include a surface-mounted tu-ning varactor diode that changes the resonant frequencyof the device. A Q-factor better than 100 has been ob-tained for a 200 MHz tuning range from 3.1 GHz. Theproposed device can be fabricated using a low-cost PCBprocess and using off-the-shelf GaAs varactor diodes. Thestructure presents a continuous tuning range of almost20% and negligible power consumption.

The obtained results are very promising compared to typi-cal planar tunable resonators [3, 5, 6], while other SIWtunable cavities based on semiconductor tuning elementshave achieved unloaded Q’s around 100 on a discretelytunable approach [10]. Moreover, losses due to the tu-ning element could be significantly reduced by using RFMEMS varactors.

A continuously tunable filter has been designed, fabrica-ted and measured in SIW technology. Main advantagesof the proposed structures are analog tuning range, lowlosses and easy integration with other planar circuits. Ad-ditionally, due to the combline topology of the devices,

a very compact implementation is also achieved with avery wide spurious-free band. This approach shows inte-resting applications for the design of low-loss tunable fil-ters based on compact high-Q SIW resonators. Moreover,the application of this technology to the post-manufac-turing tuning of narrow-band SIW filters could be studiedshowing promising results. The design, performance andmanufacturability of novel combline filters in SIW tech-nology have been demonstrated. The measured resultsshow an excellent agreement with the EM simulationswhen all involved physical effects are considered. Theproposed topology presents important advantages interms of size compactness, spurious rejection and designflexibility.

Acknowledgments

The authors would like to thank Generalitat Valenciana andMICINN (Spanish Government) for its financial supportunder projects GV/2009/007 and TEC2010-21520-C04-01.

References

[1] D. Deslandes and K. Wu, “Single-substrate integra-tion technique of planar circuits and waveguide fil-ters”, IEEE Trans. Microwave Theory & Tech., vol. 51,no. 2, pp. 593-596, Feb. 2003.

[2] D. Packiaraj, V.S. Reddy, G.J. Mello, and A.T. Kalg-hatgi, “Electronically switchable suspended substratestripline filters”, in Proc. RF and Microwave Confe-rence, pp. 64-66, Oct 2004.

[3] I.C. Hunter, and J.D. Rhodes, “Electronically tunable mi-crowave bandpass filters”, IEEE Trans. Microwave The-ory & Tech., vol. 30, no. 9, pp. 1354-1360, Sept. 1982.

[4] J. Nath, D. Ghosh, J.P. Maria, A.I. Kingon, W. Fathel-bab, P.D. Franzon, and M.B. Steer, ”An electronicallytunable microstrip Bandpass filter using thin-film ba-rium-strontium-titanate (BST) varactors”, IEEE Trans.Microwave Theory Tech., vol. 53, no. 9, pp. 2707-2712, Sept. 2005.

[5] A. Pothier, J.-C. Orlianges, E. Zheng, C. Champeaux, A.Catherinot,D. Cros, P. Blondy, and J. Papapolymerou,”Low loss two bit tunable bandpass filters using MEMS

77Waves - 2012 - year 4/ISSN 1889-8297

Figure 18. (Left): Results of two-tone third-order intermodulation (thick) intercept point IIP3. (Right): Measured filteroutput power as a function of the input carrier level.

dc contact switches”, IEEE Trans. Microwave TheoryTech., vol. 53, no. 1, pp. 354 − 360, Jan. 2005.

[6] I. C. Reines, A. R. Brown, and G. M. Rebeiz, ”1.6-2.4GHz RF MEMS tunable 3-pole suspended comblinefilter”, in IEEE MTT-S Int. Microwave Symp. Dig.,Atlanta, GA, June 2008, pp. 133-136.

[7] D.W. Winter, and R. R. Mansour, ”Tunable DielectricResonator Bandpass Filter with Embedded MEMS Tu-ning Elements”, IEEE Trans. Microwave Theory Tech.,vol. 55, no. 1, pp. 154-159, Jan. 2007.

[8] F. Huang, and R.R. Mansour, ”Tunable Compact Die-lectric Resonator”, 39th European Microwave Confe-rence, 2009, EuMC 2009, Rome, Italy, pp. 559-562,Oct. 2009.

[9] J. C. Bohorquez, B. Potelon, C. Person, E. Rius, C.Quendo, G. Tanne, and E. Fourn, “Reconfigurableplanar SIW cavity resonator and filter,” in IEEE MTT-SInt. Dig., vol. 3, pp. 947–950, Jun. 2006.

[10] M. Armendariz, V. Sekar and K. Entesari, “TunableSIW bandpass filters with PIN diodes”, Proc. 40th Eu-ropean Microwave Conference, Paris, France, pp.830-833 Oct 2010.

[11] H. Joshi, H.H. Sigmarsson, S. Moon, D. Peroulis, andW.J. Chappell, ”High-Q fully reconfigurable tunablebandpass filters”, IEEE Trans. Microwave Theory Tech.,vol. 57, no. 12, pp. 3525-3533, Dec. 2009.

[12] J.S. Hong and M.J. Lancaster, “Microstrip Filter forRF/Microwave Applications“, New York, NY: JohnWiley & Sons, Inc., 2001.

[13] H. J. Riblet, “An accurate approximation of the im-pedance of a circular cylinder concentric with an ex-ternal square tube,” IEEE Trans. Microwave TheoryTech., vol. MTT-31, no. 10, pp. 841–844, Oct. 1983.

[14] H. S. Lee and H. J. Eom, “Potential distributionthrough an annular aperture with a floating innerconductor,” IEEE Trans. Microwave Theory Tech., vol.47, no. 3, pp. 372–374, Mar. 1999.

Biographies

Stefano Sirci was born in Torino,Italy. He received the B.S. and M.Sdegrees in Electronic Engineeringfrom the University of Perugia, Peru-gia, Italy, in 2006 and 2009, respec-tively. He is current working towardthe PhD degree at the Universitat Po-litècnica de València, Valencia, Spain.

Since 2009, he has been with the Microwave ApplicationGroup (GAM) at the Institute of Telecommunications andMultimedia Applications (iTEAM) at the Universitat Poli-tècnica de València. His research interest is currently fo-cused on designing, fabrication and measurement oftunable microwave filter in substrate integrated wave-guide (SIW) technology.

Jorge D. Martínez received the In-geniero de Telecomunicación andthe Doctor Ingeniero de Telecomu-nicación degrees from the Universi-tat Politècnica de València, Valencia,Spain, in 2002 and 2008. He joinedthe Department of Electronic Engi-neering of the Universitat Politècnicade València in 2002, where he is As-

sociate Professor since 2009. He was a Visiting Researcherat XLIM (Centre National de la Recherche Scientifique /Université de Limoges) from June 2006 to September2007, working on the design and fabrication of radio-fre-quency microelectromechanical systems (RF MEMS). Hiscurrent research interests are focused on emerging tech-nologies for reconfigurable microwave components withemphasis on tunable filters and RF MEMS.

Máriam Taroncher (S’03) was bornin Lliria, Valencia, Spain, on October8, 1979. She received the Telecom-munications Engineering degreefrom the Universidad Politécnica deValencia (UPV), Valencia, Spain, in2003, and is currently working to-ward the Ph.D. degree at UPV.

From 2002 to 2004, she was a Fellow Researcher withthe UPV. Since 2004, she has been a Technical Researcherin charge of the experimental laboratory for high powereffects in microwave devices at the Research InstituteiTEAM, UPV. In 2006 she was awarded a Trainee positionat the European Space Research and Technology Centre,European Space Agency (ESTEC-ESA), Noordwijk, TheNetherlands, where she worked in the Payload SystemsDivision Laboratory in the area of Multipactor, CoronaDischarge and Passive Intermodulation (PIM) effects. Hercurrent research interests include numerical methods forthe analysis of waveguide structures and the accelerationof the electromagnetic analysis methods.

Vicente E. Boria received the Inge-niero de Telecomunicación and theDoctor Ingeniero de Telecomunica-ción degrees from the UniversitatPolitècnica de València, València,Spain, in 1993 and 1997. In 1993he joined the Universitat Politècnicade València, where he is Full Profes-sor since 2003. In 1995 and 1996

he was held a Spanish Trainee position with the EuropeanSpace research and Technology Centre (ESTEC)-EuropeanSpace Agency (ESA). He has served on the Editorial Bo-ards of the IEEE Transactions on Microwave Theory andTechniques. His current research interests include nume-rical methods for the analysis of waveguide and scatte-ring structures, automated design of waveguidecomponents, radiating systems, measurement techni-ques, and power effects in passive waveguide systems.

78 ISSN 1889-8297/Waves - 2012 - year 4