microwave and laser ranging using photonic noise … · microwave and laser ranging using photonic...
TRANSCRIPT
MICROWAVE AND LASER
RANGING USING PHOTONIC
NOISE AND SEQUENCE CODING
DANIEL KRAVITZ
SUBMITTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE MASTER'S DEGREE
IN THE FACULTY OF ENGINEERING, BAR-ILAN UNIVERSITY
RAMAT-GAN, ISRAEL 2012
THIS RESEARCH WAS CARRIED OUT UNDER THE
SUPERVISION OF DR. AVI ZADOK FROM THE FACULTY OF
ENGINEERING AT BAR-ILAN UNIVERSITY
Acknowledgments
First and foremost, I would like to thank my advisor, Dr. Avi Zadok, for the
opportunity to work on this project. Since joining his group, Avi have inspired me both
as a researcher and most importantly as a person. His dedication to research even at
bizarre hours and the close relations with his students is not something to take for
granted. His patience and encouragement were invaluable to me throughout the course of
this research. For that, I am extremely grateful.
I would also like to thank Daniel Grodensky. We have been working closely
together on this research since the first day we joined Avi’s group. Daniel's vast
knowledge and experience were a key factor for this research success. I am truly grateful
that I had the opportunity to meet such a talented researcher and a very close friend.
I wish to thank all of Avi’s group members for their advices and help
(appearance by name): Alon Lehrer, David Elooz, Eyal Preter, Idan Bakish, Ofir
Klinger, Rafi Cohen, Ran Califa, Shahar Levi, Tali Ilovitsh, and Yair Antman.
Especially I would like to thank Dr. Arkady Rudnitsky for many suggestions and much
warranted help.
I would like to express my great appreciation to the Faculty of Engineering at
Bar-Ilan University, and especially Mrs. Dina Yeminy, for her support and help since
joining the Faculty 6 years ago. I would also like to thank the German-Israeli Foundation
(GIF) under Grant No. I-2219-1978.10/2009 for financially supporting this research.
In addition, I would also like to thank Prof. Nadav Levanon for collaborating
with us and introducing us the world of Radar signals and the noncoherent compression
approach in particular. He also provided me a great amount of insight and advice.
Last but not least, I thank my family. My parents, Nelly and Vadim, my sister
Katia, and my wife Alina, for their support, encouragement and great patience
throughout my study and all times.
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Table of Contents
Page
TABLE OF CONTENTS……………………………………………………………………………………....i
LIST OF FIGURES…………………………………………………………………………………………....iii
LIST OF SYMBOLS………………………………………………………………………………………….vii
LIST OF ABBREVIATIONS………………………………………………………………………………...x
LIST OF PUBLICATIONS………………………………………………………………………………....xii
ABSTRACT…………………………………………………………………………………………………....xiii
I. INTRODUCTION......................................................................................................... 1
1.1 UWB TECHNOLOGY ......................................................................................... 1
1.1.1 UWB impulse radar……………………………… ..................................... 2
1.1.2 UWB noise radar………………………………….. .................................... 3
1.2 MICROWAVE PHOTONICS OVERVIEW AND APPLICATIONS ................................. 6
1.2.1 Electro-optic modulators and photo-detectors ............................ 6
1.2.2 Radio-over-fiber and antenna remoting…… ................................. 8
1.2.3 Microwave photonic variable delay lines ………………...……………10
1.2.4 Microwave-photonic generation of UWB waveforms ................ 11
1.2.5 Photonic generation of UWB noise waveform ............................ 13
1.3 LASER RADAR (LADAR) .............................................................................. 14
1.3.1 Incoherent and coherent LADAR systems .................................. 15
1.3.2 LADAR applications……………………………….. ................................ 16
1.3.3 Common ranging measurements techniques ............................. 16
1.3.4 Sources of noise in LADAR systems………… ................................ 19
1.3.5 Pulse Compression………………………………… ................................. 21
1.3.6 Phase codes…………………………………………. .................................. 22
1.3.7 Complementary pair……………………………… ................................. 24
1.3.8 LADAR system based on incoherent pulse Compression .......... 26
II. MICROWAVE PHOTONIC UWB RADAR SYSTEM................................................ 28
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2.1 PHOTONIC NOISE GENERATION...................................................................... 28
2.1.1 UWB noise generation via ASE of SBS in optical fiber ............... 28
2.1.2 Broadening the spectrum of the pump-laser diode ................... 30
2.1.3 UWB noise generation via ASE of EDFA… ................................... 34
2.2 SET-UP OF MICROWAVE-PHOTONIC RADAR SYSTEM ....................................... 35
2.2.1 Correlation domain analysis, match-filtering and side lobe
supression…………………………………………… ................................. 36
2.3 RADAR LINK BUDGETS .................................................................................... 38
2.4 EXPERIMENTAL RESULTS ............................................................................... 43
2.4.1 UWB noise radar range measurements based on SBS-ASE ........ 44
2.4.2 UWB noise radar range measurements based on EDFA-ASE .... 48
2.5 CONCLUDING REMARKS .................................................................................. 50
III. LADAR WITH INCOHERENT PULSE COMPRESSION ........................................ 52
3.1 CODING PRINCIPLE ......................................................................................... 52
3.1.1 Coding procedure…………………………………. ................................... 53
3.1.2 Simulated sidelobe suppression………………… .............................. 56
3.1.2.1 MPSL 82 ............................................................................ 56
3.1.2.2 MPSL 1112 ....................................................................... 58
3.1.2.3 416 complementary pair ................................................ 60
3.1.2.4 832 complementary pair ................................................ 62
3.1.3 Experimental sidelobe suppression…………… .............................. 63
3.2 LADAR LINK BUDGET .................................................................................... 68
3.3 LASER RANGE-FINDER MEASUREMENTS ......................................................... 72
3.4 CONCLUDING REMARKS .................................................................................. 74
IV. DISCUSSION AND CONCLUSIONS ...................................................................... 75
APPENDIX A .......................................................................................................... 79
A.1 MPSL 82 CODE ............................................................................................. 79
A.2 MPSL 1112 CODE ......................................................................................... 79
BIBLIOGRAPHY .......................................................................................................... 80
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List of Figures
Page
FIGURE 1. 1. SIMPLIFIED BLOCK DIAGRAM OF CORRELATION-BASED NOISE RADAR. ................. 5
FIGURE 1. 2. SIMPLIFIED BLOCK DIAGRAMS FOR 3 MAIN ROF TECHNIQUES. LEFT: RF-OVER-
FIBER TRANSPORT SCHEME. CENTER: IF-OVER-FIBER TRANSPORT SCHEME. RIGHT:
BASEBAND-OVER-FIBER TRANSPORT SCHEME. IFf : INTERMEDIATE FREQUENCY.
LOf :LOCAL OSCILLATOR FREQUENCY. CO: CENTRAL OFFICE. BS: BASE STATION. E/O:
ELECTRICAL TO OPTICAL. O/E: OPTICAL TO ELECTRICAL. ........................................... 9
FIGURE 1. 3. BLOCK DIAGRAMS OF INCOHERENT (TOP) AND COHERENT (BOTTOM) LADAR
SYSTEMS................................................................................................................. 15
FIGURE 1. 4. LEFT: BLOCK DIAGRAM OF TOF LADAR RANGE FINDER [61]. RIGHT: BLOCK
DIAGRAM OF PHASE-SHIFT LADAR RANGEFINDER [61]. D: DISTANCE. PIN: PIN PHOTO-
DIODE. OSC.RF: RF OSCILLATOR. OSC.OL: LOCAL OSCILLATOR. IFf : INTERMEDIATE
FREQUENCY. IFf : PASS-BAND FILTER BANDWIDTH. : PHASE DIFFERENCE. ............ 18
FIGURE 1. 5. BLOCK DIAGRAM OF LFM BASED LADAR RANGE FINDER . ............................. 19
FIGURE 2. 1. (TOP) DETERMINISTIC COMPONENT OF THE DIRECT-MODULATION CURRENT OF
THE PUMP LASER ( )Di t (2.8), i =7.6 MA, AND T = 100 NS. (CENTER)
CORRESPONDING RELATIVE OUTPUT POWER. (BOTTOM) SIMULATED INSTANTANEOUS
OPTICAL FREQUENCY VARIATIONS. aC = 0.25 GHZ/MA,1,thC = 0.15 GHZ/MA,
2,thC =
0.48 GHZ/MA, AND 1,2 = 20, 200 NS. .................................................................... 32
FIGURE 2. 2. SETUP FOR SBS-ASE NOISE GENERATION. FO: FIBER-OPTIC. BPF:BAND-PASS
FILTER. .................................................................................................................. 33
FIGURE 2. 3. TOP – PSD OF THE DIRECTLY MODULATED SBS PUMP WAVE. BOTTOM – PSD OF
THE DOWN-CONVERTED SBS-ASE NOISE .................................................................. 33
FIGURE 2. 4 SIMPLIFIED BLOCK DIAGRAM OF A MICROWAVE-PHOTONIC, UWB NOISE RADAR
SYSTEM. TX: TRANSMITTER. RX: RECEIVER ............................................................... 36
FIGURE 2. 5. DEFINITIONS OF PSLR, ISLR AND RESOLUTION OF THE IMPULSE RESPONSE OF
UWB NOISE RADAR WAVEFORMS. ............................................................................ 37
FIGURE 2. 6. A SIMULATED AUTO-CORRELATION OF A UWB NOISE WAVEFORM WITH A 1 GHZ-
WIDE UNIFORM PSD. ............................................................................................... 38
FIGURE 2. 7. SIMPLIFIED BLOCK DIAGRAM OF A MICROWAVE-PHOTONIC, UWB NOISE RADAR
SYSTEM. FO: FIBER-OPTIC. ....................................................................................... 40
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FIGURE 2. 8. PSLR AS FUNCTION OF ESNR FOR A GAUSSIAN DISTRIBUTED NOISE OF
1B GHZ. ............................................................................................................ 43
FIGURE 2. 9. HISTOGRAM OF THE DOWN-CONVERTED SBS-ASE NOISE (BAR), ALONGSIDE A
ZERO-MEAN GAUSSIAN DISTRIBUTION OF EQUAL VARIANCE (LINE). ............................. 45
FIGURE 2. 10. SBS-ASE IMPULSE RESPONSE ( 1B GHz 0 3.5f GHz ). ........................ 45
FIGURE 2. 11. MEASURED CORRELATION BETWEEN SBS-ASE NOISE WAVEFORM REFLECTED
FROM A METAL TARGET AND A REFERENCE REPLICA. THE DISTANCES TO THE TARGET WERE
1.5 M (BLUE, DOTTED); 1.95 M (RED, SOLID) AND 2.85 M (BLACK, DASHED). .............. 46
FIGURE 2. 12. TWO TARGETS SEPERATED BY 15 CM. ....................................................... 46
FIGURE 2. 13. CROSS-CORRELATION BETWEEN THE WAVEFORM REFLECTED FROM A PAIR OF
METALLIC TARGETS SEPARATED BY 15 CM, AND THE CENTRAL-OFFICE REFERENCE
WAVEFORM. THE TARGETS ARE RESOLVED BY THE RADAR SYSTEM. .............................. 47
FIGURE 2. 14. CROSS-CORRELATION BETWEEN THE WAVEFORM REFLECTED FROM A 20 CM-
LONG, CONCEALED METALLIC OBJECT AND THE CENTRAL OFFICE REFERENCE WAVEFORM.
............................................................................................................................. 48
FIGURE 2. 15. TOP – PSD OF THE DOWN-CONVERTED EDFA-ASE NOISE. BOTTOM –
HISTOGRAM OF THE DOWN-CONVERTED EDFA-ASE NOISE (BAR), ALONGSIDE A ZERO-
MEAN GAUSSIAN DISTRIBUTION OF EQUAL VARIANCE (LINE). ...................................... 49
FIGURE 2. 16. IMPULSE RESPONSE OF A UWB RF NOISE GENERATED THROUGH EDFA-ASE
( 2B GHz 0 3.5f GHz ). ................................................................................... 49
FIGURE 2. 17. MEASURED CORRELATION BETWEEN EDFA-ASE NOISE WAVEFORM
REFLECTED FROM A METAL TARGET AND A REFERENCE REPLICA. THE DISTANCES TO THE
TARGET WERE 1.43 M (RED, SOLID); 1.9 M (GREEN, DASHED) AND 2.16 M (BLUE,
DASHED). ............................................................................................................... 50
FIGURE 3. 1. TRANSMITTED CODE T (TOP) AND MATCHED FILTERING CODE R
(BOTTOM),CORRESPONDING TO TO THE BARKER 13 CODE: [+1 +1 +1 +1 +1 -1 -1 +1 +1
-1 +1 -1 +1]. .............................................................................................................................. 54
FIGURE 3. 2. TOP – APERIODIC AUTO-CORRELATION OF THE BARKER 13 BIPOLAR CODE:
[+++++--++-+-+]. THE CORRELATION PEAK IS 13, WHEREAS THE MAXIMAL SIDELOBE
EQUALS UNITY. CENTER – APERIODIC AUTO-CORRELATION OF A UNIPOLAR
REPRESENTATION OF THE BARKER 13 CODE: [1111100110101], SHOWING A WEAKER
CENTRAL PEAK AND INFERIOR SIDELOBE SUPPRESSION. BOTTOM – APERIODIC CROSS-
CORRELATION BETWEEN THE TRANSMITTED CODE T AND MATCHED FILTERING CODE R
CORRESPONDING TO THE BARKER 13 BIPOLAR CODE (SEE FIG. 3.1). WITH THE EXCEPTION
OF THE TWO TIME SLOTS IN THE IMMEDIATE VICINITY OF THE CENTRAL PEAK, THE
SUPPRESSION OF SIDELOBES REACHES THAT OF THE ORIGINAL BIPOLAR SEQUENCE . ......... 55
FIGURE 3. 3. TRANSMITTED CODE T (TOP) AND MATCHED FILTERING CODE R (CENTER),
CORRESPONDING TO THE MPSL 82 CODE. BOTTOM – APERIODIC CROSS-CORRELATION
BETWEEN THE TRANSMITTED CODE T AND MATCHED FILTERING CODE R . ........................ 56
v
FIGURE 3. 4. TRANSMITTED CODE T (TOP) AND MISS-MATCHED FILTERING CODE R (CENTER
),CORRESPONDING TO THE MPSL 82 CODE. BOTTOM – APERIODIC CROSS-CORRELATION
BETWEEN THE TRANSMITTED CODE T AND MATCHED FILTERING CODE R . ........................ 57
FIGURE 3. 5. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 82 PULSES-LONG
UNIPOLAR SEQUENCE WITH A SIGNAL-TO-NOISE RATIO OF -20 DB. BOTH MATCHED (LEFT,
BLUE) AS WELL AS MISMATCHED (RIGHT, BLACK) FILTERS WERE USED IN THE
COMPRESSION PROCESS. ............................................................................................................. 58
FIGURE 3. 6. CROSS-CORRELATIONS OF INCOHERENTLY COMPRESSED, 82 PULSES-LONG
UNIPOLAR SEQUENCES. BOTH MATCHED (LEFT, BLUE) AS WELL AS MISMATCHED (RIGHT,
BLACK) FILTERS WERE USED IN THE COMPRESSION PROCESS. TOP ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF +20 DB. CENTER ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF 0 DB. BOTTOM ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF -5 DB (SEE TEXT). ................................ 58
FIGURE 3. 7. TRANSMITTED CODE T (TOP) AND MATCHED FILTERING CODE
R (CENTER),CORRESPONDING TO THE MPSL 1112 CODE. BOTTOM – APERIODIC CROSS-
CORRELATION BETWEEN THE TRANSMITTED CODE T AND MATCHED FILTERING CODE R . 59
FIGURE 3.8. TRANSMITTED CODE T (TOP) AND MISS-MATCHED FILTERING CODE (CENTER)
CORRESPONDING TO THE R (CENTER), MPSL 1112 CODE. BOTTOM – APERIODIC CROSS-
CORRELATION BETWEEN THE TRANSMITTED CODE T AND MISS-MATCHED FILTERING CODE
R ................................................................................................................................................... 59
FIGURE 3. 9. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 1112 PULSES-LONG
UNIPOLAR SEQUENCE. BOTH MATCHED (LEFT, BLUE) AS WELL AS MISMATCHED (RIGHT,
BLACK) FILTERS WERE USED IN THE COMPRESSION PROCESS. TOP ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF +20 DB. CENTER ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF 0 DB. BOTTOM ROW: SIMULATED
COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF -20 DB (SEE TEXT). ............................. 60
FIGURE 3. 10. TRANSMITTED CODE T (TOP) AND MATCHED FILTERING CODE R (CENTER
),CORRESPONDING TO THE 416 BITS-LONG COMPLEMENTARY PAIR CODE. BOTTOM –
APERIODIC CROSS-CORRELATION BETWEEN THE TRANSMITTED CODE T AND MATCHED
FILTERING CODE R . .................................................................................................................... 61
FIGURE 3. 11. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 416 PULSES-LONG
COMPLEMENTARY CODE. TOP ROW: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE
RATIO OF +20 DB. CENTER ROW: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE
RATIO OF 0 DB. BOTTOM ROW: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE
RATIO OF -20 DB (SEE TEXT). ................................................................................................... 62
FIGURE 3.12. TRANSMITTED CODE T (TOP) AND MATCHED FILTERING CODE R (CENTER
),CORRESPONDING TO THE 832 BITS-LONG COMPLEMENTARY PAIR CODE. BOTTOM –
APERIODIC CROSS-CORRELATION BETWEEN THE TRANSMITTED CODE T AND MATCHED
FILTERING CODE R . .................................................................................................................... 63
FIGURE 3. 13. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 832 PULSES-LONG
COMPLEMENTARY CODE. TOP: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE RATIO
OF +20 DB. CENTER: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF 0 DB.
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BOTTOM: SIMULATED COMPRESSION WITH A SIGNAL-TO-NOISE RATIO OF -20 DB (SEE
TEXT)............................................................................................................................................ 63
FIGURE 3. 14. EXPERIMENTAL SETUP FOR LADAR MEASUREMENTS USING INCOHERENT PULSE
COMPRESSION. MZM: MACH-ZEHNDER MODULATOR. PC: POLARIZATION CONTROLLER.
EDFA: ERBIUM-DOPED FIBER AMPLIFIER. BLACK SOLID LINES DENOTE FIBER
CONNECTIONS, BLUE DASHED LINES REPRESENT ELECTRICAL CABLES, AND ORANGE DASH-
DOTTED LINES DESCRIBE FREE-SPACE PROPAGATION ............................................................. 65
FIGURE 3.15. EXPERIMENTAL SETUP OF INCOHERENT PULSE COMPRESSION ....................... 66
FIGURE 3. 16. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 1112 PULSES-LONG
UNIPOLAR SEQUENCE. BOTH MATCHED (LEFT, BLUE) AS WELL AS MISMATCHED (RIGHT,
BLACK) FILTERS WERE USED IN THE COMPRESSION PROCESS. TOP ROW: SIMULATED
COMPRESSION OF NOISE-FREE SEQUENCES. CENTER ROW: COMPRESSION OF
EXPERIMENTALLY OBTAINED LADAR ECHOES, DETECTED WITH A SIGNAL-TO-NOISE RATIO
OF +20 DB. BOTTOM ROW: COMPRESSION OF EXPERIMENTALLY OBTAINED LADAR
ECHOES, DETECTED WITH A SIGNAL-TO-NOISE RATIO OF -20 DB (SEE TEXT). .................... 67
FIGURE 3. 17. CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED, 832 PULSES-LONG
COMPLEMENTARY CODE. TOP: SIMULATED COMPRESSION OF NOISE-FREE SEQUENCES.
CENTER: COMPRESSION OF EXPERIMENTALLY OBTAINED LADAR ECHOES, DETECTED WITH
A SIGNAL-TO-NOISE RATIO OF +30 DB. BOTTOM ROW: COMPRESSION OF EXPERIMENTALLY
OBTAINED LADAR ECHOES, DETECTED WITH A SIGNAL-TO-NOISE RATIO OF -20 DB (SEE
TEXT)............................................................................................................................................ 68
FIGURE 3. 18. EXPERIMENTAL CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED,
1112 PULSES-LONG UNIPOLAR SEQUENCE FOR REFLECTOR PLACED AT 6M AWAY FROM
THE COLLIMATING LENS AT SIGNAL TO NOISE RATIO OF 20 DB. TOP: COMPRESSION USING
MATCHED FILTER. BOTTOM: COMPRESSION USING MISMATCHED FILTER. ............................ 71
FIGURE 3. 19. EXPERIMENTAL CROSS-CORRELATIONS OF AN INCOHERENTLY COMPRESSED,
1112 PULSES-LONG UNIPOLAR SEQUENCE FOR REFLECTOR PLACED AT 50M AWAY FROM
THE COLLIMATING LENS AT SIGNAL TO NOISE RATIO OF 18 DB. TOP: COMPRESSION USING
MATCHED FILTER. BOTTOM: COMPRESSION USING MISMATCHED FILTER............................. 72
FIGURE 3. 20. CROSS-CORRELATIONS OF INCOHERENTLY COMPRESSED, 1112 PULSES-LONG
UNIPOLAR LADAR ECHOES. THE DISTANCES BETWEEN THE LADAR LENS AND A RETRO-
REFLECTOR WERE 50 M (BLUE, DASHED) AND 50.025 M (RED, SOLID). THE
MEASUREMENT SNR WAS 18 DB. A MISMATCHED FILTER WAS USED IN THE COMPRESSION.
...................................................................................................................................................... 73
vii
List of symbols
IFf Intermediate frequency
LOf Local oscillator frequency
Phase difference
B Bandwidth
T Duration (in chapter 2)
M Number of bits
bt Bit duration
L Code length
B Brillouin shift frequency
0g Brillouin gain coefficient
PI Pump power
B SBS inherent linewidth
s SBS central frequency
p Pump central frequency
Planck constant
effL Fiber effective length
Loss
aC Adiabatic chirp coefficient
( )i t Injection current
viii
n Thermal time constant
( )thh t Thermal impulse response
N Noise RMS bandwidth
NB Noise bandwidth
( )Ni t Noise current
( )Di t Deterministic current
spn Spontaneous emission factor
G EDFA power gain
F EDFA noise figure
( )s t Signal
( )h t Impulse response
D Spatial resolution
c Speed of light
PDP Photo-detector noise power
RINP RIN noise power
DCP DC power
tG Antenna gain
R Distance
eA Antenna effective aperture
Target cross section
ix
tP Transmitted power
rP Received power
Wavelength
a Width dimension
b Height dimension
Angle of incident
modP Modulation power
modi Modulation current
remoteP Remoted power
T Transmitting signal (in chapter 3)
R Matched reference (in chapter 3)
R Miss-matched reference
c Binary phase code (in chapter 3)
N Code length
tD Circular receiver aperture
t Target reflectance parameter
dA Target surface area
t Laser beam angular divergence
R Target surface angular dispersion
TESTN Number of independent recordings
x
List of Abbreviations
AGC Automatic Gain Control
APD Avalanche Photo-Diode
ASE Amplified Spontaneous Emission
BS Base Station
BW Bandwidth
CO Central Office
CW Continuous Wave
DC Direct Current
DFB Distributed Feedback Laser
ESNR Electrical Signal to Noise Ratio
E/O Electrical to Optical
EDFA Erbium Doped Fiber Amplifier
FCC Federal Communication Commission
FBG Fiber Bragg Grating
FO Fiber Optic
FWHM Full Width at Half Maximum
FOV Field of View
IF Intermediate Frequency
InGaAsP Indium Gallium Arsenide Phosphide
ISLR Integrated Sidelobe Ratio
LiNbO3 Lithium Niobate
LIDAR Light Detection and Ranging
LADAR Laser Radar
LO Local Oscillator
LFM Linear Frequency Modulation
xi
MPSL Minimum Peak Sidelobe Level
NEP Noise Equivalent Power
O/E Optical to Electrical
OSNR Optical Signal to Noise Ratio
PIN P type – Intrinsic – N type
PSLR Peak to Sidelobe Ratio
PSD Power Spectral Density
RADAR Radio Detection and Ranging
RIN Relative Intensity Noise
RF Radio Frequency
RoF Radio over Fiber
RMS Root Mean Square
SNR Signal to Noise Ratio
SOA Semiconductor Optical Amplifier
SBS Stimulated Brillouin Scattering
SFDR Spurious Free Dynamic Range
ToF Time of Flight
TTD True Time Delay
UWB Ultra Wide Band
WDM Wavelength Division Multiplexing
WLAN Wireless Local Area Network
XGM Cross Gain Modulation
XPM Cross Phase Modulation
2D Two Dimensional
3D Three Dimensional
xii
List of publications
Book Chapter:
1. A. Zadok, D. Grodensky, D. Kravitz, Y. Peled, M. Tur, X. Wu and A. E.
Willner, “Ultra-Wideband Waveform Generation Using Nonlinear Propagatio in
Optical Fibers,” in Ultra Wideband Communications: Novel Trends - Antennas
and Propagation, Mohammad Matin (Ed.), Intech, 2011.
Journal papers:
1. D. Grodensky, D. Kravitz, and A. Zadok, “Ultra-Wideband Microwave-
Photonic Noise Radar Based on Optical Waveform Generation,” IEEE
Photonics Technology Letters, vol. 24, pp. 839–841, 2012.
2. D. Kravitz, D. Grodensky, N. Levanon, and A. Zadok, “High-Resolution Low-
Sidelobe Laser Ranging Based on Incoherent Pulse Compression,” IEEE
Photonics Technology Letters vol. 24, pp. 2119-2121, 2012.
Conference papers:
1. D. Grodensky, D. Kravitz, and A. Zadok, "Ultra-Wideband Noise Radar Based
on Optical Waveform Generation," Proceedings of IEEE Conference on
Microwaves, Communications, Antennas and Electronic Systems (COMCAS)
2011, Tel-Aviv, Israel, Dec. 2011.
2. D. Grodensky, D. Kravitz, and A. Zadok, “Ultra-Wideband Noise Radar Based
on Optical Waveform Generation,” Proceedings of SPIE conference on Radar
Sensor and Technology, SPIE, Defense Security and Sensing 2012, Baltimore
MD. Proc. SPIE 8361, 8361-43, 2012
xiii
Abstract
High resolution ranging systems are of great importance for both civilian and
military applications. Both radio frequency (RF) waveforms and optical waveforms (in
laser detection and ranging, or LADAR) are used for range detection purposes. Use of
each spectral range has its advantages and drawbacks. While radars are less sensitive to
alignment and atmospheric disturbance, optical waveforms can carry broader bandwidth
signals and thus give better range resolution, are inherently immune to electromagnetic
interference, and are readily integrated with fiber-optic distribution.
In both techniques, high range resolution can be obtained using short and intense
pulses. However, the transmission and processing of such pulses is difficult and
potentially unsafe. In addition, the overall signal energy falls off with the use of short
pulses, and the signal to noise ratio (SNR) is thus degraded. Instead, temporally extended
waveforms or sequences, in conjunction with proper compression techniques at the
receiver end, may be used. The auto-correlation, or matched-filtering, of long waveforms
and sequences effectively compresses their entire energy into an intense and narrow
virtual peak with low residual sidelobes. Such sequences may therefore reproduce the
high resolution and low background that are provided by a short and high-power single
pulse, with significant added values: The instantaneous power of the extended
waveforms can be orders-of-magnitude lower, making them safer and simpler to
generate in a real-world system and more difficult to intercept by an adversary.
The objective of this work is high resolution ranging measurements using
photonic techniques. To that end two distinct schemes are proposed and demonstrated,
both involving the compression of long waveforms. The first proposition is a
microwave-photonic, ultra-wideband (UWB) noise RADAR system. The system makes
xiv
use of the amplified spontaneous emission (ASE) of optical amplifiers to generate
random 'physical noise', which is later down converted to the RF domain. Photonic
techniques provide convenient integration with radio-over-fiber (RoF) distribution, a
broad bandwidth, and flexible reconfiguration. Use of noise, rather than deterministic
UWB waveforms, provides better immunity against interception and jamming by an
adversary.
An experimental demonstration of this system relied on the ASE of either
stimulated Brillouin scattering (SBS) or erbium-doped fiber amplifiers (EDFA) as
sources of photonic noise. The system comprised of a central unit, in which noise was
generated and processed, and a remote end unit where the transmitting and receiving RF
antennas were located. The central and remote units were connected by a 10 km-long
bidirectional fiber optic link. A spatial accuracy of 10 cm was obtained experimentally.
The range of the system, subject to our equipment constraints, was estimated as 15 m.
The detection of a 20 cm-long concealed metallic object was demonstrated as well.
The second technique is a LADAR system, employing an encoded sequence of
pulses and a proper post-processing to obtain high resolution ranging measurements with
low sidelobes. In most scenarios, effective compression requires phase coding, whereas
intensity coding leads to inferior performance. The measurement of phase in a photonic
system, however, involves complicated coherent receivers. Alternatively, I employ a
novel incoherent compression scheme, which was previously proposed by Prof. Nadav
Levanon of Tel-Aviv University. In this scheme, binary phase sequences are converted
to a unipolar, intensity modulation representation through a position-coding algorithm,
and then used to modulate the laser ranging source. Reflected echoes undergo simple
direct detection, followed by correlation with a bipolar reference sequence that is
digitally stored at the receiver. Even though both transmit and receive operation are
xv
incoherent, the filtered sequence nearly replicates the effective sidelobe suppression of
the original phase code.
Laser range finding using incoherent compression was also demonstrated
experimentally. Unipolar representations of two types of codes were examined: 1)
minimum peak to sidelobe (MPSL) sequences, and 2) complementary code pairs, whose
correlation sidelobes cancel each other. A ranging resolution of 3 cm was demonstrated
using this system. The range to a target could be observed at poor SNRs, as low as -20
dB. The measurement range depends on the reflectance of targets, and could reach
hundreds of meters with modest averaging times. The results provide the first successful
experimental demonstration of the incoherent compression principle.
The work is organized as follows. A brief and general introduction to UWB noise
radars, microwave photonics waveform generation, sequence compression and LADAR
systems is given in chapter I. The UWB noise microwave photonic radar system is
described in chapter II. Chapter III is dedicated to a laser ranging system based on
incoherent pulse compression, and concluding remarks are provided in chapter IV.
1
CHAPTER
1
Introduction
1.1 UWB Technology
The majority of traditional radio systems use a narrow band of signal
frequencies, modulating a sinusoidal carrier wave. The reason is simple: sine waves are
generated by elementary and widespread oscillatory systems, and their tuning and
selection are straight forward. Therefore, frequency-domain multiplexing is the basic
way of information channel division in radio engineering. The majority of radio systems
occupy a bandwidth that is much smaller than their carrier frequency. The theory and
practice of modern radio engineering are based on this property. Narrowband signals
limit the information capability of radio systems, because the amount of the information
transmitted per second scales with their bandwidth. Increasing the system’s information
capacity requires either a broader bandwidth, or simply longer durations of transmission.
An alternative paradigm for radio transmission is the transition to signals with
wide and ultra-wide bandwidths (UWBs). UWB signals are free of sine-wave carriers,
and their power spectral density is low. These characteristics provide UWB radio with
unique advantages: improved immunity to multi-path fading, increased ranging
resolution, large tolerance to interfering legacy systems, enhanced ability for penetrating
obstacles, and low electronic processing complexity at the receiver [1]. According to the
2
United States Federal Communication Commission (FCC), UWB waveforms are defined
as signals with a bandwidth that is greater than twenty percent of their central frequency,
or as signals with bandwidth more than 500MHz, whichever is less [2]. UWB
technology is considered attractive for ground-penetrating and border-security radars,
high accuracy localization, precision navigation, and through-the-wall imaging [1]. In
this thesis a UWB microwave-photonic noise radar system for ranging measurements
will be explored.
1.1.1 UWB impulse radar
In general, radar systems transmit electromagnetic energy into a specific volume
in space, in search for targets. Objects (targets) within a search volume reflect portions of
this energy, (known as radar returns or echoes), back towards the receiver of the radar
system. These echoes are then processed by the radar receiver to extract target
information such as range, velocity, angular position, and other target-identifying
characteristics.
The spatial resolution of a radar system is inversely proportional to the
bandwidth of the transmitted waveforms [3]. Therefore, UWB technology gained
widespread acceptance in radars. Many UWB radar implementations rely on impulse
radio: the transmission of tailored short pulses and their subsequent coherent detection
[4]. The distance to a target is extracted based on a time-of-flight (ToF) estimation: the
receiver calculates the time it took a single pulse to make a round trip to the target and
back.
The process of radar-tracking and surveillance with impulse UWB signals differs
considerably from similar processes when using traditional narrowband signals.
Narrowband signals (i.e., sinusoidal and quasi-sinusoidal signals) have a unique property
3
of keeping a shape identical to that of the original function, and may differ only in their
amplitude, time shift, or phase. On the contrary, the UWB signal has a non-sinusoidal
waveform that can change shape in the process of propagation and reflection, in forms
such as addition, subtraction, differentiation and integration. One example of shape
change of UWB signal occurs during pulse radiation, since the intensity of radiated
electromagnetic field varies proportionally with the derivative of the antenna current.
Another pulse shape change occurs in scattering from targets whose spatial extent might
exceed that of the incoming waveform. In addition, UWB signals are being attenuated
differently in various frequency bands when propagating through the atmosphere.
Therefore, conventional optimal processing using matched filters or correlators may not
be suitable for these signals, and they may require more sophisticated receiver
architecture (see chapter. 2 in [4]), to maximize the signal to noise ratio (SNR).
Moreover, in order to have a good Doppler or range-rate resolution, it is
necessary to extend the total duration of the signal [3]. The common approach of
extending the signal duration is by repeating it periodically. While effective radar
surveillance in a hostile setting requires covert and interference-free operation, using
periodic signals which can be easily recognized by an intelligent adversary is potentially
unsafe. A possible alternative is the transmission of modulated, broadband noise
waveforms.
1.1.2 UWB noise radar
UWB noise radar technology relies on the transmission of a random broadband
noise waveform. A simplified system block diagram is shown in Fig. 1.1. Range
information is being obtained by cross-correlating the reflected echoes with a time-
delayed replica of the transmitted signal. If the round-trip time delay to the target
4
matches the internal time delay, a peak is observed at the correlator output, whose
magnitude is proportional to the target reflectivity. The use of noise instead of
deterministic waveforms in UWB radar systems provides better immunity to interception
and jamming since it is featureless, at least in principle [5].
UWB noise radars are spectrally efficient, in the sense that many systems can
occupy the same spectral band with negligible cross-interference, as the signal from one
will not correlate with the others. Simple filters can be used to adaptively shape the
transmitted noise spectrum in order to reduce clutter and enhance detection of specific
target types. Since resolution is inversely proportional to the signal bandwidth [3],
physical processes that generate broadband and 'pure' noise are attractive platforms for
the generation of UWB radar signals. By transmitting a UWB signal with a bandwidth
of, say 1 GHz, the range resolution can be made as high as 15 cm, thereby providing
excellent ability to detect and recognize different types of small targets.
Recent years have witnessed an increasing interest in defense-oriented research
and development in the area of UWB noise radars driven by the above potential.
Examples include concealed weapon detection systems [6, 7]. Concealed weapons pose a
significant threat to military, security, and law enforcement personnel. Current
technologies to detect concealed weapons, such as X-Rays and passive radiometry, are
expensive, cumbersome, slow, and prone to false alarms [7]. These systems are primarily
oriented to detecting metal, and they require the cooperation of the person being
searched. Since a concealed weapon comes in different sizes and shapes, ultra-wideband
(UWB) radar techniques can be used to excite natural electromagnetic resonances that
characterize the size, shape, and material composition of an object. Using noise
waveforms in such systems can provide the necessary immunity to electromagnetic
interference. Furthermore, the noise spectrum can be shaped to enhance detection of
5
specific target types, prevent signal leakage into adjacent bands, or prevent in-band
spectral interference with friendly systems.
Another use of UWB noise radars is in through-wall imaging applications [5, 8].
Current building-interior imaging systems are based on short-pulse waveforms, which
require specially designed antennas to subdue unwanted ringing. UWB noise waveforms
exhibit lesser ambiguity between range variations and frequency offsets, a property that
is very useful in the imaging of moving targets. Thus, UWB waveforms are ideal
candidate sensors for covert imaging of obscured regions in hostile environments.
Simulation and preliminary experimental studies have shown detection of human activity
behind the wall and human target tracking [5].
UWB noise radar systems for ground penetration probing applications were also
developed and demonstrated. Ground-penetrating or sub-surface radar systems are
increasingly being used for a variety of military and civilian applications [8, 9]. Some of
the primary issues that merit special attention are the efficient coupling of the
electromagnetic energy into the ground, elimination of the large reflection from the air-
ground interface, achieving adequate signal penetration into lossy media, and adequate
signal bandwidth consistent with desired depth resolution. Ground-penetrating radar
systems operate over a wide range of probing depths, from close range, high-resolution
applications such as locating buried mines and hidden voids in pavements at depths of up
Noise SourceRF
Splitter
Oscilloscope
Antenna
Antenna
Received
signal
Reference
replica
Computer
Cross-correlation of time-
delayed reference replica
and received signal
Electrical noise
Waveforms
#1 and #2
Transmitted
signalAMP
AMP
Figure 1. 1. Simplified block diagram of correlation-based noise radar.
Fig. 1.1. Simplified block diagram of correlation noise Radar
6
to 50 cm, to long-range, low-resolution applications, such as probing geologic strata at
depths of over 100 m.
In one example [9], an UWB noise radar operating at the 1-2 GHz band was used
for detecting shallowly buried mine-like objects. High spatial resolution of 15 cm was
achieved due to the wide bandwidth of the transmitted signal. Two distinct receiving
antennas with complementary directions of polarization were used, in order to gather
data from a variety of buried targets at different depths and with different relative
orientations. Buried targets included metallic as well as non-metallic objects of different
sizes and shapes that mimicked land mines as well as other objects.
1.2 Microwave photonics overview and applications
Microwave photonics has been an active area of research for over thirty years
[10]. In microwave photonics, an analog radio-frequency (RF) waveform is super-
imposed on an optical carrier using a modulator, subsequently processed in a photonic
device, and finally re-converted to electrical frequencies by a photo-detector. The key
advantages of microwave-photonic setups over conventional electrical-transmission
systems, such as coaxial cables or waveguides, include reduced size and weight, low and
constant attenuation over the entire microwave and millimeter-wave modulation
frequency range, immunity to electromagnetic interference, low dispersion and high
data-transfer capacity.
1.2.1 Electro-optic modulators and photo-detectors
The fundamental elements of a microwave photonic link are devices that offer
signal modulation and detection at a broad bandwidth of microwave frequencies. The
optical carrier is modulated by electrical signals using one of two common techniques:
direct current modulation of a laser diode, or use of an external modulator.
7
Direct modulation of the laser is the simplest approach: an analog waveform is
super-imposed on the DC bias that is driving the diode. In this approach the modulation
bandwidth is restricted to below 10 GHz by relaxation oscillations of the laser and by
parasitic inductance and capacitance associated with packaging [11]. Since the invention
of semiconductor lasers diodes, much effort has been dedicated to increasing their direct
modulation bandwidth. These attempts included the incorporation of quantum wells
layers, providing a direct modulation bandwidth of over 30 GHz [12]. Other approaches
used external cavities to enhance the resonance frequency response of semiconductor
lasers [13, 14, 15].
In the second technique, the laser diode is operated at continuous-wave (CW)
mode, providing a stable optical power output. The laser’s optical output passes through
a separate device that modulates the optical carrier's intensity, namely an external
modulator. In this way, the unwanted effects of the direct modulation of the laser are
avoided, and higher modulation rates can be reached. The most popular external
modulator used is the Ti:LiNbO3 Mach-Zehnder interferometer. This modulator relies on
the electro-optic (Pockels) effect, which provides a change in the optical waveguide
refractive index that is proportional to an externally applied electric field of RF-to-
millimeter wave frequency. Incoming light is split in two paths, which recombine at the
output. Electro-optic index modulation is used to modify the optical phases that are
accumulated over the two paths. Interference at the output translates the phase
modulation into intensity variations.
High bandwidth external modulators have been greatly improved over the past
two decades. For their practical application in microwave photonics systems, it is
imperative that these devices feature a broad bandwidth, low drive voltage, high
linearity, good bias stability, high optical power-handling ability, and low optical
8
insertion loss. In addition to LiNbO3, modulators based on semiconductor and polymer
materials are being investigated as well [16, 17, 18]. Travelling-wave metallic
waveguides are used to match the propagation velocity of the driving millimeter-wave
signal to that of the optical mode [19]. There have been a number of demonstrations of
LiNbO3-based modulators with bandwidths in excess of 40 GHz and relatively low drive
voltages [20, 21].
The detection of optical signals with relatively high bandwidths in microwave
photonic systems is also required. High-speed photo-detectors operating in the
wavelength range of 1.3–1.55 μm with high responsivities of 0.35A/W, bandwidths of
up to 110GHz and high optical power-handling have been reported and are currently
available [22, 23, 24].
1.2.2 Radio-over-fiber and antenna remoting
One of the main applications of microwave photonics technologies is the
transport and distribution of radio or wireless signals over optical fiber [25]. In such
radio-over-fiber (RoF) applications, the radio signals are transported over the fiber
through either direct or external modulation of an optical carrier, to a remote base where
they are detected and retransmitted by an antenna. RoF technologies are being used in
many wireless systems including cellular networks, indoor distributed antenna systems
and wireless local area networks (WLANs), as well as fixed and mobile broadband
networks that can provide very high bandwidth services to users. RoF transmission
allows for antenna remoting: the separation of the antenna and front-end elements of a
radar system from the central office or unit, where the waveforms are generated,
processed and interpreted. Roman et al., for example, demonstrated fiber-optic antenna
remoting of an X-band radar [26].
9
There are several possible approaches to transporting radio signals over optical
fiber, each facing certain trade-off considerations. One option is to modulate the optical
carrier with the incoming radio waveform, which is often centered at a high RF, without
any preceding mixing stages (Fig. 1.2, left). It is sometimes referred to as RF-over-fiber.
This method simplifies the remote base station, however it is prone to chromatic
dispersion-induced fading [27]. RF-over-fiber systems operating at cellular and WLAN
systems (<5 GHz) can be readily implemented through the direct modulation of high-
linearity single-mode distributed feed-back (DFB) lasers.
In intermediate frequency (IF) over fiber techniques, (Fig.1.2, center), the radio
signals are first down-converted to a lower, nonzero IF, before being superimposed on
the optical carrier. IF signal transport schemes can employ readily available, mature
microwave hardware at the base station. In addition, IF radio signals transport could
allow transmission over low cost multimode fiber, as they are less susceptible to
dispersion. However, IF-over-fiber requires frequency conversion via local oscillator
(LO) at the base station, which in turn increases the complexity of its architecture,
particularly as the frequencies of wireless applications move into the millimeter wave
region.
A third technique that can be used to transport data-carrying radio signals
between the central office and the remote antenna base station is the modulation of the
Figure 1. 2. Simplified block diagrams for 3 main RoF techniques. Left: RF-over-fiber transport scheme.
Center: IF-over-fiber transport scheme. Right: baseband-over-fiber transport scheme. IFf : Intermediate
frequency. LOf :Local oscillator frequency. CO: Central office. BS: Base station. E/O: Electrical to Optical.
O/E: Optical to electrical. TX : Transmitter
11
optical carrier by base-band waveforms (Fig. 1.2, Right). At the base station detected
signals are up converted to IFs, before undergoing an additional frequency up-conversion
to the required RF band via an LO. As with IF-over-fiber, RoF systems based on
baseband transport schemes can readily exploit the use of mature and reliable RF and
digital hardware for signal processing at both central office the base station, as well as
low cost optoelectronic interfaces. Yet again, the need for frequency conversion at the
base station increases the complexity of the base station architecture as the transmitting
radio frequency band increases.
1.2.3 Microwave photonic variable delay lines
Another promising application of microwave-photonics is the optical
implementation of RF delay lines. Such delay lines are critical components for beam
steering within phased-array radar systems [28]. The variable true-time-delay (TTD) of
high-frequency waveforms using RF cables tends to be lossy and bulky, thus microwave-
photonic techniques provide a potentially attractive alternative. Numerous methods have
been proposed for realizing microwave-photonic TTD. In some examples, a TTD is
achieved using switching among fiber paths of different lengths [29, 30]. Other methods
use dispersive elements in conjunction with variable wavelength sources [31, 32], and
the combination of both switching and dispersion [33]. Recently, a new approach that is
specifically adjusted to the TTD of a particular category of radar signals, linear
frequency modulation (LFM) waveforms, was proposed and demonstrated [34]. This
approach relies on the ambiguity between temporal delays and frequency offsets in LFM
signals. In sacrificing the universality of the TTD approach, a delay of a 500-MHz wide
LFM signal by as much as 250 ns was successfully demonstrated, which is an order of
magnitude longer than previously reported TTD results.
11
1.2.4 Microwave-photonic generation of UWB waveforms
All-electrical generation and processing of UWB signals of high central
frequencies face several challenges, such as the distribution of signals to an appreciable
distance, the flexible generation and reconfiguration of signals spectra, and the
implementation of a continuously-variable TTD. Microwave-photonic generation
techniques can offer flexible tuning of high-frequency pulse shapes, inherent immunity
to electromagnetic interference, and parallel processing via wavelength division
multiplexing [35]. Driven by the promises of integration and flexibility, much research
effort has been dedicated to photonic generation of UWB waveforms in recent years.
As previously mentioned, many UWB Radar implementations rely on impulse
radio transmission. Accordingly, a variety of techniques for impulse UWB generation
using photonic means have been reported. One category of photonic UWB generation
techniques relied on the conversion of phase to intensity modulation [36, 37, 38]. This
method is simple to implement, however it offers few degrees of freedom for pulse
shaping and minimal reconfiguration. Waveforms generated using this method are
restricted to the simplest form of impulse-radio pulses: a Gaussian mono-cycle or a
Gaussian doublet shape. Bolea et al. [39, 40] proposed a method of generating higher
order pulse shapes based on microwave-photonic tapped delay line filters, with both
positive and negative coefficients. However, each additional coefficient required the use
of another laser source.
Another approach for UWB impulse generation is based on nonlinear dynamics
in optical semiconductor amplifiers (SOA) and laser diodes. Cross-gain modulation
(XGM) effects in SOAs and cross-absorption effects in electro-absorption modulators
had been used in Gaussian mono-cycle and doublet waveform generation [41, 42, 43,
44]. Relaxation oscillations in directly-modulated or externally-injected DFB lasers were
12
also demonstrated [45, 46, 47]. The technique is well suited to the FCC spectral mask
[2], which was set as a standard for UWB-based indoor wireless communication.
However, waveform generation based on relaxation oscillations is restricted to the order
of 10 GHz by the laser diode dynamics.
Another approach of generating UWB pulses used optical spectrum shaping of
the transmitted waveforms in order to maximize the transmitted power within the
constraints of the FCC mask [48, 49, 50]. In general, spectrum shaping techniques
require mode locked laser sources, free-space optics arrangement [49, 50] or complex
fiber gratings [48]. Major progress had been recently achieved, with the pulse-shaping
optical components successfully replaced by a programmable, integrated silicon-
photonic waveguide circuit [51].
UWB pulses can also be generated using nonlinear effects in optical fibers.
Cross-gain modulation in optical parametric amplifiers and cross-phase-modulation
(XPM) effects had both been used to generate mono-cycle and doublet pulse shapes [52,
53]. Zadok et al. [54, 55, 56] proposed a method of generating higher-order impulse
signals. Their technique relies on the self-phase-modulation (SPM) effect in standard
optical fiber, and on all-optical edge detection. The edge detection takes advantage of the
time-varying chirp introduced by SPM, and uses judiciously tuned optical band-pass
filters to obtain two temporally-narrowed replicas of the input pulse train. The shapes of
the narrowed replicas are then subtracted from that of the original pulse train in a
broadband, balanced differential detector. The resulting waveforms are highly
reconfigurable through adjustments of the input power and tuning of the optical filters.
High-order UWB waveforms, having a center frequency of 34 GHz and a fractional
bandwidth of 70%, were successfully generated.
13
As previously discussed, there are several disadvantages to using UWB impulse-
radio transmission for communication and radar purposes. In communication channels,
UWB architectures that are based on impulse radio require elaborate pulse shaping in
order to comply with the spectral masks imposed by the FCC standard. In radar systems,
the use of noise instead of deterministic waveforms provides better immunity to
interception and jamming.
1.2.5 Photonic generation of UWB noise waveform
UWB noise can be readily generated using electrical techniques, however optical
methods are nonetheless appealing as part of a RoF integrated system. Although the
separate potential advantages of UWB noise radars and of microwave-photonic UWB
generation are largely recognized, the generation of UWB noise using optical means
remains largely unaddressed. In one such example, Zheng et al [57] demonstrated
photonic generation of UWB noise, based on the chaotic dynamics of a laser diode in a
feedback loop. The spectrum of the noise source could be controlled by adjusting the
operating conditions and the parameters of the chaotic oscillation source, such as bias
current and feedback strength. Peled et al. [58] proposed and demonstrated the photonic
generation of UWB noise, based on the amplified spontaneous emission associated with
stimulated Brillouin scattering (SBS) in optical fibers. The noise bandwidth was
broadened via tailored direct modulation of the SBS pump source [59]. Compared with
the optical generation of impulse radio, SBS-ASE noise generation is considerably
simpler, and allows for agile reconfiguration of the noise central frequency, bandwidth
and spectrum.
In this work, a microwave-photonic, ultra-wideband (UWB) noise radar system is
proposed and demonstrated. The system takes advantage of the amplified spontaneous
14
emission (ASE) that accompanies optical amplification for the photonic generation of
UWB noise. Two gain mechanisms will be examined: SBS and erbium-doped fiber
amplifiers (EDFAs). Both will be discussed in detail in the following chapter.
1.3 Laser Radar (LADAR)
Laser radar (LADAR), or light detection and ranging (LIDAR) as it is sometimes
known, is a radar system that transmits energy at optical frequencies, four to six orders
of magnitude higher than those of RF radars. LADAR systems are limited to
wavelengths where both suitable lasers and detectors are available. There are some key
distinctions between the LADARs and radars. Producing the desired waveform in a radar
transmitter could be as simple as turning an oscillator (or amplifier) on and off. In
LADARs, on the other hand, either direct or external modulation devices must be used to
generate sophisticated waveforms. Once a signal is generated, it must be launched
towards the target. In radars, this is done through an antenna. The optical equivalent of
the antenna in a LADAR system is a telescope or an arrangement of optical lenses.
Because the optical alignment of these components is critical, care must be taken to
provide very stable bases and mounts for the optical elements.
For both radar and LADAR systems, the receiver function is to transform the
propagating energy captured by the antenna into an electrical signal that can be
processed to extract the desired information. In a radar system, the returning signal
induces current that is directly proportional to the fluctuating electromagnetic field,
whereas in a LADAR system, a photodiode is used to convert the photons to current that
is proportional to the intensity of the incident radiation. Therefore, if phase information
is required in a LADAR, then mixing with a phase-locked local oscillator (LO) is
required upon detection (i.e. a coherent receiver).
15
1.3.1 Incoherent and coherent LADAR systems
In general, there are two types of LADAR systems: incoherent, which rely on
direct detection of intensity only, and coherent, which make use of both the amplitude
and the phase information of the optical wave. The differences between the two are
illustrated in Fig. 1.3. In coherent systems, a fraction of the outgoing laser energy is split
off and redirected to the receiver detector. This energy is then aligned with the collected
LADAR echoes on a photo-detector, which is operating as a classical mixer. Generally
speaking, coherent LADARs can operate at lower SNRs than their incoherent
counterparts [11]. In addition, coherent systems allow for the use of sequence
compression techniques [3], which rely for the most part on frequency and phase coding.
Sequence compression will be addressed in much detail later in this thesis.
While prevalent in RF and microwave systems, coherent receivers in the optical
domain come at a cost of significant complexity [11]. An intermediate and highly
appealing approach, which will be explored in this research, is the direct detection of
pulse position modulated sequences. A proper post-detection processing of such
Figure 1. 3. Block diagrams of incoherent (top) and coherent (bottom) LADAR systems [60].
16
sequences could allow for highly effective compression, with SNR levels that could rival
those of coherent systems.
1.3.2 LADAR applications
LADAR systems can be catalogued into many categories. They may be grouped
according to their transmitted waveform (i.e., CW, continuously-modulated, or pulsed);
by receiver concept (coherent or direct detection); or by the intended measurement
(range, velocity, backscatter, or spectral absorption). Development of LADAR systems is
advancing rapidly [60]. Their applications include range-finders [61], 2D and 3D
imaging systems [62], Doppler vibrometers [63], and synthetic aperture imaging [63,
64]. Three-dimensional images are valuable in applications such as mapping, target
recognition and machine gesture control. A 3D image of a scene is constructed using
combined multiple range measurements taken along different lines of sight. Traditional
3D LADAR systems use scanning mirrors in order to obtain high resolution images. In
order to measure Doppler shifts, coherent receivers are needed. These coherent LADARs
are sensitive enough to measure surface vibrations on remote objects [63]. Despite this
impressive growth in such relatively recent LADAR applications, the primary objective
of LADAR has always been the measurement of the range to a target.
1.3.3 Common ranging measurements techniques
The most commonly-used LADAR scheme relies on the transmission of short
and intense isolated pulses, and time of flight (ToF) measurements of collected
reflections [60]. Typical ToF LADAR systems transmit short laser pulses using a process
called Q-switching. The Q-switching process allows for relatively large amounts of
energy to build within the laser cavity before being released over a very short period of
time. The energy is released from the laser cavity using a device such as a rotating mirror
17
or a Pockels cell crystal [65]. Laser pulses created by Q-switched systems can have
durations on the order of nanoseconds. The receiver calculates the time it took a single
pulse to make a round trip to the target and back. This time equals to the roundtrip
distance divided by the speed of light.
A block diagram of a ToF LADAR range finder is shown in Fig.1.4 (Left). This
system consists of a laser transmitter emitting pulses with a duration of few nanoseconds,
a receiver channel including a PIN or an avalanche photodiode (APD), amplifiers, an
automatic gain control (AGC) and timing discriminators. The emitted light pulse (start
pulse) triggers the time interval measurement unit, and the reflected light pulse (stop
pulse) stops it. The distance to the target is proportional to this time interval. In this
concept range accuracy and precision is limited by the length of the transmitted laser
pulse, the pulse's shape, receiver electronics and noise sources in the LADAR system. In
order to get a significant working distance, intense and short pulses are needed.
Another ranging measurement technique is based on phase-shifts. In a phase-shift
range-finder, the optical power is modulated with a constant radio frequency. The basic
operating scheme of the device is shown in Fig.1.4 (Right). A sine wave of frequency
RFf is generated by the main oscillator and modulates the DC current of the laser diode.
After reflection from the target, a photodiode collects a part of the laser beam.
Measurement of the distance D is deduced from the RF-domain phase difference
between the photo-detected current and the original out-going signal. In this technique
the accuracy and precision are limited by drifts in the intermediate frequency IFf (see
figure), cross talk between transmitter and receiver channels, and signal distortions.
18
Yet another common technique for ranging measurements is based on linear
frequency modulation (LFM) of the laser signal [66, 67] and a coherent receiver. The
basic concept of an LFM LADAR is illustrated in Fig. 1.5. The driving current to a laser
diode source is being modulated by a ramp waveform, resulting in a periodic linear
frequency chirp. The laser output is launched simultaneously towards the object and
towards a reference mirror using a beam splitter. The reflected signals are then
superimposed in a square-law detector. The beating term, which is oscillating at some
intermediate frequency IFf , is further amplified and measured with a frequency counter.
The intermediate frequency IFf is proportional to the time delay between the transmitted
and received waveforms. Thus, with the a-priori knowledge of the sweep bandwidth and
repetition rate of the LFM waveform, the distance to the target is obtained.
Due to the square law mixing process, the amplitude of the detector output at IFf
is proportional to the amplitudes (as opposed to the power levels) of the collected echo
signal and the reference, respectively. Accordingly, the dynamic range of the frequency-
swept technique is twice as large (in dB scale) as that of pulsed radars, in which the
electrical signal is proportional to the power collected from the object. The improvement
Figure 1. 4. Left: Block diagram of ToF LADAR range finder [61]. Right: Block diagram of phase-shift
LADAR rangefinder [61]. D: Distance. PIN: PIN photo-diode. Osc.RF: RF oscillator. Osc.OL: Local
oscillator. IFf : Intermediate frequency. IFf : Pass-band filter bandwidth. : phase difference.
19
in dynamic range in turn extends the total working distance, and is particularly attractive
for short range sensing.
The limiting factor of swept-frequency LADAR systems is the nonlinear
frequency response of laser diodes. The frequency modulation response of a laser diode
is, in general, non-uniform, so that a linear optical frequency sweep cannot be perfectly
realized by ramp modulation of the control current. As a consequence, deviations from
the linear sweep usually occur that, in turn, lead to variations in the intermediate
frequency IFf . Another fundamental limitation of the measurement accuracy is due to the
phase noise of the laser diode [68, 69]. Frequency-modulated laser diodes with narrow
spectral linewidths are in general preferable for highly accurate range measurements.
1.3.4 Sources of noise in LADAR systems[60]
Several phenomena contribute noise to LADAR systems. The noise sources
include statistical fluctuations in the amount of light arriving at the LADAR detector (for
example due to relative intensity noise, RIN, of the laser diode source); variations in the
photo-detected current that are due to the quantum characteristics of the detection
Figure 1. 5. Block diagram of LFM based LADAR range finder [61].
21
process (Shot noise); thermal noise due to fluctuations in current along electrical
conductors; speckle patterns, and additive stray photons.
Shot noise is inherent to the detection process. Due to the particle nature of light,
the number of photo-electrons counted during a time interval t is a random variable,
even if the incident optical intensity is entirely deterministic. The mean number of photo-
electrons is proportional to the expected number of photons as decreed by the incoming
intensity and measurement duration. The number of photo-electrons measured during the
detector integration time is characterized by Poisson statistics [11]. The standard
deviation, according to these statistics, is proportional to the square root of the mean
value. Therefore, in those systems in which shot noise is the performance-limiting
mechanism, the SNR can be improved by elevating the source power.
Thermal noise relates to the fact that at a finite temperature, electrons move
randomly in any conductor. Random thermal motion of electrons in a resistor manifests
as a fluctuating current, even in the absence of an applied voltage. The load resistor in
the front end of an optical receiver adds such fluctuations to the current generated by the
photodiode. Thermal noise is independent of the incoming power. The effect of thermal
noise is often quantified through a quantity called the noise-equivalent power (NEP):
defined as the minimum optical power per unit bandwidth required for an SNR of unity.
As in the shot noise-limited case, the SNR of a thermal noise-limited system improves
with incoming power.
The output of a semiconductor laser exhibits fluctuations in its intensity, phase,
and frequency, even when the laser is biased at a constant current with negligible
fluctuations, due to unavoidable spontaneous emission that accompanies the stimulated
emission process [11]. Phase and frequency noise could degrade the performance of
coherent LADAR systems, however they do not manifest directly in incoherent detection
21
schemes. RIN, on the other hand, has a detrimental effect on direct detection as well.
RIN is quantified in terms of the power spectral density of the intensity fluctuations,
normalized to the average power. Typical InGaAsP laser diodes operating at 1.55 µm
exhibit RIN of about -155dB/Hz [11]. By definition, SNR limitations due to RIN cannot
be improved by raising the source power.
Speckle effects are caused by interference from a large collection of independent
coherent radiators which arrive together to the photo-detector. Such interference occurs
when the laser source is reflected from a rough surface. The number of photo-electrons
subject to speckle phenomena can be modeled as a negative binomial random variable
[70]. Speckle becomes dominant for highly coherent sources [60]. Much like the case of
RIN, noise due to speckle scales with the source intensity.
Background or ambient noise is simply the radiation from sunlight or ambient
light source that falls on the receiver’s field of view. The random arrival times of the
photons from the background contribute noise to the LADAR system measurement.
Background noise can be modeled as Poisson process, and its standard deviation is
proportional to the bandwidth of the ambient light source [60]. Practical LADAR
systems use optical band pass filters, designed to block all background photons at
wavelengths other than that of the intended source.
1.3.5 Pulse Compression
Pulse compression is a technique in which the energy in a modulated, long, low-
power waveform could be integrated through matched filtering into a virtual short pulse
of high peak power [3]. The majority of pulse compression waveforms are constructed
using frequency sweeping, either linear or nonlinear, or phase-coding (bipolar and poly-
phase sequences). Probably the most popular pulse compression method is based on
22
LFM [71], as discussed earlier in the context of LADAR ranging measurements. In
LFM, rather than send a pulse with a short duration 1/ B , a bandwidth B is being
linearly swept across a pulse duration 1/T B . Matched-filtering compresses the LFM
waveform to a short duration of 1/ B [72]. The performance metrics of LFM pulses are
largely affected by their time-bandwidth product TB . Larger time-bandwidth products
result in improved resolution and more effective suppression of secondary peaks, or
sidelobes, at the output of the matched filter. The sidelobes can be further reduced using
digital spectral windowing techniques or nonlinear sweeping of the instantaneous
frequency [73, 74, 75].
1.3.6 Phase codes
Another rich family of pulse compression waveforms is based on phase codes. In
these codes, a pulse of duration T is divided into M bits of identical duration
/bt T M , and each bit is assigned with a different phase value [3]. Much like LFM
waveforms, the phase coded sequence can be cross-correlated with a replica of itself to
produce a narrow virtual peak with low sidelobes.
The quality of the compressed waveform is quantified in terms of the ratio
between the main peak power and that of the highest undesired sidelobe (peak-to-
sidelobe ratio, PSLR), and the ratio between the energy within the main peak to the
energy outside it (integrated sidelobe ratio, ISLR). The quest for efficient codes having
high PSLR and ISLR represents a major challenge to the radar community since World-
War II. Classic examples are the binary (±1) phase codes known as Barker sequences
[76]. The matched-filtered form of a Barker code of length M is characterized by a
correlation peak of magnitude M and a maximum correlation sidelobe of 1. The PSLR
23
of Barker codes is therefore M . Unfortunately, the longest Barker code reported to date
is of length 13M .
Another known class of sequences is the poly-phase Barker codes. In poly-phase
codes, the possible phase values are not restricted to 0 or , resulting in lower sidelobes
[3]. However, the search for such codes with a large number of bits requires extensive
numerical optimization techniques. Examples of such codes were found for all 45M
[3]. Although it is not possible to design phase-coded pulses with zero aperiodic
correlation sidelobes [3], the periodic correlation of a phase-coded signal can be zero for
all nonzero shifts. Phase codes having zero periodic autocorrelation sidelobes are called
perfect codes. Perfect codes are useful, for example, in radar systems that are
continuously modulated, rather than pulsed.
One example of a perfect code is the Frank code [77]. In addition to its perfect
periodic autocorrelation, the Frank code has relatively low aperiodic autocorrelation
sidelobes as well. Over the years modified versions of the Frank code were introduced
[78]. Although not perfect, they yield the same aperiodic PSLR as the original Frank
code with even lower ISLR. The main drawback of the Frank code and its modified
versions is that they only apply for codes of perfect square length 2M L . In contrast,
chirp-like perfect codes, which are applicable for any code length M , were also
introduced. Examples include the Zadoff-Chu codes and their modified versions [79, 80,
81]. These codes are characterized with higher sidelobes than those of the Frank code.
Other phase codes can be derived by sampling the instantaneous phase curve of a
nonlinear frequency-modulated (or frequency-stepped) pulse. Using nonlinear instead of
linear frequency modulation can yield lower autocorrelation sidelobes at the expense of a
broader main peak (degraded resolution). One such example is the ( , )P n k code [82].
24
The ( , )P n k code is based on step approximation of the phase function of a nonlinear
FM chirp signal. The main drawback of all chirp-like phase codes is their poly-phase
structure, leading to relatively complex transmitters and receivers designed to
approximate precise analog values. In order to ease this constraint, Ipatov suggested
perfect binary phase codes with minimum power loss at the main correlation peak [83].
Non-perfect binary codes that yield an optimized PSLR, but do not meet the
Barker condition, are often called maximum peak sidelobe level (MPSL) sequences.
Finding such codes requires exhaustive computer searches [3]. A rich library of MPSL
codes of different lengths M has been developed [3]. The sidelobes can be further
suppressed using a mismatched filtering process, in which the original sequence is cross-
correlated with a longer code, whose coefficients are not restricted to ±1. Substantial
sidelobe suppression can be obtained, at the cost of a modest degradation in the central
correlation peak power (see sec. 6.6 in [3]).
1.3.7 Complementary pair
A complementary pair of sequences satisfies the property that their out-of-phase
a-periodic autocorrelation coefficients sum to zero [84].
TABLE I
PRIMITIVE COMPLEMENTARY PAIRS N sequence a sequence b
2 ++ +-
10 ++-+-+--++ ++-+++++--
10 +++++-+--+ ++--+++-+-
20 ++++-+---++--++-+--+ ++++-+++++---+-+-++-
26 ++++-++--+-+-+--+-+++--+++ ++++-++--+-+++++-+---++---
25
In contrast to MPSL sequences, the finding of long complementary pairs is
relatively simple. The generation of a new complementary pair starts with one or two of
the primitive pairs listed in Table I, followed by the application of one of several
construction rules. The procedure can be repeated as needed. A recent detailed
description of the presently known construction rules, including proofs, appears in
section 7.3 of [85]. The most basic construction rule [84], which creates the pair {c,d}
based on a pair {a,b}, is:
, cat , ,cat ,c d a b a -b , (1.1)
where cat(a,b) stands for concatenation of the two sequences a and b. The expression –b
implies polarity reversal of the elements of sequence b. This basic construction rule
increases the length of the sequences in the new pair. The same rule can be used to create
a different pair from the same original pair:
, cat , ,cat ,-e f b a b a . (1.2)
Thus, the number of possible pairs having a longer length is larger. Table II lists the
number of such complementary pairs for all lengths up to 100 [86].
TABLE II
NUMBER OF PAIRS FOR LENGTH N < 100
PAIRS
PAIRS
26
Note that complementary pairs are not found in all lengths. It can be shown that
the possible lengths must comply with:
2 10 26 , , , 0N , (1.3)
where , and are integers. The large choice of codes, shown in Table II and equation
1.3, helps reduce the probability of intercept (LPI) of complementary pair-encoded
transmission.
In conventional coherent radars the two sequences are usually modulated on
consecutive pulses, which are then coherently processed jointly, using matched filters.
The pulse repetition interval (PRI) needs to be large enough to avoid range ambiguity.
The main drawback for radar use is the sensitivity to Doppler shift. If during the PRI the
range to the target has changed by a meaningful fraction of a wavelength, the second
pulse will exhibit an additional phase shift that will degrade or destroy the
complementary property of the pair. Direct-detection laser applications of
complementary pair coding are insensitive to phase. Thus the use of complementary
pairs in LADAR applications is immune to their main drawback.
1.3.8 LADAR system based on incoherent pulse Compression
The straight-forward application of phase codes to a LADAR system would
require a coherent receiver, which comes at the cost of significant complexity [11].
Recently, a new approach for the compression of incoherently detected pulse sequences
was introduced by Prof. Nadav Levanon of Tel-Aviv University [87]. The phase changes
of a chosen bipolar phase code are translated to a pulse position modulation through a
process called Manchester coding, which will be introduced in detail in Chapter 3. The
encoding process results in a binary, intensity modulated sequence, hence reflected
echoes can be received by simple incoherent detection. Nevertheless, the post-detection
27
processing algorithm proposed in [87] yields PSLR and ISLR performance which nearly
replicates that of the original bipolar phase code. The method provides considerable
relaxation in the LADAR receiver architecture, compared with the coherent setup that
would normally be necessary for the processing of the bipolar phase sequence, while
retaining most of the performance benefits of using that sequence.
Natural candidates for the incoherent pulse compression procedure are binary
phase codes. In this thesis, MPSL codes of 82 and 1112 bits length as well as
complementary pair codes of 416 and 832 bits length are being used. With incoherent
compression of these codes, laser ranging was demonstrated at poor SNR conditions, as
low as -20 dB [88]. The noise tolerance can be leveraged towards a longer measurement
range, lower launch power and energy consumption, reduced apertures and improved
operation at unfavorable atmospheric conditions.
28
CHAPTER
2
Microwave photonic UWB radar system
2.1 Photonic noise generation
The noise waveforms used in the microwave-photonic radar system experiments
were generated based on the amplified spontaneous emission (ASE) that is associated
with either stimulated Brillouin scattering (SBS) processes in standard fibers, or with
gain in erbium-doped fiber amplifiers (EDFA). Both mechanisms are introduced below.
2.1.1 UWB noise generation via ASE of SBS in optical fiber
In SBS, a strong pump wave and a typically weaker, counter propagating probe
wave optically interfere to generate, through electrostriction, a traveling longitudinal
acoustic wave. The acoustic wave, in turn, couples these optical waves to each other
[89]. In the absence of a seed input signal wave, SBS could still be initiated by
thermally-excited acoustic vibrations [89]. The naturally occurring vibrations scatter a
fraction of the incident pump into a preliminary probe, which is then further amplified.
In this scenario SBS acts as a generator of ASE. The frequency of SBS-ASE is lower
than that of the incident pump by the Brillouin shift of the fiber ~ 2 11 [GHz]B .
For a CW pump laser of optical frequency p and power pI , the SBS complex
gain coefficient
( )sg , in units of m-1
, is of Lorentzian shape [90].
0( ) .
1 2( ) /
p
s
p s B B
g Ig
j
(2.1)
29
Here B ~ 2 30 [MHz] denotes the narrow inherent linewidth of the SBS
process. When the pump wave is modulated to obtain a broadened power spectral density
(PSD) ( )p pI , ( )sg is given by a convolution of the pump PSD with the SBS line
shape [91]:
0 ( )( ) .
1 2( ) /
p p
s p
p s B B
g Ig d
j
(2.2)
If the spectral width of the pump PSD is much wider than B , the real part of
( )sg may be approximated by [92]:
Re[ ( )] ( ).s p s Bg I
(2.3)
The imaginary part of ( )sg can be calculated from its real part through the
Kramers–Kronig relations [91]:
2 2
Re[( ( )]2Im[ ( )] .s s
s s
s s
gg d
(2.4)
Using Eq. (2.3) and (2.4), the complex gain coefficient of the probe signal
amplification can be determined for a given arbitrary pump PSD ( )p sI . The PSD of the
obtained SBS-ASE in the absence of seed input is given by [93]:
( ) (exp[ ( ) ] 1 [1 exp( )]{exp[ ( ) ] 1}/[ ( ) ])s s s s eff s eff s effP g L L L g L g L (2.5)
Here L is the fiber length, denotes its loss coefficient in m-1
, and its effective
length is defined as [1 exp( )] /effL L . SBS-ASE is inherently narrowband: its
bandwidth for a CW pump wave is on the order of the Brillouin linewidth B . UWB
31
generation requires a substantial spectral broadening of the SBS-ASE process. Carefully
synthesized pump modulation could provide a uniform PSD within a broadened
bandwidth of interest, and consequently generate a broad and uniform PSD of SBS-ASE
[89]. Although the pump wave is often modulated in a deterministic manner, the
resulting SBS-ASE is entirely stochastic.
2.1.2 Broadening the spectrum of the pump-laser diode
The bandwidth of SBS amplification may be broadened through direct
modulation of the injection current of a distributed feedback (DFB) laser diode which is
used as an SBS pump source. The physical mechanisms underlying the DFB lasing
frequency variations in response to direct modulation were modeled in detail [94, 95]. In
general, there are three separate mechanisms that affect the instantaneous frequency:
transient, adiabatic and thermal chirp [96, 97]. Transient chirp is associated with the
mutual feedback of photon and carrier densities, following a step transition of the current
[98]. The typical time constants are on the order of 100 ps, so that the transient chirp may
be neglected for direct modulation at MHz rates used in this work. The adiabatic chirp
describes the frequency variations that are introduced by the current-related changes of
the equilibrium carrier density. The adiabatic-chirp contribution immediately follows the
instantaneous injection current ( )i t :
(2.6 ) ( ) ( ),a at C i t
where aC is typically on the order of 0.1-1 GHz/mA. The frequency variations
associated with thermal chirp are proportional to the instantaneous temperature of the
laser active region, which modifies both its refractive index and physical length [97]. The
thermal chirp is slower to evolve, and its dynamics are determined by the thermal
31
conductivity and capacitance of the various structural layers. In previous studies [94, 95]
a semi-empirical model of the thermal chirp was established in terms of convolution
integral between the current modulation waveform, and an impulse response function
that expresses the thermal time constants n of the DFB structure:
(2.7 ) ( ) ( ) ( ).th tht i t h t
,
1
( ) exp( / ).N
th n th n
n
h t C t
The constant coefficients ,n thC are on the order of 0.1-0.5 GHz/mA. With prior
knowledge of the various adiabatic and thermal chirp parameters, the functional form of
the modulation current can be designed through numerical simulations to obtain a broad
and uniform power spectral density of the SBS pump wave.
The current modulating function was chosen to be of the form:
0.7
0
mod mod( ) ( ) ( ) 0.8 ( ) ( ).D N N
t T t Ti t i t i t i i i t
T T
(2.8)
where i and T denote the magnitude and the period of a deterministic
modulation term ( )Di t respectively, 0i is the DFB laser bias current, and
( )Ni t
represents a random modulation component of Gaussian statistics, bandwidth NB , and
root-mean-squared (RMS) magnitude N . The top of Fig. 2.1 shows ( )Di t , and the
bottom of the figure shows the corresponding calculated frequency variation
( ) ( ) ( )th at t t . The instantaneous optical-frequency shift oscillates between
min[ ( )]t and max[ ( )]t , which coincide with min[ ( )]Di t and max[ ( )]Di t ,
respectively. As a consequence, the instantaneous optical frequency is minimal when the
32
intensity of the pump laser is minimal and vice versa [92]. In order to compensate for
that amplitude modulation imbalance, ( )Di t is designed to “spend more time” in the
vicinity of its minimum. The random modulation term ( )Ni t is added to further reduce
the spectral fluctuation of ( )p pI within its bandwidth
/ 2 max[ ( )] min[ ( )]t t . The increasing of N improves the spectral
uniformity of the pump, at the expense of spectral transition sharpness [92]. Based on that
tradeoff, an intermediate value is chosen for the experiments.
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2-5
0
5
Time [micro-sec]
Mo
d.c
urr
[mA
]
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 20
0.5
1
Time [micro-sec]
Rel.
po
wer
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2-2
0
2
Time [micro-sec]
Fre
q. sh
ift
[GH
z]
Figure 2. 1. (Top) Deterministic component of the direct-modulation current of the pump laser
( )Di t (2.8), i =7.6 mA, and T = 100 ns. (Center) Corresponding relative output power. (Bottom)
Simulated instantaneous optical frequency variations. aC = 0.25 GHz/mA, 1,thC = 0.15 GHz/mA,
2,thC = 0.48 GHz/mA, and 1,2 = 20, 200 ns.
A schematic drawing of an SBS-ASE noise source is shown in Fig. 2.2. Light
from a distributed feedback (DFB) laser source is directly modulated by the output of an
arbitrary waveform generator and amplified by an EDFA to 500 mW. The spectrally
broadened light is launched into a 25 km long section of standard fiber through a fiber-
optic circulator. SBS-ASE is generated in the fiber, at the opposite direction to the pump
33
propagation. Careful synthesis of the pump laser direct modulation provided a uniform
power spectral density of the SBS-ASE within a range of 1 GHz. Fig. 2.3 (top) shows a
heterodyne measurement of the PSD of the modulated DFB laser diode subject to ( )Di t
of equation (2.8), which was used as a broadened SBS pump wave. Fig. 2.3 (bottom)
shows the PSD of the down-converted, UWB RF noise that was generated through SBS-
ASE. The PSD was nearly uniform within a range of 1 GHz
Circulator
FO cable
25Km
Arbitrary
Waveform
Generator
DFB Laser
Spontaneous
Brilouin
Broadened
Pump
EDFARF
Amplifier
Optical
BPF
Generated SBS-ASE
Noise
Figure 2. 2. Setup for SBS-ASE noise generation. FO: fiber-optic. BPF: band-pass filter
2000 2500 3000 3500 4000 4500 5000 5500 6000-80
-60
-40
PS
D[d
Bm
/MH
z]
Frequency[MHz]
-3000 -2000 -1000 0 1000 2000 3000-80
-70
-60
Optical Frequency Offset [MHz]
PS
D[d
Bm
/MH
z]
Figure 2. 3. Top – PSD of the directly modulated SBS pump wave. Bottom – PSD of the down-
converted SBS-ASE noise
34
2.1.3 UWB noise generation via ASE of EDFA
Erbium-doped fiber amplifiers (EDFA) make use of rare-earth elements as their
gain medium by doping the fiber core during the manufacturing process [11]. EDFAs
became a mainstay of optical communication links since they operate in the wavelength
region near 1.55 μm. Their deployment in WDM systems revolutionized the field of
fiber-optic communications. In addition to the stimulated process, in which a pumping at
a suitable wavelength provides gain through population inversion, a spontaneous
emission of photons also occurs in random manner. These randomly emitted photons
may interact with other dopant ions and thus lead to ASE noise. The gain bandwidth of
EDFAs typically exceeds 35 nm. In order to generate UWB noise waveform at GHz-
scale bandwidth, a small portion of the broad gain spectrum has to be filtered. In this
work the filtering is done by fiber-Bragg-gratings (FBG), which retain only about 10
GHz (~0.1nm) of the EDFA-ASE bandwidth.
The ASE total power at the output of an EDFA can be approximately written as
[99]:
2 ( 1) ( 1) .ASE spP n G h F G h (2.9)
where spn is the spontaneous emission factor that mainly dependent on the degree of
population inversion, h is the Plank constant, is the central optical frequency, G is the
EDFA power gain, and is the noise bandwidth. The noise figure of the EDFA is
given by 2 spF n . While EDFAs provide broader noise waveforms, the use of SBS-
ASE allows for flexible adjustments of the UWB spectral width and shape through
synthesis of the pump wave.
35
2.2 Set-up of microwave-photonic radar system
A simplified block diagram of the proposed microwave-photonic UWB noise
radar system is shown in Fig. 2.4. A source of optical noise in a central office unit is
coupled together with a CW local oscillator (LO), which is spectrally detuned form the
noise central frequency by a few GHz. The combined optical field is split in two
branches. The light in one path is detected by a broadband photo-diode within the central
office. Upon detection, beating with the LO down-converts the optical noise to the RF
domain. Heterodyne detection is necessary to spectrally separate the interference term of
interest from base-band fluctuations due to the optical noise intensity. In addition,
heterodyne detection provides freedom to select a central radio frequency of choice. The
resulting electrical noise is sampled and stored as a reference.
Light in the other branch is transmitted over fiber towards a remote antenna unit,
where heterodyne beating is used once again to generate a replica of RF noise that is
stored at the central office. The resulting electrical waveform is amplified and
transmitted by a directive antenna towards potential targets. Reflected signals are
collected by a second directive antenna, amplified and carried back over fiber towards
the central office via direct modulation of a DFB laser diode source. The received
waveform is detected, sampled and correlated against the stored reference using digital
signal processing. The distances of targets are recovered based on the timing of
correlation peaks.
36
Figure 2. 4 Simplified block diagram of a microwave-photonic, UWB noise radar system. TX:
transmitter. RX: receiver
2.2.1 Correlation domain analysis, match-filtering and side lobe suppression
At the UWB noise radar receiver, the returning echoes s t are cross-correlated
with a replica of the launched waveform h t to obtain an impulse response:
( ) ( ) ( ) ( ) .oS s t h t s t h t dt
(2.10)
The cross-correlation produces a peak at a delay which corresponds to the
round-trip propagation time from the noise source to the target and back to the
processing unit. Any off-peak residual correlation manifests as background noise in the
radar trace. Use of the launched waveform itself as reference, known as matched
filtering, is known to maximize the ratio of intended peak strength to that of the
background in the presence of additive white Gaussian noise in the channel [3]. The
temporal width of the correlation peak determines the spatial resolution of the radar
system. The resolution is often quantified as the full width at half maximum of the
correlation main lobe (see Fig. 2.5).
37
As discussed in the introduction with respect to sequence coding schemes, the
reduction of correlation sidelobes is critical to the detection of weak or multiple targets.
The sidelobe strength is sensitive to the specifics of the waveform being used. The
sidelobe suppression performance of a radar system is quantified in terms of two primary
figures of merit: the peak-to-sidelobe-ratio (PSLR), and the integrated-sidelobe-ratio
(ISLR) (see Fig. 2.5 and chapter 1).
Figure 2. 5. Definitions of PSLR, ISLR and resolution of the impulse response of UWB noise radar
waveforms.
For example, a matched filter for a waveform of rectangular PSD shape provides
a sinc-shaped auto-correlation function with a PSLR of -13dB (see Fig. 2.6). In various
radar applications further suppression of the sidelobes is necessary, in order to improve
the dynamic range and contrast. The correlation side-lobes can be suppressed by
applying an amplitude weighting function to the waveform in the frequency domain [73].
Other techniques, such as CLEAN and PROJECTION [100, 101], identify the strongest
peaks of the correlation function, and then employ iterative procedures such as least-
squares methods to find the optimal filter that will further reduce the sidelobes [100]. As
opposed to windowing techniques, the CLEAN and PROJECTION techniques require
manual intervention in setting a threshold and selecting the target peaks based on some
a-priori knowledge. Nevertheless, in some systems these techniques are the only viable
ones.
38
0.99 0.995 1 1.005 1.01
x 104
-15
-13
-10
-5
0
Rel
ativ
e C
orr
leat
ion
Mag
.[d
B]
Samples
Figure 2. 6. A simulated auto-correlation of a UWB noise waveform with a 1 GHz-wide uniform
PSD. The central radio frequency of the waveform in the particular example was 3.5 GHz.
2.3 Radar link budgets
In this section, the expected range and resolution of a UWB noise radar system
are estimated, taking into consideration the noise and saturation of its various constituent
components. As discussed earlier, the spatial resolution of the system is inversely
proportional to the operating bandwidth [3]:
2
cD
B
. (2.11)
where c is the speed of light in vacuum and B is the total bandwidth of the
transmitting source. The SBS-ASE source used in our experiments, for example,
produces noise of 1 GHz bandwidth that corresponds to a spatial resolution of about 15
cm.
The ranging accuracy is related to the signal bandwidth and the acquisition SNR
according to [102]:
2 2
R
c
B SNR
. (2.12)
39
For example, a noise source of 1 GHz bandwidth at an SNR of 20 dB yields
ranging accuracy of about 1 cm.
In order to estimate the optical signal to noise ratio (OSNR) and the electrical
signal to noise ratio (ESNR), the noise contributions of various mechanisms need to be
evaluated (see chapter 1 for the introduction of noise mechanisms). Noise contributions
in the electrical and radio-frequency parts of the system include the thermal and
electronic noise of the RF amplifier and the sampling error of the digitizing oscilloscope.
The contribution of the RF amplifier is quantified in terms of its noise figure NF = 7
dB. The noise figure represents the degradation in the ESNR going through the
amplifier. The noise floor of the oscilloscope, while operated at maximum sensitivity, is
on the order of 0.2 mV.
Noise in the optical parts consists of the photo-detectors thermal noise, and
intensity noise of both the local oscillator used in heterodyne detection and the DFB laser
used for transmission from the remote antenna unit towards the central office. Detector
noise is often quantified in terms of its noise-equivalent power or NEP (see section 1.3.4
for more details):
PDP NEP BW (2.13)
The NEP of our photo-detector equals 1010W
Hz
. I assume a measurement
bandwidth that matches the spectral width of the SBS-ASE source, BW = 1 GHz. The
thermal noise is therefore equivalent to the photo-current that is generated by the
detection of optical power PDP = -25 dBm. The fluctuations in optical power due to RIN
are given by:
/1010RIN
RIN DCP BW P (2.14)
41
where RIN is typically on the order of -155dB/Hz in InGaAsP laser diodes
operating at 1.55 µm, and DCP is the operating DC optical power. For a DFB laser
operating at an average output power DCP of +5 dBm, RINP is on the order of -27 dBm.
The contributions of thermal noise band RIN are therefore comparable. The overall
optical noise, combining the two contributions, can be estimated as -25 dBm.
Next, the strength of the signal and the noise power will be estimated at different
points along the path of the system, leading eventually to the sampling oscilloscope (see
Fig. 2.7). The maximum optical power that of SBS-ASE incident upon the photo-
detector in the remote antenna base is restricted to 0 dBm by its saturation. Therefore, the
OSNR of the photo-generated UWB noise, prior to its launch towards the remote antenna
base, is 25 dB at best, and the corresponding ESNR is 50 dB.
PC
1x2 Coupler
FO cable 10kmTunable Laser (LO)
TX Antenna
RX Antenna
BPF
BPF
Oscilloscope
DFB Laser Detector
Detector
Detector 1x2 Splitter
Remote Antenna Base
FO cable 10km
RF
Amplifier
Central OfficeUpstream
Downstream
Photonic
Noise
Generator
RF
Amplifier
Figure 2. 7. Simplified block diagram of a microwave-photonic, UWB noise radar system. FO: fiber-
optic.
The photo-detected UWB RF noise is amplified at the transmitter to an electrical
power level tP of 10 dBm. The electrical power of the collected RF echoes at the output
of the receiver antenna is given by the radar equation [103]:
2 2(4 )
t tr e
P GP A
R
(2.15)
41
Here tG = 20 dBi is the directional gain of the antenna, is the cross section of
the target in units of m2 (to be defined shortly), R is the one way distance to target and
eA is the antenna effective aperture which equals:
2
4
te
GA
(2.16)
In equation (2.16), = 8.5 cm is the central wavelength of the optically-
generated RF UWB noise. Given the parameters of our experiment as detailed above, the
effective aperture equals 0.06 m2. The radar cross section of a rectangular flat plate
can be calculated using the following equation (see 2.5.6 in [103]):
2 2
2 2
2
4 sin( sin )( ) (cos )
sin
a b ak
ak
(2.17)
Here ,a b are the length and the width of the target, 2k , and is the angle
between the surface normal of the plate and the direction to the incoming radar beam. In
the experiments, a 40 cm x 40 cm metal target was located at an angle 15 with
respect to the transmitted beam. The cross-section represented by the target equals 0.67
m2. Substituting the target cross section and the effective aperture into the radar equation
(2.13), the following simple expression is obtained for the received electrical power as a
function of distance, for our experimental parameters:
4
0.254[mW]rP
R (2.18)
where the distance is given in meters. The received power is amplified by 16 dB
at the receiver. Therefore, the electrical power that is used in modulating the DFB laser
transmitter is approximately given by:
mod 4
10[mW]P
R (2.19)
42
At the maximum experimental range, the power attenuation according to the
radar equation exceeds the gain provided by the transmitting RF amplifier. In these
conditions, the contribution of the transmitting amplifier to the overall noise becomes
negligible, and the RF noise is dominated by that of the receiving-end amplifier (see Fig.
2.7). Hence the ESNR of the modulating signal is degraded by the noise figure of 7 dB,
to 43 dB.
Given the 50 impedance of the DFB laser transmitter, the UWB modulating
current for down-link transmission back towards the central office is given by:
mod 2
20[mA]i
R (2.20)
where the distance to the target is again given in m. The slope efficiency of the
DFB is on the order of 0.1 mW / mA. The optical power is further attenuated by 3 dB
due to losses over 10 km of optical fiber and connectors. Therefore, subject to our
experimental conditions, the optical power of the received UWB noise waveform that
reaches the central office is related to the target as:
2
1[mW]remoteP
R (2.21)
The OSNR of the down-link waveform is 21.5 dB. This is the maximum OSNR
attainable with our setup.
In order to set an upper bound on the measurement range, numerical simulation
of the impulse response function of the UWB noise waveform, in the presence of
additive noise, were carried out. These simulations attempted to identify the ESNR for
which the sidelobe suppression metrics of the impulse response function begin to
degrade. The results suggest that the PSLR degrades considerably for an ESNR below 0
dB (see Fig. 2.8). Therefore, the longest measurement distance is that for which the
43
incoming signal power remoteP equals the noise equivalent power of the central office
photo-detector, which was previously estimated to be on the order of -25 dBm. The
signal power drops to that limiting level for R = 15 m. Lastly, the responsivity of the
amplified photo-detector is on the order of 1000 [V/W]. Therefore, the voltage
magnitude of the photo-detected signal for the maximum target range is on the order of 3
mV, exceeding the noise of the oscilloscope.
-20-15-10-5051015200
5
10
13
SNR[dB]
PS
LR
[dB
]
Figure 2. 8. Simulated PSLR as function of ESNR for a Gaussian distributed noise of 1B GHz.
2.4 Experimental results
A list of equipment used in the experiments is given in Table III.
44
TABLE III
LIST OF EQUIPMENT USED IN MICROWAVE PHOTONIC UWB NOISE RADAR EXPERIMENTS
2.4.1 UWB noise radar range measurements based on SBS-ASE
The above optical UWB noise source was employed in a microwave photonic
radar system (see Fig. 2.2). The experimental values of the various system parameters
COMPONENT MANUFACTURER MODEL
AWG 5GS/S TEKTRONIX AWG7051
RF AMPLIFIER MINI-CIRCUITS ZHL-42W
DFB LASER 1550 nm ORTEL 3740A-001-002
EDFA KEOPSYS
CIRCULATOR
25 KM SMF
PHOTODETECTOR 12
GHZ NEW FOCUS 1544-A
RF BPF MINI-CIRCUITS VBF-1445+
DIRECTIVE ANTENNA
20 dBi PASTERNACK
ENTERPRISES PE9862
OSCILLOSCOPE 6 GHZ AGILENT 90604A
TUNABLE LASER NEW FOCUS TLB-6600
3 nm OPTICAL BAND
PASS FILTER 1530-
1570 nm
OEFTF-100
RF SPECTRUM
ANALYZER ANRITSU MS2724B
OPTICAL SPECTRUM
ANALYZER 600-1700
nm YOKOGAWA AQ6370B
POWER SUPPLY
RF SMA CABLES
FIBER OPTIC FC/APC
CABLES
RF M/F CONNECTORS
OPTICAL CONNECTORS
PC WITH MATLAB
45
-5 -4 -3 -2 -1 0 1 2 3 4 5-60
-50
-40
-30
-20
-10
0
Imp
uls
e r
esp
. M
ag
. [d
B]
Time [ns]
were provided in the preceding section. Figure 2.9 shows a histogram of the UWB SBS-
ASE noise, which is of Gaussian statistics as expected.
-0.1 -0.05 0 0.05 0.1 0.150
0.5
11
Voltage
Rel.
Occu
ran
ce
Figure 2. 9. Histogram of the down-converted SBS-ASE noise (bar), alongside a zero-mean
Gaussian distribution of equal variance (line).
Fig. 2.10 shows the experimental impulse response of the SBS-ASE noise
waveform, with 1B GHz and 0 3.5f GHz . The PSLR of the generated waveform was
-13dB, in agreement with the expectations .The full width at half maximum (FWHM) of
impulse response main lobe was 0.5 ns, which is equivalent to a ranging resolution of 15
cm, in agreement with the noise bandwidth.
Figure 2. 10. SBS-ASE impulse response ( 1B GHz 0 3.5f GHz ).
46
0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5-25
-20
-15
-10
-5
0
5
10
15
20
25
Distance [m]
Rela
tive C
orr
. M
ag. [L
in.]
1.5 m
1.95 m
2.85 m
Fig. 2.11 shows the measured cross-correlations between the reflections from a
40 cm x 40 cm metallic target and the reference waveform stored at the central office, for
several target distances. The full width at half maximum (FWHM) of the main auto-
correlation peak was equivalent to a ranging resolution of 15 cm, in agreement with the
noise bandwidth. The received power varied between measurements due to the manual
placement of antennas and target. Fig. 2.13 shows the experimentally obtained cross-
correlation trace between the reflected waveform from a pair of nearby metallic targets,
and the central office reference. The two targets were separated by 15 cm (see Fig. 2.12).
The two distinct targets are readily resolved.
Figure 2. 11. Measured correlation between SBS-ASE noise waveform reflected from a metal target
and a reference replica. The distances to the target were 1.5 m (blue, dotted); 1.95 m (red, solid) and
2.85 m (black, dashed).
47
0.5 1 1.5 2 2.5
-20
-15
-10
-5
0
Distance [m]
Rela
tiv
e C
orr
ela
tio
n M
ag
. [d
B]
Figure 2. 13. Cross-correlation between the waveform reflected from a pair of metallic targets
separated by 15 cm, and the central-office reference waveform. The targets are resolved by the radar
system.
Fig. 2.14 shows the cross-correlation between the reflections from a concealed,
20-cm long metallic object, and the reference waveform. The object was placed about 3
m from the antennas.
Figure 2.12. Two targets separated by 15 cm.
48
0 1 2 3 4 5 6-30
-25
-20
-15
-10
-5
0
Rela
tiv
e C
orr
ela
tio
n.
Mag
. [d
B]
Distance [m]
Figure 2. 14. Cross-correlation between the waveform reflected from a 20 cm-long, concealed
metallic object and the central office reference waveform.
2.4.2 UWB noise radar range measurements based on EDFA-ASE
A UWB noise source generated by EDFA-ASE was also employed in a
microwave photonic radar system. Here too, the transmission and reception horn
antennas at the remote unit had a directive gain of 20 dBi, and the RF power of the
transmitted waveform was 10 dBm. The PSD and histogram of the RF UWB noise that
was generated using EDFA-ASE are shown at Fig. 2.15. A 10 GHz-wide FBG was used
for slicing the optical spectrum of the ASE, and the RF waveform was further filtered to
accommodate the 2 GHz bandwidth of the antennas. Once more, the generated noise is
of Gaussian statistics. In this experiment, a pair of 10-km-long fibers was used to
connect between the central and remote units. Fig. 2.16 shows the experimental
matched-filtered form of the EDFA-ASE noise waveform, with 2B GHz and
0 3.5f GHz . The PSLR of the generated waveform was -14dB. The full width at half
maximum (FWHM) of impulse response main lobe was 0.3 ns which is equivalent to a
ranging resolution of 10 cm, in agreement with the noise bandwidth.
49
-0.8 -0.4 0 0.4 0.80
0.5
1
Voltage
Rel.
Occu
ran
ce
2000 2500 3000 3500 4000 4500 5000-80
-60
-40
Frequency[MHz]
TX
PS
D [
dB
m/M
Hz.]
Figure 2. 15. Top – PSD of the down-converted EDFA-ASE noise. Bottom –histogram of the down-
converted EDFA-ASE noise (bar), alongside a zero-mean Gaussian distribution of equal variance
(line).
-5 -4 -3 -2 -1 0 1 2 3 4 5-60
-50
-40
-30
-20
-10
0
Imp
uls
e r
es.
Mag
. [d
B]
Time [ns]Figure 2. 16. Impulse response of a UWB RF noise generated through EDFA-ASE
( 2B GHz 0 3.5f GHz ).
51
0 0.5 1 1.5 2 2.5 3 3.5 4-60
-40
-20
0
20
Rela
tiv
e C
orr
ela
tio
n M
ag
. [d
B]
Distance [m]
2.16 m
1.9 m
1.43 m
Fig. 2.17 shows the measured cross-correlations between the reflections from a
40 cm x 40 cm metallic target and the reference waveform stored at the central office, for
several target distances. The ranging accuracy was on the order of 10 cm.
Figure 2. 17. Measured correlation between EDFA-ASE noise waveform reflected from a metal
target and a reference replica. The distances to the target were 1.43 m (red, solid); 1.9 m (green,
dashed) and 2.16 m (blue, dashed).
2.5 Concluding remarks
In this chapter, a microwave-photonic, ultra-wideband (UWB) noise radar system
was proposed and demonstrated. The system brings together photonic generation of
UWB waveforms and fiber-optic distribution. The system relies on the ASE noise that is
associated with SBS in a standard optical fiber, or with an EDFA. The system combines
the fiber-optic antenna remoting capabilities, broad bandwidth and flexibility of
microwave-photonic processing, together with the immunity against interception and
jamming of noise radars. The system resolution is not affected by the fiber link, and can
be improved with a broader bandwidth of antennas and RF amplifiers.
51
Ranging measurements were carried out for both SBS and EDFA noise sources.
The statistics of the optically-generated RF UWB noise waveforms are Gaussian. The
widths of the main-lobe of impulse response functions are inversely proportional to the
signals bandwidths. The detection of a concealed metallic object and the resolving of two
targets with the anticipated ranging resolution had also been demonstrated. The upper
bound on the ranging accuracy, for the 2 GHz bandwidth of noise waveforms used in the
experiments and at high SNR, is estimated as 0.5 cm but was not evaluated
experimentally. The systems link budget suggests a maximum operating distance of 15
m, limited by the detector noise floor and the transmitted RF power. The operating
distance may be expanded with higher transmitting power and active RF amplifier
control. The sidelobes of the correlation function may be suppressed using signal
processing methods at either the receiver or the transmitter end.
Microwave photonics allows for agile reconfiguration of the RF carrier
frequency. This property may be utilized to build a multipurpose radar system, using
several UWB noise source bands in order to detect and sort different targets with
different resonant frequency responses. Such systems would require broader bandwidth
of antennas and amplifiers, and a very stable local oscillator.
52
CHAPTER
3
LADAR with incoherent pulse compression
This chapter addresses the compression of an incoherently detected train of
unipolar pulse sequences, and its application to a laser range-finder system. The
compression principle relies on a unipolar representation of known bipolar phase codes,
such as maximum peak-to-sidelobe (MPSL) sequences and complementary pairs. After
introducing the underlying principle, simulations of the incoherent compression of
unipolar derivatives of MPSL codes, 82 bits and 1112 bits in length, as well as
complementary pair codes,416 bits and 832 bits in length, are reported. Simulations are
carried out for different SNRs. Next, the incoherent compression of the 1112 bits-long
MPSL sequence and the 832 bits-long complementary pair are demonstrated
experimentally, using a simple optical link: the sequences are used to drive an electro-
optic amplitude modulator, and they are recovered through simple direct detection. The
sidelobes of the compressed waveform are suppressed by as much as 46 dB and 42 dB
for MPSL and complementary code respectively, with respect to the main correlation
peak power. Lastly, the principle is used in a laser range-finder setup demonstration. A
spatial resolution of 3 cm is achieved.
3.1 Coding principle
The primary motivation for incoherent pulse compression is to try and obtain the
sidelobe suppression performance that can be provided by phase-coded pulse sequence,
53
while employing simple direct detection that is fundamentally phase-insensitive. The
principle is of particular consequence in LADAR schemes, since it circumvents the need
for complicated optical coherent receivers.
3.1.1 Coding procedure
Consider a bipolar code of length N :
[ ]c n , such as an MPSL sequence or other
(see chapter 1 for a discussion of different phase-coded pulses sequences), where
1....n N . A unipolar code of length 2N is generated based on [ ]c n by applying
Manchester coding: if [ ] 1c n , then [2 1] 1T n and [2 ] 0T n . For
[ ] 1c n ,
[2 1] 0T n and [2 ] 1T n are chosen instead [87]. Manchester coding converts the
bipolar phase information into pulse-position modulation, and it is used in optical
communication [104]. The code T would later be used in intensity modulation of the
LADAR light source.
A bipolar matched filtering sequence R of length 2N is constructed in a similar
manner: [ ]R k is set to 1 if [ ] 1T k and equals -1 if [ ] 0T k , 1...2k N [87]. The code
R is digitally stored at the receiver for post-detection processing. Using a matched
bipolar reference signal instead of a unipolar signal results in a cross-correlation
(between T and R ) with an average value of zero. Since the sequence R is used only
digitally, its bipolar nature does not overburden the LADAR setup.
As an example, the construction of the T and R codes corresponding to the
Barker 13 bipolar sequence is illustrated in Fig. 3.1. The aperiodic cross-correlation
between these two codes is shown in Fig. 3.2 (bottom), alongside the aperiodic auto-
correlation of the original Barker 13 sequence itself (top). With the exception of the two
sidelobes immediately adjacent to the main correlation peak, the cross-correlation
replicates the sidelobe suppression of the original bipolar code [87]. In contrast, the
54
cross-correlation of a unipolar representation [ ]c n of the Barker code itself, in which -1
symbols are simply replaced by 0, exhibits inferior sidelobe suppression performance
(Fig. 3.2, center).
The cross-correlation sidelobes can be further suppressed using a mismatched
filtering process, in which the sequence R is replaced by a longer code R whose
coefficients are not restricted to 1. Substantial sidelobe suppression can be obtained, at
the cost of a modest degradation in the central correlation peak power [3]. The sequence
R can be designed to maximize the ISLR, according to principles described in sec. 6.6
of [3].
0 5 10 15 20 25-1
0
1
bit
Mag
nitu
de
0 5 10 15 20 25-1
0
1
bit
Mag
nitu
de
Figure 3. 1. Transmitted code T (top) and matched filtering code R (bottom),corresponding to
the Barker 13 code: [+1 +1 +1 +1 +1 -1 -1 +1 +1 -1 +1 -1 +1].
55
In coherent receivers, negative sidelobes (such as in Fig 3.2. bottom) can cause
two problems [87]: 1) They could mask-out the main lobe of a nearby weaker target; and
2) Due to possible phase change of the carrier frequency of the reflected signal, negative
sidelobes can change their sign. Together with measurement noise, the negative
sidelobes could lead to false alarms or miss-detections. In incoherent receivers, the
current that is directly provided by a photo-detector is phase insensitive, so problem 2 in
unlikely to happen. The strong negative sidelobes can still mask a weaker target, but
only if the delay difference between the two targets matches the duration of a single code
bit. For a relatively wide reflection targets, the negative sidelobes can be differentiated,
emphasizing edges while still maintaining the system resolution [87]. Without loss of
generality, the two negative sidelobes immediately adjacent to the main correlation peak
will be neglected for the rest of this chapter.
0 5 10 15 20 25-5
5
15
bit
Cor
r.
0 5 10 15 20 25-5
5
15
bit
Cor
r.
0 10 20 30 40 50-5
5
15
bit
Cor
r.
Figure 3. 2. Top – aperiodic auto-correlation of the Barker 13 bipolar code: [+++++--++-+-+]. The
correlation peak is 13, whereas the maximal sidelobe equals unity. Center – aperiodic auto-correlation
of a unipolar representation of the Barker 13 code: [1111100110101], showing a weaker central peak
and inferior sidelobe suppression. Bottom – aperiodic cross-correlation between the transmitted code
T and matched filtering code R corresponding to the Barker 13 bipolar code (see Fig. 3.1). With the
exception of the two time slots in the immediate vicinity of the central peak, the suppression of
sidelobes reaches that of the original bipolar sequence.
56
3.1.2 Simulated sidelobe suppression
In order to evaluate the proposed method for incoherent pulse compression, its
performance was simulated for two Manchester-code MPSL codes of 82 and 1112 bits
long respectively, and two complementary pair codes of 416 and 832 bits long
respectively. The MPSL codes themselves are provided in Appendix A.
3.1.2.1 MPSL 82
First an 82 bits-long MPSL code was used. The transmitted, Manchester coded
sequence T is illustrated in Fig. 3.3 (top), alongside its matched reference R (center).
The noise-free aperiodic cross-correlation between these two codes is shown in Fig. 3.3
(bottom), with a projected PSLR of 26.24 dB. The miss-matched reference R , specially
designed for the sequence T , is illustrated in Fig. 3.4 (center), alongside the noise free
aperiodic cross-correlation between T and R . Note that the sequence R is three time
longer than R , and that its values are not restricted to 1. The PSLR is improved by 8
dB, while the main lobe power is attenuated by 0.9 dB (Fig. 3.4, bottom).
0 10 20 30 40 50 60 70 80-0.5
0
0.5
1
1.5
Time / chip duration
Sig
nal
Manchester-coded MPSL 82
0 10 20 30 40 50 60 70 80
-1
0
1
Time / chip duration Matc
hed
refe
ren
ce
0 500 1000 1500 2000 2500 3000 3500-50
-26.24
0
Samples
Rel.
Co
rr. [d
B]
Figure 3. 3. Transmitted code T (top) and matched filtering code R (center), corresponding to the
MPSL 82 code. Bottom – aperiodic cross-correlation between the transmitted code T and matched
filtering code R .
57
Next, additive Gaussian distributed noise was introduced to the simulation. The
incoherently compressed forms of the 82 bits-long MPSL sequence are shown in Fig.
3.5, for different SNR values and for both matched and mismatched filtering. At a high
SNR of 20 dB, the PSLR of the match-filtered sequence was 26 dB, and a mismatched
filter further improved the PSLR to 32 dB, with a 1 dB attenuation of the main lobe (top
row). When the noise and signal power levels are equal (center row), the simulated
PSLR was 16.4 dB, and the sidelobe suppressions obtained with matched and
mismatched filters were practically equal. At a negative SNR of -5 dB (bottom row), the
main correlation peak cold still can be recovered at a PSLR of about 6.7 dB. Here too,
the mismatched filter gave no benefit in sidelobe suppression. Sidelobe suppression was
further tested in an extremely noisy condition, at which the noise power is 100 times
larger than that of the signal (Fig 3.6). In this scenario, the main correlation peak can no
longer be recovered.
0 50 100 150 200-0.5
0
0.5
1
1.5
Time / chip duration
Sig
nal
Manchester-coded MPSL 82
0 50 100 150 200-2
0
2
Time / chip duration
Mis
s. r
efe
ren
ce
0 1000 2000 3000 4000 5000 6000 7000-50
-34.6
-0.9085
Samples
Rel.
Co
rr. [d
B]
Figure 3. 4. Transmitted code T (top) and miss-matched filtering code R (center ),corresponding to
the MPSL 82 code. Bottom – aperiodic cross-correlation between the transmitted code T and matched
filtering code R .
58
3.1.2.2 MPSL 1112
Similar simulations were also carried out for a 1112 bits-long MPSL code. The
transmitted Manchester coded sequence T is illustrated in Fig. 3.7 (top), alongside its
matched reference R (center). The noise-free aperiodic cross-correlation between R and
T is illustrated in Fig. 3.7 (bottom) with a PSLR of 33.3 dB.
Here too, the sidelobes can be further suppressed using a specially designed
mismatched sequence R (Fig 3.8 center). The noise-free aperiodic cross-correlation
300 400 500 600 700 800 900 1000
0
-26
Rel.
Co
rr. [d
B]
Samples
SNR +20 dB
0 200 400 600 800 1000
0
-32
Rel.
Co
rr. [d
B]
Samples
SNR + 20 dB
300 400 500 600 700 800 900 1000
0
-16.4
Rel.
Co
rr. [d
B]
Samples
SNR 0 dB
0 200 400 600 800 1000
0
-16
Rel.
Co
rr. [d
B]
Samples
SNR 0 dB
300 400 500 600 700 800 900 1000
0-6.7
Rel.
Co
rr. [d
B]
Samples
SNR -5 dB
0 200 400 600 800 1000
0-6
Rel.
Co
rr. [d
B]
Samples
SNR -5 dB
Figure 3. 5. Cross-correlations of incoherently compressed, 82 pulses-long unipolar sequences.
Both matched (left, blue) as well as mismatched (right, black) filters were used in the compression
process. Top row: simulated compression with a signal-to-noise ratio of +20 dB. Center row:
simulated compression with a signal-to-noise ratio of 0 dB. Bottom row: simulated compression
with a signal-to-noise ratio of -5 dB (see text).
300 400 500 600 700 800 900 1000-60
-50
-40
-30
-20
-10
0
Rel
. Cor
r. [d
B]
Samples
SNR -20 dB
0 200 400 600 800 1000-60
-50
-40
-30
-20
-10
0
Rel
. Cor
r. [d
B]
Samples
SNR -20 dB
Figure 3. 6. Cross-correlations of an incoherently compressed, 82 pulses-long unipolar sequence
with a signal-to-noise ratio of -20 dB. Both matched (left, blue) as well as mismatched (right,
black) filters were used in the compression process.
59
between R and T is illustrated in Fig. 3.8 (bottom). The PSLR in this case equals 51.6
dB, with a 0.9dB attenuation to the main lobe.
Sidelobe suppression at +20 dB, 0 dB, and -20 dB SNR levels are illustrated in
Fig. 3.9. At a high SNR of 20 dB, the PSLR of the matched filtered sequence reached
2000 2020 2040 2060 2080 2100 2120 2140 2160 2180 2200-0.5
0
0.5
1
1.5
Time / chip duration
Sig
nal
Manchester-coded MPSL 1112
2000 2020 2040 2060 2080 2100 2120 2140 2160 2180 2200-2
0
2
Time / chip duration
Mis
s. R
efer
ence
0 2 4 6 8 10 12
x 104
-51.58
-0.8892
Samples
Rel
. C
orr
. [d
B]
400 420 440 460 480 500 520 540 560 580 600-0.5
0
0.5
1
1.5
Time / chip duration
Sig
nal
Manchester-coded MPSL 1112
400 420 440 460 480 500 520 540 560 580 600
-1
0
1
Time / chip duration Mat
ched
Ref
eren
ce
0 0.5 1 1.5 2 2.5 3 3.5 4 4.5
x 104
-50
-33.32
0
Samples
Rel
. C
orr
. [d
B]
Figure 3. 7. Transmitted code T (top) and matched filtering code R (center),corresponding to the
MPSL 1112 code. Bottom – aperiodic cross-correlation between the transmitted code T and matched
filtering code R
Figure 3.8 Transmitted code T (top) and miss-matched filtering code (center) corresponding to the R
(center), MPSL 1112 code. Bottom – aperiodic cross-correlation between the transmitted code T and
miss-matched filtering code R
61
33.3 dB (top row), and a mismatched filter further improved the PSLR to 48.8 dB with 1
dB attenuation of the main lobe. At an SNR level of 0 dB, the PSLR for matched and
mismatched filtering reached 25.2 dB and 24.6 dB respectively, with a miss-match-
induced loss of 1 dB to the main lobe. This time, long length of the sequence allowed for
a recovery of the main lobe even at poor SNR levels as low as -20 dB, with a PSLR of 9
dB. Generally speaking, mismatch filters provide no added value for SNR levels below 0
dB.
3.1.2.3 416 bits-long complementary pair
This time, 416 bits-long complementary pair codes were simulated. The code
was generated by applying 1.1 (see section 1.3.7) to the 26 element primitive pair of
Table I (see section 1.3.7) 4 times. The transmitted Manchester coded sequence T is
illustrated in Fig. 3.10 (top), alongside its matched reference R (center). The noise-free
aperiodic cross-correlation between R and T is illustrated in Fig. 3.10 (bottom) with a
PSLR of 58 dB. The ideal zero-sidelobes correlation property of the complementary pair
4000 6000 8000 10000 12000 14000
0
-31.8
Rel
. C
orr
. [d
B]
Samples
SNR +20 dB
2000 4000 6000 8000 10000 12000 14000
0
-43
Rel
. C
orr
. [d
B]
Samples
SNR +20 dB
4000 6000 8000 10000 12000 14000
0
-25.2
Rel
. C
orr
. [d
B]
Samples
SNR 0 dB
0 5000 10000 15000
0
-24.6
Rel
. C
orr
. [d
B]
Samples
SNR 0 dB
2000 4000 6000 8000 10000 12000 14000
0
-8.2
Rel
. C
orr
. [d
B]
Samples
SNR -20 dB
4000 6000 8000 10000 12000 14000
0
-9.1
Rel
. C
orr
. [d
B]
Samples
SNR -20 dB
Figure 3. 9. Cross-correlations of an incoherently compressed, 1112 pulses-long unipolar
sequence. Both matched (left, blue) as well as mismatched (right, black) filters were used in the
compression process. Top row: simulated compression with a signal-to-noise ratio of +20 dB.
Center row: simulated compression with a signal-to-noise ratio of 0 dB. Bottom row: simulated
compression with a signal-to-noise ratio of -20 dB (see text).
61
is nearly preserved by the Manchester encoding, with a PSLR of 1/(2N) instead of zero
where N is the length of the each code in the pair [105] .
Sidelobe suppression at +20 dB, 0 dB, and -20 dB SNR levels are illustrated in
Fig. 3.11. At a high SNR of 20 dB, the PSLR reached 46.7 dB (top row). At an SNR
level of 0 dB, the PSLR reached 25.7 dB (center). At low SNR level of -20 dB the PSLR
is degraded to 8 dB (bottom row), though main lobe remains discernible.
0 500 1000 1500 2000 2500 3000 3500-0.5
00.5
11.5
Sign
al
Samples
Manchester-coded 416 complementary pair
0 500 1000 1500 2000 2500 3000 3500-2
0
2
Mat
ched
ref
eren
ce
Samples
4000 4500 5000 5500
-58
0
Rel
. Cor
r. [
dB]
Samples
Figure 3. 10. Transmitted code T (top) and match-filter code R (center) corresponding to the 416
bits-long complementary pair code. Bottom – aperiodic cross-correlation between the transmitted
code T and the matched-filter code R.
62
4000 4500 5000 5500
-46.7
0
Rel
. Cor
r. [d
B]
Samples
SNR +20 dB
4000 4500 5000 5500
-25.7
0
Rel
. Cor
r. [d
B]
Samples
SNR 0 dB
4000 4500 5000 5500
-8
0
Rel
. Cor
r. [d
B]
Samples
SNR -20 dB
3.1.2.4 832 bits-long complementary pair
Lastly, 832 bits-long complementary pair codes were simulated. The code was
generated by applying 1.1 to the 26 element primitive pair of Table I 5 times. The
transmitted Manchester coded sequence T is illustrated in Fig. 3.12 (top), alongside its
matched reference R (center). Following the enlargement of the code length by factor of
2, the noise-free aperiodic cross-correlation between R and T has a PSLR of 64 dB
(Fig 3.12 bottom).
Again, sidelobe suppression at +20 dB, 0 dB, and -20 dB SNR levels are
illustrated in Fig. 3.13. At a high SNR of 20 dB, the PSLR reached 50 dB (top). At an
SNR level of 0 dB, the PSLR reached 29 dB (center). At low SNR level of -20 dB the
PSLR is degraded to 13 dB (bottom).
Figure 3. 11. Cross-correlations of an incoherently compressed, 416 pulses-long complementary
code. Top row: simulated compression with a signal-to-noise ratio of +20 dB. Center row:
simulated compression with a signal-to-noise ratio of 0 dB. Bottom row: simulated compression
with a signal-to-noise ratio of -20 dB (see text).
63
0 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 5500-0.5
0
0.5
1
1.5
SamplesS
ignal
Manchester-coded 832 complementary pair
0 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 5500-1
0
1
Samples
Sig
nal
5000 5500 6000 6500 7000 7500 8000-64
0
Samples
Rel
. C
orr
. [d
B]
3.1.3 Experimental sidelobe suppression
The scheme and components list for LADAR measurements using incoherent
pulse compression are shown in Fig. 3.14 and in Table IV respectively. Light from a
laser diode at 1550 nm wavelength passed through a Mach-Zehnder electro-optic
Figure 3.12 Transmitted code T (top) and matched filtering code R (center), corresponding to the
832 bits-long complementary pair code. Bottom – aperiodic cross-correlation between the
transmitted code T and matched filtering code R .
4000 5000 6000 7000 8000 9000 10000 11000 12000
-50
0
Rel
. Cor
r. [d
B]
Samples
SNR +20 dB
4000 5000 6000 7000 8000 9000 10000 11000 12000
-29
0
Rel
. Cor
r. [d
B]
Samples
SNR 0 dB
4000 5000 6000 7000 8000 9000 10000 11000 12000
-13
0
Rel
. Cor
r. [d
B]
Samples
SNR -20 dB
.
Figure 3. 13. Cross-correlations of an incoherently compressed, 832 pulses-long complementary
code. Top: simulated compression with a signal-to-noise ratio of +20 dB. Center: simulated
compression with a signal-to-noise ratio of 0 dB. Bottom: simulated compression with a signal-to-
noise ratio of -20 dB (see text).
64
intensity modulator (MZM), driven by an arbitrary waveform generator programmed to
the transmission sequence T . The sequence was constructed from a 1112 bits-long
maximum peak-to-sidelobe ratio (MPSL) bipolar code or from 832 bits-long
complementary pair code, following the above procedure. The coding symbol duration
was 200 ps. The measurement SNR was controlled by the addition of ASE of variable
power from an EDFA.
The modulated waveform was amplified by a second EDFA and launched
towards a movable retro-reflector via a fiber circulator (55 dB isolation) and a
collimating lens. Reflections were partially collected by the lens, directly detected by a
photo-diode with 12 GHz bandwidth, and sampled by a digitizing oscilloscope of 6 GHz
bandwidth. The detected sequences were compressed through matched and mismatched
filtering, carried out using offline signal processing.
In a first set of experiments, the retro-reflector was placed a short distance (tens
of cm) from the lens (see Fig 3.15), and the detection SNR was varied through adjusting
the power of both the laser diode and ASE noise source. In this manner the reflected
signal remained above the thermal noise of the photo-detector, and the SNR could be
quantified by switching the ASE on and off.
First, 1112 bits-long MPSL code was used as the transmitting signal T . The
cross-correlations of incoherently compressed LADAR are shown in Fig. 3.16, alongside
the simulated correlations of compressed noise-free sequences. At a high SNR of 20 dB,
the PSLR of the experimentally obtained sequence following matched filtering reached
33 dB, in agreement with the design prediction. A mismatched filter further improved
the PSLR to 46 dB, while the peak power of the main correlation lobe was only 1 dB
lower than that obtained with a matched filter. The results come close to the simulated
48.8 dB PSLR of the mismatched MPSL code (see Fig 3.9 top row)
65
TABLE IV
LIST OF COMPONENTS FOR INCOHERENT PULSE COMPRESSION LADAR SYSTEM
COMPONENT MANUFACTURER MODEL
AWG 5GS/S TEKTRONIX AWG7051
RF AMPLIFIER MINI-CIRCUITS ZHL-42W
MACH-ZENDER
INTENSITY MODULATOR
THORLABS LN56S-FC
POLARIZATION
CONTROLLER
TUNABLE LASER NEW FOCUS TLB-6600
EDFA KEOPSYS
EDFA REDC
CIRCULATOR
PHOTODETECTOR 12GHZ NEW FOCUS 1544-A
OSCILLOSCOPE 6 GHZ AGILENT 90604A
COLLIMATING LENS
25.4MM F=75MM
THORLABS AC254-075-C-ML
RETRO-REFLECTOR
50MM
THORLABS PS973M-C
MOUNTS & HARDWARE
FOR OPTIC ALIGNMENT
THORLABS
POWER SUPPLY
RF SMA CABLES
Arbitrary
waveform
generator
MZM
Laser diode
Noise EDFA
EDFA
PC
Attenuator
Circulator
Broadband
detector
Digital
processingOscilloscope
Lens Reflector
Figure 3.14. Experimental setup for LADAR measurements using incoherent pulse
compression. MZM: Mach-Zehnder modulator. PC: polarization controller. EDFA: erbium-
doped fiber amplifier. Black solid lines denote fiber connections, blue dashed lines represent
electrical cables, and orange dash-dotted lines describe free-space propagation
66
FIBER OPTIC FC/APC
CABLES
RF M/F CONNECTORS
OPTICAL CONNECTORS
PC WITH MATLAB
Incoherent compression could still be carried out even when the measurement
SNR was drastically degraded to -20 dB (Fig. 3.11, bottom row) in a good agreement
with the simulation results (see Fig. 3.9 bottom row). The results demonstrate the
potential of the incoherent compression scheme at poor SNR conditions.
Figure 3.15. Experimental setup of incoherent pulse compression
67
0 1000 2000 3000 4000
0
-33.3
Simulation
Samples
Rel
. C
orr
. [d
B]
4000 5000 6000 7000 8000
0
-51.6
Simulation
Samples
Rel
. C
orr
. [d
B]
-300 -200 -100 0 100 200 300
0
-8.1
Time [ns]
Rel
. C
orr
. [d
B]
SNR -20 dB
-300 -200 -100 0 100 200 300
0
-6.8
Time [ns]
Rel
. C
orr
. [d
B]
SNR -20 dB
-300 -200 -100 0 100 200 300
0
-46
Time [ns]
Rel
. C
orr
. [d
B]
SNR +20 dB
-300 -200 -100 0 100 200 300
0
-33
Time [ns]
Rel
. C
orr
. [d
B]
SNR +20 dB
Figure 3. 16. Cross-correlations of an incoherently compressed, 1112 pulses-long unipolar sequence.
Both matched (left, blue) as well as mismatched (right, black) filters were used in the compression
process. Top row: simulated compression of noise-free sequences. Center row: compression of
experimentally obtained LADAR echoes, detected with a signal-to-noise ratio of +20 dB. Bottom row:
compression of experimentally obtained LADAR echoes, detected with a signal-to-noise ratio of -20 dB
(see text).
Next, 832 bits-long complementary pair code was used as the transmitting signal
T . The cross-correlations of incoherently compressed LADAR are shown in Fig. 3.17,
alongside the simulated correlations of compressed noise-free sequences. At a high SNR
of 20 dB, the PSLR of the experimentally obtained sequence following matched filtering
reached 42 dB (Fig 3.17, center). Note, that the comparable incoherent compression of
the MPSL sequence required a precise mismatched filter of 3336 coefficients [88]. Once
again, compression could still be carried out at SNR level of -20 dB. The PSLR in this
case degraded to 10 dB (Fig 3.17, bottom) in a good agreement with the simulation
results (see Fig. 3.13 bottom).
68
-1500 -500 0 500 1500-64
0
SamplesR
el. C
orr.
[dB
]
Simulation
-300 -200 -100 0 100 200 300
-42
0
Rel
.Cor
r. [d
B]
Time [ns]
SNR +30 dB
-300 -200 -100 0 100 200 300
-10
0
Rel
. Cor
r. [d
B]
Time [ns]
SNR -20 dB
Figure 3. 17. Cross-correlations of an incoherently compressed, 832 pulses-long complementary code.
Top: simulated compression of noise-free sequences. Center: compression of experimentally obtained
LADAR echoes, detected with a signal-to-noise ratio of +20 dB. Bottom row: compression of
experimentally obtained LADAR echoes, detected with a signal-to-noise ratio of -20 dB (see text).
The experimental compression is restricted by the noise and distortion of the RF
amplifiers used to match the output voltage of the waveform generator to the levels
necessary for the MZM electrical input port. The full width at half maximum of the main
correlation lobe, signifying resolution, is 200 ps as expected.
3.2 LADAR link budget
The expected range and resolution of the proposed system are calculated using
the same steps described in section 2.3. This time the system is much simpler, involving
fewer components and transitions from the optical to the electrical domain.
The spatial resolution is proportional to the duration of a single transmitted
symbol [3]. The arbitrary waveform generator used in our experiment, for example,
produces of 200 ps-long symbols. Using equation 2.11 we get an estimated resolution of
3 cm. The ranging accuracy for an SNR of 20 dB is estimated as 2 mm (see equation
2.12)
69
The overall SNR in the system is limited by source of noise in the optical parts,
such as the thermal noise of the photo-detector and the amplified spontaneous emission
of the EDFA. These noise sources overshadow the contributions of RF amplifiers and
quantization errors of the waveform generator and oscilloscope.
The optical power that is equivalent to the detector thermal noise (see section
2.3), for a bandwidth of 5 GHz, is -21.5 dBm. The EDFA amplifies the transmitted
signal power by 25 dB, up to 25tP dBm. Photo-current noise in the detection of
amplified signals stems from both the interference between signal and ASE, and the
detection of ASE power [11]. In the calculations I assume the inclusion of an optical
filter, whose bandwidth matches that of the signal, for maximum noise suppression.
Subject to this assumption, the ASE power can be reduced to the order of -25 dBm, well
below that of the signal. The electrical SNR due to the beating between signal and ASE
is also very high in this case, on the order of 50 dB. At the range limitation, the optical
power of collected laser echoes are very weak, and fall well below the noise power (as
discussed in the previous section and will be addressed in detail shortly). In this
condition, the setup is limited primarily by thermal noise.
The optical power of the collected echoes rP at the input of the photo-detector is
given by the LADAR range equation [60]:
2
2 2( )
t t tr
R t
D dA PP
R R
(3.1)
Here tD is the circular receiver aperture, and t is the target reflectance
parameter. Typical values for this parameter range from as little as 2% to as high as 25%.
In my calculations I assumed a high reflectance target of t = 25%. dA is the target
surface area, t is the laser beam angular divergence angle, R is the target surface
71
angular dispersion (see below for both angles), and R is the one-way distance to the
target. The beam divergence t is given by [60]:
1.22
t
tD
(3.2)
In the experiments the wavelength and aperture were 1550 nm and
25.4tD mm respectively. Given these parameters the angular beam divergence equals
0.07 mrad. For the case in which the illuminating beam in the target plane is smaller than
the target surface area, dA is simply the projected area of the beam at the target, as
expressed by the following equation:
2 2
4
t RdA
. (3.3)
The solid angle R over which radiation is dispersed generally takes on either the
value of π steradians for Lambertian targets, or the same value as t for mirrored
surfaces. Values between these limits are possible for surfaces that are not entirely rough
or polished, but these cases represent the extremes. I will assume a Lambertian target.
Substituting the above parameters into the LADAR range equation (3.1), the
following simple expression is obtained:
2
0.0127rP
R [mW], (3.4)
where distance is given in meters. The experimental results described above (see
subsection 3.1.3) suggest that lowest SNR in which the correlation peak could still be
observed is -20 dB. Therefore, the longest measurement distance is that for which the
incoming signal power rP is 20 dB below the noise equivalent power of the photo-
detector, previously estimated to be on the order of -21.5 dBm. The signal power drops
to that limiting level for R = 14 m. The working distance can be further increased with
71
0 6 10 20 30 40 50 60 70
0
-32.3
SNR +20 dB
Distance [m]
Rel.
Corr
. [d
B]
0 6 10 20 30 40 50 60 70
-1
-37.6
Distance [m]
Rel.
Corr
. [d
B]
SNR +20dB
Figure 3. 18. Experimental Cross-correlations of an incoherently compressed, 1112 pulses-long
unipolar sequence collected from a reflector that was placed 6 m away from the collimating lens at
an optical signal to noise ratio of 20 dB. Top: compression using matched filter. Bottom:
compression using mismatched filter.
averaging: The SNR (in dB) in which the main correlation peak remains discernible is
improved by a factor of 10*log10( )TESTN , where TESTN is the number of independent
recordings to be averaged. If, for example, the transmitted code is being generated each 2
µs and the sampling window is one second-long, than each sampling window contains
500,000 independent measurements. In these conditions, the lowest SNR in which the
main lobe remains discernible is lowered by another factor of 28.5 dB. The signal power
drops to that limiting level for R = 300 m.
Another option for increasing the measurement range is the amplification of the
collected reflections by an EDFA prior to detection. Assuming an amplifier gain of 30
dB, and weak reflected echoes in the nW range, the optical noise due to ASE-ASE
interference and due to signal-ASE interference remains below the noise equivalent
power of the detector thermal noise. Therefore, the maximal range may be extended by a
factor of G , where G is the amplifier power gain. The range could reach hundreds of
meters without averaging, and towards several km with averaging.
72
0 10 20 30 40 50 60 70
0
-31.7
SNR +18 dB
Distance [m]
Rel.
Co
rr.
[dB
]
0 10 20 30 40 50 60 70
-1
-39.2
Distance [m]
Rel.
Co
rr.
[dB
]
SNR +18 dB
Figure 3. 19. Experimental Cross-correlations of an incoherently compressed, 1112 pulses-long
unipolar sequence collected from a reflector that was placed 50 m away from the collimating lens at
an optical signal to noise ratio of 18 dB. Top: compression using matched filter. Bottom: compression
using mismatched filter.
3.3 Laser range-finder measurements
Preliminary ranging performance was illustrated by placing the retro-reflector
several distances away from the collimating lens. The 1112 bits long MPSL code was
used. The average transmitted power was 100 mW, the collimating lens aperture was
25.4 mm, the symbol duration was 200 ps and the receiver sampling interval was 50 ps.
First, the retro-reflector was placed 6 meters away from the collimating lens. The
SNR of the collected reflection was 20 dB. Both matched and mismatched filters were
used in the pulse compression. Fig 3.18 (previous page) displays the compressed
waveforms as function of absolute distance. The PSLR for the matched and mismatched
filters were 32.3 dB and 36.6 dB respectively with mismatched induced loss of 1 dB.
Next, the retro-reflector was placed 25 meters away from the collimating lens.
The SNR of the collected reflection was again 20 dB. The PSLR for matched and
mismatched filters were 31.5 dB and 34.7 dB respectively, with 1 dB induced mismatch
loss.
73
In the following experiment, the retro-reflector was placed 50 meters away from
the collimating lens, and the SNR of the collected reflection was 18 dB. Fig 3.19
displays the compressed waveforms as function of absolute distance. The PSLRs this
time were 31.7 dB and 38.2 dB respectively, with 1 dB induced mismatch loss.
Lastly the range precision of the system, defined by its ability to recover relative
changes in the distance of a single target, was evaluated by placing the reflector 50 m
away from the collimating lens, and changing its position by 2.5 cm. Fig. 3.20 displays
the compressed waveforms as function of delay for the two reflector positions. A
mismatched filter was used in the compression process. The full width of the main lobe
at 70 dB below the peak is approximately 4 cm, in agreement with the pulse duration and
sampling rate. The PSLR of both curves is above 35 dB. The two peaks are
approximately 2.4 cm apart, in agreement with the reflector position change.
49.6 49.8 50 50.2 50.4-70
-60
-50
-40
-30
-20
-10
0
Relative distance [m]
Rel
ativ
e co
rrel
atio
n [
dB
]
Figure 3. 20. Cross-correlations of incoherently compressed, 1112 pulses-long unipolar
LADAR echoes. The distances between the LADAR lens and a retro-reflector were 50 m (blue,
dashed) and 50.025 m (red, solid). The measurement SNR was 18 dB. A mismatched filter was
used in the compression.
74
3.4 Concluding remarks
In this chapter, a LADAR system based on incoherent pulse compression was
proposed and demonstrated. The system relies on simple intensity modulation and direct
detection of dense position-coded MPSL sequences and complementary code pairs. The
compression is achieved through cross-correlating the received echoes with matched or
mismatched filters stored at the receiver. The principle is of particular consequence for
photonic applications, in which coherent detection is more difficult to implement. The
high-SNR delay response exhibit very low sidelobes: a PSLR of 46 dB for mismatched
filtered, 1112 bits long MPSL code, and 42 dB for matched filtered 832 bits long
complementary pair. The spatial resolution of the experimental demonstration is
estimated as 3 cm. A change in range of 2.5 cm was accurately recovered in a high-SNR
measurement. The ranging accuracy for a high SNR is estimated as 2 mm, and was not
validated experimentally. The systems link budget suggests a maximum operating
distance of 14 m (without averaging), limited primarily by the detector noise and the
maximum transmitted power. Averaging can further extend the maximum operating
distance to hundreds of meters.
75
CHAPTER
4
Discussions and Conclusions
In this research, high resolution microwave and laser ranging was proposed and
demonstrated using two distinct approaches. In the first approach, ranging measurements
were obtained using a microwave-photonic UWB noise radar system. The UWB noise
waveform was generated using the ASE associated with SBS or an EDFA. Waveforms
of more than 1 GHz bandwidth and arbitrary radio-frequency carriers were generated,
and distributed over 10km fiber to a remote antenna unit. The collected echoes were
carried back over fiber towards the central office where they were detected, sampled and
correlated against the stored reference using digital signal processing. The system
combines the fiber-optic antenna remoting capabilities, broad bandwidth and flexibility
of microwave-photonic processing, together with the immunity against interception and
jamming of noise radars. The detection of a concealed metallic object and the resolving
of two targets with the anticipated ranging resolution had been demonstrated. Photonic
noise generation could scale towards high frequencies of electrical noise in the
millimeter waves range, provides a high degree of randomness and spectral shaping
based on a physical process, and readily integrates into a RoF system. The ranging
resolution is not affected by the length of the fiber link.
In a second approach, a high-resolution laser ranging system with strong sidelobe
suppression was demonstrated experimentally. The sidelobe suppression was obtained
through an incoherent pulse compression process that was previously proposed by Prof.
Nadav Levanon of Tel-Aviv University. The compression relies on the transmission and
76
direct detection of Manchester coded, unipolar representations of a chosen binary phase
sequence, and its matched or mismatched filtering by a stored bipolar reference on
receive. The extent of sidelobe suppression nearly replicates that of the original bipolar
sequence, even though phase is not maintained and a simple incoherent receiver is
employed. This principle is of particular consequence for photonic applications, in which
coherent detection is more difficult to implement. The principle was experimentally
implemented using MPSL sequences, with a PSLR of 46 dB achieved through
mismatched filtering, and using complementary code pairs, for which matched filters
yielded a PSLR of 42 dB.
I had shown through simulation and experiments that a range to the target could
be measured in the presence of additive noise, at SNRs as low as -20 dB. The noise
tolerance can be leveraged towards a longer measurement range, lower launch power and
energy consumption, reduced apertures and improved operation at unfavorable
atmospheric conditions.
It is important to note that the spatial resolution is determined by the bandwidth
of the waveform used, rather than by its temporal shape. However, use of clever
sequences helps reduce the sidelobes of the correlation function, for comparable
bandwidth. The employment of 'pure' noise waveforms provides better immunity to
interception and jamming.
The results obtained in both approaches were limited by several factors. The
UWB microwave photonic radar system resolution was restricted to the order of 10-15
cm by the bandwidths of the noise waveforms. Their bandwidths could not be increased
beyond 2 GHz due to the limitations of the RF antennas. The measurement range was
limited to the order of 15 m by the maximal current modulation of the laser diode used
77
for transmission towards the central office on the one hand, and by the thermal noise
floor of the detector and relative intensity noise of the light source on the other hand.
In the second approach, the system resolution was proportional to the duration of
a single sub-pulse, and reached 3 cm. This limitation stems from equipment constraints
and is not fundamental. The theoretical sidelobe suppression scales with the length of the
sequences used. Even though some sequences (such as complementary pairs) are
scalable to arbitrary lengths through simple procedure, the sidelobe suppression is
limited in practice by additive noise from signal generators, RF amplifiers, EDFAs and
photo-detectors. In addition, the temporal shapes of the transmitted and received sub-
pulses deviate from that of an ideal square, whereas those of the digitally stored
reference pulses are undistorted. These differences in shape could further restrict the
extent of sidelobe suppression. In future work the profile of the reference pulses could be
corrected. The range of system, without averaging, is restricted to the order of 15 m by
the thermal noise of the photo detector. The range can be increased, in principle, towards
hundreds of meters with optical amplification of reflected echoes.
An interesting future work, with respect to microwave-photonic UWB noise
radars, would be the construction of a multi-purpose radar system. Such a system would
employ several UWB noise source bands in order to detect and sort different targets with
various resonant frequency responses, sizes and shapes. The system would require
broader bandwidth of antennas and amplifiers, and a very stable local oscillator. It will
be also worthwhile to measure the antennas impulse responses using literature
procedures [108], and correct for them in the correlation process in order to further
reduce the sidelobes. These measurements might require an anechoic chamber. Further
work in the area of incoherent LADAR systems would extend their ranging
78
measurements to longer distances, and employ them in 3D imaging. Acquisitions of 3D
images would require a fast and precise angular scanning capability in two dimensions.
79
Appendix A
A.1 MPSL 82 code [106]
T = [1 1 1 1 -1 -1 1 -1 1 1 -1
-1 1 -1 -1 1 -1 1 1 1 -1 1
-1 -1 1 1 1 -1 -1 -1 -1 -1 -1
-1 1 1 -1 -1 1 1 1 -1 -1 -1
1 1 1 -1 1 1 -1 1 1 1 -1
1 1 1 -1 1 1 -1 -1 1 -1 1
-1 1 1 -1 1 -1 -1 1 -1 1 -1
1 1 1 1 1 ];
A.2 MPSL 1112 code[107]
This code was found by Ron Ferguson and has never been published, thus only
its autocorrelation will be given below.
0 500 1000 1500 2000
24
1112
Autocorrelation of MPSL 1112
bit
Corr
elat
ion
81
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תקציר
תקציר
חשיבות רבה ליישומים הינן בעלות של סנטימטרים בודדים מדידות טווח ברזולוציה גבוהה
בתחום וגלים מחד ( מ"במערכות מכ)גלים -ומיקרוגלי רדיו . צבאייםאזרחיים כמו גם ליישומים
ישנם יתרונות וחסרונות בשימוש . למדידות טווח יוםכ משמשים מאידך (במערכות לייזר)אופטי ה
הפרעות למ הינו פחות רגיש לדיוקי הרכבה ו"מכבעוד .בכל אחד מהתחומים הספקטראלים הללו
יותר ובכך לאפשר גדוליםרוחבי סרט בעליאותות לשאת גלים אופטיים יכולים, אטמוספריות
ניתנים ליישום , מיסודם חסינים להפרעות אלקטרומגנטיות, מדידות ברזולוציות גבוהות יותר
.ויכולים להיות מופצים בקלות באמצעות סיבים אופטיים, יחסית קומפקטייםבמכלולים
קצרים פולסיםשידור והה באמצעות ברזולוציה גב טווח ניתן להשיג מדידות, י השיטותתבש
יצירתם ושידורם של פולסים קצרים ובעלי הספק גבוה קשים למימוש , ואולם .עוצמה-ורבי
קצרים גורם לירידה בפולסיםהשימוש , בנוסף. במערכות מעשיות ועלולים להוות סיכון בטיחותי
. ק הרעש מתקלקלהספהאות לבין הספק יחס הכתוצאה מכך ו, בהספק הכולל של האות המשודר
עיבוד באמצעות . טכניקות דחיסהאותות ארוכים או סדרות בצירוף ניתן להשתמש ב, לחילופין
את כל האנרגיה לדחוס עשוי וסדרות ממושכיםשל אותות , מתואמתאו מסננת , יהצקורל-אוטו
טכניקה זו ,משום כך. עם אונות צד נמוכות, עוצמה-תוך אונה מרכזית צרה ורבתלשלהם
, צרבשימוש בפולס ק יםמתקבלהגבוהה ורעש הרקע הנמוך ההרזולוציה לשחזר אתשרת מאפ
כמה סדרי נמוך בהספק הרגעי של סדרות ארוכות יכול להיות ה: בתוספת יתרונות משמעותיים
ר המאפשר יצירה פשוטה ובטיחותית של אותות אלו במערכות אמיתיות ובנוסף מקשה דב, גודל
.עוייןעל האזנה מצד גורם
מטרת עבודה זו הינה מדידות טווח ברזולוציה גבוהה באמצעות קידוד פוטוני של אותות
. בעבודה זו יודגמו ,אשר משלבות דחיסה של אותות ארוכים, שונות מערכות שתי. מיקרוגל ולייזר
פליטה במערכת זו משתמשת . פוטוני-רדיורחב סרט מ רעש "המערכת הראשונה הינה מכ
שלאחר , אקראירחב סרט 'פיזיקלי'במגברים אופטיים על מנת לייצר רעש תית המתרחשנהספונט
הפצה נוחה של האותות למרחק שרירותי מחוללי רעש פוטוניים מאפשרים. רדיו ימכן מומר לתדר
. בעיצוב צפיפות ההספק הספקטרלית של האותוגמישות , רוחבי סרט גדולים, על גבי סיבים
האזנה ושיבוש על ידי משופרת כנגד מאפשר חסינות דועותבסדרות יהשימוש באות רעש במקום
. גורמים עויינים
תקציר
ברילואן או לפיזור הנלוויתית נעל אפקט ההגברה הספונטההדגמה הניסיונית מסתמכת
המערכת כוללת מרכזייה אשר בה אותות הרעש נוצרים . למגברי סיבים מאולחי ארביום
המרכזייה ויחידת הקצה . ם האנטנות ומגברי הרדיואשר בה נמצאי ניידת ידת קצהיחו, ומעובדים
בניסוי זה הושגה רזולוציה מרחבית של . מ"ק 01רות ביניהם באמצעות סיב אופטי באורך ובמח
הושגבנוסף . 'מ 01 -מוערכת בכ, בכפוף למגבלות הציוד, טווח המדידה של מערכת זו. מ"ס 01
.מוסתרתמתכתית גילוי של מטרה
המשתמשת בסדרות מקודדות , דגם הינה מערכת מד טווח לייזרהמערכת השנייה שתו
ברוב . ועיבוד מתאים על מנת לאפשר מדידות טווח ברזולוציה גבוהה עם אונות צד נמוכות
קידוד עוצמה נותן ביצועים כאשר, שימוש בקודי פאזה מחייבותשיטות דחיסה יעילות , המקרים
. קוהרנטיים מסובכים במקלטיםשימוש הדידת הפאזה במערכות אופטיות מצריכמ .פחותים
' לאחרונה על ידי פרופ אשר פותחה, משתמש בשיטת דחיסה לא קוהרנטית חדשנית אני, לחילופין
-חד קוטביים מומרים לקודי עוצמה-די פאזה דוקו, בשיטה זו. נדב לבנון מאוניברסיטת תל אביב
. ים לאיפנון מקור הלייזרמשמשולאחר מכן ,מיקום לקידודבאמצעות אלגוריתם ,קוטביים
ולאחר מכן עוברים קורלציה עם , נדגמים על ידי גלאי עוצמה פשוטממטרות האותות המוחזרים
אף על פי שתהליך השידור והקליטה .המקלט באופן ספרתי בזכרוןקוטבי אשר שמור -קוד דו
כמעט לחלוטין את אונות הצד הנמוכותתוצאת הקורלציה משחזרת , אינם קוהרנטיים
. המקוריקוטבי -הדוקוד הפאזה בבשימוש המתקבלות
-ייצוג חד. הודגם אף הוא באופן נסיוני דחיסה לא קוהרנטיתעל המבוססלייזר המד טווח
זוג( 2 -ו ,אונות צד מינימליות קוד המתוכנן לקבלת( 0: נבחן נסיוניתקודים סוגי קוטבי של שני
רזולוציות המדידה שהושגה היתה . ד של השניאשר מבטלים את אונות הצד אח, קודים משלימים
טווח המדידה . dB 20- -עד ל, אות לרעש נמוכים ייחסגם ב נמדד בהצלחההמרחק למטרה . מ"ס 3
פני זמנים -עלצעות מיצוע מויכול להגיע למאות מטרים בא, תלוי ברפלקטיביות של המטרה
של עיקרון הדחיסה הלא לאה מתוצאות אלו מציגות לראשונה הדגמה ניסיונית . קצרים יחסית
. תקוהרנטי
יצירת אותות , רחב סרט מ רעש"הקדמה מתומצתת למכ. העבודה מאורגנת באופן הבא
מ "מכ I. בפרק ניתנתומערכות מד טווח לייזר , דחיסת אותות ארוכים, פוטוניים-באמצעים רדיו
מבוססות על שטווח לייזר וקדש למערכות מד מ IIIפרק . IIבפרק נדוןפוטוני -רדיורחב סרט רעש
תקציר
בפרק מופיעיםעבודה ההמשך גבי סיכום ודיון ל לבסוף. דחיסה לא קוהרנטית של אותות ארוכים
IV.
אבי ר"ד של הנחייתו תחת הבוצע המחקר עבודת
אילן-בר באוניברסיטת להנדסה מהפקולטה צדוק
של אופטי קידוד באמצעות טווח מדידות
ולייזר גלמיקרו אותות
קרביץ דניאל
מוסמך תואר קבלת לשם מהדרישות כחלק מוגשת זו עבודה
אילן-בר אוניברסיטת של להנדסה בפקולטה
ג"תשע גן רמת