FILTERS

consists of a set of parallel-grounded resonators loaded bythe variable capacitors madeof tuning screws. Combline fil-

ter can be realized on different transmis-sion lines. Suspended stripline provideshigh quality factor (Q) of approximately500, stability over a wide temperaturerange, and high impedance range.6 In thehigh-Q suspended stripline (Fig.1b), the par-allel strips are printed on both sides of asubstrate in a symmetrical configuration.Plated through holes (vias) provide elec-trical connection between top and bot-tom conductors. When dual-center con-ductors are located symmetrically withrespect to each other, they are excited inphase, causing most of the electromag-netic field to propagate in the air dielec-tric. Therefore, substrate dielectric lossesand dielectric constant variations havenegligible effects on the attenuation andphase velocity of the transmission media.

Suspended stripline resonators areplaced between two parallel ground planes.Adjacent suspended stripline resonators arecoupled by the fringing fields. The typicallength of the combline filter resonators isbetween 0/16 and 0/6, where 0 is thecenter guide wavelength at the resonator.For this resonator length, magnetic cou-

Assemble A TunableL-Band Preselector

Microstrip and suspended-striplinetransmission techniques can be com-bined to create a compact electricallytunable preselector filter for L-bandapplications.

lectrically tunable preselectors are key elements in com-

munications, avionics, and radar receivers. Narrowband RF

and microwave preselectors prevent large off-channel sig-

nals from overdriving a receiver front end. Microstrip

combline and interdigital tunable filters have been

described by several authors.1-5 By combining a suspended-

stripline bandpass filter (BPF), microstrip low-noise

amplifier (LNA), and input/output match-ing networks, an electrically tunable L-band preselector can be assembled with typ-ically 3-dB bandwidth from 18 to 24 MHz.

The tunable BPF is split to provide par-tial selectivity with minimum insertionloss prior to amplification for improvedinput noise figure. The first two-polefilter before LNA prevents undesirablesignals from overdriving the LNA. Thesecond three-pole filter after the LNA pro-vides selectivity against receiver imageand spurious frequencies. The three-pole BPF placed after the LNA has anegligible effect on the overall prese-lector input noise figure.

A conventional combline filter (Fig.1)

eLEO G. MALORATSKY Principal Engineer Rockwell Collins, P.O. Box 1080,Melbourne, FL 32902; (321) 953-1729,e-mail:lgmalora@rockwellcollins.com.

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Table 1: Comparing combline filter configurationsBANDPASS FILTER PARAMETERS

Insertion loss at 1030 MHz (dB)Bandwidth at 3-dB level (MHz)Return loss (dB)Second-harmonicattenuation at 2060 MHz (dB)Third harmonicattenuation at 3090 MHz (dB)

TWO POLES

1.241.020.3

THREE POLES

2.129.023.5

FIVE POLES

3.425.020.794.8

94.6

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pling predominates.7 The minimum prac-tical length of the resonators is limited bythe Q. Practical Q values are dependentupon the ground-plane spacing (base),the frequency of operation, the finish ofthe ground surface, the plating material ofthe printed-circuit board (PCB), and thesuspended-stripline structure.

Short resonators yield a compact struc-ture with excellent stopband performance.With a resonator length, l, of l =0/8, thesecond passband will appear at betterthan four times the fundamental operat-ing frequency, while at l = 0/16, the sec-ond passband will occur at more thaneight times the fundamental frequency.Combline filter trade-offs for various res-onator lengths are described in ref. 4.

The bandwidth of a combline filter isa function of the ground-plane spacing, b,to wavelength ratio, b/0 and the spacing,S, between resonators. The bandwidthincreases with higher S and b/0. For

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INPUT0 1 2 3

n – 1 n n + 1

OUTPUTL

C0 C1 C2 C3 Cn – 1 Cn + 1Cn

b

S01

S12

S23 Sn – 1, n

Sn + 1, n

(a)

(b)

via

W0

W1W2

W3Wn – 1 Wn + 1

Wn

1. These two representations show the (a) plane view and (b) cross-sectional viewof suspended-stripline resonators.

MWsept03_82.ps 9/2/03 8:28 PM Page 82

combline filters, bandwidths of 2 to 50 per-cent can be achieved.

The spacing (b) between two groundplanes (cover and housing) defines res-onator impedances and lengths and max-imum power and Q. Resonator impedancesrange from 70 to 140 Ω at frequencies (f)of f 1 GHz. Large bases (ground planes)lead to higher power handling and increasedQ, but also to an increase in resonatorlength and housing height.

Spacing S (between resonators) is pro-portional to base b. For a distance betweenprinted resonators and ground planes ofb/2 = 200 mils, with a substrate thickness(h) of 10 mils, the base should be equal tob + h = 410 mils. For these conditions, theresonator impedance is approximately100 Ω (an admittance of 0.01 Ω–1.

The loading capacitance for eachcombline resonator is1:

where:

C YTOT i= ( )cot / ( )Θ0 0 1ω

YI = the admittance of the ith resonatorwhen the (i – 1)th and (i + 1)th resonatorsare shorted and 0 = (2l)/0 = the elec-trical length at the center frequency.

For 1-GHz suspended-stripline res-onators with the above dimensions, the guidewavelength is equal to 0 = 28.8 cm.According to Eq. 1, the total capacitance,CTOT = 2.75 pF for a resonator length of0 = 30 deg. (l = /12).

Usually, capacitors are also used astunable elements to compensate for pro-duction tolerances. The use of capacitorsis especially critical for narrow bandwidth.At low frequencies, capacitor Q’s are high-er than the Q’s of resonators. At microwavefrequencies and for higher capacitancevalues, capacitor Q’s can be lower than res-onator Q’s, dominating performance whenfilter losses are calculated.

Table 1 compares experimental resultsfor tunable combline filters with /12-long suspended-stripline resonators and withair-dielectric trimmer capacitors (Giga-

Trim products from Johanson Corp.,Boonton, NJ) with capacitance range of0.4 to 2.5 pF.

Figure 2a shows an electrically tun-able BPF consisting of suspended-striplineresonators grounded at one end, high-QGaAs varactor diodes, and lumped-ele-ment loading capacitors between theground plane and the other end of each res-onator. The tunable BPF was realized withreverse-biased varactor diodes D1, D2,D3, D4, and D5 which were used as tun-ing elements to adjust the combline pass-band over the full frequency range. Tun-ing is performed by altering the bias of thevaractor diodes. A two-pole BPF fabri-cated with trimmers provided a 3-dB BWof 41 MHz with 1.2 dB insertion loss. Asimilar two-pole BPF filter with varactorsyielded a 63.8-MHz BW with 1.85-dBinsertion loss.

Preselector selectivity depends on fil-ter Q. The total Q of the tunable BPF istaken as the combination of the resonator,

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loading capacitor, and varactordiode Q’s. For a low-loss L-bandcombline filter, the varactordiode Q is an very importantparameter. The Q of the bestvaractor diodes is lower thanthat of the suspended stripline res-onators and loading capacitors,and is a dominating factor whencalculating filter losses.

To increase diode Q, GaAsvaractors can be used. For exam-ple, model MA46617 GaAsabrupt varactor diodes providea Q of 210 at 1 GHz and serveas effective variable capacitors.For the MA46617 diodes, the totalcapacitance ratio, CT0/CT45 =4.4 to 6.9, where CT0 and CT45

are the varactor capacitances at0 and 45 V, respectively. Varactortuning voltages are controlledby a microprocessor to precise-ly tune the filter response.

The total loaded capacitanceof combline tunable filter is givenby:

where:Cj = the varactor diode junc-

tion capacitance andC = the parallel lumped-ele-

ment capacitance (C1 = C2 = C3 = C4 = C5

= C). The center frequency of the tunablepreselector is determined by lengths ofresonators and the total loaded capaci-tance, CTOT.

For the 1-GHz tunable filter with res-onator length of l = L/12, the total capac-itance (from Eq. 1) is CTOT = 2.75 pF. Fora 25-percent tuning range, varactor diodeswith a junction capacitance of Cj = 1.3 pFat –4 VDC can be used. In this case, thelumped-element capacitance (accordingto Eq. 2) should be C = 2.75 – 1.3 = 1.45pF. To allow for biasing, capacitors C6,C7, C8, C9, and C10 are added (Fig. 2a).

These capacitors provide an RF short forthe varactors and an open circuit for thebias currents.

The tunable preselector is based on acombination of suspended-stripline andmicrostrip transmission lines. Each ofthese two transmission-line formats offerscertain strengths.6 Microstrip, for exam-ple, supports a high level of integration andexcellent heat dissipation, since a goodground connection (with minimal reactance)is needed for each device. The tunablepreselector includes a microstrip LNAbetween the first and second suspended-stripline BPF. The LNA is based on a

monolithic-microwave-integrated-circuit(MMIC) MGA-85563 device from Agi-lent Technologies.8 The RF layout of theLNA is shown in Fig. 2b.

The LNA operates from a +3-VDCbias supply and draws nominal current of15 mA. For the lowest noise-figure per-formance, the amplifier’s input port shouldbe matched with the output of the two-polecombline filter and the input of the three-pole combline filter. To match the inputof the LNA to the 50-Ω two-pole BPFoutput, inductor L1 is placed in series withthe input of the LNA. DC blocking capac-itor C12 is placed at the output of theMMIC LNA to isolate the amplifier fromthe three-pole BPF. Inductor L2 and capac-itor C13 isolate RF from the DC supply.

A circuit that includes the MMIC LNAand its input and output matching networkswas realized on a microstrip line usingthe same BPF dielectric substrate. The

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Table 2: A summary of L-band preselector performance FREQUENCY(MHZ)96210901213

3-dB BANDWIDTH(MHz)

18.120.824.4

INPUT RETURNLOSS (dB)23.720.626.3

INPUT NOISEFIGURE (dB)4.63.84.0

GAIN(dB)4.56.07.0

FourthPoleTune

FifthPoleTune

R6

R2R1

Microstrip LNA

Suspended striplineThree-pole BPF

RF output

D1 D2

FirstPoleTune

SecondPoleTune

ThirdPoleTune(a)

RF input

Suspended striplineTwo-pole BPF

BiasC13

C12

C1 C2

C11

C6

C7

L2

L1

MGA85563

R4R3

D3 D4

C3 C4

D5

C5

C8

C9R5

C10

OUTPUT

Microstrip line Housing(c)

Dielectricsubstrate

Suspended striplineCover

INPUT2-pole BPF LNA 3-pole BPF

2. The experimental L-band electrically tun-able preselector isshown in (a) schematicform, (b) PCB top andbottom layouts, and (c)side view.

(b)

C C CTOT j= + ( )2

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combine-filter input/output network pro-vides a transition from the microstrip-lineLNA to the suspended-stripline BPF aswell as matching between the LNA input/out-put ports and the suspended stripline res-onators. For low-frequency application(less than 1 GHz), a straight connectionbetween microstrip and suspended striplinecan be used. This connection includes thestep between center conductors of twolines and ground plane step (Fig. 2c) to pro-vide a 50-Ω impedance for both lines. Forhigher-frequency applications, the specialtransition between the two lines can be used.6

Matching sections at the filter inputand output match the resonators with the50-Ω microstrip lines. Each matching net-work (Fig. 3)consists of two high-impedancesuspended-stripline series printed induc-tors (L1 and L2) and low-impedance sus-pended stripline (shunt capacitance Cm).

This matching net-work is equivalentto the T-section ofthe LPF (Fig. 3b).

The main param-eters of the matchingnetwork can be deter-mined using thewave-matrix meth-od. This matchingcircuit is a two-portnetwork, which isequivalently repre-sented in the formof four cascade-connected elementarytwo-port networks (Fig. 3b). The result-ing transfer matrix of the equivalent two-port network is equal to the product of thetransfer matrices of the above componenttwo-port networks written down in the orderof energy flow.

Multiplying the transfer matrices forthe center frequency yields:

For the case of perfect matchingof input port 1, element S11 of thescattering matrix, which has the phys-ical meaning of reflection coefficient,must be equal to zero, S11 = T21/T11

= 0; therefore, T21 = 0. If Z1 = Z3 = Z,it is possible to obtain from Eq. 3:

From Eq. 4, it is possible to findthe value of the capacitance match-ing element and its dimensions.Using these techniques, the preselector

was fabricated on 10-mil-thick dielectricsubstrate TLE-95™ from Taconic Advanced

Dielectric (Germantown, NY) with dielec-tric constant of 2.95. The PCB was sus-pended over a silver-plated aluminummachined housing. The depth of the hous-ing and cover was 0.200 in. (0.508 cm).The total dimensions of the preselectorare 12.7 5.08 1.27 cm.

Figure 4 shows the frequency responseof the preselector for various varactortuning voltages. As the filter tunes from962 to 1213 MHz, its 3-dB bandwidth variesfrom 18.1 to 24.4 MHz. Table 2 summa-rizes the preselector’s performance.

REFERENCES1. G.L. Matthaei, “Comb-Line Band-Pass Filters of Narrow orModerate Bandwidth,” Microwave Journal, Vol. 6, August1963, pp. 82-91.

2. R.M. Kurzrok, “Design of Combline Band Pass Filters,” IEEETheory and Techniques, Vol. MTT-14, July 1966, pp. 351-353.

3. I.C. Hunter and J.D. Rhodes, “Electrically TunableMicrowave Bandpass Filters,” IEEE Trans. on Microwave The-ory and Techniques, Vol. 30, September 1982, pp. 1354-1360.

4. R.M. Kurzrok, “Tunable Combline Filter using 60 DegreeResonators,” Applied Microwave & Wireless, Vol. 12, Novem-ber 2000, pp. 98-100.

5. A.R. Brown and G.M. Rebeiz, “A Varactor Tuned RF Filter,”submitted review as a short paper to the IEEE Transactions onMicrowave Theory & Techniques, October 29, 1999.

6. L.G. Maloratsky, “Reviewing the Basic of SuspendedStriplines,” Microwave Journal, Vol. 45, No. 10, October 2002,pp. 82-98.

7. L.G. Maloratsky, “Design Regular- And Irregular-Print Cou-pled Lines,” Microwaves & RF, Vol. 39, No. 9, September2000, pp. 97-106.

8. Hewlett Packard, Technical Data, “3-volts, Low NoiseAmplifier for 0.8-6 GHz Application,” 1998.

MRF

MICROWAVES & RF 88 SEPTEMBER 2003

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Cover

PCBb/2 Housing

Suspended substrate linecross-section

(a)

L1

I1

L2

W1

BPFIn/Out

(Connectionwith 50-microstrip

line)

Cm

b/2

BPF Out/In(Connectionwith 100-

BPF resonator)

L11

1

2

2

L2

Cm

TII TIII TIV

(b)

50 50 50 50 50 50 50 50 100 100

50 100

Z1 Z1Z3

y2 y2

TTOT TI

3.(a) This printed-circuit configurationshows the combline BPF’s matching circuitry.

3.(b) This equivalent circuit represents the impedance-matching cir-cuitry used in the combline BPF.

4. Preselector response was tested from 800 to 1400 MHz.

Am

plit

ud

e—d

B

0

Frequency—MHz

10

20

30

40

50

60

70

800 1100 140080

T Z Z Y

Z Y Z Y21 1 3 2

1 2 3 2

2

2 1 3

= + −+ − + ( )

Y Z Z2 2 1 2 4= +( ) −( )/ ( )

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