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www. masoncom. com

Mason Communica tions Ltd 2001

WCDMA Radio Planning Course4 Network Design4.1 Link Budgets

Mason Communications Training: WCDMA Radio Planning Course

Module 4: Network Design

Section 4.1: Link Budgets


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4.1.2 Mason Communica tions Ltd 2001

Where are We Now?Introduction

UMTSOverview

AccessTechnologies

WCDMAIntroduction

ModelArchitecture

UMTSStandards

Mobile RadioChannel

NarrowbandChannel

WidebandChannel

Local MeanSignal

Path Loss

Diversity

DesignElements

Basic Radio

Principles

Antennas andFeeders

Interference

MatchedFilters and

Rake Receivers

WCDMAPhysical Layer

NetworkDesign

OperatorsDesign Guides

The PlanningProcess

Polygons

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

CourseOverview

ConventionalOptimisation

3GOptimisation

Radio ResourceManagement

Optimisation

CourseWash Up

LinkBudgets

Where are We Now?

The Course Map shows which section we are now on.


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What is in This Section?

Introduction

Classical 2G Link Budgets

UMTS Link Budget How It Differs

Summary

UMTS Uplink Link Budget

UMTS Downlink Link Budget

UMTS Link Budget Analysis

NetworkDesign


The PlanningProcess

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

Polygons

LinkBudgets

What is in This Section?

The Radio Planning section will concentrate upon the current industry and academic approaches in

analysing the UMTS or WCDMA Link Budget. The aim of this section is to demonstrate the issues,

and limitations in the UMTS Link Budget, when compared to conventional Link Budget analysis forGSM or TDMA. As a result of these limitations the need for more sophisticated approaches is

introduced. The use of Static and Dynamic simulation techniques for Link Budget and Detailed Radio

Planning, using RF Planning Tools (such as Aircoms Asset), are presented in the next Section, The

Radio Planning Process.


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Why is this Section Important to You?

Very Important that the UMTS Link Budget is understood

The UMTS Link Budget will be used in dimensioning anetwork or an area of a network

The Link Budget is the precursor to using a Network PlanningTool

The Network Planning Tool uses the Link Budget and isextremely dynamic in UMTS

Why is this Section Important to You?


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How Will You Learn?

Discussion

Worked Examples Exercises

Demonstrations

The differences betweenGSM and UMTS LinkBudgets

That the confidence ofthe UMTS Link Budget islimited when we considerall the UMTS parameternetwork interactions

The need for Simulationsto predict UMTSnetwork performance

How Will You Learn?


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Where Are We Now?

Introduction



Summary

NetworkDesign


The PlanningProcess

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

Polygons

LinkBudgets





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Approaches to Radio Planning

Generally there have been two key approaches to Radio Planning (i.eGSM Planning, or more precisely TDMA Planning):

Network Dimensioning using Link Budget Analysis

Network Planning using Radio Planning Tools

UMTS Presents New Challenges for the Radio Planner and RadioPlanning Process

Conventional Approaches are limited when considering UMTS RadioPlanning

There are classically two approaches to Radio Planning.

Network Dimensioning using Spreadsheet based Link Budget analysis has been used to provide a

basic understanding of Cell Range for GSM and TDMA networks. Business Plans for country wide

networks often use the Spreadsheet Link Budget to estimate the quantity of Base Stations required toprovide Coverage and Capacity, and the sensitivities behind the Link Budget and the overall site count

estimate.

Network Planning using Radio Planning Tools offer a detailed estimate of specific coverage and

service levels over specific areas for GSM and TDMA networks. Planning tools use Terrain and

Buildings information of the specific area and aim to estimate coverage levels at a resolution at the

order of the resolution of the Terrain data (e.g. 25m resolution). This allows collections of Base

Station locations, heights, and antenna configurations to be optimised or engineered to best meet the

coverage and capacity expectations for the area.

GSM or TDMA based Radio Planning is relatively straightforward. The Link Budget and RadioCoverage Levels are predictable, since all of the parameters which make up the GSM or TDMA

Radio Link are not dependant upon one another. In UMTS or WCDMA, there are many parameters of

the Radio Link which are inter-dependent upon one another. The classic WCDMA parameter

dependency is that of Cell Breathing, where the Cell Loading, or number of users per cell increases

the Interference Levels within that cell, which in turn reduce the potential range. These

interdependencies present new challenges to the Radio Planning process and approaches to UMTS

Planning.


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The Link Budget is used in dimensioning exercises, and tounderstand basic coverage principles.

Link Budget equations are normally developed using Spreadsheetformulas.

There are many different ways of presenting the same Link Budget.

A Link Budget should generally have 3 sections:

Tx Parameters Element

Rx Parameters Element

Propagation Parameters Element

The Link Budget

The Link Budget is one of the fundamental tools a Radio Planner should be familiar with. Radio Link

budgets are used in dimensioning networks, and more importantly can give valuable insight to how a

radio link might behave, in terms of Coverage Range, and Capacity.

There are many ways of presenting a Link Budget. Vendors, Operators, Standards bodies,consultancies often present the link budget in different ways. There is no universally accepted

approach to presenting a link budget. However, a Link Budget should be made up of three elements;

Transmission, Receiver, and Propagation Parameters. We shall use a certain approach, which largely

reflects the Link Budgets shown in [1].


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Link Budgets track Power levels along a transmission pathfrom Transmitter Output Port to Receiver Input Port

Two Distinct Links in a Duplex Communications Channel:

Mobile (transmit) to Base Station (receive) - uplink

Base Station (transmit) to Mobile (receive) - downlink

Tx Rx

Rx Tx

UPLINK

DOWNLINK

The Link Budget

Link Budgets strictly track Power Levels along a Transmission Path or Link from a Transmitters

Output Port to the Receiver Input Port. They can also be re-arranged in many ways, but normally Link

Budgets are re-arranged in a manner which provides information, such as the Maximum Acceptable

Propagation Loss, given all other Parameters. This Propagation Loss can then be used to estimate

Range, and hence estimate Coverage.

There are always two Link Budgets in a Duplex Communications Channel. A Simplex Channel (such

as TV Broadcast has one Link Budget). We often refer to these links as Uplink and Downlink. In the

US (and technically correct), theses are the Reverse and Forward Channels.


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Duplex Communication Link having two links are ideallybalanced in terms of their link budgets

Uplink: Low Power Mobile Transmission to HighSensitivity Base Receiver

Downlink: High Power Base Station to Low SensitivityMobile Receiver

Tx Rx

Rx Tx

UPLINK

DOWNLINK

The Link Budget

Having a Power Balanced Link is ideal in terms of Range, or Coverage. However, the Uplink and

Downlink power budgets are not reciprocal, as the Mobile Stations are limited in terms of the Power

they can Transmit, and the Mobile Stations must be made for a Mass Market and hence the Receiver

Sensitivity they can afford, can not be highly engineered. This presents a potential link imbalance,

which is counted by having a more highly engineered Base Station receiver, and hence Receiver

Sensitivity, and a higher power can be transmitted from a Base Station. Generally speaking the Uplinkoften presents the limiting case.

In GSM or TDMA planning, there are a number of parameters at the RF engineers disposal to

improve the overall Link Budgets, or to attempt to balance the Link Budgets. These parameters are

discussed later in this section, but include the use of Base Station Receive Antenna Diversity, and the

use of LNAs at the Base Station Receiver.


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Where Are We Now?

Introduction



Summary

NetworkDesign


The PlanningProcess

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

Polygons

LinkBudgets





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The 2G Link Budget

BTS

+30dBm

-104dBm

Tx Parameters Rx ParametersEnvironment Parameters

Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Path

Loss

LogNormal

FadeMargin

Antenna

Gain

Diversity

Gain

Feeder

Losses

Rx

Power

Tx

Power

RxPower

Rx

Sensitivity

RxSensitivity

The three parametric elements which make up the Link Budget are shown; Transmission,

Environment, and Receiver Parameters.

The Link Budget shows the Power Received at the Base Station Receiver (Uplink) given the all the

Link Parameters. If the Log Normal Fade Margin is set to zero, then the Rx Power calculated is theAverage Rx Power received for all locations (at the same distance, and in the same Environment of

course). In other words the actual Rx power can vary above (50%) and below (50%) this Average Rx

Power, due to location variability. The shaded/blurred elements represent the Locations Variability in

a certain environment.

The Log Normal Fade Margin is added such that the Rx Power represents the Rx Power Received for

a certain percentage of locations. This might be 90% of locations. That is Rx Power represents the

minimum Rx Power expected for 90% of locations variability.

The Link Budget aims to ensure that the Rx Power (for a certain %locations, given the LN Fade

Margin), does not fall below the Minimum Rx Sensitivity, and the slide indicates the MaximumAllowable Propagation Path Loss acceptable to maintain the Radio Link.


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The 2G Link Budget

BTS

+30dBm

-90dBm


Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Path

Loss

LogNormal

FadeMargin

Antenna

Gain

Diversity

Gain

Feeder

Losses

Rx

Power

Tx

Power

Rx

Power

RxSensitivity

Rx

Sensitivity

This slide simply shows the same Power Link Budget as before, but with a smaller Path Loss. The Rx

Power in this slide represents the Rx Power exceeded for 90% of locations (assuming that a Log

Normal Fade Margin for 90% Locations is added). In this example this Rx Power (90% Locations) is

well above the minimum Rx Sensitivity Threshold.

What would happen in reality is that the Mobile Tx Power would reduce in a GSM system such that

the Rx Power (90% locations) is closer to the Rx Sensitivity Threshold, thereby conserving Battery

Power, reducing unnecessary interference (to other cells), and still maintaining the Link Budget.

If one was to imagine an animation of the above slide, the Path Loss would vary, as the distance

varied, between MS and BS. Also the instantaneous Rx power level due to location variability would

dance around (obeying a Log Normal Probability distribution the shaded red parts) which would

result in Tx Power Variations, such that the Rx Power was => Rx Sensitivity Threshold.


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The 2G Link Budget

The Rx Power can be calculated given Path Loss and the Link BudgetParameters

The Max Path Loss can be calculated given the Minimum RxSensitivity and Link Parameters

TxPower

AntennaGain

BodyLosses

PenetrationLosses

PathLoss

Log

NormalFade

Margin

AntennaGain

DiversityGain

FeederLosses+ - - - - + + -

RxPower =

Path

Loss

Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Log

Normal

FadeMargin

Antenna

Gain

Diversity

Gain

Feeder

Losses+- - +-=Rx

Power - + +-

MaxPath

Loss

Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

LogNormal

FadeMargin

Antenna

Gain

Diversity

Gain

Feeder

Losses+- - +-=Rx

Sensitivity- + +-

The Link Budget can be re-arranged such that the Maximum Allowable Path Loss can be calculated.

The first equation is that shown before in the previous examples.

The second equation is the first re-arranged to represent Path Loss.

The Third equation is the second equation, but using Rx Sensitivity = Rx Power (for 90% locations) todetermine the Maximum Path Loss.

This common re-arrangement is used to find out Maximum Path Loss, and hence Cell Range.

The Rx Sensitivity for TDMA and GSM Systems is a function of:

Noise Power Bandwidth

Receiver Noise Figure

SNR Margin above Noise+Noise Figure such that a Minimum acceptable decoded BER for

a Service (such as Coded Speech) is maintained

The SNR Margin varies for different Services, Speeds, and Environments (Typical Urban, Hilly, Bad

Urban, etc.).

Environments give rise to different Multipath Models, and hence Fading Dynamics.

The Bit/Frame Interleaving mechanisms used in GSM assist the Error Correction algorithms (by

randomising errors) in the Multipath Channel.

The Worst Case speed, Environment, and acceptable BER is usually taken resulting in the Worst Case

SNR. GSM Specifications stipulate that the Base Station Receiver should have a minimum Rx

Sensitivity of 104dBm.


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2G Uplink

BTS

Tx Power Rx Sensitivity

(TU50 or

RA130)

Mobile Tx Power = 30dBm (1 Watt) variable due to manufacturer variances

Mobile Antenna Gain = 0 dBi variable due to polarisation and near-field effects

Body Loss = 4 dB variable due to orientation, body position wrt BTSPenetration Loss = 5 dB dependent upon location (i.e building or direction of car)

LN Fade Margin = 5 dB based upon % location probability of coverage over cell area

Path Loss = ? dB The maximum path loss can be calculated to achieve % loc prob.

Base Station Antenna Gain = 18 dBi variable due to (azimuthal) antenna pattern

Base Station Diversity Gain = 4 dB variable due to extent of multipath de-correlation

Base Station Feeder Loss = 2 dB dependent upon quality and size of feeder, and length

Base Station Splitter Loss = 2 dB dependent upon how the signal is shared between multiple TRXs

Int. Degradation margin = 3 dB with interference limited design

Rx Sensitivity = -104 dBm Function of:

(residual BER = 0.2%) noise floor in 200kHz (-120 dBm)

receiver noise figure (8 dB)

Eb/No in fading environment (8 dB)

Path Loss = 30+0+(-4)+(-5)+(-5)+18+4+(-2)+(-2)+(-3)-(-104) = 135dB

Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Path

Loss

LogNormal

Fade

Margin

Antenna

Gain

Diversity

Gain

Feeder

Losses

In this Uplink Link Budget Example, real figures are used for a GSM/TDMA scenario.

The Slide also illustrates the variability of each Link Budget parameter.

Generally speaking parameters such as:

Tx Power

Rx Sensitivity

Antenna Gains

Feeder Losses

are very deterministic, and vary very little we dont worry too much about these.

Parameters such as:

Penetration Loss

Body Loss

Fade Margin

Diversity Gain

can be quite variable quantities, and are usually quoted as statistical limits. These can vary depending

upon specific MS-BS Orientation, diversity schemes used, etc. Penetration Loss can vary enormously

for buildings, ranging from a few dB to 25dB. This variability is seen between buildings, and the

locations variability within the buildings. Ideally the penetration margin should represent the

penetration losses experienced for a certain % of locations within all buildings, e.g. 95% rather thanan average penetration loss.


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2G Downlink

BTS

Tx power Rx Sensitivity(TU50 or

RA130)

Base Station Tx Power = 40dBm (10 Watts) variable due to manufacturer variances

Base Station Combiner Loss = 4 dB dependent upon how the TRXs are shared with common antennas

Base Station Feeder Loss = 2 dB dependent upon quality and size of feeder, and lengthBase Station Antenna Gain = 18 dBi variable due to (azimuthal) antenna pattern

Path Loss = ? dB The maximum path loss can be calculated to achieve % loc prob.

LN Fade Margin = 5 dB based upon % location probability of coverage over cell area

Penetration Loss = 5 dB dependent upon location (i.e building or direction of car)

Body Loss = 4 dB variable due to orientation, body position wrt BTS

Mobile Antenna Gain = 0 dBi variable due to polarisation and near-field effects

Int. degredation margin = 3 dB with interference limited design

Rx Sensitivity = -102 dBm Function of:(residual BER = 0.2%) noise floor in 200kHz (-120 dBm)

receiver noise figure (10 dB)

Eb/No in fading environment (8 dB)

Path Loss = 40+(-6)+18+(-5)+(-5)+(-4)+0+(-3)-(-102) = 137dB

Tx

Power

Antenna

Gain

Feeder/

CombinerLosses

Penetration

Losses

Path

Loss

LogNormal

Fade

Margin

Antenna

Gain

Body

Losses

The example shown above applies to the previous example, but for the Downlink channel.

The same variability and issues apply as discussed in the previous slide.

There is no receive diversity on the downlink for GSM/TDMA.

The Downlink can suffer a slightly greater Path Loss than the uplink channel in this example.


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2G Link Budget Calculation

Given Link Parameters, a maximum path loss can be calculated, in ourUplink example this is:

Loss = 30+0+(-4)+(-5)+(-5)+18+4+(-2)+(-2)+(-3)-(-104) = 135dB

Given environment and appropriate path loss model a maximumdistance, hence cell size can be calculated to a % locations probability

For example, assume a countryside environment with in-carpenetration and maximum path loss of 135dB, then using a stochasticmodel, such as Hata, we can work out the cell radius

Maximum Path Loss or Propagation Loss can be translated to a Maximum Cell Radius or Range when

applied to a suitable Empirical loss equation, such as a Hata Model.

A more exact range can be computed for all locations within a cells area if we were to use a Radio

Planning Tool. The Radio Planning tool computes the specific path loss for every point to the BaseStation. All points-BS links which fall below a the Maximum acceptable Path Loss means that

Coverage Service is available.

If a very accurate terrain/buildings model is used with the planning tool then the Locations Variability

can be resolved. In this case we would not need to consider the Log-Normal Fade Margin in the

Maximum Path Loss.


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2G Link Budget Calculation

100

105

110

115

120

125

130

135

140

145

150

0 2 4 6 8 10 12 14 16

Hb

= 50 m

Hm

= 1.5 m

f = 900MHz

Path Distance (km)

Pa

thLos

s(dB)

HATA Model Rural (Quasi-Open)Environment

For a 135dB Path Loss, a Cell radius of ~ 9km can be

achieved with 90% Area Locations Probability, given

system parameters in example for in-car penetration in a

rural environment

Applying the previous GSM Maximum Path Loss to a Hata Empirical Equation, for Quasi-Open Rural

Environment, a Base Station height of 50m, and a MS height of 1.5m at 900MHz, we could achieve

about 9km. This means that service is available to 9km which maintains that the Rx Power is above

the minimum Rx Sensitivity Threshold for 90% Locations at the cell edge, or at 9km.


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2G Link Budget Sensitivity

We shall examine the effect of an additional 2dB margin ona link budget

With previous example a path loss of 135dB corresponded to

a cell radius of 9km Using same path loss model and system parameters, except

that the uplink now has an extra 2dB of link margin,therefore making a maximum path loss of 137dB. What cellradius can be achieved now?

If we were to gain an extra 2dB on the Uplink Maximum Path Loss, such that it now balances with the

Downlink Maximum Path Loss, we could examine the potential new range, using Hata. This

improvement might be achieved through improved engineering of the Uplink, by the inclusion of

LNAs and/or low loss feeder at the Base Station.


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We can now achieve a cell radius of ~ 10.5km

increase in cell area from 9km2 to 10.5km2

increase in cell area from 254km2 to 346km2

36% Area Increase !

100

105

110

115

120

125

130

135

140

145

150

0 2 4 6 8 10 12 14 16

Hb

= 50 m

Hm

= 1.5 m

f = 900MHz

Path Distance (km)

Pa

thLos

s(dB)

HATA Model Rural (Quasi-Open)

Environment2dB

A 2dB improvement on the Uplink can give a new range of 10.5km, which means an improvement of

36% in service area.

This is quite surprising, and demonstrates the sensitivity of the Link Budget with respect to estimation

of quantities of sites. This is only 2dB. When we consider that parameters such as Penetration Margin,Body Losses, and diversity Gain have variability over many dBs we can begin to understand the

difficulty, or the limitations in the confidence a Spreadsheet Link Budget approach to site count

estimation gives.


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When considering network build an extra couple of dBs in alink margin can have significant effect on site numbers andultimately cost of network!

Rural (Quasi-Open) Suburban Urban Urban Indoor

Cell Area Coverage increase due to extra 2dB Link margin

The key message for ensuring that a network gets the most out of the Link Budget, is to engineer the

Links such that extra Path Loss can be tolerated. This can result in fewer sites for coverage.


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URBAN

0dB + 2dB

SUBURBAN

0dB + 2dB

RURAL

0dB + 2dB

Cell

Radius (km2)

Effective Cell

Area (km2)

Area to be

Covered (km2)

No. of CellSites

Relative Cost

per Cell Site (k)

Network Build

Costs(M)

0.84 0.95 1.35 1.53 6.74 7.62

1.84 2.34 4.73 6.12 118.2 153

1,500 1,500 6,000 6,000 100,000 100,000

815 641 1268 980 846 653

300 324 300 324 200 208

245 208 380 318 169 136

168M lower Network Build Costs for +2dB link margin (1997)

This slide shows a simple demonstration of how the Link Budget can improve Network Costs. The

example applies to GSM, and is purely for illustrative purposes only. The same argument would apply

to UMTS, in that extra coverage can be achieved through Link Budget improvement. Alternatively, in

UMTS this Link Budget improvement can be traded for extra capacity. This will be demonstrated

later in this section. Also see p167 in Reference [1].


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2G Link Budget Optimisation

The need to balance uplink and downlink budgets

Uplink budget usually the limiting case

Uplink can benefit from additional budget over downlinkthrough:

Minimise Feeder Losses

Base Station Receive Diversity Gain

Higher Base Station Receive Sensitivity

Use of Head Amplification and LNAs

We have shown that Link Budget improvement can be used to improve the Network Build

requirement. Generally speaking, the Uplink is the limiting case for GSM/TDMA systems. In UMTS

this is also generally the case, but the Link Imbalance can vary much more due to cell loading,

Asymmetric loading, and the level of WCDMA Interference from other cells and Mobiles connected

to other cells. We shall show this later in this section.

The Uplink can be improved through various methods. Four methods are discussed in the following

slides, and relate to Uplink Link Budget improvement through:

The use of low loss feeder at the Base Station

The use of Base Station Receive Diversity Reception

The use of better Base Station sensitivities (better Noise Figures)

The use of Low Noise Amplifiers (LNAs) at the Base Station Receiver


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2G Link Budget Optimisation - Feeder Losses

Feeder Losses can be minimised through use of:

Short Feeder Runs

Minimum number of connectors

Use of high quality and thick feeder cable

FEEDER LOSS AT 960 MHz

7.51 dB/100m 4.2 dB/100m 3.1 dB/100m

11 2 "dia"dia 7 8"dia 1 4

The slide shows examples of Feeder loss at 960MHz, I.e for GSM. The feeder losses at

1900/2100MHz would be even greater, and more reason to choose a High Quality, Thick Feeder

Cable. Such cable could only be used on Tower or Macro type base stations, and probably not on

Micro Base Stations. Micro Base Stations can incorporate the Transceiver(s) directly behind an

Antenna, rather than Antenna-Feeder-Transceiver arrangement. In this case, of course the Feeder run

losses are negligible.

Connectors inherently introduce losses, through two mechanisms

Contact losses the fact that conductivity across two surfaces is not as good as

continuous cable

Reflection losses impedance mismatches result in power reflections and hence

reduced power transfer


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2G Link Budget Optimisation Diversity Gain

Diversity Combining on Uplink path can give extra link margin

Relies on sufficiently separated receive antennas at Base Stationsuch that the transmission channels from mobile to each antenna arede-correlated in terms of fast fading

Fast Fading Signal at

Antenna 1

Fast Fading Signal atAntenna 2

Diversity CombinedSignal

D

Base Station Receive Diversity is often called Micro Diversity, since the difference in signals is due to

microscopic fading, or fast fading phase differences, which occur over a number of wavelengths.

Diversity attempts to increase the median signal strength at the receiver, by the reducing the

probability of deep fades. Section x.x.x discusses the theory, practice, and benefits offered by

Diversity in much more detail.

Diversity Gain is not really an Active Gain product, but is the difference between:

Power Level at Receiver for a Certain BER with Diversity

Power level at Receiver for same BER without Diversity

The BER is usually the minimum acceptable BER for a certain Multipath Channel (e.g. TU50). The

diversity gain represents the equivalent dB improvement had there not been any Diversity.

Diversity Gain will therefore vary depending upon BER, Multipath Channel, and Separation ofreceive antennas.


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2G Link Budget Optimisation BS Specification

GSM Specification was put togetherin 1980s

Radio Receiver Technology can now

offer Base Station ReceiverSensitivities below -110dBm

Nortel claim -117dBm

Equipment maturity and supplierissues must be considered

GSM SpecificationRx Sensitivity = -104dBm

Rx Sensitivity = -108dBm

1991

2001

GSM Vendors have recognised the importance of increasing the Link Budget. As a result there has

been huge investment in attempting to better the fundamental Receiver Sensitivity. This improvement

is by improving the Noise Figure of the receivers.

When GSM Specifications were first released the Base Station Receiver Sensitivity (for a certainBER, and Multipath channel) was 104dBm. Vendors have steadily improved on this, at almost 1dB a

year, and now below 110dBm is common.

There is a fundamental limit to this improvement. Assuming a Noise Figure of 0dB, and a Gaussian

Channel, then the fundamental Receiver Sensitivity might be

Rx Sensitivity = 10Log10(kTB) + Noise Figure (dB) + Eb/No (BER=10-3)

Rx Sensitivity = 10Log10(1.38x10-23 x 290 x 200x106) + 0dB + 5dB

= -121dBm + 0dB + 5dB

= -116dBm

Where Eb/No = 5dB represents the decoded Bit Error Rate for a Gaussian Channel (estimate).

When Nortel claim 117dBm, they are claiming the fundamental figure, in Gaussian Channel, and

possibly lower Noise Temperature.

This value is academic since it assumes no additional Noise introduced by the Receiver, and it


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2G Link Budget Optimisation MHA/LNAs

Receiver amplification provided at antenna

Receiver front-end stage is situated directly at antenna

Effect is though whole receiver at antenna therefore removingany feeder losses

Improvement is in overall receive system Noise Figure

Feeder

AntennaHead

BS Rx Feeder

AntennaHead +

LNA

BS Rx

If we remove the Feeder losses by situating the Base Station receiver directly at the Antenna, we

improve the link as discussed earlier, e.g. the micro base station receiver. However, it can be

impractical to do this for Macro or Tower Base Stations, as we would have to mount all the equipment

at say 20m on a head frame, This presents some challenges for maintenance, access, and reliability.

A Receiver is usually made up of various Amplification stages, as discussed in section x.x.x. If we

move some of the amplification stages, or at least the first stage to the antenna, we can achieve a

similar improvement in the link by essentially removing the feeder losses, whilst maintaining a

practical deployment solution.

Depending upon the Active Gain(s) of the first amplification stages, we afford improvement in the

link budget. This is shown through the Cascade Equation.


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NF1 NF2 NFn

G1 G2 G3

NF NFNF

G

NF

G G

NF

G G GSYS

n

n= +

+

+ +

1

2

1

3

1 2 1 2 1

1 1 1...

...

Cascade Equation

NFSYS

- Overall Noise Figure of Receiving System

NFx

- Noise Figure for each Receiver stage/elementG

x- Gain of each Receiver stage/element

NF and G input as Linear quantities, not logarithmic (dBs)

The slide illustrates a number of amplification stages as might be expected in a Receiver. The Overall

Noise Figure (in linear terms) for a Receiver (NFSYS)is shown in the slide, and is a composite sum of

Gains and individual Noise Figures of the stages as shown. We can represent the Feeder in this

diagram as having less than unity Gain, and a Noise Figure equal to its attenuation. So a 3dB Feeder

Loss would have a Gain of 0.5, and a NF of 2.

The next two slides present an example of calculating the overall Noise Figure assuming no head

amplification and with head amplification, to demonstrate the improvement in the overall Noise

Figure and hence the link budget.


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NF1=3dBNF1 = 2

NF2=6dBNF2=4

G1=-3dB

G1=0.5

NF NFNF

G

NF

G GSYS = +

+

1

2

1

3

1 2

1 1

Attenuator

Amplifier

NF3=6dBNF3=4

G2=10dB

G2=10

G3=10dB

G3=10

NFS YS = +

+

24 1

0 5

4 1

0510. . .

NFSYS = + + = =2 6 0 6 8 6. . 9 . 3 4 d B

The example shows a conventional Receiver Deployment, with Antenna-Feeder-Receiver

arrangement. In this example the Receiver has an effective Noise Figure of 9.34dB.


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NF2=3dBNF2 = 2

NF1=6dBNF1=4

G2=-3dB

G2=0.5

NF NFNF

G

NF

G GSYS = +

+

1

2

1

3

1 2

1 1

Attenuator

Amplifier

NF3=6dBNF3=4

G1=10dB

G1=10

G3=10dB

G3=10

NFS YS = +

+

42 1

10

4 1

0510. .

NFSYS = + + = =4 0 1 0 6 4 7. . . 6 . 7 2 d B

2.6dB increase in Link

Margin over feederthen amplifier

cascade

The example shows a Receiver Deployment with Head Amplification, with Antenna-LNA-Feeder-rest

of Receiver arrangement. In this example the Receiver has an effective Noise Figure of 6.72dB, a

2.6dB increase in Link Budget.

If we moved all the receiver, i.e. all of the amplification stages to the antenna, an improvement of 3dB(=Feeder Losses) would be seen. The example above demonstrates also that the first amplification

stage has the largest influence on overall Noise Figure. Because of this, the first amplifier stage is

often the best engineered, in terms of reducing its Noise Figure, and the use of Low Noise Amplifiers

(LNAs) are typically used.


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Where Are We Now?

Introduction



Summary

NetworkDesign


The PlanningProcess

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

Polygons

LinkBudgets





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UMTS Radio Planning - Differences

In TDMA, the Noise (or Interference)Level with which a Mobile or Base Stationmust operate remains essentially Constant.

In TDMA, there is no real concept of avariable Processing Gain.

In WCDMA, the Interference Level variesdue to Loading of the Cell, which in turnaffects Maximum Path Loss, and hencecoverage.

In WCDMA, there are many Services, whichhave different Datarates, which give rise todifferent Processing Gains.

There are key differences between WCDMA and TDMA/FDMA link budgets

When we move from our GSM or TDMA Link Budget to UMTS we need to consider a number of

new, and variable Link Budget Parameters. The slide shows the key differences in GSM and UMTS

Link Budget Parameters.

In GSM/TDMA systems the design is Interference Limited, that is Cell Frequencies, and Time Slotsare re-used, such that a predictable, and reasonably steady state of Interference is present. The idea of

introducing an Interference Margin in the GSM Link budget (shown earlier) represents the fact that

the wanted signal in the Link Budget competes against Noise and Interference. In WCDMA or UMTS

however, the Interference levels vary much more widely since the same downlink Carrier Frequency

is re-used in every cell and the same uplink frequency by every mobile. As a result the Interference

levels in an Uplink Link Budget, for example will vary with the number of active mobiles, both in the

home cell (Intracell interference) and the number of active mobiles and their positions in other cells

(Intercell interference).

In WCDMA/UMTS many different services can be supported which demand/consume different Data

Rates, Latency, and Throughput. These result in different DS-CDMA Processing Gains, as well as

adding different Interference Levels to the Cell. These different datarates give rise to different link

budgets, and hence range. In GSM/TDMA, there is essentially a limited subset of services, i.e. EFR

Voice, Data (14.4kbps), and some GPRS data rates where error correction coding is traded for extra

datarate capacity at the expense of Receiver Sensitivity.


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UMTS Radio Planning - Differences

In TDMA, the Service is normally alwaysvoice, which dictates a certain Eb/No, andhence Rx Sensitivity, based upon a worstcase Environment, and minimum acceptable

BER is a Constant (e.g. 104dBm for GSM).

In TDMA, there is Hard Handover whichhas no influence on the Radio Link Budget.

In TDMA, there is simple slow PowerControl, which preserves Battery Life, andensures that the MS-BS AverageInterference power is kept in order.

Rx Sensitivity is a function of Eb/No, whichis dependent upon actual service type,datarate, speed, Multipath environment,diversity schemes and RAKE Receiver

Algorithms

In WCDMA, soft handover is possible, whichgives rise to Macro-Diversity Gains againstLog Normal Fading

WCDMA requires that all MS Powersreceived at the BS are equal. To achieve thisWCDMA employs fast power control tocounter Rayleigh fading. A Fast PowerControl Margin (or Headroom is needed forMobiles at the Cell Edge)

There are key differences between WCDMA and TDMA/FDMA link budgets

In GSM/TDMA a certain Bit Energy to Noise Power Density (Eb/No) is required for a certain BER

for say EFR Voice in a certain Multipath channel, which leads to a certain reference Receiver

Sensitivity. Eb/No varies with Service, and Multipath Channel. This might be a decoded EFR voice

stream at 10-3 BER, in Bad Urban 50km/h. Normally in GSM 104dBm is used as the reference

sensitivity. In UMTS Eb/No also varies as above, but is also variable with datarate, data service (10 -6

might be needed for data), specific RAKE Receiver Scheme (no. of RAKE Fingers), and effect of FastFading Power Control, which also varies with speed. This leads to a much wider spectrum of Eb/No

values relating to different environments, and services.

In GSM/TDMA there is only Hard Handover which has no influence on the Link Budget. In

UMTS/WCDMA soft handover is possible, which can afford a Macro Diversity Gain against Log

Normal Fading, over the non-soft handover case. Macro diversity reduces the influence of fading

(good for reducing stress on Error Correction/Interleaving), and allows a higher median received

signal.

Fast Fading Power Control is available in UMTS for both Uplink and Downlink. This allows the

Eb/No to be effectively reduced in a Multipath environment. However, at the cell edge the Mobile is

Power limited, and will not be able to fully negate deep fades in the channel. As a result the Eb/No at

the cell edge deteriorates. In order for the Link Budget to be consistent a Fast Fading Margin (or

Power Control Headroom) is added to represent this limitation.


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Where Are We Now?

Introduction



Summary

NetworkDesign


The PlanningProcess

Site Placement

AntennaPlacement

FrequencyPlanning

ForwardCapacity

Planning

Polygons

LinkBudgets





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BTS


Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Path

Loss

LogNormal

FadeMargin

Antenna

Gain

Diversity

Gain

Feeder

Losses

Rx

Power


TxPower

AntennaGain

BodyLosses

PenetrationLosses

PathLoss

Log

NormalFade

Margin

AntennaGain

DiversityGain

FeederLosses

Processing

Gain

InterCell

Int.

IntraCell

Int.

Eb/No

Target

Rx

Sensitivity

RxSensitivity

RxPower

BTS

LogNormal

FadeMargin

Soft

HandoverGain

Fast FadeMargin

If we refer back to our depiction of the Link Budget, the additional UMTS Parameters can be thought

of as influencing the Rx Sensitivity as shown.

The actual Rx Power remains the same. This can not change of course.

1. The variable Intercell and Intracell Interference quantities add to the Noise Power and limit the RxSensitivity.

2. The Processing Gain (Service Dependant) influences the equivalent Rx Sensitivity.

3. The Eb/No (Dependant upon Service, Datarate, Speed, and Multipath Channel) will influence the

equivalent Rx Sensitivity.

4. The Soft Handover Gain will reduce the Log Normal Fade Margin needed, or can be thought of in

the Link Budget as Log-Normal Fade Margin (without Soft Handover) + Soft Handover Gain. Since

the Link Budget is often used to find the Maximum Range, this must be considered (assuming a

continuum of cells).

5. The Fast Fading Margin represents the limit in Power Control for Mobiles at the Cell Edge. It

represents the deterioration in Eb/No due to not being able to adequately follow the fast fading

because of Power Limiting. Since the Link Budget is often used to find the Maximum Range, this

must be considered. The Fast Fading Margin can be considered as reducing the effective Rx

Sensitivity, or included as part of the overall Fade Margin (i.e. with Log Normal Fade Margin, and

Soft Handover Gain).

The Link Budget becomes dynamic, changing every time:

A Mobile user moves within the cell (the Interference to other cells will change)

A Mobile user in another cell moves (the Interference to the home cell will change)

A Mobile user becomes admitted/handed-off/removed to/from the home cell

A Mobile user becomes admitted/handed-off/removed to/from other cells

A Mobile user changes datarate (say for VBR Service Type)


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+21dBm

-110dBm

Tx

Power

Rx

Power

Rx

Sensitivity


Tx

Power

Antenna

Gain

Body

Losses

Penetration

Losses

Path

Loss

LogNormal

Fade

Margin

Antenna

Gain

Diversity

Gain

Feeder

Losses

ProcessingGain

InterCellInt.

IntraCellInt.

Eb/NoTarget

Rx

Sensitivity

Rx

PowerBTS

Log

NormalFade

Margin

SoftHandover

Gain

Fast FadeMargin

This slide simply shows the same Power Link Budget as before, but with a smaller Path Loss. The Rx

Power in this slide represents the Rx Power exceeded for 90% of locations (assuming that a Log

Normal Fade Margin for 90% Locations is added). In this example this Rx Power (90% Locations) is

well above the minimum Rx Sensitivity Threshold.

What would happen in reality is that the Mobile Tx Power would reduce in a GSM system such that

the Rx Power (90% locations) is closer to the Rx Sensitivity Threshold, thereby conserving Battery

Power, reducing unnecessary interference (to other cells), and still maintaining the Link Budget.

If one was to imagine an animation of the above slide, the Path Loss would vary, as the distance

varied, between MS and BS. Also the instantaneous Rx power level due to location variability would

dance around (obeying a Log Normal Probability distribution the shaded red parts) which would

result in Tx Power Variations, such that the Rx Power was => Rx Sensitivity Threshold.


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MaxPathLoss

TxPower

AntennaGain

BodyLosses

PenetrationLosses

LogNormalFadeMargin

AntennaGain

DiversityGain

FeederLosses+- - +-=

RxSensitivity- + +-

ProcessingGain

InterCellInt.

IntraCellInt.

Eb/NoTarget

ThermalNoisePower

NoiseFigure+ + + - +

LogNormalFade

Margin

SoftHandover

Gain-

Eb/No

Target

Fast

FadeMargin-

Similar to the GSM/TDMA Link Budget, the UMTS Link Budget can be re-arranging such that the

Maximum Path Loss can be calculated. We have to pay particular attention to the variable UMTS

parameters if we are to estimate Maximum Path Loss.

The key variables are:

Intracell Interference

Intercell Interference

Processing Gain

Eb/No

Fast Fading Margin.

We shall briefly look at each of these UMTS specific Link Budget variables.


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UMTS Uplink Link Budget InterCell Interference

MaxPathLoss

TxPower

AntennaGain

BodyLosses

PenetrationLosses

LogNormalFadeMargin

AntennaGain

DiversityGain


RxSensitivity- + +-

ProcessingGain

InterCellInt.

IntraCellInt.

Eb/NoTarget

ThermalNoisePower


LogNormalFadeMargin

SoftHandover

Gain-

Eb/NoTarget

FastFadeMargin

-

Pj.PLjIntercell Interference =j = mobiles in other cells, PL = Path Loss, P = Power

The Intercell Interference experienced is the Sum of the received powers (at the Cell Base Station)

from all Mobiles in all other cells. This is dependant upon many factors, which include:

Position (and hence distance of Mobile to Cell) and Powers of other Mobiles in other cells

Quantity and service rates of other Mobiles in other cellsCell Antenna Downtilt

Base Station Cell Sectorisation

Macro or Micro cell (if micro then there may not be any Intercell Interference)

It will be shown that the Intercell Interference has a significant effect on the usable available Capacity

in the cell, dictated by what is called the Pole Capacity. It will be shown for example that a Micro

Cell with one Transceiver can have twice as much capacity as a Macro Cell with one Transceiver. A

Micro cell has no, or little Intercell Interference as they are normally deployed in isolation, or within

confined spaces, allowing isolation from adjacent cell Interference. The Macro Cell will normally be

in a sea of Interference from its neighbouring cells.

If we use downtilt, cell sectorisation, or careful site positioning we can minimise the Intercell

Interference experienced in the Macro Cell case. For example we would not want the situation where

high traffic demand is along the boundary of two cells (e.g. Football Ground). In this case there would

be many mobiles on high power communicating with say Cell1, this would also present a very high

Intercell Interference to Cell2 since the mobiles on Cell1 are at there nearest point to Cell2.

This prompts the question of should we use 4-sector sites or 3-sector sites?. A 4-sector site will

offer more capacity per Base Station, and possibly more other-cell isolation, but most existing sites

are geared up for 3-sector deployments and head frames.Discussion Point.

We shall look at Intercell Interference later and demonstrate that we experience at the Home Cell an

effective Noise Rise per Interfering Subscriber in other cells. This Noise Rise influences the

Intercell Interference, but not the Intracell Interference when we consider a design.


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UMTS Uplink Link Budget InterCell Interference

MaxPathLoss

TxPower

AntennaGain

BodyLosses

PenetrationLosses

LogNormalFadeMargin

AntennaGain

DiversityGain


RxSensitivity- + +-

ProcessingGain

InterCellInt.

IntraCellInt.

Eb/NoTarget

ThermalNoisePower


LogNormalFadeMargin

SoftHandover

Gain-

Eb/NoTarget

FastFadeMargin

-

Pj.PLjIntracell Interference =j = mobiles in own cell, PL = Path Loss, P = Power

The Intracell Interference experienced is the the Sum of the received powers at the Base Station cell

from all mobiles within the cell. As the number of mobiles increases, or the capacity loading on the

cell increases, the Intracell Interference increases. From this statement the rate of increase would

appeargradual(first order) in nature, but as more mobiles are added each mobile has to increase its

power to overcome the increased noise rise at the Base Station cell, which in turns adds more

Interference. This produces asecond orderrate of increase, such that a theoretical infinite Interferenceis reached this is termed the Pole Capacity.

It is normal to impose a hard limit on the number of mobiles, or more precisely a hard limit on the cell

capacity, to avoid Intracell Interference rising above a certain level. This allows the range and

capacity of the cell to become more deterministic. A Cell Load of 50% means 50% of Pole Capacity,

results in an Intracell Interference of 3dB. A Cell Load of 75%, results in an Intracell Interference of

6dB. This Interference reduces Link Budget margin, and Path Loss, and hence potential range.


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UMTS Uplink Link Budget Processing Gain

A

B

A

B

Mobile 1

Mobile 2

Rx 1

Rx 2

In the UMTS Link Budget Processing Gain represents the effective improvement in power from a

wanted Signal carrying information (which has been produced by multiplying the information with a

Pseudo-Noise Scrambling code running at 3.84Mcps) to the resulting signal power of the signal

produced through decoding or decorrelation (i.e. multiplying again by the same Scrambling

sequence).

The decoding or correlation process produces a narrowband Baseband signal at the datarate of the

DPDCH Channel, from the Wideband original signal. What is happening is a process of trading

Bandwidth (Wide in the original signal, to low in the decoded signal) for Power (Low in the original

signal, High in the decoded Signal). There exists a Power-Bandwidth Conservation (rather like

conservation of Momentum in Physics), and the Processing Gain is always equal to {Chip

Rate/Information Rate}, where the Chip Rate > Information or Data Rate.

Processing Gain will vary depending upon Information Bandwidth (Service Datarate), For UMTS the

following Processing Gains are available:

Strictly speaking the WCDMA Processing Gain is equal to (Chip Rate/Channel datarate) and not

(Chip Rate/User Information Rate), since it is the DPDCH Physical Channel which receives WCDMAspreading. In fact we use the (Chip Rate/User Information Rate) to loosely define overall Processing

Gain since there is effective gain from Channel Coding, and Interleaving. Different services may

have different coding, interleaving, etc. and therefore their Processing Gains may be different for the

Service Information Rate Chip Rate Linear Log

8kbps (Voice) 3.84Mcps =3840/8 480 26.8dB12.2kbps (Voice) 3.84Mcps =3840/12.2 314 25.0dB

64kbps (LCD Data) 3.84Mcps =3840/64 60 17.8dB144kbps (LCD Data) 3.84Mcps =3840/144 26.7 14.2dB

384kbps (LCD Data) 3.84Mcps =3840/384 10 10.0dB

2Mbps(LCD Data) 3.84Mcps =3840/2000 1.92 2.8dB

Processing Gain in Rx


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UMTS Uplink Link Budget Eb/No

W = 3.84MHz

Eb

No

R x Eb

R x No

R bps 1 Hz

Linear Scaled Graphs

Frequency Domain

Noise Power = PSD x Bandwidth = 10-13 Watts

Wanted Signal Power = PSD x Bandwidth = 10 -14 Watts

Despread

Signal

Power

=

10 -14 W

Despread

Noise

Power

=

1.7.10-15 W

PowerSpectraldensity(W/Hz)=Energy(J)

RF SNR = 0.1 = -10dBDespread SNR =

RF SNR x W/R =

0.1 x 3.84/0.64 =

6 = 7.8dB

Eb/No = 7.8dB

Energy

(in 1Hz)

A Bit of Theory First - Eb/No is one of those terms which can be confusing! In UMTS we

use Ec/No and Eb/No. They are related to one another as Eb/No = Ec/No x Spreading

Factor.

The Energy in a User Information Bit (Eb) comes from the summing or Integration of theEnergies in every chip (Ec) during a bit duration through the de-spreading process in the

receiver. If we use the same Scrambling code in the receivers de-spreading process as used

in the spreading process in transmission we achieve voltage Integration on the received

signal. Noise power on the other hand is de-correlated, and in fact averages, when we sum

the chips. The voltage of the wanted signal is integrated and the noise component averages

which produces integration of Energy. Theprobability that a bit will be received in Error is

a function of the Energy in the Bit (Joules) and the Average Noise Energy (or Noise Power

Spectral Density in W/Hz = Joules). We have aprobability since actual Noise Power (even

after integration) can vary around an average value, which follows Gaussian statistics in

general. The greater the ratio of Bit Energy to Noise Energy (Eb/No) the lowerprobability that the Bit will be received in Error.

The bits in this case are the User Information Bits. In UMTS these bits undergo channel

coding with Error correction coding schemes, and as a result the Eb/No for the channel

(coded) bits can be lower than the Eb/No for the User Information bits at the expense of

reduced datarate, since channel coding adds redundancy. Generally speaking for low User

Bit error probabilities, channel coding can offer a better User Bit error probability for the

same user datarate as the uncoded channel case.

Therefore if we have a slow bit rate we have many chips per bit and hence can achieve ahigher Eb than for a higher user datarate. Hence we can state that Eb = Ec x Spreading

Factor. Alternatively we can achieve the same Eb for different user bitrates by varying Ec.


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Eb/No is a parameter to definethe Energy per User InformationBit divided by the Noise PowerSpectral Density.

There are Waterfall curves to

characterise the trade-offbetween Eb/No against Bit ErrorRate (BER) for differentModulation Schemes.

SNR against BER (a more tangiblequantity) can be derived fromSNR = Eb/No x Bit Rate/NoiseBandwidth.

Eb/No is a notional quantity itcan not be directly measured.

Eb/No is theoretically

independent of datarate

Probability of Bit Error (or BER) for QPSK Modulation and

Coherent Detection at Receiver

1.00E-17

1.00E-16

1.00E-15

1.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-031.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

Eb/No is defined as the (Energy per User Information Bit) divided by (Noise Power Spectral Density),

required to yield a specified bit error probability. Different user services require different BER or Bit

Error Probabilities. Voice may require 10-3 BER, MPEG2 Video 10-4, and FTP Data Transfer 10-6.

Ec/No is the same for a single Chip.

In order to send more information (bits) without increasing the Bandwidth, Baseband informationstreams are split into groups of say 2,3, or 4 bits and these groups are sent as different modulation

states over the radio channel. Each group of bits is a Symbol. Different modulationstates are different

Amplitude and/or Phases of the RF Signal, which can be differentiated at the Receiver. QPSK can

represent 4 changes in phase, 64QAM can represent 64 changes in Phase and Amplitude. The rate of

change of this Modulation state, represents the Symbol Rate, which dictates the RF Bandwidth

Occupancy. E.g. an RF Signal changing Phase and/or Amplitude at 3.84Million times per second will

occupy about 3.84MHz. In UMTS we dont modulate User Information bits in this way but modulate

Chips. A number of chips of course represents a User Bit depending upon the Spreading Factor used.

If we consider a QPSK modulation with state changes (or Symbol Rate) at 1MSymbols/sec. As each

Modulation State can represent two bits (or one complex bit), we effectively have a channel bitratethroughput of 2Mbps. Why dont we send more chips for every Modulation State, and get a higher

user bit rate throughput? Well we could, we would need say 8-PSK modulation to increase our

information throughput twofold. We now have less differentiatingspace between phase states, and

there is an increased risk in decoding the wrong phase state at the receiver (I.e. increase in BER). In

order to combat this we need to increase the S/N ratio. This is the fundamental trade-off of

Information Rate against S/N (related through Shannons Theorem). We would need a whole family

of graphs to represent Bit Error Rate against SNR for all bitrates. To avoid this we use the notional

parameter of Eb/No, where we assign a notional Energy for each bit, although bits are physically

transferred in blocks of 2 as a complex bit.


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The Eb/No Waterfall curvesshown previously assumes aGaussian Radio Channel. Thatis Perturbations of the Signal(due to Noise or Fading) follow

Gaussian Statistics. We can generate these

Waterfall Curves fromSimulation (Eye Diagrams, andConstellation Diagrams).

Diagram shows theConstellation Diagram forQPSK Signal with SNR=11dB(Eb/No = 8dB). This gives us aBER of about 0.0002 (1 BitError in 5000).

Demonstration

Signal

Voltage

Noise

Voltage

Example of non-WCDMA QPSK Modulated Waveform Constellation.

SNR = 11dBEb/No = 8dB

2 bits/symbol

This is not a UMTS/WCDMA Waveform

Eb/No are therefore strictly independent of Datarate, and hence there are Waterfall Curves to

Characterise the BER against Eb/No used to represent the performance at all information datarates.

We can convert between Eb/No and the more familiar SNR (S/N) for UMTS through the following

equation:

Where;

B = Bandwidth (Hz). I.e. 3.84MHz

W = Chip Rate (cps). I.e. 3.84Mcps

R = User Information Datarate (bps), e.g. 8kbps, 12.2kbps, 64kbps, 144kbps, etc.

To help illustrate why and how errors occur in a digital modulated channel we can use the ideas of a

Constellation Diagram. The Constellation diagram is a useful tool to represent the symbol decisionstates for say in QPSK Modulation. Each symbol representing bits of 00, 01, 10, and 11. The distance

from the Origin represents the Signal Strength, and the Angle represents the Phase. Alternatively the

Amplitude and Phase are shown as I and Q on the Constellation Diagram, as shown above. The

addition of Noise or Channel Fading will add another vector (Noise Power) which has Random

(Rectangular Distribution) Phase and Random (Gaussian Distributed) Amplitude.

Given enough Noise Power the a QPSK Modulated Symbol may end up nearer another QPSK state,

and be incorrectly decoded, that is we get a Symbol Error or Bit Error(s).

BR

NE

NS

o

b =RW

NE

NE

o

c

o

b =


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Diagram shows the TemporalEye Diagram for QPSK Signalwith SNR=11dB (Eb/No = 8dB).This gives us a BER of about0.0002 (1 Bit Error in 5000).

The Four Colours correspondto the 4 phase states asshown in the ConstellationDiagram.

-1.50

-1.00

-0.50

0.00

0.50

1.00

1.50

0

0.4

0.8

1.2

1.6

2

2.4

2.8

3.2

3.6

4

4.4

4.8

5.2

5.6

6

6.4

Likewise the Eye Diagram can be used to help illustrate the concept of Symbol Errors, or

Bit Errors for a Signal in Noise.

The above images represent the symbol decision boundaries for a QPSK modulated signal

perturbed by Noise which obeys Gaussian Statistics. This means that the received signal issimply the Original Transmitted QPSK signal is added to Noise (Voltage Terms). This

represents the case when we have a signal at a receiver and the power of the signal is only a

few dBs higher than the Thermal Noise Power the signal is competing with. The above

diagrams illustrate the case when we have a SNR of 8dB. Given the statistics of Noise there

exists a (albeit a low) probability that the Noise Voltage will be high, which will perturb the

QPSK signal such that a symbol is incorrectly decoded and a but error(s) results. The above

constellation represents about 11dB SNR and corresponds to a BER of 0.0002.


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Rayleigh Fading Radio Channelhas significant impact uponBER performance.

Since Fast Fading goesthrough deep fades which arein fact Phase Reversals thiscan flip a Symbols State.

Fast Fading alwaysexperiences Phase Reversals,regardless of SNR, or Eb/No.This results in the signalalways experiencing someerrors, which are irreducible.



1.00E-17

1.00E-16

1.00E-15

1.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-031.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

Rayleigh Channel: Severely degraded BER

Gaussian Channel: Normal Eb/No vs. BER

Not all Transmission Channels are Gaussian. Cables, Waveguides, and Satellite-Earth

Radio links are very much Gaussian and the Waterfall Curves for BER against Eb/No

apply. However, in the mobile communications world we have a Rayleigh or Fast Fading

Transmission Channel. We dont just have perturbations in Signal Strength around an

Average Signal Power, but also rapid Phase reversals of the composite signal due to theMulti-path channel.

To help imagine the mechanism of the Fast Fading channel on the QPSK Constellation

diagram we wouldnt see small clouds of decoded constellation points around 4 phase

states. We would see the four clouds but also lots of decoded constellation points around

the origin, as the instantaneous signal has undergone a deep fade, and hence phase reversal.

As datarate slows down we can imagine a Symbol may be at one Phase State (say 45o) on

the constellation diagram, then a deep fade occurs and appears at 215o, before the next

information symbol comes along! The decoder wouldnt know what the Symbol should be.Therefore one would have a family of curves to represent different datarates. Eb/No is then

no longer independent of datarate, and we could now use SNR.

As a Fast Fading signal always encounters phase reversal (transitions through or near the

origin), increasing the SNR (or Eb/No) will not significantly improve BER. It can be shown

that for a Fast Fading Signal, and certain datarate there exists an Irreducible BER figure.


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Eb/No Waterfall Curves canbe improved by introducingError Correction Schemes atthe expense of reducedthroughput.

The effect of adding ErrorCorrection (part of theChannel Coding) is tointroduce a Knee in theWaterfall Curve such thatbeyond a certain Eb/NoErrors can not be correctedand the information collapses.



1.00E-17

1.00E-16

1.00E-15

1.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-031.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

Rayleigh ChannelBER Without

Error Correction

Gaussian ChannelBER Without

Error Correction

Gaussian ChannelBER With

Error Correction

Rayleigh ChannelBER With

Error Correction

In most communication systems the Information is normally protected with various Error

Correction schemes. The introduction of Error Correction Coding will in effect add

redundancy of information, and reduce the overall throughput given a constant bit rate

channel. However, the Error correction schemes will be able to correct a certain percentage

of randomly occurring errors.

The raw Channel still encounters errors (as per the usual Waterfall Curves), but the

Information or decoded channel will appear errorless due to error correction and

information restoration.

If we add Error Correction coding the decoded channel Waterfall curve becomes more of

a two stage curve: the lower part representing complete recovery of information for Bit

Errors, up to a certain Channel Bit Error Rate, and the upper portion representing rapid

deterioration of information since there are too many errors to try and correct.

In the Fast Fading Channel we saw the Waterfall curve for Eb/No against BER reach a

point where there is Irreducible Bit Error rate performance. As long as these errors are

randomised and can be handled by Error Correction Routines then we can achieve a similar

decoded channel Waterfall curve as above.


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Typical Uplink and Downlink Eb/NoValues are shown left.

Eb/No varies with:

Up/Down Link

Datarate Channel Type (and Speed)

QoS for Service

Fast Fading Power ControlLimits

Micro-diversity schemes(such as Spatially separatedantennas)

Eb/No values are determinedthrough experiment orsimulations.

Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian

8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No

Urban Suburban Rural

Source: Nortel Networks


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 1.7 0.4 1.7 0.4 2.2 1.2UDD64 1.7 0.4 1.7 0.4 2.2 0.7

UDD144 0.9 0.7 0.9 0.7 2.1 1.5UDD384 0.6 -0.7 0.6 -0.7 1.2 -0.5

Rural

Mobile Eb/No

Urban Suburban

Eb/No values are shown above for different Datarates, Services, and Speeds for the Uplink

and Downlink.

Eb/No values can only be realistically derived from simulations or trials A theoretical

approach would be too complex. In the UMTS system the Receivers are constantly takingmeasurements of BER, and adjusting the Target Eb/No such that the Service Quality (or a

certain minimum BER) is maintained. One can imagine the Target Eb/No varying as the

mobile terminal movement speeds up/down, encounters interference from another cell(s),

or changes datarate (for a variable bit rate service).

A lower Eb/No can be achieved when the mobile can effectively compensate for the Fast

Fading Radio Channel (shown in the next few slides). However, the Eb/No Target will also

ramp up as the Maximum Power is reached on the Mobile Terminal. As the Mobile reaches

Full Power, it can not effectively compensate for Fast Fading. This results in the need for a

higher Eb/No at the Cell Edge.

LCD = Low Constrained Delay data (low latency, high QoS, such as Voice, ISDN, or

Video streaming type services)

UDD = Unconstrained Delay Data (variable latency, variable QoS, such as FTP, Web

Access, email, and other non time critical services)

LCD and UDD are terms used to generally describe the Service Container. All Services can

be mapped to UDD and LCD together with QoS Targets, BER, FER, mi nimum bandwidth,

maximum bandwidth, latency, throughput, etc. Different QoS, BER, and FERs can bedesigned through use of coding, interleaving, etc.


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Generally Speaking Eb/No ishigher for:

Delay intolerant services

Lower Datarate Services

Higher Mobile Speeds

Higher Power DelaySpread Environment

Circuit Switched (LCD)services over the samedatarate PacketSwitched (UDD) services

The Uplink (only forPacket services)


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No


Source: Nortel Networks


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 1.7 0.4 1.7 0.4 2.2 1.2UDD64 1.7 0.4 1.7 0.4 2.2 0.7

UDD144 0.9 0.7 0.9 0.7 2.1 1.5UDD384 0.6 -0.7 0.6 -0.7 1.2 -0.5

Rural

Mobile Eb/No

Urban Suburban

The same Eb/No values are shown above for different Datarates, Services, and Speeds for

the Uplink and Downlink. Why do we get variability in Eb/No values. The general rules are

shown in the slide and are discussed in the next few slides.


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Delay Intolerant Servicesrequire a higher Eb/No than asimilar more delay tolerantQoS service. Assuming allother factors are the same.

Not many service examplesbut could be Voice comparedto Voice Messaging Service.

Not shown on table but theabove applies for 64kbpsbeing more delay tolerant than8kbps Voice (assuming theyalso had the same QoS BER)


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No


Probability of Bit Error (or BER) for QPSK Modulation andCoherent Detection at Receiver

1.00E-17

1.00E-16

1.00E-151.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-03

1.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

BER With

Error Correction and80ms Interleaving

BER WithError Correction and

20ms Interleaving

Eb/No = 5dB

BER = 10 -3Eb/No = 3.5dB

BER = 10 -3

Delay intolerant services such as Conversational Voice require a higher Eb/No over a

more delay tolerant, similar datarate and QoS target service. This is because services such

as Conversational Voice can only receive a 20ms interleaving depth of data during

Physical Layer, Transport Sub-Layer processing. In fact Conversational Voice always

uses a 20ms interleaving depth. Higher interleaving depths mean greater randomisation ofbit errors and hence better Error Coding performance at the expense of greater processing

delays. A Service such as a non-conversational Voice Messaging service, or Audio

Streaming Radio Service could in principle use a deeper interleaving depth and benefit

from a lower Eb/No.

The Eb/No table does not show this but the values ringed could in principle represent a

Voice service and a 64kbps service, where the 64kbps service has the same QoS BER

target, and use a deeper interleaving depth. A 64kbps LCD service could be used to

transport an Audio Streaming service. The lower graph illustrates the Eb/No vs. BER

performance for two services, one Conversational Voice, and the other Non-Conversational Voice where the latter uses a deeper interleaving depth. The graph is

purely illustrative and the curves are not based upon any real simulations or measurements.


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Lower Datarate Servicesgenerally require a higherEb/No than a similar delaytolerance, and QoS service.Assuming all other factors arethe same.

Service examples include sayFTP using different datarates.

384kbps LCD requires higherEb/No than 144kbps. DuringTransport Formattingpuncturing is used to ratematch the 384kbps service,whereas repetition is used for

144kbps service.


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No



1.00E-17

1.00E-16

1.00E-151.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-03

1.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

BER with codinggain due to 64kbps

information rate

Eb/No = 3dB

BER = 10 -4

Eb/No = 2dB

BER = 10 -4

BER with greater coding

gain due to 144kbpsinformation rate

Lower datarate services such as 64kbps LCD require a higher Eb/No than similar QoS

higher datarate services (such as 144kbps LCD). This is because higher datarate services

carry more bits per Transmission Time Interval (TTI) during Physical Layer, Transport

Sub-Layer processing and as such the Error Coding can perform better when applied on a

greater bit length. There can be differences, e.g. the 384kbps uplink LCD service in theabove table needs a slightly higher Eb/No than 144kbps uplink LCD service. This is

because the Reference Physical Channels for the above tables correspond to the

Reference Transport Formats detailed in Module 3, section 3.5 (WCDMA Physical

Layer, and in 3GPP TS25.101) where puncturing is applied to the 384kbps channel during

rate matching whereas repetition is applied to the 144kbps uplink channel.

The Eb/No shows this with typical values ringed and could represent two similar services at

different datarates. The lower graph illustrates the Eb/No vs. BER performance for two

services, one FTP at 64kbps, and the other FTP at 144kbps where the latter benefits

from more efficient error correction perfromance. The graph is purely illustrative and thecurves are not based upon any real simulations or measurements.


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Services at higher mobilespeeds require a higher Eb/Nothan a the same service at alower speed.

Service examples include sayany two services at twodifferent mobile speeds.

Fast Power Control providessome equalisation of a FadingChannel seen at the BaseStation. A fading channel canbe equalised when at lowmobile speeds. Lessequalisation occurs at higher

speeds.


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No



1.00E-17

1.00E-16

1.00E-151.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-03

1.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

Eb/No = 6dB

BER = 10 -3Eb/No = 5dB

BER = 10 -3

Voice at 3km/h

Voice at 50km/h

Mobiles at low speeds can effectively combat the fast fading channel through fast power

control. As a result the channel appears Gaussian at the receiver and Equalisation (at least

in terms of power fluctuation) is achieved. A mobile travelling at a higher speed will not be

able to combat the fading channel as efficiently and the channel appears more Rayleigh at

the Base Station receiver. This is discussed in more detail a few slides ahead.

The Eb/No shows this with typical values ringed and could represent two similar services at

different mobile speeds. The lower graph illustrates the Eb/No vs. BER performance for

two voice services, one with a mobile travelling at 3km/h and the other with a mobile

travelling at 50km/h where the latter benefits from more efficient Fast Power Control and

equalisation of the fading channel. The graph is purely illustrative and the curves are not

based upon any real simulations or measurements.


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Services in Higher Power DelaySpread Environments require ahigher Eb/No than a the sameservice in a less dispersivechannel.

Service examples include say anytwo services in two differentMultipath environments.

In a less dispersive channel lesschip energy is spread in time.Although the Rake Receiverrecovers energy for each chipspread in time, any chip energyspread over over chip periodsintroduces inefficiencies.


8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4

LCD144 1.7 0.5 1.7 0.5 2.2 0.5

LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2

UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2

Base Station

Eb/No



1.00E-17

1.00E-16

1.00E-151.00E-14

1.00E-13

1.00E-12

1.00E-11

1.00E-10

1.00E-09

1.00E-08

1.00E-07

1.00E-06

1.00E-05

1.00E-04

1.00E-03

1.00E-02

1.00E-01

1.00E+00

-5 0 5 10 15 20

Eb/No

Pb(BER

Eb/No = 6dB

BER = 10 -3Eb/No = 5dB

BER = 10 -3

Voice in Sub-urban

Voice in Rural

The greater the time dispersion of the channel the more energy per chip is spread over other

chip periods (the Wideband Channel). Using a Rake receiver with a number of fingers

aligned at delayed received energy peaks attempts to recover the total energy per chip.

However, not all energy will be recovered due to the finite number of Rake Receiver

fingers. Also, any single finger and hence delayed chip sequence will in principle have alow cross-correlation with other users chip sequences, but not zero cross correlation, from

the properties of the Scrambling codes used by each user in the Uplink. As a result chip

sequence energy recovery in a Rake Receiver finger will never be perfect. If all the energy

of each chip of a users chip sequence falls within one chip period we do not need to use

Rake Receiver fingers, and as a result better receiver performance can be achieved or

alternatively seen as an improvement in Eb/No. The actual values of Eb/No in different

Multipath environments will be a function of the Rake Receiver performance and in

particular the number of Rake Receiver fingers. The Rake Receiver and number of fingers

is not specified by the 3GPP specifications and is left to the vendor to engineer. Refer also

to section 3.2 (Rake Receiver and Matched Filters) of the Course notes for further

information.

The Eb/No table shows this with typical environments leading to different dispersive

characters ringed and could represent two similar services in different environments. The

lower graph illustrates the Eb/No vs. BER performance for two voice services, one in a

Rural environment and the other in a

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