Non-Load-Balance-Dependent High EfficiencySingle-Inductor Multiple-Output (SIMO)

DC-DC ConvertersY. H. Ko, Y. S. Jang, S. K. Han, S. G. Lee

Korea Advanced Institute of Science and Technology

Abstract—A single-inductor multiple-output (SIMO) DC-DCconverter providing buck and boost outputs with a new controltopology is presented. In the proposed switching sequence, whichdoes not require any special blocks, energy delivery is alwaysaccomplished by flowing energy through an inductor, which leadsto high conversion efficiency regardless of the balance betweenthe buck and boost output loads. Implemented in 0.35-µm CMOS,the proposed SIMO DC-DC converter achieves high conversionefficiency regardless of the load balance between the outputs. Themeasured maximum efficiency reaches 82 % under heavy loads.

I. INTRODUCTION

Along with the growth of the battery-powered portable de-vice market, the development of efficient power managementintegrated circuits (PMIC) is currently a popular and activeresearch area. A DC-DC converter, which is an essentialblock in PMIC, provides a regulated output voltage fromlinearly discharging battery voltage in portable devices. Inmany portable device systems which require multiple regulatedoutput voltages from only one battery supply, including mul-tiple DC-DC converters can be a viable means of generatingdifferent output voltage levels. However, this approach leadsto an increase in cost and area requirements due to theuse of multiple off-chip inductors, which is not suitable forportable devices. To circumvent this issue, the single-inductormultiple-output (SIMO) DC-DC converter has been proposed.It literally requires only one inductor to regulate multipleoutput voltages. Various works [1]-[4] have focused on SIMODC-DC converters that adopt a single energizing cycle perswitching period, leading to performance levels with greateraccuracy characterized by smaller output voltage ripples andfaster control loops [5]. Several works [1]-[3] have proposedcontrol topologies, each with their own advantages. However,their applications are limited by lack of flexibility as regardsbuck and boost implementations. One study [4] presented aSIMO DC-DC converter that provides both buck and boostoutputs simultaneously with a reduced number of powerswitches. To overcome a stability problem that arises duringunbalanced output loads, a hysteresis mode was proposed[4]. However, implementing the hysteresis mode not onlyrequires special blocks such as a power comparator and adelta-voltage generator circuit, but it also degrades the powerconversion efficiency because the current flows not throughthe inductor but through the resistive freewheeling switch asthe the buck output is charged. Thus, this paper proposes

VOA < VIN < VOB

VINVOA

VOB

V NVOA

VOB

V NVOA

VOB

V NVOA

VOB

VINVOA

VOB

SW1

SW2 SW3

SW

SW5

SW1

SW2 SW3

SW

SW5

SW1

SW2 SW3

SW

SW5

SW1

SW2 SW3

SW

SW5

SW1

SW2 SW3

SW

SW5

(1)

(2-1)

(2-2)

(3) (4)

IL

s 2 s 3 s 4 s ime

Ip

(1)

(2-1)

(3)

( )

(1)

(2-1) (3)

( )(1)

2-2)

3)

( )

PWM PWMBuck < Boost Buck > Boost

Operation modeLoad condition

A = IOA X TSB = IOB X TS

BABABA

Fig. 1. Control sequence of a proposed SIMO DC-DC converter

a new control topology. A switching sequence is presentedto generate multiple buck and boost outputs without specialblocks while achieving high conversion efficiency.

This paper is organized as follows. Section II introduces thenewly proposed control topology. Then, the implementationdetails of a DC-DC converter that adopts the newly proposedcontrol sequence are shown in Section III and the measurementresults are given in Section IV. The conclusions are made inSection V.

II. PRINCIPLE OF THE PROPOSED CONTROL TOPOLOGY

A new control sequence is proposed, as shown in Fig.1, to resolve the problems in the hysteresis mode [4]. Theoperation of a SIMO DC-DC converter with one buck andone boost output, adopting the proposed control sequence, canbe described as follows. During phase-(1), switches SW1 andSW4 are turned on to charge the buck output. The inductorcurrent IL then ramps up at a rate of (VIN − VOA)/L.The next phase is then determined between phase-(2-1) andphase-(2-2) depending on output load condition. When thebuck output finishes being charged to a desired voltage levelbefore the inductor current reaches a peak-current Ip, whichcan be controlled by previously reported techniques, phase-(2-1) is selected. Switches SW1 and SW3 are then turnedon to energize the inductor during phase-(2-1), which makesIL increase with a slope of VIN/L until IL reaches Ip.On the other hand, phase-(2-2) is chosen if more energy isrequired to regulate the buck output when IL reaches Ip.During phase-(2-2), Switches SW2 and SW4 are turned onto deliver more charge to the buck output while IL rampsdown at a rate of −VOA/L. After phase-(2-1) or phase-(2-2)

978-1-4673-1556-2/12/$31.00@2012 IEEE

TABLE ICONTROL SEQUENCES DEPENDING ON LOAD CONDITIONS

Load conditions Control sequencesBuck < Boost 1 → 2-1 → 3 → 4Buck > Boost 1 → 2-1 → 3 → 4 → 1 → 2-2 → 3 → 4

Peak-CurrentControl Circuit

Peak-CurrentSensor Freewheeling

Current Sensor

VOK

VOT

VSWT

VSWK

VSWN1VSWN2

VSWI

VINL

COT

COK

IOT

IOK

BGR

SlopeCompensation

Circuit

VREF

VREF

VOK

VOT

Iref

I-V Converter

VSWN1VSWN2

VSWI

VSWT VSWK

If

Ifs

ILVX VY

Ilps

CMR

CMT

CMK

VOCP

VINVX VREF

SW1

SW2 SW3SW4

SW5

DeadTime

ControlLogic

&Driver

ControlLogic

Clock

VSWIVSWN1VSWN2VSWTVSWK

VCLK

Ilp

VREF

Fig. 2. Block diagram of the proposed SIMO DC-DC converter with non-load-balance-dependent high efficiency

is finished, phase-(3) begins and then IL ramps down at arate of (VIN − VOB)/L with switches SW1 and SW5 turnedon. As soon as the boost output reaches thee desired voltage,phase-(3) becomes phase-(4) which is a freewheeling period.In phase-(4), switches SW2 and SW3 are turned on to holdIL constant. Upon the end of the switching period, the routinereturns to phase-(1). In summary, according to output loadcondition, two types of control sequences are used to regulatethe buck and the boost outputs, as shown in Table I. Note that,high conversion efficiency can be achieved because energydelivery is always accomplished by flowing energy throughthe inductor. Moreover, the decision process between phase-(2-1) and phase-(2-2) can be implemented digitally withoutany special blocks.

III. IMPLEMENTATION DETAILS

The block diagram of the proposed SIMO DC-DC converterwith non-load-balance-dependent high efficiency is illustratedin Fig. 2. The proposed control sequence, which was explainedin Section II, can be implemented by a control logic block.The required control signals (CMR, CMT , and CMK) aregenerated from other sub-blocks. A peak-current Ip, whichis one of the factors that determines the duty cycle for eachphase, can be controlled by adopting a freewheeling currentfeedback control scheme [2]. The output voltages are regulateddirectly by comparators in the freewheeling current feedbackcontrol scheme. A single compensation is sufficient for afreewheeling feedback loop. The principle of the freewheelingcurrent feedback control scheme is to control the peak currentby comparing the average of the freewheeling current with

(a) (b)Fig. 3. Inductor current waveforms for (a) case-I (Buck < Boost) and (b)case-II (Buck > Boost)

a reference current. Even though the basic concept of thefreewheeling current feedback control is used, the DC andAC characteristics of the proposed SIMO DC-DC converterare different from [2] because the control sequences are dis-similar. There are two cases for the DC and AC characteristicsaccording to the load condition. Each case is analyzed in thenext parts.

A. Case-I : Buck < BoostWhen the power of boost output is larger than that of buck

output, the current waveform is shown in Fig. 3(a). The dutycycle of each phase is given by

dk1 =fsmK· (ik1 − ifw1) (1)

dn1 =fsmN1

· (ip − ik1) (2)

dt1 =fsmT· (ip − ifw1) (3)

df1 = 1− dk1 − dn1 − dt1, (4)where mK , mN1, and mT are (VIN − VOA)/L, VIN/L,and −(VIN − VOB)/L, respectively. The average of thefreewheeling current, which should be regulated to Iref , isgiven by

〈if1〉T = df1 · ifw1. (5)The averaged supplied currents to the buck and boost outputsare given by

〈iok〉T =1

2· (ik1 + ifw1) · dk1 (6)

〈iot〉T =1

2· (ip + ifw1) · dt1. (7)

From the above equations (1)-(7), the quiescent operatingpoint is determined. At the operating point, the small signalgain G1(s) from ip to if1 can be calculated by constructingperturbation variable matrices.

B. Case-II : Buck > BoostWhen the power of boost output is smaller than that of buck

output, the current waveform is shown in Fig. 3(b). The dutycycle of each phase is given by

dk2 =fsmK· (ik2 − ifw3) (8)

dn2 =fsmN1

· (ip − ik2) (9)

dt2 =fsmT· (ip − ifw2) (10)

df2 = 1− dk2 − dn2 − dt2 (11)

Fig. 4. Small signal gain G0 versus load currents Iok and Iot

VP

Cc

Rc

Iref

Ifs

VREF

Fig. 5. Peak-current control circuit

dk3 =fsmK· (ip − ifw2) (12)

dn3 =fsmN2

· (ip − ik3) (13)

dt3 =fsmT· (ik3 − ifw3) (14)

df3 = 1− dk3 − dn3 − dt3, (15)where mN2 is VOA/L. The averaged freewheeling current,which should be equal to Iref , is given by

〈if2〉T =1

2· (df2 · ifw2 + df3 · ifw3). (16)

The averaged supplied currents to the buck and boost outputsare given by

〈iok〉T =1

2· (ik2 + ifw3) · dk2 (17)

=1

2· (ip + ifw2) · dk3 + (ip + ik3) · dn3(18)

〈iot〉T =1

2· (ip + ifw2) · dt2 (19)

=1

2· (ik3 + ifw3) · dt3. (20)

The above equations (9)-(21) determine the quiescent operat-ing point. At the operating point, the small signal gain G2(s)from ip to if2 can be obtained by using perturbation variablematrices. Fig. 4 shows the calculated gain G0 versus buck andboost load currents where G0 can be G1,0 or G2,0 accordingto the load condition. The design parameters are chosen asL = 10 µH, VIN = 3.7 V, VOK = 1.8 V, VOT = 5 V,fs = 1 MHz, and Iref = 30 mA. The gain G0 is consideredto compensate for the freewheeling feedback loop in a peak-current control circuit.

C. Peak-Current Control Circuit

The peak-current control circuit, which is an integrator withone pole, is shown in Fig. 5. The value of Iref is chosen forthe freewheeling periods (Df1, Df2 and Df3) to be larger

Fig. 6. Micrograph of the proposed SIMO DC-DC converter

(a) (b)

Fig. 7. Inductor current waveforms for (a) case-I (IOK=10 mA and IOT =80mA) and (b) case-II (IOK=80 mA and IOT =10 mA)

Fig. 8. Maximum output ripples for IOK=80 mA and IOT =80 mA

than 0.1 for accurate operation because a freewheeling currentsensor takes time to transition from an off-state to an on-state.Adequate freewheeling periods also guarantee good cross-regulation performance. Ilps (=Ilp/N1) and Ifs (=If/N2) arethe scaled-down current from the peak-current sensor and thefreewheeling current sensor, respectively. The loop gain of thefreewheeling feedback loop is given by

T (s) =N1

N2 ·RP·G(s) ·A(s)

=1

RP· G0 ·Rc

1 + sRc · Cc, (21)

where N1 = N2 = N and RP is the value of the resistor in theI-V converter (I-V conversion ratio). Generally, the unity gainfrequency of a feedback loop, which should be sufficientlylower than the Nyquist rate, is set at one tenth of the switchingfrequency. Because the effective frequency of the inductorcurrent waveform for case-II is one half of the switchingfrequency fs, the unity gain frequency (fu = G0/(2π·RP ·Cc))of the freewheeling feedback loop is designed to be aboutfs/20.

(a) (b)

Fig. 9. Load transient response for (a) IOK changes of 10 mA to 80 mAunder IOT =10 mA and (a) IOT changes of 10 mA to 80 mA under IOK=10mA

Fig. 10. Line transient response for line variations of 3 V to 4 V withIOK=40 mA and IOT =40 mA

Fig. 11. Conversion efficiency of the proposed SIMO DC-DC converter

IV. MEASUREMENT RESULTS

The proposed SIMO DC-DC converter is implemented in a0.35-µm CMOS technology. The chosen design parameterswere as follower: COK = COT = 22 µF, L = 10 µH,VIN = 3.7 V, VOK = 1.8 V, VOT = 5 V, and fs = 1MHz. Fig. 7 shows the measured inductor current waveforms.The operation mode changes according to the output loadcondition. Fig. 8 shows the maximum output ripples for themaximum output load currents. The maximum output ripplesof the buck and boost outputs are 5.6 mV and 4.8 mV,respectively. Because the operation of the proposed schemeis even stable for the small value of ESR, output voltageripples can be minimized. Fig. 9 shows the load transientresponse when pulse loads of 10 mA to 80 mA are applied.The load regulations of the buck and boost outputs for theload current variations are 10 mV and 32 mV, respectively.Fig. 10 shows the line transient response for step line changesbetween 3 V and 4 V. The line regulations of the buck and

TABLE IIPERFORMANCE SUMMARY

Process 0.35-µm CMOSChip area 1460 µm x 1250 µm

Input voltage 3.7 V (3 - 4 V)Switching frequency 1 MHz

Inductor 10 µH (400 mΩ DCR)Output capacitor 22 µF (10 mΩ ESR)Converter type Buck (VOK ) Boost (VOT )Output voltage 1.8 V 5 V

Maximum load current 80 mA 80 mAOutput ripples 5.6 mV 4.8 mV

Load regulation 0.142 mV/mA 0.457 mV/mALine regulation 7 mV/V 7 mV/V

boost outputs for the line variations are 7 mV and 7 mV,respectively. The power conversion efficiency of the proposedSIMO DC-DC converter is shown in Fig. 11. It is verifiedthat high conversion efficiency is achieved regardless of theoutput load condition. There is no conversion efficiency dropaccording to mode change at the load balance boundary. Theproposed scheme achieves the maximum conversion efficiencyof 82 % under heavy load conditions. Table II summarizes thekey performances of the proposed SIMO DC-DC converter.

V. CONCLUSION

In this paper, a single-inductor multiple-output (SIMO) DC-DC converter capable of regulating buck and boost outputswith non-load-balance-dependent high efficiency is proposedand implemented. By adopting the newly proposed controlsequence, input energy can be delivered to the outputs througha lossless inductor continually without a current accumulationproblem. Thus, the proposed SIMO DC-DC converter canachieve high efficiency unrelated to the load condition. Im-plemented in a 0.35-µm CMOS technology, the measurementresults of the proposed SIMO DC-DC converter show even aconversion efficiency of 82% under heavy load conditions.

ACKNOWLEDGMENT

This research was financially supported by the Ministry ofEducation, Science Technology (MEST) and Korea Institutefor Advancement of Technology (KIAT) through the HumanResource Training Project for Regional Innovation.

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