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lI2~ United States PatentFurst et al. (Io) Patent No.: US 7,899,196B2(45) Date of Patent: Mar. 1, 2011(54) DIGITAL MICROPHONE (56) References Cited(75) Inventors: Claus Erdmann Furst, Roskilde(DK);

Henrik Thomsen, Dyssegard (DK); IgorMucha, Bratislava (SK)(73) Assignee: Audioasics A/S, Roskilde (DK)( * ) Notice: Subject to any disclaimer, the term of thispatent is extended or adjusted under 35U.S.C. 154(b) by 1203 days.(21) Appl. No 4 10/597,552(22) PCT Filed: Feb.9,2005(86) PCT Noz PCT/DK2005/0000S6

) 371 (c)(I),(2), (4) Date: Sep. 13,2006

U.S. PATENT DOCUMENTS5,339,285 A5,357,214 A5,861,778 A6,104,821 A6,160,450 A6,275,109 Bl *6,516,069 Bl6,573,785 Bl *6,583,658 Bl6,661,217 B2 *2002/0106091 Al2003/0045252 Al2003/0059060 Al2003/0235315 Al

8/199410/19941/19998/200012/20008/20012/20036/20036/200312/20038/20023/20033/200312/2003

StrawHeyl et al.Louagie et al.Husung et al.Eschauzier et al.TallgTakeuchi et al.Callicotte et al.....Kern et al.Kimbal1 et al.Furst et al.NamEastty et al.ReesorFOREIGN PATENT DOCUMENTS

WO WO-2005/003941 4/2005* cited by examiner

.. 330/261... 330/9

324/76.41

(87) PCT Pub. No 4 WO2005/076466PCT Pub. Date: Aug. 1S,2005

(65) Prior Publication DataUS 2009/0316935 Al Dec. 24, 2009

(30) Foreign Application Priority DataOct. 6, 2004 (DK) ... PA 2004 01531

(51) Int. Cl.HO4R 3IOO (2006.01)HO3/((I 3IOO (2006.01)

(52) U.S. Cl. . 3S1/111; 341/143(58) Field of ClassiTication Search ..................81/111,381/121, 120, 122; 330/252, 253, 260, 297,330/303, 306; 341/143, 155See application file for complete search history.

Related U.S.Application Data(60) Provisional application No. 60/542,305, filed on Feb.9, 2004.

Primary Examiner Curtis KuntzAssistant Examiner Hai Phan(74) Attorney, Agent, or Firm Kenyon & Kenyon LLP(57) ABSTRACT

20 Claims, 11Drawing Sheets

An integrated circuit, configured to process microphone sig-nals, where the integrated circuit comprises: a preamplifier(306) with an amplifier section (301)which has a first input($) and a second input ($*) and an output ($), and with afeedback filter network (Z1; Z1, Z1*, Z2) coupled betweenthe output ($ ; $ , $*)and the second input ($'); where the firstinput ($) to the amplifier section (301)has an input imped-ance which by means of the input impedance of the amplifiersection is substantially isolated from the feedback networkwith respect to input impedance; and where the preamplifierhas a frequency-gain transfer function which suppress lowfrequencies; and an analogue-to-digital converter coupled toreceive an anti-aliasing filtered input signal and providing adigital output signal (Do).

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U.S.Patent Mar. 1, 2011 Sheet 9 of 11 US 7,899,196B2T5

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US 7,899,196B21

DIGITAL MICROPHONECROSS REFERENCE TO PRIOR APPLICATION

The above-referenced application is a National Phase ofInternational Patent Application of PCT/DK2005/000086filed on Feb. 9, 2005, and claims the benefit ofU.S. Provi-sional Application No. 60/542,305 filed Feb. 9, 2004 andDanish Patent Application No. PA 2004 01531, filed Oct. 6,2004. The International Application was published on Aug.18, 2005 as WO 2005/076466.The present invention relates to an integrated circuit com-prising a microphone preamplifier and an analogue-to-digitalconverter to provide a digital output signal. Additionally, itrelates to a digital microphone comprising a microphoneelement and an integrated circuit as set forth above.

BACKGROUNDA sound pressure detected by the microphone element willcause its membrane to move and consequently change thecapacitance of the capacitor formed by its membrane and a

so-called back plate of the microphone element. When thecharge on the capacitor formed by these two members is keptconstant, the voltage across the two capacitor members willchange with the sound pressure acting on the membrane. Asthe charge on the microphone capacitor has to be kept con-stant to maintain proportionality between sound pressure andvoltage across the capacitor members, it is important not toload the microphone capacitance with a resistive load. Aresistive load will discharge the capacitor and therebydegrade or ruin the capacitor' performance as amicrophone.There fore, in order to pick up a microphone signal from thecapacitor, amplifiers configured with the primary objective ofproviding high input resistance are preferred in order to bufferthe capacitor from circuits which are optimized for otherobjectives. The amplifier connected to pick up the micro-phone signal is typically denoted a preamplifier or a bufferamplifier or simply a buffer. The preamplifier is typicallyconnected physically very close to the capacitor within adistance of very few millimeters or fractions ofmillimeters.For small sized microphones the active capacitance is verysmall (typically I to 10pF).This further increases the require-ment ofhigh input resistance and capacitance. Consequently,the input resistance of preamplifiers for small sized micro-phones has to be extremely high in the magnitude of Gigaohms. Additionally, the input capacitance of this amplifier hasto be very small in order to achieve a fair sensitivity to soundpressure.Especially so-called telecom microphones with an inte-

grated preamplifier are sold in high volumes and at very lowprices. As cost of an amplifier for a telecom microphone isdirectly related to the size of the preamplifier chip die it isimportant, for the purpose of reducing price, that the pream-

p

chi die is as small as possible. Therefore extraordinaryattention is drawn to compact circuits and such circuits are invery high demand. However, it is important in this respect toprovide circuits with a low noise level. Low noise is importantas noise can be traded for area i.e.if the circuit has low noiseand a noise lower than required, this noise level can be tradedfor lower chip die area and it is thus possible to manufacturethe preamplifier at lower cost.When designing a preamplifier in CMOS technology for amicrophone there is normally three noise sources. Thesesources are noise from a bias resistor, I/f noise from an inputtransistor, and white noise from the input transistor. Weassume that input transistor noise dominates. Both white

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noise and I/f noise can beminimized by optimizing the lengthand the width of the input transistor(s). This applies for anyinput stage e.g. a single transistor stage or a differential stage.The noise from the bias resistor can also be minimized. If thebias resistor is made very large then the noise from the resistorwill be high-pass filtered and the in-band noise will be verylow. This has the effect though that the lower bandwidth limitof the amplifier will be very low. This can be a problem as theinput of the amplifier will settle at a nominal value only aftera very long period of time after power up. Additionally, sig-nals with intensive low frequency content arising form egslamming of a door or infra sound in a car can overload theamplifier. Another related problem is small leakage currentsoriginating from mounting of the die inside a microphonemodule. Such currents will due to the extreme input imped-ance establish a DC offset. This will reduce the overloadmargin of the amplifier.In addition to the above there is a demand for digital micro-phones comprising a microphone element and an integratedcircuit with a preamplifier and an analogue-to-digital con-verter to provide a digital output signal. Since typically atelecom microphone is integrated in a consumer electronicsdevice where a substantial amount ofdigital signal processingis performed by a mainly digital integrated circuit chips it is ingeneral preferred that signals from sensors (such as micro-phones) are provided as digital signals. This introduces newchallenges in respect of signal processing in the integratedcircuits embedded with the microphones and especially inrespect of distortion in the digital domain.In recent years so-called sigma-delta modulators havebecome very popular for implementing A/D converters. Theyexhibit many virtues, which among others are: no need forhigh precision components; high linearity; and for so-calledsingle loop modulators also the advantages of small die area,low voltage operation and possibly very low power consump-tion. These are advantages which makes sigma delta modu-lators very suitable for single chip implementations.A special class of sigma delta modulators are I-bit quan-tized sigma delta modulators. This type of modulators areespecially suited for low cost implementations as the com-

plexity of the analogue part of the A/D converter is minimalcompared to other types ofA/D converters. A complete I-bitsigma delta converter consists of a I-bit analogue sigma deltamodulator and a digital decimation filter only. The normallyrequired higher order anti-aliasing filter can be implementedby a simple RC-filter. This is due to the fact that heavyover-sampling is used and thus the digital decimation filterperforms the job of anti-aliasing filtering.I -bit sigma delta modulators are very simple to implementin the analogue domain. Thus they are very suitable for lowcost miniature digital microphones. Unfortunately they doalso have disadvantages. Especially I -bit sigma delta modu-lators exhibit the so called idle mode tones, which are lowlevel tones in the audio band caused by low frequency or DClevels at the input of the modulator. This is the reason whyI-bit sigma delta modulators has been abandoned by manydespite of its many virtues. One can use dither to remove thisproblem or design chaotic modulators: but all of these solu-tions has the effect that the complexity of the design increasesdramatically. Thus both power consumption and die areaincreases dramatically.This idle-mode-tones effect has caused sigma delta modu-lators less suitable for high quality audio applications. Appar-ently, this may seem to be of little concern in consumer/telecom applications. But as the demand for low cost digitalmicrophones increases higher demands of performance,which may almost equal the performance of high quality

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US 7,899,196B2audio, will follow. Consequently, the idle-mode-tones effectwill become an increasing problem also for telecom applica-tions.In order to achieve high performance from the digitalmicrophone, the preamplifier of the digital microphone ASIChas to have as high performance as possible i.e.Iow noise, lowdistortion, high dynamic range etc. According to presentlyavailable technology, CMOS technology is a prerequisite toachieve low noise performance and it can be shown that theinput stage of the amplifier can be optimized in respect tonoise. Also the input impedance should be as large as possiblein order to minimize the noise. This is especially dominant fornew and thinner types of telecom microphones which has amuch lower sensitivity and cartridge capacitance than previ-ously experienced.Unfortunately this has the consequence that the preampli-fier becomes capable of amplifying low frequency signals

arising from the sound pressure of a door slamming, carrumbling or just changes in sensitivity of the microphoneelement due to humidity changes. This adds to the aboveexplained problem of idle tone modes if a I-bit sigma deltamodulator is used. In fact also 2-bit and modulators with evenmore levels will exhibit such behaviour when exposed to suchlow frequency signals.Additionally, these low frequency signals reduces the

dynamic range and creates inter-modulation distortion as thelow frequency signals can be excessive in amplitude.The problem is worsened as the telecom microphones arebecoming smaller and thinner and thus more gain is requiredfrom the preamplifier. However, normally the disturbing lowfrequency signals do not become smaller in amplitude. Thusthe relative effect of the disturbance will increase.So there is a need for a configuration of a preamplifier andanA/D converter which is suited for thin ECM cartridges witha very low cartridge sensitivity and capacitance. Additionally,the configuration should provide a very high performance onnoise, dynamic range and distortion. Moreover, it shall befeasible to implement the configuration on a single chip diewith a very small area in combination with few or noneexternal components.In the below description, the term audio band is used. In theprior art this term have various definitions depending on itscontext. However, in the below it will be used to designate afrequency band which typically has a lower corner frequencyof 20 Hz to 500 Hz and an upper corner frequency of 5 KHzto 25 KHz. The specific definition of the band represents adesign criterion, but for the below description it should beread with this broad definition.

RELATED ARTA so-called two stage preamplifier configuration, in whicha simple buffer amplifier is followed by filter, has two disad-vantages: as it has two stages, it is noisier and because there isno gain in the first stage the physical size of the filter has to berelatively large. The size of the filter could be minimized byincreasing the gain of the first stage, but the amplifier wouldbe sensitive to overload because of low frequency compo-nents which are not diminished until in the subsequent filter.

Thus such a solution, originally developed for hearing aidmicrophones, will be far from optimal for new high sensitivetelecom microphones. The area of the amplifier die wouldsimply be too large and the device consequently too costly.Since, the chip area occupied by the preamplifier must beas small as possible to obtain relatively low cost, the pream-plifier must be as small as possible. Therefore, since amplifierconfigurations known from hearing aids are generally not

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optimised for chip area to the same extent as telecom micro-phones, these configurations are not applicable for telecomapplications. Further, one should bear in mind that buffers oramplifiers applied in hearing aids are not configured to pro-vide such high gain levels as are required for the low-sensi-tivity microphones used in telecommunication applications.U.S. Pat. No. 6,583,658-BI discloses a balanced circuitarrangement for converting an analogous input signal from afirst terminal of a capacitor microphone into a symmetricaloutput signal. A first operational amplifier is coupled as avoltage follower and its output is provided as a first outputsignal of the symmetrical output signal. A second terminal ofthe capacitor microphone is coupled directly to the output ofa second operational amplifier, which thereby provides asecond output signal of the symmetrical output signal. Thesecond operational amplifier' non-inverting input is coupledto a ground reference whereas its inverting input is coupled toa voltage divider which provides a voltage midway betweenthe second and first output signal. The symmetrical outputsignal is provided to an analogue-to-digital converter of thesigma-delta type which provides a binary output signal.The disclosed configuration is expedient in that relativelyhigh voltage levels from the capacitor microphone can behandled while providing low noise. However, the capacitormicrophone in combination with the input impedance of theamplifier will form a filter with only a very slowly decayingimpulse response. When the microphone is exposed to tran-sient sounds with large amplitude or low-frequent signals thisin turn will generate very slowly varying signal componentswhich are input to the analogue-to-digital converter of thesigma-delta type. In the sigma-delta type analogue-to-digitalconverter the signal components will generate so-called idlemode tones in the binary output signal. Further, the configu-ration is sensitive to overload caused by a largely unevendistribution of the signal spectrum generated by the micro-phone.US 2002/0106091-Al discloses a microphone unit with aninternal analogue-to-digital converter. The unit comprises asound transducer (a capacitor microphone with an electretmember), a pre-amplifier coupled at its input to the soundtransducer and at its output coupled to provide a signal inputto the analogue-to-digital converter. A high-pass filter situ-ated between the pre-amplifier and the analogue-to-digitalconverter is configured to block DC signals and reduce thenoise level. Additionally, a low-pass filter is configured as ananti-aliasing filter which is situated between the pre-amplifierand the analogue-to-digital converter.Although this configuration addresses important signalprocessing aspects, basic, but key implementation problemsrelated to performance in terms of cost and noise are leftunsolved. The disclosed configuration involves several signalprocessing stages. Each contributes to increasing the noiselevel. Additionally, the several stages occupy a large chip areawhich implies higher costs. Further, the configuration is sen-

sitive to overload caused by a largely uneven distribution ofthe signal spectrum generated by the microphone.U.S. Pat. No. 5,339,285 discloses a preamplifier for apiezoelectric sensor. The preamplifier is configured as a fullydifferential amplifier with its inputs coupled to the piezoelec-tric sensor and its outputs coupled to e.g. an analogue-to-digital converter. The preamplifier comprises a common-mode filter feedback configuration that in combination withthe capacitance of the sensor realizes a high-pass filter inte-grated with the pre-amplifier. This configuration is small insize, it has relatively low noise and its differential configura-tion makes e.g. silicon substrate noise (when the pre-ampli-fier is implemented on chip) appear as a common-mode sig-

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US 7,899,196B2nal which can be removed subsequently. However, thisconfiguration is not suitable since the gain of the amplifier isdependent on the microphone. Further, noise of the pream-plifier cannot be optimized independently of the microphoneelement.

SUMMARY OF THE INVENTIONThus, it is an objective of the present invention to providea preamplifier with the lowest possible input capacitance,lowest possible noise, largest output signal swing and at thesame time exhibiting the lowest smallest possible chip area.It is an objective of the present invention to provide a

preamplifier having a large power supply rejection and lowdistortion.It is an objective of the present invention to provide anamplifier which is able to handle slowly varying signals withrelatively large amplitude at its input terminal while at thesame time being able to amplify a low level signal with ahigher frequency with low distortion.It is an objective of the present invention to provide anamplifier which performance is very insensitive towards leak-age and parasitic couplings connected to the input.Additionally it is an objective to provide a digital outputsignal with low distortion.Further, it is an obj ective to provide a preamplifier configu-ration which can be optimized independently of the micro-phone circuit.There is provided an integrated circuit, configured to pro-cess microphone signals, where the integrated circuit com-prises: a preamplifier with an amplifier section which has afirst input and a second input and an output, and with afeedback filter network coupled between the output and thesecond input. The first input to the amplifier section has aninput impedance which by means of the input impedance ofthe amplifier section is substantially isolated from the feed-back network with respect to input impedance; and thepreamplifier has a frequency-gain transfer function whichsuppress low frequencies. Additionally it comprises an ana-logue-to-digital converter coupled to receive an anti-aliasingfiltered input signal and providing a digital output signal.The anti-aliasing filtered input signal is provided either byan anti-aliasing filter coupled to receive the output signalfrom the preamplifier or it is provided as a result of thefrequency-gain transfer function being configured as a band-pass filter where its upper stop-band prevents anti-aliasing.Irrespective of the implementation of the anti-aliasing, thepreamplifier is configured to provide a determined frequency-gain transfer function, which remains substantiallyunchanged irrespective of the frequency-impedance charac-teristic of a microphone circuit that is coupled to provide aninput signal to the preamplifier. This is an important improve-ment since often the design of the preamplifier and the micro-phone circuit, with the microphone element itself, yields con-tradictions. Especially, since the microphone element is amechanical component and often more difficult to controlwith respect to its electrical properties, the independence ofthe microphone element on the frequency-gain transfer func-tion is expedient. This applies both for condenser microphoneelements which are located as a unit separate from the chipcarrying the integrated circuit and for MEMS microphoneelements which are located as a micro mechanical portion ofthe MEMS device. Since the preamplifier typically has adifferential input stage (of an operational amplifier) a largehigh impedance is realized. This high input impedance is notdestroyed by the feedback filter, and consequently the ampli-fier does not load the microphone circuit.

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Further, inter-modulation distortion introduced by fre-quency components at low frequencies, outside the audioband, will be very low. The loop-gain characteristic providedby the feed-back configuration provides among other thingslower distortion.It should be noted that the preamplifier can be embodied asa single-ended amplifier or as a differential amplifier or adifferential difference amplifier or other amplifier with sev-eral inputs and outputs.However, in preferred embodiments the preamplifier isconfigured as a differential amplifier. Thus, expediently the

preamplifier is configured to provide a differential outputsignal by a first and a second amplifier section, where thepreamplifier has a differential mode transfer function whichcomprises a band-pass characteristic. The preamplifier com-prises a feedback filter network which establishes filter feed-back paths which couple outputs to respective invertinginputs of the amplifier sections and establishes a filter inter-connection path, which interconnects the inverting inputs.The differential configuration of the preamplifier providesgenerally a large common-mode rejection ratio and a veryhigh input impedance. Due to the configuration of the feed-back filter simple circuit control of the frequency-gain trans-fer function is achieved. Thereby, the transfer function can betrimmed or manipulated with a larger degree of freedom.Further, the differential configuration and the feedback filterin combination allows for utilizing the common-mode rejec-tion ratio in a frequency dependent way.Preferably, a lower cut-off frequency of the filter realized

by the preamplifier is located below the lower corner fre-quency of an audio band. Thereby, an expedient compromisebetween a downwardly broad audio band and a sufficientlyshort decay time of the impulse response of the microphonecircuit and the preamplifier in combination is achieved. Theshort decay time of the impulse response is expedient in thatthe effect of low frequent pulses, be it either from soundpulses or electrical disturbances, is reduced. Otherwise suchlow frequent pulses could overload the amplifier and subse-quent signal processing circuits and hence generate unpleas-ant non-linear distortion. In an expedient embodiment, thecut-off frequency is located about 10Hz.Expediently, the preamplifier has a differential mode trans-fer function which comprises a band-pass characteristic withan upper cut-off frequency located below half the samplingfrequency of the analogue-to-digital converter. Thereby anefficient implementation, with respect to chip area, of ananti-aliasing filter is provided by the preamplifier. Forinstance the sampling frequency may be about 2.4 MHz andan upper cut-off frequency of about 40-70 KHz may be cho-sen. Further, circuit control of the cut-off frequency, whichmay coincide with the band limit of the amplifier sectionsthemselves, is introduced.In an embodiment, the preamplifier has a differential modetransfer function which comprises a band-pass characteristic,which has a nominal pass-band and a gain plateau band,where the nominal pass-band extends over audio band fre-quencies and where the gain plateau band extends over fre-quencies above the audio band up to an upper cut-off fre-quency. Thereby, noise components, arising from eitheracoustic/mechanical sources or electrical sources, locatedabove the audio band is damped. Thereby it is possible tosafeguard the preamplifier from being overloaded by noisesignals and gain/amplitude effects of a resonance peak of themicrophone circuit. Such a peak may have an amplitude ofe.g. about 6 dB.The preamplifier can have a common-mode transfer func-tion which comprises a low-pass characteristic. Thereby, it is

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US 7,899,196B2possible to set the common-mode DC output level of thepreamplifier at the input of the preamplifier, while the com-mon-mode rejection ratio at audio band frequencies is uti-lized. Since the input impedance of the preamplifier is veryhigh it is possible to set the DC input level via a pull-upresistor with a very high ohmic impedance (e.g. 1-20 GOhmimplemented e.g. by CMOS transistors in their weak inver-sion mode and in linear region) which will not load the micro-phone circuit substantially.Further, the preamplifier can have a common-mode trans-fer function which comprises a stop-band characteristic, andwhere a flat gain response is provided for low frequencies.

Thereby, the DC input setting is preserved while the pream-plifier acts to provide a common-mode signal at two fre-quency bands: at low frequencies (below the audio band) andat high frequencies (above the audio band). Thereby efficientdamping ofundesired frequency components, with respect toaudio sound reproduction, is achieved. The stop-band willencompass the audio band.In an embodiment, the preamplifier has a common-modetransfer function and a differential mode transfer functionwhere the preamplifier is configured such that its common-mode gain prevails at low frequencies, whereas its differentialmode gain prevails at audio band frequencies.Further, the common-mode gain may prevail at frequenciesabove an upper cut-off frequency of the band-pass character-istic.In an expedient embodiment, a phase-shifter is cross-coupled between the output of a first amplifier section and aninput of a second amplifier section. This is an efficient con-figuration which ensures that the second amplifier sectionoperates at or close to 180 degrees out of phase with the firstamplifier section when a dominating differential mode gain isdesired e.g. at audio band frequencies. Additionally, thephase-shifter may be configured to control the DC level ofinput to the second amplifier section. This is achieved whenthe phase-shifter comprises a resistive path between the inputand output of the respective amplifiers.Alternatively, or additionally, a phase-shifter is coupledbetween respective inputs ofthe respective amplifier sections.This configuration is also capable of providing the 180degrees phase shift and optionally the DC level of input to thesecond amplifier section.Preferably, the preamplifier comprises a DC off-set circuitintegrated with the feedback filter to provide a DC shift at the

output ofthe preamplifier. This integration can be provided bya voltage divider coupled to aAC feedback resistor where thevoltage divider has substantial lower impedance than the ACfeedback resistor e.g. lower by a factor of about '/s, '/8 or '/io.Alternatively, the DC shift can be implemented by activecurrent sources.Further, a DC off-set circuit can be integrated with thefeedback filter to provide a differential mode DC shift at the

output of the preamplifier. The differential mode DC shift isdetermined by a difference in DC off-sets provided by firstand second off-set circuit. Thereby so-called idle mode tonesof the analogue-to-digital converter of the sigma-delta modu-lator type can be controlled. The location of the idle modetone is proportional to the differential mode DC shift (and aconstant determined by a sampler of the sigma-delta modu-lator and half of the sampling frequency).Preferably, the analogue-to-digital converter comprises asigma-delta modulator. The sigma-delta modulator providesnoise power spectrum which (in practical implementations) isdistributed with a flat and relatively low noise floor for lowerfrequencies, but with an increasing noise level above a corner

frequency. Since heavy over-sampling is applied the corner

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frequency appears well above the audio band. Expediently,the modulator provides a serial output signal.The sigma-delta modulator may comprise a switch-capaci-tor sampler, which samples the differential signal provided bythe preamplifier to provide a single ended input signal for thesigma-delta A/D conversion, and samples a DC voltage levelsuch that the single ended input signal is superimposed on thesampled DC voltage level. This provides for easier optimiza-tion of sigma-delta modulator since idle-mode tone controltakes place in the modulator. Since the sampler (and controlof it) is already available for sampling the signal from thepreamplifier the sampling of the DC level can be achievedwith only a slightly added complexity. Further, the preampli-fier is not loaded with common-mode DC overhead whichultimately reduces output AC swing.Preferably, the sampler comprises a summing amplifierwhich is an integrated portion of the sampler and the sigma-delta modulator loop. The sigma-delta modulator loop iswell-known to a person skilled in the art, but for complete-ness, it comprises an integrator filter of a given order which iscoupled to provide an integrated error signal to a quantizerthat quantifies the signal into discrete levels e.g. two, three orfour levels.Further, the summing amplifier may be provided with anintegration error feedback signal of the sigma-delta modula-tor via a first series capacitor and the DC voltage level isprovided to the summing amplifier via a second series capaci-tor. Thereby the idle mode tones can be controlled by the ratiobetween the values of the f irst and second series capacitor.The location of the idle mode tone is determined by theexpression

F,ai,=(VDco// rssfVmsss) (Csi/Cn) tzFswhere F,~is the location of the idle mode tone, C and Care the values of the fi rst and second capacitor, Fs is thesampling frequency, V~c +~is the sampled DC voltageand V~~F~~ is an internal reference ofquantizer in the sigma-delta modulator.When the analogue-to-digital converter comprises asigma-delta modulator, and when a DC off-set voltage levelinput to the sigma-delta modulator is chosen such that alow-frequent pulse input to and processed by the preamplifierprovides idle-mode tones above the audio band, a substantialreduction of non-linear distortion in a digital microphone isachieved. The DC off-set voltage level is provided by thepreamplifier as a differential mode DC signal or by the sam-pler as described above. The temporal duration of the pulseresponse of the combination ofthe microphone circuit and thepreamplifier is limited by the high-pass filter function of thepreamplifier, this further reduces the sensitivity to generationof idle mode tones.Additionally, there is provided a microphone comprisingan integrated circuit as set forth in the above and a condensermicrophone element configured to provide a microphone sig-

nal, responsive to a sound pressure on the microphone ele-ment, to the input of the microphone preamplifier. The con-denser microphone element may be a microphone with anelectret layer (ke. an electret condenser microphone, ECM) ora DC biased condenser microphone.Moreover, there is provided a microphone comprising anintegrated circuit as set forth in the above and a Micro ElectroMechanical System, MEMS, microphone element to providea microphone signal, responsive to a sound pressure on theMEMS microphone element, to the microphone preamplifier.

BRIEF DESCRIPTION OF THE DRAWINGThe invention will be described in more detail and withreference to a preferred embodiment, in which:

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US 7,899,196B2 10FIG. 1 shows a digital microphone comprising a micro-

phone element, a preamplifier with a high-pass filter function,an anti-aliasing filter and an analogue-to-digital converter;FIG. 2 shows a digital microphone comprising a micro-phone element, a preamplifier with an integrated band-passfilter function and an analogue-to-digital converter;FIG. 3 shows a microphone comprising a differentialpreamplifier with a filter function and a phase-shifter in a first

configuration;FIG. 4 shows a microphone comprising a differentialpreamplifier with a filter function and a phase-shifter in asecond configuration;FIG. 5 shows a first phase-shifter;FIG. 6 shows a second phase-shifter;FIG. 7 shows a shows a four-port high-pass feedback net-work providing in an amplifier a low-pass filter function;FIG. S shows a four-port band-stop feedback network pro-viding in an amplifier a band-pass filter function;FIG. 9 shows in greater detail a preamplifier with a differ-ential mode band-pass filter function and a common-modelow-pass filter function;FIG. 10 shows a preamplifier with a differential modeband-pass filter function and a common-mode low-pass filter

function, where a differential DC shift is provided;FIG. 11 shows a first frequency-gain transfer function of apreamplifier;FIG. 12 shows a second frequency-gain transfer functionof a preamplifier;FIG. 13 shows a differential preamplifier followed by aswitch-capacitor; sampler integrated with a sigma-delta con-verter;FIG. 14 shows a first configuration of a digital microphone;FIG. 15 shows a second configuration of a digital micro-phone;FIG. 16 shows a single-ended preamplifier with a filterfunction and an analogue-to-digital converter;FIG. 17 shows a schematic view of a microphone with anintegrated circuit and a microphone element;FIG. 1S shows a schematic view of a microphone with an

integrated circuit and a MEMS microphone element.DETAILED DESCRIPTION

FIG. 1 shows a digital microphone comprising a micro-phone element, a preamplifier with a high-pass filter function,an anti-aliasing filter and an analogue-to-digital converter.The microphone element Cm, 105 comprises a first memberin the form of a membrane or diaphragm that moves inresponse to a sound pressure acting on the membrane. Themembrane moves relative to a second member typically aso-called back plate or simply a microphone casing whichalso serves for holding the movable membrane. One of themembers, typically the second member is coupled to a groundreference whereas the other member, typically the membrane,is biased via a bias resistor R b, 104 that is coupled to a DCbias voltage V b. Thereby an electrical charge is provided onthe membrane or movable member of the microphone ele-ment 105, Cm. Since the amount of charge is kept constant(for very low frequencies and up), an electrical microphonesignal is provided by the membrane when it moves inresponse to a sound pressure acting on it. The microphonesignal caused by the movements of the membrane is super-posed on a DC signal caused by the biasing. The circuitcomprising the microphone element 105, the bias resistor 104and the DC blocking capacitor is comprised by the micro-phone circuit 107.

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The microphone signal is provided to the input, $, of apreamplifier 101 via a DC blocking capacitor 106 whichprevents the DC bias signal from reaching the input of thepreamplifier.The preamplifier 101 is characterised by having a high-pass gain characteristic with relatively low gain for frequen-cies below the audio band and a relatively high gain forfrequencies in the audio band. Preferably, the gain character-istic descents as a I",2, 3', 4'r higher order below theaudio band. Preferably the low-pass gain characteristic has acut-off frequency of about 10 Hz. In addition thereto theamplifier is characterised by processing a low frequencymicrophone signal as a common-mode signal and a highfrequency microphone signal as a differential mode signal.Thereby low frequency components are effectively sup-pressed. For an audio band with a lower corner frequency of20 Hz, low frequencies are frequencies below about 5-20 Hzand higher frequencies are frequencies above 10-30Hz.Output of the preamplifier 101 is provided as a differential

signal at output ports, $ and $*, to a sigma-delta A/D con-verter 103 via an anti-aliasing filter, AAF, 102. The sigma-delta converter provides an over-sampled I-bit output signaldesignated Do. The sigma-delta converter operates at a sam-pling frequency of about 2.4 MHz or greater.The digital microphone of FIG. 1 may be exposed tosounds that occur in its environment with a relatively largeamplitude and with a relatively steep amplitude slope rate inthe time-domain. Such sounds may originate from slamminga car door, dropping the telecom device on a table etc. and willgenerate input signals which comprise low-frequent signalswith a large amplitude. When the microphone is connected toits power supply the electrical circuit of the microphone willbe exposed to a step-like power supply pulse. This is alsodenoted a start-up or power-up pulse and it will likewisegenerate a signal comprising low-frequent (input) signalswith a large amplitude.In the analogue domain the bias resistor R b and thecapacitance of the microphone element forms a high-passfilter with an impulse response that has a relatively shortattack time, but a relatively long decay time. Consequently, avery low frequent pulse (considered almost DC) stays on atthe input of the preamplifier. Unless this impulse response ismodified, a severe distortion of the microphone's output sig-nal in the digital domain is the consequence.In the analogue domain noise considerations make it gen-erally desired to design the bias resistor Rmb to have a largeohmic value e.g. in the order of several hundred mega Ohmsto Giga or Tera Ohms. A typical capacitance value of themicrophone generates in combination with the large biasresistor a lower cut-off frequency of about 0.01 Hz. Thiscorresponds to a decay time of the impulse response which isin the order of several minutes. Thereby a large sound impulsegenerates aDC potential at the input ofthe preamplifier whichdecays, but prevails for several minutes. Further, since thepreamplifier provides a gain above 0 dB e.g. 6 dB this gainwould establish an even higher DC level at the output of theamplifier if it was not for the high-pass filter function of thepreamplifier.The high-pass filter function of the preamplifier introducesa lower cut-off frequency of about 10 Hz. This filter of thepreamplifier makes the combined impulse response of themicrophone element and the preamplifier decaying fastertypically faster than 0.5 seconds. Thereby, the input of thesigma-delta converter is not exposed to a slowly decaying DClevel, but only to fast decaying impulses.The sigma-delta converter provides an output signal, D~,in the digital domain. This output signal has a power spectrum

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US 7,899,196B2 12which comprises a flat noise floor. Above a noise cornerfrequency the noise level gradually increases with higherfrequencies. The nature of the sigma-delta converter makesthe converter sensitive to the DC level of the low frequentsignals described above in that dominant tones occur in powerspectrum of the digital output signal. These tones give asevere distortion ofthe reproduction of the sound signal in thedigital output signal. These idle mode tones must be con-trolled to provide good sound reproduction in the digitaldomain.FIG. 2 shows a digital microphone comprising a micro-phone element, a preamplifier with an integrated band-passfilter function and an analogue-to-digital converter. Themicrophone element 105 is comprised by the microphonecircuit 107 and operates as described above.The preamplifier 201 is configured as a band-pass filter

providing a gain of more than 0 dB in its pass-band. Thepass-band has a lower cut-off frequency ofabout 10Hz and anupper cut-off frequency located below half the sampling fre-quency of the sigma-delta converter 103. For a samplingfrequency, fofabout 2.4MHz the upper cut-o ff frequency ise.g. 40-70 KHz.Thus, the preamplifier is characterised by having a fre-quency-gain characteristic with a relatively low gain for fre-quencies below the audio band, a relatively high gain forfrequencies in the audio band, and a relatively low gain forfrequencies above the audio band. Preferably, the frequency-gain characteristic descents as a I",2, 3', 4', or higherorder below and above the audible range. In addition theretothe preamplifier is characterised by processing a low fre-quency microphone signal as a common-mode signal and ahigh frequency microphone signal as a differential mode sig-nal. Thereby low frequency components are effectively sup-pressed. Thus, the preamplifier implements a band-pass filter.In the following, the preamplifier is also designated a differ-ential preamplifier.The function of the band-pass filter amplifier is to suppresssignals at frequencies below the pass-band ie the audio bandto avoid overload of the input stage of the differential ampli-fier and to suppress frequencies above approximately half the

Nyquest frequency to avoid aliasing problems in subsequentsampling and digitalizing of the signal output from the band-pass filter amplifier. Thereby, a separate anti-aliasing filtercan be avoided.FIG. 3 shows a microphone comprising a differentialpreamplifier with a filter function and a phase-shifter in a firstconfiguration. The differential preamplifier 306 is shownwith an input terminal, $ , and output terminals $ and $*.Thepreamplifier is connected to the microphone circuit 107wherefrom it receives a microphone signal at its input termi-nal, $.The preamplifier is configured as an instrumentation-likeamplifier with a filter feedback network comprised by theimpedances Z1, Z2 and Z2* designated 305, 303 and 304 andwith a phase-shifter network 307.The impedance Z2, 303 iscoupled between the output terminal $ and the inverting inputof the operational amplifier 301.The impedance Z2*, 304 iscoupled as a like feedback for the operational amplifier 302.The impedance Z1 is coupled between the inverting and non-inverting input of amplifiers 301 and 302, respectively. Thefilter feedback network establishes a high-pass filter functionor a band-pass filter function ofthe differential preamplifier inits differential mode. The impedances Z1, Z2, and Z2* areimplemented by components, available for chip implementa-tion, which provides a capacitive or resistive behaviour or acombination thereof.

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The phase-shifter network PD(f), 307 has an input portdesignated (a) and an output port designated (b). The inputport (a) is coupled to the inverting input ofthe first operationalamplifier 301 and the output port (b) is coupled to the invert-ing input of the second operational amplifier 302. The phase-shifter network realizes a frequency dependent phase shiftshifting the phase about ~180degrees at high frequencies andabout 0 degrees at low frequencies. Thereby it is ensured thatthe differential mode output signal behaves as a commonmode signal at low frequencies whereas it behaves as a truedifferential mode signal at high frequencies. This phase shiftprovides an efficient high-pass filtering since the output sig-nal is forced to be a common mode signal at low frequencies,but only at low frequencies.This configuration of the preamplifier is expedient in thatthe input of the phase-shifter is coupled to the input of thepreamplifier via the inverting input of operational amplifier301.In an alternative configuration however, the input of thephase-shifter is coupled to the circuit node establishedbetween the microphone block 107 and the input of thepreamplifier (at the non-inverting input of the operationalamplifier 301.It should be noted that the feedback filters can be imple-mented by means of active filters and/or an active DC servo.FIG. 4 shows a microphone comprising a differentialpreamplifier with a filter function and a phase-shifter in asecond configuration. This configuration corresponds closelyto the one shown in FIG. 3, but the phase-shifter 307 iscoupled between the output terminal of the first operationalamplifier 301 and the non-inverting input terminal of thesecond operational amplifier.Thus, the phase-shifter 307 shown in FIGS. 3 and 4 and asdescribed in the above is cross-coupled between the twocommon mode amplifiers of the differential preamplifier.The amplifiers 301 and 302 are generally also designatedamplifier sections since they may comprise several amplifierstages.FIG. 5 shows a first phase-shifter. The phase-shifter 307 isrealized in combination with an operational amplifier toestablish a first order low-pass filter which provides a phaseshift of about 0 degrees at low frequencies and about 180degrees at high frequencies.The phase-shifter 307 forces the outputs, $ and $*,of thepreamplifier to act in common-mode at low frequencies andin differential mode at higher frequencies. Thereby, thepreamplifier, with the phase-shifter, network provides an efft-cient differential-mode suppression of low frequencies. Pref-erably, the cut-off frequency of this differential mode high-pass filter is located about 10Hz, but it could be located in therange up to about 30 or 50 Hz.FIG. 6 shows a second phase-shifter. In this embodiment,the phase-shifter 401 is realized in combination with anoperational amplifier to establish a band-pass filter whichprovides a phase shift of about 0 degrees at low frequencies,about 180 degrees at intermediate frequencies and about 0degrees at higher frequencies. Intermediate frequencies aredefined as frequencies comprising the audio band and fre-quencies up to an upper cut-off frequency of a band-passfilter. The upper cut-off frequency is typically designed to belocated about the pole which is introduced by the operationalamplifiers. Frequencies above this upper cut-off frequency isdenoted higher frequencies.The phase-shifter 401 forces the outputs, $ and $*,of thepreamplifier to act in common-mode at low frequencies and athigher frequencies. At intermediate frequencies the phase-shifter allows the preamplifier to act in differential mode.

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13 US 7,899,196B2 14Thereby, the preamplifier, with the phase-shifter, networkprovides an efficient differential-mode suppression of lowfrequencies and high frequencies. Thereby, an efficient anti-aliasing filter is additionally provided.The phase-shifter comprises a capacitor 603 and a resistor601 coupled in parallel and forming a signal path between theinput port (a) and the output port (b).A series connection of acapacitor 602 and a resistor 604 provides a signal pathbetween the output port (b) and ground.The phase-shifter PD(fj, irrespective ofwhether it is imple-mented as a first order or second order or higher order phase-shifter, ensures that the differential mode gain at low and highfrequencies, but not intermediate frequencies, is very low.The phase-shifter provides such a very low gain by introduc-ing at least one zero at 0 Hz (DC) and a pole, at a frequencyabove the audio band, in the transfer function of the pream-plifier filter.FIG. 7 shows a four-port high-pass feedback network pro-viding in an amplifier a low-pass filter function. The feedbacknetwork comprises ports (a), (b), (c) and (d). Feedback pathsfrom (b) to (a) and from (d) to (c)are established by capacitorsC2, 701 and C2*, 704, respectively.Between the ports (a) and (c) a series connection of acapacitor and a resistor is established by C1,703 and R1, 702.This series connection connects the inputs of the amplifiersections 301 and 302.FIG. S shows a four-port band-stop feedback network pro-

viding in an amplifier a band-pass filter function. The feed-back network comprises ports (a), (b), (c) and (d). Feedbackpaths from (b) to (a) and from (d) to (c) are established bycapacitor S02 coupled in parallel with resistor S01, respec-tively capacitor S05 coupled in parallel with resistor S06.Between the ports (a) and (c) a series connection of acapacitor and a resistor is established by C1,S04 and R1, S03.This series connection connects the inputs of the amplifiersections 301 and 302.However, it should be noted that other feedback filter con-

figurations can be provided comprising active filters and/orhigher order filters.FIG. 9 shows in greater detail a preamplifier with a differ-ential mode band-pass filter function and a common-modelow-pass filter function. The preamplifier 901 comprises anAC feedback network, corresponding to the one shown inFIG. S, and a phase-shifter network, corresponding to the oneshown in FIG. 5 in a configuration as shown in FIG. 4. Thefeedback network and the phase-shifter establish a differen-tial mode high-pass filter function. Integrated therewith, aDCfeedback network provides a common mode low-pass filterfunction, which provides a definite common-mode gain forDC and low frequencies. The preamplifier is configured suchthat the impedance of the DC feedback network dominates atDC (low frequencies) whereas the impedance of the AC feed-back network dominates at higher frequencies. Thereby theDC level at which the differential signal is provided can becontrolled with very limited circuitry while a desired AC filterfunction is realized.A reference DC voltage level, Vb, is applied to the input ofthe preamplifier which by means of its common-mode defi-nite DC gain establishes a definite DC level at which thedifferential signal is provided.The input, $, of the preamplifier 901 is coupled to themicrophone circuit 107 and a controlled DC voltage level isapplied to the input via a resistor Rb, 902 by means of avoltage source (not shown). This DC voltage level is set inaccordance with a desired common-mode DC output leveland the common-mode DC gain of the preamplifier. To thisend it should be noted that the power supply to a microphone

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typically is single ended i.e. between the nominal powersupply voltage level, V~~, and a ground reference. In order toprovide a symmetrical limitation of the AC output swing atdifferential-mode, the common-mode DC output level shouldbe close to half ofthe nominal power supply voltage. Thereby,a maximum differential-mode AC voltage swing is achiev-able. Special emphasis is given to this aspect as this limitationis the most restrictive for obtaining a large AC gain. Further,since noise in the digital microphone signal decreases as thepreamplifier gain increases, a large preamplifier gain (and inturn room for a large AC voltage swing) is preferred. It shouldbe noted that a larger AC signal swing improves signal-to-noise ratio since the noise from the sigma-delta converter isconstant in respect of different amplitudes.Since, typically, the input stage of a preamplifier saturatesif it is exposed to input DC levels which are relatively largecompared to the supply voltage, the output DC level must beachieved by a DC preamplifier gain ofmore than 0 dB.Iffor instance the nominal power supply voltage level is 1.5volts, the common-mode DC output level should be approxi-mately half of 1.5 volts, equal to 0.75 volts. For a powersupply voltage of 1.5 volts and a typical input stage of thePMOS differential pair type, the input should be able tohandle an input DC voltage level of up to approximately 0.4volts. In order to set the DC level at the output, the DC gainshould be at least about two times. A DC output level of 0.75volts and a DC gain of two times would require a DC refer-ence voltage Vb equal to 0375 volts which is less than 0.4volts.The preamplifier is implemented as a differential pream-plifier made from two operational amplifiers 903 and 904.The DC gain of the differential preamplifier is realized bya DC feedback network around each of the two operationalamplifiers. The DC feedback network senses the respectiveoutput signal by means of a voltage divider. The voltagedivider is implemented by means of resistors 906 and 907 forthe operational amplifier 903 and by means of resistors 909and 90S for the operational amplifier 904.The AC feedback network is configured as it is shown inFIG. S. Since however, it is integrated with the DC network,the resistive feedback path of the AC network comprises theresistors 910 and 906 and resistors 909 and 913 for amplifiers903 and 904, respectively. The capacitors 905 and 914 arecoupled in parallel with the resistive feedback paths. Thecapacitor 911 and the resistor 912 are coupled in seriesbetween the inverting inputs of the amplifiers 903 and 904.The phase-shifter is configured as shown in FIG. 5. Sincehowever, it is desired to integrate it with the DC network, itreceives its input from the circuit node established by thevoltage divider 906 and 907.At DC the impedance of the DC feedback network domi-nates. Thus, looking at a DC equivalence diagram, the voltageprovided by voltage divider 906, 907 is virtually identical tothe voltage fed back to inverting input of the amplifier 903.Consequently, the DC gain of amplifier 903 is determined bythe voltage divider. A DC gain of two times is achieved e.g. bytwo 100KOhm resistors. Likewise, the DC gain of the ampli-fier 904 is determined by the resistors 909 and 90S.This DCgain is selected to match that of the amplifier 903.Further, to force the amplifier 904 to provide an outputvoltage at the same level as that of the amplifier 903, theresistor 912of the phase-shifter is coupled to the circuit nodeestablished by the voltage divider 906, 907. Thereby, theoutput level ofamplifier 904 follows the output level ofampli-fier 903.As an alternative embodiment, the DC offset or DC biascan be introduced in the input stage of the amplifier 903 e.g.

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15 US 7,899,196B2 16by shifting the source voltage level (and consequently the gateand drain voltage) of a differential pair PMOS devices in theinput stage. This embodiment can be based on DC and ACfeedback networks similar to the ones described above. In thiscase Vb is connected to ground in order to set the DC biaslevel. It should be noted that combinations of the two con-f igurations is also possible.At AC the impedance of the AC feedback network domi-nates. Thus, looking at an AC equivalence diagram, the feed-back filter and the phase-shifter operates as described in con-nection with FIGS. 4, 5 and S.However, i t is recalled that thephase-shifter network comprising resistor 512 and capacitor513 is configured to provide a gradually shifting phase of thesignal input to the phase-shifter. This ensures that the pream-plifier provides a common-mode output signal at low fre-quencies, including DC, and a differential mode signal athigher frequencies, including the audio band and frequenciesabove the audio band.The phase shift between the one side, constituted aroundoperational amplifier 903, of the differential amplifier and theother side, constituted around operational amplifier 904 isimplemented by capacitor 916 and resistor 915. Thus, thephase shift is obtained by a phase shifter. It should be notedthat the resistor 915 in series with the capacitor 916 estab-lishes a pole (F~~) and a zero (F) confer the following.FIG. 10 shows a preamplifier with a differential modeband-pass filter function and a common-mode low-pass filterfunction, where a differential DC shift is provided. In theconfiguration of a preamplifier described in connection withFIG. 9, the common-mode DC output is provided by a DCvoltage reference provided either at the input terminal of thepreamplifier or at the input stage of one or both of the opera-tional amplifiers 903, 904. In combination therewith the com-mon-mode DC output is determined by the DC voltage ref-erence and the common-mode DC gain of the preamplifier.The preamplifier 1001shown in FIG. 10however, providesthe common-mode DC shift by means of two active currentsources 1001and 1002.Additionally, a differential-mode DCshif t is provided by a difference in the currents drawn by thetwo current sources. Since a sigma-delta converter is coupledto detect the differential signal provided by the preamplifier1001, the idle mode tones can be controlled by the differen-tial-mode DC shift provided in the preamplifier.When the DC shift is provided by the active current sources1001,1002 irrespective ofwhether it is a differential-mode orcommon-mode DC shift or both, the input terminal should beset to a reference level e.g.by coupling the input to ground viathe resistor 902 (he. V1=0Volts; however for practical imple-mentations Vb should be at least about 100mV). However, itis an option to apply another DC reference in combinationwith the active current sources.Iffor instance the power supply voltage V~~ is single-ended1.5 volts and the reference Vb is set to 0 Volts, a common-mode DC voltage of yiV~~=0.75 volts can be achieved bydrawing a DC current through resistor 906 which establish aDC voltage of0.75volts across it. Say resistor 906has a valueof 100 KOhms, a current of 7.5 pAmps will provide thedesired DC voltage at the output (output terminal $).Further, if it is desired, with respect to optimization of theidle mode tones, to establish a differential-mode DC shift of15mVolts and the resistor 909 has a value substantially equalto its counter part 906, a current of 735 pAmps will providea DC voltage at the output of the amplifier 904 (output termi-nal $*)of 0.735 volts. Thus establishing a differential-modeDC output of 15 mVolts as desired.From an AC point ofview, the phase-shifter, PD(fj is estab-lished by means ofresistor 1003 and capacitor 1004 as shown

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in FIG. 5 and it is embedded in the preamplifier as showngenerally in FIG. 3. Thus, the resistor 1003 is coupled to theinverting terminal of the operational amplifier 903 and to thenon-inverting input of the operational amplifier 904. Thecapacitor 1004 is coupled between the non-inverting input ofamplifier 904 and the ground reference.The preamplifier is thereby coupled to provide a common-mode DC level for maximization of the output AC voltageswing and a differential-mode DC level for control of idle-mode tones while at AC low frequency output signals areforced to appear as a common-mode signal and at higherfrequencies the output signal is forced to appear as a differ-ential mode signal.In an alternative embodiment, the differential-mode DCshift can be realized by means of a voltage divider coupledbetween the output terminal, $, and ground. The voltagedivider provides a divided output voltage at a circuit node towhich the resistor 1003 is coupled instead of being coupled tothe inverting input of amplifier 903.FIG. 11 shows a first frequency-gain transfer function of apreamplifier. The frequency-gain is shown for the preampli-fier as it is acting in common-mode (A~~=curve 1) and dif-ferential mode (A~~=curve 2). The response is shown as astraight line approximation on a logarithmic frequency axisand a logarithmic gain axis.Curve 2 illustrates the common-mode operation. From DCand up to the location of the pole F~,, a flat response isprovided. This flat response provides typically a gain of orabove 0 dB, but the amount of gain depends on whichembodiment of the DC off-set that is chosen and the desiredDC level. Above F~,, up to F, the response passes over to alower gain level. This lower gain level is flat above F,. Thiscommon-mode response is provided by the phase-shiftershown in FIG. 5 when it operates in combination with thepreamplifier.Curve 1 illustrates the differential-mode operation. At DCat least one pole establishes a positive slope of the transferfunction which continues up to the location of the pole F~,.Thereby differential mode DC signals are effectively sup-pressed. From the pole F~, to the pole F~~, a flat response isprovided. Preferably the audio band is comprised by the fre-quency range ofthis flat response. Above F~~ and up to F theresponse passes over to a lower gain plateau. The purpose ofthe gain plateau is to suppress noise sources above the audioband e.g. sound noise, electronic noise and to diminish gaineffects of resonance peaks of the microphone element. Thelevel of the gain plateau is determined such that the noisesources and gain effects do not limit the output swing (gain)of the preamplifier. At the pole F~3 the gain function begins todescent for higher frequencies. The pole F~3may be designedto be located about a pole introduced by the operationalamplifiers as such or the pose F~3 may be introduced by theoperational amplifiers as such.The series resistor C1, 702; S04 in the feedback networkintroduces the pole-zero pair (F~~, F).This pole-zero pair istypically located about 50-60 KHz and establishes the gainplateau from F to F~3 located about 500 KHz where theamplifier itself introduces at least one pole and thereby anegative slope.The audio band is illustrated by means of a lower cornerfrequency F~z and an upper corner frequency F~~.FIG. 12 shows a second frequency-gain transfer functionof a preamplifier. Again, the frequency-gain is shown for thepreamplifier as it is acting in common-mode (A~~=curve 1)and differential mode (A~~=curve 2).Curve 2 illustrates the common-mode operation. The com-mon-mode transfer function behaves like shown in FIG. 11,

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17 US 7,899,196B2 18but for frequencies above the transfer function zero F,.Above F, a positive slope begins. This ensures that thepreamplifier starts to act as a common-mode amplifier forfrequencies above F,. Thereby the suppression of signalcomponents in the upper stop band of the differential modeband-pass filter is further suppressed by the gradually prevail-ing common mode operation. This common-mode responseis provided by the phase-shifter shown in FIG. 6 when itoperates in combination with the preamplifier.Curve 1 i llustrates the common-mode operation. Thistransfer function illustrates a true band-pass filter function

without the gain plateau introduced by the series resistor R1,702; 803 in the feedback network of the preamplifier.FIG. 13 shows a differential preamplifier followed by aswitch-capacitor sampler integrated with a sigma-delta con-verter. The differential preamplifier 201 receives a micro-phone signal at its input $ and provides the dual outputs $ and$*.The signal from the preamplifier provided by means ofthese outputs is sampled differentially by means of a switch-capacitor detector integrated with the sigma-delta converter103.The switch-capacitor detector is build around an opera-tional amplifier 1301.The differential sampling is realized byan input series capacitor 1305 and a feedback capacitor 1304which are coupled between two circuit configurations bymeans of switches S1-S4.The input series capacitor 1305 is, at its input side, con-nected to the outputs $ and $*of the preamplifier by means ofrespective switches S1 and S2. The feedback capacitor 1304is coupled as a feedback path by means of switch S3. Theswitch S4 is coupled in parallel with the series connection ofthe capacitor 1304 and the switch S3.The switches S1-S4 are controlled to be either closed oropen according to the scheme shown in the bottom rightcorner i.e. switches S1 and S3 operate in unison and theswitches S2 and S4 operate in unison, but 180 degrees phaseshifted relative to S1 and S3. The switches S1-S4 are con-trolled by means of a clock frequency e.g. the sampling fre-quency of the sigma-delta converter. A switch-capacitor sam-pling ofdifferential signals is known to a person skilled in theart and will not be described in greater detail, but it is shownto illustrate interconnection of the differential preamplifierand the sigma-delta converter.It should be noted that the amplifier 1301 is coupled bymeans of capacitor 1303 to realize the summing amplifier ofthe sigma-delta feedback loop. A person skilled in the art willknow how a sigma-delta modulator in general is configured.The skilled person will know that the summing amplifiercompares the input signal to a feedback signal obtained fromthe quantizer that provides the digital output signal D~. Out-put of the summing amplifier is coupled to an integrator(irrespective of its order) which provides its output signal tothe quantizer. The feedback signal is provided to the summingamplifier 1301by means of the capacitor 1303.In addition to switched capacitor sampling of a differentialsignal, an embodiment ofa DC shift is realized. This embodi-ment of a DC shift is an alternative to the differential DC shiftprovided in the preamplifier and is configured to control idlemode tones of the sigma-delta converter.The DC shift can at this stage of the digital microphone beimplemented as a single-ended DC shift. It is implemented bysampling a DC voltage reference V~c +~by means of aseries capacitor 1306which is alternately coupled to eitherthe DC voltage reference or to a ground reference. Thecapacitor is alternately coupled by means of switches S5 andS6. The switching scheme for the switches S5 and S6 iscontrolled by a logic network coupled to the output of thequantizer in the sigma-delta converter.

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In this illustration an anti-aliasing filter is implemented bymeans of the upper cut-off frequency implemented by theband-pass filter of the preamplifier. It is required to removespectral components above a sampling frequency of the con-verter.Thus, the summing amplifier can be provided with an inte-gration error feedback signal of the sigma-delta modulatorvia a first series capacitor and the DC voltage level is providedto the summing amplifier via a second series capacitor.

Thereby the idle mode tones can be controlled by the ratiobetween the values of the first and second series capacitor.The location of the idle mode tone is determined by theexpressionFd=(VDc II,~~IV~~~~)~(C/30~C/3Q4)"/W~

where F,~ is the location of the idle mode tone, C$306 andC are the values of the first and second capacitor, F~ is the1304sampling frequency, V~c +~is the sampled DC voltageand V~~F~~ is an internal reference ofquantizer in the sigma-delta modulator.When the analogue-to-digital converter comprises asigma-delta modulator, and when a DC off-set voltage levelinput to the sigma-delta modulator is chosen such that alow-frequent pulse input to and processed by the preamplifierprovides idle-mode tones above the audio band, a substantialreduction of non-linear distortion in a digital microphone isachieved. The DC off-set voltage level is provided by thepreamplifier as a differential mode DC signal or by the sam-pler as described above. The temporal duration of the pulseresponse of the combination ofthe microphone circuit and thepreamplifier is limited by the high-pass filter function of thepreamplifier, this further reduces the sensitivity to generationof idle mode tones.FIG. 14 shows a first configuration ofa digital microphone.For a condenser microphone implementation, the digitalmicrophone is enclosed by a capsule 1401which encloses anintegrated circuit in the form of a chip 1402. The chip 1402comprises terminals Tcl, Tc2, Tc3, Tc4 for coupling to themicrophone element 1408, the bias voltage, a ground refer-ence potential and a supply voltage, respectively. TerminalTc6 provides the digital microphone output signal, D~, fromthe A/D converter. Via terminal Tc5 a clock signal is providedto the A/D converter. The power supply voltage to the ampli-fier 1405 and the A/D converter can be provided via theterminal Tc6, in which case the terminal Tc4 can be omitted.For the condenser microphone capsule implementation themicrophone element 1408 is a condenser microphone whichrequires a DC bias supply to provide the proper electricalcharge on one of the microphone members. This DC bias isprovided via resistor 1403.A DC blocking capacitor 1404prevents the DC bias level from reaching the input stage ofthepreamplifier 905. In an alternative embodiment the micro-phone element 1408 is an Electret Condenser Microphone,ECM. Thereby the microphone element 1408 is coupleddirectly to the input of the preamplifier 1405 and the biasresistor and the DC blocking capacitor is not needed.For a Micro Electro Mechanical System, MEMS, imple-mentation the digital microphone is implemented as aMEMSdevice which comprises a micro electro circuit portion and amicro mechanical portion, which implements the micro-phone element 1408.For a circuit perspective the microphoneelement changes circuit position with the DC blockingcapacitor. The micro electro circuit portion or the chip unit, asthe case may be, comprises a preamplifier 1405with a band-pass filter function and an A/D converter 1407.FIG. 15 shows a second configuration of a digital micro-phone. It should be noted that the below description applies to

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19 US 7,899,196B2 20an implementation of the digital microphone by means of achip and a condenser microphone. However, based on thedescription given in connection with FIG. 14 a person skilledin the art will be able to realize a MEMS implementation.The digital microphone 1501 comprises an integrated cir-cuit 1502with a DC voltage regulator 1503which provides aregulated voltage to the amplifier 1509 and the sigma-deltaconverter 1011.The microphone bias voltage is provided byan on-chip voltage up-converter 1504which receives an off-chip oscillating signal with a voltage amplitude; in responsethereto the up-converter provides an output oscillating signalwith a larger voltage amplitude. This output signal is low passfiltered by low pass filter 1505 and is provided via a seriesresistor 1506 to the microphone element 1508.A capacitor1507 blocks the DC bias voltage from reaching the input ofthe preamplifier 1509 with the transfer function mentionedabove. Output ofthe preamplifier 1509is provided to a sigma-delta converter 1511.The voltage up-converter or voltage pump, UPC, 1004 canbe in the form of a so-called Dickson-converter. The voltagepump is operated by an oscillator which preferably providesa square-wave oscillator signal to the voltage pump. Othersignals, eg sine waves or filtered square waves, with lowercontents of harmonics may be used to obtain lower noise. Inan alternative embodiment, the oscillator is embedded on thechip 1502.It is shown that the up-converter and the sigma delta con-verter shares the same oscillator/clock signal as provided viaterminal Tc4. It should be noted that the signal may be dividedto obtain different oscillating/clock signal frequencies to theUPC and sigma-delta converter.FIG. 16 shows a single-ended preamplifier and an ana-logue-to-digital converter. In this embodiment the micro-phone circuit 107 provides a signal to the single-endedpreamplifier 1602. The output of the preamplifier 1602 isprovided to an analogue-to-digital converter 103 of thesigma-delta modulator type.The preamplifier 1602 comprises an amplifier section 160.Preferably, this amplifier section is an operational amplifierwith a differential input. The amplifier section receives the

signal from the microphone circuit 107 on its non-invertinginput (+),whereas a feedback filter 1603 receives a the outputsignal from the amplifier section 1601 and provides a feed-back signal at the inverting input () of the amplifier section1601.The frequency-gain characteristic of the feedback filter1603 has a low-pass characteristic which in turn realizes a

high-pass filter characteristic of the preamplifier with thefeedback filter. Preferably, the pass-band (low frequencies) ofthe feedback filter provides a substantial flat gain responsetowards DC.And for higher frequencies, above its gain tran-sition band, a flat response for higher frequencies. This isillustrated by the feedback filter.However, the frequency-gain transfer function can be con-figured in line with the transfer functions of the differentialamplifier with the necessary changes.FIG. 17 shows a schematic view of a microphone with anintegrated circuit and a microphone element. The microphoneis shown as a cartridge with a microphone member, compris-ing the microphone membrane and an integrated circuit.FIG. 1S shows a schematic view of a microphone with anintegrated circuit and a MEMS microphone element. Themicrophone 1902 comprises a MEMS microphone member1903 integrated on a first substrate and the preamplifier cir-cuitry 1901integrated on a second substrate. The preamplifiercircuitry comprises one of the different embodiments dis-closed above i.e. comprising a preamplifier with a feedback

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circuit and e.g. a voltage pump and/or a feedback circuit,where the preamplifier is a differential amplifier or a single-ended amplifier.It should be noted that the MEMS microphone member1903 and the microphone preamplifier 1901can be integratedon a single semiconductor substrate.Generally it should be noted that the preamplifier can beimplemented as a single-ended amplifier, a differential ampli-fier or other amplifier types such as a differential differenceamplifier. In case preamplifiers with several inputs and out-puts are used different realisations ofmultiple feedback filterpaths are possible to realize a desired frequency transferfunction.A condenser microphone consists of a very light dia-phragm and back plate to which is applied a polarizing volt-age. Thereby a constant charge (for relevant frequencies isprovided). The principle of operation is that sound wavesimpinging on the diaphragm cause the capacitance between itand the back plate to change in sympathy. This in turn inducesan AC voltage on the back plate.The electret condenser microphone, ECM, operates in asimilar manner except that it has a permanent charge voltageimplanted in an electret material to provide the polarizingvoltage. This can be effected in three ways, the most commonbeing when the diaphragm is the electret material, in this caseone side is metalized. This is known as the foil or diaphragmtype. The electret material does not make the best diaphragmand where higher performance is required the diaphragm ismade ofother material and the electret material applied to theback plate. This is known as the back type. A more recentvariation is the so called front type. Here the electret materialis applied to the inside of the front cover of the microphoneand the metalized diaphragm connected to the input of thepreamplifier.The audio band can be defined to be any band within thetypical definition ofan audio band. A typical definition can be20 Hz to 20 KHz. Exemplary lower cut-off frequencies for anaudio bandcanbe: 20Hz,50Hz, 80Hz, 100Hz, 150Hz,200Hz, 250 hz. Exemplary upper corner frequencies of the anaudio band could be 3 KHz, 5 KHz, 8 KHz, 10KHz, 18KHz,20 KHz. By substantial flat is meant gain response variationswithin approximately +/ dB; +/3 dB; +/4 dB; +/ dB.However, other additional values of variation can be used todefine the term 'substantial flat'.In the above different preamplifier configurations havebeen disclosed. These configurations comprise differentinput/output terminal configurations e.g. a two-terminal con-figuration. However, it should be noted that three, four ormore terminals can be provided for input/output of signals tomicrophone and preamplifier. Especially, it should be notedthat separate terminals can be provided for supply voltage (ata first terminal) and preamplifier output (at a second termi-nal). In case of a differential preamplifier output two termi-nals for the output signals can be provided in addition to aterminal for power supply. A separate terminal is provided fora ground reference. This ground reference is typically, but notalways, shared by the power supply and output signal.The invention claimed is:1.An integrated circuit, configured to process microphonesignals, where the integrated circuit comprises:a preamplifier with an amplifier section which has a differ-

ential input comprising a first input (+) and a secondinput () and an output ($; $*),and with a feedback filternetwork coupled between the output ($ ; $*) and thesecond input ();where the first input (+)to the amplifiersection is coupled to an input ($))of the preamplifier forreceiving a microphone signal; and where the preampli-

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21 US 7,899,196B2 22fier has a frequency-gain transfer function which sup-presses low frequencies in a stop band relative to higherfrequencies in a pass band; and where the preamplifier isconfigured to provide a common-mode differential out-put signal in the stop band and a differential-mode dif-ferential output signal in the pass band; and

an analogue-to-digital converter coupled to receive the dif-ferential output signal, as an anti-aliasing filtered signal,from the preamplifier and to provide a digital outputsignal.2. An integrated circuit according to claim 1, where the

preamplifier is configured to provide a differential outputsignal ($, $*)by a first and a second amplifier section,where the preamplifier has a differential mode transfer

function which comprises a band-pass characteristic(A~~), andwhere the preamplifier comprises a feedback filter networkwhich establishes filter feedback paths (a-b; c-d) whichcouple outputs to respective inverting inputs of theamplifier sections, and which establishes a filter inter-connection path (a-c), which interconnects the invertinginputs.3.An integrated circuit according to claim 1,where a lowercut-off frequency (F~,)of the filter realized by the preampli-fier is located below the lower corner frequency of an audioband.4. An integrated circuit according to claim 1, where thepreamplifier has a differential mode transfer function (A~~)which comprises a band-pass characteristic with an uppercut-off frequency (F~~; F~~) located below half the samplingfrequency (F~) of the analogue-to-digital converter.5. An integrated circuit according to claim 1, where thepreamplifier has a differential mode transfer function (A~~)which comprises a band-pass characteristic, which has a

nominal pass-band (F~,-F~~) and a gain plateau band (F-F~~), where the nominal pass-band extends over audio bandfrequencies and where the gain plateau band extends overfrequencies above the audio band up to an upper cut-offfrequency (F~~).6. An integrated circuit according to claim 1, where thepreamplifier has a common-mode transfer function (A~~)which comprises a low-pass characteristic.7. An integrated circuit according to claim 1, where thepreamplifier has a common-mode transfer function (A~~)which comprises a stop-band characteristic (F,, F,-F,),and where a flat gain response is provided for low frequencies(DC-F~i ).S. An integrated circuit according to claim 1, where thepreamplifier has a common-mode transfer function (A~~)and a differential mode transfer function (A~~) which areconfigured such that its common-mode gain (A~~) prevails atlow frequencies (DC-F~,,) whereas its differential mode gain(A~~) prevails at audio band frequencies (F~-F~~).

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9.An integrated circuit according to claim 1,where addi-tionally the common-mode gain (A~~) prevails at frequen-cies above an upper cut-off frequency (F~~, F~~)of the band-pass characteristic.10.An integrated circuit according to claim 1, where aphase-shifter is cross-coupled between the output of a firstamplifier section and an input of a second amplifier section.11.An integrated circuit according to claim 1, where aphase-shifter is coupled between respective inputs () of therespective amplifier sections.12.An integrated circuit according to claim 1, where thepreamplifier comprises a DC off-set circuit integrated withthe feedback filter (Z1; Z1,Z1*,Z2) to provide a DC shift atthe output of the preamplifier.13.An integrated circuit according to claim 1, comprisinga DC off-set circuit integrated with the feedback filter andconfigured to provide a differential mode DC shift at theoutput of the preamplifier.14.An integrated circuit according to claim 1, where theanalogue-to-digital converter comprises a sigma-delta modu-lator.15.An integrated circuit according to claim 14, where thesigma-delta modulator comprises a switch-capacitor sam-pler, which samples the differential signal ($, $*)provided bythe preamplifier to provide a single ended input signal for thesigma-delta A/D conversion, and samples a DC voltage level(V~,~~) such that the single ended input signal is superim-posed on the sampled DC voltage level.16.An integrated circuit according to claim 15, where thesampler comprises a summing amplifier which is an inte-grated portion of the sampler and the sigma-delta modulatorloop.17.An integrated circuit according to claim 16, where thesumming amplifier is provided with an integration error feed-back signal of the sigma-delta modulator via a first seriescapacitor and where the DC voltage level is provided to thesumming amplifier via a second series capacitor.1S.An integrated circuit according to claim 1, where theanalogue-to-digital converter comprises a sigma-delta modu-lator, and where a DC off-set voltage level input to the sigma-delta modulator is chosen such that a low-frequent pulse inputto and processed by the preamplifier provides idle-modetones above the audio band.19.A microphone comprising an integrated circuit as setforth in claim 1 and further comprising a condenser micro-phone element configured to provide a microphone signal,responsive to a sound pressure on the microphone element, tothe input ($) of the microphone preamplifier.20.A microphone comprising an integrated circuit as setforth in claim 1 and further comprising aMEMS microphoneelement to provide a microphone signal, responsive to asound pressure on the MEMS microphone element, to themicrophone preamplifier.

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