rameters unchanged. The impedance characteristics of the antennafor different values of a (i.e., Xf) are shown in Figure 2(a).

The input resistance component of the resonant antenna Ra hasa simple dependence on the distance D. In the orthogonally prox-imity-coupled structure, the coupled resistance of the radiatingpatch per unit length is almost uniform along the feedline, and,therefore, the total coupled resistance can be easily controlled bysimply varying the distance D while keeping the feedline dimen-sions unchanged. As the distance D goes up, the input resistance Ra

increases and, therefore, the diameter of the �-shaped impedancelocus increases. The impedance characteristics of the antenna fordifferent values of D are shown in Figure 2(b).

As illustrated in Figure 2, the proposed antenna presents asimple and easy way to match its impedance to an arbitrary tagchip impedance. In the prototype antenna the distance D and theself-reactance of the feedline Xf were tuned to maximize the 3-dBreturn loss bandwidth with some margin for fabrication tolerances.The measured impedance locus is shown in Figure 2(a), whichagrees well with the simulated one. The measurement was carriedout with the antenna placed at the center of a 400 � 400 mm2

metal plate. Figure 3 shows the simulated and measured returnlosses of the antenna with respect to Zc. The measured 3-dB returnloss bandwidth is 44 MHz which is a little wider than the simulatedvalue of 35 MHz. From the result, we can see that the prototypeantenna totally covers the 26 MHz bandwidth requirement in

North America. Figure 4 shows the simulated radiation efficiencyof the antenna considering the ohmic and dielectric losses, and anefficiency of 28–66% is obtained in the 902–928 MHz band. Thesimulated directivity of the antenna on an infinite metal surface isabout 6 dBi. Figure 5 presents the measured co- and cross-polar-ization radiation patterns of the antenna placed at the center of the400 � 400 mm2 metal plate. The antenna shows good cross-polarization performance, with the cross-polarization level of �20dB.

4. CONCLUSION

A low-cost, wideband patch antenna using an orthogonally prox-imity-coupled feed has been proposed for RFID tags on metallicsurfaces. The antenna can be directly matched to an arbitrarycomplex impedance of a tag chip and has wideband characteristics.The prototype of the antenna was fabricated and measured, and 44MHz bandwidth at 3-dB return loss has been achieved. The an-tenna also shows good cross-polarization performance.

REFERENCES

1. M. Hirvonen, P. Pursula, K. Jaakkola, and K. Laukkanen, Planar in-verted-F antenna for radio frequency identification, Electron Lett 40(2004), 848–850.

2. L. Ukkonen, L. Sydanheimo, and M. Kivikoski, Patch antenna withEBG ground plane and two-layer substrate for passive RFID of metallicobjects, In: IEEE AP-S International Symposium, Monterey, USA, June2004, pp. 93–96.

3. W.-K. Choi, H.-W. Son, J.-H. Bae, G.-Y. Choi, C.-S. Pyo, and J.-S.Chae, An RFID tag using a planar inverted-F antenna capable of beingsuck to metallic objects, ETRI J 28 (2006), 216–218.

4. H.-W. Son, G.-Y. Choi, and C.-S. Pyo, Design of wideband RFID tagantenna for metallic surfaces, Electron Lett 42 (2006), 263–265.

5. D.M. Pozar and B. Kaufman, Increasing the bandwidth of a microstripantenna by proximity coupling, Electron Lett 23 (1987), 368–369.

6. H.-W. Son and C.-S. Pyo, Design of RFID tag antennas using aninductively coupled feed, Electron Lett 41 (2005), 994–996.

7. D.M. Pozar, Microwave Engineering, Addison-Wesley, Wokingham,1993, pp. 198–200.

© 2007 Wiley Periodicals, Inc.

STOPBAND-IMPROVED DUAL-MODEBANDPASS FILTER USING SIDE-SLITPATCH RESONATOR

Rui Li and Lei ZhuSchool of Electrical and Electronic Engineering, NanyangTechnological University, Nanyang Avenue, Singapore 639798

Received 8 August 2006

ABSTRACT: A rectangular patch resonator with four equal slitsetched on sides is presented to make up a stopband-improved dual-modefilter. By stretching these four side slits towards the patch center, thefirst resonant frequency gets a gradual decrease while its second onekeeps almost unchanged. This feature is attractive in widening upperstopband while miniaturizing overall size in filter design. To realize arelatively wide passband with DC-blocking, the narrowed feed lines aredeeply inserted into their respective slits, resulting in the enhanced ca-pacitive coupling degree. Our studies imply that the spacing between thefirst two resonant frequencies of this resonator itself is increased byabout 50% while the first spurious resonant frequency is fully rejectedby introducing a pair of open-circuited stubs in the feed lines. In final, adual-mode filter is optimally designed with a fractional bandwidth of4.5% at 1.60 GHz and predicted results are confirmed by experiment.

Figure 5 Measured normalized radiation patterns at 915 MHz. (a) x–zplane; (b) y–z plane

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 3, March 2007 717

© 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett 49: 717–720,2007; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22221

Key words: microstrip dual-mode filter; patch resonator; harmonic sup-pression; size miniaturization

1. INTRODUCTION

MODERN wireless communication systems require high perfor-mance bandpass filters with desired features, such as narrow band-width, miniaturized size, low transmission loss and harmonicsuppression. Since the concept of a microstrip dual-mode bandpassfilter was firstly presented in 1972 [1], extensive works have beendone so far to construct various types of bandpass filters byexciting the two degenerate modes of ring resonators [2–5] andpatch resonators [6–9]. By properly installing some perturbationelements such as cuts or stubs in either of diagonal planes, thesetwo degenerate modes can be simultaneously excited and they canbe employed as two resonators in filter design. Hence, the overallsize of dual-mode resonator filters can be dramatically reduced byabout 50% or more, thus making up a compact filter configuration.

Over the past several years, research works have been exten-sively made to improve several performances of ring resonatorbased dual-mode filters, such as harmonic suppression and sizeminiaturization [2–5]. In Ref. 2, the stepped-impedance ring res-onator and spur-line loaded feed lines are utilized together to putoff and suppress the lowest spurious resonant mode, respectively.Later on, four open-ended stubs [3] and four capacitive smallpatches [4] are etched in the four corners of a rectangular ringresonator to achieve simultaneous size reduction and harmonicsuppression, using their attractive slow-wave properties. In Ref. 5,a called periodic stepped-impedance ring resonator is proposedbased on the similar concept to explore a super small and wide-stopband dual-mode ring filter.

As an alternative resonator, a square patch with crossed slots isinitially proposed in Ref. 6 to make up a new type of dual-modefilters with high power-handling efficiency [7]. With the properchoice of two perpendicular crossed slots in length, this dual-modefilter can achieve simultaneous size and loss reduction. These twoaspects of performances get the further improvement by formingan inductively loaded cross-slotted patch resonator [8]. Recently,the work in Ref. 9 pays its attention on the existence of narrowupper stopband in the initial cross-slotted resonator [6, 8] andproposes a modified resonator configuration by rotating the crossslots in parallel to the two perpendicular feed lines so as to solvethis problem. Also, it should be pointed out that the filter structurein Ref. 4 is actually the same as that in Ref. 8 in geometry andprinciple.

In this article, a modified patch resonator with four equal slitsetched on the four sides is proposed to construct a new dual-modepatch filter with improved stopband performances. Its configura-tion is illustrated in Figure 2. Four outer-oriented slits in this patchfacilitates a tight coupling with external feed lines, so the consti-tuted dual-mode filter is capable of realizing the good DC-blockingbehavior [9] and widen the dominant passband [4] to a certaindegree. After a single-mode patch resonator is investigated to showits improved stopband behavior and increased Q factor, a dual-mode filter in Figure 1 is designed and fabricated. Predicted resultsare in final confirmed in experiment to exhibit improved upper-stopband and size-miniaturized performances.

2. SIDE-SLIT PATCH RESONATOR

Figure 2(a) shows the configuration of the proposed single-modepatch resonator with four identical slits, etched on the four sides

and stretched to the patch center. In this design, the width of fourslits is fixed at 0.6 mm and their length (S) is properly selected.The used substrate is the Duroid 6010.2LM with 50-mil thicknessand �r � 10.2 and the simulation is carried out using AgilentMomentum software. Intuitively speaking, the etched slits serve asthe similar purpose as the cross slots in Ref. 3 to perturb theelectric current flow on patch and lengthen its route for the fun-damental resonant mode (TM10), aiming at lowering the firstresonant frequency (f0). But, unlike the cross slots in Ref. 3, theyhardly affect the current flow of the second resonant mode (TM20).It is mainly attributed to the varied geometry that the horizontalslits are parallel to the electric current flow of TM20 mode whilethe vertical slits are placed at the symmetrical plane (that ismagnetic wall for TM20 mode) of this patch. This viewpoint can beevidently convinced, using the simulated frequency responsesshown in Figure 2(b). As the slit length (S) increases from 0, 4–8mm, the first resonant frequency (f0) drops off from 2.34 to 1.62GHz, whereas the second resonant frequency (f1) almost stays still.Hence, the concerned frequency ratio (f1/f0) gets a large increment.Figure 2(c) shows the extracted ratio (f1/f0) and unloaded Q or Q0

at f0 as a function of slit length (S). It can be seen here that f1/f0increases gradually up to 2.9 as S increases to 8 mm. In simulation,the Q0 factor evaluated with zero and actual loss tangents, i.e., tan� � 0 and 0.0025, to exhibit the radiation loss reduction andcumulative loss factor of the proposed resonator. As shown inFigure 2(c), the Q0 factor with tan � � 0 (no substrate/conductorlosses) rises up from 163 to 312, implying that the radiation lossget a dramatic reduction with S similar to the initial cross-slottedpatch case [6].

Next, let’s utilize a stub-loaded parallel coupled structure [8] atthe input/output ports to achieve an enhanced coupling degree at f0while suppressing a spurious resonance at f1 so as to make up astopband-improved and DC-blocking dual-mode filter. As de-picted in Figure 3(a), as the narrowed feed lines with the length (L)are deeply inserted into the respective slits, the magnitude of S21 atf0 is gradually increased from �27.6, �4.3 to �1.4 dB while itkeeps below �10.0 dB at f1. It implies that the above inquiredfactors are achieved to a certain extent. To further improve them,

Figure 1 Configuration of the proposed dual-mode bandpass filter

718 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 3, March 2007 DOI 10.1002/mop

open-ended stubs are placed in the feed lines in proximity to thepatch periphery to strengthen the coupling degree at f0 whileallocating the transmission zero of such an open-ended stub at f1.Figure 3(b) is readily plotted to confirm this property. As L � 9

mm is selected, S21-magnitide around 4.5–5.0 GHz is split into thetwo parts relying on the transmission zero, thereby reducing itsvalue from �15 to �25 dB at 4.8 GHz. As illustrated in Figure 1,this modified patch resonator can be used for dual-mode filterdesign if two ports are placed at the two orthogonal positions andperturbation elements are properly introduced.

3. DUAL-MODE PATCH FILTERS

To produce the two transmission zero in the skirts outside thedominant passband, the dual-mode bandpass filter in Figure 1 isconstituted by forming two square cuts along the diagonal planeB–B� instead of A–A�. Of course, two short stubs, placed in theA–A�, can reach the same purpose [3] as well. As well known, thecut dimension primarily control the coupling degree between thetwo degenerate modes and it can be adjusted to achieve thespecified passband performance. All the dimensions of the opti-mized filter are indicated in Figure 1. The predicted and measuredresults, as depicted in Figure 4, show an agreement with each otherquite well. With the help of Figure 2, we can further find that thedominant passband is tremendously reduced from 2.54 (normalsquare patch) to 1.60 GHz under the fixed patch dimensions of17.6 � 17.6 mm2. As a consequence, if the above two patches aredesigned with the same operating frequency (f0), the proposed

Figure 2 Side-slit patch resonator. (a) Configuration; (b) frequencyresponses; (c) frequency ratio (f1/f0), and unloaded Q factor or Q0 at f0

Figure 3 Frequency responses of side-slit patch resonator circuits. (a)Versus inserted feed line length (L); (b) versus stub length (t)

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 3, March 2007 719

patch filter can achieve a size reduction by 60%. Furthermore,Figure 4(a) allows us to observe that the spurious passband in the4.5–5.0 GHz range is fully suppressed, resulting in a widenedupper stopband. Figure 4(b) shows that the filter achieves the goodpassband behavior, with the insertion loss less than 2.2 dB andfractional bandwidth of 4.5%. In addition, the S21-magnitude in thewhole low range (DC to the dominant passband) is kept below�20 dB, showing a good DC-blocking performance.

4. CONCLUSION

In this article, a stopband-improved compact dual-mode bandpassfilter is presented by employing the side-slit patch resonator. Ourstudies at first confirm that the first resonant frequency is largelylowered while its second counterpart frequency keeps unchangedas the slits on four sides of a patch are introduced. As the stub-loaded feed lines are inserted into side slits, it is shown that the firstspurious passband can be effectively suppressed. After the pro-posed resonator circuit is investigated, a dual-mode filter using thisresonator is designed and fabricated. The filter is exhibited insimulation and measured to achieve several advantageous features,such as widened stopband, miniaturized size, DC-blocking, sharprejection skirts, and controllable dominant passband.

REFERENCES

1. I. Wolff, Microstrip bandpass filter using degenerate modes of a mi-crostrip ring resonator, Electron Lett 8 (1972), 163–164.

2. U. Karacaolu, D.S. Hernandez, I.D. Robertson, and M. Guglielmi,Harmonic suppression in microstrip dual-mode ring-resonator bandpassfilters, IEEE Int Microw Symp Dig San Francisco, CA (1996), 1635–1638.

3. A. Gorur, A novel dual-mode bandpass filter with wide stopband usingthe properties of microstrip open-loop resonator, IEEE Microw Wire-less Compon Lett 12 (2002), 386–388.

4. S.-W. Fok, P. Cheong, K.-W. Tam, and R.P. Martins, Microstrip dual-mode bandpass filter design with simultaneous size reduction and spu-rious response suppression, IEEE MTT-S Int Microw Symp Dig 4(2005), 2175–2178.

5. J.-T. Kuo and C.-Y. Tsai, Periodic stepped-impedance ring resonator(PSIRR) bandpass filter with a miniaturized area and desirable upperstopband characteristics, IEEE Trans Microwave Theory Tech 54(2006), 1107–1112.

6. L. Zhu, P.-M. Wecowski, and K. Wu, New planar dual-mode filter usingcross-slotted patch resonator for simultaneous size and loss reduction,IEEE Trans Microwave Theory Tech 47 (1999), 650–654. 7

7. A. Cassinese, F. Palomba, G. Pica, and A. Andreone, Dual modecross-slotted filters realized with superconducting films, Appl Phys Lett77 (2000), 4407–4409.

8. L. Zhu, B.C. Tan, and S.J. Quek, Miniaturized dual-mode bandpassfilter using inductively loaded cross-slotted patch resonator, IEEE Mi-crowave Wireless Compon Lett 15 (2005), 22–24.

9. W.-H. Tu and K. Chang, Miniaturized dual-mode bandpass filter withharmonic control, IEEE Microwave Wireless Compon Lett, 15 (2005),838–840.

© 2007 Wiley Periodicals, Inc.

LINEAR TUNABLE DISPERSIONCOMPENSATION DEVICE USINGSELECTIVE STRETCHING IN CHIRPEDFIBER BRAGG GRATING

R. Romero and O. FrazaoINESC Porto—UOSE, 4169–007 Porto, Portugal

Received 17 July 2006

ABSTRACT: A linear tunable dispersion compensation device basedon a chirped grating (CFBG) mounted on a stretching mechanical struc-ture is described. The tunability is obtained changing the CFBG band-width without modification of the total length structure. © 2007 WileyPeriodicals, Inc. Microwave Opt Technol Lett 49: 720–722, 2007;Published online in Wiley InterScience (www.interscience.wiley.com).DOI 10.1002/mop.22245

Key words: tunable dispersion compensation; chirped gratings

1. INTRODUCTION

Chromatic dispersion is one of the basic characteristics of anoptical fiber, and can be described as the effect that with slightlydifferent frequency travels down the fiber at a slightly differentspeed. This phenomenon has been studied for a number of years asit is one of the major impediments for signals propagating throughlong distances. Although it is possible to manufacture fiber withzero chromatic dispersion, it should be emphasized that such fiberis incompatible with the deployment of wavelength division mul-tiplexing (WDM) systems, since nonlinear effects would becomeimportant. As long as WDM is employed, chromatic dispersionmust exist, and therefore must be compensated.

Figure 4 Comparison of simulated and measured S-parameters for theproposed dual-mode bandpass filter. (a) Over a wide frequency range(0.1–5.0 GHz); (b) over a narrow frequency range (1.0–2.2 GHz)

720 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 3, March 2007 DOI 10.1002/mop