the continuous-cathode (emitting-sole) crossed-field amplifier · 2015-07-29 · the backward-wave...

330 PROCEEDINGS OF THE IEEE, VOL. 61, NO. 3, MARCH 1973

[lo21 S. V. Yadavalli, “On the performance of a class of hybrid tubes,” Proc. IRE (Corresp.), vol. 42, p. 263, Feb. 1960.

[lo31 A. J. Lichtenberg, “Prebunched beam traveling-wave tube studies,” IRE Trqns. Electron Devices, vol. ED-9, pp. 345-351, July 1962.

[lo41 A. D. LaRue and R. 5. Rubert, “Multi-megawatt hybrid TWTs at S-band and C-band, presented to the IEEE Electron Devices

[lo51 T. Roumbanis, “Centipede Twystron amplifiers and traveling wave Meet., Washington, D. C., Oct. 1964.

tubes for broadband, high-efficiency, super-power amplification,” in Proc. 7th Int . Conf. on Microwave and Optical Generation and Am- blifrcation (Hamburg. Germanv. SeDt. 1 6 2 0 . 1968): see also _ _ Nachrichtenkch. Faciberichte, vol. 35; pp. 110-114, VDE Verlag

11061 M. R. Boyd, R. A. Dehn. J. S. Hickey, and T. G. Mihran, ‘The GmbH 1 Berlin 12, 1968.

. . multiple-beam klystron,” ZRE Trans. Electron Devices, vol. ED-9,

(1071 G. M. Branch et al., “High-power traveling-wave multiple-beam pp. 247-252, May 1962.

klystron,” Final Rep., General Electric Co., Tech. Rep. ECOM- 00007-F, Oct. 1967.

[lo81 W. J. Pohl, “The design and demonstration of a wide-band multiple-beam traveling-wave klystron,” IEEE Trans. Electron De-

[lo91 A. Staprans, “A study of space-charge wave propagation in periodic vices, vol. ED-15, pp. 351-368, June 1965.

electron beams,” Ph.D. dissertation, Dep. Elec. Eng., Univ. of

ASTIA Doc. AD 212224. California, Berkeley, Jan. 1959; also WADC Tech. Rep. 59-160,

[110] R. H. Pantell, “Backward-wave oscillations in an unloaded wave-

[l l l] R. B. Dyott and M. C. Davis, “Interaction between an electron guide,” Proc. I R E (Corresp.), vol. 47, p. 1146, June 1959.

beam of periodically varying diameter and EM waves in a cylindrical guide,” IEEE Trans. Electron Devices, vol. ED-13, pp. 374- 376, Mar. 1966.

[112] R. M. Phillips, “The Ubitron, a high-power traveling-wave tube based on a periodic beam interaction in unloaded waveguide,” I R E Trans. Electron Devices, vol. ED-7, pp. 231-241, Oct. 1960.

[113] C. E. Enderbyand R. M. Phillips, “The Ubitron amplifier-A high- power millimeter-wave TWT,” Proc. IEEE (Corresp.), vol. 53, p. 1648, Oct. 1965.

The Continuous-Cathode (Emitting-Sole)

Crossed-Field Amplifier

JOHN F. SKOWRON

Invited Paper

Absfract-The crossed-field amplifier (CFA) is an outgrowth of the magnetron. Several varieties exist including the forward-wave/ backward-wave types, and the injected-beam continuous-cathode (emitting-sole) types. Of these, the continuous-cathode type has found the most usage and has been established in many systems. Emphasis is placed upon this type. The CFA interaction process has similarities to a synchronous generator. From this, many of the characteristic features can be deduced or understood. The size, efficiency, range of performance, phase stability, and operating parameters are a feature of the crossed-field interaction process which distinguishes the CFA from other microwave tubes. Its greatest usage is in lightweight transportable radar systens, although not restricted to these. Its future lies in the extension of its gain capability and in the incorporation of technological advances such as new materials.

A I . INTRODUCTION

S AN ESTABLISHED member of the microwave tube family, the crossed-field amplifier (CFX) has found a place in applications where power per unit weight and

volume and high efficiency are important. These features have allowed the CFA to be used in a variety of systems ranging from low-power high-reliability space communications to multimegawatt high-average-power coherent pulsed radar.

While the CFA is a well-known member of the tube family, much of its underlying performance capability is not as well

The Editor. This incited p a p n is one of a series planned on topics of general intnest-

Manuscript received September 29, 1972; revised November 27, 1972. The author is with the Microwave and Power Tube Division, Ray-

theon Company, Waltham, Mass.

known, nor is its usage. This appears to stem from the fact that the crossed-field interaction process is sufficiently complex tha t a reduction to a good analytical form from which to make accurate performance predictions is not possible, even though a good qualitative understanding exists. Design approaches and appraisals of performance capability therefore tend to rely heavily upon empirical data rather than upon calculations. Certain phenomena such as the secondary emission capability of a cold cathode produce surprising performance attributes for the CFA, yet this would be difficult to forecast analytically because of the complications of integrat- ing this factor into the rest of the CFA interaction process. Other desirable attributes such as high gain and efficiency together have not materialized, even though this could rea- sonably be expected. The CFA has therefore not realized its full potential as a microwave device, which means only tha t more work is needed for a full understanding of i t .

The CFA contains elements in common with other microwave tubes, such as slow-wave circuits, cathodes, microwave windows, matching transformers, vacuum envelopes, etc. Much has been published on each of these topics, particularly where they are amenable to good analytical examination, so it does not seem fruitful to repeat or summarize them as they apply to the CFA. What is less known is performance capability in the finer features of performance and the knowledge gained in recent field experience. Finally, i n most reviews of the CFA, equal attention is given to the varied concepts, novel approaches, and variations on the basic device. Yet the

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SKOWRON: CROSSED-FIELD AMPLIFIER 33 1

practical significance currently tends to lie in only a few versions that have survived the test of time and have established themselves as useful and available devices. The discussion that follows therefore places most emphasis on those technical areas that are of current significance and on information that is not readily available in prior published form. I t does not at- tempt a full and comprehensive survey of the entire topic because of its scope.

11. HISTORICAL BACKGROUND

A . Technical Evolution

The CFA is the outgrowth of an evolutionary sequence of high-frequency generators which started in 1924 with the split anode magnetron [l]. This device produced oscillations in the ultrahigh-frequency region. I t was a negative resistance type of oscillator that required a static magnetic field oriented crosswise to a static electric and radio-frequency electric field interacting with electrons to produce a negative resistance.

The split anode magnetron was followed by the cyclotron frequency magnetron oscillator. This type depended upon the coincidence of the cyclotron frequency of electrons in a magnetic field to the radio frequency of an electric field superposed upon a static electric field between concentric cathode and anode. Efficiency was not high, but microwave performance was obtained, and a feature of all crossed-field interaction, the selective removal of noncontributing electrons from the interaction region, was noted. These were driven toward the negative electrode, the cathode.

The cyclotron oscillator was followed by the traveling- wave magnetron oscillator which became the microwave power source for most radar systems in the early forties. The R F electric field of the traveling-wave magnetron was primarily transverse to the dc and magnetic fields rather than superposed between the anode and the cathode. The electrons, on their trip from the cathode to the anode, traveled tangen- tially with the R F field and “locked-in” with a traveling tangential component wave of t h e R F field [2, pp. 24-29, 2701. The lock-in feature arose from the removal of nonproductive electrons, leaving only the useful ones, phased favorably, to continue their trip to the anode. The electron bunches so formed traveled at the same, or synchronous, velocity as the R F traveling wave and finally dissipated their kinetic energy on the anode. By selecting the parameters of the tube so that tight cycloidal motion prevailed within the bunches, or “spokes,” the synchronous kinetic energy dissipated on the anode could be made small in relation to the dc potential applied to the anode and the amount of energy transferred to the R F signal. The result was surprisingly high dc to R F conversion efficiency. The magnetron achieved electronic conversion efficiencies well above 60 percent and ultimately reached 90 percent a t D band.’

The reliable, consistent, and noncritical performance of the traveling-wave magnetron led to consideration of the crossed-field interaction concept for use as an amplifier. At the time, the recently developed linear beam traveling-wave tube (TWT) suffered from the shortcoming of low efficiencv. al-

use the continuously emitting cathode of the magnetron and arrange an input and output isolated from each other. The second was to introduce an injected beam into the interaction region by means of a special crossed-field injection gun [3]. In most of this early CFA work, conditions for gain and regeneration in other modes dominated and the devices oscillated. The backward-wave mode2 of oscillation was identified even if other resonance induced oscillatory conditions were not present. The backward-wave mode was tunable with respect to the anode-sole3 voltage. A useful device evolved in the form of the M-type backward-wave oscillator (M-BWO) [4] which was electronically tunable over the entire operating frequency band, a feature the magnetron did not have. On this base [SI the entire family of M-BWO devices were developed and put into service as rapidly tuned generators for electronic counter- measures (ECM) applications.

The first practically successful CFA was introduced in 1953 [6]. This device operated in a circular configuration similar to the traveling-wave magnetron,. except that the microwave circuit was broken electrically to provide input and output connections while the electron stream continuity re- mained uninterrupted. The operation was in the backward- wave mode and the frequency was in the passband region of a strapped-vane network, a structure typical of a magnetron. The success of this concept, in contrast to earlier approaches to a CFA, resided in two main attributes-the range of oscillation-free operation when R F driven was sufficient to be practically useful without the introduction of mode suppressing attenuation, and the interaction efficiency equaled o r exceeded that of the traveling-wave magnetron. Its bandwidth was comfortably 10 percent and its phase stability very good, making it a natural candidate for the final stage in an amplifier chain for a radar transmitter. Its principal deficiency was its gain which was 10 dB. Higher gain was possible, but not without the penalties of reduced bandwidth and stability.

In an effort to improve the utility of the CFA, the forward- wave emitting-sole amplifier with a drift space for reentering electrons and a cutoff electrode was developed [7] . This approach capitalized on a feature observed in magnetrons and in the backward-wave amplifier-that the cathode emission could be supplied entirely by a cold secondary emitter. With cathodeanode voltages applied, the CFA could be started by the RF dr ive signal. This permitted consideration of methods to shut off emissions by collecting electrons after the R F drive signal was discontinued. In this concept, the forward-wave feature permitted the tube to be operated from a constant dc voltage power source. The cutoff electrode permitted modulation of the tube with a low-energy modulator. Finally, a bandwidth design restriction was eliminated since the drift space removed the phase information on reentering electrons. Effi- ciency was good, reaching 60 percent a t D band.

In recent years, the forward-wave continuous emitting- sole tube type has been extended to achieve RF turnon and turnoff or full R F keyed performance, and the injected-beam type of CFA (IB-CFA), similar in configuration to the M- BWO, has been developed for special purposes.

though it displayed other attractive lfeatures such as good 2 The U b a c b r d - w v e refers to an electric field configuration gain and bandwidth. on the anode microwave slow-wave structure having a traveling compo-

There were two concepts under consideration, one was to nent with phase velocity opposite the main energy travel. A feedback of energy between circuit and electron beam is thus possible, creating an oscillation whose frequency is determined by the beam velocity.

a The sole refers to the negative electrode of the CFA. It may or may In this paper, frequency band designations follow the new letter not be used as the electronic emission surface depending upon whether or

_ I

code. not separate beam injection is used.


332 PROCEEDINGS OF THE IEEE, MARCH 1973


S K O m O N : CROSSEDFIELD AMPLIFIER 333

wide application in transmitters requiring high peak and average power with bandwidths on the order of 10 to 15 percent. The method of pulsing and the variation of electrical properties over the frequency band for a given application influence the selection of forward- or backward-wave mode of operation. This type of CFA has also been used in CW applications. I t has been used with frequency modulation or with no modulation.

The injected-beam CFA has found its principal application in relatively low peak power, wide bandwidth uses characteristic of ECM usage. With a broad-band forward-wave circuit, performance capability approaching an octave bandwidth and power in the kilowatt range is feasible. The efficiency of the injected-beam tube has not approached that of the continuous-cathode type, remaining at approximately half that typically obtained with continuous-cathode tubes. Although high peak power designs of the IB-CFA have been reduced to practice and have been shown to be feasible [13], they have seen much less application than has the CC-CFA. For this reason, the discussion that follows places emphasis upon the continuous-cathode emitting-sole type and deriva- tions of it.

Work on the basic CFA's has been performed to extend their performance and to enhance their features with respect to competing devices. The effort has been in two general areas -improvements in the methods of pulsing, and extensions of the power-frequency capability.

In the modulation or pulsing area, several variations in the method of beam injection into the interaction area have been investigated. These include axial injection [I41 using an electron gun similar to that of a magnetron injection gun used in beam tubes to create hollow beams and in voltage tunable magnetrons to control the current. The beam enters the interaction region with a spiraling motion synchronized with the phase velocity on the circuit and is collected on the opposite end of the tube. Other variations include combinations of injected beams and emitting soles. At this time, the degree of successful practical utilization of these approaches has been limited.

Of more immediate and continuing interest, however, is the concept of low-energy modulation of a CFA. In the ideal case, the concept of self-pulsing has attracted considerable interest. This attractive feature springs from the natural property of crossed-field interaction in which, during the interaction process, some of the electrons acquire energy from the R F field and bombard the cathode. In so doing, new electrons are created through the process of secondary emission, which results in a cumulative buildup of emission to levels that are large. Indeed, in many cases, the entire current of the tube is produced by such secondary emission without any thermionic or primary electrons. In this process, it has been established that emission can be initiated with R F drive power alone. That is, with full dc voltages applied to a tube having a non- thermionic cold cathode, no current will flow until R F energy is introduced into the microwave structure. When this occurs, some ions in the tube will produce free electrons that are ac- celerated by the R F drive signal fields, and their bombardment of the cathode triggers off the emission process. Very clearly, then, the turnon of R F interaction and power can be initiated exclusively by the driving R F signal. I t has been observed that, with a proper arrangement of emission properties, R F levels, circuit interaction impedance, and other design features, the current flow can be made to stop when the R F drive signal is removed [lj]. By this method, the tube and its

amplification process, and therefore its power output, can be entirely controlled by the incoming signal. CVith a tube design having a forward-wave circuit for which the phase velocity and thus the tube voltage are substantially constant over the operating frequency band, one has the very desirable feature of a device that is passive except when a suitable R F drive or input pulse is applied. I t t hus requires no modulation circuitry with its associated energy loss.

The feature of R F keying in the device is achieved a t the expense of efficiency and requires a somewhat critical combination of the design features listed in the foregoing. Achiev- able values of efficiency are approximately 30 percent, and noncritical operation occurs a t gains below 10 dB.

With respect to extensions of power and frequency, the technical approach has been to increase the interaction surface area by paralleling interaction structures and working out the signal driving technique, the matching, and the multimode problem that invariably results [16], [17]. This approach is needed because the general design of the C,FA is rather critically limited by the microwave circuit and its size and cooling capability. I t is not possible to increase its electrical length indefinitely because of accumulation of insertion loss and the resultant reduction of efficiency. Of the several attempts to make a significant increase in power-frequency performance, none have been fully successful in reaching their objectives, mostly because of problems arising from multimode interference.

111. DESIGN APPROACH: INTERACTION CONCEPT As with all microwave tube types, the interaction or

energy conversion process of the CFA involves the motion of electrons and the method by which energy is acquired from the dc power sou'rce and transferred to the RF signal. In the CFA, the interaction process differs from that of linear beam types, such as the TLi'T, in that the electrons make the conversion of dc potential energy to RF without acquiring much kinetic energy from the dc field in the interim.

In linear beam interaction, the electron stream is first ac- celerated by an electron gun to the full dc velocity; the dc velocity approximates the phase velocity of t h e R F wave on the circuit. After interaction occurs, the spent electron stream leaves the interaction region with a lower average velocity. The difference in velocity is accounted for by the RF energy created on the microwave circuit (see Fig. 3).

In the CFA, the presence of the magnetic field causes electron flow to be generally transverse to the dc fields along dc equipotentials. After synchronism is reached, the electron stream is then capable of traveling through the dc potential field without significant increase in velocity and produces a conversion of energy directly to the microwave field on the circuit. The spent beam strikes the anode of the tube at approximately the synchronous velocity (see Fig. 4).

The events that occur in the interaction process, in their fundamentals, reduce to the movements of electrons while under the influence of several forces. In the CFA, the electron is exposed to the dc electric field force, the magnetic field force, and the electric field force of t h e R F field. Its accelera- tion and resultant motion is a self-consistent pattern arising out of these three forces. A fourth force, the space-charge force from other electrons, also exists. I t has the effect of de- pressing the potentials, skewing the spokes, and producing some random bunching of electrons. This, in turn, causes the true spoke to depart from an oversimplified ideal form. In the direction orthogonal to the principal plane of the electron



KINETK ENERGY CONVERTED TO RF

AT COLLECTOR KINETK ENERGY

CATHODE DISTANCE - t

ACCELERATING ANODE

COLLECTOR

ENERGY DlSTRlWlON DIAGRAM

Fig. 3. Physical schematic and energy distribution diagram of a TWT.

-I++ PHYSICAL SCHEMATIC

/ POTENTIAL ENERGY CONVERTED TO RF

I I /

M I N I M U M KINETIC

f” I STANCE ___c

CATHODE ANODE

ENERGY DISTRIBUTION DIAGRAM

Fig. 4. Physical schematic and energy distribution diagram of a CFA.

motion, the space-charge force tends to eject electrons out of the interaction region. This force is normally not considered in analytical approaches because of its complexity. In practice, this force is neutralized through the use of appropriate static electric or magnetic field shaping to confine the charge to the region of interest.

Under the influence of the three forces, the electrons travel in spiral trajectories in a direction tending along equipotentials. The ’exact motion has been subject to much analysis by means of a computer. However, the “simplest” case, t h a t of the magnetron, has yet to be solved with enough rigor and freedom from approximations to produce accurate predictions

of performance. A good description of the interaction process can be obtained from consideration of the derivation of the Hartree equation.

The Hartree equation is an expression giving the dc voltage between the anode and the emitting cathode under conditions allowing the electrons to leave the cathode at zero energy and reach the anode at a specified “synchronous” energy, having accomplished the transit while locked into a fixed transverse velocity. The general characterization is that of a spoke of charge locked into synchronous angular motion.

The Hartree expression has been derived in numerous references [2, pp. 196-1981, [lo, pp. 323-3321, [18]. A rearrangement of terms of the expression helps to display its physical significance:

for cylindrical coordinates, where

f RF frequency, B magnetic field, k mode number or number of space-charge spokes, r. anode radius, re cathode radius, m electron mass, e electronic charge.

With the following successive substitutions:

1 = r, - r,

0, = raw

angular velocity of spoke,

spoke length.

mean radius of spoke,

mean spoke velocity,

velocity of spoke at anode radius,

algebraic rearrangement yields

l m 2 e

v = Blv, - - - ua2.

This expression can now be subjected to a simple physical interpretation with the aid of Fig. 5 . The first term‘ is the “back EMF” induced into a conductor cutting magnetic flux lines while moving a t a mean velocity v,. The second term is the kinetic energy, in electronvolts, of an electron striking the anode at radius r,, and velocity u,,.

The Hartree voltage is derived under the assumption that the electrons are confined to a “spoke” or “conductor” traveling at constant angular velocity o. The difference between the behavior of an electron confined to a conductor and one which is free is that the confined electron can move only in one dimension down the length of the conductor, whereas the free electron can execute cycloidal motion in two dimensions. The inclusion of the second term in the expression accounts for this difference, since the electron must start from rest at the

of radial forces acting on the electron. I t is also the well-known 8 = E / B relationship which defines a balance


SKOWRON : CROSSED-FIELD AMPLIFIER 335

Fig. 5. Schematic diagram showing a space-charge spoke moving angularly between anode and cathode.

cathode and reach the spoke angular velocity through a cycloidal trajectory. In short, the crossed-field interaction process has the features of a synchronous ac generator. The driving force for the spoke, or “rotor,” is the dc field. The retarding force is t h e R F field, and the voltage drop across the spoke, cathode-to-anode, is the “back EMF” produced by cutting magnetic flux lines. As long as synchronism is preserved, one would expect the performance of both devices t o be orderly and predictable. If synchronism is disrupted for any reason, one would expect both devices to ‘unlock” and performance to deteriorate.

In most tube designs, the second term is small in relation to the first term because it represents wasted energy. For an efficient design, the second term must be about one-tenth as large as the first term. This means that the gross behavior of the crossed-field tube depends mostly on the first term. From this, one can immediately deduce the following four general properties of the crossed-field tube.

1) The plate voltage required for operating the tube will be proportional to the magnetic field.

2) I t will be proportional to the anode-cathode spacing, or spoke length.

3) I t will be proportional to the R F frequency (contained in the mean spoke velocity).

4) I t will be inversely proportional to the number of spokes, or the mode number.

Items 2) and 4) are dimensional features built into the tube and cannot be changed externally. Items 1) and 3) are externally controllable.

In forward-wave types, the circuit is so designed that the frequency and the mode number-items 3) and 4)-vary in proportion to one another. The phase velocity is thus constant. In such a case, the plate voltage will remain constant when frequency is varied. In magnetrons, with a fixed mode number, and in backward-wave devices, with variable phase velocity, the plate voltage will vary in proportion to frequency. In both types, the voltage will be proportional to magnetic field.

The Hartree expression identifies the lowest dc voltage at which the first electron reaches the anode while constrained to synchronism with a traveling wave on the circuit. It is the point at which current would first be expected t o flow as the voltage is increased. Note that the assumed radial velocity is zero and that all kinetic energy contained in the first electron is transverse and equal in magnitude to the synchronous energy. Note also that the dc voltage or electronvolt energy may be an order of magnitude higher than the transverse kinetic energy, depending on choice of design parameters. This distinction forms the basis of the intrinsically high efficiency of the CFA process.

The maximum energy available for conversion to R F is the dc voltage minus the synchronous voltage. The upper limit to efficiency is

V d c - V synchronous

v d c

which occurs in the limiting case of negligible current and can be made arbitrarily high by selecting an operating point high in relation to the synchronous voltage. The penalty for this lies in increased voltage gradients and the risk of voltage breakdown if carried to extreme, and in the need for a higher magnetic field.

The presumption up to this point is that no radial energy of motion is present in the sy’nchronous electron. In practice, when finite current flows and finite power is generated, radial electron flow must occur. The ,onset of current therefore co- incides with the need for radial motion and an additional component to the velocity, which, in turn, introduces an additional component of loss at the anode. The added loss reduces the efficiency. The general trend is therefore for a declining efficiency with increasing current and power output.

If the motion of electrons in response to the RF fields is examined, it will be found that this motion is along lines of R F equipotential. Analysis of this motion has been made by various simplified approaches and by computerized techniques. If the problem is shifted to moving coordinates, it has the effect of deleting the Hartree dc field needed to produce tangential flow. In these coordinates, the form of the electron flow can be visualized as shown in Fig. 6 . Note that the spoke is polarized to match the RF field bopndary and, therefore, can be located only in the retarding side of t h e R F cycle. Had it been positioned in the opposite phase, its transverse polarity would not have matched the R F field. This is the region in which electrons are driven back to the cathode.

The moving coordinate system displays the important point that radial electron flow cannot occur without tangential electric fields in the interaction space. If current, in fact, is flowing, it is an indication that R F energy exists on the anode. In the CFA driven from an independent external R F drive source, the current will be influenced by the drive energy and will affect the slope of the V-I characteristic.

The distribution of current within a spoke has been measured in an analog tube test vehicle having conditions imposed which simulate current flow in moving coordinates. Such a distribution is shown in Fig. 7 , which clearly shows the phase of the current spoke with respect to the R F voltage. From such data it is possible to calculate the efficiency of interaction for various conditions of R F voltage, magnetic field, dc voltage, and current.

The contours of performance which occur in consideration of the factors discussed in the foregoing can be plotted on voltage-current coordinates to display the characterization of efficiency and other properties. A typical case is shown in Fig. 8 plotted in normalized fashion to a set of parameters called VO, Bo, 10, and PO. Note that the region of high efficiency is a t high voltage and low current. Contours of maximum current, as derived from the secondary emission ratio 6 for the cathode material, are also shown. Note that a low-voltage high-power operating point requires a high secondary ratio emitting material for the cathode. The normalization factors are defined for cylindrical coordinates as follows:

l m 2 e

vo = - - ra2 (Ty Authorized licensed use limited to: Princeton University. Downloaded on December 8, 2008 at 15:31 from IEEE Xplore. Restrictions apply.


/

Fig. 6. Sketch of operating CFA rotating coordinate system.

Av = v -v a h

0

Fig. 7. Typical voltage and current distribution in moving coordinates (simulated in analog test vehicle with stepped voltages).

I n = - l - l

Po = IOVO

where all parameters are defined as before except for the addition of a, which is a multiplier equal approximately to unity, and h, which is the axial length of the interaction space. cis the free-space dielectric constant [2, pp. 41-17].

The physical interpretation of these parameters is quite straightforward and convenient to use in the design of a CFA for a particular performance requirement. VO is the electronvolt equivalent of the synchronous energy at the anode radius.

Bo is the magnetic field needed to make the Hartree voltage equal to VO. This is a condition where no spoke can exist. The electron executes one-half of a cycloid after leaving the cathode and intersects the anode at grazing incidence. I t co- incides with the Hull cutoff condition5 [I91 for a static cylindrica! magnetic diode for the specific field BO.

10 is the current that would be expected to flow when VO and Bo are applied to the diode. I t is somewhat less than the nonmagnetic space-charge-limited current that would flow and has been calculated by Allis to allow for the presence of the magnetic field.

5 The Hull cutoff condition is given by the expression

or

v = - - , l e 8 m .zBz [l - (1)*]'

= vo [$I' in normalized form.


SKOWRON : CROSSED-FIELD AMPLIFIER

$ = ANODE/CATHODIDIMENSION*LP*TIO = 1 . 8 4 c

0 0 1.0 2.0

I/ I, 3.0

Fig. 8. performance chart derived from analog test vehicle data.

Po is normalized power and is a measure of the size of the tube insofar as its power generating capability is concerned.

A . The Microwave Circuit The selection of the microwave circuit for the CFA is

dominated by the form factor and the suitability for cooling, because the circuit serves the dual function of electron collector and slow-wave network. Additional requisites are sufficient bandwidth, sufficient impedance, sufficient freedom from velocity coincident high-impedance interfering modes, and dimensional stability under power. The general study of periodic lines has evolved a large assortment of types, both forward wave and backward wave, for the various kinds of microwave tubes [20], [21]. When the constraints for CFA usage are imposed on these, they reduce to only a few practically suitable types, especially when appreciable power is in- volved.

In the forward-wave category, the requirement for good bandwidth and flat dispersion limits the selection to some version of the helix or of the meander line. The adaptation of either of these types to the CFA form factor usually will compromise either the bandwidth or its power handling capability because of the necessity to deform the basic circuit to accommodate either conductive or liquid cooling. The helix-coupled vane and its counterpart, the stub-mounted helix, have provided satisfactory compromises by retaining the good electrical features of the helix and providing cooling through the vane or stub. Ceramic-mounted circuits have provided a method for conduction cooling through the ceramic and are particularly suited to the higher frequencies.

In the backward-wave category, the strapped bar line represents a satisfactory compromise of both requirements.

337

Other backward-wave circuits, such as the interdigital line, are electrically satisfactory but more difficult to cool. A variation of the interdigital line, the split-folded waveguide, provides thermally conductive paths to the back wall and the top and bottom of the circuit, which makes i t a desirable circuit for higher frequencies.

IV. CAPABILITIES AND PERFORMANCE FACTORS A . Secondary Emission

The feature of the CFA which segregates favorable from unfavorable electrons in the interaction process and creates the spokelike pattern of space charge also causes some electrons to be driven toward the cathode or the negative electrode. The energy of these bombarding electrons is substantial, amounting to about 5 to 10 percent of the total dc input energy. This is sufficient, in many cases, to heat a primary emitting cathode and maintain thermionic emission without the use of or need for heater power. This energy represents a loss and detracts from the total efficiency.

However, in cases where a cold secondary emitter is used, this energy loss then makes a CFA, in its properties, quite distinct from other microwave tubes. I t creates secondary electrons emanating from the negative electrode. In the injected-beam CFA, these additional electrons are objectionable because they upset the beam optics and the performance of the tube to such an extent that special precautions are needed to inhibit emission from the sole, either by grooving the metal to trap the secondaries or by special coatings. In the continuous-cathode CFA, these additional electrons can provide the major (in some cases the entire) source of current for the tube and may be critically needed for good performance [26].

The cathode current in the latter case can be triggered or turned on by the R F drive signal if dc voltage is present and if the cathode is cold and has no thermionic emission as discussed earlier. This means that the turnon of the tube can be caused entirely by the turnon of the RF drive, independently of the application of dc voltage. I t also means that the total available current will not follow the rules and limitations of primary emitting cathodes, and that the stress imposed upon the cathode as measured by bombardment power is a major factor in the selection of cathode material for life and reliability. Thus the total available current in a CFA will be a property of the cathode secondary emission ratio as well as a property of the interaction space design.

A useful expression that predicts the maximum current to be expected from a given secondary emitting material and geometry in cylindrical coordinates is

I 1.61 X le6(& - 1) h -- v3/2

-

[l - ctyI’rt+ 11

- 1.

where

I anode current in amperes, V anode voltage in volts, 7, cathode radius, r. anode radius, h length of cylinder, 6 secondary emission ratio.

All physical constants are lumped into the numerical constant. This expression is derived on the basis of space-charge-

limited current flow and has the familiar 3/2-power relationship. In practice, it is found to be quite useful in identifying Authorized licensed use limited to: Princeton University. Downloaded on December 8, 2008 at 15:31 from IEEE Xplore. Restrictions apply.


2.0

3.0

2.0

6

1.0

0

' I

0 1 1 vp Wi

0 400 800 I200 1600 2003 2400 2800 32W 3600 4CUI

Fig. 9. Secondary emission ratio for platinum as a function of primafy energy.

0 e = 00 (NORMAL INCIDENCE) u e : m O

e : 700 A 8 : I K ) O

TARGET WATER1 AL: ROW TMPERAWRE, A IR OXIDIZED ALUINUM

500 lo00 1 5 0 0 2ooo Vp (VOLTS)

Fig. 10. Secondary emission ratio versus primary energy for aluminum target for various angles of inadence.

design limits for a given cathode material. Furthermore, it gives the correct voltampere relationship. The current is dependent upon the voltages; it will continue to rise with voltage until some other restriction, such as arcing, causes a limitation. This is in contrast to the current available from a thermionic emitter which will present a roughly constant current upper limit that is independent of voltage.

The maximum current available is seen to be dependent upon the secondary emission ratio of the cathode material, the interaction space area, the cathodeanode spacing, and the choice of operating voltage. For a given power output, the design choice for operating voltage is strongly dependent upon the secondary emission ratio of the cathode material if the physical dimensions are independently constrained by such factors as frequency. For a low-voltage design, a high ratio is needed.

Secondary emitters fall into two broad categories, metals and oxides [23]. The oxides of barium, aluminum, thorium, magnesium, beryllium, etc., produce secondary emission ratios

that are relatively high, that is, more than 2.5. Secondary emission ratios of the pure metals are relatively low, tha t is, below about 1.9. Of the materials suitable and available for vacuum tube use, the best of the metals, platinum, has a maximum measured ratio of 1.85. While oxides can reach values as high as 20, a more important issue is the resistance of the oxide to the stress of electron bombardment. The means to preserve the oxide, particularly under high-duty-cycle operation, is an integral and dominating factor in the tube design.

The general character of the secondary emission ratio is shown in Figs. 9 and 10 as a function of bombardment energy. The ratio peaks up broadly at a few hundred volts and then trails off gradually. Both the level of the peak and its location vary with the material, but the peak is rarely above 1500 V. The secondary emission ratio is also a function of the angle of incidence of the bombarding electrons. Measurements indicate an increasing value with angular incidence, as shown in Fig. 10. The implication of the secondary emitting nature of materials is, first, that at low bombardment energies, say below


SKOWRON: CROSSEDFIELD AMPLIFIER 339

several hundred volts, the ratio drops off rapidly and sufficient emission is not possible. This is corroborated, in practice, by observing that CFA’s intended for operation below roughly 10- kW peak require primary emitters to insure dependable cathode turnon and sufficient emission. Second, one would expect that some upper limit in power would exist, since the ratio drops off a t high values of bombarding energy, and that there would be an offsetting factor by the angle of incidence in the CFA resulting from cycloidal motion and cathode impinge- ment not always normal. In practice, no firm upper limit has been found with platinum, at least up to 20-MW peak power output a t 150-kV cathode-anode potential. One can conclude that there is a self-correcting phenomenon that insures that the bombardment energy never becomes excessive. This factor could be the space-charge density that may adjust itself to produce conditions suitable for the required current. In magnetrons with primary emitting heated cathodes, it has been noted that the back bombardment energy, as measured by efficiency changes, will drop when heater power is applied and vice versa. There appears to be a broad self-regulating effect that insures enough bombardment to produce the necessary emission.

In the CFA, the turnon of current can be initiated by the turnon or application of R F drive, that is, t h e R F field on the anode triggers the emission. The phenomenon is initiated by the ionization of residual gas in the tube followed by accelera- tion of the free electrons by the R F field resulting in bombardment of the cathode and the initiation of emission. The cumulative effect may be likened to multipactor, except that, with a crossed magnetic field, trajectories cannot retrace themselves and accumulate in the usual manner associated with multipactor. However, with a reentrant device, this could be expected to repeat on a one revolution around basis and cumulate. In any case, there is clearly a buildup time or time constant associated with turnon of emission following the application of R F drive. The factors that bear on the buildup time are a t least the R F drive magnitude, the secondary emission ratio, the dc operating voltage, the R F frequency (as a scale factor), and the gas pressure in the tube. If any one of these is too low, the emission will not buildup a t all, or there will be a substantial time required for the buildup.

While it is difficult to assign numbers individually to each of these factors because of their interaction with such indi- vidual tube design parameters as impedance, bandwidth, etc., i t has been observed empirically that if the RF input is more than 10 kW, the dc voltage more than 15 kV, the delta 1.8, the frequency D band or higher, and gas pressure 10-8 mmHg, then the turnon delay is not likely to be greater t h a n 10 ns and the time jitter will be less than 3 ns. Conversely, with R F input on the order of 1 kW or lower, turnon delays of micro- seconds can occur, depending upon the mix of the other factors. Finally, a t about 100 W, i t is unlikely that any combination of parameters will achieve turnon. A design intended for this type of operation must use a primary emitter.

Even when conditions are adjusted deliberately to create a large delay in emission buildup, the leading edge of the current pulse is observed to be jitter-free. The time jitter is less than 3 ns and probably as low as 1 ns. This feature is shown in Fig. 11. Observations of current pulses for examination of buildup and jitter have been made with cold oxide cathodes, cold platinum cathodes, and with both forward-wave and backward-wave tubes. The results have been uniformly good and have been limited entirely by the stability of the instru-

TIME (20nr/div) - Fig. 11. Oscillogram showing the leading edge of an R F drive and

output pulse on a dc operated forward-wave CFA.

mentation and power sources. However, misinterpretation of the buildup process can occur if, during the pulse buildup time, the current passes through oscillating modes which themselves have oscillation startup jitter.

A dependence of current from secondary emitting cathodes upon the pulse duration has not been observed. Once the current starts, there is no decline or decay as a function of time. This contrasts sharply with primary emitters whose peak emission may decline during the pulse. The secondary emission process is stable and unchanging with time. This leads to stable phase characteristics over the pulse duration. Test information exists on CW and pulsed tubes up to 500 ps, on forward-wave and backward-wave tubes, and on platinum and beryllium oxide cathodes.

In addition to the insensitivity of the emission to pulse duration, if the cathode is cooled well enough there is also an insensitivity to duty cycle. The secondary properties of most materials do not change significantly below 400 or 500°C. If the operating temperature of the cathode is maintained well below this level by liquid or other cooling, then there is no sensitivity to duty cycle, and high- and low-duty operating modes can be freely interchanged to provide versatile performance.

B . Oscillation Buildup as Compared to &he Driven Amplifying Case

As in any microwave amplifier, the gain, bandwidth, dynamic range, etc., of the CFA are functions of the field configuration and relative magnitude in the interaction region. The design is tailored for this mode pattern. However, the intended beam-circuit interaction may be only one of hundreds of interaction possibilities distributed throughout the frequency spectrum, each of which could be encountered under the same conditions of voltage and current desired for the intended mode. Instability and interference can result from un- wanted excitation of these modes. In practice, it is necessary to introduce into the device some means for discrimination in favor of the desired mode and suppression of the others. For- tunately, the process of optimizing the conditions for desired mode operation will have the effect of de-emphasizing, or suppressing, most other interfering possibilities, so that the problem usually consists of dealing with one or a few specific modes whose starting conditions approximate those of the amplifying mode.

Of the many modes present in the microwave circuit, the most troublesome usually are those in the same passband a t or near the cutoff frequency of the circuit. At these points, the interaction coupling impedance is high and the losses are low because of practical difficulties in achieving a good match at


340

the band edges. Fig. 1 indicates common problem areas for a forward-wave and a backward-wave circuit as displayed on the omega-beta diagram.

Difficulties with cutoff oscillation tend to be somewhat worse with the forward-wave circuit than with the backward wave circuit because of the similarity of phase velocity, hence dc voltage, of the cutoff frequency to that of the operating frequency. The backward-wave circuit has a natural phase velocity and voltage separation that improves the discrimination between the oscillating mode and the amplifying mode.

Oscillation at cutoff has the general character of a magnetron oscillation, except that it will not necessarily be precisely at the pi-mode. I t may be a t t he first resonance into the passband which is created by the imperfect junction match. Also, the power generated will be split between the input and the output ports in a ratio determined by the gain, the magnitude of the junction reflections, and the direction of interaction. The input port may be an exit for substantial power at the oscillating frequency, requiring that the input RF circuitry and components be capable of withstanding this power from the view of R F breakdown and average power dissipation at the oscillating frequency.

An oscillating mode can be looked upon as an amplifying mode with feedback whose gain is high enough to produce oscillation. I t has a set of running conditions, such as a particular voltage, frequency, efficiency, and dynamic range, similar to those of an amplifying mode, but with one excep- tion. I t has a buildup time associated with the time needed to store energy in the feedback system which can be stated in terms of a Q.

The buildup time provides a useful means to discriminate against, or totally avoid, oscillation in pulsed tubes by using a fast voltage rise on the leading edge of the applied pulse. T o guarantee reliable starting on each pulse, an oscillating magnetron must have a rise time slow enough to encompass the oscillation buildup time as prescribed mostly by the frequency and the Q. The exact converse is true of the CFA. Since the CFA amplifying mode is a driven mode and is not dependent upon feedback, its buildup time is on the order of a transit time through the tube, or a few multiples thereof, which is shorter than the response of most circuitry attached to the CFA. The CFA is not the limiting factor. A fast rise, therefore, is advantageous because it will inhibit any interfering oscillation but will have no effect upon the amplifying mode.

A fast fall at the end of the pulse is also advantageous be- because the dwell time a t a possible synchronous oscillation voltage is reduced. With dc operation and pulsed R F drive, the oscillating conditions are less likely to arise because the anode-cathode voltage never traverses the oscillation value.

The incremental voltage above the Hartree voltage is the additional energy needed to produce a radial flow of electrons and therefore operating current. The slope of this voltage as a function of current is usually referred to as the dynamic impedance of the Gauss line, the voltagecurrent characteristic for a fixed magnetic field. The precise operating voltage of the CFA will then depend upon this slope and can influence the separation of modes. The loading presented to a modulator when current flows will depend upon this slope and, in turn, will influence such things as transients and pulse ringing. Finally, the voltage separation between the operating point and a competing mode will be determined by the corresponding Gauss-line slope of the competing mode.

PROCEEDINGS OF THE IEEE. MARCH 1973

\ I LOW DRIVE 148kWI \ IGH DRNE (5% LW) H Q EQUIVALENT)

SUPPRESSED BY HIGH DRNE LOWER MODE NOT EXISTENT,

CIRCUIT CUTOFF FREQUENCY LOWER VOLTAGE MODE AT

35 Y d, = .005 f = 3000 h4Hr

30 1 , 0 IO M 3 0 4 0 5 0 6 0 70

ANODE CURRENT (amp)

Fig. 12. QKS622 V-Z characteristics under high-drive and low-drive conditions.

The slope of the Gauss line is a direct function of the RF voltage present and on the circuit. This can be demonstrated on a magnetron by a Rieke6 [2, pp. 40-421 diagram pulling test and a comparison of the Gauss-line slopes for loaded and unloaded test conditions. The counterpart of this in the CFA is a measurement of the slope under conditions of high and low R F drive power. The heavily loaded case for the magnetron and the low drive case for the CFA are similar and result in low R F voltage for a given power on the circuit and, in turn, high-dynamic Gauss-line impedance. The lightly loaded magnetron resembles the high-drive low-gain case of the CFA and produces a low dynamic impedance (Fig. 12). The performance of these tubes in a given modulator can therefore be influenced by the external R F conditions imposed upon them.

C. Power Distribution Input to Output Since the current in a CFA impinges upon the anode, which

is also the microwave circuit, and on the cathode i n the form of back bombardment, the average power, peak power, and pulse duration capabilities of a tube are interrelated with the design and cooling of these elements. The distribution pattern of power dissipated on the anode and cathode is thus important. The distribution of energy from input to output, and the consequent current distribution, have been approximated by various calculations. These show a generally rising value from input to output, with the precise pattern varying according to the approximations employed in the analysis. The precision of these calculations is insufficient to account for some observable features, and, again in practice, it is safer to depend upon measurements if the power density in any one region is be- lieved to be critical.

The factors under consideration are the average power density from the view of temperature rise and cooling method, and the peak power density in combination with the pulse duration from the view of vane surface temperature rise and erosion. Both of these density factors are fully dependent upon the power distribution throughout the circuit structure.

The power density distribution has been measured by

6 Polar coordinate phase diagram for displaying sensistivity to output microwave load magnitude and phase.


SKOWRON: CROSSED-FIELD AMPLIFIER 34 1

COOLANT METAL INTERFACE

Fig. 13. RK8129 anode power dissipation.

means of specially instrumented CFA models. An example of such a measurement is shown in Fig. 13. The technique consists of running coolant through each vane of a CFA and individually measuring the thermal energy picked up by the coolant while the tube is running. Assuming that the efficiency of interaction for each vane is roughly constant, and knowing that the dc voltage is, in fact, constant, the power dissipation is therefore a direct measure of the current intercepted by each vane. In any case, the dissipation energy is real, with the only uncertainty being the distribution along the length of each vane. This can be estimated by presuming that the R F signal has a half-wave distribution over the length.

The results of such measurements made on various types of tubes indicate that one or a few vanes near, but not at, the output receive power approximately twice the overall average. With additional allowance for longitudinal variation, about a three-to-one distribution factor may occur. For safe cooling and negligible erosion of vane material during life, the design must allow for the power distribution factor. Why the power reaches a peak and then declines at the output is not fully understood. This pattern has been observed and measured on forward-wave reentrant, backward-wave reentrant, and non- reentrant forward-wave tubes.

The effects of power density on the cathode also need to be considered, particularly when the cathode is a metal oxide because of the chemical dissociation of oxygen from the metal with electron bombardment. Deterioration of the oxide greatly reduces the value of the secondary emission ratio and can cause tube failure. In general, a replenishment mechanism must be built into the tube to replace the oxide destroyed by bombardment. In the classic magnetron and in the earlier CFA’s, the replenishment technique was the use of temperature to vaporize the metal remaining after loss of oxygen and expose fresh oxide, or to replace the surface oxide from within the cathode. Metal film oxides have been preserved through the use of a partial pressure of oxygen within the tube en- velope. In any case, the operating stress on the cathode for a given objective determines the selection of the appropriate cathode material.

For a given power density, i t is possible to calculate the cooling requirements and the transient temperature rise of the vane surface. The average power affects the temperature of the vanes because of the temperature gradient through the vane material and the interface drop to the coolant. Standard

FACE OF VANE SHOE =l;;;;f-. Fig. 14. Model used for solution of one-dimensional heat flow equation.

techniques are available for determining this, and, indeed, the whole problem in its extreme can be reduced to how much coolant can be passed through the geometry consistent with other metallurgical and physical constraints. In practice, power densities of several kilowatts per square centimeters are not unusual, and data beyond 12 kW/cmz have been achieved to illustrate this point [%I, [25].

Electron bombardment of the anode during the pulse results in a transient temperature rise a t the surface being bombarded. The amount of energy stored during the pulse is related directly to the thermal capacity of the metal. During the interpulse period, the stored energy diffuses away from the bombarded surface toward the heat sink.

This problem can be handled by programming a digital computer to solve the one-dimensional diffusion equation using finite difference techniques.

The one-dimensional heat flow equation is

aT aT - a2 - - - at ax2

where

a =K/pc, K thermal conductivity of copper, p density of copper, G specific heat of copper, T temperature, t time, x distance into material from surface bombarded.

The model used for the solution is given in Fig. 14.

tions are The boundary conditions imposed for the particular solu-

a m , o W ) a x K

-=-

T(x , 0) = To

where Pd is the peak power dissipation density impinging on the face of the vane during the pulse, and TO is the ambient temperature of the vane before application of the first pulse.

The following simplifying assumptions have been made. 1) Dissipation is uniform across the face of the vane. 2) All the energy of the electrons is converted to heat a t

3) Heat flow is one-dimensional. A sample solution was carried aut, using a 200-ps pulse,

100 pulsesis, and a peak power density of 67 850 W/cmZ. Re- sults are illustrated in Figs. 15 and 16. I n Fig. 15, the temperature at the face of the vane is plotted as a function of time for one complete period T. These results show that stabilization

the surface of the vane.


342 PROCEEDINGS OF TEE IEEE, MARCH 1973

Fig. 15. Vane tip temperature as a function of time.

0 1 I 0 .01 .I .03 .M .05 .# .07

V U E OISTUCE ( inches)

Fig. 16. Temperature as a function of distance into the vane tip (time as a parameter).

occurs within one period; the temperature a t t he face of the vane returns to nominal before the start of the next pulse. Maximum temperature at the vane face (495OC) occurs a t t h e end of the applied pulse. In Fig. 16, temperature as a function of distance into the vane is plotted with time as a parameter. These results show that the temperature pulsations fall off very rapidly as a function of distance into the vane. In other words, the vane acts like a filter in the sense t h a t it smooths out the temperature pulsations so that, at the coolant-metal interface, steady (dc) heat flow conditions exist.

The peak temperature penetrates only a short distance into the material. With this as a further approximation, the solution to the differential equation can be reduced to a simple formula as follows [2, p. 523, sec. 12.61.

where

AT temperature rise in OC, PB peak power density at surface, K thermal conductivity of material, c thermal capacity of material, p density of material, r pulse duration in seconds.

For copper this reduces to

A T = 0.314P~d;

with P B in watts per square centimeters. The dependence of temperature upon power density is di-

rect, and on pulse duration it is as the square root. If a constant energy per pulse is desired, the temperature rise of the vane surface will be less with a longer pulse and lower peak power.

D. Breakdown Phenomena The arcing and voltage breakdown phenomena, character-

istic of all high-voltage vacuum devices, apply equally well to the CFA, along with the additional factors of the crossed magnetic field and the occurrence of R F voltages in the same regions as the dc voltages. These additional factors introduce both favorable and unfavorable conditions for breakdown. The crossed magnetic field enables the CFA to withstand higher gradients and helps to diminish the persistence of an arc. On the other hand, the RF voltages in conjunction with dc fields in regions having appreciable bombardment, namely, the vane surfaces, accentuate the tendency for breadkown.

In the magnetron, the forerunner of the CFA, conditions were quite often favorable for breakdown. These included high R F fields arising from the high Q of the resonant circuit, dc gradients as high as 2 0 0 kV/cm a t the higher microwave frequencies, and cathode emission densities of 15 to 20 A/cmz.


SKOWRON: CROSSED-FIELD AMPLIFIER 343

Acceptable operation of this tube under such severe conditions depended upon relatively short pulse operation and the ability to withstand appreciable arcing without permanent damage.

The reputation of the magnetron as an arcy device is some- times extended to the CFA despite significant and favorable distinctions. The CFA is a matched transmission line with moderate impedance and therefore has relatively low R F voltages on the vane tips. The cathode loading is designed around the secondary emission capability of the cathode, and, in the cold-cathode case, the emission is virtually independent of pulse duration. Finally, the presence of the RF dr ive signal at the beginning of the pulse causes both the emission and the interaction process to occur without the normal oscillation buildup time of the magnetron, diminishing the risk of voltage overshoot and consequent arcing. These distinctions have enabled the CFA to operate with a large improvement factor with respect to breakdown and allow performance a t pulse durations of up to several milliseconds.

When an anode-cathode arc does occur, it characteristically short-circuits not only the dc voltage but also the R F transmission line. This is identified by the collapse of the feedthrough signal coincident with the arc. By contrast, a high- voltage arc in other regions, such as the high-voltage leads or bushing, will still permit the R F drive signal to pass through and be visible in the output viewing.

The occurrence of multipactoring in the R F system within the tube has been observed when appropriate conditions of spacing, frequency, and power prevail [22]. This phenomenon is not usually troublesome in regions where crossed RF and static magnetic fields occur, as between the vanes, because the magnetic field prevents cumulative bombardment in a local- ized region unless coincidence occurs with a cyclotron resonance condition. However, it can be a serious factor in the connecting transmission lines and matching sections which may have fields in line with the magnetic field and thus focus the multipactoring electrons. The evidence of this lies in the increase in insertion loss a t specific values of feedthrough power and coincident heating and gas pressure buildup. Sensitivity of insertion loss to magnetic field is further evidence. If the high-level insertion loss is equal to the low-level cold loss, this is very convincing evidence that little or no multipactoring is occurring in the tube.

The detrimental effects of multipactoring, in addition to excessive insertion loss, occur when the multipactor phenomenon degenerates into a gas discharge sustained by gas evolution from the heated adjacent materials. This can be severe enough to melt metal because substantially all of the R F power present is absorbed by the discharge and must be dissipated in the adjacent material. LYith the onset of a discharge, the regions of the R F structure on the outboard side are protected because the power reaching that point is reduced, while the regions on the inboard side are stressed further by reflection from the discharge. The trend is for a discharge to settle on the vacuum side of the input waveguide window, which will very likely be fractured even with R F drive energy alone. A source of gas from HV arcing or from a leak in some other region of the tube is thus likely to resuit in input window fracture when RF drive is turned on if the average power of the drive exceeds the power the window is able to absorb without fracture. For example, an output window fracture caused by external factors in the output wave-

guide would probably cause input window fracture immediately following. The reverse situation, failure of input window causing output window fracture, is impossible because the output window is protected by the input failure.

E. Physical Size Factors The physical size of a CFA is dominated by the wavelength

of the operating frequency and the peak power to be generated. The frequency is a self-evident constraint imposed upon all microwave tubes. In the CFA, it places a specific limit upon the longitudinal length of interaction space. The length of the circuit vane, in general, cannot be greater than a half-wavelength. Only the midregion of the vane has sufficient impedance ( R F fields) to be useful. This declines to zero a t the ends of the vane. Paralleling of circuits to increase the useful length of the interacting vane has been proposed and tried to some degree. This has not been successful in practice because of the multiplicity of new modes added by multiple circuits which often have phase velocities similar to that of the desired mode in the intended region of operation.

The length of the interaction region is also subject to a tradeoff with the magnetic field. Since the magnet gap is at least as long as the interaction space, the size and weight of the magnetic circuit enter the problem as tradeoffs. This is mostly a technological issue bearing on any specific requirement. In recent years, improvements in magnetic materials have had a favorable effect upon the tradeoff.

The other significant physical dimension of the CFA is the pitch of the slow-wave structure in the direction of travel of t h e R F signal along the circuit. By reason of the phase velocity, voltage, and current, the design peak power tends to dictate this dimension. Since the phase velocity or synchronous velocity is appreciably less than the velocity equivalent of the dc voltage (about 10 to 1 as measured in electronvolts), the periodic pitch of the microwave circuit is smaller than in the corresponding case of a beam tube. In the latter case, the phase velocity, in volts, approximates the dc beam velocity rather than being much less as in the CFA.

The total current is roughly proportional to the anode surface area which, in terms of perveance, can be relatively high for a given operating voltage. The overall consequence of the dimensional factors is that a CFA designed for high peak power will generally have substantial average capability but will be small as compared to a beam tube of the same peak and average power. Conversely, the design of C W tubes, which are basically low peak and high average power devices, is less feasible and tends to be limited to certain restricted cases where appropriate compromises can be made. The utility of a CFA thus principally falls into pulsed applications where size, weight, and efficiency are paramount considerations.

F . Gain-Bandwidth Relation The power output of a CFA is determined primarily by

the dc input power to the tube. The R F drive power serves mostly to control or lock the operation of the tube. Unlike a linear beam tube, there is no real linear portion of the operating range over which the power output of a continuous-cathode CFA is proportional to the drive. The operation is always in a saturated condition. In addition, the definition of bandwidth and gain is different from tha t of a beam tube. A “3-dB” bandwidth, as such, does not exist. Instead, the bandwidth is



POWER OUTFUl

M dB /

I , POWER DRIVE

Fig. 17. CFA gain characteristics.

the range of frequency over which the tube will operate stably with a specified dc input and RF drive power, including allowance for practical variabilities.

With a prescribed RF drive power, the CFA can operate over a range of dc current, and hence a range of power output. There are an upper limit and a lower limit to this range, as shown on the V-I plot of Fig. 12. The upper limit can be caused by one of three factors: 1) a limitation in the available cathode current as produced by the secondary emission ratio or by primary emission, as the case may be; 2) the onset of a competing oscillation, because the voltage has entered a region of synchronism for another mode; or 3) a limitation in gain of the main amplifying mode for which the drive is no longer able to retain lock at this higher amount of power output. In the last instance, the current range can be increased by increasing the drive power; however, this increase does not occur proportionally. The result is that, at the increased operating point, the ratio of power output to power input, or 'gain," is reduced. A plot of this upper boundary is shown in Fig. 17 on power-in-power-out coordinates. With appropriate design steps, the first two items can be eliminated as a limitation, leaving the third as the intrinsic limit because this relates only to the basic interaction process.

The lower limit of operation is created by the onset of oscillation caused by entering a voltage region that is synchronous with a competing lower voltage mode or by an oscillation due to residual feedback. As with an upper voltage competing mode, design steps can be taken to diminish or eliminate the lower competing mode. The intrinsic limit then arises out of stray feedback between the input and the output, either along the circuit or through the electron stream, or in any other incidental way. In the lower current region of operation, the locking drive power is low in relation to the power output, as shown in Fig. 17. The gain is thus high. The gain in this region may be 25 to 30 dB or more and is high enough to enable residual reflections due to imperfect matching to cause feedback, regeneration, and oscillation. If these are removed, either by improving the junction matching or by the introduction of attenuation such that no frequency determining ele- ment in the circuit exists, then the tube will produce a broad noisy output. In the case of a backward-wave tube, the center frequency of noise output will be determined to a large degree by the dc voltage, because the backward-wave phase velocity, and hence frequency, are proportional to each other and a

I

2.

2446 <

A 1 W DRIVE

I Z W DIM

0 4 W D I M

10 20 30 40 sa 60 m b (-AI

Fig. 18. Plate characteristics for the QKSI 119 CFA.

t

voltage tunable noisy output results. In the forward-wave case, because the phase velocity is nearly constant over the interacting frequency range, the output will be a broad-band noise output. Crossed-field noise generators are built on this concept.

An example of this behavior is shown in Figs. 18 and 19. These are data obtained on a C W 75-W backward-wave tube whose performance, because of the low power, requires a primary emitting cathode. This enabled an assessment of i ts behavior driven and undriven exclusive of any cathode emission or transient factors.

When the tube is operated undriven, it generates a noisy oscillation that is voltage tunable and whose range of frequency performance is prescribed by intrinsic electronic feedback within the tube perturbed slightly by circuit junction reflections. When the tube is driven, its operating voltage as- sumes the value coincident with its free-running value at that frequency. The current (and power output) available from



I UPPER MODE REGION I

FOR SPECIFIED BAND AND FIXED POWER SWPLY SETTING

L

I 2.90 2.95 3.0 3.05 3.10

FREWDICY (Wr)

Fig. 20. Performance of QKSl267 backward-wave CFA.

the tube increases as the drive power is increased. I n addition, a suppression of oscillation occurs with increased drive in the vicinity of the driven operating voltage. The gain in the vicinity of the free-running current is more than 20 dB and of course reaches infinity at the free-running point.

An interesting sidelight on this point is the behavior of the signal and the noise output when plotted as a function of the drive signal. As the drive is increased from low levels, the out- pu t signal a t the driven frequency increases nearly linearly. At higher levels, however, the drive signal suppresses the noise and the output reaches a limiting value determined by the dc power input. Stated another way, the performance shifts from linear to saturated with increasing drive, but when it is linear i t is also simultaneously producing noise. The utility of such a mode of operation is therefore questionable.

Over the operating band, the composite effects of circuit and beam coupling impedance, circuit phase velocity, electrical length, optimum anodecathode spacing, and, in some cases, reentrant phasing of electrons become mutually incon- sistent with the required operating gain resulting in a limitation of bandwidth. Practically, this will show as the intersec- tion of either lower or upper boundary with the operating point (see Fig. 20). Beyond this, the tube will not operate under rated conditions. If conditions are altered, however, as by readjustment of current or drive power, operation can be restored and the frequency band can be extended. I n this con- text, a gain-bandwidth relation exists; it cannot be measured by dropoff in performance but by whether the tube runsor does not run under the specified conditions.

Since mode interference, practically, may prescribe the limits of useful operation of the CFA, it is necessary to consider the factors affecting mode discrimination in the tube. A

VOLTAGE ANODE

"a

ST Rt IN T

ANT

I a ANODE CURRENT

Fig. 21. CFA plate characteristics.

mode interference is actually an oscillation that takes time to build up. In cathode pulsed operation, as discussed earlier, the rate of rise of the voltage in relation to the time constant of oscillation buildup has a significant effect upon the mode boundary. The modulator load line also significantly affects the boundary. The combination, which relates entirely to the properties of the modulator, may therefore entirely dictate the successful or unsuccessful operation of the tube.

The effect of load lines is illustrated in Fig. 21, which shows a variation from a stiff regulated case to a current regulated case. The voltage regulated case is characteristic of a hard- tube modulator, and the current regulated case is more characteristic of a soft-tube line-type modulator. With appropriate switch tube grid input, of course, i t is possible to synthesize the features of either type.

From the diagram, it is clear that the load line slope determines the likelihood of coincident mode possibilities and the available current range in the amplifying mode.

The transient properties of the modulator and their interaction with the various CFA4 modes may also significantly affect the quality of performance. For example, a fast rate of rise which prevents a lower mode from starting may also result in an unloaded modulator until main signal mode current is conducted. Depending upon the details of the modulator transient character, this could cause the voltage to greatly ex- ceed the running mode voltage and intersect the region of an upper mode, resulting in complete failure to run.

Obviously, these considerations do not apply to dc or C u ' operation, since the voltage is maintained constant with time. There are, hourever, transient phenomena in dc operated tubes which are R F keyed or cutoff-electrode modulated. The transient effects arise out of sudden increase in current and loading of the dc supply. Connecting lead inductance may prohibit fast response of current and may induce faults.

The onset of oscillation, and the current a t which it stops, are affected by many factors, as has already been noted. These determine a regenerative gain-feedback relation that exceeds infinity when oscillation starts. The difference between finite gain and infinity which produces oscillation is a small distinction and is independently sensitive to each of the factors con- tributing to oscillation. This means that precise control of the mode boundary by construction uniformity of the tubes is not


346

practically feasible. Allowance for this variability must be made in specifying the bandwidth and operating range.

In operation over the frequency band, it has been noted that the range of current prescribes the useful bandwidth. If the current is permitted to change or vary over the band, the bandwidth may be actually increased or decreased according to the particular shape of the mode boundaries. In the forward-wave tube, the phase velocity, and hence the dc voltage, is nearly constant over the frequency band. With a fixed voltage, the current will be roughly constant over the band. A t the band edges where the phase velocity versus frequency de- parts from linearity, the current will deviate from a constant value. Within the nondispersive range, the frequency can be changed or swept with essentially fixed power supply settings.

In the backward-wave tube, the phase velocity and the voltage are roughly proportional to frequency (see Figs. 19 and 1). In sweeping the frequency band, the voltage and current will vary, resulting in a sloping power output characteristic over the band according to a power source load line and efficiency contour of the tube. While this may be objectionable, the magnitude of the variation is not as large as might be supposed because, in practice, the load line of the modulator introduces a compensating characteristic that will hold flatness of output to less than 1 dB. The mode boundaries should provide clearance for this sloping feature.

I f the rate of change of frequency is high, as in the case of a pulse compression application in which the entire band may be swept within one pulse, the transient response of the power source must be considered because there will be a rapid change in current or voltage, or both, during the time of the pulse. The modulator must be capable of maintaining a constant impedance under rapid changes in voltage to enable the anode-cathode voltage to adjust to the frequency. This requirement is particularly necessary with the backward-wave tube because of its voltage tunable nature, although the same is true of the forward-wave tube, but to a lesser degree. If fast response is not provided, the modulator will prohibit the voltage from shifting in response to frequency. The current will then change and cause a change in the R F power output. The frequency modulation during the pulse will thus produce amplitude modulation of the output. The discharging line- type modulator overcomes this because, basically, its pulse line and transformer have video response appropriate to fast changes and it can provide a matched source impedance during such changes. It thus acts like a constant energy source which tends to hold power constant over the band despite some changes in voltampere ratio.

I t has been noted that the CFA has regions of operation a t high gain. These are at operating points of relatively low current. Stated another way, on a normalized basis, the tube is running in a region where i t is large in relation to its power output capability. To operate usefully in this region, the'input and output junctions need to be well matched, as do the load and the drive source. Without attenuation in the tube, the reflections can create regenerative and degenerative feedback as described in Fig. 22, creating oscillation at worst and some phase cycling at best. This can be minimized by the deliberate insertion of attenuation in the circuit, as in a TWT. However, a penalty is imposed in the form of reduced efficiency and reduced average power capability because of the losses and poor thermal properties of attenuating materials. The at- tempt a t high-gain operation thus discredits the most useful features of the CFA, namely, high power in a small package

PROCEEDINGS OF THE IEEE, MARCH 1913

0 INPUT - I = AMPLIFIER X 0LIIPL.n

LOAD

REFLECTION INPUT OUlPUT

REFLECTION '1 '2

voltage gain of an amplifier arising out of reflections rl and

voltage gain of the amplifier without reflection, rt located on each end of the amplifier,

reflection coefficients of input and output reflections, respec- tively, phase length (inclusive of the amplifier) between input and output reflections.

Fig. 22. Effect of input and output reflections on the gain of an amplifier.

TABLE I CC-CFA MODULATION: PRINCIPAL FEATURES

Modulation Method

DC Operation

Circuit Type Pulsing Electrode RF Keyed On-Off Cathode Control

Backward wave suitable

and

not suitable with frequency agility

Forward equivalent suitable with suitable with regulated wave regulated power supply; not suit-

power supply able with feedthrough mode

with high efficiency. This tradeoff so far has not been particularly fruitful.

G. Modulation

The CFA is basically a diode, having a cathode and an anode. .If the cathode is a cold secondary emitter, then the tube has one more performance feature for which no counterpart exists in other tube types-specifically, that its current can be turned on by the RF signal and not by the application of anodecathode voltage. In this respect, the RF signal per- forms the function of the grid in a power grid tube or in an electron gun. The methods of modulation, therefore, have to give consideration to this property, and indeed may even capitalize on it for more advantageous performance.

The various possible methods of modulation are summa- rized in Table I . These methods show the alternatives that exist and how they may fit the forward-wave and backward- wave types. The injected-beam type of CFA lies in a different category because the current is produced in an electron gun and controlled by the gun electrodes. The method of modulation is therefore clearly grid controlled.

The CFA diode may be modulated by the anode-cathode voltage. This can be accomplished with a switch tube or a line-type soft-tube modulator. In any case, the anode- cathode potential starts from zero and travels through all values until it reaches the running voltage. The reverse is true on the down slope at the end of the pulse. In the case of a primary emitting cathode, current is available at all times



and, unless the RF dr ive signal is on prior to the voltage pulse, self-oscillation may occur during the entire pulse. In the case of a cold second emitter, the RF drive can be on either before or after the voltage pulse since emitting current does not flow until drive is applied. However, a complicating factor is the load line property of the modulator which may permit overshoot of voltage during the undriven time at the beginning of the pulse and cause breakdown and arcing. The determination of R F drive timing depends heavily on the modulator type. Delayed R F is not feasible with a line-type modulator because of the large overshoot that always results. With a regulated hard-tube modulator, delay is feasible. The leading edge of the drive signal can then be used for a time reference.

Both the forward-wave and the backward-wave types can be cathode pulsed. If cathode pulsing is used for any reason, then the distinction between the forward-wave and the backward-wave types becomes academic because the cathode pulsed modulator is or can be basically a constant current source and the voltage flatness over the band is not an issue.

DC operation, in which the current is turned on by R F drive and turned off by pulsing an auxiliary electrode, has been shown to be feasible. The advantage of this concept is that the pulser is a small low-energy modulator as contrasted to a cathode pulser. To be useful over a band of frequencies without power loss caused by deliberate softening of the power supply load line, this approach requires a forward-wave circuit.

The basic concept is shown schematically in Fig. 23. The application of pulses and timing are shown in Fig. 24. The principle relates to the reentrant feature of the electron stream in a circular format CFA in which electrons are required to reenter the interaction region after traversing a drift region between the input and the output to maintain operation during the on-time of the pulse. If the electrons are prevented from reentering the interaction region by removing or collecting them, the single transit through the system is not sufficient to maintain the tube running current and the tube will shut off. The method of accomplishing this is to place an electrode mounted within the cathode, but insulated from it, and pulse it positive during the shutoff cycle. The electron current will be collected by this electrode. The amount of current is about 25 percent of the running current, but its duration is for a short time, on the order of nanoseconds. The energy requirements for the pulser are low because of the short interval of current flow.

An alternative configuration for this ‘cutoff electrode” approach is to remove the electrons from the interaction region in the longitudinal direction rather than circumferen- tially. This can be achieved by electrically separating the end shields from the cathode and pulsing these positive. This has been tried experimentally and found to be feasible; however, the p is low because of the form factor of the anode-cathode space. The fields created by pulsing the end shields do not penetrate far into the interaction space.

The most ideal form of modulation from the view of the user is to operate in a fully R F keyed mode. In this method, the current and the amplification mechanism of the tube are actuated by the R F drive signal, both for turnon and for turnoff. One then has the capability for self-modulation by the R F signal but with fu l l tube efficiency since no dc power is flowing and wasted during the R F off interval.

The simplest way to visualize this mode of modulation is to consider the reentrant tube with the cutoff electrode with

El3 + -

RF ENERGY

ANODE SLOW WAVE ELECTRONS

SECOWDARY EMITTER

CUTOFF ELECTRODE IN OR1 FT SPACE

Fig. 23. Schematic of CFA with cutoff electrode.

a steady bias voltage applied to the electrode rather than a pulsed signal. Noting that when the electrode is at cathode potential, the tube is in a running state because all electrons passing by are not interrupted in their passage, and when the electrode is pulsed positive, all current is collected forcing shutdown, there is clearly an in-between region of potential for the electrode in which one would expect sufficient reentrance to maintain operation while drive is on, but be unable to maintain it with drive off. With appropriate bias on such an electrode, R F keying has been achieved.

The reentrance condition can be removed by so designing the tube that its electrical length is long enough to sustain operation without the reentrance condition. This can be built in a linear format [ IS] . The emission near the input end of the tube is triggered on or off by the incident signal. The electrons cumulatively turn on additional current down the tube until the entire length is functioning [27].

H . Phase Performance The spokelike charge character of crossed-field interaction

would lead to the expectation that the RF drive signal would have a good phase lock onto the output signal and thus result in highly stable phase properties. On the other hand, the noisy free-running nature of the undriven CFA would raise the question of over what range of performance stable phase lock can be achieved.

The CFA in its normal regime of operation, that is, a t gains of 10 to 20 dB, exhibits very good phase stability. This feature, together with its short electrical length, has made the CFA particularly suitable for multitube usage in which phase coherence among tubes is important for signal quality.

Phase characteristics of interest which warrant discussion are the following: 1) phase sensitivity to changes in dc input power; 2) phase sensitivity to changes in RF drive power; 3) phase linearity with frequency; 4) response of output phase to fast changes of input signal phase; 5) response of output frequency to fast changes of input signal frequency; and 6) behavior under multiple signal input drive frequencies.

The phase sensitivity to dc input power can be referenced to a change of either dc voltage or dc current and can be trans- lated from one to the other through the dynamic impedance. Fig. 25 shows. a plot of phase sensitivity referenced to l-percent change of anode current for two backward-wave ampli-



RF DRIVE PULSE (detected)

I I I I

RF OUTPUT PULSE (detected)

I I I I I I 1

CFA ANODE (BEAM) CURRENT

( i b ) -1 1 :

I I 1 '

I ! I

I

POSITIVE VOLTAGE PULSE APPLl ED TO CUTOFF ELECTRODE (ecr) I RATED e,, FOR SnUToFF

(with respect to cathode) I I

I I

I

I

CURRENT FLOW TO AND FROM THE CUTOFF ELECTRODE ( ice) < 40 IUOSECOIDS SlUlUFF

(posit ive current corresponds t o n e t e l e c t r o n f l o w & t h e C.E. from the interact ion space) L M I S S I O I CURRUT KEPOFF al-1

( 2 11 i,) ( 2 n I,)

Fig. 24. Normal phasing of CFA pulses.

2 1 , l,

20-

19 -

18 -

17 -

.I 0

I

16 -

= I S - 4 m -

I 4 ~

13 -

12 -

I I -

IO -

QKS1267 /'

60.0 LW PULSED OUTPUT / 2900 MHz

/

F - B A N D BW-CFA / //

/

QK1300 25.0 W C W OUTPUI 2280 MHz

E-BAND BW-CFA

1 . . . . 1 . . " I . ' " . ~ ' ~ ~ ~ ~ " " ' ' ' " ' ' ' ~ ~ ~ ~ ' 0 . 5 1.0 1.5 2.0 2.5 3.0 3.5

PHASE S H l F l DEGREES/% A i b

Fig. 25. Phase sensitivity versus gain. Authorized licensed use limited to: Princeton University. Downloaded on December 8, 2008 at 15:31 from IEEE Xplore. Restrictions apply.

SKOWRON: CROSSED-FIELD AMPLIFIER 349

0.3

0.2 -

tpd = lops prr = loo0

I tpf = 9 p s pd = 8.5 kW 0.1 - 1

I I

OL!, , I , I I , , , I I , , , 0 .5 1.0 I

PHASE SHIFT DEGREES/%Aib

Fig. 26. Phase sensitivity to dc input power versus operating with RF drive constant.

tp = 250 B = 1238 6 f = I325 MHz CONSTANT VOLTAGE (10.5 kV)

+ 2 RELATIVE 0

PHASE - 2

I" - y DEQREES

- 6

PEAK RF DRIVE - kw t u

12 REUTI YE

PHASE

DEGREES 111 - 2

- 6 9.5 10.0 10.5 11.0

DC M O D E VOLTAGE - kV

Fig. 27. QICSlSl9 D-band forward-wave CFA; phase sensitivity to drive and voltage.

fiers. One is a 25-W CW S-band type having a hot primary emitting cathode, and the other is a 60-kW pulsed S-band tube operating on secondary emission. In both tests, the dc input power was maintained constant while the R F drive power was varied to obtain the data points as a function of gain.

The high-drive low-gain regions clearly exhibit better phase sensitivity, indicating that the phase lock on the interaction process improves with larger driving signal as one might expect. Fig. 26 shows phase sensitivity to dc input power as a function of operating current while holding R F drive constant. Again, the implication is that the phase lock improves when the locking power increases relative to the interaction power.

Phase sensitivity to dc data has been obtained on forward- wave tubes with substantially similar results. The phase sensitivity of a dc operated forward-wave tube with cutoff electrode is shown in Fig. 27 as a function of operating voltage and R F drive.

The linearity of phase with frequency is determined almost entirely by circuit reflections and gain of the tube. The fraction of the cold circuit passband that is used for the bandwidth of the tube is small enough that phase deviation contributed

by the passband shape is negligible. The impact of a given circuit reflection on linearity, of course, depends upon the electrical length of the tube and its operating gain. In general, the CFA is short in wavelengths by comparison to a linear beam tube. The backward-wave type is generally shorter than the forward-wave type. I t ranges from a low of f wavelength to a high of 10 wavelengths for the interaction network. The forward-wave types extend typically to 15 or 20 wavelengths. The phase linearity is directly related to junction match quality and fabrication uniformity and, in principle, can be improved arbitrarily. In practice, however, the short electrical length of the CFA is quite advantageous, resulting in easily achievable deviation from linearity of less than f 5 " with routine manufacturing methods.

Of some interest is the hot-to-cold phaselength of the tube. In the normal operating regime of the tube, the space charge behaves as a capacitive reactance (as well as a negative resistance) to the microwave circuit. In the magnetron, this results in a lower hot running frequency than the cold resonant frequency. In the CFA, it produces additional phase shift, causing the electrical length to increase by roughly 20 to 40" from the cold case. When the anode current is increased, the capacitive space-charge loading and the electrical length diminish. Experiments indicate that this effect continues until the tube unlocks at the upper mode boundary. At this point, the hot and cold phase shifts are equal, or the incremental hot-to-cold difference in phase is zero.

The response of the output phase and frequency to fast changes in these on the input signal is important to pulse coding and pulse compression modes of operation. I f the input signal makes a sudden phase reversal or a sudden frequency jump, the cold circuit network will follow such changes as long as i t has sufficient bandwidth to accommodate the spectrum components arising from the change. The time factor is the transit time through the tube as determined by the group velocity and circuit length.

A question that often arises is whether there is a time factor, or an "inertia" factor, associated with the shifting of the space charge from one phase configuration to another. T o answer this, tests were conducted on a 100-kW reentrant forward-wave D-band tube that was 15 wavelengths long, by step changing the phase or the drive signal 180" in a time period of 20 ns [28].

The 20-ns figure was the speed of a phase reversal switch used in the test. The jitter observed during the reversal, which included the reversal switch, the TWT driver, and the CFA, was less than 1 ns. The transit time through the slow-wave circuit of the CFA was approximately 30 ns. This indicates that the space charge can reorient itself in less than a transit around the tube and probably less than that as judged by the low jitter. A further implication of this test is that one could expect to change frequency within bandwidth restrictions in an equivalent amount of time.

Frequency sweeping and jumping modes have been employed using CFA's but the rate of change has not been sufficiently fast to approach the limiting capability of the CFA, and the limit currently lies in the circuitry producing the change. The backward-wave CFA is capable of instantaneous bandwidth, frequency sweeping, and jumping performance with the provision that the anodecathode voltage source have a response time consistent with the rate of change of frequency to avoid FM to AM conversion because the voltage of the backward-wave tube varies with frequency.

Because the CFA is an amplifier "locked" to the drive Authorized licensed use limited to: Princeton University. Downloaded on December 8, 2008 at 15:31 from IEEE Xplore. Restrictions apply.


FW CFA DUAL SIGNAL PERFORMANCE S I ot 1258 MHz HELD CONSTAM

S2 at 1337 MHz VARIED

00-

1258 MHz S I DRNE POWER

= 3.6 kW COMTANT

x ) t Fig. 28. Forward-wave CFA dual-signal performance.

signal, it is not a linear amplifier. Its performance is most suited to frequency modulation. As with any saturated amplifier, it cannot accept multiple signals simultaneously and re- produce these without distortion or suppression. Fig. 28 shows the effects of driving a forward-wave CFA with two signals. The data display the behavior of one signal as a widely spaced (in frequency) signal is added. Fig. 29 shows similar data for closely spaced frequencies on a backward-wave tube, including one of the intermodulation products.

The introduction of a second signal has the effect of suppressing the first signal; however, the overall efficiency remains roughly the same. The general effect of suppression apparently applies to a competing oscillation as though it were an independent signal introduced into the interaction. For example, free-running noise can be suppressed by the gradual increase of a coherent drive signal until the noise close to the signal (intraspectral) is very low.

I. Noise The CFA, as an offshoot of the magnetron, has been looked

upon as a noisy device. The technical basis for this view has been that the rotating electron cloud is a multivelocity stream ranging from zero at the cathode to synchronous velocity or higher at the anode, which leads to the view that random currents will be induced on the anode and result in generated noise. There is substantiation for this view since noise generators and voltage tunable crossed-field devices have been built which base their performance on interaction of the stream with a broad-band circuit.

40

A OUTPUT POWER OF S1 o OUTPUT P O W E R OF S2

OUTPUT POWER OF IMERMODUIATIO

1 PRODUCT AT fS - 1 MHz

16 10 14 18 22 26

S2 INPUT POWER ( d h )

Fig. 29. Q K S l O S l E-band backward-wave CFA; variation of output power spectrum as a function of input power of SI while SI is held constant at 24 dBm.

If the circuit and the electron stream do not have any frequency determining elements, then one would expect some form of noise production. If the cathode emission is restricted by some means such as thermionic limitation, then the frequency determining feature is the applied voltage and the output will tend to be a voltage tunable signal.

If, however, the operation of the tube is locked in and controlled by an incident drive signal, then the performance of the CFA is remarkably quiet. The locked-in characteristic of cycloidal motion tends to suppress noise generation. This has been demonstrated experimentally by taking a device meant to operate as a noise generator (undriven) and then progressively increasing the drive signal while observing the reduction of noise.

T o place a correct perspective on the noise created by the CFA, i t is necessary to distinguish between the several possibilities that exist. There are three general types: spurious signals; interpulse with voltage on, current off; and intrapulse with voltage on, current on.

Spurious signals arise out of oscillations occurring at some time during the pulse. This type of problem relates to the design quality of the tube and tradeoffs existing between such factors as average power and attenuator capability. I t may also relate to the microwave environment. This type of problem is controllable, and it can be completely eliminated if sufficient latitude prevails in other specification areas.

The usual manifestation of spurious signals is on the rise and fall of the voltage pulse, as indicated in Fig. 1. With good attenuation, or matching of these regions, oscillation can be minimized or avoided. Such oscillation can be eliminated by a fast voltage rise and fall, as could occur in a hard-tube modulator. Any assessment of this type of spurious problem must be made in conjunction with the actual modulator that is to be used with a given tube.


SKOWRON: CROSSED-FIELD AMPLIFIER

TABLE I1 CFA INTRASPECTRUM PM NOISE MEASUREMENTS

35 1

~

Po (kW)

&/Nan BWb SO/NQ' t P PRF D. SINd (dB) (Hz) (dB/MHz) ocs) (kHz) (dB) (dB/MHz)

D band 100 71.8 10 21.8 11 D band 100 67.0 D band 100 74.4 150 37.2 30 D band 100 73.0

1.3 18.38 40.2 24.0 11 1.3 18.38 42.4

1.0 15.22 52.4 29.0 10 1.0 20.0 49.0

50

50

G band 500 69.0 200 32.0 1 5.0 23.0 55.0 G band 6 60 G band 600

78.0 3 21.7 20 0.160 25 . O 46.7 78.9 3 22,6 20 0.150 25.2 47.8

F band 750 F band 750 F band 60 F band 60 F band 60 F band 60 F band 666 F band 60 F band 60

83.0 88.0 71 . O 76.0 70.0 77 .0 79.0 76.7 60.9

50 50 44.0

39.0

10 21 .o 10 26.0 10 19.0 10 3

26.0

10 22.0 25.7

50 16.0

* Measured noise power relative to spectral line power.

a Measurements normalized for 1-MHz bandwidth. Filter bandwidth for measurement.

Equivalent CW signal-to-noise power density ratio.

In the case of a dc operated tube, the voltage is always equal to or above the running voltage (as in the unloaded interpulse interval). The voltage is never depressed to the values coincident with lower voltage mode oscillations. The onset of this problem is therefore less severe in the dc operated tubes. Note, however, that the dc tubes are primarily forward- wave types that have less running voltage separation from the band-edge oscillation voltage.

A type of spurious signal can arise during the flat part of the pulse when an oscillation is excited a t the running voltage and occurs coincidently with the amplifying signal. Here, again, this can be traced to a specific mode or resonance, and appropriate steps can be taken to remove it. A sidelight on the identification of such a mode is that intermodulation products occurring between it and the main signal produce misleading signals quite removed from the true spurious signal. By changing the drive signal to a new frequency, i t is possible to distinguish between the intermodulation products and the oscillation signal.

Measurements of noise during the time when voltages are applied but the tube is inactive because no R F drive is applied have been made on cold-cathode models. The tube appears to be a totally passive object with noise approaching thermal levels. Values as low as - 115 dBm/MHz have been measured. A typical specification for this item is - 106 dBm/MHz.

LVith a hot-cathode device, this measurement is not applicable since, obviously, primary current would flow and create activity. \iTith no voltage applied, however, the hot- cathode interpulse noise is commensurate with the temperature of t h e R F circuit.

Varied intrapulse noise measurements on CFA's have been made in recent years. Values obtained have been normalized with reference to an equivalent CW value for ready comparison among various types on which, necessarily, the tests have been made with quite diverse values of repetition rate and pulse duration. The best values obtained by this method for both AM and Phl noise are -58 dBIMHz. The spread of values ranges down to the thirties. Table I1 shows an assortment of cases which include both hot and cold cathodes, and forward-wave and backward-wave tubes.

10 10

4.0 14.0 4.0 14.0

5.5 1.25 21.6 5.5 1.25 21.0 5 .5 5.5

1.5 20.8 1.5 20.8

29.5 0.189 10

22.5 0.585 23.3

33.4 0.585 16.6

53.0

42.6 58.0

40.0 47.6

47.0 44.5 49.0 32.6

J . Anomalous Effects The CFA may occasionally display anomalous perfor-

mance that is not necessarily predictable or observable under standard test conditions. Such effects can arise from interface factors relating to particular microwave and socket environ- ments. The nature of these effects may be quite subtle and warrants discussion to avoid confusion.

The first effect relates to the cathode mounting. Since, necessarily, the cathode is insulated from the anode for high- voltage reasons, and since the cathode is centrally located within the microwave interaction region, there is the possibility that R F energy can be coupled from within the tube by means of the cathode structure. Virtually any frequency can be so transmitted by this route if i t is coupled to the cathode structure because the cathodeanode structure forms a coaxial transmission line. The main interaction modes of the anode do not couple strongly to the cathode because of symmetry. I n terms of the impact on tube performance, this negligible coupling may consist of only a small fraction of the total power. However, in terms of impact on other circuitry and instrumentation, this could be harmful. Such cathode energy could adversely affect video circuitry, particularly protection circuits, and create false shutdowns. I t can also reach low-level R F circuits a t the input end of the system and create instability and system oscillation. The high-voltage bushing region of the tube, therefore, should be shielded from other circuits to avoid such possible interaction.

The second effect relates to the occurrence of resonant modes in the high-voltage bushing enclosure or the high- voltage connecting leads. Since the cathode and anode form an open circuit, the probability is that any resonant mode in the high-voltage connecting structure will have a high- impedance point between the cathode and the anode in the electron interaction region, and is thus subject to excitation by the electron stream. Such modes are likely to be in the A-band frequency region by reason of dimensions. After excitation, such a mode will, in effect, modulate the anode-cathode dc voltage a t a low frequency rate, producing modulation side- bands and intermodulation components. Some of these will be in the microwave region in places where circuit mismatches


35 2 PROCEEDINGS OF THE IEEE, MARCH 1973

occur and may cause arcing in the microwave structure. Others will be in low-frequency regions and will deliver energy to the high-voltage system, which again may cause arcing and breakdown, but this time in the high-voltage leads. The corrective step is to attenuate or damp the original resonant mode in the high-voltage connections to avoid excitation. I t is good practice to examine the high-voltage socket from the tube terminals for possible resonances by means of cold test techniques for positive identification.

A third effect arises from the possibility of high-Q resonant modes in the waveguide system which occur at frequencies where the interconnecting waveguide is propagating, but where the tube slow-wave circuit is cut off. Such a mode will be attentuated through the slow-wave circuit, but will display a ?r-mode R F field pattern to the electron stream over a wide range of velocity. While the coupling to the electron stream is weak, the high Q provides sufficient field strength to cause excitation. The direct result is microwave transmission line breakdown and severe arcing. The corrective step is to remove the source of the reflection in the waveguide system that creates the resonant cavity or to introduce a small amount of loss into the waveguide system to lower the Q of the resonant mode. Since the Q is very high, only a small amount of loss, on the order of 0.1 dB, is needed to reduce the Q significantly.

Note that in this discussion the modes under consideration are those arising from the combination of tube and external connections. A retest of the tube under altered external conditions might not display these modes. In general, the excitation of such modes, as well as any spurious mode within the tube which occurs coincident with the amplifying mode, will be more likely to occur at higher anode currents than at lower. The starting current relates to the running current of the tube. I t is thus possible to have no difficulty under one set of operating conditions and power output, and then to run into difficulty under a higher set of conditions.

Since the CFA is a nonlinear device, the occurrence of any other signal at the same time as the main amplifying signal will produce intermodulation products and modulation side- bands of various types. The analysis of any problem relating to extraneous signals must be done in consideration of these extra signals.

K . Performance Prediction The equations governing the motion of electrons under the

influence of the various electric and magnetic fields are well understood. The relativity corrections are not necessary because even in 20-MW tubes operating at dc voltages of more than 100 kV the electron velocities are only around 10 kV or so. However, the solutions to equations applicable to the CFA problem, which correctly include factors such as space charge, emission velocity at the cathode, end effects, higher order space harmonics on the circuit, and secondary emission, are very difficult. With the advent of the computer, the outlook for predictable designs has improved but has not reached the degree of utility that similar effort on beam tube gun designs and beam tube interaction behavior has achieved.

The approximations that make a problem tenable are also those that remove the information really needed for good performance prediction. For example, the assumption of a single traveling-wave component for the interaction process would seem to be a reasonable and valid approximation of the practical problem. Yet, in practice, measurements of power and thus current intercepted by each vane indicate that the

flow of electrons responds to the local total field (standing wave) and, that the exposure time to the fields of the vanes may be less than a transit through the tube. This implies that the traveling-wave hypothesis may be invalid.

The achievable gain as a function of electrical length is one of the important pieces of information one would like to obtain from a computed prediction to save the cost of major design changes if done empirically. In the emitting-sole' CFA, the performance is remarkably independent of length for changes as great as two to one. Even in the injected-beam CFA, which is more amenable to analysis, the accuracy of a gain versus length prediction remains relatively crude.

Finer design features such as the inclusion of interaction space tapering, electron bombardment distribution, and efficiency enhancement are not possible to predict with sufficient precision to avoid the construction of an operating model although useful trends can be so established. Finally, the upper and lower limits to the performance range, which often are determined by the onset of an interfering oscillation and identified during a first hot test, are not possible to predict if the conditions leading to start oscillation are not programmed in the analytical design approach.

These factors virtually force the designer to depend on empirical information together with analytical trends to predict performance features of a new design. While this is not desirable, i t is nevertheless the most expedient approach if one considers the nature of the CFA electron interaction problem.

V. APPLICATIONS A . Device Selection

The selection of a particular microwave tube type should be made on the basis of its suitability for the requirement. Each system requirement is unique, and very often a specific need dominates and overrides all other considerations in the selection of the most appropriate tube type. The CFA is char- acterized by low or moderate gain, moderate bandwidth, high efficiency, saturated amplification, small size, low weight, and high perveance. Enhancement of any one feature will invariably degrade another, resulting in a penalty that undercuts its overall suitability. For this reason, the usage of the CFA has been most successful in those applications in which its natural character is essential for the system objective.

The usage of the CFA should be considered as complemen- tary to all other available types rather than, necessarily, a substitute for them. Its most frequent application has been in coherent amplifier configurations in final stages where efficient and multimode performance is required. The driver stages, which are at reduced levels of power, are served best by lower power, higher gain devices such as the linear beam tubes.

With respect to the selection of a specific type of CFA, the same reasoning applies. The several types have their individ- ual features, any one of which may be the dominant factor that determines the final selection. The injected-beam type is more suitable for low peak power broad-band operation as exemplified by ECM usage. The emitting-sole type is more suitable to moderate bandwidth, high peak and average power operation as exemplified by pulsed radar usage.

In the present time frame, a more immediate factor in the selection of the appropriate tube is its development status and availability, which tends to dominate in the short term. In the long term, the technical performance potential dominates.



B. Application Suitability b y Performance Attribute The low gain, high efficiency, small size, and lightweight

features of the CFA make it particularly suitable for mobile systems. The sizeweight limitations in mobile systems have an impact not only on the size and weight of the tube itself, but, more importantly, on the peripheral equipment. With an efficient final high-power stage, savings accrue to the power supply, the primary fuel quantity, the size and capacity of the cooling system, and the additional incidental power requirements such as filament and solenoid. Advantages further accrue if the system requires multipower modes that can be achieved by omission of pulsing to the final stage and allowing lower level drive power to feed through the final stage. The system efficiency does not degrade with such operation.

In mobile systems, it is common to subdivide the various radar units into transportable modules. A system composed of a chain of amplifier stages naturally adapts itself to the mobile concept with minimal interstage connecting circuitry. I t further permits an “add on” module concept that can either upgrade or derate a given chain to fit varied requirements without major modification. Typical configurations that have been established successfully in field use are shown in Fig. 30.

The features that make a CFA attractive for mobile use, of course, are those that may be inconsequential to a large fixed installation. The availability of unlimited primary power would make high efficiency unimportant to such a requirement. Similarly, large permanent floor area would render size and weight of no importance. In requirements of this type, other performance attributes would dominate the problem and the tube selection would, of course, be different.

A factor that may enter the picture in all types of systems is the operating voltage of a high-power microwave tube. The CFA is characteristically a high-perveance device, which means that the operating voltage is low for a given peak power output. This factor has an impact on the size and cost of d c components in the modulator and power supply. The cost impact arises out of initially costlier components and/or later reliability and replacement costs. Obviously, in mobile systems, the space occupied by higher voltage components is a further significant factor. In the short run, the added cost of higher voltage components may not dominate the problem because the cost of a new system is largely in the engineering design and software rather than in the initial hardware. In the long term, however, the maintenance and replacement costs dominate. These relate directly to life and reliability. T o estimate this, it is necessary to perform tests or to use available experience to make cost projections. In general, available experience and technical assessments result in projections having higher confidence factors with lower voltage devices.

Fine-grain electrical performance features of the CFA which may influence its selection for a given requirement are mostly in the area of phase and noise properties. LYhile phase stability of the CFA has been well established, i t is only recently that noise information bearing on signal quality of a radar system has been obtained. All data indicate that, with regard to such factors as phase coherence, jitter, fast rate frequency sweeping, and phase and frequency discontinuities, the CFA does not introduce an intrinsic limitation. With regard to intrapulse noise, the value of -55 dB/MHz below a C W carrier appears to be satisfactory for most applications.

The reproducibility of the CFA adapts itself readily to applications having multiparallel amplifier stages, as in a transmitter driving a phased array. The modes of operation

I < CASCADE C H A I N WITH INTERPULSE FEEDTHROUGH CAPABILITY

10 dB

PARALLEL FINAL STAGE FOR UPGRADED POWER CAPABILITY

CORPORATE FEED WITH PHASE SHIFTERS

Fig. 30. Typical CFA configurations.

available are with the tubes either in parallel with power com- bined at the outputs, or separately feeding the antenna. The parallel feature also makes possible designs in which flexibility of approach exists by added combinations of stages.

C. Environmental Factors The CFA has had field experience in fixed, land mobile,

shipborne, airborne, and space applications. Environmentally, the principal design consideration has revolved around the shock and vibration resistance of the cathode structure and the magnet-the cathode because it commonly has a cantiliver construction, and the magnet because of its weight. Neverthe- less, designs have been achieved which have met these varied environmental requirements together with the necessary thermal and electrical requirements with high reliability and good field experience. The small-size and low-voltage features of CFA interaction have enabled tubes to have physical designs that readily conform to environmental requirements with the main design effort concentrated on the two areas noted.

D . Field Experience Experience with the CFA in field use goes back approxima-

tely 12 years. The number of tube models manufactured in quantity lots and delivered to systems in the field is estimated, in total, a t about 3000 units during this time period. The approximate number of tube models, delivered as a function of the operating frequency, is shown in Fig. 31. Most of these are high-power cathode-pulsed radar types operating between 100 kh’and 5 MW of peak power output and more than l-klt’ average power output.


354

FREQUENCY (GHr) 1 .o 10.0

Fig. 31. Quantities of CFA’s delivered to system sockets in field use.

10 OOO

Y i

E P

f 1 .o 10.0

TIME (Yean since quantity me started) 100.0

Fig. 32. Recent average tube life as a function of years in service.

The average tube life in systems experienced in recent years is shown in Fig. 32 as a function of years in service. The lower frequency types have been in service longer, as indicated on the curve, and have acquired more history than the higher frequency types. The best life, as with all microwave tube types, also correlates with the lower frequencies where the tubes are large and understressed. Noteworthy, however, is the average life at 3 GHz where the tubes are relatively small but the peak and average power are appreciable. All data are for high-voltage operating time since filament life is not applicable to operation of most CFA’s.

The projection of future life for the basic CFA based upon this history is quite optimistic. Most of the types indicated have used oxide cathodes of various types. The mode of failure

PROCEEDINGS OF THE IEEE, MARCH 1973

is usually cathode related in that the loss of oxide (and emission) with life creates various stability failures. Except for the cathode, the integrity of the rest of the tube body has been sufficiently good to commonly permit full use of the structure for rebuilding. Ultimate tube life is determined largely by random failure and cathode wearout. Random failures relate mostly to manufacturing quality and field application problems. Cathode wearout is a design consideration. Toward this end, the more recent CFA’s employ cold cathodes of several varieties. There are either pure metal or cold oxides. The pure metal type, such as platinum, has no wearout mechanism other than erosion. Estimates of this factor indicate extremely long life outlook exceeding 100 000 h. The cold oxide types wear out through dissociation of the oxide. However, replenishment mechanisms that can be either continuous or periodically applied are under development, and these enhance the outlook for preserving the oxide for an indefinitely long time. The outlook for large life expectancy improvement of the CFA by the elimination of its principal wearout mechanism is extremely good: 100 000-h tubes should be achievable.

VI. FUTURE TRENDS The outlook for the future of the CFA lies in two general

areas-technological and functional.

A . Technological The technological area of improvement of the CFA lies in

the incorporation.of newly developed materials and processes to enhance the performance or utility of the device. Where the needs lie and where the materials exist are not necessarily consistent with each other. One such area is in cathode materials.

For attractive low-voltage CFA designs and for low-level operation, cathode materials having a high secondary emission ratio are desirable. Oxides are the presently known best materials that fall into this category. For operation under dc and R F keying conditions, materials that are stable and unchanging with life, with electron bombardment, and with power level are required. Pure metals fall into this category. Mate- rials tha t fall into both categories at this time do not exist. Work has been performed in attempts to combine both features through compositions and matrix approaches. While these have shown promise, they have not yet demonstrated all things at one time. The biggest shortcoming lies in the stability of emission properties under high-power conditions and with life.

Despite these problems, however, film oxides on metal have performed well when the design avoided severe stress on the cathode surface and when an oxide replenishment mechanism was provided. Composition cathodes have also functioned well but require cyclic replenishment of the oxide by heating. With further research, these limitations are expected to be overcome. In the meantime, the tube designs need to be tailored around the cathode to insure practical devices.

Historically, the magnetic field required by M-type tubes has always created a problem. To the user, it has been a weight and proxmity problem. To the tube designer, it has been a constraint upon internal dimensions, limiting flexibility in R F design features.

The magnetic field required by a crossed-field tube is roughly proportional to frequency. Xumerically, a C-band tube will require about 600 G, an F-band tube about 2500 G, and G - and I-band tubes from 4000 to 6000 G. The magnetic field usually has been provided by Alnico magnets, although a t C band the large size and low field make solenoids competi-



INTERNAL SmCo POLE

MAGNET \

VACUUM

/ ALTERNATE LOCATION FOR FIELD SHAPING MAGNETS \

ANODE FIELD SHAPING MAGNETS

Fig. 34. Section of a forward-wave CFA having internal main Sm-Co magnet and auxiliary external field shaping magnets.

positioned in a region that is closest to the place where the magnet field is needed. The efficiency is thus high and the fringing field extending externally to the tube is negligible. Fig. 34 shows the interior of an experimental model built with samarium-cobalt magnets positioned inside the cold cathode of a forward-wave amplifier.

Recent improvements in the mounting and cooling of ferrites have provided a means to build directive attenuation or frequency-selective attenuation within the tube. The outlook for more mode stability and less sensitivity to reflections is thus enhanced without the expense of efficiency loss, as would be introduced by ordinary attenuating materials having

(C) two-way insertion loss. As noted earlier, the useful gain of a Fig. 33. Examples of computer field plots. (a) QR1640 magnet plot, CFA can be quite high if interference from competing modes

straightening field. (b) QR1640 magnet plot, bowing field. (c) Mag- is eliminated. Ferrites provide an outlook for such gain im- netic arcuit with external bucking magnets. provement.

tive on a size-weight-power basis. The low coercive force of Alnico requires that magnets have a long length and leads to the horseshoe or bowl configurations. This also leads to substantial leakage paths that diminish the efficiency and add weight.

Two recent advances relating to magnets promise appreciable improvements in tube designs, particularly when related to the cold cathode. The first is the development of computerized magnetic-field design methods [29] which include the properties of the magnetic material in the problem. The second is the development of rare-earth high-coercive-force materials

The first advance makes possible magnet designs having shapes specifically intended to reduce leakage flux. An example of this is a radially gaussed cylindrical shape. Another example is the use of multiple magnets in configurations of opposing polarity which can shape the field pattern more effectively than by pole shaping alone. Fig. 33 shows several examples of field plots made in this way.

The second advance allows a large reduction in the size and weight of the magnet. The size reduction makes possible the placement of magnets in otherwise unsuitable places, such as inside a cold cathode. In the latter case, the magnet is

[301.

B. Functional Areas In the functional areas of CFA operation, the dc operated

R F keyed device holds the most appeal to the user. The ideal of a passive device connected to a dc power supply and actuated only by the R F drive signal has been shown to be feasible by several routes. What is still needed, however, is the demonstration of consistent, repeatable, long-term performance under conditions characteristic of typical systems. The shortcoming lies in the cathode material area. The need is for a high-i material whose properties remain stable with time or stress. The film oxides have not.been fully satisfactory in this respect, necessitating some combination of materials which places oxides in regions of low stress and plain metals in regions of high stress.

The failure modein dc operated CFA’s lies in failure to either turn on or turn off. This is nearly always traceable to variability in the cathode secondary emission. The failure to turn on results from the emission buildup time for the given drive and other conditions being longer than permissible. The failure to turn off means that the space-charge cloud has not been fully quenched at the end of the RF drive pulse, resulting in the tube going “long” pulse or CW in a n oscillating mode which then activates a protect circuit.



T h e R F drive is introduced, typically, into the anode, which is the R F circuit whose fringing fields extend toward the cathode where they are the weakest. In this respect, the RF drive, which is expected to control the entire tube operation, is introduced in precisely the wrong place because the controlling signal is weakest in the region where i t must exercise most positive control over the emission. Conceptually, what is needed is some feature analogous to a grid in an electron gun whose controlling signal approximates the energy being controlled if positive, uncritical performance is to be expected. New investigations along this line are now under consideration and hopefully will produce improvements and advances for the future.

The cold cathode, once quenched, will not restart until R F drive is reapplied. This feature leads to additional modulation concepts that are extensions of the cutoff electrode type of quenching. The cutoff electrode collects the electron stream when it is pulsed positive a t the end of the R F drive pulse. The same result can be achieved by pulsing the entire cathode at a less negative potential a t the end of the R F drive pulse. Experimental work indicates that the quench pulse applied to the entire cathode can be short, at least no longer than the fall time of t h e R F drive pulse. The pulsing energy is somewhat higher than the equivalent cutoff electrode, but this method eliminates the complexity of an additional insulated electrode in the tube.

VII. CONCLUSION The crossed-field amplifier has several versions of which

the continuous-cathode emitting-sole type is the most common and best established. I t has been in field service for about a decade and field life of up to 10 000 h average has been achieved. Other versions of the CFA, such as the injected- beam type and the self-keyed dc operated type, have been developed with special performance features. The IB-CFA has potential application for ECM because of its wide-band performance capability. I t is also under development for radar use at high peak power. The self-keyed dc operated types are available in several bands and are just beginning to acquire field experience.

The CFA electron interaction process is comparatively simple in concept, but very complex in detail. Analytical solutions have been helpful in establishing performance trends of a design but not with complete precision. However, the gross features of the CFA process can be understood from its similarity to a synchronous generator. In this respect, the CFA interaction is markedly different from other types of interaction. This also accounts for its high efficiency, good phase stability, low operating voltage, and the novel feature of self- keying.

The CFA tends to be small and lightweight. I t is particularly suitable for lightweight transportable and mobile radar systems where size, weight, and fuel consumption dominate the requirements. Various configurations of amplifier chains, which can operate with feedthrough modes and into corporate feed combinations, are possible. The CFA is most suitable for the final stages of such chains because its efficiency can be used most advantageously there.

Future performance of the CFA will lie in the use of newer materials, such as ferrites, rare-earth magnets, and composite cathodes, which should improve performance attributes as well as physical form and weight factors. I t will also lie in de-

sign concepts that should improve or extend the attractive features of self-keying and improve the life and reliability of such tubes.

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a generator of centimeter waves,” BellSyst. Tech. J . , vol. 25, pp. 178- 182, Apr. 1946.

[2] G. B. Collins, Microwave Magnetrons (M.I.T. Radiation Laboratory

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magnetron-type traveling-wave amplifier tube, Proc. IRE, vol. 38, pp. 486-495, May 1950.

[4] B. Epsztein, paper presented at the IRE-AIEE Electron Tube Conf., Ottawa, Canada, 1952; also P. Guenard, 0. Doehler, R. Warnecke, and B. Epsztein, “New UHF oscillator valves with wide electronic tuning band,” C. R. Acad. Sc i . , vol. 235, p. 236, 1952.

[SI R. R. Warnecke, P. GuCnaid, 0. Doehler, and B. Epsztein, “The “‘-type Carcinotron tube, Proc. IRE, vol. 43, pp. 413-424, Apr. 1955.

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[7] H. L. McDowell and J. Feinstein, “The S-F-D crossed field amplifier,” in Proc. 4th I n f . Congr. Microwave Tubes (Scheveningen, The

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. . York: Dover, 1953, chl IX.

[12] 0. Doehler, A. Dubois, and D. Maillart, “An M-type pulsed amplifier,” Proc. Inst. Elec. Eng., vol. 105, no. 10, p. 454, 1958.

[13] R. E. Edwards, “Development of TPOMI-type 10 Mw L-band forward wave amplifier,” Rome Air Develop. Cent. Contract AF30- (602)1830, QK751, Final Rep., RADC-TDR-62-383.

[14] G. Bernstein, H. McDowell, and T. Schultz, “Feasibility model of an advanced crossed-field amplifier for space communications systems,” NASA Tech. Rep. CR-86334, Jan. 1970.

(151 G. E. Pokorny, A. E. Kushnick, and J. F. Hull, “The DEMATRON- A new crossed-field amplifier,” IRE Trans. Electron Devices, vol.

[16] W. C. Brown and G. E. Dombrowski, “Analysis of a new crossed- field amplifier, the ‘EAL,’ or electromagnetic amplifying lens,” IRE Trans. Electron Devices, vol. ED-9, pp. 221-228, Mar. 1962.

[17] H. L. Thal, “Superpower millimeter wave tube,” Signal Corps Con- tract DA36-039 AMC-00031 (E) (continuation of Contract DA36- 039 SC-88960), Rep. 9, Final Rep., July 16, 1962-Sept. 15, 1963.

[18] K. R. Spangenberg, Vacuum Tubes. New York: McGraw-Hill, 1948, pp. 660-665.

[19] W. W. Harman, Fundamentals of Electron Motion. New York:

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[22] ‘Final engineering report on the multipactor study program,” Sperry Gyroscope Co., U. S. Navy Office of Naval Res. Contract NOnr-217(01), Rep. 5270-6144, Oct. 1953.

[23] K. Dudley, “Secondary emission form power tube materials,” in Proc. 7th Nat. Conf. Tube Techniques (Sept. 1964), vol. 2, pp. 643- 688.

[24] W. C. Brown and G. Perloff, “High power C W X-band amplitron,” in IRE Int . Conv. Rec., pt. 3, pp. 52-55, Mar. 1960.

[25] J. F. Skowron, y. C. Brown, and G. H. MacMaster, “The super power amplitron, Microwave J., vol. 7, pp. 65-69, Oct. 1964.

[26] R. L. Jepson a2d M. W. Muller, “Enhanced emission from magnetron cathodes, J. Appl. Phys. , vol. 22, pp. 1196-1207, Sept.;951.

[27] J. R. M. Vaughan, “Beam build-up in a dematron amplifier, presented at the Int. Electron Devices Meeting, Session 17.1, Oct. 31,

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