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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 49, NO. 10, OCTOBER 2001 1357

On the Gain of a Reconfigurable-Aperture AntennaElliott R. Brown, Fellow, IEEE

AbstractA full-wave analysis based on the method of mo-ments (MoM) is carried out for a reconfigurable-aperture antennaconsisting of a two-dimensional (2-D) array of filamentarymicrostrip-dipoles interconnected by lossy microelectrome-chanical-system (MEMS) switches. Activation of specific MEMSswitches allows the dipoles to be maintained near the halfwave-res-onant length as the frequency is reduced in octave incrementsbetween 16 and 2 GHz. This keeps the real part of the dipoleself-impedance much higher and the imaginary part much lowerthan in a dipole having a fixed length of at 16 GHz. Hence,the array-antenna gain and aperture efficiency remain muchhigher with frequency than in an array of fixed dipoles. Broadside aperture efficiencies of 3.9, 6.0, 9.5, and 10.6 dB arepredicted for , , , and recap dipole arraysat frequencies of 16, 8, 4, and 2 GHz, respectively, for MEMSswitches having 0.5 dB insertion loss. In contrast, fixed-element

-separated arrays operating at the same frequencies have

predicted efficiencies of 3.9, 24.2, 45.0, and 63.0 dB,respectively.

Index TermsAntenna gain, aperture antennas, reconfigurableantennas.

I. INTRODUCTION

SEVERAL research programs have been started recently to

develop electronic antennas by a new approach called a

reconfigurable aperture, or recap for short. As suggested by

the title, a recap is fundamentally different than the traditional

electronic antennas that have been developed over the years, in-

cluding the family of single-beam phased arrays, multiple-beam

aperture antennas (e.g., Rotman lens), and the growing familyof switchable-element smart antennas [1]. The distinguishing

feature of a recap is its ability to alter the RF current distribution

within a planar-radiating aperture. In the language of phased ar-

rays, a recap can change its element pattern in addition to its

complex-array factor.

To see this distinction more precisely, recall that the electric

(far) field from a traditional-phased array can be written [ 2]

(1)

where

;

distance (radial unit vector) between the center

of the array and the measurement point;

Manuscript received August 5, 2000; revised November14, 2000. This workwas supported by the Defense Advanced Research Projects Agency under theReconfigurable Aperture Program.

The author is with the University of California, Los Angeles, Los Angeles,CA 90095-1594 USA (e-mail: [email protected]).

Publisher Item Identifier S 0018-926X(01)06368-2.

vector between the th element and the measure-

ment point;permeability of free space;

propagation constant ;

element factor;

array factor;

total number of elements in the array.

In traditional phased-array antennas, the element factor is fixed

by design and cannot be altered electronically or otherwise. The

far-field pattern is changed only by variation of the complex

coefficients in phase, amplitude, or both. In a recap one can

also change the element factor.

Optoelectronic [3] and MEMS switches [4] have already been

used to change the length of single-element antennas, such as

dipoles. Similar devices have been proposed to change the el-ement factor in arrays [5], but it is difficult to tell at this point

which will work best. However, two benefits of a recap, inde-

pendent of the core technology, should be the antenna gain as a

function of beam pointing and as a function of frequency shift.

The pointing issue has already been addressed through the de-

velopment of a broad-side/end-fire switchable antenna based on

an integrated leaky-mode/YagiUda structure [6] and willnotbe

addressed here. Instead, this paper focuses on the issue of fre-

quency shift and how a recap array of resonant elements can

maintain high gain over a wide (reconfiguration) bandwidth by

varying the element length and interelement separation to stay

at or near resonance at each frequency. This does not imply an

increase in the conventional gain-bandwidth product in whichthe bandwidth must be instantaneous. But for some applica-

tions this distinction may not be so significant if the reconfigura-

tion time is sufficiently small. For example, with contemporary

MEMS switches this time would be s, which is adequate

for many communications systems.

The pervasiveness of military and commercial satellite com-

munications in Ku band (1218 GHz) and the explosive growth

of personal communications services (PCS) just above 2 GHz

define an interesting application of a reconfigurable aperture as

an electronically-steerable antenna that can link to space-based

or terrestrial transceivers. This application will define the spe-

cific frequency range and other parameters in the simulation de-

scribed below.

II. CANDIDATE ARCHITECTURE

The primary purpose of this paper is to analyze the gain and

impedance characteristics of a recap antenna in comparison to

the diffraction limit and to a fixed-element antenna having iden-

tical architecture and materials properties, but lacking the recon-

figurability. To facilitate the analysis, the simple recap architec-

ture shown in Fig. 1(a) was chosen. It consists of a square lat-

tice of rectangular microstrip elements, each having

0018926X/01$10.00 2001 IEEE

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1358 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 49, NO. 10, OCTOBER 2001

(a) (b)

Fig. 1. Recap architecture consisting of microstrip elements that can be connected by MEMS switches or to a balanced transmission line to form an array ofplanar-dipole antennas. (a) Maximum-frequency ( ) configuration where the gaps between microstrip elements are alternately opened and connected to abalanced line to form an array of approximately -long, -spaced dipoles. (b) First subharmonic configuration where 50% of the gaps are closed (by aMEMS switch), 25% are opened, and 25% connected to a balanced line to form an array -long, -spaced dipoles at .

length along the -axis and filamentary width (i.e., )

along the -axis, and each separated from its neighbors along

the -axis by an infinitesimal gap (i.e., ). Located at

each gap is a series-connected MEMS switch and a shunt con-

nected balanced transmission line. The figure shows a top view

of this array in the maximum-frequency configuration

in which all the MEMS switches are left open and every other

gap is coupled to RF so that the radiating aperture consists of

a square lattice of microstrip dipoles having length .

The unit cell of the square lattice is defined by the dashed box

(width ). In principle, this array can be quite efficient and

electronically steerable over wide angles under the conditionof half-wave resonance, , where is the

wavelength in free space, and is the

permittivity of the dielectric substrate material (assumed loss-

less). Hence, the maximum frequency configuration is defined

by .

Fig. 1(b) shows the array in the first subharmonic

configuration in which half of the MEMS switches are closed

and half of the remaining gaps are driven with balanced RF

so that the radiating aperture consists of -long microstrip

dipoles lying on a square lattice (unit cell width ). Be-

cause the length of the dipoles has approximately doubled com-

pared to those in Fig. 1(a), this configuration should be reso-

nant at a frequency . Furthermore,if the number of microstrip elements is large, lower subhar-

monic configurations can be produced by judiciously switching

some gaps and coupling RF to others. In each case, the recap

array is reconfigured as a square lattice of half-wave dipoles

with approximately half-wave center-to-center element sepa-

ration. The lowest subharmonic frequency that yields an elec-

tronically-steerable array, and the one called the minimum-fre-

quency configuration, is the array defined by

.

As the recap is configured for subharmonic frequencies, the

number of activated switches must increase in a manner

described analytically by the expression .

TABLE IPROPERTIES OF THE AND THREE SUBHARMONIC CONFIGURATIONS OF

RECAP ARRAY of 32 16 MICROSTRIP ELEMENTS ON A SUBSTRATEHAVING AND THICKNESS mm

For example, , 6, 14, and 30 for the 1st, 2nd, 3rd, and

4th subharmonics, respectively. The switches have a small butsignificant value of insertion and return loss that is ultimately

an important factor in the useful bandwidth of a recap antenna.

III. SIMULATION PARAMETERS AND METHODOLOGY

As suggested above, an interesting application of a recap is

an electronically-steerable antenna that can link to terrestrial or

space-based transceivers over a range between an of 16

GHz and an (3rd-subharmonic) configuration of 2 GHz.

From the relations given above, an array of microstrip

elements will cover this range by providing a

dipole array at 16 GHz and a dipole array at 2 GHz.

At the intermediate subharmonic frequencies of 8 and 4 GHz,and arrays will be available that also satisfy the

-length, -spacing of the dipoles. The characteristics

of the maximum frequency and three subharmonic configura-

tions are listed in Table I, along with the number of switches

per dipole required in each configuration.

The next parameters chosen were the permittivity and

thickness of the substrate material. Through extensive research

conducted in the 1980s, it was shown that high- substrates are

usually deleterious to the performance of microstrip antennas

(dipoles and patches in particular), because of their propensity

for surface modes [7], [8]. It was also shown that there is

always at least one surface mode present (the TM ) and that

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BROWN: ON THE GAIN OF A RECONFIGURABLE-APERTURE ANTENNA 1361

TABLE IIISELF-IMPEDANCE AND NEAREST NEIGHBOR (COLLINEAR) MUTUAL IMPEDANCE VALUES OF PLANAR MICROSTRIP DIPOLES IN FIXED AND RECAP ARRAYS

effect of the ground plane, which at 8 GHz is now 50% closer

to the antenna (relative to ) than it is at 16 GHz. These

trends continue at the lower subharmonics, culminating in a

fixed element, self-impedance at 2 GHz of 0.01212180 and

a recap self-impedance of 1.352.69 .

After obtaining the elements, the current components

were computed by matrix inversion of (2) using standard Matlab

routines. Then the antenna gain was computed from (11) using

(8)(10) as input. Plotted in Fig. 2 is the resulting gain for

(16 GHz) and each subharmonic configuration in comparison

to the diffraction-limited gain at 16, 8, 4, and 2 GHz and to

the fixed-array (dipole length cm) gain at the same fre-

quencies. The corresponding gain values are listed in Table IV.

The recap gain is parameterized by the MEMS switch insertion

loss, ranging between 0.0 and 3.0 dB. For zero MEMS loss, the

gain of the recap array falls at approximately the same rate with

frequency as the diffraction limit, namely . Hence, the

aperture efficiency , remains around dB. At

first this appears surprising in light of the fact that the real part

of the recap self-impedance drops steadily from 61.6 to about

1 between 16 and 2 GHz. However, while the real part drops,

the mutual impedance elements remain relatively unchanged, as

displayed through the nearest-neighbor mutual-impedance term

in Table III. Hence, the mutual coupling between adjacent ele-ments increases, and an increasing fraction of the input power

to a given element is radiated by its neighbors.

In contrast to the recap behavior, decreasing the fre-

quency in a fixed-element array decreases the real part of the

self-impedance while increasing the (capacitive) imaginary

part at a comparable rate. This is evident in Table III where

at all subharmonic frequencies the self-impedance term is

dominated by a large capacitive reactance. Not only does this

present a larger return loss to a 50- generator than a recap

element, but it is much less favorable for mutual coupling. This

point is also demonstrated in Table III where one sees that the

self-impedance terms for the fixed-element array dominate

the mutual impedance terms (collinear nearest neighbors) atall frequencies, making mutual coupling rather ineffective in

transferring power from one element to its neighbors.

Another interesting aspect of Fig. 2 pertains to the effect of

MEMS switch loss on the gain and aperture efficiency. It is

remarkable that no level of switch loss simulated here can re-

duce the recap gain to the fixed-element gain. Thus, an inter-

esting figure of merit is the level of switch loss at which the

recap gain and the fixedaperture gain become equal. Using

the definitions given above, the switch loss will be given by

the expression , where

is the gain of the recap configuration with zero switch loss.

Solving for the insertion loss per switch, one finds

Fig. 2. Antenna gain as a function of frequency for the recap array, afixed-element array, and a diffraction-limited aperture ( ) with a

cm . The gain for the recap and fixed-element arrays are computedonly at the maximum frequency (16 GHz) and the first three subharmonics (8,4, and 2 GHz). The lines connecting the data points are drawn only as a visualaid. The recap gain is parameterized by the MEMS switch insertion loss ( )that ranges between 0 and 3.0 dB. The MEMS return loss is assumed to benegligible.

. The resulting values for are 9.6, 6.4,

and 4.2 dB at 8, 4, and 2, GHz, respectively.

V. SUMMARY

This paper has analyzed two-dim arrays of planar microstrip

dipoles as the frequency is varied in octave increments between

16 and 2 GHz. Two different array structures were considered:

1) a fixed-element aperture in which the dipole length is con-

stant at the maximum frequency (16 GHz)

and 2) a recap in which the dipole length is maintained near

over several octaves of bandwidth by judicious activation

of MEMS switches between the elements. In both structures, the

inter-element separation is maintained at . For the fixed-el-

ement aperture, the gain and aperture efficiency are found to de-crease rapidly with frequency because of a rapid increase in the

return loss arising from impedance mismatch between the gen-

erator(s) and dipole elements. The impedance mismatch is as-

sociated with a rapid drop in the real part of the self-impedance

and a large (capacitive) increase in the imaginary part. And be-

cause the mutual impedance elements all remain relatively

small, there is little radiation into free space by either a driven

element or its neighbors.

In contrast, for low switch insertion loss, the recap array

maintains a high gain with reduced frequency that nearly tracks

the diffraction limit ( ). This is because the switches

maintain the length near the resonance, which keeps the



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1362 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 49, NO. 10, OCTOBER 2001

TABLE IVANTENNA GAIN FOR THE FIXED ELEMENT AND RECAP ARRAYS VERSUS FREQUENCY AND PARAMETERIZED BY MEMS SWITCH INSERTION LOSS ( )

real part of the self-impedance relatively high and the imagi-

nary part relatively low. In addition, the mutual impedance to

neighboring elements, particularly the collinear term is

relatively large and approximately constant with decreasing

frequency. Hence, the recap maintains high efficiency with

reduced frequency largely because each dipole maintains an

acceptable return loss and couples increasingly to its neighbors

in a constructive way, at least for the broadside radiation pat-

tern. Future research will analyze these effects in the presence

of electronic beam steering.

In conclusion, it should be noted that the critical choice ofsubstrate permittivity ( ) in this analysis was driven

by performance and economic considerations. Higher- mate-

rials, such as high-resistivity Si or SIGaAs, would likely yield

inferior performance of the recap antenna compared to that of

Fig. 2 because of deleterious surface-wave effects. However,

MEMS switches are now being developed primarily on such

high- materials. Hence, application of the low- substrate

simulated here would require that semiconductor-based MEMS

be bonded by flip-chip or similar packaging technology. This

approach may be feasible over the simulated frequency range

where the size of the MEMS die should be a small fraction of a

wavelength and, therefore, have little effect on the behavior of

the planar antennas.

ACKNOWLEDGMENT

The author would like to thank T. Itoh and Y. Rahmat-Samii

of UCLA for helpful discussions on this subject.

REFERENCES

[1] K. L. Virga and Y. Rahmat-Samii, Low-profile enhanced-bandwidthPIFA antennas for wireless communications packaging, IEEE Trans.

Microwave Theory Tech., pt. 2, vol. 45, pp. 187988, Oct. 1997.[2] R. J. Mailloux, Phased Array Antenna Handbook. Norwood, MA:

Artech House, 1994.[3] J. L. Freeman, B. J. Lamberty, and G. S. Andrews, Optoelectri-

cally reconfigurable monopole antenna, Electron. Lett., vol. 28, pp.15021503, July, 1992.

[4] J. H. Schaffner, R. Y. Loo, A. E. Schmitz, H. Tsung-Yuan, D. J. Hyman,A. Walston, B. A. Warneke, G. L. Tangonan, and S. W. Livingston, RFMEMS switches for tunable filters and antennas, in Proc. 3rd Int. Conf.

Micro-Opto-Electro-Mechanical Systems, Mainz, Germany, 1999, pp.2449.

[5] E. R. Brown, RF-MEMS switches for reconfigurable integrated cir-cuits, IEEE Trans. Microwave Theory Tech., vol. 46, pp. 18681880,Nov. 1998.

[6] Y. Qian, B. C. C. Chang, M. F. Chang, and T. Itoh, Reconfigurableleaky-mode/multifunction patch antenna structure, Electron. Lett., vol.35, no. 21, pp. 104105, Jan. 1999.

[7] I. E. Rana and N. G. Alexopoulos, Current distribution and inputimpedance of printed dipoles, IEEE Trans. Antennas Propagat., vol.AP-29, pp. 99105, January 1981.

[8] P. B. Katehi and N. G. Alexopoulos, On the effect of substrate thick-ness and permittivity on printed circuit dipole properties, IEEE Trans.

Antennas Propagat., vol. AP-31, pp. 3439, Jan. 1983.[9] N. G. Alexopoulos, P. B. Katehi, and D. B. Rutledge, Substrate

optimization for integrated circuit antennas, IEEE Trans. MicrowaveTheory Tech., vol. MTT-31, pp. 550557, July 1983.

[10] R. S. Elliott and G. Stern, The design of microstrip dipole arrays in-cluding mutual couplingPart I: Theory, IEEE Trans. Antennas Prop-agat., vol. Ap-29, pp. 757760, Sept. 1981.

[11] D. M. Pozar, Analysis of finite phased arrays of printed dipoles, IEEETrans. Antennnas Propagat., vol. AP-33, pp. 10451053, Oct. 1985.

[12] A. Malczewski, S. Eshelman, B. Pillans, J. Ehmke, and C. L. Goldsmith,X-band RF MEMS phase shifters for phased array applications, IEEE

Microwave Guided Wave Lett., vol. 9, no. 12, pp. 51719, Dec. 1999.[13] Matlab 5.3. Natick, MA: Mathworks, Inc.[14] D. M. Pozar, Improved computational efficiency for the moment

method solution of printed dipoles and patches, Electromagn., vol. 3,

pp. 299309, 1983.[15] R. C. Hansen, Phased Array Antennas. New York: Wiley, 1998.

Elliott R. Brown (M92SM97F00) received theM.S. and Ph.D. degrees in applied physics from Cali-fornia Institute of Technology, Pasadena, in 1985 and1981, respectively.

He is a Professor of Electrical Engineering at theUniversity of California, Los Angeles (UCLA) and iscurrently conducting research projects in RF powerelectronics and thermal management, RF recon-figurable antennas, MEMS ultrasonic transducersfor biomedical imaging, shot noise suppression

in semiconductor devices, and THz electronics and optoelectronics. Beforejoining UCLA, he was a Program Manager at DARPA in Arlington, VA. Prior

to DARPA, he was with Massachusets Institute of Technology (MIT), LincolnLaboratory, Lexington, MA, where he conducted and managed research insolid-state science and technology.

Dr. Brown is a member of the American Physical Society.

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