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1

A Planar End-fire Circularly Polarized

Complementary Antenna with Beam in Parallel

with Its Plane

Wen-Hai Zhang, Wen-Jun Lu and Kam-Weng Tam

Abstract—Operation principle and design approach of a novel

planar, end-fire circularly polarized (CP) complementary

antenna is proposed. A vertically polarized printed magnetic

dipole and a horizontally polarized printed dipole are combined

on the same substrate, and a planar CP antenna with end-fire

beam in parallel with its plane is thus designed. Prototype

antennas centered at 5.80 GHz are then fabricated and measured

to validate the operation principle and the design approach. The

experimental prototype reported that the impedance bandwidth

(20log|S11| < -10 dB) is about 1.90%, from 5.75 to 5.86 GHz and

the 3-dB axial ratio (AR) bandwidth is about 14.48%, from 5.19 to

6.00 GHz. Therefore, the proposed design is applicable as a

low-profile, hand-held reader antenna in radio frequency

identification (RFID) systems.

Index Terms—Circularly polarized, end-fire, planar antenna,

complementary antenna, hand-held RFID reader.

I. INTRODUCTION

he circularly polarized (CP) antennas have been intensively

studied since 1940’s [1, 2]. The design approaches to

generate CP radiation can be basically categorized into five

distinctive types, according to their operation principles. The

first type is to use the superposition of complementary dipoles

[1-5], such as the combination of dipole/monopole and loop [1],

slotted cylindrical dipoles [3], printed strips and slots [4].

Generally, both broadside [1-4] and end-fire [5] CP radiation

beams can be realized by using these techniques. The second

one is to use the superposition of identically orthogonal dipoles.

The use of two orthogonally crossed dipole elements fed with

equal magnitude and 90° phase difference is a straightforward

way to generate broadside, CP radiations [6-9]. The third way

to generate CP radiation is to introduce the turnstile structures,

including the helices [10, 11] and the spirals [12-15]. In this

case, end-fire CP radiation beams can be obtained by using

relatively electrically large turnstile structures [10-15]. With

reference to planar spirals or helices configuration, the end-fire

beam is perpendicular to the plane of the antenna [13-15]. The

forth type is to excite two degenerate modes within a single

radiator, e.g., a patch [16, 17] or a dielectric resonator [18, 19],

by employing 90° hybrid couplers/dividers [16] or using

perturbations [17-19]. As similar to the previously described

three distinctive types of CP antenna design methods, the

Manuscript received June 24, 2015. This work is supported in part by FDCT

project 015/2013/A1, One-time Special Fund for PhD Student Support of

University of Macau and National Natural Science Foundation of China under grant no. 61471204.

W. H. Zhang and K. W. Tam are with the Department of Electrical and

Computer Engineering, Faculty of Science and Technology, University of Macau, Macao SAR, China (email:[email protected]; [email protected]).

W. J. Lu is with the Jiangsu Key Laboratory of Wireless Communications,

Nanjing University of Posts and Telecommunications, Nanjing 21003, China (email: [email protected]).

resulted radiation patterns of this method are also broadside.

The final one utilizes the traveling-wave or periodical

structures, such as the waveguide-fed horn antennas [20, 21],

the substrate integrated waveguide antennas [22-25], and the

metamaterial-based leaky wave antennas [26, 27]. Most of

them are electrically large and exhibit broadside CP beams. By

comparing the five types of CP antenna design approaches

according to their basic operation mechanisms and radiation

behaviors, it is found that to design a planar CP antenna with an

end-fire beam in parallel with its plane while keeping a planar

antenna structure, is a challenging task. Although an end-fire

CP beam in parallel with the major plane of the antenna can be

realized [25, 27], nonplanar configurations will be introduced

due to the inherent feed structure [5] or introducing pairs of

additional orthogonal elements [25, 27] .

On the other hand, due to broadside radiation characteristic,

the antennas are often assembled perpendicularly to the reader

to achieve front-directional radiation for handheld RFID

readers, which significantly increases the whole profile of

handheld reader [28, 29]. Recently, a new approach to design

planar, end-fire CP antenna having a beam in parallel with its

plane is studied [30]. Different from combining a pair of

orthogonally magnetic dipole together [30], a new planar

complementary antenna with a simpler structure is proposed in

this paper. The proposed antenna is combined with an aperture

and a printed dipole. To the best of our knowledge, this is the

first time using the complementary dipoles to achieve a planar

end-fire CP antenna with its beam in parallel with the plane.

This paper is organized as follows: In Section II, an equivalent

sources model is presented to deduce the dipole-aperture

combined antenna configuration, and the design guideline is

addressed as well. In Section III, the proposed antenna is

parametrically studied and optimal parameters are obtained. In

Section IV, prototypes are fabricated, measured and compared

to validate the design approach.

II. THEORY AND DESIGN GUIDELINE

Conceptual configuration consisting of a pair of paralleled

magnetic and electric dipole is shown in Fig.1 (a), which aligns

with the y-axis. The aperture element is equivalent to a virtual,

time-harmonic magnetic dipole. The electric-field at an

arbitrary observation point P in the far field can be obtained,

according to [31].

ˆ ˆcos cos sin0 04

jkrapertureI l

E E E j er

(1)

Where I0 is the amplitude, l is the length of dipole, η = 0 0

is the wave impedance, μ0 and ε0 are the permittivity and the

permeability of free space, respectively. On the other hand, the far field pattern of y-oriented electric

dipole is [31, 32]

ˆ ˆcos sin cos0 04

jkrdipoleI l

E E E j er

(2)

When the aperture and printed electric dipole are excited

with equal amplitude with a distance of d about 3λ/8 (λ is the

guided-wave wavelength, k = 2π/λ is the wave number). Here δ0

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indicates the temporal phase that caused by the current flowing

from the aperture to the printed dipole, and it equals to kd =

3π/4. The total far field of the complementary configuration

will be

( sin cos )

ˆ cos cos sin ( , )

ˆ cos sin cos ( , )

0

0 0

4

j kd

total aperture dipole

jkr

E E e E

fI lj e

r f

(3)

Where

( , ) cos sin cos sin sin cos3 3 3 3

4 4 4 4

f j (4)

When ,90 0 , the E-field along the +x-axis is

ˆ ˆ0 04

jkr

x

I lE j j e

r (5)

From (5) it is seen perfect circularly polarized radiation

characteristic can be obtained at the +x-direction, i.e., the

end-fire direction of the planar, aperture-dipole combined

antenna.

To more clearly reveal the mechanism for CP generation of

the proposed antenna, radiation patterns of a single microstrip

magnetic dipole at the resonant frequency is simulated in Fig. 2.

It is observed that the antenna produces an “8”-shaped radiation

pattern in the azimuth plane (xy-plane, H-plane), while an

“O”-shaped radiation pattern in the elevation plane (xz-plane,

E-plane). The result proves that the antenna resonates as a

vertically polarized (i.e., z-polarized) magnetic dipole

operating at its dominant, one-half wavelength mode. It

exhibits radiation maxima at the end-fire, +x-direction [33].

For a y-polarized electric dipole operating at its dominant mode,

it should have a complementary radiation pattern and

orthogonal polarization with the magnetic dipole. Therefore, if

90-degree phase difference is provided, end-fire, CP

characteristic can be achieved while a planar structure is

maintained.

x

Yz

Magnetic dipole Electric dipole E-field

P

r

Aperture

d

o

X

Y

Z

φ

θ

Dipole

(a)

x

Y

Feed point

Top layer Bottom layer

λ/4

Substrate

Shorting wall

Aper

ture D

ipole

(b)

Shorting

wall

Coaxial cable

xy

z

Top layer

Bottom layer

Substrate

(c)

Fig. 1. Planar end-fire complementary CP antenna, (a) equivalent sources

model, (b) conceptual design, and (c) sectional view of the proposed antenna.

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0

XY

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0

XZ

(a) (b)

Fig. 2. Simulated radiation patterns at 5.8 GHz. (a) xy-plane; (b) xz-plane.

Based on the theoretical and numerical analysis, the basic

structure of the planar CP complementary antenna can be

deduced: In Fig. 1(b) and (c), three edges of magnetic

microstrip dipole are shorted with one left opened [33] to form

an aperture. The aperture serves as a virtual magnetic

microstrip dipole which operates at one-half wavelength mode

with width of one-quarter wavelength [34]. A coaxial cable

probe is used to excite the magnetic dipole and is located near

its E-current’s antinode [30]. A pair of 3λ/8, broadside coupled

stripline [35] is placed between the aperture and printed electric

dipole to introduce a proper phase difference. In this way, a

planar, end-fire CP beam in parallel with the substrate’s plane

will be resulted in.

III. ANTENNA GEOMETRY AND PARAMETRIC STUDY

The geometry of the proposed antenna is shown in Fig. 3.

The proposed antenna with a total size of 38 mm×33.5 mm is

designed on substrate with relative permittivity of εr=2.65, tan δ

= 0.001, thickness h=2.0 mm.

The direction of CP of the proposed antenna can be

determined by the probe current’s path/loop [30]. Suppose the

probe is fed to the top layer of the magnetic dipole, and the

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current flows from the probe, then to the top layer of the

broadside coupled stripline, the top arm of the printed dipole.

To maintain the continuity of the current path/loop, it is

supposed the displacement current flows to the +x-direction,

and then turns back to the bottom arm printed dipole, the

bottom layer of the broadside coupled stripline, and finally

flows into the bottom layer of the magnetic dipole. If such a

virtual probe current's path obeys a right handed helix direction

with thumb point at +z-direction, a right handed circularly

polarized (RHCP) antenna is obtained. Otherwise, a left handed

circularly polarized (LHCP) one is resulted instead. In this

paper, a RHCP antenna is designed and investigated.

In order to increase the bandwidth, a printed fatter dipole is

adopted [36]. The initial parameters are determined according

to the design guideline and the empirical results in [30, 33] and

tabulated in Table I. In order to simultaneously achieve the

optimized impedance bandwidth and axial ratio characteristics,

a parametric studies are carried out to determine the optimal

parameters. The antenna is modelled by using Zeland’s IE3D

v12.0 with method-of-moment codes and finite substrate is

considered in simulations. When one parameter is studied, the

others are kept unchanged.

L

W1

L1

L2

W2

g

d

α

D

g1

W

X

Y

Z

Feed point

Metallic via

Top layer

Bottom layer

Fig. 3. Geometry of the proposed antenna.

TABLE I

ANTENNA PARAMETERS

Parameters Initial/Optimal Parameters Initial/Optimal

L(mm) 36.0/38.0 g(mm) 0.6/0.4

W(mm) 33.5/33.5 L1(mm) 11.9/13.5

W1(mm) 7.9/7.9 W2(mm) 7.9/16.0

D(mm) 0.6/0.6 L2(mm) 4.0/6.0

d(mm) 2.0/2.0 α 80°/84°

g1(mm) 0.3/0.3

5.2 5.4 5.6 5.8 6.0 6.2

-40

-35

-30

-25

-20

-15

-10

-5

0

|S11|(

dB

)

Frequency(GHz)

α=80° α=82° α=84° α=86°

Fig. 4. Effect of the angle of magnetic dipole α.

The effect of angle α on the antenna performance is studied

firstly in Fig. 4. It is observed that it has a significant effect on

impedance characteristic, the center frequency shifts up for a

larger α. Fig. 5 shows the effect of the distance of feed point.

As we can see better impedance characteristic can be achieved

when d equals to 2 mm. However, both of them have slight

effect on 3-dB AR bandwidth.

Fig. 6 shows the effect of length of broadside coupled

stripline on the performance of the proposed antenna. It is

observed that with the increasing of L1, it improves 3-dB

bandwidth significantly and the minimized AR shifts up

gradually. The effect of the width and the length of the printed

dipole are also studied. In Fig. 7, with the increasing of L2, the

minimized AR shifts down and better 3-dB AR bandwidth is

achieved when L2 equals to 6 mm. However, If L1 and L2 decrease continuously, AR will become worse. Fig. 8 shows

that when the length increases, 3-dB AR bandwidth becomes

much wider and minimized AR shifts down gradually.

In order to show mechanism for CP generation, simulated

surface current distributions at 5.8 GHz are investigated and

shown in Fig. 9. It is seen both of the magnetic and electric

dipoles are operating at their dominant mode, as theoretically

predicted in the previous section. When a 90-degree phase

difference properly introduced, end-fire CP radiation

characteristic can be achieved.

5.0 5.2 5.4 5.6 5.8 6.0

-20

-15

-10

-5

0

|S11|(

dB

)

Frequency(GHz)

d=1 mm

d= 1.5 mm

d= 2 mm

d= 2.5 mm

Fig. 5. Effect of the distance of feed point d.

5.0 5.2 5.4 5.6 5.8 6.0

0

1

2

3

4

5

6

Axia

l R

atio

(dB

)

Frequency(GHz)

L1=11.5 mm

L1=13.5 mm

L1=15.5 mm

Fig. 6. Effect of the length of broadside coupled stripline L1.

5.0 5.2 5.4 5.6 5.8 6.0

0

1

2

3

4

5

6

7

8

Axia

l R

atio

(dB

)

Frequency(GHz)

L2=4 mm

L2=6 mm

L2=8 mm

Fig. 7. Effect of the width of printed electric dipole L2.

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5.0 5.2 5.4 5.6 5.8 6.0

0

1

2

3

4

5

6

7

8

Axia

l R

atio

(dB

)

Frequency(GHz)

W2=14 mm

W2=16 mm

W2=18 mm

Fig. 8. Effect of the length of printed electric dipole W2.

Fig. 9. Surface current distribution at 5.8 GHz.

IV. SIMULATED AND EXPERIMENTAL RESULTS

Based on the parametric studies, prototypes of the proposed

antenna, shown in Fig. 10, are fabricated and measured to

verify the design approach. The reflection coefficient is

measured by using an Agilent N5230A vector network analyzer.

The radiation patterns, gain, efficiency and AR are measured in

a Satimo’s Starlab near-field antenna measurement system.

Fig. 11 shows the measured and simulated reflection

coefficient of the proposed antenna. The measured -10-dB

reflection coefficient bandwidth is from 5.75 to 5.86 GHz,

about 1.90% in fractional, while the simulated one is from 5.70

to 5.91 GHz, about 3.62% in fractional. The mismatch is caused

by the fabrication errors and measurement errors caused by the

non-ideal probe and the SMA connector. Fig. 12 shows the

measured and simulated 3-dB AR bandwidth of the proposed

antenna. The measured AR bandwidth is from 5.18 to 6.00 GHz,

which is larger than 14.48% in fractional, while the simulated

one is from 5.18 to 5.98 GHz, which is about 14.34%. Both of

them are wider than the corresponding impedance bandwidth.

The measured and simulated gain at +x-direction and efficiency

of the proposed antenna are illustrated in Fig. 13 and Fig. 14,

respectively. It is observed that the proposed antenna exhibits

stable gain at +x-direction of about 2.3 dBic within its

impedance bandwidth. The measured gain is in good agreement

with the simulated one in the low frequency regime and some

discrepancies can be observed in the high frequency band, i.e.,

above 5.80 GHz. The average measured efficiency is about

78%, slightly lower than the simulated one. The measured and

simulated results, including the reflection coefficient, AR, gain

and efficiency are in good agreement with each other.

The measured and simulated radiation patterns in xz-plane

and xy-plane at 5.65 GHz (the minimized AR) and 5.80 GHz

(the center frequency of impedance band) are plotted in the Fig.

15 and Fig. 16, respectively. It can be observed that the

simulated results are consistent with the measured ones. The

proposed antenna has a +x-direction, end-fire CP radiation

pattern, as predicted in the above. In both principal planes,

symmetrical radiation patterns and wide angle AR

characteristics are obtained. The measured half-power beam

widths of the proposed antenna is 155° and 75° in xz-plane and

xy-plane, respectively, while the corresponding 3-dB AR beam

width is 110° and 80° at 5.80 GHz, which is desirable for

wide-coverage RFID applications [37]. The measured and

simulated half-power and 3-dB AR beam widths of both

frequencies are summarized in Table II.

(a) (b)

Fig. 10. Top and bottom-view photographs of the prototype. (a) Top view; (b)

bottom view.

5.0 5.2 5.4 5.6 5.8 6.0

-20

-15

-10

-5

0

|S11|(

dB

)

Frequency(GHz)

Simulated

Measured

Fig. 11. Measured and simulated reflection coefficient of the proposed antenna.

5.0 5.2 5.4 5.6 5.8 6.0

1

2

3

4

5

Axia

l R

atio

(dB

)

Frequency(GHz)

Simulated

Measured

Fig. 12. Measured and simulated axial ratio at +x-direction (θ=90°, φ=0°) of

the proposed antenna.

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5.0 5.2 5.4 5.6 5.8 6.0

-8

-6

-4

-2

0

2

4

Gai

n(d

Bic

)

Frequency(GHz)

Simulated

Measured

Fig. 13. Measured and simulated gain of the proposed antenna.

5.0 5.2 5.4 5.6 5.8 6.0

0

20

40

60

80

100

Eff

icie

ncy

(%)

Frequency(GHz)

Simulated

Measured

Fig. 14. Measured and simulated efficiency of the proposed antenna.

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0

LHCP,simulated RHCP,simulated

LHCP,measured RHCP,measured

XZ

(a)

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0



X

Y

(b)

Fig. 15. Measured and simulated radiation patterns at 5.65 GHz. (a) xz-plane;

(b) xy-plane.

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0



XZ

(a)

-40

-30

-20

-10

0

0

30

60

90

120

150

180

210

240

270

300

330

-40

-30

-20

-10

0



X

Y

(b)

Fig. 16. Measured and simulated radiation patterns at 5.80 GHz. (a) xz-plane;

(b) xy-plane.

A comprehensive comparison with other typical CP antennas

is also presented. The comparisons in terms of operation

principles, beam directions, geometry and 3-dB AR bandwidth

are tabulated in Table III. It is seen that most planar antennas

can only achieve CP beam which is perpendicular to the plane

except [30], and the new complementary antenna has an

end-fire, CP beam in parallel with its plane. Furthermore, the

CP operation bandwidth is increased to 14.48% compared to its

planar counterpart using orthogonally combined magnetic

dipoles [30]. This is possibly caused by the mutual cancellation

of reactance characteristics of the complementary dipoles [38].

TABLE II

BEAM WIDTHS OF THE ANTENNA

Frequency

Half-power beam width

(Simulated/Measured)

3-dB AR beam width

(Simulated/Measured)

xz-plane xy-plane xz-plane xy-plane 5.65 GHz 177°/156° 93°/75° 103°/106° 74°/87°

5.80 GHz 165°/155° 68°/75° 104°/110° 94°/80°

TABLE III FIGURE OF MERIT COMPARISONS

V. CONCLUSION

In this work, a novel planar dipole-aperture combined

antenna has been proposed, fabricated and tested. An end-fire

CP beam in parallel with its substrate plane is obtained. Our

study shows that the proposed antenna can achieve an

impedance bandwidth of 1.90% and a 3-dB AR bandwidth of

Reference Operation principle Beam Direction Beam relative to the substrate’s

plane Geometry 3dB-AR Bandwidth

[3] Complementary dipoles Broadside Perpendicular Non-planar Not available

[5] Complementary dipoles End-fire In parallel with Non-planar 12.50%

[10] Turnstile structures End-fire Perpendicular Non-planar Not available

[14] Turnstile structures End-fire Perpendicular Planar 6.70%

[21] Orthogonal modes within a radiator Broadside Perpendicular Non-planar Not available [22] Orthogonal modes within a radiator Broadside Perpendicular Non-planar 17.39%

[26] Orthogonal modes within a radiator End-fire Perpendicular Planar Low band:＜0.56%

Upper band:＜0.51% [27] Orthogonal dipoles End-fire In parallel with Non-planar 33.00% [30] Orthogonal dipoles End-fire In parallel with Planar 9.24%

[39] Orthogonal dipoles Broadside Perpendicular Non-planar 39.00%

[40] Orthogonal modes within a radiator Broadside Perpendicular Planar 0.65%(2dB-AR

bandwidth)

This work Complementary dipoles End-fire In parallel with Planar 14.48%

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14.48%. The measured results exhibit good agreement with the

predicted ones. The principle of end-fire CP beam realization

has been demonstrated and experimentally validated. Therefore,

the proposed antenna is advantageous for low-profile, handheld

RFID reader applications .

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