antenna effects on indoor obstructed wireless channels and...

International Journal of Wireless h~formation Networks, Vol. I, No. 1, 1994

Antenna Effects on Indoor Obstructed Wireless Channels and a Deterministic Image-Based Wide-Band Propagation Model for In-Building Personal Communication Systems

C . M . P e t e r H o , 1 T h e o d o r e S . R a p p a p o r t , 1'2 a n d M . P r a b h a k a r K o u s h i k I

In this paper, the effects of antenna pattern and antenna polarization on indoor obstructed (OBS) wireless channels are investigated experimentally. Our results show that linearly polarized (LP) directional antennas are more effective in combating multipath components than other types of antennas in OBS channels, The measurement results are verified with a deterministic wide-band propagation model based on image theory that takes into account the effects of building geometry, antenna pattern, and antenna polarization. Preliminary prediction results show that the propagation model holds promise for accurate and efficient in-building wireless channel prediction.

KEY WORDS: Indoor radio propagation: ray tracing algorithm: wide-band channel measurement; path loss.

1. I N T R O D U C T I O N

Interest in high data rate in-building wireless communication systems has led to many experiments char- acterizing in-building wide-band channels [1-8]. Based on measurement results, typical path loss and rms delay spread values in different indoor environments are known. However, the effects of antenna pattern and antenna polarization on indoor wide-band channels are less well understood. The transmit or receive antenna acts as a spatial filter to different multipath components, which have different propagation path lengths and thus different propagation delays and signal strength. The departure and arrival angles of a multipath component depend on the geometry of the environment. Thus, the transmit or receive antenna attenuates or amplifies each multipath component based on its departure/incident angle and polarization state. In other words, different types of antennas can induce different channel impulse responses

~Mobile and Portable Radio Research Group, Bradley Department of Electrical Engineering, Virginia Tech, Blacksburg, Virginia.

2Correspondence should be directed to Prof. Theodore S. Rappaport, MPRG, Department of Electrical Engineering, Virginia Tech, 340 Whittemore Hall, Blacksburg, Virginia 24061-0111.

in the same environment. For example, directional cir- cularly polarized (CP) antennas always reduce rms delay spread when compared to omnidirectional and directional linearly polarized (LP) antennas in LOS channels [7]. This is because significant late arriving components in LOS channels are primarily single-hop components, with nearly orthogonal polarization states as the CP transmitted signal. Hence, a CP receiver (with the same sense as the CP transmitter antenna) can filter most late arriving components. This paper reports wide-band channel measurements made in a modern office at 2.45 GHz using 25 antenna combinations. Measurements results are used to analyze the effects of antenna pattern and antenna polarization on indoor OBS channels. The measurement procedure and results are presented in Sections 2 and 3, respectively.

Although typical channel characteristics are known, and a few useful statistical indoor propagation models are available [10-12], more accurate signal prediction techniques are necessary to provide optimum capacity and maximum coverage for future indoor wireless communication systems [9]. Site-specific channel impulse response prediction methods, which can incorporate geometrical information about the environment, can be a viable alternative because measurements are usually ex-

61 1068-960519410100-0061507.0010 �9 1994 Plenum Publishing Corporation

62 Ho, Rappaport, and Koushik

pensive and time-consuming. Research in site-specific propagation models includes both microcellular [13-17] and in-building environments [18-27]. Based on the reported results, site-specific propagation models show promise for accurate signal prediction. This paper presents a theoretical propagation model to verify the measurement results presented here.

Different techniques are in use to implement ray tracing. "Brute force" ray tracing methods, which sim- ply launch rays in all directions in three dimensions and exhaustively trace their paths, are used in the works of Seidel and Rappaport [18] and Honcharenko et al. [24] to account for all possible propagation paths for in- building propagation predictions. An alternative approach to implementing ray tracing is based on image theory [14, 16, 17, 19, 20, 22]. With the exception of the latter two, most models ignore antenna polarization or assume vertical polarization. As shown by Rappaport and Hawbaker [7] and by Ho and Rappaport [8], antenna pattern and polarization have significant effects on path loss and rms delay spread of wireless channels. Ex- cept for Refs. 18 and 20, none of the previous works have compared predicted and measured channels on a location-by-location basis. Here we present a three-dimensional image-based ray tracing algorithm that takes into account the effects of surrounding walls and buildings, obstacles inside buildings, antenna patterns, and antenna polarizations for predicting path loss, delay spread, etc. The model computes the magnitude and the phase of each multipath component and is capable of tracing them up to any number of reflections. The algorithm has been used to predict channel impulse responses inside a modem office building using different antennas, and the predicted and measured channels have been compared on a location-by-location basis. Sections 2 and 3 present the measurement procedure and the results obtained. Sections 4 and 5 discuss the ray tracing algorithm and the predicted results. Preliminary results show good agreement between measured and predicted results.

2. M E A S U R E M E N T P R O C E D U R E

Traditionally, wide-band channels have been char- acterized using signal attenuation (path loss) and time dispersion parameters such as rms delay spread as a function of transmitter-receiver (T-R) separation, the type of indoor environment, and the topography of the measurement link. As shown previously [33, 34], the bit error performance of indoor wireless communication systems is a strong function of rms delay spread. A possible solution to minimize delay spread is to employ dif-

ferent antenna patterns and polarizations [7]. One of the goals of the present measurement campaign is to determine optimal antenna combinations in obstructed indoor channels.

The measurements reported here were performed on the second floor of a modem two-story office building of Teledyne. The measurement location included many soft partitions, some concrete walls, and standard office furniture. The distance between the ceiling (standard drop made of foam tiles) and the floor is approximately 3 m. The building layout and the measurement locations are given in Fig. 1. The receiver was mounted on a cart and was moved to different locations, while the transmitter was fixed within a soft-partitioned office. For all results presented in this paper, there was no direct line-of-sight between the transmitter and the receiver at each measurement location. Typical obstacles between the transmitter and the receiver were soft partitions and office furniture.

The measurement setup is a bi-static time-domain channel sounder and is similar to that described previously [7]. A pulse with an absolute width of 33 ns (rms width 13 ns) is mixed with a 2.45-GHz carrier, ampli- fied, and transmitted by the antenna. A relatively simple square law detector is used at the receiver. Both the transmitter and the receiver antennas were positioned 1.8 m above ground. The main beams of the directional antennas were lined up on boresight as if a line-of-sight path existed. The antennas used in the measurements are listed in Table I along with single letter codes which will be used in the rest of the paper for reference purposes. For example, fb represents the combination of a CP helical antenna at the transmitter and a VP omnidirectional discone antenna at the receiver. A total of 25 such combinations is possible since 5 different antennas have been used.

Measurements were performed at six different receiver locations (locations B, H, J, K, L, and M in Fig. 1) and T-R separation ranged from 15 to 45 m. Data from one measurement location were discarded due to an apparent error in the recorded attenuator setting of the measurement. For each antenna combination at each location, the receiver was moved over a measurement track of 1 m, and 10 power delay profiles were recorded at equally spaced points on the l-m track. Although the data are limited for broad-based conclusions, the preliminary results and evaluation criteria of antenna performance provide some direction for future research in indoor wireless system design.

The noise threshold is assumed to be the maximum value of the last 10% of the measured multipath profile. Moreover, a peak signal-to-noise ratio of 10 dB is re-

Antenna Effects on Indoor Obstructed Wireless Channels 63

,, Receiver Locations aG

Measurement Locations ~

It L

Using Multiple Antenna Combinations: B, H, Y. K, L, M -- ~ ' ~

Transrm

On (0,0,0)

Fig. 1. Three-dimensional view of Teledyne's second-floor office.

quired for a profile to be deemed suitable for data pro- cessing. Channel parameters such as path loss, rms delay spread, and excess delay are computed from each instantaneous measured profile at each location [7].

persion parameters computed from both methods are very similar [7]. We shall refer to the static rms delay spread herein after, bearing in mind that we are referring to instantaneous profiles and not a spatially averaged power delay profile.

3. M E A S U R E M E N T RESULTS

The measured results are processed by a method known as parameter averaging [7] which can provide information about the local behavior (small-scale variation) of the channel, Channel parameters at each measurement location are computed by averaging the mean excess delay computed from individual measured profiles taken within a local area rather than the more tra- ditional spatial averaging used previously [11]. The latter method provides information about how bad the channel spreads the transmitted data [35] and the rms delay spread computed from this method has been re- ferred to as the static rms delay spread. For indoor channels, it has been shown experimentally that the time dis-

3.1. Delay Spread Results

3.1.1. rms Delay Spread (crT) and Maximum Excess Delay (MED) < 10 dB

The distributions of rms delay spread for different antenna combinations were found to be somewhat similar; hence, the maximum excess delay (MED) values (10 dB down from maximum signal level) for different antenna combinations are also presented to give additional insights into the multipath mechanisms and the time delays of significant multipath energy. The MED is defined as the maximum excess delay at which a multipath component is received with an amplitude within 10 dB of the peak of the profile. The descriptive statistics of rms delay spread and MED for all antenna corn-

Antenna Antenna code

Table I. Antennas Used in the Measurements

Measured Polarization gain (dB)

HPBW azimuth/vertical

(degrees) Remark

Discone b VP 2 omni/60 Discone c HP 2 60/omni Comer reflector d VP 8 50/NA" Comer reflector e HP 8 NA/50

Helical f CP 12 35/35

Rotated antenna b 90 degrees

Rotated antenna d 90 degrees

"NA = not available.

64

binations are given in Tables II and III, respect ively.

Rms delay spread for all antenna combina t ions at all

locations ranges from 17 to 71 ns. M E D for all antenna

combinat ions ranges f rom 51 to 364 ns. In open-p lanned

office envi ronments , strong late arriving componen t s are

unlikely because signal componen t s must penetrate

through office cubicles and " w a v e - g u i d e " - l i k e effects

do not exist . Therefore , the spread o f the rms delay

spread value is rather small. Antenna combina t ions bf ,

cf, dd , df , and f f have the smallest average rms delay

spread over all locations. Except for antenna combina-

tion i f , all antenna combina t ions with small rms delay

spread values also have small mean excess delay values.

Moreover , antenna combinat ions c f a n d d d also have a

small standard deviat ion of rms delay spread. To con-

firm that antenna combina t ions c f a n d d d offer bet ter de-

lay spread per formance , the M E D values are studied

closely. From Tables II and III, it is found that antenna

combinat ions with small rms delay spread values also

have small M E D values. The differences be tween M E D

values for different antennas are larger than the differ-

ences be tween the rms delay spread values. The C D F

curves o f rms delay spread and M E D for some selected

Table II. Descriptive Statistics for rms Delay Spread of All Antenna Combinations

Standard Antenna Average deviation Minimum Maximum

combination (ns) (ns) (ns) (ns) Median

(ns)

bb 37.78 7.23 19.40 50.30 37.20 bc 40.55 6.67 23.50 52.60 41.40 bd 34.94 8.09 23.20 47.20 34.00 be 42.86 7.71 29.70 57.50 44.10 bf 33.95 5.62 23.20 45.90 32.60 cb 44.13 9.41 30.60 71.20 41.30 cc 38.37 7.04 23.30 54.60 38.50 cd 35.23 8.21 19.10 56.10 35.00 ce 34.85 6.17 18.10 48.50 34.20 cf 31.29 5.62 17.80 40.90 30.90 db 36.51 9.40 20.60 56.40 36.20 dc 39.71 6.37 25.10 52.50 39.90 dd 33.64 5.30 21.50 43.40 34.10 de 36.62 6.77 24.00 49.90 35.20 df 33.14 8.00 18.50 50.30 33.80 eb 41.66 7.55 27.60 59.60 40.00 ec 38.81 7.03 22.00 52.40 39.60 ed 37.03 9.26 18.10 57.70 36.70 ee 36.50 6.21 23.10 49.00 36.20 ef 38.84 6.80 27.70 57.70 38.00 Jb 41.22 7.35 23.20 55.50 42.40 fc 45.07 8.77 29.20 64.50 45.30 fd 37.65 5.55 22.00 48.10 38.20 fe 37.75 9.72 20.00 58.10 38.10 ff 32.17 8.87 17.00 49.40 31.10

Ho, Rappaport, and Koushik

Table III. Maximum Excess Delay (MED) (10 dB Down) Results (All Locations)

Standard Antenna Average deviation Minimum Maximum Median

combination (ns) (ns) (ns) (ns) (ns)

bb 129.81 38.39 79.60 260.70 123.50 bc 144.43 38.62 86.40 256.80 144.00 bd 123.97 39.98 60.50 200.20 ! 11.30 be 136.68 32.11 72.30 208.50 127.40 bf 102.90 25.55 65.90 152.80 93.90 cb 149.95 37.61 78.60 248.50 133.80 cc 113.77 37.75 53.70 220.70 110.40 cd 114.13 47.40 64.00 304.20 100.10 ce 102.59 25.91 61.00 170.90 102.10 cf 93.62 23.10 54.70 168.90 90.30 db 130.22 57.88 50.80 304.70 131.30 dc 152.51 49.39 89.80 275.40 131.30 dd 99.45 37.77 50.80 229.00 92.30 de 122.23 34.27 89.40 232.40 112.80 df 107.36 31.76 59.10 181.60 101.10 eb 145.44 43.21 57.10 288.60 138.70 ec 123.29 48.08 52.70 267.60 127.90 ed 116.02 41.68 51.30 259.30 106.90 ee 119.36 42.37 56.60 279.30 120.60 ef 121.23 43.46 58.60 268.10 111.30 ./b 135.70 44.10 57.10 244.10 126.50 fc 154.91 52.32 73.20 363.80 150.40 fd 107.54 32.39 51.30 195.80 110.80 fe 127.99 53.46 59.10 270.50 114.30 ff 108.76 44.29 55.20 244.60 99.60

antenna combina t ions are given in Figs. 2 and 3, respectively. In Fig. 3, it is shown that antenna combinations cfand dd have 3 to 6 ns smaller rms delay spread than when copolarized omnidirectional antennas are used at both the transmitter and the receiver, for a fixed percentage o f time. Antenna combination fc has more than 10 ns rms delay spread larger than that o f the best antenna combinations, for a fixed percentage o f time. Sire-

1.0

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m et [ ~ .... . . i ~ . . ......... ddl I'' f ," ~r ~

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6 12 18 24 30 36 42 48 54 60 RMS Deloy Spread (ns)

CDF of rms delay spread over a]] locations for some selected antenna combinations.


'~ I :j J

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0.8 ./ /

~"~0.7 :'" t" :'"

0.5 I ~,~ 0.4 I-

~0.3 0.2

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0.0 O 40 80 120 160 200 240 280 320 360 400

Maximum Excess Delay (I OdB down) ~dED (ns)

Fig. 3. CDF of MED over all locations for some selected antenna combinations.

ilarly, antenna combinations c fand dd have 20 to 40 ns smaller MED than bb.

Antenna combination f f has a small rms delay spread, but the standard deviation of the rms delay spread values is large. This can be attributed to the fact the reflections and transmissions can depolarize CP waves. It appears that the CP signal is vulnerable to depolarization through penetrations of obstacles, while the LP signal is not. Hence, even though antenna f has the smallest half-power beamwidth (HPBW) [32] in the azimuth direction, the rms delay spread and MED for f fa re about the same as that of antenna combination dd. This is because late arriving components can be copolarized with respect to the receiver while there are polarization mismatches for the early arriving components. Thus, antenna combination f f cannot give better delay spread performance than other directional antenna pairs, despite the fact that CP antennas are more directive than LP directional antennas in our measurements. The fact that MED f o r f f i s small, but the standard deviation of MED is relatively large, further strengthens this observation. Therefore, use of directional CP antennas at both the transmitter and the receiver in OBS channels cannot reduce rms delay spread although it can combat multipath in LOS channels [7].

From the values of rms delay spread and MED listed in Tables II and III, it can be observed that an omnidirectional transmitter with a directional receiver has a smaller delay spread than a directional transmitter with an omnidirectional receiver. This implies that the channel is not symmetrical. (This does not violate the reciprocity principle, which states that the received volt- age is the same if the roles of the transmitter and the receiver are switched if the antenna locations and ori- entations are the same.) More scatterers are located on one side of the channel than the other. An example of an asymmetrical channel is shown in Fig. 4. Figure 4a

Transmitter Receiv~ Reflector

Reflector

F ig . 4. An example of an asymmetrical channel.

depicts a multipath component arriving at the back of the receiver antenna (omnidirectional), resulting in a strong multipath component. Figure 4b shows the same configuration as in Fig. 4a, but with the antennas inter- changed. The directional receiver antenna greatly attenuates the strong multipath component arriving at the receiver. Therefore, delay spread in this case is smaller than that of Fig. 4a.

The average rms delay spread for antenna combinations bb and cc are almost the same. Although antenna c is slightly directive in the azimuth plane, it has two main " l o b e s , " and therefore the directivity does not help to reduce the rms delay spread. The average MED value for cc is slightly lower than that of bb. An- tenna c can be modeled as a misaligned directional antenna since there are two main lobes in the azimuth direction, and therefore it is impossible to align the transmitter and receiver antennas perfectly to give small delay spread. There does not seem to be much difference for linear polarization for one sense (VP) over the other (HP). Similar results were found in Refs. 1 and 7.

When an omnidirectional discone antenna is used at the transmitter (regardless of vertical or horizontal orientation), a helical antenna at the receiver always gave the smallest average rms delay spread. This can be explained as follows: The directivity of the CP directional antenna is slightly higher than the LP directional antenna. If most significant scatters do not generate cross-polarization, the received signal for a LP transmitted signal will always be linearly polarized. (How- ever, the orientation of the LP received signal can be different from that of the transmitted one.) Since all LP waves can be represented by two orthogonal CP waves, the polarization mismatch factor for each LP multipath component at a CP receiver antenna is always 0.5. In other words, if scatterers do not generate cross-polarization, then the polarization mismatch for all multipath components will be the same. Therefore, for a helical antenna at the receiver, the antenna pattern but not the polarization can significantly affect the relative strength of multipath components.

If a discone antenna is used at the transmitter, a

66 H o , R a p p a p o r t , and K o u s h i k

cross-polarized discone receiver antenna always gives higher delay spread than when the same omnidirectional discone is used at both the transmitter and the receiver. For cross-polarized antenna pairs, multiple-reflected late arriving components become copolarized with the receiver, and hence the rms delay spread and MED values for cross-polarized discone antenna pairs (bc or cb) are high.

Antenna combinations using antenna d at the transmitter generally have smaller average rms delay spread and MED values than antenna combinations using antenna e at the transmitter. A possible reason for this is the E-plane HPBW is larger than the H-plane HPBW for antenna d. (Since antenna e is obtained by rotating antenna d by 90 degrees, the HPBW in the azimuth direction is larger than HPBW in the vertical direction for antenna e.) Since the ceiling height is much smaller than the T - R separation, most multipath components depart from the transmitter and arrive at the receiver in the azimuth direction. Azimuth directivity can reduce some late arriving components, but vertical directivity cannot. Similarly, the average MED value fo r fd is smaller than that of fe. Since the polarization mismatches between CP and HP, and between CP and VP, are the same, different average rms delay spread and MED values for antenna combination f d and fe are due to the difference in antenna patterns between d and e.

From our measurements, LP directional antennas can combat indoor multipath more effectively than other antenna combinations. However, the performance of the directional antennas seems to be quite sensitive to the alignment of the antennas. A similar observation is found for the simulation for directional antennas in Sec- tion 5. Figure 5 shows the local variation of MED for antenna combination dd at locations L and B. The MED values for dd are less than 100 ns for the first few pro-

files. However, MED increases to more than 150 ns as the receiver moved along the I-m track. This may be attributed to the fact that antenna alignment was good at the beginning of the measurements. As the receiver moved, the alignment of the antennas became worse, causing higher MED.

3.1.2. Consistency of Delay Spread Performance at Different Locations

Overall rms delay spread and MED results for different antennas were discussed. This section presents consistency of performance of different antenna combinations at different locations. Table IV gives delay spread results of some antenna combinations at measurement location M; a~ is the average rms delay spread at location M. The average is computed from the instantaneous rms delay spread values of each of the ten profiles. The ~,(-) operator compares a time dispersion parameter for a particular antenna combination with the overall average of all antenna combinations. For example, ~,(a~) is a measure of how far o~ of each antenna combination is from the average rms delay spread computed over all antenna combinations at location M, and is given by

X ( a ~ ) = ( o , - #x~.)/a~. ( I )

where X(a,) is the normalized distance of the averaged rms delay spread from the average over all antenna combinations at location M, ~a=t is the average of the averaged rms delay spread of all antenna combinations at location M, and trot t is the standard deviation of the averaged rms delay spread of all antenna combinations at location M.

Max MED in the fourth column of Table IV is the largest maximum excess delay (10 dB down) of the ten

180

160

~ 140- "~120

1~ I

~ 80 ~ �9 .~ 60'

4o'

20

I

- d = L='~ I Locotion L

I

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I .......... / ] ""'....%

t j , \ l \\

I I

4 5 8 7 8 g Io Profile Number Fig. 5. Variation of MED within a local area for antenna combination

dd.

Table IV. Delay Spread Results of Some Antenna Combinations at Location M (T-R Separation -- 24 m)

Max Antenna MED ;k(Max MED

combination o,(ns) h(a,) (ns) MED) (ns) ;k(MED)

bb 39 0.2 196 0. I 132 cc 32 - 0 . 8 142 - 0 . 7 79 cf 27 - 1.6 98 - 1.4 76 dd 35 - 0 . 4 124 - 1.0 97 ee 38 0.1 138 - 0 . 8 116 f f 36 - 0 . 2 245 0.9 118 Maximum 51 364 192 Minimum 27 98 76 Average 38 186 122

Standard deviation 6 65 30

0.3 - I . 4 - 1 . 5 - 0 . 8 - 0 . 2 - 0 . 1


profiles for each antenna combination at location M. Hence, column four is the worst-case MED of all profiles of each antenna combination. ;k (Max MED) is the normalized distance of Max MED of each antenna combination from the average largest maximum excess delay over all antenna combinations and is obtained in a similar fashion as k(a~). MED in column six is the averaged maximum excess delay (10 dB down) of all profiles for each antenna combination, k (MED) in column seven is a normalized distance computed from column six via a similar method as k ( a , ) and k (Max MED). If a number in the third, fifth, and seventh columns is negative, then it is below the ensemble average of columns two, four, or six, respectively. For good channels (with small intersymbol interference), we want rms delay spread and maximum excess delay (10 dB down) to be as small as possible and hence the numbers in columns three, five, and seven to be as negative as possible. To find good antenna combinations, we want to determine if there are any antenna combinations which cause columns three, five, and seven of Table IV to be consistently negative while keeping path loss at a minimum, and therefore we need to look at results from all locations.

It turns out that a few antenna pairs have small delay spread [i.e., k(aT) < --0.3 and k(Max MED) < - 0 . 3 and k (MED) < - 0 . 3 ] at all locations. They are antenna combinations bf, ce, cf, dd, and fd. Although the average rms delay spread f o r f f i s small, the performance of f f at all locations is not consistent. The rms delay spread and maximum excess delay (10 dB down) of cfare not always the smallest at any location, but this antenna combination provides delay spreads which are consistently close to the minimum at each location. Hence from our results, it seems that an omnidirectional HP transmitting antenna and a directional CP receiving antenna is the antenna combination that provides the most consistency in minimizing delay spread for all obstructed locations. From our limited data, it is clear that some antenna combinations work consistently better than others at mitigating delay spread. Based on the measurement results, antenna combinations bf, cf, dd, and f d give the smallest rms delay spread and MED over all locations. Therefore, it appears that these antenna combinations are more robust in combating muttipath than other antenna combinations.

3.2. Path Loss (PL') Results

In general, the system path loss PL ' (which is defined as the difference between the received power and the transmitted power) for most of the antenna combi-

nations is greater than the path loss of an isotropic radiator in free space separated by the same T - R separation distance. This is sensible since the signal undergoes attenuation through obstacles and hence much energy is lost during propagation despite the gain offered by the directional antennas. Antenna combinations dd and ee, which use directional antennas of the same polarization at both the transmitter and receiver, provide consistently lower system path loss for all locations. Quantitatively, the path loss values on links using these two antenna combinations are approximately one standard deviation below the average path loss over all antenna combinations. Path loss results of location M for some antenna combinations are given in Table V. The descriptive statistics of system path loss values for all antenna combinations over all locations are given in Table VI, while the complementary CDF of system path loss values for some antenna combinations are presented in Fig. 6.

3.2.1. Effects o f Beam Pattern on Path Loss

Table VII summarizes some comparisons between the power received for co-polarized antenna pairs, the antenna patterns of which may be omnidirectional or directional. The first two cases in Table VII consider the effect of the receiving antenna pattern (with a fixed transmitting antenna) or the received power, while the last two cases consider the effect of the transmitting antenna pattern (with a fixed receiving antenna) on the received power. Consider as an example the VP antenna in case i of Table VII. The differences between the received powers of bb and bd were calculated for all locations. An average value of PWR (bd) - PWR (bb) over all the measurement locations is computed and is shown in the fourth column of the table (3.0 dB). A similar average value is computed for HP antennas for case i and is shown in the fourth column of the table (4.3 dB). An overall average (both HP and VP), based

Table V. Path Loss Results at Location M (Free Space Path Loss with Isotropic Radiator = 68 dB)

Antenna combination PL' (dB) k(PL')

bb 77 bc 82 cc 72 cf 69 dd 69 ee 68 ff 73 Maximum 82 Minimum 68 Average 74 Standard deviation 4

0.6 1.9

-0.5 -1.2 -1.2 -1.4 -0.2

68 H o , R a p p a p o r t , a n d K o u s h i k

Table VI. System Path Loss (PL') Results (All Locations)

Standard Antenna Average deviation Maximum Minimum Median

combination (dB) (dB) (dB) (dB) (dB)

bb 72.14 3.51 79.82 66.91 71.53 bc 76.75 4.29 84.54 71.83 74.63 bd 68.88 3.38 75.77 63.59 68.44 be 74.11 4.73 83.07 65.91 73.84 bf 70.41 1.76 73.91 67.79 70.61 cb 77.75 3.10 83.00 70.93 78.39 cc 72.28 3.10 78.96 67.49 71.17 cd 72.43 3.17 78.31 66.34 72.49 ce 67.53 2.92 72.03 63.28 68.08 cf 70.16 2.05 73.93 66.67 70.27 db 69.44 1.79 72.54 66.78 69.37 dc 75.33 1.53 78.66 72.52 74.97 dd 65.40 3.13 71.02 60.43 66.39 de 71.61 1.95 76.48 68.15 71.93 df 67.26 2.59 70.45 62.59 67.82 eb 75.63 2.30 79.88 69.42 76.16 ec 70.31 3.35 76.51 63.66 71.53 ed 71.35 2.61 78.50 68.43 70.56 ee 64.15 3.59 70.19 57.09 65.20 ef 68.67 1.99 72.70 65.68 68.53 ./b 72.31 1.96 76.62 67.69 72.05 fc 73.86 3.18 80.41 67.29 73.64 fd 68.60 2.49 74.20 64.81 68.16 fe 67.13 3.49 76.77 62.38 66.21 ff 67.60 3.09 75.47 63.07 66.92

on the values in column four of case i, is computed for case i and is shown in the fifth column of the table (3.8 dB).

When the transmitting antenna is fixed, i.e., case i and ii, the signals arriving at the receiving antenna will be the same for antenna combination #1 and #2 because the same transmitter antenna is used for both antenna combinations, and comparisons are performed at the same measurement location. If signals arrive only in the

1 . 0 , " . . . . . . �9 I I

0 9 "'... "~

. . .I 4 : : =o. i 5 ........

~-o.2 "., !

o l "" \ o'o

50 55 60 65 70 75 80 85 90 95 1 O0 Poth Loss (dEI)

Fig. 6. Complementary CDF of system path loss (PL') values (all locations) for some antenna combinations.

Table VII. Comparisons Between Co-polarized Antenna Pairs with Different Beam Patterns

PWR(#2) - Overall Antenna Antenna PWR(#1) average

Case combination #1 combination #2 (dB) (dB)

i omni/omni omni/dir VP: bb VP: bd 3.0 3.8 HP: cc HP: ce 4.3

ii dir/omni dir/dir VP: db VP: dd 4.4 5.2 HP: ec HP: ee 5.6

iii omni/omni dir/omni VP: bb VP: db 5.4 3.9 HP: cc HP: ec 2.5

iv omni/dir dir/dir VP: bd VP: dd 4.4 3.9 HP: ce HP: ee 3.6

boresight direction of the receiving antenna, the difference between the received power for antenna combinations #1 and #2 in case i will be roughly equal to the difference of the directive gains of directional and omnidirectional antennas in the boresight direction, i.e., 10 dB. On the other hand, we expect no difference between the received power of the two antenna systems if the signals arrive from all directions. The overall averaged difference of received power between antenna combination #1 and #2 is 3.8 dB for case i and 5.2 dB for case ii. Our results suggest that most signal components arrive uniformly in the azimuth direction. This agrees closely with the difference between the mean effective gain (MEG) values of the two antennas, assuming signal components arrive uniformly in the azimuth direction [23]. The MEG incorporates the spatial distri- bution of the incident signals to determine the "average" antenna gain value [29]. A directional transmitting antenna only localizes the signals arriving within the

main beam of the receiver by less than 2 dB, when compared with an omnidirectional transmitting antenna, in indoor obstructed channels.

At location B, the power received with b b is better than that of bd . Therefore, the gain of the directional transmitting antenna does not always help to improve the power received at the receiver. Similar observations were found at other locations. On average, directional receiving antennas pointed on boresight on the hypo- thetical LOS path provides more power at the receiver than omnidirectional receiver antennas using the same co-polarized transmitting antenna, but the improvement (4.5 dB) is less than the difference of gains between the omnidirectional antenna and the directional antenna.

The effect of the antenna pattern of the transmitting


antenna for a fixed receiving antenna is studied for case iii and iv in Table VII. The link improvement by using a directional antenna at the transmitter over an omnidirectional antenna at the transmitter is about 4 dB if co- polarized antennas are used at the receiver. The power improvement using a directional antenna at the transmitter over an omnidirectional antenna at the transmitter, when using a co-polarized receiving antenna, is about the same (4 dB) as the improvement offered by a directional antenna at the receiver over an omnidirectional antenna at the receiver, when a co-polarized transmitter antenna is used. The 4-dB improvement closely matches the difference between the MEG values of the omnidirectional and directional antennas. From the above analysis, it appears that signal components depart from the transmitter and arrive at the receiver uniformly in the azimuth direction.

3.2.2. Effects of Polarization on Path Loss

Table VI shows that co-polarized directional antenna pairs dd and ee provide consistently low path loss values at all measurement locations. On the other hand, ffgives fairly high path loss values at two measurement locations. This may be due to the fact that the sense of polarization of a CP wave is changed for every reflection and hence some reflected components could not be detected by the CP directional receiver.

When the transmitter is a linearly polarized directional antenna, a receiver antenna with the same beam pattern and polarization as the transmitter always provides the highest received power. The received power of a linearly polarized directional transmitting antenna with a CP directional receiving antenna is, on average, 3 dB smaller than the highest received power of all antenna combinations with the same transmitting antenna. The 3-dB difference agrees with the theoretical polarization discrimination between LP and CP, when no depolarization occurs in the channel.

The average difference between path loss o f f f and fd (or fe) is small and is, on average, less than 1 dB. Assuming no depolarization, the difference between the received power of fd andffwould be around 3 dB since most signal components arrive in the azimuth direction. The above observation shows that depolarization of a CP signal is fairly large in indoor OBS channels. This can be explained physically by the fact that a CP signal changes its sense after each reflection.

The average cross-pol discrimination (XPD) (defined as the ratio of the signal level at the output of a receiving antenna that is nominally co-polarized to the output of receiving antenna of the same gain but orthog-

onally polarized to the transmitting antenna, when they are aligned on boresight [7]) of the HP and VP waves is 4.5 and 4.6 dB, respectively. The XPD for directional antennas is higher than that of omnidirectional antennas. The overall average and standard deviations of XPD of all measurements are 4.5 and 2.8 dB. The overall average is slightly higher than that reported previously [7] due to the fact that directional antennas were employed in our measurements at both the transmitter and the receiver.

3.3. Path Loss vs. rms Delay Spread

Linear regression analysis is performed for path loss vs. rms delay spread for different antenna combinations. For co-polarized omnidirectional antenna, PL' is uncor- related with rms delay spread as reported previously [3]. Antenna combinations with an HP directional transmitter provide some correlation between PL' and rms delay spread for all receiving antennas. From our experimen- tal results, it is found that directional antenna pairs are more likely to provide high correlation between PL' and rms delay spread. This is because a directional receiver can receive the strong early arriving components resulting in low values of path loss and rms delay spread. Antenna combination cc has the highest correlation (R 2 = 0.83) between path loss and rms delay spread. Other antenna combinations that have some correlation (R larger than 0.50) between PL' and rms delay spread include ce, dc, eb, ec, ed, ee, andfe. For all other antenna combinations, PL' appears to be uneorrelated with rms delay spread.

Antenna combinations that have a high correlation between PL' and rms delay spread are desired for system design because minimizing both path loss and delay spread are important for future PCN systems. If the correlation between PL' and rms delay spread is high, then two parameters of the linear regression model, y-inter- cept and slope, can describe the relation between PL' and rms delay spread fairly well.

4. IMAGE-BASED RAY TRACING ALGORITHM

Since buildings have different sizes and different structures, site-specific propagation models can be val- uable for efficient installations of indoor PCN systems. The models can give accurate results if the size of the obstacle is much larger than a wavelength and the observation point is many wavelengths from the scatterer. Alternatively speaking, the prediction provides an asymptotic solution of the problem when the wave-


length vanishes or the frequency approaches infinity. In this section, a site-specific ray tracing algorithm that predicts channel impulse responses inside buildings is presented. An overview of this prediction model is given in Fig. 7.

The proposed model accepts the building geometry and parameters such as antenna patterns and antenna locations as input and performs ray tracing up to a speci- fied level. That is, the program can include all first-order, second-order, and up to the Nth order reflection paths. Multipath sources are determined using image theory. For example, a single-hop multipath component as shown in Fig. 8 has an image located at location IM. The reflection point P can be determined by the inter- section between the object and the line from IM to the receiver. As shown in Fig. 8, incident and reflected electric field can be decomposed into parallel and per- pendicular polarizations. Second-order reflection paths can be determined by secondary images, which are cre- ated by considering primary images as sources. Simi- larly, multiple reflection paths can be determined by creating images recursively. The methodology for de- termining multiple reflection paths is explained in detail by Ho [23]. Since all objects are finite in extent, inter- section tests must be performed to check if physical reflection paths exist or not. For fixed transmitter locations, the same images can be used to determine channel characteristics at multiple receiver locations.

After a propagation path is determined, geometrical information is used to compute the received power using a geometrical optics (GO) assumption which takes into account amplitude, time delay, phase, and polarization. Fixed transmission losses are then added to each multipath component which is transmitted through an object. At the frequencies of interest (2.45 GHz), the surfaces of the objects can be assumed to be smooth. At

objects data, transmitter location and antenna receiver location and aate.n.na

1

! De.minion or" I Ganmel~rical I [ Propagati~ Paths Analysis

Compute Received Power using GO

and Fro,me[ Fcmul~ i Computation of Received ,~. ~ Power and Absolute Ttme

[ Transmission Losse~ i Delay

. . . . . . . . . . . . . . ~ ~ : ; ~ ~ R~po~e

Fig. 7. Image-based ray tracing algorithm.

TX RX

A

I ,- t / " ! % :.qh ~.l,c" , ~ , ~ , o , ~ , =

Q ..... ; p

i "" range and with a certain [ ...." dieTectric constant t r i .." , ..." i. . .

a IM : IMAGE

Fig. 8. A single-hop muhipath component.

higher frequencies, a roughness factor should be included in the modeling. Further, a diffuse scattering component [31] should be include into the analysis if the surface is " rough . " Most obstacles in indoor environments can be modeled as a dielectric slab with pre- defined thickness, boundaries, and dielectric constant. In the proposed algorithm, all objects (walls, soft partitions) are assumed to be finite rectangular planes. Al- though walls and other obstacles have finite width, reflection coefficients are approximated by the Fresnel coefficients here. Since the received signal components are dominated by reflected or transmitted components [21, 27], diffraction is not considered in this paper. In addition, far field conditions for the antennas are assumed. The received power of each component can be determined [23] using Eq. (4). The polarization characteristics at reflecting interfaces_ are described by the dyadic reflection coeff• Y'j [32]

e -j ~L K - F r ( O r , C~T)FR(OR, ~R)/1.1.

L

�9 F i " F2 . . . . . FN" f i r (2) - - ^ p ^ Fj = Fll j i[ l j f l l j -t- F • j l • j r x j (3)

PR = PT[KI 2 (GTGRX2/167r2) (4)

The notation is explained in Table VIII.

5. P R E D I C T I O N INSIDE T E L E D Y N E ' S B U I L D I N G

The Teledyne building was represented using AutoCAD as a database manager. The office contains about 200-300 objects, each of which is either a


Table VIII. Symbols in Eqs. (2)-(4)

Symbols Explanations

FT/F R Transmitter/receiver antenna pattern GT/G R Antenna gains llT/l~tr Antenna polarization vectors L Total propagation path length /3 Phase constant (= 27r/X) ~j Dyadic reflection coefficient at thejth

interface I',lj/r a.) Fresnel reflection coefficients PTIPr Transmitted/received power il~, i ~j, #h~, :'.~j Defined in Fig. 8

2-m-high soft partition or a concrete wall between the floor and ceiling. Because of the difficulty of manually creating such a large database, only 52 objects repre- senting important characteristics of the environment have been defined in the objects data file and are shown in Fig. 1, which is a reasonable approximation to the geometry of the environment. The floor is 43.3 m by 82.3 m and the height of the ceiling from the floor is 3 m. The transmitter was placed at the location (9.5, 32.0) meters for all measurements.

Comparisons between measured and predicted values have been done using various parameters in the lit- erature. Path loss is a common measure of the accuracy of a propagation model and has been used previously [18, 21, 24]. However, time dispersion parameters are more appropriate to determine the validity of a propagation model. Therefore, comparisons between measured and predicted rms delay spread (aT) values are also presented in this paper. Two profiles can have different shapes, yet have identical path loss and rms delay spread values. Qualitative comparisons between shapes of measured and predicted power delay profiles provide insights into the actual propagation mechanisms, and two such examples are presented in this paper while more extensive comparisons can be found in the work of Ho [23].

To compare the measured and the predicted results, the predicted channel impulse response is convolved with a 33-ns pulse to obtain the corresponding predicted power delay profile, In this section, predictions for three antenna combinations are presented. The predicted results are compared with measured results on a location- by-location basis using the image-based model and parameters defined below. The following assumptions have been made in the propagation model [23]:

�9 Figure 1 is a reasonably good approximation of the measurement location.

�9 The ceiling is modeled as a rough surface and does not contribute significant signal components to the receiver [24].

�9 Transmission loss through each object is 2 dB for all incidence angles [26].

�9 The operating frequency is 2.45 GHz. �9 The dielectric constant for each object is 15.0

(arbitrarily chosen). Other work shows the prediction error can be minimized by optimizing the dielectric constant [27].

Figure 9 shows the measured and predicted power delay profiles at location F (OBS) using co-polarized omnidirectional discone antennas at both the transmitter and the receiver (30 m apart). The measured profile shows multipath components with most of the energy arriving between absolute time delay intervals of 150 and 270 ns. A similar behavior is found for the predicted profiles. There are also significant amounts of signal components near 300 and 375 ns. The predicted power delay profiles successfully capture these two significant late arriving components. The early part of the predicted power delay profiles is not as " smooth" as that of the measured power delay profile because of the limited number of reflected rays in the model. However, the predicted results seem to be fairly reasonable for the assumptions made in the modeling. The measured path loss is 77 dB, while the predicted path loss is 78 and 77 dB using second- and third-order ray tra_cing, respectively. The measured mean excess delay (z) and the predicted mean excess delays using second-order and third- order ray tracing are 78, 43, and 61 ns, respectively. The predicted rms delay spreads are within 20 ns of the measured. Thus, the predicted profiles agree reasonably well with the measured profile at this location.

o~ PU Measured 78~5 60ns 77dB

levd--2 43ns 40ns 78dB level--3 61n.s 49ns 77dB

- 7 5 . 0 ~ Measured A J - - fobb2 J - 7 8 . 5 ......... f obb3

-82~ / ~ i " , I I - 8 5 . 5

i - 8 9 . 0 ' i : Measured Profile

n. - 9 6 . 0

- 9 9 . 5 ~ L c I 2 ~ Pr~ Most Ener l~ ~+r . - 10.3.0 : ~.., Predicted

-;o6.5 I ~ l ' h ~ T t,-,,--+~ - 1 1 0 . 0 , I , , , i .:, I , . I , : . h - I L l [, H I ' , . t , :. ,

54 1 0 8 162 2 1 6 270 3 2 4 378 4 3 2 4 8 6 5 4 0 absolute time delay (ns)

Fig. 9. Predicted and measured profiles at location F (antenna combination = bb).

72 Ho, Rappaport , and Koushik

-60 - 6 4

-6B ~-72 J - 7 6 i-80

-84 - 8 8

-92 -96

-1 O0

M ~ . ~ - e d SOre ~ 69dB

levd--?, 22aa 27t~ ?OdB levd-~ ~ a s 1 5 ~ 69dB

I ~ Measured I modd2 modd3

f l l l . . . . . h . ,

54 108 162 216 270 324 378 432 486 540 �9 al~olul~ lime delay (m)

Fig. tO. Measured and predicted profiles at location M (antenna combination = dd).

•I•II•M inlsur~l Profile

I ~ 2 n d level) Ig ~ Pre4~l P~l~e

I I N A t ~ O n l l e v i |

. . . . . . . . .

Figure 10 shows the measured vs. predicted results for the directional antenna combination dd at location M. Assuming perfect antenna alignment, the predicted path loss and rms delay spread values are very small ( < 10 ns). The actual performance of directional antennas in obstructed channels is not as good as the predicted performance with perfect antenna alignment due to possible alignment errors and shadowing effects caused by some unmodeled objects along the direct path. Hence, an alignment error has been used in the simulation. At an off-boresight angle of 0.5 radians (29~ the path loss prediction error is almost unbiased. While this much larger than the actual alignment error in the

measurement, it helps offset the effects of unmodeled objects and has been assumed for predictions with this antenna combination. Measurement results show that directional antennas can give smaller path loss and rms delay spread than omnidirectional antennas. However, performance gain using directional antennas with perfect antenna alignment can be too optimistic [30]. In Fig. 10, the predicted profiles given reasonable esti- mates of the strong components at 130 ns, and the relatively weaker component at 190 ns. Path loss prediction error is within 1 dB. The predicted rms delay spread values are within 7 ns of the measured value.

The rms delay spread, mean excess delay, and the wide-band path loss values for the measured and the predicted profiles using third-order ray tracing at eigh- teen locations are given in Table IX. (Some measurement results for antenna combination bb at additional measurement locations are not reported in previous sections, but they are used here for comparisons between predicted and measured channels.) The path loss error is nearly unbiased, i.e., the prediction error is symmetrical about 0 dB. The standard deviation of the prediction error is 4.6 dB for second-order ray tracing and 4.7 dB for third-order ray tracing over a 20-dB dynamic range. The worst-case path loss error is 9 dB and is close to those reported by Seidel and Rappaport [18] and by Honcharenko et al. [24]. Moreover, the difference between the predicted path loss values using second-order or third-order ray tracing is negligible. This is because

Table IX. Measured and Third-Order Prediction Results

Receiver Predicted a, location Predicted r (ns) (ns)

(topogrdphy) Measured ~ (ray tracing Measured a, (ray tracing Antenna combination (ns) level = 3) (ns) level = 3)

Measured path loss

(dB)

Predicted path loss (dB)

(ray tracing level = 3)

A (LOS) bb 50 28 34 40 64.7 64.8 B (OBS) bb 62 40 37 29 69.2 64.0 C (LOS) bb 47 78 29 85 61. I 70. I D (LOS) bb 51 68 36 76 64.3 69.8 E (OBS) bb 59 39 43 20 74.6 73.5 F (OBS) bb 78 61 60 49 76.8 77. I

G (LOS) bb 53 50 48 62 78.7 76.3 H (LOS) bb 50 4 2 41 57 71.3 69.4 I ffJBS) bh 101 42 74 42 82.4 73. I L (OBS) bb 59 34 38 33 71.0 68.9 M (OB.S) bb 65 31 37 29 76.8 73.6 B (f)BS/ cc 6(l 24 411 23 73 66 K (OBS) ~'r." 64 48 40 45 76 68 1, fOBS) ~:~." 55 29 41 31 72 75 M fOBS) t.'c 47 24 32 26 69 70 B (OilS) dd 50 31") 33 45 66 61 I. taBS) dd 47 19 34 15 62 65 M (OBS) dd 50 31"1 34 35 69 69


the probability of a significant triple or multiple reflection path is low. In corridors, where the radio channel behaves like a rectangular waveguide, higher-order reflection paths can contribute significantly toward the received power. Assuming the predicted mean excess delay or rms delay spread should be within a factor of two relative to the measured data for " g o o d " prediction, most data points can be classified as good predictions. However, there are large delay spread prediction errors at a couple of locations, since rms delay spread is sensitive to small changes of power delay profile, it is not necessarily a good measure of prediction accuracy. Sim- ilar prediction problems for rms delay spread are also found in Refs. 18 and 20. It appears that a better method to quantify time dispersion is required. Qualitative comparisons of predicted and measured power delay profiles show that the shapes of the profiles agree reasonably well at most locations. If the computation is performed over the entire office area, a map of the channel param-

eters can be obtained. The predicted path loss for the entire office superimposed on the simplified blueprint is given in Fig. 11. Such a path loss map can be very useful for coverage, interference, and capacity analysis.

6. CONCLUSIONS

The effects of antenna patterns and polarizations on path loss and delay spread in obstructed wireless channels are presented in this paper based on a series of experiments. It is found that an omnidirectional transmitting antenna with a directional CP receiving antenna provides the lowest rms delay spread and the lowest maximum excess delay (10 dB down) among all antenna combinations. However, the system path loss for this combination is fa i ry large at two locations. Comer re- flectors used at both the transmitter and the receiver are able to provide consistently high received power at all


measurement locations irrespective o f the polarization.

Co-polar ized directional CP antennas are not effective for combat ing multipath or achieving high received

power in obstructed channels since CP signals are vulnerable to depolarization in obstructed channels. Hence ,

the results o f our exper iment show that using vertically polarized directional antennas at both the transmitter and

the receiver can give relatively low delay spread and low path loss. Moreover , the fact that a high correlation exists between the path loss and rms delay spread for

this antenna combinat ion is an advantage for system design o f future P C N systems. However , s imulation re-

sults show that the performance o f directional antennas is sensitive to their a l ignment . For practical implemen-

tation, an adaptive mechanism is needed to line up the beams i f the antennas are directional. Using adaptive

antennas at both the mobi le and the base station can give the best power and min imum delay spread, while re- ducing the interference from other users. Therefore , use

o f adaptive antennas may provide good system perfor-

mances with high capacity. Table X summarizes some

of the key measurement results o f this paper. The results o f this measurement program provide some prel iminary

guidelines for antenna design inside buildings.

In the second half o f the paper, an image-based propagation prediction algori thm applicable to any type

o f in-building envi ronment using different transmitter

and receiver antennas has been described. Compar isons between predicted and measured results for the Tele-

dyne office building at 2.45 GHz for three antenna combinations have been presented. Path loss errors are

shown to be symmetr ical about 0 dB with a standard

deviat ion of 4 .7 dB. Predicted rms delay spread values agree reasonably well with the measured values at most locations. The shapes o f the measured power delay pro-

files also agree reasonably well with the predicted power

delay profiles. Further, predictions for a VP corner re-

flector combinat ion were found to be too optimist ic if the antennas are assumed to be perfectly al igned due to

Table X. Summary of Antenna Combinations that Have Small Delay Spread or Small Path Loss or High Correlation Between PL'

and v, at All Measurement Locations

Anlenna combinations

Small delay Small path High correlation spread loss between PL' and a,

b f ce ce tit' ~J'dd dd ee eb ed

fd ~f J'e dd

possible al ignment errors in the measurement and shad-

owing caused by unmodeled objects along the direct path. It appears that unmodeled furniture and office

equipment can be a significant source o f predict ion errors. Although comparison statistics are not large enough

to give a broad-based conclusion on the effect iveness o f

the prediction model , the image-based ray tracing algori thm shows promise for accurate and efficient in-

building site-specific propagat ion prediction.

A C K N O W L E D G M E N T S

The work was sponsored by Teledyne Inc. , the

M P R G Industrial Affiliates, and A R P A .

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C. M. Peter Ho was born in Hong Kong on October 13, 1970. He received the B.S. and M.S. degrees in electrical engineering from Virginia Tech in 1991 and 1993, respectively. In August 1991, he joined the Mobile and Portable Radio Research Group (MPRG) at Virginia Tech where his graduate research focuses on indoor wireless channel characterization and prediction. He is currently employed by Hand Held Products Inc. in Charlotte, North Carolina.

Theodore S. Rappaport was born in Brooklyn, New York, on November 26, 1960. He received B.S.E.E., M.S.E.E., and Ph.D. degrees from Purdue University in 1982, 1984, and 1987, respectively. In 1988, he joined the Electrical Engineering faculty of Vir- ginia Tech, Blacksburg where he is an associate professor and director of the Mobile and Portable Radio Research Group. Prof. Rappaport


conducts research in mobile radio communication system design and RF propagation prediction through measurements and modeling. He guides a number of graduate and undergraduate students in mobile radio communications, and has authored or co-authored more than 70 technical papers in the areas of mobile radio communications and propagation, vehicular navigation, ionospheric propagation, and wide- band communications. He holds a U.S. patent lbr a wide band antenna and is co-inventor of SIRCIM. an indoor radio channel simu- lator that has been adopted by over 100 companies and universities. In 1990, he received the Marconi Young Scientist Award for his con- tributions in indoor radio communications, and was named a National Science Foundation Presidential Faculty Fellow in 1992. He is an active member of the IEEE, and serves as senior editor of the IEEE Journal on Selected Areas in Communications. He is a registered pro- fessional engineer in the State of Virginia and is a fellow of the Radio Club of America.

M. Prabhakar Koushik was born in India. He received the B.S. degree in electronics and communications from Bangalore Uni- versity in 1982. He worked with the Indian Space Research Organi- zation from 1982 to 1985. He received the M.S. degree in computer science in 1987 from Virginia Tech. He is currently associated with the Mobile and Portable Radio Research Group in Blacksburg, Vir-

ginia.

antenna effects on indoor obstructed wireless channels and...

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