efficiency improvement of high frequency inverter for ... an active series reactive power...
TRANSCRIPT
IEEE PEDS 2017, Honolulu, USA
12 - 15 December 2017
Efficiency Improvement of High Frequency Inverterfor Wireless Power Transfer System Using a Series
Reactive Power Compensator
Jun OsawaGraduate School of Pure and Applied Sciences
University of Tsukuba
Ibaraki 305–8573, Japan
Takanori Isobe, Hiroshi TadanoFaculty of Pure and Applied Sciences
University of Tsukuba
Ibaraki 305–8573, Japan
Abstract—This paper proposes a wireless power transfersystem using a series compensator GCSC as a primary sidecapacitor. The GCSC is a circuit module that equivalently func-tions as a series variable capacitor by controlling semiconductorswitches. The advantage of applying the GCSC to a primaryside capacitor is giving a controllability of power factor for ahigh frequency inverter. Therefore, the optimum operation ofthe high frequency inverter can be achieved regardless of thecoil parameters. Experimental results with an 1 kW laboratoryprototype confirmed that the proposed system can achieve anoptimum operation and high efficiencies of the high frequencyinverter.
I. INTRODUCTION
In recent years, wireless power transfer (WPT) systems
are activity studied. Especially, WPT systems for Electric
Vehicles (EVs) or Plug-in Hybrid Electrical Vehicles (PHEVs)
are highly paid attention since it can improve safety and
convenience of charge operation.
A magnetic resonant coupling is the leading circuit topology
of WPT for EVs or PHEVs since it enables high-power and
high-efficiency power transfer with a large air gap. In this
topology, inductive reactance of the coils are compensated by
using resonant capacitors to reduce current or voltage applied
to the coils and the power source. However, there is one
challenge that the reactance of the coil is not fixed because the
air gap between the primary and secondary coils is expected
to change according to the parking position. So the optimum
capacitance for the resonant capacitors can vary, and fixed
resonant capacitors cannot achieve the optimum operation
constantly in that situation.
To address this problem, this paper proposes a WPT system
using an active series reactive power compensator named
GCSC (Gate Controlled Series Capacitor)[1][2] instead of
the fixed resonant capacitors. The GCSC is a circuit module
connected in series that equivalently functions as a variable
capacitor. In the proposed system, the optimum operation
can be realized by controlling the equivalent capacitance of
the GCSC according to the reactance of the coil. The coil
efficiency improvement using the GCSC on the secondary
side has been proposed in [3]. This paper proposes an inverter
Fig. 1. Circuit configuration of the gate controlled series capacitor (GCSC).
optimum operation by using the GCSC on the primary side.
This paper reports experimental verifications of the proposed
system with an 1 kW laboratory prototype and discusses the
efficiencies of the high frequency inverter and the GCSC.
II. SERIES COMPENSATOR GCSC
A. Circuit Configuration and Features
Gate controlled series capacitor (GCSC) is one of FACTS
(flexible ac transmission system) devices proposed as a series
reactive compensator applied for ac power transmission lines.
Fig. 1 shows the configuration of the GCSC. The GCSC is
consists of two full-controlled reverse conductive semiconduc-
tor switches and a capacitor. The GCSC is controlled by line
frequency switching, and the two switches share their source
potential therefore the gate drive circuit can be simple. The
GCSC can achieve soft-switching in all the operating range
and there is no conduction loss when current is conducting
the capacitor. Therefore, loss in the semiconductor switches is
comparatively low.
B. Operation Principles
Switch S1 and S2 are complementary turned on with the
line frequency and a half duty ratio. The phase angle difference
between the line current phase and the gate signals, δ, is given
for control the equivalent capacitance of the GCSC , CGCSC.
Fig. 2 shows the schematic waveforms of the GCSC and Fig. 3
shows possible current paths. The GCSC has two types of
current paths, one is conducting the capacitor as shown in (a)
(b) (d) (e), and the other is conducting semiconductor switches
as shown in (c) (f). The capacitor voltage waveform has a zero
voltage period which is achieved by modes (c) and (f), and its
978-1-5090-2364-6/17/$31.00 c©2017 IEEE
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0
1
-1
0
1
0
1
Cu
rren
tC
apac
ito
rS
2 s
ign
alv
olt
age
(a)
-1
0
1
-1
0
1
0
1
Cu
rren
tC
apac
ito
rS
2 s
ign
alv
olt
age
δ
(b)
-1
0
1
-1
0
1
0
1
Cu
rren
tC
apac
ito
rS
2 s
ign
alv
olt
age
δ
(c)
Fig. 2. Three waveform modes of the GCSC with different control phase angle conditions of (a)δ = 0, (b)0 < δ < 90, and (c)δ = 90.
Fig. 3. Possible current paths of the GCSC.
duration can be controlled by δ. When δ = 0 , flowing current
conducts only the semiconductor switches and the capacitor is
not charged, that means CGCSC = ∞. By increasing δ, flowing
current conducts the capacitor and generated voltage in the
capacitor increases. Finally, when δ = 90, flowing current
conducts only the capacitor, that means CGCSC = C, where
C is the capacitance of the equipped capacitor. Therefore, the
GCSC can vary its equivalent capacitance from C to ∞ by
controlling δ.The equivalent reactance of the GCSC can be derived by
the similar way discussed in [4]. The injecting voltage, vgcsc,
within a half fundamental cycle (0 < θ < π) can be derived
as
vgcsc=
√2XcI (sin θ−cos δ)
(
π
2− δ<θ< π
2+ δ
)
0 (else) ,(1)
where the pure sinusoidal current as
i =√2I cos θ, (2)
is assumed. The rms value of the fundamental component, V1,
can be derived as
V1 =2√2π
∫
π
0
vgcsc(θ) sin θdθ
= IXc
(
2δ
π− sin 2δ
π
)
, (3)
and then, the equivalent reactance, XGCSC, can be derived as
XGCSC = V1/I
= Xc
(
2δ
π− sin 2δ
π
)
. (4)
The current conducting semiconductor switches, isw, within
a half fundamental cycle(0 < θ < π) can be derived as
isw=
0(
π
2− δ<θ< π
2+ δ
)
√2I cos θ (else) ,
(5)
The loss of the GCSC, PGCSC, is derived as
PGCSC =1
π
∫
π
0
roni2sw(θ) + VFisw(θ)dθ
=1
π(ronI
2(π − 2δ + sin (π − 2δ))
+√2VFI sin (
π
2− δ))), (6)
where ron is the on resistance of the MOSFETs, VF is the
forward voltage of the free-wheeling diodes.
III. INVERTER LOSS REDUCTION BY APPLYING GCSC
A. Optimum Operation of High Frequency Inverter
Generally, the inverter of WPT systems for EV is operated
at high switching frequency, for instance 85 kHz; therefore, the
switching loss reduction is important to improve the system
efficiency. A voltage-source type high frequency inverter with
fundamental frequency switching, which means the output
current frequency is same as the switching frequency, is used
for the purpose.
A lagging power factor at the inverter output, as shown in
Fig. 4(a), is attractive for this topology, in which the turn-
off is performed with mitigated dv/dt achieved by a snubber
capacitor and/or device parasitic capacitance, and turn-on is
performed with conducting free-wheeling diode, so that turn-
on is complete soft-switching. On the other hand, the leading
power factor, as shown in Fig. 4(b), is not attractive since
it results in shorting the capacitors and turn-off of the free-
wheeling diode that causes reverse recovery, and the switching
993
(a)
(b)
Fig. 4. Schematic waveforms of a switching device of the high frequencyinverter with (a)Lagging power factor, (b)Leading power factor.
loss increases. And same phenomena occurs if the power factor
is too high as the current crosses the zero during the dead time.
Therefore, the high frequency inverter should be operated
at lagging power factor that delays current beyond the dead
time. At the same time, in order to minimize the voltage and
current at turn-off and to reduce the turn-off losses, the power
factor should be as high as possible. Therefore, operating the
high frequency inverter at slightly lagging power factor which
the current crosses zero at the end of dead time is optimum
and can minimize the switching loss.
B. Selection of Resonant Capacitor
In the WPT system in series-series circuit topology [5]
which is shown in fig. 5, the secondary side capacitor can
improve the efficiency of the coils by compensating for the
secondary side self inductance and minimizing the primary
current to achieve the same power transfer. Efficiency of the
coils is maximized when the secondary side capacitor fully
compensates for the self inductance of the secondary coil. The
primary side capacitor improves the output power factor of
the high frequency inverter. The capacitance of the primary
capacitor is selected to operate the high frequency inverter at
the slightly lagging power factor.
When the secondary side capacitor fully compensates for
the self inductance of the secondary coil, the power factor at
L1 L2
C1 C2
RL
M
Fig. 5. An equivalent circuit diagram of the WPT system of the series-seriestopology in the fundamental frequency.
0 50 1000
0.5
1
C1=11.68 nF
po
wer
fac
tor
Mutual inductance [μH]
C1=12.79 nF
C1=13.49 nF
Fig. 6. Example of the resulting power factor with varying mutual inductance,M , and fixed primary capacitance, C1.
0 50 10011
12
13
14
pf=0.9
Mutual inductance [μH]
Pri
mar
y c
apac
itan
ce [
nF
]
pf=0.8
pf=1
Fig. 7. Required primary capacitance, C1, to achieve constant power factorsfor varying mutual inductance, M .
the inverter output can be derived as
pf = cos
(
arctan(ωL1 − 1/ωC1)RL
(ωM)2
)
, (7)
where L1 is the self inductance of the primary coil, C1 is
the capacitance of the primary side series capacitor, RL is the
load equivalent resistance, and M is the mutual inductance
between the coils. To operate the high frequency inverter at
a lagging power factor, C1 should be selected appropriately
to compensate for L1 partially. However, the power factor is
influenced by the possible mutual inductance changing in WPT
applications. Therefore, fixed capacitors can not achieve the
optimum operation for varying mutual inductance.
In conventional WPT systems using a fixed capacitor, the
994
(a) (b)
Fig. 8. Coils used in the experiment at (a)primary side (b)secondary side.
TABLE ISPECIFICATIONS OF COILS
Primary Secondary
Winding type Spiral typeSize (W / D / H) 300 mm / 300 mm / 30 mmWire specification Litz wire (φ0.08 mm × 30)Number of Parallels 30 14Number of Turns 24 45Core PC95 (ferrite)Resistance 0.21 Ω 0.59 Ω
primary capacitor should be designed to achieve a slightly
lagging power factor when mutual inductance is the largest
value, to achieve the ZVS turn-on for all the range of varying
mutual inductance.
For example, the resulting power factor with fixed L1
(300 µH), fixed RL (100 Ω), and varying M is shown in
Fig. 6. As can be seen from the figure, one capacitance can
achieve unity power factor regardless of varying M ; however,
the other values of capacitance can not achieve constant power
factor as function of M . Therefore, fixed capacitance can not
achieve a lagging constant power factor for varying M .
Fig. 7 shows required C1 to achieve constant power factors
for varying M . As can be seen from the figure, varying C1 is
needed to maintain the power factor at a certain value except
for unity power factor. This is the reason for the need of the
active compensator for the WPT applications with possible
miss alignment. In the proposed system, the power factor can
be kept constant by controlling the equivalent capacitance of
the GCSC connected in the primary side.
IV. EXPERIMENTAL VERIFICATIONS
A. Experimental Setup
To confirm the above discussion, experiments were con-
ducted. Fig. 8 shows the picture of coils used in the ex-
periments. The coil shape is a spiral winding type and the
diameter is 300 mm. Litz wires were used to reduce the loss
increased by skin effect and proximity effect, and ferrite cores
are placed on the back side of the winding to increase the
coupling coefficient and quality factor. In order to suppress
the increase of the primary copper loss, which is the dominant
loss component of the coils, the primary coil was designed to
reduce resistance by increasing the number of parallels of the
winding.
0 40 80 1200
200
400
0
600
1200
Horizontal gap [mm]
L2 [
μH
]
L1,M
[μ
H]
L1
L
Fig. 9. Inductance characteristics of the coils used in the experiments asfunction of the horizontal gap.
M
Vout
P1 P2 P3 P4
Fig. 10. The proposed circuit configuration for the WPT using the GCSCas a primary side capacitor.
TABLE IIEXPERIMENTAL CONDITIONS
Fixed cap. A Fixed cap. B with GCSC
Frequency 85 kHzRated power 1 kWDead time 800 ns
Vout 350 V
C – – 32.09 nFC1 11.28 nF 12.29 nF 17.42 nFC2 3.09 nF
Table I lists specifications of the coils. In the experiments,
the vertical gap between the primary coil and secondary coil
was set constant at 100 mm and the horizontal gap was
changed from 0 mm to 120 mm. Fig. 9 shows varying L1,
L2 and M of the coils as function of the varying horizontal
gap. It can be seen from the figure that the self inductance
L1, L2 are almost constant; on the other hand, the mutual
inductance M highly varies with the gap change.
Fig. 10 shows the circuit diagram of the proposed system
using the GCSC as a primary side capacitor, which were evalu-
ated in the experiments. Table II shows the circuit parameters
of the experiments. The power supply voltage was adjusted
to achieve a constant output power of 1 kW for varying
condition of the horizontal gap. The secondary side capacitor
was selected to achieve full compensation for L2.
Three cases with different primary side capacitors were
evaluated as shown in the table. The first condition refereed as
fixed capacitor A was given to achieve full compensation for
L1. The second condition referred as fixed capacitor B was set
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0 10 20
-500
0
500
Cap
acit
or
Cu
rren
t [A
]
time [μs]
vo
ltag
e [V
]
(a)
-10
0
10
0 10 20
-500
0
500
Cap
acit
or
Cu
rren
t [A
]
time [μs]
vo
ltag
e [V
]
(b)
-10
0
10
0 10 20
-500
0
500
Cap
acit
or
Cu
rren
t [A
]
time [μs]
vo
ltag
e [V
]
(c)
Fig. 11. Measured waveforms of the current flowing through the GCSC (top), and the capacitor voltage of the GCSC (bottom) when the horizontal gap was(a)0 mm, (b)60 mm, (c)120 mm.
0 40 80 1200
1
Po
wer
fac
tor
Horizontal gap [mm]
Capacitor A
Capacitor B
GCSC
Fig. 12. Measured power factors at the output of the high frequency inverterwith varying horizontal gap .
0 40 80 12090
95
100
Eff
icie
ncy
[%
]
Horizontal gap [mm]
Capacitor A
Capacitor B
GCSC
Fig. 13. Measured efficiencies of the high frequency inverter with varyinghorizontal gap.
to achieve partial compensation for L1 to achieve a slightly
lagging power factor when the mutual inductance becomes
maximum. The third condition is using the GCSC, which is
the proposed one in this paper.
In the proposed system using the GCSC, the GCSC and
0 40 80 1200
10
Horizontal gap [mm]
Curr
ent
[A]
Capacitor A
Capacitor B
GCSC
Fig. 14. Measured output current magnitude in rms of the high frequencyinverter with varying horizontal gap.
a fixed capacitor were connected in series and divided the
voltage in order not to exceed the rated voltage of the GCSC.
The capacitance of the GCSC and the fixed capacitor were
designed to achieve full compensation for the self inductance
of the primary coil with the maximum compensation degree of
the GCSC with some margin. At the same time, the voltage
sharing of the GCSC should be high as possible within an
available voltage rating of the device in order to reduce
the conduction loss of the semiconductor switches. For the
experiments, the capacitor in the GCSC was designed to be
applied up to 800 V with considering the use of 1200 V SiC-
MOSFET. Then, the equivalent reactance in the fundamental
frequency of the GCSC was controlled to obtain a slightly
lagging power factor that delays current beyond the dead time.
In the experiment, the power flows between components,
P1, P2, P3, and P4, shown in Fig. 10, were measured and
losses were evaluated.
B. Experimental Results
Fig. 11 shows the waveforms of the voltage of the GCSC
and line current when the horizontal gap was set at 0 mm,
60 mm and 120 mm. As can be seen from the figure, the GCSC
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0
200
400
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5
10MOSFET Voltage
Free-Wheeling Diode current
MOSFET CurrentV
olt
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[V]
time [μs]
Cu
rren
t [A
]
(a)
0 5 10-200
0
200
400
-5
0
5
10MOSFET Voltage
Free-Wheeling Diode current
MOSFET Current
Vo
ltag
e [V
]
time [μs]C
urr
ent
[A]
(b)
0 5 10-200
0
200
400
-5
0
5
10MOSFET Voltage
Free-Wheeling Diode current
MOSFET Current
Vo
ltag
e [V
]
time [μs]
Cu
rren
t [A
]
(c)
Fig. 15. Measured current and voltage waveforms of the semiconductordevices of the inverter when the horizontal gap was 0 mm. (a)With fixedcapacitor A. (b)With fixed capacitor B. (c)GCSC was used as a primary sidecompensator.
worked as an equivalent variable capacitor since the capacitor
conduction section varied according to the horizontal gap. The
current conducted the semiconductor switches only in a short
section and the semiconductor loss was generated only in this
section.
Fig. 12 shows the output power factor of the high frequency
inverter. In the case of fixed capacitor A, output power factor
of the high frequency inverter was kept at almost unity power
factor. In the case of fixed capacitor B, a slightly lagging
power factor was achieved when the horizontal gap was
0 mm; however, the power factor was significantly decreased
according to the increase of the horizontal gap. On the other
hands, in the case of the GCSC, the slightly lagging power
factor was achieved regardless of the horizontal gap.
Fig. 13 shows the efficiencies of the high frequency inverter.
The fixed capacitor A achieved lowest efficiencies than the
others regardless of the horizontal gap. The fixed capacitor
B achieved higher efficiencies than the capacitor A; however,
0 5 10-100
0
100
200
-10
0
10
20
time [μs]
Vo
ltag
e [V
]
Cu
rren
t [A
]
MOSFET Voltage
Free-Wheeling Diode Current
MOSFET Current
(a)
0 5 10-100
0
100
200
-10
0
10
20
time [μs]
Vo
ltag
e [V
]
Cu
rren
t [A
]
MOSFET Voltage
Free-Wheeling Diode Current
MOSFET Current
(b)
0 5 10-100
0
100
200
-10
0
10
20
time [μs]
Vo
ltag
e [V
]
Cu
rren
t [A
]
MOSFET Voltage
Free-Wheeling Diode Current
MOSFET Current
(c)
Fig. 16. Measured current and voltage waveforms of the semiconductordevices of the inverter when the horizontal gap was 120 mm. (a)With fixedcapacitor A. (b)With fixed capacitor B. (c)GCSC was used as a primary sidecompensator.
the efficiency was remarkably decreased by the horizontal gap
increase. In the case of the GCSC, inverter efficiencies were
always high and the efficiency decrease in high horizontal gap
conditions was mitigated.
Fig. 14 shows the output current of the high frequency
inverter in rms. The output current of the high frequency
inverter, which is therefore same as the primary coil current,
to achieve the same output power was almost same regardless
of the primary compensation. Therefore, the conduction loss
of the MOSFETs can be same for all three conditions and the
sole difference for the inverter efficiency is switching losses.
Fig. 15 and Fig. 16 show the waveforms of current and
voltage applied to the MOSFETs and external free-wheeling
diodes when the horizontal gap was 0 mm and 120 mm,
respectively. With the fixed capacitor A, turn-on was hard-
switching and the switching loss can be though to be high
since the parasitic capacitance was charged before the turn-
on, and it was shorted by the turn-on. With the fixed capacitor
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0 40 80 1200
100
Horizontal gap [mm]
Lo
ss [
W]
InverterTransformer
Rectifier
(a)
0 40 80 1200
100
Horizontal gap [mm]
Lo
ss [
W]
Inverter
Transformer
Rectifier
(b)
0 40 80 1200
100
Horizontal gap [mm]
Lo
ss [
W]
GCSC
Transformer
Rectifier
Inverter
(c)
Fig. 17. Loss breakdown of the overall system with (a)Fixed capacitor A, (b)Fixed capacitor B, and (c)GCSC.
0 40 80 12090
95
100
Eff
icie
ncy
[%
]
Horizontal gap [mm]
Capacitor A
Capacitor B
GCSC
Fig. 18. Measured efficiencies of the overall system including the inverter,coils, rectifier, and compensators.
B, a completely soft turn-on and a turn-off at low current
were achieved when the horizontal gap was 0 mm. However,
the current at the turn-off was increased when the horizontal
gap was 120 mm due to the lower power factor at the inverter
output. That can increase the turn-off loss. On the other hands,
in the case of the GCSC, the slightly lagging power factor
achieved complete soft-switching turn-on and turn-off, and at
the same time, the current at the turn-off could be maintained
as low regardless of the horizontal gap.
Fig. 17 shows the overall system losses. The system losses
were divided into losses of the inverter, the coil part including
compensators (capacitors and GCSC) and the rectifier by
taking differences between the measured power, P1, P2, P3,
and P4, shown in Fig. 10. Then the losses of the GCSC were
calculated by using equation 6 and separated from the coil
losses. As can be seen from the figure, the loss of the coils
was almost same regardless of the compensation at the primary
side. The possible loss increase due to harmonic components
generated by the GCSC is considered to be negligible. The
loss of the GCSC was less than 3.5% of the overall loss and
was much smaller than other loss components. The inverter
loss using the GCSC was reduced by 8.4 W compared with
the system using fixed capacitors A and B at a maximum.
Fig. 18 shows the overall system efficiencies. The proposed
system using the GCSC achieved high efficiency compare
with the system using capacitor A in the whole area. When
the horizontal gap was larger than 80 mm, the proposed
system achieved high efficiency compare with the system
using the capacitor B since the loss reduction of the inverter
was remarkable and it exceeds the additional loss of the GCSC.
V. CONCLUSION
This paper proposed the WPT system using series com-
pensator named GCSC as a primary side capacitor. The
proposed system can control the output power factor of the
high frequency inverter and minimize its switching loss. The
main advantage of using the GCSC than the other topologies
like a full-bridge converter as a series compensator is low
semiconductor losses.
Experiments were conducted with an 1 kW laboratory
prototype. The proposed system achieved the efficiency im-
provement of the high frequency inverter, and considerably low
loss of the GCSC. The loss reduction of the inverter exceeded
the loss increase by the introduce of the GCSC; therefore,
the system overall efficiency was improved when the variation
degree of the coil parameters was large. The proposed system
reduced maximum loss by 8.4 W compared with the system
using fixed capacitors as a primary side capacitor.
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