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High Performance Narrow-Band Transceiver IC
Data Sheet ADF7021-N
Rev. B Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
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FEATURES Low power, narrow-band transceiver Frequency bands using dual VCO
80 MHz to 650 MHz 842 MHz to 916 MHz
Programmable IF filter bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz
Modulation schemes: 2FSK, 3FSK, 4FSK, MSK Spectral shaping: Gaussian and raised cosine filtering Data rates supported: 0.05 kbps to 24 kbps 2.3 V to 3.6 V power supply Programmable output power
−16 dBm to +13 dBm in 63 steps Automatic power amplifier (PA) ramp control Receiver sensitivity
−130 dBm at 100 bps, 2FSK −122 dBm at 1 kbps, 2FSK
On-chip image rejection calibration
On-chip VCO and fractional-N PLL On-chip, 7-bit ADC and temperature sensor Fully automatic frequency control loop (AFC) Digital received signal strength indication (RSSI) Integrated Tx/Rx switch 0.1 µA leakage current in power-down mode
APPLICATIONS Narrow-band, short range device (SRD) standards
ARIB STD-T67, ETSI EN 300 220, Korean SRD standard, FCC Part 15, FCC Part 90, FCC Part 95
Low cost, wireless data transfer Remote control/security systems Wireless metering Wireless medical telemetry service (WMTS) Home automation Process and building control Pagers
FUNCTIONAL BLOCK DIAGRAM
Figure 1.
Tx/RxCONTROL
AFCCONTROL
2FSK3FSK4FSK
DEMODULATOR
CLOCKAND DATARECOVERY
RSSI/
7-BIT ADC
GAIN
DIV R
RFOUT
LNA
PFDCP
OSC1 OSC2
N/N + 1DIV P
TEMPSENSOR
OSC CLKDIV
CLKOUT
TEST MUX
VCOIN CPOUT
LDO(1:4)
MUXOUTRSET CREG(1:4)
RLNA
RFIN
RFINB
CE
TxRxCLK
SWD
TxRxDATA
SERIALPORT
SLE
SDATA
SREAD
SCLK
IF FILTER
Σ-ΔMODULATOR
PA RAMP
L1 L2
LOG AMP
MUX
2FSK3FSK4FSK
MOD CONTROL
GAUSSIAN/RAISED COSINE
FILTER
3FSKENCODING
AGCCONTROL
MUX
÷1/÷2
VCO1
VCO2
÷2
0724
6-00
1
ADF7021-N* PRODUCT PAGE QUICK LINKSLast Content Update: 02/23/2017
COMPARABLE PARTSView a parametric search of comparable parts.
EVALUATION KITS• ADF7021-N Evaluation Boards
DOCUMENTATIONApplication Notes
• AN-0987: Designing a Wireless Transceiver System to Meet the Wireless M-Bus Standard
• AN-1182: Understanding and Optimizing the AFC Loop on the ADF7021 for Minimum Preamble
• AN-1258: Image Rejection Calibration on the ADF7021, ADF7021-N, and ADF7021-V
• AN-1285: ADF7021-N Radio Performance for Wireless Meter-Bus (WM-Bus), Mode N
• AN-1389: Recommended Rework Procedure for the Lead Frame Chip Scale Package (LFCSP)
• AN-771: ADSP-BF533 EZ-KIT Lite and ADF70xx Interface
• AN-772: A Design and Manufacturing Guide for the Lead Frame Chip Scale Package (LFCSP)
• AN-852: Using the Test DAC on the ADF702x to Implement Functions Such as Analog FM DEMOD, SNR Measurement, FEC Decoding, and PSK/4FSK Demodulation
• AN-859: RF Port Impedance Data, Matching, and External Component Selection for the ADF7020-1, ADF7021, and ADF7021-N
• AN-915: CDR Operation for ADF7020, ADF7020-1, ADF7021, and ADF7025
Data Sheet
• ADF7021-N: High Performance Narrow-Band Transceiver IC Data Sheet
SOFTWARE AND SYSTEMS REQUIREMENTS• ADF70xx Evaluation Software
• ADIismLINK Development Platform
TOOLS AND SIMULATIONS• ADIsimSRD Design Studio
REFERENCE MATERIALSSolutions Bulletins & Brochures
• Emerging Energy Applications Solutions Bulletin, Volume 10, Issue 4
Technical Articles
• Low Power, Low Cost, Wireless ECG Holter Monitor
• RF Meets Power Lines: Designing Intelligent Smart Grid Systems that Promote Energy Efficiency
• Smart Metering Technology Promotes Energy Efficiency for a Greener World
• The Use of Short Range Wireless in a Multi-Metering System
• Understand Wireless Short-Range Devices for Global License-Free Systems
• Wireless Short Range Devices and Narrowband Communications
• Wireless Technologies for Smart Meters: Focus on Water Metering
DESIGN RESOURCES• ADF7021-N Material Declaration
• PCN-PDN Information
• Quality And Reliability
• Symbols and Footprints
DISCUSSIONSView all ADF7021-N EngineerZone Discussions.
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ADF7021-N Data Sheet
Rev. B | Page 2 of 65
TABLE OF CONTENTS Features .............................................................................................. 1 Applications ....................................................................................... 1 Functional Block Diagram .............................................................. 1 Revision History ............................................................................... 3 General Description ......................................................................... 4 Specifications ..................................................................................... 5
RF and PLL Specifications ........................................................... 5 Transmission Specifications ........................................................ 6 Receiver Specifications ................................................................ 7 Digital Specifications ................................................................. 10 General Specifications ............................................................... 11 Timing Characteristics .............................................................. 12 Timing Diagrams ........................................................................ 13
Absolute Maximum Ratings .......................................................... 16 ESD Caution ................................................................................ 16
Pin Configuration and Function Descriptions ........................... 17 Typical Performance Characteristics ........................................... 19 Frequency Synthesizer ................................................................... 23
Reference Input ........................................................................... 23 MUXOUT .................................................................................... 24 Voltage Controlled Oscillator (VCO) ...................................... 25 Choosing Channels for Best System Performance ................. 26
Transmitter ...................................................................................... 27 RF Output Stage .......................................................................... 27 Modulation Schemes .................................................................. 27 Spectral Shaping ......................................................................... 29 Modulation and Filtering Options ........................................... 30 Transmit Latency ........................................................................ 30 Test Pattern Generator ............................................................... 30
Receiver Section .............................................................................. 31 RF Front End ............................................................................... 31 IF Filter ........................................................................................ 31 RSSI/AGC .................................................................................... 31
Demodulation, Detection, and CDR ....................................... 33 Receiver Setup ............................................................................. 35 Demodulator Considerations ................................................... 37 AFC Operation ........................................................................... 37 Automatic Sync Word Detection (SWD) ................................ 38
Applications Information .............................................................. 39 IF Filter Bandwidth Calibration ............................................... 39 LNA/PA Matching ...................................................................... 40 Image Rejection Calibration ..................................................... 41 Packet Structure and Coding .................................................... 43 Programming After Initial Power-Up ..................................... 43 Applications Circuit ................................................................... 46
Serial Interface ................................................................................ 47 Readback Format ........................................................................ 47 Interfacing to a Microcontroller/DSP ..................................... 49 Register 0—N Register ............................................................... 50 Register 1—VCO/Oscillator Register ...................................... 51 Register 2—Transmit Modulation Register ............................ 52 Register 3—Transmit/Receive Clock Register ........................ 53 Register 4—Demodulator Setup Register ............................... 54 Register 5—IF Filter Setup Register ......................................... 55 Register 6—IF Fine Cal Setup Register ................................... 56 Register 7—Readback Setup Register ...................................... 57 Register 8—Power-Down Test Register .................................. 58 Register 9—AGC Register ......................................................... 59 Register 10—AFC Register ....................................................... 60 Register 11—Sync Word Detect Register ................................ 61 Register 12—SWD/Threshold Setup Register ........................ 61 Register 13—3FSK/4FSK Demod Register ............................. 62 Register 14—Test DAC Register ............................................... 63 Register 15—Test Mode Register ............................................. 64
Outline Dimensions ....................................................................... 65 Ordering Guide .......................................................................... 65
Data Sheet ADF7021-N
Rev. B | Page 3 of 65
REVISION HISTORY 3/16—Rev. A to Rev. B Changed CP-48-3 to CP-48-5 ..................................... Throughout Changes to Features Section ............................................................ 1 Changes to General Description Section ....................................... 4 Changes to Figure 10 ...................................................................... 17 Updated Outline Dimensions ........................................................ 65 Changes to Ordering Guide ........................................................... 65 10/14—Rev. 0 to Rev. A Changes to Table 8 .......................................................................... 17 Changes to Figure 37 ...................................................................... 25 Change to Post Demodulator Filter Setup Section ..................... 35 Change to When to Use a Fine Calibration Section ................... 40
Change to Battery Voltage/ADCIN/Temperature Sensor Readback Section ............................................................................ 48 Change to Register 4—Demodulator Setup Register Section .... 54 Change to Register 6—IF Fine Cal Setup Register Section ........ 56 Change to Register 7—Readback Setup Register Section .......... 57 Change to Register 10—AFC Register Section ........................... 57 Changes to Fine Filter Calibration Description .......................... 44 Changes to POST_DEMOD_BW Calculation Description ..... 38, 59 Changes to Fine Filter Calibration Tone Timing ........................ 62 Change to AFC Range Description .............................................. 66 Changes to Temperature Readback Formula ........................ 52, 63 2/08—Revision 0: Initial Version
ADF7021-N Data Sheet
Rev. B | Page 4 of 65
GENERAL DESCRIPTION The ADF7021-N is a high performance, low power, narrow-band transceiver based on the ADF7021. The ADF7021-N has IF filter bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz, making it ideally suited to worldwide narrowband standards and particularly those that stipulate 12.5 kHz channel separation.
It is designed to operate in the narrow-band, license-free ISM bands and in the licensed bands with frequency ranges of 80 MHz to 650 MHz and 842 MHz to 916 MHz. The part has both Gaussian and raised cosine transmit data filtering options to improve spectral efficiency for narrow-band applications. It is suitable for circuit applications targeted at the Japanese ARIB STD-T67, the European ETSI EN 300 220, the Korean short range device regulations, the Chinese short range device regulations, and the North American FCC Part 15, Part 90, and Part 95 regulatory standards. A complete transceiver can be built using a small number of external discrete components, making the ADF7021-N very suitable for price-sensitive and area-sensitive applications.
The range of on-chip FSK modulation and data filtering options allows users greater flexibility in their choice of modulation schemes while meeting the tight spectral efficiency requirements. The ADF7021-N also supports protocols that dynamically switch among 2FSK, 3FSK, and 4FSK to maximize communication range and data throughput.
The transmit section contains two voltage controlled oscillators (VCOs) and a low noise fractional-N PLL with an output resolution of <1 ppm. The ADF7021-N has a VCO using an internal LC tank (421 MHz to 458 MHz, 842 MHz to 916 MHz) and a VCO using an external inductor as part of its tank circuit (80 MHz to 650 MHz). The dual VCO design allows dual-band operation where the user can transmit and/or receive at any frequency supported by the internal inductor VCO and can also transmit and/or receive at a particular frequency band supported by the external inductor VCO.
The frequency-agile PLL allows the ADF7021-N to be used in frequency-hopping, spread spectrum (FHSS) systems. Both VCOs operate at twice the fundamental frequency to reduce spurious emissions and frequency pulling problems.
The transmitter output power is programmable in 63 steps from −16 dBm to +13 dBm and has an automatic power ramp control to prevent spectral splatter and help meet regulatory standards. The transceiver RF frequency, channel spacing, and modulation are programmable using a simple 3-wire interface. The device operates with a power supply range of 2.3 V to 3.6 V and can be powered down when not in use.
A low IF architecture is used in the receiver (100 kHz), which minimizes power consumption and the external component count yet avoids dc offset and flicker noise at low frequencies. The IF filter has programmable bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz. The ADF7021-N supports a wide variety of programmable features including Rx linearity, sensitivity, and IF bandwidth, allowing the user to trade off receiver sensitivity and selectivity against current consumption, depending on the application. The receiver also features an automatic frequency control (AFC) loop with programmable pull-in range that allows the PLL to track out the frequency error in the incoming signal.
The receiver achieves an image rejection performance of 56 dB using an IR calibration scheme that does not require the use of an external RF source.
An on-chip ADC provides readback of the integrated tempera-ture sensor, external analog input, battery voltage, and RSSI signal, which provides savings on an ADC in some applications. The temperature sensor is accurate to ±10°C over the full operating temperature range of −40°C to +85°C. This accuracy can be improved by performing a 1-point calibration at room temperature and storing the result in memory
Data Sheet ADF7021-N
Rev. B | Page 5 of 65
SPECIFICATIONS VDD = 2.3 V to 3.6 V, GND = 0 V, TA = TMIN to TMAX, unless otherwise noted. Typical specifications are at VDD = 3 V, TA = 25°C. All measurements are performed with the EVAL-ADF7021-NDBxx using the PN9 data sequence, unless otherwise noted.
RF AND PLL SPECIFICATIONS
Table 1. Parameter Min Typ Max Unit Test Conditions/Comments RF CHARACTERISTICS See Table 9 for required VCO_BIAS and
VCO_ADJUST settings Frequency Ranges (Direct Output) 160 650 MHz External inductor VCO
842 916 MHz Internal inductor VCO Frequency Ranges (RF Divide-by-2 Mode) 80 325 MHz External inductor VCO, RF divide-by-2 enabled
421 458 MHz Internal inductor VCO, RF divide-by-2 enabled Phase Frequency Detector (PFD) Frequency1 RF/256 24 MHz
PHASE-LOCKED LOOP (PLL) VCO Gain2
868 MHz, Internal Inductor VCO 67 MHz/V VCO_ADJUST = 0, VCO_BIAS = 8 426 MHz, Internal Inductor VCO 45 MHz/V VCO_ADJUST = 0, VCO_BIAS = 8 426 MHz, External Inductor VCO 27 MHz/V VCO_ADJUST = 0, VCO_BIAS = 3 160 MHz, External Inductor VCO 6 MHz/V VCO_ADJUST = 0, VCO_BIAS = 2
Phase Noise (In-Band) 868 MHz, Internal Inductor VCO −97 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8 433 MHz, Internal Inductor VCO −103 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8 426 MHz, External Inductor VCO −95 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 9.84 MHz, VCO_BIAS = 3 Phase Noise (Out-of-Band) −124 dBc/Hz 1 MHz offset, fRF = 433 MHz, PA = 10 dBm,
VDD = 3.0 V, PFD = 19.68 MHz, VCO_BIAS = 8 Normalized In-Band Phase Noise Floor3 −203 dBc/Hz PLL Settling 40 µs Measured for a 10 MHz frequency step to within
5 ppm accuracy, PFD = 19.68 MHz, loop bandwidth (LBW) = 100 kHz
REFERENCE INPUT Crystal Reference4 3.625 24 MHz External Oscillator4, 5 3.625 24 MHz Crystal Start-Up Time6
XTAL Bias = 20 µA 0.930 ms 10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V XTAL Bias = 35 µA 0.438 ms 10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V
Input Level for External Oscillator7 OSC1 0.8 V p-p Clipped sine wave OSC2 CMOS levels V
ADC PARAMETERS INL ±0.4 LSB VDD = 2.3 V to 3.6 V, TA = 25°C DNL ±0.4 LSB VDD = 2.3 V to 3.6 V, TA = 25°C
1 The maximum usable PFD at a particular RF frequency is limited by the minimum N divide value. 2 VCO gain measured at a VCO tuning voltage of 0.7 V. The VCO gain varies across the tuning range of the VCO. The software package ADIsimPLL™ can be used to model this
variation. 3 This value can be used to calculate the in-band phase noise for any operating frequency. Use the following equation to calculate the in-band phase noise performance
as seen at the power amplifier (PA) output: −203 + 10 log(fPFD) + 20 logN. 4 Guaranteed by design. Sample tested to ensure compliance. 5 A TCXO, VCXO, or OCXO can be used as an external oscillator. 6 Crystal start-up time is the time from chip enable (CE) being asserted to correct clock frequency on the CLKOUT pin. 7 Refer to the Reference Input section for details on using an external oscillator.
ADF7021-N Data Sheet
Rev. B | Page 6 of 65
TRANSMISSION SPECIFICATIONS
Table 2. Parameter Min Typ Max Unit Test Conditions/Comments DATA RATE
2FSK, 3FSK 0.05 18.51 kbps IF_FILTER_BW = 18.5 kHz 4FSK 0.05 24 kbps IF_FILTER_BW = 18.5 kHz
MODULATION Frequency Deviation (fDEV)2 0.056 28.26 kHz PFD = 3.625 MHz
0.306 156 kHz PFD = 20 MHz Deviation Frequency Resolution 56 Hz PFD = 3.625 MHz Gaussian Filter BT 0.5 Raised Cosine Filter Alpha 0.5/0.7 Programmable
TRANSMIT POWER Maximum Transmit Power3 +13 dBm VDD = 3.0 V, TA = 25°C Transmit Power Variation vs.
Temperature ±1 dB −40°C to +85°C
Transmit Power Variation vs. VDD ±1 dB 2.3 V to 3.6 V at 915 MHz, TA = 25°C Transmit Power Flatness ±1 dB 902 MHz to 928 MHz, 3 V, TA = 25°C Programmable Step Size 0.3125 dB −16 dBm to +13 dBm
ADJACENT CHANNEL POWER (ACP) 426 MHz, External Inductor VCO PFD = 9.84 MHz
12.5 kHz Channel Spacing −50 dBc Gaussian 2FSK modulation, measured in a ±4.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ARIB STD-T67
25 kHz Channel Spacing −50 dBc Gaussian 2FSK modulation, measured in a ±8 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ARIB STD-T67
868 MHz, Internal Inductor VCO PFD = 19.68 MHz 12.5 kHz Channel Spacing −46 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ETSI EN 300 220
25 kHz Channel Spacing −43 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ETSI EN 300 220
433 MHz, Internal Inductor VCO PFD = 19.68 MHz 12.5 kHz Channel Spacing −50 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ETSI EN 300 220
25 kHz Channel Spacing −47 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ETSI EN 300 220
OCCUPIED BANDWIDTH 99.0% of total mean power; 12.5 kHz channel spacing (2.4 kbps PN9 data, 1.2 kHz frequency deviation); 25 kHz channel spacing (9.6 kbps PN9 data, 2.4 kHz frequency deviation)
2FSK Gaussian Data Filtering 12.5 kHz Channel Spacing 3.9 kHz 25 kHz Channel Spacing 9.9 kHz
2FSK Raised Cosine Data Filtering 12.5 kHz Channel Spacing 4.4 kHz 25 kHz Channel Spacing 10.2 kHz
3FSK Raised Cosine Filtering 12.5 kHz Channel Spacing 3.9 kHz 25 kHz Channel Spacing 9.5 kHz
4FSK Raised Cosine Filtering 19.2 kbps PN9 data, 1.2 kHz frequency deviation 25 kHz Channel Spacing 13.2 kHz
Data Sheet ADF7021-N
Rev. B | Page 7 of 65
Parameter Min Typ Max Unit Test Conditions/Comments SPURIOUS EMISSIONS
Reference Spurs −65 dBc 100 kHz loop bandwidth HARMONICS4 13 dBm output power, unfiltered conductive/filtered conductive
Second Harmonic −35/−52 dBc Third Harmonic −43/−60 dBc All Other Harmonics −36/−65 dBc
OPTIMUM PA LOAD IMPEDANCE5 fRF = 915 MHz 39 + j61 Ω fRF = 868 MHz 48 + j54 Ω fRF = 450 MHz 98 + j65 Ω fRF = 426 MHz 100 + j65 Ω fRF = 315 MHz 129 + j63 Ω fRF = 175 MHz 173 + j49 Ω
1 Using Gaussian or raised cosine filtering. Choose the frequency deviation to ensure that the transmit-occupied signal bandwidth is within the receiver IF filter bandwidth. 2 For the definition of frequency deviation, refer to the Register 2—Transmit Modulation Register section. 3 Measured as maximum unmodulated power. 4 Conductive filtered harmonic emissions measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one capacitor). 5 For matching details, refer to the LNA/PA Matching section.
RECEIVER SPECIFICATIONS
Table 3. Parameter Min Typ Max Unit Test Conditions/Comments SENSITIVITY Bit error rate (BER) = 10−3, low noise amplifier (LNA)
and power amplifier (PA) matched separately 2FSK
Sensitivity at 0.1 kbps −130 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz
Sensitivity at 0.25 kbps −127 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz
Sensitivity at 1 kbps −122 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz
Sensitivity at 9.6 kbps −115 dBm fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW = 18.5 kHz
Gaussian 2FSK Sensitivity at 0.1 kbps −129 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz Sensitivity at 0.25 kbps −127 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz Sensitivity at 1 kbps −121 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz Sensitivity at 9.6 kbps −114 dBm fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz GMSK
Sensitivity at 9.6 kbps −113 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW = 18.5 kHz
Raised Cosine 2FSK Sensitivity at 0.25 kbps −127 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz Sensitivity at 1 kbps −121 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz Sensitivity at 9.6 kbps −114 dBm fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz
ADF7021-N Data Sheet
Rev. B | Page 8 of 65
Parameter Min Typ Max Unit Test Conditions/Comments 3FSK
Sensitivity at 9.6 kbps −110 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW = 18.5 kHz, Viterbi detection on
Raised Cosine 3FSK Sensitivity at 9.6 kbps −110 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz, alpha = 0.5, Viterbi detection on 4FSK
Sensitivity at 9.6 kbps −112 dBm fDEV (inner) = 1.2 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz
Raised Cosine 4FSK Sensitivity at 9.6 kbps −109 dBm fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_FILTER_BW = 13.5 kHz, alpha = 0.5 INPUT IP3 Two-tone test, fLO = 860 MHz, F1 = fLO + 100 kHz,
F2 = fLO − 800 kHz Low Gain Enhanced Linearity
Mode −3 dBm LNA_GAIN = 3, MIXER_LINEARITY = 1
Medium Gain Mode −13.5 dBm LNA_GAIN = 10, MIXER_LINEARITY = 0 High Sensitivity Mode −24 dBm LNA_GAIN = 30, MIXER_LINEARITY = 0
ADJACENT CHANNEL REJECTION 868 MHz Wanted signal is 3 dB above the sensitivity point
(BER = 10−3); unmodulated interferer is at the center of the adjacent channel; rejection measured as the difference between the interferer level and the wanted signal level in dB
12.5 kHz Channel Spacing 40 dB 9 kHz IF_FILTER_BW 25 kHz Channel Spacing 39 dB 18.5 kHz IF_FILTER_BW
426 MHz Wanted signal is 3 dB above the reference sensitivity point (BER = 10−2); modulated interferer (same modulation as wanted signal) at the center of the adjacent channel; rejection measured as the difference between the interferer level and reference sensitivity level in dB
12.5 kHz Channel Spacing 40 dB 9 kHz IF_FILTER_BW, compliant with ARIB STD-T67 25 kHz Channel Spacing 39 dB 18.5 kHz IF_FILTER_BW, compliant with ARIB STD-T67
CO-CHANNEL REJECTION Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is 3 dB above the sensitivity point (BER = 10−3), modu-lated interferer
868 MHz −5 dB IMAGE CHANNEL REJECTION Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is
10 dB above the sensitivity point (BER = 10−3); modu-lated interferer (2FSK, 9.6 kbps, ±4 kHz deviation) is placed at the image frequency of fRF − 200 kHz; the interferer level is increased until BER = 10−3
868 MHz 26/39 dB Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C 450 MHz, Internal Inductor
VCO 29/50 dB Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C
BLOCKING Wanted signal is 10 dB above the input sensitivity level; CW interferer level is increased until BER = 10−3
±1 MHz 69 dB ±2 MHz 75 dB ±5 MHz 78 dB ±10 MHz 78.5 dB
SATURATION (MAXIMUM INPUT LEVEL)
12 dBm 2FSK mode, BER = 10−3
Data Sheet ADF7021-N
Rev. B | Page 9 of 65
Parameter Min Typ Max Unit Test Conditions/Comments RSSI
Range at Input2 −120 to −47 dBm Linearity ±2 dB Input power range = −100 dBm to −47 dBm Absolute Accuracy ±3 dB Input power range = −100 dBm to −47 dBm Response Time 390 µs See the RSSI/AGC section
AFC Pull-In Range 0.5 1.5 × IF_
FILTER_BW kHz The range is programmable in Register 10
(R10_DB[24:31]) Response Time 64 Bits Accuracy 0.5 kHz Input power range = −100 dBm to +12 dBm
Rx SPURIOUS EMISSIONS3 Internal Inductor VCO −91/−91 dBm <1 GHz at antenna input, unfiltered conductive/filtered
conductive −52/−70 dBm >1 GHz at antenna input, unfiltered conductive/filtered
conductive External Inductor VCO −62/−72 dBm <1 GHz at antenna input, unfiltered conductive/filtered
conductive −64/−85 dBm >1 GHz at antenna input, unfiltered conductive/filtered
conductive LNA INPUT IMPEDANCE RFIN to RFGND
fRF = 915 MHz 24 − j60 Ω fRF = 868 MHz 26 − j63 Ω fRF = 450 MHz 63 − j129 Ω fRF = 426 MHz 68 − j134 Ω fRF = 315 MHz 96 − j160 Ω fRF = 175 MHz 178 − j190 Ω
1 Calibration of the image rejection used an external RF source. 2 For received signal levels < −100 dBm, it is recommended to average the RSSI readback value over a number of samples to improve the RSSI accuracy at low input powers. 3 Filtered conductive receive spurious emissions are measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one
capacitor).
ADF7021-N Data Sheet
Rev. B | Page 10 of 65
DIGITAL SPECIFICATIONS
Table 4. Parameter Min Typ Max Unit Test Conditions/Comments TIMING INFORMATION
Chip Enabled to Regulator Ready 10 µs CREG (1:4) = 100 nF Chip Enabled to Tx Mode 32-bit register write time = 50 µs
TCXO Reference 1 ms XTAL 2 ms
Chip Enabled to Rx Mode 32-bit register write time = 50 µs, IF filter coarse calibration only
TCXO Reference 1.2 ms XTAL 2.2 ms
Tx-to-Rx Turnaround Time 390 µs + (5 × tBIT) Time to synchronized data out, includes AGC settling (three AGC levels)and CDR synchronization; see the AGC Information and Timing section for more details; tBIT = data bit period
LOGIC INPUTS Input High Voltage, VINH 0.7 × VDD V Input Low Voltage, VINL 0.2 × VDD V Input Current, IINH/IINL ±1 µA Input Capacitance, CIN 10 pF Control Clock Input 50 MHz
LOGIC OUTPUTS Output High Voltage, VOH DVDD − 0.4 V IOH = 500 µA Output Low Voltage, VOL 0.4 V IOL = 500 µA CLKOUT Rise/Fall 5 ns CLKOUT Load 10 pF
Data Sheet ADF7021-N
Rev. B | Page 11 of 65
GENERAL SPECIFICATIONS
Table 5. Parameter Min Typ Max Unit Test Conditions/Comments TEMPERATURE RANGE (TA) −40 +85 °C POWER SUPPLIES
Voltage Supply, VDD 2.3 3.6 V All VDD pins must be tied together TRANSMIT CURRENT CONSUMPTION1 VDD = 3.0 V, PA is matched into 50 Ω
868 MHz VCO_BIAS = 8 0 dBm 20.2 mA 5 dBm 24.7 mA 10 dBm 32.3 mA
450 MHz, Internal Inductor VCO VCO_BIAS = 8 0 dBm 19.9 mA 5 dBm 23.2 mA 10 dBm 29.2 mA
426 MHz, External Inductor VCO VCO_BIAS = 2 0 dBm 13.5 mA 5 dBm 17 mA 10 dBm 23.3 mA
RECEIVE CURRENT CONSUMPTION VDD = 3.0 V 868 MHz VCO_BIAS = 8
Low Current Mode 22.7 mA High Sensitivity Mode 24.6 mA
433MHz, Internal Inductor VCO VCO_BIAS = 8 Low Current Mode 24.5 mA High Sensitivity Mode 26.4 mA
426 MHz, External Inductor VCO VCO_BIAS = 2 Low Current Mode 17.5 mA High Sensitivity Mode 19.5 mA
POWER-DOWN CURRENT CONSUMPTION Low Power Sleep Mode 0.1 1 µA CE low
1 The transmit current consumption tests used the same combined PA and LNA matching network as that used on the EVAL-ADF7021-NDBxx evaluation boards.
Improved PA efficiency is achieved by using a separate PA matching network.
ADF7021-N Data Sheet
Rev. B | Page 12 of 65
TIMING CHARACTERISTICS VDD = 3 V ± 10%, DGND = AGND = 0 V, TA = 25°C, unless otherwise noted. Guaranteed by design but not production tested.
Table 6. Parameter Limit at TMIN to TMAX Unit Test Conditions/Comments t1 >10 ns SDATA to SCLK setup time t2 >10 ns SDATA to SCLK hold time t3 >25 ns SCLK high duration t4 >25 ns SCLK low duration t5 >10 ns SCLK to SLE setup time t6 >20 ns SLE pulse width t8 <25 ns SCLK to SREAD data valid, readback t9 <25 ns SREAD hold time after SCLK, readback t10 >10 ns SCLK to SLE disable time, readback t11 5 < t11 < (¼ × tBIT) ns TxRxCLK negative edge to SLE t12 >5 ns TxRxDATA to TxRxCLK setup time (Tx mode) t13 >5 ns TxRxCLK to TxRxDATA hold time (Tx mode) t14 >¼ × tBIT µs TxRxCLK negative edge to SLE t15 >¼ × tBIT µs SLE positive edge to positive edge of TxRxCLK
Data Sheet ADF7021-N
Rev. B | Page 13 of 65
TIMING DIAGRAMS Serial Interface
Figure 2. Serial Interface Timing Diagram
Figure 3. Serial Interface Readback Timing Diagram
2FSK/3FSK Timing
Figure 4. TxRxDATA/TxRxCLK Timing Diagram in Receive Mode
Figure 5. TxRxDATA/TxRxCLK Timing Diagram in Transmit Mode
SCLK
SLE
DB31 (MSB) DB30 DB2DB1
(CONTROL BIT C2)SDATA
DB0 (LSB)(CONTROL BIT C1)
t6
t1 t2
t3 t4
t5 0724
6-00
2
t8
t3
t1 t2
t10
t9
X RV16 RV15 RV2
SCLK
SDATA
SLE
SREAD
REG7 DB0(CONTROL BIT C1)
RV1 X
0724
6-00
3
TxRxCLK
DATATxRxDATA
±1 × DATA RATE/32 1/DATA RATE
0724
6-00
4
TxRxCLK
DATATxRxDATA
SAMPLEFETCH
1/DATA RATE
0724
6-00
5
ADF7021-N Data Sheet
Rev. B | Page 14 of 65
4FSK Timing
In 4FSK receive mode, MSB/LSB synchronization is guaranteed by SWD in the receive bit stream.
Figure 6. Receive-to-Transmit Timing Diagram in 4FSK Mode
Figure 7. Transmit-to-Receive Timing Diagram in 4FSK Mode
Rx SYMBOLMSB
Rx SYMBOLLSB
Rx SYMBOLMSB
Rx SYMBOLLSB
Tx SYMBOLMSB
Tx SYMBOLLSBTxRxDATA
TxRxCLK
SLE
Rx MODE Tx MODE
REGISTER 0 WRITE
SWITCH FROM Rx TO Tx
Tx/Rx MODE
Tx SYMBOLMSB
t11 t12
t13
tBIT
tSYMBOL
0724
6-07
4
Tx SYMBOLMSB
Tx SYMBOLLSB
Tx SYMBOLMSB
Tx SYMBOLLSB
Rx SYMBOLLSB
Rx SYMBOLMSBTxRxDATA
TxRxCLK
SLE
Tx MODE Rx MODE
REGISTER 0 WRITE
SWITCH FROM Tx TO Rx
Tx/Rx MODE
t15
t14
tBIT
tSYMBOL
0724
6-07
5
Data Sheet ADF7021-N
Rev. B | Page 15 of 65
UART/SPI Mode
UART mode is enabled by setting R0_DB28 to 1. SPI mode is enabled by setting R0_DB28 to 1 and setting R15_DB[17:19] to 0x7. The transmit/receive data clock is available on the CLKOUT pin.
Figure 8. Transmit Timing Diagram in UART/SPI Mode
Figure 9. Receive Timing Diagram in UART/SPI Mode
Tx BIT Tx BIT Tx BIT Tx BITTxRxCLK
(TRANSMIT DATA INPUTIN UART/SPI MODE.)
CLKOUT(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.NOT USED IN UART MODE.)
Tx MODETx/Rx MODE
TxRxDATA(RECEIVE DATA OUTPUT
IN UART/SPI MODE.) HIGH-Z
Tx BIT
tBIT
FETCH SAMPLE
0724
6-08
2
Rx BIT Rx BIT Rx BIT Rx BIT
TxRxCLK(TRANSMIT DATA INPUT
IN UART/SPI MODE.)
CLKOUT(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.NOT USED IN UART MODE.)
Rx MODETx/Rx MODE
TxRxDATA(RECEIVE DATA OUTPUT
IN UART/SPI MODE.)
HIGH-Z
Rx BIT
tBIT
FETCH SAMPLE
0724
6-07
8
ADF7021-N Data Sheet
Rev. B | Page 16 of 65
ABSOLUTE MAXIMUM RATINGS TA = 25°C, unless otherwise noted.
Table 7. Parameter Rating VDD to GND1 −0.3 V to +5 V Analog I/O Voltage to GND −0.3 V to AVDD + 0.3 V Digital I/O Voltage to GND −0.3 V to DVDD + 0.3 V Operating Temperature Range
Industrial (B Version) −40°C to +85°C Storage Temperature Range −65°C to +125°C Maximum Junction Temperature 150°C MLF θJA Thermal Impedance 26°C/W Reflow Soldering
Peak Temperature 260°C Time at Peak Temperature 40 sec
1 GND = CPGND = RFGND = DGND = AGND = 0.
Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability.
This device is a high performance RF integrated circuit with an ESD rating of <2 kV and it is ESD sensitive. Take proper precautions for handling and assembly.
ESD CAUTION
Data Sheet ADF7021-N
Rev. B | Page 17 of 65
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 10. Pin Configuration
Table 8. Pin Function Descriptions Pin No. Mnemonic Description 1 VCOIN The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO).
The higher the tuning voltage, the higher the output frequency. 2 CREG1 Regulator Voltage for PA Block. Place a series 3.9 Ω resistor and a 100 nF capacitor between this pin and
ground for regulator stability and noise rejection. 3 VDD1 Voltage Supply for PA Block and VCO cores. Place decoupling capacitors of 0.1 µF and 100 pF as close as
possible to this pin. Tie all VDD pins together. 4 RFOUT The modulated signal is available at this pin. Output power levels are from −16 dBm to +13 dBm. Impedance
match the output to the desired load using suitable components (see the Transmitter section). 5 RFGND Ground for Output Stage of Transmitter. Tie all GND pins together. 6 RFIN LNA Input for Receiver Section. Input matching is required between the antenna and the differential LNA
input to ensure maximum power transfer (see the LNA/PA Matching section). 7 RFINB Complementary LNA Input. (See the LNA/PA Matching section.) 8 RLNA External Bias Resistor for LNA. Optimum resistor is 1.1 kΩ with 5% tolerance. 9 VDD4 Voltage Supply for LNA/MIXER Block. Decouple this pin to ground with a 10 nF capacitor. 10 RSET External Resistor. Sets charge pump current and some internal bias currents. Use a 3.6 kΩ resistor with 5% tolerance. 11 CREG4 Regulator Voltage for LNA/MIXER Block. Place a 100 nF capacitor between this pin and GND for regulator
stability and noise rejection. 12, 19, 22 GND4 Ground for LNA/MIXER Block. 13 to 18 MIX_I, MIX_I,
MIX_Q, MIX_Q, FILT_I, FILT_I
Signal Chain Test Pins. These pins are high impedance under normal conditions; leave the pins unconnected.
20, 21, 23 FILT_Q, FILT_Q, TEST_A
Signal Chain Test Pins. These pins are high impedance under normal conditions; leave the pins unconnected.
24 CE Chip Enable. Bringing CE low puts the ADF7021-N into complete power-down. Register values are lost when CE is low, and the part must be reprogrammed after CE is brought high.
25 SLE Load Enable, CMOS Input. When SLE goes high, the data stored in the shift registers is loaded into one of the four latches. A latch is selected using the control bits.
26 SDATA Serial Data Input. The serial data is loaded MSB first with the four LSBs as the control bits. This pin is a high impedance CMOS input.
27 SREAD Serial Data Output. This pin is used to feed readback data from the ADF7021-N to the microcontroller. The SCLK input is used to clock each readback bit (for example, AFC or ADC) from the SREAD pin.
28 SCLK Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into the 32-bit shift register on the CLK rising edge. This pin is a digital CMOS input.
29 GND2 Ground for Digital Section.
0724
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6
NOTES1. THE EXPOSED PAD MUST BE CONNECTED TO GND.
VCOINCREG1
VDD1RFOUTRFGND
RFINRFINB
RLNAVDD4RSET
CREG4GND4
CLKOUT
TxRxDATATxRxCLK
SWDVDD2CREG2ADCINGND2SCLKSREADSDATASLE
CVC
OG
ND
1L1 G
ND
L2 VDD
CPO
UT
CR
EG3
VDD
3O
SC1
OSC
2M
UXO
UT
MIX
_IM
IX_I
MIX
_QM
IX_Q
FILT
_IFI
LT_I
GN
D4
FILT
_QFI
LT_Q
GN
D4
TEST
_A CE
1234567
242322212019181716151413
4445464748 43 42 41 40 39 38 37
ADF7021-NTOP VIEW
(Not to Scale)
252627282930313233343536
89
101112
ADF7021-N Data Sheet
Rev. B | Page 18 of 65
Pin No. Mnemonic Description 30 ADCIN Analog-to-Digital Converter Input. The internal 7-bit ADC can be accessed through this pin. Full scale is 0 V to
1.9 V. Readback is made using the SREAD pin. 31 CREG2 Regulator Voltage for Digital Block. Place a 100 nF capacitor between this pin and ground for regulator
stability and noise rejection. 32 VDD2 Voltage Supply for Digital Block. Place a decoupling capacitor of 10 nF as close as possible to this pin. 33 SWD Sync Word Detect. The ADF7021-N asserts this pin when it has found a match for the sync word sequence
(see the Register 11—Sync Word Detect Register section). This provides an interrupt for an external microcontroller indicating that valid data is being received.
34 TxRxDATA Transmit Data Input/Received Data Output. This is a digital pin, and normal CMOS levels apply. In UART/SPI mode, this pin provides an output for the received data in receive mode. In transmit UART/SPI mode, this pin is high impedance (see the Interfacing to a Microcontroller/DSP section).
35 TxRxCLK Outputs the data clock in both receive and transmit modes. This is a digital pin, and normal CMOS levels apply. The positive clock edge is matched to the center of the received data. In transmit mode, this pin outputs an accurate clock to latch the data from the microcontroller into the transmit section at the exact required data rate. In UART/SPI mode, this pin is used to input the transmit data in transmit mode. In receive UART/SPI mode, this pin is high impedance (see the Interfacing to a Microcontroller/DSP section).
36 CLKOUT A divided-down version of the crystal reference with output driver. The digital clock output can be used to drive several other CMOS inputs such as a microcontroller clock. The output has a 50:50 mark-space ratio and is inverted with respect to the reference. Place a series 1 kΩ resistor as close as possible to the pin in applications where the CLKOUT feature is being used.
37 MUXOUT Provides the DIGITAL_LOCK_DETECT signal. This signal is used to determine if the PLL is locked to the correct frequency. It also provides other signals such as REGULATOR_READY, which is an indicator of the status of the serial interface regulator (see the MUXOUT section for more information).
38 OSC2 Connect the reference crystal between this pin and OSC1. A TCXO reference can be used by driving this pin with CMOS levels and disabling the internal crystal oscillator.
39 OSC1 Connect the reference crystal between this pin and OSC2. A TCXO reference can be used by driving this pin with ac-coupled 0.8 V p-p levels and by enabling the internal crystal oscillator.
40 VDD3 Voltage Supply for the Charge Pump and PLL Dividers. Decouple this pin to ground with a 10 nF capacitor. 41 CREG3 Regulator Voltage for Charge Pump and PLL Dividers. Place a 100 nF capacitor between this pin and ground
for regulator stability and noise rejection. 42 CPOUT Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The
integrated current changes the control voltage on the input to the VCO. 43 VDD Voltage Supply for XTAL and bandgap core. Decouple this pin to ground with a 10 nF capacitor. 44, 46 L2, L1 External VCO Inductor Pins. If using an external VCO inductor, connect a chip inductor across these pins to set
the VCO operating frequency. If using the internal VCO inductor, these pins can be left floating. See the Voltage Controlled Oscillator (VCO) section for more information.
45, 47 GND, GND1 Grounds for VCO Block. 48 CVCO Place a 22 nF capacitor between this pin and CREG1 to reduce VCO noise. 49 EPAD Exposed Pad. The exposed pad must be connected to GND.
Data Sheet ADF7021-N
Rev. B | Page 19 of 65
TYPICAL PERFORMANCE CHARACTERISTICS
Figure 11. Phase Noise Response at 900 MHz, VDD = 2.3 V
Figure 12. RF Output Power vs. PA Setting
Figure 13. PA Output Harmonic Response with T-Stage LC Filter
Figure 14. Output Spectrum in 2FSK and GFSK Modes
Figure 15. Output Spectrum in 2FSK and Raised Cosine 2FSK Modes
Figure 16. Output Spectrum in 4FSK and Raised Cosine 4FSK Modes
FREQUENCY OFFSET (kHz)
PHA
SE N
OIS
E (d
Bc/
Hz)
–150
–140
–130
–120
–110
–100
–90
–80
–70
1 10 100 1000 10000
RF FREQ = 900MHzVDD = 2.3VTEMPERATURE = 25°CVCO_BIAS = 8VCO_ADJUST = 3ICP = 0.8mA
ICP = 1.4mA
ICP = 2.2mA
0724
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0
–40–36–32–28–24–20–16–12
–8–4
048
1216
0 4 8 12 16 20 24 28 32 36 40 44 48 52 56 60
PA SETTING
RF
OU
TPU
T PO
WER
(dB
m)
PA_BIAS = 5µA
PA_BIAS = 11µA
PA_BIAS = 9µA
PA_BIAS = 7µA
0724
6-05
1
VBW 100HzSTART 300MHzRES BW 100Hz SWEEP 385.8ms (601pts)
STOP 3.5GHz
RF FREQ = 440MHzOUTPUT POWER = 10dBmFILTER = T-STAGE LC FILTERMARKER Δ = 52.2dB
1R
1
0724
6-05
0CENTER 869.5 25MHzRES BW 300Hz SWEEP 2.118s (601pts)
SPAN 50kHz
DR = 9.6kbpsDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz
VBW 300Hz
GFSK
2FSK
0724
6-04
7
CENTER 869.5 25MHzRES BW 300Hz SWEEP 2.118s (601pts)VBW 300Hz
SPAN 50kHz
DR = 9.6kbpsDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz
RC2FSK
2FSK
0724
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8
VBW 300HzCENTER 869.493 8MHzRES BW 300Hz SWEEP 4.237s (601pts)
SPAN 100kHz
SR = 4.8ksym/sDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz
RC4FSK
4FSK
0724
6-04
9
ADF7021-N Data Sheet
Rev. B | Page 20 of 65
Figure 17. Output Spectrum in 3FSK and Raised Cosine 3FSK Modes
Figure 18. Output Spectrum in Maximum Hold
for Various PA Ramp Rate Options
Figure 19. 2FSK Sensitivity vs. VDD and Temperature, fRF = 868 MHz
Figure 20. 2FSK Sensitivity vs. VDD and Temperature, fRF = 135 MHz
Figure 21. 3FSK Sensitivity vs. VDD and Temperature, fRF = 440 MHz
Figure 22. 4FSK Sensitivity vs. VDD and Temperature, fRF = 420 MHz
RES BW 300HzCENTER 869.5MHz VBW 300Hz SPAN 50kHz
SWEEP2.226s (401pts)
REF 15dBmSAMP LOG 10dB/ ATTEN 25dB
VAVG 100V1 V2S3 FC
DR = 9.6kbpsDATA = PRS9fDEV = 2.4kHzRF FREQ = 869.5MHz
3FSK
RC3FSK
0724
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0
FREQUENCY OFFSET (kHz)
OU
TPU
T PO
WER
(dB
m)
0
–10
10
–20
–30
–40
–100 –50 500 100
–50
–60
RAMP RATE:CW ONLY256 CODES/BIT128 CODES/BIT64 CODES/BIT32 CODES/BIT
TRACE = MAX HOLDPA ON/OFF RATE = 3HzPA ON/OFF CYCLES = 10,000VDD = 3.0V
0724
6-06
8
–8
–7
–6
–5
–4
–3
–2
–1
0
–122 –120 –118 –116 –114 –112 –110 –108 –106 –104
3.0V, +25°C
3.6V, –40°C2.3V, +85°C
RF INPUT POWER (dBm)
LOG
BER
DATA RATE = 9.6kbpsfDEV = 4kHzRF FREQ = 868MHzIF BW = 25kHz
0724
6-05
2
–8
–7
–6
–5
–4
–3
–2
–1
0
–130 –128 –126 –124 –122 –120 –118 –116 –114 –112 –110 –108
3.6V, –40°C
3.0V, +25°C2.3V, +85°C
RF INPUT POWER (dBm)
LOG
BER
DATA RATE = 1kbpsfDEV = 1kHzRF FREQ = 135MHzIF BW = 12.5kHz
0724
6-05
3
–8
–7
–6
–5
–4
–3
–2
–1
0
–120 –115 –110 –105 –100 –95
RF INPUT POWER (dBm)
LOG
BER
2.3V +25°C3.0V +25°C3.6V +25°C2.3V –40°C3.0V –40°C3.6V –40°C2.3V +85°C3.0V +85°C3.6V +85°C
3FSK MODULATIONDATA RATE = 9.6kbpsfDEV = 2.4kHzMOD INDEX = 0.5RF FREQ = 440 MHz
0724
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5
–8
–7
–6
–5
–4
–3
–2
–1
0
–120 –115 –110 –105 –100 –95
RF INPUT POWER (dBm)
LOG
BER
2.3V +25°C3.0V +25°C3.6V +25°C2.3V –40°C3.0V –40°C3.6V –40°C2.3V +85°C3.0V +85°C3.6V +85°C
DATA RATE = 19.6kbpsSYMBOL RATE = 9.8ksym/sfDEV (inner) = 2.4kHzMOD INDEX = 0.5RF FREQ = 420MHzIF BW = 12.5kHz
0724
6-06
6
Data Sheet ADF7021-N
Rev. B | Page 21 of 65
Figure 23. Wideband Interference Rejection
Figure 24. Digital RSSI Readback Linearity
Figure 25. Image Rejection, Uncalibrated vs. Calibrated
Figure 26. Variation of IF Filter Response with Temperature
(IF_FILTER_BW = 9 kHz, Temperature Range is −40°C to +90°C in 10° Steps)
Figure 27. 2FSK Sensitivity vs. Modulation Index vs. Correlator Discriminator Bandwidth
Figure 28. 3FSK Receiver Sensitivity Using Viterbi Detection and
Threshold Detection
FREQUENCY OFFSET (MHz)
BLO
CK
ING
(dB
)
–10
0
10
20
30
40
50
60
70
80
90
–22 –18 –14 –10 –6 –2 0 2 6 10 14 18 22
RF FREQ = 868MHz WANTED SIGNAL (10dB ABOVE SENSITIVITY POINT) = 2FSK, fDEV = 4kHz, DATA RATE = 9.8kbpsBLOCKER = 2FSK,
fDEV = 4kHz, DATA RATE = 9.8kbpsVDD = 3.0VTEMPERATURE = 25°C
0724
6-05
9
–140
–120
–100
–80
–60
–40
–20
–122.5 –112.5 –102.5 –92.5 –82.5 –72.5 –62.5 –52.5 –42.5
ACTUAL RF INPUT LEVEL
RF INPUT (dBm)
RSS
I LEV
EL (d
Bm
)
RSSIREADBACK LEVEL
0724
6-05
5
–10
0
10
20
30
40
50
60
70
429.80 429.85 429.90 429.95 430.00 430.05 430.10 430.15 430.20
RF FREQ = 430MHzEXTERNAL VCO INDUCTORDATA RATE = 9.6kbpsTEMPERATURE = 25°C, VDD = 3.0V
RF FREQUENCY (MHz)
BLO
CK
ING
(dB
)
CALIBRATED
UNCALIBRATED
0724
6-05
4
0072
46-0
91
2.50
–2.5–5.0–7.5
–10.0–12.5–15.0–17.5–20.0–22.5–25.0–27.5–30.0–32.5–35.0–37.5
90 92 94 96 98 100 102 104 106 108 110
ATT
ENU
ATI
ON
(dB
)
IF FREQUENCY (kHz)
–40°C
+90°C
MODULATION INDEX
SEN
SITI
VITY
PO
INT
(dB
m)
–118
–116
–114
–112
–110
–108
–106
–104
–102
–100
0 0.2 0.4 0.6 0.8 1.0 1.2
RF FREQ = 860MHz2FSK MODULATIONDATA RATE = 9.6kbpsIF BW = 25kHzVDD = 3.0VTEMPERATURE = 25°C
DISCRIMINATOR BANDWIDTH =1× FSK FREQUENCY DEVIATION
DISCRIMINATOR BANDWIDTH =2× FSK FREQUENCY DEVIATION
0724
6-05
8
–120 –118 –116 –114 –112 –110 –108 –106 –104 –102 –100
3FSK MODULATIONVDD = 3.0V, TEMP = 25°CDATA RATE = 9.6kbpsfDEV = 2.4kHzRF FREQ = 868MHzIF BW = 18.75kHz
INPUT POWER (dBm)
LOG
BER
–7
–6
–5
–4
–3
–2
–1
0
VITERBI DETECTION
THRESHOLD DETECTION
0724
6-06
2
ADF7021-N Data Sheet
Rev. B | Page 22 of 65
Figure 29. 4FSK Receiver Eye Diagram Measured Using the Test DAC Output
Figure 30. 3FSK Receiver Eye Diagram Measured Using the Test DAC Output
Figure 31. Receive Sensitivity vs. LNA/IF Filter Gain and Mixer Linearity Settings (The input IP3 at each setting is also shown)
REC
EIVE
R S
YMB
OL
LEVE
L
+1
+3
–1
–3
0
RF I/P LEVEL = –70dBmDATA RATE = 9.7kbpsfDEV (inner) = 1.2kHz
22452 ACQS M 50µs
IF BW = 25kHzPOST DEMOD BW = 12.4kHz
0724
6-06
4
+1
–1
0
4
20834 ACQS M 20µs C13 1.7V
RF I/P LEVEL = –70dBmDATA RATE = 10kbpsfDEV = 2.5kHz
IF BW = 12.5kHzPOST DEMOD BW = 12.4kHz
REC
EIVE
R S
YMB
OL
LEVE
L
0724
6-06
3
LNA GAIN, FILTER GAIN
SEN
SITI
VITY
(dB
m)
–130
–120
–110
–100
–90
–80
–70
3, 72(LOW GAIN MODE)
10, 72(MEDIUM GAIN MODE)
30, 72(HIGH GAIN MODE)
HIGH MIXERLINEARITY
DEFAULTMIXER
LINEARITY
MODULATION = 2FSKDATA RATE = 9.6kbpsfDEV = 4kHzIF BW = 12.5kHzDEMOD = CORRELATORSENSITIVITY @ 1E-3 BER
IP3 = –3dBm
IP3= –5dBm
IP3 = –9dBm
IP3 = –13.5dBm
IP3 = –24dBm
IP3 = –20dBm
0724
6-06
9
Data Sheet ADF7021-N
Rev. B | Page 23 of 65
FREQUENCY SYNTHESIZER REFERENCE INPUT The on-board crystal oscillator circuitry (see Figure 32) can use a quartz crystal as the PLL reference. Using a quartz crystal with a frequency tolerance of ≤10 ppm for narrow-band applications is recommended. It is possible to use a quartz crystal with >10 ppm tolerance, but to comply with the absolute frequency error specifications of narrow-band regulations (for example, ARIB STD-T67 and ETSI EN 300 220), compensation for the frequency error of the crystal is necessary.
The oscillator circuit is enabled by setting R1_DB12 high. It is enabled by default on power-up and is disabled by bringing CE low. Errors in the crystal can be corrected by using the automatic frequency control feature or by adjusting the fractional-N value (see the N Counter section).
Figure 32. Oscillator Circuit on the ADF7021-N
Two parallel resonant capacitors are required for oscillation at the correct frequency. Their values are dependent on the crystal specification. Choose them to ensure that the series value of capacitance added to the PCB track capacitance adds up to the specified load capacitance of the crystal, usually 12 pF to 20 pF. Track capacitance values vary from 2 pF to 5 pF, depending on board layout. When possible, choose capacitors that have a very low temperature coefficient to ensure stable frequency operation over all conditions.
Using a TCXO Reference
A single-ended reference (TCXO, VCXO, or OCXO) can also be used with the ADF7021-N. This is recommended for applications having absolute frequency accuracy requirements of <10 ppm, such as applications requiring compliance with ARIB STD-T67 or ETSI EN 300 220. The following are two options for interfacing the ADF7021-N to an external reference oscillator.
An oscillator with CMOS output levels can be applied to OSC2. Disable the internal oscillator circuit by setting R1_DB12 low.
An oscillator with 0.8 V p-p levels can be ac-coupled through a 22 pF capacitor into OSC1. Enable the internal oscillator circuit by setting R1_DB12 high.
Programmable Crystal Bias Current
Bias current in the oscillator circuit can be configured between 20 μA and 35 μA by writing to the XTAL_BIAS bits (R1_DB [13:14]). Increasing the bias current allows the crystal oscillator to power up faster.
CLKOUT Divider and Buffer
The CLKOUT circuit takes the reference clock signal from the oscillator section, shown in Figure 32, and supplies a divided-down, 50:50 mark-space signal to the CLKOUT pin. The CLKOUT signal is inverted with respect to the reference clock. An even divide from 2 to 30 is available. This divide number is set in R1_DB[7:10]. On power-up, the CLKOUT defaults to divide-by-8.
Figure 33. CLKOUT Stage
To disable CLKOUT, set the divide number to 0. The output buffer can drive up to a 20 pF load with a 10% rise time at 4.8 MHz. Faster edges can result in some spurious feedthrough to the output. A series resistor (1 kΩ) can be used to slow the clock edges to reduce these spurs at the CLKOUT frequency.
R Counter
The 3-bit R counter divides the reference input frequency by an integer between 1 and 7. The divided-down signal is presented as the reference clock to the phase frequency detector (PFD). The divide ratio is set in R1_DB[4:6]. Maximizing the PFD frequency reduces the N value. This reduces the noise multiplied at a rate of 20 log(N) to the output and reduces occurrences of spurious components.
Register 1 defaults to R = 1 on power-up.
PFD [Hz] = XTAL/R
Loop Filter
The loop filter integrates the current pulses from the charge pump to form a voltage that tunes the output of the VCO to the desired frequency. It also attenuates spurious levels generated by the PLL. A typical loop filter design is shown in Figure 34.
Figure 34. Typical Loop Filter Configuration
Design the loop so that the loop bandwidth (LBW) is approximately 100 kHz. This provides a good compromise between in-band phase noise and out-of-band spurious rejection. Widening the LBW excessively reduces the time spent jumping between frequencies, but it can cause insufficient spurious attenua-tion. Narrow-loop bandwidths can result in the loop taking long periods to attain lock and can also result in a higher level of power falling into the adjacent channel. Use the loop filter design on the EVAL-ADF7021-NDBxx for optimum performance.
OSC1
CP1CP2
OSC2
0724
6-08
3
DVDD
CLKOUTENABLE BIT
CLKOUTOSC1 DIVIDER1 TO 15
÷2
0724
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8
CHARGEPUMP OUT
VCO
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0
ADF7021-N Data Sheet
Rev. B | Page 24 of 65
The free design tool ADI SRD Design Studio™ can also be used to design loop filters for the ADF7021-N (see the ADI SRD Design Studio web site for details).
N Counter
The feedback divider in the ADF7021-N PLL consists of an 8-bit integer counter (R0_DB[19:26]) and a 15-bit, sigma-delta (Σ-Δ) fractional_N divider (R0_DB[4:18]). The integer counter is the standard pulse-swallow type that is common in PLLs. This sets the minimum integer divide value to 23. The fractional divide value provides very fine resolution at the output, where the output frequency of the PLL is calculated as
+×= 152
__
NFractionalNInteger
RXTALfOUT
When RF_DIVIDE_BY_2 (see the Voltage Controlled Oscillator (VCO) section) is selected, this formula becomes
+××= 152
_0.5 NFractionalInteger_NR
XTALfOUT
The combination of Integer_N (maximum = 255) and Fractional_N (maximum = 32,768/32,768) gives a maximum N divider of 255 + 1. Therefore, the minimum usable PFD is
[ ]( )1255
Hz+
=FrequencyOutputRequiredMaximum
PFDMIN
For example, when operating in the European 868 MHz to 870 MHz band, PFDMIN = 3.4 MHz.
Figure 35. Fractional_N PLL
Voltage Regulators
The ADF7021-N contains four regulators to supply stable voltages to the part. The nominal regulator voltage is 2.3 V. Regulator 1 requires a 3.9 Ω resistor and a 100 nF capacitor in series between CREG1 and GND, whereas the other regulators require a 100 nF capacitor connected between CREGx and GND. When CE is high, the regulators and other associated circuitry are powered on, drawing a total supply current of 2 mA. Bringing the CE pin low disables the regulators, reduces the supply current to less than 1 µA, and erases all values held in the registers.
The serial interface operates from a regulator supply. Therefore, to write to the part, the user must have CE high and the regulator
voltage must be stabilized. Regulator status (CREG4) can be monitored using the REGULATOR_READY signal from the MUXOUT pin.
MUXOUT The MUXOUT pin allows access to various digital points in the ADF7021-N. The state of MUXOUT is controlled in Register 0 (R0_DB[29:31]).
REGULATOR_READY
REGULATOR_READY is the default setting on MUXOUT after the transceiver is powered up. The power-up time of the regulator is typically 50 µs. Because the serial interface is powered from the regulator, the regulator must be at its nominal voltage before the ADF7021-N can be programmed. The status of the regulator can be monitored at MUXOUT. When the regulator ready signal on MUXOUT is high, programming of the ADF7021-N can begin.
Figure 36. MUXOUT Circuit
FILTER_CAL_COMPLETE
MUXOUT can be set to FILTER_CAL_COMPLETE. This signal goes low for the duration of both a coarse IF filter calibration and a fine IF filter calibration. It can be used as an interrupt to a microcontroller to signal the end of the IF filter calibration.
DIGITAL_LOCK_DETECT
DIGITAL_LOCK_DETECT indicates when the PLL has locked. The lock detect circuit is located at the PFD. When the phase error on five consecutive cycles is less than 15 ns, lock detect is set high. Lock detect remains high until a 25 ns phase error is detected at the PFD.
RSSI_READY
MUXOUT can be set to RSSI_READY. This indicates that the internal analog RSSI has settled and a digital RSSI readback can be performed.
Tx_Rx
Tx_Rx signifies whether the ADF7021-N is in transmit or receive mode. When in transmit mode, this signal is low. When in receive mode, this signal is high. It can be used to control an external Tx/Rx switch.
VCO
4\N
THIRD-ORDERΣ-Δ MODULATOR
PFD/CHARGE
PUMP
4\R
INTEGER_NFRACTIONAL_N
REFERENCE IN
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REGULATOR_READY (DEFAULT)
DIGITAL_LOCK_DETECT
RSSI_READY
Tx_Rx
LOGIC_ZERO
TRISTATE
MUX CONTROL
DGND
DVDD
MUXOUT
FILTER_CAL_COMPLETE
LOGIC_ONE
0724
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9
Data Sheet ADF7021-N
Rev. B | Page 25 of 65
VOLTAGE CONTROLLED OSCILLATOR (VCO) The ADF7021-N contains two VCO cores. The first VCO, the internal inductor VCO, uses an internal LC tank and supports 842 MHz to 916 MHz and 421 MHz to 458 MHz operating bands. The second VCO, the external inductor VCO, uses an external inductor as part of its LC tank and supports the RF operating band of 80 MHz to 650 MHz.
To minimize spurious emissions, both VCOs operate at twice the RF frequency. The VCO signal is then divided by 2 inside the synthesizer loop, giving the required frequency for the transmitter and the required local oscillator (LO) frequency for the receiver. A further divide-by-2 (RF_DIVIDE_BY_2) is performed outside the synthesizer loop to allow operation in the 421 MHz to 458 MHz band (internal inductor VCO) and the 80 MHz to 325 MHz band (external inductor VCO).
The VCO needs an external 22 nF capacitor between the CVCO pin and the regulator (CREG1 pin) to reduce internal noise.
Figure 37. Voltage Controlled Oscillator (VCO)
Internal Inductor VCO
To select the internal inductor VCO, set R1_DB25 to Logic 0, which is the default setting.
VCO bias current can be adjusted using R1_DB[19:22]. To ensure VCO oscillation, the minimum bias current setting under all conditions when using the internal inductor VCO is 0x8.
Recenter the VCO, depending on the required frequency of operation, by programming the VCO_ADJUST bits (R1_DB[23:24]). This is detailed in Table 9.
External Inductor VCO
When using the external inductor VCO, the center frequency of the VCO is set by the internal varactor capacitance and the combined inductance of the external chip inductor, bond wire, and PCB track. The external inductor is connected between the L2 and L1 pins.
A plot of the VCO operating frequency vs. total external inductance (chip inductor + PCB track) is shown in Figure 38.
Figure 38. Direct RF Output vs. Total External Inductance
The inductance for a PCB track using FR4 material is approxi-mately 0.57 nH/mm. Subtract this from the total value to determine the correct chip inductor value.
Typically, a particular inductor value allows the ADF7021-N to function over a range of ±6% of the RF operating frequency. When the RF_DIVIDE_BY_2 bit (R1_DB18) is selected, this range becomes ±3%. At 400 MHz, for example, an operating range of ±24 MHz (that is, 376 MHz to 424 MHz) with a single inductor (VCO range centered at 400 MHz) can be expected.
The VCO tuning voltage can be checked for a particular RF output frequency by measuring the voltage on the VCOIN pin when the part is fully powered up in transmit or receive mode.
The VCO tuning range is 0.2 V to 2 V. Choose the external inductor value to ensure that the VCO is operating as close as possible to the center of this tuning range. This is particularly important for RF frequencies <200 MHz, where the VCO gain is reduced and a tuning range of <±6 MHz exists.
The VCO operating frequency range can be adjusted by programming the VCO_ADJUST bits (R1_DB[23:24]). This typically allows the VCO operating range to be shifted up or down by a maximum of 1% of the RF frequency.
To select the external inductor VCO, set R1_DB25 to Logic 1. Set up the VCO_BIAS depending on the frequency of operation (as indicated in Table 9).
VCOLOOP FILTER
VCO_BIASR1_DB(19:22)
22nF
CVCO PIN
RF_DIVIDE_BY_2R1_DB18
÷2÷2
MUX
TON DIVIDER
TO PA
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0 5 10 15 20 25 30200
250
300
350
400
450
500
550
600
650
700
TOTAL EXTERNAL INDUCTANCE (nH)
FREQ
UEN
CY
(MH
z)
750
fMIN (MHz)
fMAX (MHz)
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ADF7021-N Data Sheet
Rev. B | Page 26 of 65
Table 9. RF Output Frequency Ranges for Internal and External Inductor VCOs and Required Register Settings
RF Frequency Output (MHz)
VCO to Be Used
RF Divide by 2
Register Settings VCO_INDUCTOR R1_DB25
RF_DIVIDE_BY_2 R1_DB18
VCO_ADJUST R1_DB[23:24]
VCO_BIAS R1_DB[19:22]
870 to 916 Internal L No 0 0 11 8 842 to 870 Internal L No 0 0 00 8 440 to 458 Internal L Yes 0 1 11 8 421 to 440 Internal L Yes 0 1 00 8 450 to 650 External L No 1 0 XX 4 200 to 450 External L No 1 0 XX 3 80 to 200 External L Yes 1 1 XX 2
CHOOSING CHANNELS FOR BEST SYSTEM PERFORMANCE An interaction between the RF VCO frequency and the reference frequency can lead to fractional spur creation. When the synthesizer is in fractional mode (that is, the RF VCO and reference frequencies are not integer related), spurs can appear on the VCO output spectrum at an offset frequency that corresponds to the difference frequency between an integer multiple of the reference and the VCO frequency.
These spurs are attenuated by the loop filter. They are more noticeable on channels close to integer multiples of the reference where the difference frequency may be inside the loop bandwidth; thus, the name integer boundary spurs. The occurrence of these spurs is rare because the integer frequencies are around multiples of the reference, which is typically >10 MHz. To avoid having very small or very large values in the fractional register, choose a suitable reference frequency.
Data Sheet ADF7021-N
Rev. B | Page 27 of 65
TRANSMITTER RF OUTPUT STAGE The power amplifier (PA) of the ADF7021-N is based on a single-ended, controlled current, open-drain amplifier that has been designed to deliver up to 13 dBm into a 50 Ω load at a maximum frequency of 950 MHz.
The PA output current and consequently, the output power, are programmable over a wide range. The PA configuration is shown in Figure 39. The output power is set using R2_DB[13:18].
Figure 39. PA Configuration
The PA is equipped with overvoltage protection, which makes it robust in severe mismatch conditions. Depending on the appli-cation, users can design a matching network for the PA to exhibit optimum efficiency at the desired radiated output power level for a wide range of antennas, such as loop or monopole antennas. See the LNA/PA Matching section for more information.
PA Ramping
When the PA is switched on or off quickly, its changing input impedance momentarily disturbs the VCO output frequency. This process is called VCO pulling, and it manifests as spectral splatter or spurs in the output spectrum around the desired carrier frequency. Some radio emissions regulations place limits on these PA transient-induced spurs (for example, the ETSI EN 300 220 regulations). By gradually ramping the PA on and off, PA transient spurs are minimized.
The ADF7021-N has built-in PA ramping configurability. As Figure 40 illustrates, there are eight ramp rate settings, defined as a certain number of PA setting codes per one data bit period. The PA steps through each of its 64 code levels but at different speeds for each setting. The ramp rate is set by configuring R2_DB[8:10].
If the PA is enabled/disabled by the PA_ENABLE bit (R2_DB7), it ramps up and down. If it is enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up and turns hard off.
Figure 40. PA Ramping Settings
PA Bias Currents
The PA_BIAS bits (R2_DB[11:12]) facilitate an adjustment of the PA bias current to further extend the output power control range, if necessary. If this feature is not required, the default value of 9 µA is recommended. If output power of greater than 10 dBm is required, a PA bias setting of 11 µA is recommended. The output stage is powered down by resetting R2_DB7.
MODULATION SCHEMES The ADF7021-N supports 2FSK, 3FSK, and 4FSK modulation. The implementation of these modulation schemes is shown in Figure 41.
Figure 41. Transmit Modulation Implementation
IDAC
2
6R2_DB(13:18)
R2_DB7
R2_DB(11:12)
+
RFGND
RFOUT
FROM VCO
R0_DB2707
246-
013
DATA BITS
PA RAMP 0(NO RAMP)
PA RAMP 1(256 CODES PER BIT)
PA RAMP 2(128 CODES PER BIT)
PA RAMP 3(64 CODES PER BIT)
PA RAMP 4(32 CODES PER BIT)
PA RAMP 5(16 CODES PER BIT)
PA RAMP 6(8 CODES PER BIT)
PA RAMP 7(4 CODES PER BIT)
1 2 3 4 ... 8 ... 16
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VCO
÷N
THIRD-ORDERΣ-Δ MODULATOR
PFD/CHARGE
PUMP
REF
INTEGER_NTx_FREQUENCY_DEVIATION
TOPA STAGE
1 – D2 PRSHAPING
4FSKBIT SYMBOL
MAPPER
MUX
TxDATA
2FSK
4FSK
GAUSSIANOR
RAISED COSINEFILTERING
PRE-CODER
3FSK
÷2LOOP FILTER
FRACTIONAL_N
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ADF7021-N Data Sheet
Rev. B | Page 28 of 65
Setting the Transmit Data Rate
In all modulation modes except oversampled 2FSK mode, an accurate clock is provided on the TxRxCLK pin to latch the data from the microcontroller into the transmit section at the required data rate. The exact frequency of this clock is defined by
32____ ××=
DIVIDECLKCDRDIVIDECLKDEMODXTALCLKDATA
where: XTAL is the crystal or TCXO frequency. DEMOD_CLK_DIVIDE is the divider that sets the demodulator clock rate (R3_DB[6:9]). CDR_CLK_DIVIDE is the divider that sets the CDR clock rate (R3_DB[10:17]).
Refer to the Register 3—Transmit/Receive Clock Register section for more programming information.
Setting the FSK Transmit Deviation Frequency
In all modulation modes, the deviation from the center frequency is set using the Tx_FREQUENCY_DEVIATION bits (R2_DB[19:27]).
The deviation from the center frequency in Hz is as follows:
For direct RF output,
162__]Hz[ DEVIATIONFREQUENCYTxPFDfDEV
×=
For RF_DIVIDE_BY_2 enabled,
162__5.0]Hz[ DEVIATIONFREQUENCYTxPFDfDEV
××=
where Tx_FREQUENCY_DEVIATION is a number from 1 to 511 (R2_DB[19:27]).
In 4FSK modulation, the four symbols (00, 01, 11, 10) are transmitted as ±3 × fDEV and ±1 × fDEV.
Binary Frequency Shift Keying (2FSK)
Two-level frequency shift keying is implemented by setting the N value for the center frequency and then toggling it with the TxDATA line. The deviation from the center frequency is set using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27].
2FSK is selected by setting the MODULATION_SCHEME bits (R2_DB[4:6]) to 000.
Minimum shift keying (MSK) or Gaussian minimum shift keying (GMSK) is supported by selecting 2FSK modulation and using a modulation index of 0.5. A modulation index of 0.5 is set up by configuring R2_DB[19:27] for an fDEV = 0.25 × transmit data rate.
3-Level Frequency Shift Keying (3FSK)
In 3-level FSK modulation (also known as modified duobinary FSK), the binary data (Logic 0 and Logic 1) is mapped onto three distinct frequencies: the carrier frequency (fC), the carrier frequency minus a deviation frequency (fC − fDEV), and the carrier frequency plus the deviation frequency (fC + fDEV).
A Logic 0 is mapped to the carrier frequency while a Logic 1 is either mapped onto the fC − fDEV frequency or the fC + fDEV frequency.
Figure 42. 3FSK Symbol-to-Frequency Mapping
Compared to 2FSK, this bits-to-frequency mapping results in a reduced transmission bandwidth because some energy is removed from the RF sidebands and transferred to the carrier frequency. At low modulation index, 3FSK improves the transmit spectral efficiency by up to 25% when compared to 2FSK.
Bit-to-symbol mapping for 3FSK is implemented using a linear convolutional encoder that also permits Viterbi detection to be used in the receiver. A block diagram of the transmit hardware used to realize this system is shown in Figure 43. The convolu-tional encoder polynomial used to implement the transmit spectral shaping is
P(D) = 1 − D2
where: P is the convolutional encoder polynomial. D is the unit delay operator.
A digital precoder with transfer function 1/P(D) implements an inverse modulo-2 operation of the 1 − D2 shaping filter in the transmitter.
Figure 43. 3FSK Encoding
fCfC – fDEV fC + fDEV
RF FREQUENCY
0+1–1
0724
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7
PRECODER1/P(D)
CONVOLUTIONALENCODER
P(D)
FSK MODCONTROL
ANDDATA FILTERING
Tx DATA0, 1
0, +1, –1
0, 1
TON DIVIDER
fCfC + fDEVfC – fDEV
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Data Sheet ADF7021-N
Rev. B | Page 29 of 65
The signal mapping of the input binary transmit data to the 3-level convolutional output is shown in Table 10. The convolutional encoder restricts the maximum number of sequential +1s or −1s to two and delivers an equal number of +1s and −1s to the FSK modulator, thus ensuring equal spectral energy in both RF sidebands.
Table 10. 3-Level Signal Mapping of the Convolutional Encoder TxDATA 1 0 1 1 0 0 1 0 0 1
Precoder Output 1 0 0 1 0 1 1 1 1 0
Encoder Output +1 0 −1 +1 0 0 +1 0 0 −1
Another property of this encoding scheme is that the transmitted symbol sequence is dc-free, which facilitates symbol detection and frequency measurement in the receiver. In addition, there is no code rate loss associated with this 3-level convolutional encoder; that is, the transmitted symbol rate is equal to the data rate presented at the transmit data input.
3FSK is selected by setting the MODULATION_SCHEME bits (R2_DB[4:6]) to 010. It can also be used with raised cosine filtering to further increase the spectral efficiency of the transmit signal.
4-Level Frequency Shift Keying (4FSK)
In 4FSK modulation, two bits per symbol spectral efficiency is realized by mapping consecutive input bit-pairs in the Tx data bit stream to one of four possible symbols (−3, −1, +1, +3). Thus, the transmitted symbol rate is half of the input bit rate.
By minimizing the separation between symbol frequencies, 4FSK can have high spectral efficiency. The bit-to-symbol mapping for 4FSK is gray coded and is shown in Figure 44.
Figure 44. 4FSK Bit-to-Symbol Mapping
The inner deviation frequencies (+fDEV and −fDEV) are set using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27]. The outer deviation frequencies are automatically set to three times the inner deviation frequency.
The transmit clock from Pin TxRxCLK is available after writing to Register 3 in the power-up sequence for receive mode. Clock the MSB of the first symbol into the ADF7021-N on the first transmit clock pulse from the ADF7021-N after writing to Register 3. Refer to Figure 6 for more timing information.
Oversampled 2FSK
In oversampled 2FSK, there is no data clock from the TxRxCLK pin. Instead, the transmit data at the TxRxDATA pin is sampled at 32 times the programmed rate.
This is the only modulation mode that can be used with the UART mode interface for data transmission (refer to the Interfacing to a Microcontroller/DSP section for more information).
SPECTRAL SHAPING Gaussian or raised cosine filtering can be used to improve transmit spectral efficiency. The ADF7021-N supports Gaussian filtering (bandwidth time [BT] = 0.5) on 2FSK modulation. Raised cosine filtering can be used with 2FSK, 3FSK, or 4FSK modulation. The roll-off factor (alpha) of the raised cosine filter has programmable options of 0.5 and 0.7. Both the Gaussian and raised cosine filters are implemented using linear phase digital filter architectures that deliver precise control over the BT and alpha filter parameters, and guarantee a transmit spectrum that is very stable over temperature and supply variation.
Gaussian Frequency Shift Keying (GFSK)
Gaussian frequency shift keying reduces the bandwidth occupied by the transmitted spectrum by digitally prefiltering the transmit data. The BT product of the Gaussian filter used is 0.5.
Gaussian filtering can only be used with 2FSK modulation. This is selected by setting R2_DB[4:6] to 001.
Raised Cosine Filtering
Raised cosine filtering provides digital prefiltering of the transmit data by using a raised cosine filter with a roll-off factor (alpha) of either 0.5 or 0.7. The alpha is set to 0.5 by default, but the raised cosine filter bandwidth can be increased to provide less aggressive data filtering by using an alpha of 0.7 (set R2_DB30 to Logic 1). Raised cosine filtering can be used with 2FSK, 3FSK, and 4FSK.
Raised cosine filtering is enabled by setting R2_DB[4:6] as outlined in Table 11.
Tx DATA
SYMBOLFREQUENCIES
f
t
+3fDEV
+fDEV
–fDEV
–3fDEV
0 0 0 1 1 0 1 1
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ADF7021-N Data Sheet
Rev. B | Page 30 of 65
MODULATION AND FILTERING OPTIONS The various modulation and data filtering options are described in Table 11.
Table 11. Modulation and Filtering Options Modulation Data Filtering R2_DB[4:6] BINARY FSK
2FSK None 000 MSK1 None 000 OQPSK with Half Sine Baseband Shaping2
None 000
GFSK Gaussian 001 GMSK3 Gaussian 001 RC2FSK Raised cosine 101 Oversampled 2FSK None 100
3-LEVEL FSK 3FSK None 010 RC3FSK Raised cosine 110
4-LEVEL FSK 4FSK None 011 RC4FSK Raised cosine 111
1 MSK is 2FSK modulation with a modulation index = 0.5. 2 Offset quadrature phase shift keying (OQPSK) with half sine baseband shaping
is spectrally equivalent to MSK. 3 GMSK is GFSK with a modulation index = 0.5.
TRANSMIT LATENCY Transmit latency is the delay time from the sampling of a bit/symbol by the TxRxCLK signal to when that bit/symbol appears at the RF output. The latency without any data filtering is one bit. The addition of data filtering adds a further latency as outlined in Table 12.
It is important that the ADF7021-N be left in transmit mode after the last data bit is sampled by the data clock to account for this latency. Maintain the ADF7021-N in transmit mode for a time equal to the number of latency bit periods for the applied modulation scheme. This ensures that all of the data sampled by the TxRxCLK signal appears at RF.
The figures for latency in Table 12 assume that the positive TxRxCLK edge is used to sample data (default). If the TxRxCLK is inverted by setting R2_DB[28:29], an additional 0.5 bit latency can be added to all values in Table 12.
Table 12. Bit/Symbol Latency in Transmit Mode for Various Modulation Schemes Modulation Latency 2FSK 1 bit
GFSK 4 bits RC2FSK, Alpha = 0.5 5 bits RC2FSK, Alpha = 0.7 4 bits
3FSK 1 bit RC3FSK, Alpha = 0.5 5 bits RC3FSK, Alpha = 0.7 4 bits
4FSK 1 symbol RC4FSK, Alpha = 0.5 5 symbols RC4FSK, Alpha = 0.7 4 symbols
TEST PATTERN GENERATOR The ADF7021-N has a number of built-in test pattern generators that can be used to facilitate radio link setup or RF measurement.
A full list of the supported patterns is shown in Table 13. The data rate for these test patterns is the programmed data rate set in Register 3.
The PN9 sequence is suitable for test modulation when carrying out adjacent channel power (ACP) or occupied bandwidth measurements.
Table 13. Transmit Test Pattern Generator Options Test Pattern R15_DB[8:10] Normal 000 Transmit Carrier 001 Transmit + fDEV Tone 010 Transmit − fDEV Tone 011 Transmit 1010 Pattern 100 Transmit PN9 Sequence 101 Transmit SWD Pattern Repeatedly 110
Data Sheet ADF7021-N
Rev. B | Page 31 of 65
RECEIVER SECTION RF FRONT END The ADF7021-N is based on a fully integrated, low IF receiver architecture. The low IF architecture facilitates a very low external component count and does not suffer from powerline-induced interference problems.
Figure 45 shows the structure of the receiver front end. The many programming options allow users to trade off sensitivity, linearity, and current consumption to best suit their application. To achieve a high level of resilience against spurious reception, the low noise amplifier (LNA) features a differential input. Switch SW2 shorts the LNA input when transmit mode is selected (R0_DB27 = 0). This feature facilitates the design of a combined LNA/PA matching network, avoiding the need for an external Rx/Tx switch. See the LNA/PA Matching section for details on the design of the matching network.
Figure 45. RF Front End
The LNA is followed by a quadrature downconversion mixer, which converts the RF signal to the IF frequency of 100 kHz. An important consideration is that the output frequency of the synthesizer must be programmed to a value 100 kHz below the center frequency of the received channel. The LNA has two basic operating modes: high gain/low noise mode and low gain/low power mode. To switch between these two modes, use the LNA_MODE bit (R9_DB25). The mixer is also configurable between a low current and an enhanced linearity mode using the MIXER_LINEARITY bit (R9_DB28).
Based on the specific sensitivity and linearity requirements of the application, it is recommended to adjust the LNA_MODE bit and MIXER_LINEARITY bit as outlined in Table 15.
The gain of the LNA is configured by the LNA_GAIN bits (R9_DB[20:21]) and can be set by either the user or the automatic gain control (AGC) logic.
IF FILTER IF Filter Settings
Out-of-band interference is rejected by means of a fifth-order Butterworth polyphase IF filter centered on a frequency of 100 kHz. The bandwidth of the IF filter can be programmed to 9 kHz, 13.5 kHz, or 18.5 kHz by R4_DB[30:31]; it is recommended to be chosen as a compromise between interference rejection and attenuation of the desired signal.
If the AGC loop is disabled, the gain of the IF filter can be set to one of three levels by using the FILTER_GAIN bits (R9_DB[22:23]). The filter gain is adjusted automatically if the AGC loop is enabled.
IF Filter Bandwidth and Center Frequency Calibration
To compensate for manufacturing tolerances, calibrate the IF filter after power-up to ensure that the bandwidth and center frequency are correct. Coarse and fine calibration schemes are provided to offer a choice between fast calibration (coarse calibration) and high filter centering accuracy (fine calibration). Coarse calibration is enabled by setting R5_DB4 high. Fine calibration is enabled by setting R6_DB4 high.
For details on when it is necessary to perform a filter calibration, and in what applications to use either a coarse calibration or fine calibration, refer to the IF Filter Bandwidth Calibration section.
RSSI/AGC The RSSI is implemented as a successive compression log amp following the baseband (BB) channel filtering. The log amp achieves ±3 dB log linearity. It also doubles as a limiter to convert the signal-to-digital levels for the FSK demodulator. The offset correction circuit uses the BBOS_CLK_DIVIDE bits (R3_DB[4:5]); set these bits between 1 MHz and 2 MHz. The RSSI level is converted for user readback and for digitally controlled AGC by an 80-level (7-bit) flash ADC. This level can be converted to input power in dBm. By default, the AGC is on when powered up in receive mode.
Figure 46. RSSI Block Diagram
RSSI Thresholds
When the RSSI is above AGC_HIGH_THRESHOLD (R9_DB[11:17]), the gain is reduced. When the RSSI is below AGC_LOW_THRESHOLD (R9_DB[4:10]), the gain is increased. The thresholds default to 30 and 70 on power-up in receive mode. A delay (set by AGC_CLK_DIVIDE, R3_DB[26:31]) is programmed to allow for settling of the loop. A value of 13 is recommended to give an AGC update rate of 7.7 kHz.
SW2 LNA
RFIN
RFINB
Tx/Rx SELECT(R0_DB27)
LNA_MODE(R9_DB25)
LNA_BIAS(R9_DB[26:27])
MIXER LINEARITY(R9_DB28)
LO
I (TO FILTER)
Q (TO FILTER)
LNA_GAIN(R9_DB[20:21])
LNA/MIXER_ENABLE(R8_DB6) 07
246-
017
1
IFWR IFWR IFWR IFWR
LATCHA A A
R
CLK
ADC
OFFSETCORRECTION
RSSI
FSKDEMOD
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ADF7021-N Data Sheet
Rev. B | Page 32 of 65
The user has the option of changing the two threshold values from the defaults of 30 and 70 (Register 9). The default AGC setup values are adequate for most applications. The threshold values must be more than 30 apart for the AGC to operate correctly.
Offset Correction Clock
In Register 3, set the BBOS_CLK_DIVIDE bits (R3_DB[4:5]) to give a baseband offset clock (BBOS CLK) frequency between 1 MHz and 2 MHz.
BBOS CLK [Hz] = XTAL/(BBOS_CLK_DIVIDE)
where BBOS_CLK_DIVIDE can be set to 4, 8, 16, or 32.
AGC Information and Timing
AGC is selected by default and operates by setting the appropriate LNA and filter gain settings for the measured RSSI level. It is possible to disable AGC by writing to Register 9 if the user wants to enter one of the modes listed in Table 15. The time for the AGC circuit to settle and, therefore, the time it takes to measure the RSSI accurately, is typically 390 µs. However, this depends on how many gain settings the AGC circuit has to cycle through. After each gain change, the AGC loop waits for a programmed time to allow transients to settle. This AGC update rate is set according to
AGC Update Rate [Hz] = DIVIDECLKAGC
DIVIDECLKSEQ__
[Hz]__
where: AGC_CLK_DIVIDE is set by R3_DB[26:31]. A value of 13 is recommended. SEQ_CLK_DIVIDE = 100 kHz (R3_DB[18:25]).
By using the recommended setting for AGC_CLK_DIVIDE, the total AGC settling time is
[Hz][sec]
RateUpdateAGCChangesGainAGCofNumber
TimeSettlingAGC =
The worst case for AGC settling occurs when the AGC control loop has to cycle through all five gain settings, which gives a maximum AGC settling time of 650 µs. RSSI Formula (Converting to dBm)
The RSSI formula is
Input Power [dBm] = −130 dBm + (Readback Code + Gain Mode Correction) × 0.5
where: Readback Code is given by Bit RV7 to Bit RV1 in the Register 7 readback register (see Figure 58 and the Readback Format section). Gain Mode Correction is given by the values in Table 14.
The LNA gain (LG2, LG1) and filter gain (FG2, FG1) values are also obtained from the readback register, as part of an RSSI readback.
Table 14. Gain Mode Correction LNA Gain (LG2, LG1)
Filter Gain (FG2, FG1)
Gain Mode Correction
H (1, 0) H (1, 0) 0 M (0, 1) H (1, 0) 24 M (0, 1) M (0, 1) 38 M (0, 1) L (0, 0) 58 L (0, 0) L (0, 0) 86 Introduce an additional factor to account for losses in the front-end-matching network/antenna.
Table 15. LNA/Mixer Modes
Receiver Mode LNA_MODE (R9_DB25)
LNA_GAIN (R9_DB[20:21])
MIXER_LINEARITY (R9_DB28)
Sensitivity (2FSK, DR = 4.8 kbps, fDEV = 4 kHz)
Rx Current Consumption (mA)
Input IP3 (dBm)
High Sensitivity Mode (Default)
0 +30 0 −118 +24.6 −24
Enhanced Linearity High Gain
0 +30 +1 −114.5 +24.6 −20
Medium Gain +1 +10 0 −112 +22.1 −13.5 Enhanced Linearity
Medium Gain +1 +10 +1 −105.5 +22.1 −9
Low Gain +1 +3 0 −100 +22.1 −5 Enhanced Linearity
Low Gain +1 +3 +1 −92.3 +22.1 −3
Data Sheet ADF7021-N
Rev. B | Page 33 of 65
DEMODULATION, DETECTION, AND CDR System Overview
An overview of the demodulation, detection, and clock and data recovery (CDR) of the received signal on the ADF7021-N is shown in Figure 47.
The quadrature outputs of the IF filter are first limited and then fed to either the correlator FSK demodulator or to the linear FSK demodulator. The correlator demodulator is used to demodulate 2FSK, 3FSK, and 4FSK. The linear demodulator is used for frequency measurement and is enabled when the AFC loop is active. The linear demodulator can also be used to demodulate 2FSK.
Following the demodulator, a digital post demodulator filter removes excess noise from the demodulator signal output. Threshold/slicer detection is used for data recovery of 2FSK and 4FSK. Data recovery of 3FSK can be implemented using either threshold detection or Viterbi detection.
An on-chip CDR PLL is used to resynchronize the received bit stream to a local clock. It outputs the retimed data and clock on the TxRxDATA and TxRxCLK pins, respectively.
Figure 47. Overview of Demodulation, Detection, and CDR Process
Correlator Demodulator
The correlator demodulator can be used for 2FSK, 3FSK, and 4FSK demodulation. Figure 48 shows the operation of the correlator demodulator for 2FSK.
Figure 48. 2FSK Correlator Demodulator Operation
The quadrature outputs of the IF filter are first limited and then fed to a digital frequency correlator that performs filtering and frequency discrimination of the 2FSK/3FSK/4FSK spectrum.
For 2FSK modulation, data is recovered by comparing the output levels from two correlators. The performance of this frequency discriminator approximates that of a matched filter detector, which is known to provide optimum detection in the presence of additive white Gaussian noise (AWGN). This method of FSK demodulation provides approximately 3 dB to 4 dB better sensitivity than a linear demodulator.
PO
ST
DE
MO
D F
ILT
ER
I
Q
LIMITERS
VITERBIDETECTION
MUX
CLOCKAND
DATARECOVERY
TxRxDATA
TxRxCLK
FREQUENCY CORRELATOR
LINEARDEMODULATOR
MUX
3FSK
THRESHOLDDETECTION
2/3/4FSK
0724
6-08
0
IF – fDEV IF + fDEV
I
IFQ
LIMITERS
R4_DB(10:19)
R4_DB7DOT_PRODUCT
FREQUENCY CORRELATOR
DISCRIMINATOR_BW
R4_DB9Rx_INVERT
DISCRIM BW
2FSK = +1, –13FSK = +1, 0, –1
4FSK = +3, +1, –1, –3
OUTPUT LEVELS:
0724
6-07
9
ADF7021-N Data Sheet
Rev. B | Page 34 of 65
Linear Demodulator
Figure 49 shows a block diagram of the linear demodulator.
Figure 49. Block Diagram of Linear FSK Demodulator
A digital frequency discriminator provides an output signal that is linearly proportional to the frequency of the limiter outputs. The discriminator output is filtered and averaged using a combined averaging filter and envelope detector. The demodulated 2FSK data from the post demodulator filter is recovered by slicing against the output of the envelope detector, as shown in Figure 49. This method of demodulation corrects for frequency errors between transmitter and receiver when the received spectrum is close to or within the IF bandwidth. This envelope detector output is also used for AFC readback and provides the frequency estimate for the AFC control loop.
Post Demodulator Filter
A second-order, digital low-pass filter removes excess noise from the demodulated bit stream at the output of the discriminator. The bandwidth of this post demodulator filter is programmable and must be optimized for the user’s data rate and received modulation type. If the bandwidth is set too narrow, performance degrades due to intersymbol interference (ISI). If the bandwidth is set too wide, excess noise degrades the performance of the receiver. The POST_DEMOD_BW bits (R4_DB[20:29]) set the bandwidth of this filter.
2FSK Bit Slicer/Threshold Detection
2FSK demodulation can be implemented using the correlator FSK demodulator or the linear FSK demodulator. In both cases, threshold detection is used for data recovery at the output of the post demodulation filter.
The output signal levels of the correlator demodulator are always centered about zero. Therefore, the slicer threshold level can be fixed at zero, and the demodulator performance is independent of the run-length constraints of the transmit data bit stream. This results in robust data recovery that does not suffer from the classic baseline wander problems that exist in the more traditional FSK demodulators.
When the linear demodulator is used for 2FSK demodulation, the output of the envelope detector is used as the slicer threshold, and this output tracks frequency errors that are within the IF filter bandwidth.
3FSK and 4FSK Threshold Detection
4FSK demodulation is implemented using the correlator demodulator followed by the post demodulator filter and threshold detection. The output of the post demodulation filter is a 4-level signal that represents the transmitted symbols (−3, −1, +1, +3). Threshold detection of 4FSK requires three threshold settings, one that is always fixed at 0 and two that are programmable and are symmetrically placed above and below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).
3FSK demodulation is implemented using the correlator demodu-lator, followed by a post demodulator filter. The output of the post demodulator filter is a 3-level signal that represents the transmitted symbols (−1, 0, +1). Data recovery of 3FSK can be implemented using threshold detection or Viterbi detection. Threshold detection is implemented using two thresholds that are programmable and are symmetrically placed above and below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).
3FSK Viterbi Detection
Viterbi detection of 3FSK operates on a four-state trellis and is implemented using two interleaved Viterbi detectors operating at half the symbol rate. The Viterbi detector is enabled by R13_DB11.
To facilitate different run length constraints in the transmitted bit stream, the Viterbi path memory length is programmable in steps of 4 bits, 6 bits, 8 bits, or 32 bits by setting the VITERBI_PATH_MEMORY bits (R13_DB[13:14]). Set this equal to or longer than the maximum number of consecutive 0s in the interleaved transmit bit stream.
When used with Viterbi detection, the receiver sensitivity for 3FSK is typically 3 dB greater than that obtained using threshold detection. When the Viterbi detector is enabled, however, the receiver bit latency is increased by twice the Viterbi path memory length.
Clock Recovery
An oversampled digital clock and data recovery (CDR) PLL is used to resynchronize the received bit stream to a local clock in all modulation modes. The oversampled clock rate of the PLL (CDR CLK) must be set at 32 times the symbol rate (see the Register 3—Transmit/Receive Clock Register section). The maximum data/symbol rate tolerance of the CDR PLL is determined by the number of zero-crossing symbol transitions in the transmitted packet. For example, if using 2FSK with a 101010 preamble, a maximum tolerance of ±3.0% of the data rate is achieved. However, this tolerance is reduced during recovery of the remainder of the packet where symbol transi-tions may not be guaranteed to occur at regular intervals. To maximize the data rate tolerance of the CDR, some form of encoding and/or data scrambling is recommended that guarantees a number of transitions at regular intervals.
PO
ST
_DE
MO
D_
FIL
TE
R
EN
VE
LO
PE
DE
TE
CT
OR SLICER
2FSK
FREQUENCY
IFLEVELI
Q
LIMITER
LINEARDISCRIMINATOR
R4_DB(20:29)FREQUENCYREADBACKAND AFC LOOP
+ 2FSK RxDATA
RxCLK
0724
6-07
3
Data Sheet ADF7021-N
Rev. B | Page 35 of 65
For example, using 2FSK with Manchester-encoded data achieves a data rate tolerance of ±2.0%.
The CDR PLL is designed for fast acquisition of the recovered symbols during preamble and typically achieves bit synchro-nization within 5-symbol transitions of preamble.
In 4FSK modulation, the tolerance using the +3, −3, +3, −3 preamble is ±3% of the symbol rate (or ±1.5% of the data rate). However, this tolerance is reduced during recovery of the remainder of the packet where symbol transitions may not be guaranteed to occur at regular intervals. To maximize the symbol/data rate tolerance, construct the remainder of the 4FSK packet so that the transmitted symbols retain close to dc-free properties by using data scrambling and/or by inserting specific dc balancing symbols that are inserted in the transmitted bit stream at regular intervals such as after every 8 or 16 symbols.
In 3FSK modulation, the linear convolutional encoder scheme guarantees that the transmitted symbol sequence is dc-free, facilitating symbol detection. However, Tx data scrambling is recommended to limit the run length of zero symbols in the transmit bit stream. Using 3FSK, the CDR data rate tolerance is typically ±0.5%.
RECEIVER SETUP Correlator Demodulator Setup
To enable the correlator for various modulation modes, refer to Table 16.
Table 16. Enabling the Correlator Demodulator Received Modulation DEMOD_SCHEME (R4_DB[4:6]) 2FSK 001 3FSK 010 4FSK 011
To optimize receiver sensitivity, the correlator bandwidth must be optimized for the specific deviation frequency and modulation used by the transmitter. The discriminator bandwidth is controlled by R4_DB[10:19] and is defined as
( )310400
_×
×=
KCLKDEMODBWTORDISCRIMINA
where: DEMOD CLK is as defined in the Register 3—Transmit/Receive Clock Register section. K is set for each modulation mode according to the following:
For 2FSK,
×=
DEVfRoundK
310100
For 3FSK,
××
=DEVf
RoundK2
10100 3
For 4FSK,
××
=DEV
FSK fRoundK
410100 3
4
where: Round is rounded to the nearest integer. Round4FSK is rounded to the nearest of the following integers: 32, 31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3. fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is the frequency deviation used for the ±1 symbols (that is, the inner frequency deviations).
To optimize the coefficients of the correlator, R4_DB7 and R4_DB[8:9] must also be assigned. The value of these bits depends on whether K is odd or even. These bits are assigned according to Table 17 and Table 18.
Table 17. Assignment of Correlator K Value for 2FSK and 3FSK K K/2 (K + 1)/2 R4_DB7 R4_DB[8:9] Even Even N/A 0 00 Even Odd N/A 0 10 Odd N/A Even 1 00 Odd N/A Odd 1 10
Table 18. Assignment of Correlator K Value for 4FSK K R4_DB7 R4_DB[8:9] Even 0 00 Odd 1 00
Linear Demodulator Setup
The linear demodulator can be used for 2FSK demodulation. To enable the linear demodulator, set the DEMOD_SCHEME bits (R4_DB[4:6]) to 000.
Post Demodulator Filter Setup
Set the 3 dB bandwidth of the post demodulator filter according to the received modulation type and data rate. The bandwidth is controlled by R4_DB[20:29] and is given by
CLKDEMODf
BWDEMODPOST CUTOFF××=
π2__
11
where fCUTOFF is the target 3 dB bandwidth in Hz of the post demodulator filter. Round up POST_DEMOD_BW to the nearest integer value.
Table 19. Post Demodulator Filter Bandwidth Settings for 2FSK/3FSK/4FSK Modulation Schemes Received Modulation
Post Demodulator Filter Bandwidth, fCUTOFF (Hz)
2FSK 0.75 × data rate 3FSK 1 × data rate 4FSK 1.6 × symbol rate (= 0.8 × data rate)
ADF7021-N Data Sheet
Rev. B | Page 36 of 65
3FSK Viterbi Detector Setup
The Viterbi detector can be used for 3FSK data detection. This is activated by setting R13_DB11 to Logic 1.
The Viterbi path memory length is programmable in steps of 4, 6, 8, or 32 bits (VITERBI_PATH_MEMORY, R13_DB[13:14]).
Set the path memory length equal to or greater than the max-imum number of consecutive 0s in the interleaved transmit bit stream.
The Viterbi detector also uses threshold levels to implement the maximum likelihood detection algorithm. These thresholds are programmable via the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).
These bits are assigned as follows:
3FSK/4FSK_SLICER_THRESHOLD =
×
×× 310100
75KDeviationrequencyTransmit F
where K is the value calculated for correlator discriminator bandwidth.
3FSK Threshold Detector Setup
To activate threshold detection of 3FSK, set R13_DB11 to Logic 0. Set the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]) as outlined in the 3FSK Viterbi Detector Setup section.
3FSK CDR Setup
In 3FSK, a transmit preamble of at least 40 bits of continuous 1s is recommended to ensure a maximum number of symbol transitions for the CDR to acquire lock.
The clock and data recovery for 3FSK requires a number of parameters in Register 13 to be set (see Table 20).
4FSK Threshold Detector Setup
The threshold for the 4FSK detector is set using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]). Set the threshold according to
3FSK/4FSK_SLICER_THRESHOLD =
×
×× 310100
87 KDeviationTxOuter4FSK
where K is the value calculated for correlator discriminator bandwidth.
Table 20. 3FSK CDR Settings Parameter Recommended Setting Purpose PHASE_CORRECTION (R13_DB12) 1 Phase correction is on 3FSK_CDR_THRESHOLD (R13_DB[15:21])
×
×× 310100
62KDeviationrequencyTransmit F
where K is the value calculated for correlator discriminator bandwidth.
Sets CDR decision threshold levels
3FSK_PREAMBLE_TIME_VALIDATE (R13_DB [22:25]) 15 Preamble detector time qualifier
Data Sheet ADF7021-N
Rev. B | Page 37 of 65
DEMODULATOR CONSIDERATIONS 2FSK Preamble
The recommended preamble bit pattern for 2FSK is a dc-free pattern (such as a 10101010… pattern). Preamble patterns with longer run-length constraints (such as 11001100…) can also be used but result in a longer synchronization time of the received bit stream in the receiver. The preamble needs to allow enough bits for AGC settling of the receiver and CDR acquisition. A minimum of 16 preamble bits is recommended when using the correlator demodulator and 48 bits when using the linear demod-ulator. When the receiver uses the internal AFC, the minimum recommended number of preamble bits is 64.
The remaining fields that follow the preamble header do not have to use dc-free coding. For these fields, the ADF7021-N can accommodate coding schemes with a run length of greater than eight bits without any performance degradation. Refer to Application Note AN-915 for more information.
4FSK Preamble and Data Coding
The recommended preamble bit pattern for 4FSK is a repeating 00100010… bit sequence. This 2-level sequence of repeating −3, +3, −3, +3 symbols is dc-free and maximizes the symbol timing performance and data recovery of the 4FSK preamble in the receiver. The minimum recommended length of the preamble is 32 bits (16 symbols).
Construct the remainder of the 4FSK packet so that the trans-mitted symbols retain close to a dc-free balance by using data scrambling and/or by inserting specific dc balancing symbols in the transmitted bit stream at regular intervals, such as after every 8 or 16 symbols.
Demodulator Tolerance to Frequency Errors Without AFC
The ADF7021-N has a number of options to combat frequency errors that exist due to mismatches between the transmit and receive crystals/TCXOs.
With AFC disabled, the correlator demodulator is tolerant to frequency errors over a ±0.3 × fDEV range, where fDEV is the FSK frequency deviation. For larger frequency errors, the frequency tolerance can be increased by adjusting the value of K and thus doubling the correlator bandwidth.
Calculate K as
××
=DEVf
RoundK2
10100 3
Recalculate the DISCRIMINATOR_BW setting in Register 4 using the new K value. Doubling the correlator bandwidth to improve frequency error tolerance in this manner typically results in a 1 dB to 2 dB loss in receiver sensitivity.
The linear demodulator (AFC disabled) tracks frequency errors in the receive signal when the receive signal is within the IF filter bandwidth. For example, for a receive signal with an occupied bandwidth = 9 kHz, using the 18.5 kHz IF filter bandwidth allows the linear demodulator to track the signal at an error of ±4.75 kHz with no increase in bit errors or loss in sensitivity.
Correlator Demodulator and Low Modulation Indices
The modulation index in 2FSK is defined as
RateDataf
IndexModulation DEV×=
2
The receiver sensitivity performance and receiver frequency tolerance can be maximized at low modulation index by increasing the discriminator bandwidth of the correlator demodulator. For modulation indices of less than 0.4, it is recommended to double the correlator bandwidth by calculating K as follows:
×
=DEVf
RoundK2
1003
Recalculate the DISCRIMINATOR_BW in Register 4 using the new K value. Figure 27 highlights the improved sensitivity that can be achieved for 2FSK modulation, at low modulation indices, by doubling the correlator bandwidth.
AFC OPERATION The ADF7021-N also supports a real-time AFC loop that is used to remove frequency errors due to mismatches between the transmit and receive crystals/TCXOs. The AFC loop uses the linear frequency discriminator block to estimate frequency errors. The linear FSK discriminator output is filtered and averaged to remove the FSK frequency modulation using a combined averaging filter and envelope detector. In receive mode, the output of the envelope detector provides an estimate of the average IF frequency.
Two methods of AFC supported on the ADF7021-N are external AFC and internal AFC.
External AFC
Here, the user reads back the frequency information through the ADF7021-N serial port and applies a frequency correction value to the fractional-N synthesizer-N divider.
The frequency information is obtained by reading the 16-bit signed AFC readback, as described in the Readback Format section, and by applying the following formula:
Frequency Readback [Hz] = (AFC READBACK × DEMOD CLK)/218
Although the AFC READBACK value is a signed number, under normal operating conditions, it is positive. In the absence of frequency errors, the frequency readback value is equal to the IF frequency of 100 kHz.
ADF7021-N Data Sheet
Rev. B | Page 38 of 65
Internal AFC
The ADF7021-N supports a real-time, internal, automatic frequency control loop. In this mode, an internal control loop automatically monitors the frequency error and adjusts the synthesizer-N divider using an internal proportional integral (PI) control loop.
The internal AFC control loop parameters are controlled in Register 10. The internal AFC loop is activated by setting R10_DB4 to 1. A scaling coefficient must also be entered, based on the crystal frequency in use. This is set up in R10_DB[5:16]; calculate it using
×=
XTALRoundFACTORSCALINGAFC
5002__
24
Maximum AFC Range
The maximum frequency correction range of the AFC loop is programmable on the ADF7021-N. This is set by R10_DB[24:31]. The maximum AFC correction range is the difference in frequency between the upper and lower limits of the AFC tuning range. For example, if the maximum AFC correction range is set to 10 kHz, the AFC can adjust the receiver LO within the fLO ± 5 kHz range.
However, when RF_DIVIDE_BY_2 (R1_DB18) is enabled, the programmed range is halved. Account for this halving by doubling the programmed maximum AFC range.
The recommended maximum AFC correction range is ≤1.5 × IF filter bandwidth. If the maximum frequency correction range is set to be >1.5 × IF filter bandwidth, the attenuation of the IF filter can degrade the AFC loop sensitivity.
The adjacent channel rejection (ACR) performance of the receivers can be degraded when AFC is enabled and the AFC correction range is close to the IF filter bandwidth. However, because the AFC correction range is programmable, the user can trade off correction range and ACR performance.
When AFC errors are removed using either the internal or external AFC, further improvement in receiver sensitivity can be obtained by reducing the IF filter bandwidth using the IF_FILTER_BW bits (R4_DB[30:31]).
AUTOMATIC SYNC WORD DETECTION (SWD) The ADF7021-N also supports automatic detection of the sync or ID fields. To activate this mode, the sync (or ID) word must be preprogrammed into the ADF7021-N. In receive mode, this preprogrammed word is compared to the received bit stream. When a valid match is identified, the external SWD pin is asserted by the ADF7021-N on the next Rx clock pulse.
This feature can be used to alert the microprocessor that a valid channel has been detected. It relaxes the computational requirements of the microprocessor and reduces the overall power consumption.
The SWD signal can also be used to frame the received packet by staying high for a preprogrammed number of bytes. The data packet length can be set in R12_DB[8:15].
The SWD pin status can be configured by setting R12_DB[6:7]. R11_DB[4:5] are used to set the length of the sync/ID word, which can be 12, 16, 20, or 24 bits long. A value of 24 bits is recommended to minimize false sync word detection in the receiver that can occur during recovery of the remainder of the packet or when a noise/no signal is present at the receiver input. The transmitter must transmit the sync byte MSB first and the LSB last to ensure proper alignment in the receiver sync-byte-detection hardware.
An error tolerance parameter can also be programmed that accepts a valid match when up to three bits of the word are incorrect. The error tolerance value is assigned in R11_DB[6:7].
Data Sheet ADF7021-N
Rev. B | Page 39 of 65
APPLICATIONS INFORMATION IF FILTER BANDWIDTH CALIBRATION Calibrate the IF filter on every power-up in receive mode to correct for errors in the bandwidth and filter center frequency due to process variations. The automatic calibration requires no external intervention once it is initiated by a write to Register 5. Depending on numerous factors, such as IF filter bandwidth, received signal bandwidth, and temperature variation, the user must determine whether to carry out a coarse calibration or a fine calibration.
The performance of both calibration methods is outlined in Table 21.
Table 21. IF Filter Calibration Specifications Filter Calibration Method
Center Frequency Accuracy1
Calibration Time (Typ)
Coarse Calibration 100 kHz ± 2.5 kHz 200 µs Fine Calibration 100 kHz ± 0.6 kHz 8.2 ms
1 After calibration.
Calibration Setup
IF Filter calibration is initiated by writing to Register 5 and setting the IF_CAL_COARSE bit (R5_DB4). This initiates a coarse filter calibration. If the IF_FINE_CAL bit (R6_DB4) has already been configured high, the coarse calibration is followed by a fine calibration, otherwise the calibration ends.
Once initiated by writing to the part, the calibration is performed automatically without any user intervention. Calibration time is 200 µs for coarse calibration and a few milliseconds for fine calibration, during which time the ADF7021-N must not be accessed. The IF filter calibration logic requires that the IF_FILTER_DIVIDER bits (R5_DB[5:13]) be set such that
kHz50__[Hz]
=DIVIDERFILTERIF
XTAL
The fine calibration uses two internally generated tones at certain offsets around the IF filter. The two tones are attenuated by the IF filter, and the level of this attenuation is measured using the RSSI. The filter center frequency is adjusted to allow equal attenuation of both tones. The attenuation of the two test tones is then remeasured. This continues for a maximum of 10 RSSI measurements, at which stage the calibration algorithm sets the IF filter center frequency to within 0.6 kHz of 100 kHz.
The frequency of these tones is set by the IF_CAL_LOWER_ TONE_DIVIDE (R6_DB[5:12]) and IF_CAL_UPPER_TONE_ DIVIDE (R6_DB[13:20]) bits, outlined in the following equations:
Lower Tone Frequency (kHz)
2VIDEER_TONE_DIIF_CAL_LOW
XTAL
×
Upper Tone Frequency (kHz)
2VIDEER_TONE_DIIF_CAL_UPP
XTAL
×
It is recommended to place the lower tone and upper tone as outlined in Table 22.
Table 22. IF Filter Fine Calibration Tone Frequencies IF Filter Bandwidth
Lower Tone Frequency
Upper Tone Frequency
9 kHz 78.1 kHz 116.3 kHz 13.5 kHz 79.4 kHz 116.3 kHz 18.5 kHz 78.1 kHz 119 kHz
Because the filter attenuation is slightly asymmetrical, it is necessary to have a small offset in the filter center frequency to give near equal rejection at the upper and lower adjacent channels. The calibration tones given in Table 22 give this small positive offset in the IF filter center frequency.
In some applications, an offset may not be required, and the user may wish to center the IF filter exactly at 100 kHz. In this case, the user can alter the tone frequencies from those given in Table 22 to adjust the fine calibration result.
The calibration algorithm adjusts the filter center frequency and measures the RSSI 10 times during the calibration. The time for an adjustment plus RSSI measurement is given by
IF Tone Calibration Time = SEQCLK
TIMEDWELLCALIF ___
It is recommended that the IF tone calibration time be at least 800 µs. The total time for the IF filter fine calibration is given by
IF Filter Fine Calibration Time = IF Tone Calibration Time × 10
When to Use Coarse Calibration
It is recommended to perform a coarse calibration on every receive mode power-up. This calibration typically takes 200 µs. The FILTER_CAL_COMPLETE signal from MUXOUT can be used to monitor the filter calibration duration or to signal the end of calibration. The ADF7021-N must not be accessed during calibration.
ADF7021-N Data Sheet
Rev. B | Page 40 of 65
When to Use a Fine Calibration
In cases where the receive signal bandwidth is very close to the bandwidth of the IF filter, it is recommended to perform a fine filter calibration every time the unit powers up in receive mode.
Perform a fine calibration if
OBW + Coarse Calibration Variation > IF_FILTER_BW
where: OBW is the 99% occupied bandwidth of the transmit signal. Coarse Calibration Variation is 2.5 kHz. IF_FILTER_BW is set by R4_DB[30:31].
The FILTER_CAL_COMPLETE signal from MUXOUT (set by R0_DB[29:31]) can be used to monitor the filter calibration duration or to signal the end of calibration. A coarse filter calibration is automatically performed prior to a fine filter calibration.
When a fine calibration is executed, the LNA is temporarily detached from the receive chain to ensure that external signals do not affect the calibration.
When to Use Single Fine Calibration
In applications where the receiver powers up numerous times in a short period, it is only necessary to perform a one-time fine calibration on the initial receiver power-up.
After the initial coarse calibration and fine calibration, the result of the fine calibration can be read back through the serial interface using the FILTER_CAL_READBACK result (refer to the Filter Bandwidth Calibration Readback section). On subsequent power-ups in receive mode, the filter is manually adjusted using the previous fine filter calibration result. This manual adjust is performed using the IF_FILTER_ADJUST bits (R5_DB[14:19]).
Only use this method if the successive power-ups in receive mode are over a short duration, during which time there is little variation in temperature (<15°C).
IF Filter Variation with Temperature
When calibrated, the filter center frequency can vary with changes in temperature. If the ADF7021-N is used in an application where it remains in receive mode for a considerable length of time, the user must consider this variation of filter center frequency with temperature. This variation is typically 1 kHz per 20°C, which means that if a coarse filter calibration and fine filter calibration are performed at 25°C, the initial maximum error is ±0.5 kHz, and the maximum possible change in the filter center frequency over temperature (−40°C to +85°C) is ±3.25 kHz. This gives a total error of ±3.75 kHz.
If the receive signal occupied bandwidth is considerably less than the IF filter bandwidth, the variation of filter center frequency over the operating temperature range may not be an issue. Alternatively, if the IF filter bandwidth is not wide enough to tolerate the variation with temperature, a periodic filter calibration can be performed or, alternatively, the on-chip temperature sensor can be used to determine when a filter cali-bration is necessary by monitoring for changes in temperature.
LNA/PA MATCHING The ADF7021-N exhibits optimum performance in terms of sensitivity, transmit power, and current consumption, only if its RF input and output ports are properly matched to the antenna impedance. For cost-sensitive applications, the ADF7021-N is equipped with an internal Rx/Tx switch that facilitates the use of a simple, combined passive PA/LNA matching network. Alternatively, an external Rx/Tx switch such as the ADG919 can be used, which yields a slightly improved receiver sensitivity and lower transmitter power consumption.
Internal Rx/Tx Switch
Figure 50 shows the ADF7021-N in a configuration where the internal Rx/Tx switch is used with a combined LNA/PA matching network. This is the configuration used on the EVAL-ADF7021-NDBxx evaluation board. For most applications, the slight performance degradation of 1 dB to 2 dB caused by the internal Rx/Tx switch is acceptable, allowing the user to take advantage of the cost saving potential of this solution. The design of the combined matching network must compensate for the reactance presented by the networks in the Tx and the Rx paths, taking the state of the Rx/Tx switch into consideration.
Figure 50. ADF7021-N with Internal Rx/Tx Switch
The procedure typically requires several iterations until an acceptable compromise has been reached. The successful imple-mentation of a combined LNA/PA matching network for the ADF7021-N is critically dependent on the availability of an accurate electrical model for the PCB. In this context, the use of a suitable CAD package is strongly recommended. To avoid this effort, a small form-factor reference design for the ADF7021-N is provided, including matching and harmonic filter components. The design is on a 2-layer PCB to minimize cost. Gerber files are available at www.analog.com.
PA
LNA
PA_OUT
RFIN
RFINB
VBAT
L1
ADF7021-N
OPTIONALBPF OR LPF
LA
CA
C1
CB
ZIN_RFIN
ZOPT_PA
ZIN_RFIN
ANTENNA
0724
6-02
2
Data Sheet ADF7021-N
Rev. B | Page 41 of 65
External Rx/Tx Switch
Figure 51 shows a configuration using an external Rx/Tx switch. This configuration allows an independent optimization of the matching and filter network in the transmit and receive path. Therefore, it is more flexible and less difficult to design than the configuration using the internal Rx/Tx switch. The PA is biased through Inductor L1, while C1 blocks dc current. Together, L1 and C1 form the matching network that transforms the source impedance into the optimum PA load impedance, ZOPT_PA.
Figure 51. ADF7021-N with External Rx/Tx Switch
ZOPT_PA depends on various factors, such as the required output power, the frequency range, the supply voltage range, and the temperature range. Selecting an appropriate ZOPT_PA helps to minimize the Tx current consumption in the application. Application Note AN-764 and Application Note AN-859 contain a number of ZOPT_PA values for representative conditions. Under certain conditions, however, it is recommended to obtain a suitable ZOPT_PA value by means of a load-pull measurement.
Due to the differential LNA input, the LNA matching network must be designed to provide both a single-ended-to-differential conversion and a complex, conjugate impedance match. The network with the lowest component count that can satisfy these requirements is the configuration shown in Figure 51, consisting of two capacitors and one inductor.
Depending on the antenna configuration, the user may need a harmonic filter at the PA output to satisfy the spurious emission requirement of the applicable government regulations. The harmonic filter can be implemented in various ways, for example, a discrete LC pi or T-stage filter. The immunity of the ADF7021-N to strong out-of-band interference can be improved by adding a band-pass filter in the Rx path. Alternatively, the ADF7021-N blocking performance can be improved by selecting one of the enhanced linearity modes, as described in Table 15.
IMAGE REJECTION CALIBRATION The image channel in the ADF7021-N is 200 kHz below the desired signal. The polyphase filter rejects this image with an asymmetric frequency response. The image rejection performance of the receiver is dependent on how well matched the I and Q signals are in amplitude and how well matched the quadrature is between them (that is, how close to 90° apart they are). The uncalibrated image rejection performance is approximately 29 dB (at 450 MHz). However, it is possible to improve on this performance by as much as 20 dB by finding the optimum I/Q gain and phase adjust settings.
Calibration Using Internal RF Source
With the LNA powered off, an on-chip generated, low level RF tone is applied to the mixer inputs. The LO is adjusted to make the tone fall at the image frequency where it is attenuated by the image rejection of the IF filter. The power level of this tone is then measured using the RSSI readback. The I/Q gain and phase adjust DACs (R5_DB[20:31]) are adjusted and the RSSI is remeasured. This process is repeated until the optimum values for the gain and phase adjust are found that provide the lowest RSSI readback level, thereby maximizing the image rejection performance of the receiver.
PA
LNA
PA_OUT
RFIN
RFINB
VBAT
L1
ADF7021-N
ADG919
OPTIONALBPF
(SAW)
OPTIONALLPF
LA
CA
C1
CB
ZIN_RFIN
ZOPT_PA
ZIN_RFIN
ANTENNA
Rx/Tx – SELECT
0724
6-02
1
ADF7021-N Data Sheet
Rev. B | Page 42 of 65
Figure 52. Image Rejection Calibration Using the Internal Calibration Source and a Microcontroller
Using the internal RF source, the RF frequencies that can be used for image calibration are programmable and are odd multiples of the reference frequency.
Calibration Using External RF Source
IR calibration can also be implemented using an external RF source. The IR calibration procedure is the same as that used for the internal RF source, except that an RF tone is applied to the LNA input.
Calibration Procedure and Setup
The IR calibration algorithm available from Analog Devices, Inc., is based on a low complexity, 2D optimization algorithm that can be implemented in an external microprocessor or microcontroller.
To enable the internal RF source, set the IR_CAL_SOURCE_ DRIVE_LEVEL bits (R6_DB[28:29]) to the maximum level. Set the LNA to its minimum gain setting, and disable the AGC if the internal source is being used. Alternatively, an external RF source can be used.
The magnitude of the phase adjust is set by using the IR_PHASE_ ADJUST_MAG bits (R5_DB[20:23]). This correction can be applied to either the I channel or Q channel, depending on the value of the IR_PHASE_ADJUST_DIRECTION bit (R5_DB24).
The magnitude of the I/Q gain is adjusted by the IR_GAIN_ ADJUST_MAG bits (R5_DB[25:29]). This correction can be applied to either the I or Q channel, depending on the value of
IR_GAIN_ADJUST_I/Q bit (R5_DB30), whereas the IR_GAIN_ADJUST_UP/DN bit (R5_DB31) sets whether the gain adjustment defines a gain or an attenuation adjust.
The calibration results are valid over changes in the ADF7021-N supply voltage. However, there is some variation with temperature. A typical plot of variation in image rejection over temperature after initial calibrations at −40°C, +25°C, and +85°C is shown in Figure 53. The internal temperature sensor on the ADF7021-N can be used to determine if a new IR calibration is required.
Figure 53. Image Rejection Variation with Temperature After Initial
Calibrations at −40°C, +25°C, and +85°C
INTERNALSIGNALSOURCE
MUX
RFIN
RFINB
LNA
ADF7021-N
POLYPHASEIF FILTER
PHASE ADJUST
GA
IN A
DJU
ST
I Q
FROM LO
GAIN ADJUSTREGISTER 5
PHASE ADJUSTREGISTER 5
SERIALINTERFACE
4
MICROCONTROLLER
4
RSSI/LOG AMP
7-BIT ADC
RSSI READBACK
I/Q GAIN/PHASE ADJUST ANDRSSI MEASUREMENT
ALGORITHM
0724
6-07
2
0
10
20
30
40
50
60
–60 –40 –20 0 20 40 60 80 100
VDD = 3.0VIF BW = 25kHz
WANTED SIGNAL:RF FREQ = 430MHzMODULATION = 2FSKDATA RATE = 9.6kbps, PRBS9fDEV = 4kHzLEVEL= –100dBm
INTERFERER SIGNAL:RF FREQ = 429.8MHzMODULATION = 2FSKDATA RATE = 9.6kbps, PRBS11fDEV = 4kHz
TEMPERATURE (°C)
IMA
GE
RE
JEC
TIO
N (
dB
)
CAL AT +25°C
CAL AT +85°CCAL AT –40°C
0724
6-06
7
Data Sheet ADF7021-N
Rev. B | Page 43 of 65
PACKET STRUCTURE AND CODING The suggested packet structure to use with the ADF7021-N is shown in Figure 54.
Figure 54. Typical Format of a Transmit Protocol
Refer to the Receiver Setup section for information on the required preamble structure and length for the various modulation schemes.
PROGRAMMING AFTER INITIAL POWER-UP Table 23 lists the minimum number of writes needed to set up the ADF7021-N in either Tx or Rx mode after CE is brought high. Additional registers can also be written to tailor the part
to a particular application, such as setting up sync byte detection or enabling AFC. When going from Tx to Rx or vice versa, the user needs to toggle the Tx/Rx bit and write only to Register 0 to alter the LO by 100 kHz.
Table 23. Minimum Register Writes Required for Tx/Rx Setup Mode Registers Tx Reg 1 Reg 3 Reg 0 Reg 2 Rx Reg 1 Reg 3 Reg 0 Reg 5 Reg 4 Tx to Rx and Rx to Tx Reg 0
The recommended programming sequences for transmit and receive are shown in Figure 55 and Figure 56, respectively. The difference in the power-up routine for a TCXO and XTAL reference is shown in these figures.
PREAMBLESYNCWORD
IDFIELD DATA FIELD CRC
0724
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3
ADF7021-N Data Sheet
Rev. B | Page 44 of 65
Figure 55. Power-Up Sequence for Transmit Mode
POWER-DOWNCE LOW
XTALREFERENCE
TCXOREFERENCE
CE HIGHWAIT 10µs (REGULATOR POWER-UP)
WRITE TO REGISTER 1 (TURNS ON VCO)WAIT 0.7ms (TYPICAL VCO SETTLING)
WRITE TO REGISTER 0 (TURNS ON PLL)WAIT 40µs (TYPICAL PLL SETTLING)
WRITE TO REGISTER 2 (TURNS ON PA)WAIT FOR PA TO RAMP UP (ONLY IF PA RAMP ENABLED)
WAIT FOR Tx LATENCY NUMBER OF BITS(REFER TO TABLE 12)
WRITE TO REGISTER 2 (TURNS OFF PA)WAIT FOR PA TO RAMP DOWN
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
CE HIGHWAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
CE LOWPOWER-DOWN
Tx MODE
OPTIONAL. ONLY NECESSARY IF PARAMP DOWN IS REQUIRED. 07
246-
086
Data Sheet ADF7021-N
Rev. B | Page 45 of 65
Figure 56. Power-Up Sequence for Receive Mode
POWER-DOWNCE LOW
WRITE TO REGISTER 5 (STARTS IF FILTER CALIBRATION)WAIT 0.2ms (COARSE CAL) OR WAIT 8.2ms
(COARSE CALIBRATION + FINE CALIBRATION)
WRITE TO REGISTER 11 (SET UP SWD)WRITE TO REGISTER 12 (ENABLE SWD)
WRITE TO REGISTER 6 (SETS UP IF FILTER CALIBRATION)
CE LOWPOWER-DOWN
Rx MODE
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
WRITE TO REGISTER 4 (TURNS ON DEMOD)
WRITE TO REGISTER 10 (TURNS ON AFC)
OPTIONAL.
WRITE TO REGISTER 0 (TURNS ON PLL)WAIT 40µs (TYPICAL PLL SETTLING)
CE HIGHWAIT 10µs (REGULATOR POWER-UP)
WRITE TO REGISTER 1 (TURNS ON VCO)WAIT 0.7ms (TYPICAL VCO SETTLING)
CE HIGHWAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
XTALREFERENCE
TCXOREFERENCE
OPTIONAL:ONLY NECESSARY IFAFC IS REQUIRED.
OPTIONAL:ONLY NECESSARY IFSWD IS REQUIRED.
OPTIONAL:ONLY NECESSARY IFIF FILTER FINE CAL IS REQUIRED.
0724
6-08
7
ADF7021-N Data Sheet
Rev. B | Page 46 of 65
APPLICATIONS CIRCUIT The ADF7021-N requires very few external components for operation. Figure 57 shows the recommended application circuit. Note that the power supply decoupling and regulator capacitors are omitted for clarity.
For recommended component values, refer to the ADF7021-N evaluation board data sheet and the AN-859 Application Note accessible from the ADF7021-N product page. Follow the reference design schematic closely to ensure optimum performance in narrow-band applications.
Figure 57. Typical Application Circuit (Regulator Capacitors and Power Supply Decoupling Not Shown)
48 47 46 45 44 43 42 41 40 39 38 37
ADF7021-N
VCOIN
CREG1
VDD1
RFOUT
RFGND
RFIN
RFINB
RLNA
VDD4
RSET
CREG4
GND4
CV
CO
GN
D1
L1
GN
D L2
VD
D
CP
OU
T
CR
EG
3
VD
D3
OS
C1
OS
C2
MU
XO
UT
1
2
3
4
5
6
7
8
9
10
11
12
13 14 15 16 17 18 19 20 21 22 23 24
MIX
_I
MIX
_I
MIX
_Q
MIX
_Q
FIL
T_I
FIL
T_I
GN
D4
FIL
T_Q
FIL
T_Q
GN
D4
TE
ST
_A
CE
CLKOUT
TxRxDATA
TxRxCLK
SWD
VDD2
CREG2
ADCIN
GND2
SCLK
SREAD
SDATA
SLE
36
35
34
33
32
31
30
29
28
27
26
25
VDD
VDD
VDD
TCXO
VDD
VDD
VDDANTENNA
CONNECTION
TOMICROCONTROLLERTx/Rx SIGNALINTERFACE
TOMICROCONTROLLERCONFIGURATIONINTERFACE
T-STAGE LCFILTER
MATCHING
LOOP FILTER
CVCOCAP
EXT VCO L*
REFERENCE
RSETRESISTOR
RLNARESISTOR
CHIP ENABLETO MICROCONTROLLER
*PIN 44 AND PIN 46 CAN BE LEFT FLOATING IF EXTERNAL INDUCTOR VCO IS NOT USED.
NOTES1. PINS [13:18], PINS [20:21], AND PIN 23 ARE TEST PINS AND ARE NOT USED IN NORMAL OPERATION. 07
246-
084
Data Sheet ADF7021-N
Rev. B | Page 47 of 65
SERIAL INTERFACE The serial interface allows the user to program the 16-/32-bit registers using a 3-wire interface (SCLK, SDATA, and SLE). It consists of a level shifter, 32-bit shift register, and 16 latches. Signals must be CMOS compatible. The serial interface is powered by the regulator, and, therefore, is inactive when CE is low.
Data is clocked into the register, MSB first, on the rising edge of each clock (SCLK). Data is transferred to one of 16 latches on the rising edge of SLE. The destination latch is determined by the value of the four control bits (C4 to C1); these are the bottom 4 LSBs, DB3 to DB0, as shown in Figure 2. Data can also be read back on the SREAD pin.
READBACK FORMAT The readback operation is initiated by writing a valid control word to the readback register and enabling the READBACK bit (R7_DB8 = 1). The readback can begin after the control word has been latched with the SLE signal. SLE must be kept high while the data is being read out. Each active edge at the SCLK pin successively clocks the readback word out at the SREAD pin, as shown in Figure 58, starting with the MSB first. The data appearing at the first clock cycle following the latch operation must be ignored. An extra clock cycle is needed after the 16th readback bit to return the SREAD pin to tristate. Therefore, 18 total clock cycles are needed for each read back. After the 18th clock cycle, bring the SLE low.
AFC Readback
The AFC readback is valid only during the reception of FSK signals with either the linear or correlator demodulator active. The AFC readback value is formatted as a signed 16-bit integer comprising Bit RV1 to Bit RV16 and is scaled according to the following formula:
FREQ RB [Hz] = (AFC_READBACK × DEMOD CLK)/218
In the absence of frequency errors, FREQ RB is equal to the IF frequency of 100 kHz. Note that, for the AFC readback to yield a valid result, the downconverted input signal must not fall outside the bandwidth of the analog IF filter. At low input signal levels, the variation in the readback value can be improved by averaging.
RSSI Readback
The format of the readback word is shown in Figure 58. It comprises the RSSI-level information (Bit RV1 to Bit RV7), the current filter gain (FG1, FG2), and the current LNA gain (LG1, LG2) setting. The filter and LNA gain are coded in accordance with the definitions in the Register 9—AGC Register section. For signal levels below −100 dBm, averaging the measured RSSI values improves accuracy. The input power can be calculated from the RSSI readback value as outlined in the RSSI/AGC section.
Figure 58. Readback Value Table
READBACK MODE
AFC READBACK
DB15
RV16
X
X
RV16
0
RSSI READBACK
BATTERY VOLTAGE/ADCIN/TEMP. SENSOR READBACK
SILICON REVISION
FILTER CAL READBACK
READBACK VALUE
DB14
RV15
X
X
RV15
0
DB13
RV14
X
X
RV14
0
DB12
RV13
X
X
RV13
0
DB11
RV12
X
X
RV12
0
DB10
RV11
LG2
X
RV11
0
DB9
RV10
LG1
X
RV10
0
DB8
RV9
FG2
X
RV9
0
DB7
RV8
FG1
X
RV8
RV8
DB6
RV7
RV7
RV7
RV7
RV7
DB5
RV6
RV6
RV6
RV6
RV6
DB4
RV5
RV5
RV5
RV5
RV5
DB3
RV4
RV4
RV4
RV4
RV4
DB2
RV3
RV3
RV3
RV3
RV3
DB1
RV2
RV2
RV2
RV2
RV2
DB0
RV1
RV1
RV1
RV1
RV107
246-
029
ADF7021-N Data Sheet
Rev. B | Page 48 of 65
Battery Voltage/ADCIN/Temperature Sensor Readback
The battery voltage is measured at Pin VDD4. The readback information is contained in Bit RV1 to Bit RV7. This also applies to the readback of the voltage at the ADCIN pin and the temperature sensor. From the readback information, the battery or ADCIN voltage can be determined using
VBATTERY = (BATTERY VOLTAGE READBACK)/21.1
VADCIN = (ADCIN VOLTAGE READBACK)/42.1
The temperature can be calculated using
Temp [°C] = 469.5 - (7.2 × TEMP_READBACK)
Silicon Revision Readback
The silicon revision readback word is valid without setting any other registers. The silicon revision word is coded with four quartets in BCD format. The product code (PC) is coded with three quartets extending from Bit RV5 to Bit RV16. The revision code (RC) is coded with one quartet extending from Bit RV1 to Bit RV4. The product code for the ADF7021-N reads back as PC = 0x211. The current revision code reads as RC = 0x1.
Filter Bandwidth Calibration Readback
The filter calibration readback word is contained in Bit RV1 to Bit RV8 (see Figure 58). This readback can be used for manual filter adjust, thereby avoiding the need to do an IF filter calibration in some instances. The manual adjust value is programmed by R5_DB[14:19]. To calculate the manual adjust based on a filter calibration readback, use the following formula:
IF_FILTER_ADJUST = FILTER_CAL_READBACK − 128
Program the result into R5_DB[14:19] as outlined in the Register 5—IF Filter Setup Register section.
Data Sheet ADF7021-N
Rev. B | Page 49 of 65
INTERFACING TO A MICROCONTROLLER/DSP Standard Transmit/Receive Data Interface
The standard transmit/receive signal and configuration interface to a microcontroller is shown in Figure 59. In transmit mode, the ADF7021-N provides the data clock on the TxRxCLK pin, and the TxRxDATA pin is used as the data input. The transmit data is clocked into the ADF7021-N on the rising edge of TxRxCLK.
Figure 59. ADuC845/ADuC847/ADuC848 to ADF7021-N Connection Diagram
In receive mode, the ADF7021-N provides the synchronized data clock on the TxRxCLK pin. The receive data is available on the TxRxDATA pin. Use the rising edge of TxRxCLK to clock the receive data into the microcontroller. Refer to Figure 4 and Figure 5 for the relevant timing diagrams.
In 4FSK transmit mode, the MSB of the transmit symbol is clocked into the ADF7021-N on the first rising edge of the data clock from the TxRxCLK pin. In 4FSK receive mode, the MSB of the first payload symbol is clocked out on the first negative edge of the data clock after the SWD and must be clocked into the microcontroller on the following rising edge. Refer to Figure 6 and Figure 7 for the relevant timing diagrams.
UART Mode
In UART mode, the TxRxCLK pin is configured to input transmit data in transmit mode. In receive mode, the receive data is available on the TxRxDATA pin, thus providing an asynchronous data interface. The UART mode can only be used with oversampled 2FSK. Figure 60 shows a possible interface to a microcontroller using the UART mode of the ADF7021-N. To enable this UART interface mode, set R0_DB28 high. Figure 8 and Figure 9 show the relevant timing diagrams for UART mode.
Figure 60. ADF7021-N (UART Mode) to
Asynchronous Microcontroller Interface
SPI Mode
In SPI mode, the TxRxCLK pin is configured to input transmit data in transmit mode. In receive mode, the receive data is available on the TxRxDATA pin. The data clock in both transmit and receive modes is available on the CLKOUT pin. In transmit mode, data is clocked into the ADF7021-N on the positive edge of CLKOUT. In receive mode, the TxRxDATA data pin are sampled by the microcontroller on the positive edge of the CLKOUT.
Figure 61. ADF7021-N (SPI Mode) to Microcontroller Interface
To enable SPI interface mode, set R0_DB28 high and set R15_DB[17:19] to 0x7. Figure 8 and Figure 9 show the relevant timing diagrams for SPI mode, while Figure 61 shows the recommended interface to a microcontroller using the SPI mode of the ADF7021-N.
ADSP-BF533 Interface
The suggested method of interfacing to the Blackfin® ADSP-BF533 is given in Figure 62.
Figure 62. ADSP-BF533 to ADF7021-N Connection Diagram
MISO
ADuC84x ADF7021-N
MOSISCLOCK
SSP3.7
P3.2/INT0P2.4P2.5
TxRxDATA
TxRxCLK
CESWDSREADSLE
P2.6P2.7
SDATASCLK
GPIO
0724
6-02
6
UART
ADF7021-NTxRxCLKTxRxDATA
TxDATARxDATA
CESWDSREADSLESDATASCLK
GPIO
MICROCONTROLLER
0724
6-08
5
SPI
ADF7021-NTxRxCLK
TxRxDATAMISO
MOSI
CESWDSREADSLESDATASCLK
GPIO
MICROCONTROLLER
SCLK CLKOUT
0724
6-07
6
MOSI
ADSP-BF533 ADF7021-N
MISOPF5
RSCLK1DT1PRIDR1PRI
RFS1PF6
SDATA
SLE
TxRxDATA
SWDCE
SCK SCLK
SREAD
TxRxCLK
0724
6-02
7
ADF7021-N Data Sheet
Rev. B | Page 50 of 65
REGISTER 0—N REGISTER
Figure 63. Register 0—N Register Map
The RF output frequency is calculated by the following:
For the direct output
152
__ NFractionalNIntegerPFDRFOUT
For the RF_DIVIDE_BY_2 (R1_DB18) selected
152
__5.0 NFractionalNIntegerPFDRFOUT
In UART/SPI mode, the TxRxCLK pin is used to input the Tx data. The Rx Data is available on the TxRxDATA pin. FILTER_CAL_ COMPLETE in the MUXOUT map in
Figure 63 indicates when a coarse or coarse plus fine IF filter calibration has finished. DIGITAL_ LOCK_DETECT indicates when the PLL has locked. RSSI_READY indicates that the RSSI signal has settled and an RSSI readback can be performed.
Tx_Rx gives the status of DB27 in this register, which can be used to control an external Tx/Rx switch.
TR1 Tx/Rx
0 TRANSMITRECEIVE1
M3 M2 M1 MUXOUT
0 REGULATOR_READY (DEFAULT)FILTER_CAL_COMPLETE0
0 DIGITAL_LOCK_DETECT0 RSSI_READY
1 Tx_Rx1 LOGIC_ZERO1 TRISTATE1
0011
0011
0101
0101 LOGIC_ONE
U1 UART_MODE
0 DISABLED1 ENABLED
N8 N7 N6 N5 N4 N3 N2 N1
0 230 24
.
.
.1 253
1 254
1
00
.
.
.1
1
1
00
.
.
.
.
.
.1
1
1
11
1
1
1
.
.
.
01
1
1
1
.
.
.
10
1
1
1
.
.
.
10
0
1
1
.
.
.
.
.
.
10
1
0
1 255
FRACTIONAL_NINTEGER_NTx/
Rx
UA
RT
_MO
DE
MUXOUTADDRESS
BITS
N5
N4
N8
M5
M6
M7
M8
M12
M13
M15N1
N2
N3
M14
M9
M10
M11
M4
M3
TR
1
U1
M1
M3
M2
C2
(0)
C1
(0)
C3
(0)
C4
(0)
M1
M2
N7
N6
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
FRACTIONAL_NDIVIDE RATIO
012...32764327653276632767
M15
000...1111
M14
000...1111
M13
000...1111
...
...
...
...
...
...
...
...
...
...
...
M3
000...1111
M2
001...0011
M1
010...0101
0724
6-03
0
INTEGER_NDIVIDE RATIO
Data Sheet ADF7021-N
Rev. B | Page 51 of 65
REGISTER 1—VCO/OSCILLATOR REGISTER
Figure 64. Register 1—VCO/Oscillator Register Map
• The R_COUNTER and XTAL_DOUBLER relationship is as follows:
If XTAL_DOUBLER = 0, PDF = COUNTERRXTAL
_
If XTAL_DOUBLER =1, PFD = COUNTERR
XTAL_
2×
• CLOCKOUT_DIVIDE is a divided-down and inverted version of the XTAL and is available on Pin 36 (CLKOUT).
• Set XOSC_ENABLE high when using an external crystal. If using an external oscillator (such as TCXO) with CMOS-level outputs into Pin OSC2, set XOSC_ENABLE low. If
using an external oscillator with a 0.8 V p-p clipped sine wave output into Pin OSC1, set XOSC_ENABLE high.
• Set the VCO_BIAS bits according to Table 9.
• The VCO_ADJUST bits adjust the center of the VCO operating band. Each bit typically adjusts the VCO band up by 1% of the RF operating frequency (0.5% if RF_DIVIDE_BY_2 is enabled).
• Setting VCO_INDUCTOR to external allows the use of the external inductor VCO, which gives RF operating frequencies of 80 MHz to 650 MHz. If the internal inductor VCO is being used for operation, set this bit low.
R3 R2 R100...1
12...7
10...1
01...1
X1 XOSC_ENABLE0 OFF1 ON
VA2 VA1VCO CENTERFREQ ADJUST
0 NOMINAL0 VCO ADJUST UP 11 VCO ADJUST UP 21
0101 VCO ADJUST UP 3
D1XTAL_DOUBLER
0 DISABLEENABLED1
CP2CP1RSET
ICP (mA)3.6kΩ
0 0 0.30 1 0.91 0 1.51 1 2.1
VB4 VB3 VB2 VB1VCO_BIASCURRENT
0 0.25mA0 0.5mA.1
10.1
01.1
00.1 3.75mA
CL4 CL3 CL2 CL1CLKOUT_DIVIDE RATIO
0 OFF00...1
010...1
24...
001...1
000...1 30
VCO_BIAS CP_
CU
RR
ENT
RF_
DIV
IDE_
B
Y_2
XOSC
_EN
AB
LE
VCO
_EN
AB
LE ADDRESSBITSXT
AL_
DO
UB
LER
XTAL_BIASVC
O_
AD
JUST
VA1
VB4
CL1
CL2
CL3
CL4
CP1
CP2
RFD
1
VB1
VB2
VB3
VE1
X1XB1
XB2
D1
R3
C2
(0)
C1
(1)
C3
(0)
C4
(0)
R1
R2
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
VA2
DB
24
DB
1
DB
0
DB
2
DB
3
XB2 XB1XTAL_BIAS
0 20µA0 25µA1 30µA1
0101 35µA
RFD1 RF_DIVIDE_BY_20 OFF
ON1
LOOPCONDITIONVCO OFFVCO ON
VE101
DB
25VC
L1 V
CO
_IN
DU
CTO
R
VCO_INDUCTORINTERNAL L VCOEXTERNAL L VCO
VCL101
R_COUNTERCLKOUT_DIVIDE
0724
6-03
1
RF R_COUNTERDIVIDE RATIO
ADF7021-N Data Sheet
Rev. B | Page 52 of 65
REGISTER 2—TRANSMIT MODULATION REGISTER
Figure 65. Register 2—Transmit Modulation Register Map
• The 2FSK/3FSK/4FSK frequency deviation is expressed by the following:
Direct output
Frequency Deviation [Hz] =
162__ PFDDEVIATIONFREQUENCYTx ×
With RF_DIVIDE_BY_2 (R1_DB18) enabled
Frequency Deviation [Hz] = 0.5 ×
162__ PFDDEVIATIONFREQUENCYTx ×
where Tx_FREQUENCY_DEVIATION is set by R2_DB[19:27] and PFD is the PFD frequency.
• In the case of 4FSK, there are tones at ±3 × the frequency deviation and at ±1 × the deviation.
• The power amplifier (PA) ramps at the programmed rate (R2_DB[8:10]) until it reaches its programmed level (R2_DB[13:18]). If the PA is enabled/disabled by the PA_ENABLE bit (R2_DB7), it ramps up and down. If it is enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up and turns hard off.
• R-COSINE_ALPHA sets the roll-off factor (alpha) of the raised cosine data filter to either 0.5 or 0.7. The alpha is set to 0.5 by default, but the raised cosine filter bandwidth can be increased to provide less aggressive data filtering by using an alpha of 0.7.
P60000..1
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.1
P20011..1
P10101..1
0 (PA OFF)1 (–16.0 dBm)23.. 63 (13 dBm)
TFD90000.1
TFD30000.1
...
...
...
...
...
...
...
TFD20011.1
TFD10101.1
0123.511
Tx_FREQUENCY_DEVIATION POWER_AMPLIFIERTxDATA_INVERT PA_BIAS PA_RAMP
MODULATION_SCHEME
ADDRESSBITSPA
_EN
AB
LE
PE101
PA_ENABLEDOFFON
PA20011
PA10101
PA_BIAS5µA7µA9µA11µA
DI20011
DI10101
TxDATA_INVERTNORMALINVERT CLKINVERT DATAINV CLK AND DATA
S300001111
S200110011
MODULATION_SCHEME2FSKGAUSSIAN 2FSK3FSK4FSKOVERSAMPLED 2FSKRAISED COSINE 2FSKRAISED COSINE 3FSKRAISED COSINE 4FSK
S101010101
PR300001111
PR200110011
NO RAMP256 CODES/BIT128 CODES/BIT64 CODES/BIT32 CODES/BIT16 CODES/BIT8 CODES/BIT4 CODES/BIT
PR1 PA_RAMP RATE01010101
TFD
5
TFD
4
TFD
8
PR1
PR2
PR3
PA1
P3P4P6TFD
1
TFD
2
TFD
3
P5 PA2
P1P2 PE1
S3TFD
9
DI1
DI2
C2
(1)
C1
(0)
C3
(0)
C4
(0)
S1S2TFD
7
TFD
6
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
R-C
OSI
NE_
ALP
HA
DB
30N
RC
1
NRC1
01
R-COSINE_ALPHA
0.70.5 (Default)
fDEV
POWER_AMPLIFIER
0724
6-03
2
Data Sheet ADF7021-N
Rev. B | Page 53 of 65
REGISTER 3—TRANSMIT/RECEIVE CLOCK REGISTER
Figure 66. Register 3—Transmit/Receive Clock Register Map
• Baseband offset clock frequency (BBOS CLK) must be greater than 1 MHz and less than 2 MHz, where
DIVIDECLKBBOSXTALCLKBBOS
__=
• Set the demodulator clock (DEMOD CLK) such that 2 MHz ≤ DEMOD CLK ≤ 15 MHz, where
DIVIDECLKDEMODXTALCLKDEMOD
__=
• For 2FSK/3FSK, the data/clock recovery frequency (CDR
CLK) needs to be within 2% of (32 × data rate). For 4FSK, the CDR CLK needs to be within 2% of (32 × symbol rate).
DIVIDECLKCDRCLKDEMOD
CLKCDR__
=
• The sequencer clock (SEQ CLK) supplies the clock to the digital receive block. It is recommended to be as close to 100 kHz as possible.
DIVIDECLKSEQXTALCLKSEQ
__=
• The time allowed for each AGC step to settle is determined by the AGC update rate. It is recommended to be set close to 8 kHz.
DIVIDECLKAGCCLKSEQ
RateUpdateAGC__
[Hz]=
FS800.11
FS700.11
FS300.11
...
...
...
...
...
...
FS201.11
FS110.01
CDR_CLK_ DIVIDE12.254255
BK20011
BK10101
BBOS_CLK_DIVIDE481632
SK800.11
SK700.11
SK300.11
...
...
...
...
...
...
SK201.11
SK110.01
SEQ_CLK_DIVIDE12.254255
OK200...1
OK101...1
DEMOD_CLK_DIVIDEINVALID1...15
SEQ_CLK_DIVIDEAGC_CLK_DIVIDE CDR_CLK_DIVIDE
BB
OS_
CLK
_D
IVID
E
DEMOD_CLK_DIVIDE
ADDRESSBITS
GD600...1
GD500...1
GD300...1
GD400...1
GD200...1
GD101...1
AGC_CLK_DIVIDEINVALID1...63
SK8
SK7
FS1
FS2
FS3
FS4
FS8
SK1
SK3
SK4
SK5
SK6
SK2
FS5
FS6
FS7
OK
2
OK
1
OK
4
OK
3
C2
(1)
C1
(1)
C3
(0)
C4
(0)
BK
1
BK
2
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
GD
6
GD
5
GD
1
GD
2
GD
3
GD
4
DB
24
DB
25
DB
28
DB
27
DB
26
DB
29
DB
30
DB
31
DB
1
DB
0
DB
2
DB
3
OK300...1
00...1
OK4
0724
6-03
3
ADF7021-N Data Sheet
Rev. B | Page 54 of 65
REGISTER 4—DEMODULATOR SETUP REGISTER
Figure 67. Register 4—Demodulator Setup Register Map
To solve for DISCRIMINATOR_BW, use the following equation:
DISCRIMINATOR_BW = 310400 KDEMODCLK
where the maximum value = 660.
For 2FSK,
DEVfRoundK
310100
For 3FSK,
DEVf
RoundK2
10100 3
For 4FSK,
DEV
FSK fRoundK
410100 3
4
where: Round is rounded to the nearest integer.
Round4FSK is rounded to the nearest of the following integers: 32, 31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3. fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is the frequency deviation used for the ±1 symbols (that is, the inner frequency deviations).
Rx_INVERT (R4_DB[8:9]) and DOT_PRODUCT (R4_DB7) need to be set as outlined in Table 17 and Table 18.
CLKDEMODf
_BWPOST_DEMOD CUTOFF
π211
where the cutoff frequency (fCUTOFF) of the post demodulator filter is typically be 0.75 × the data rate in 2FSK. Round up POST_DEMOD_BW to the nearest integer value. In 3FSK, set it equal to the data rate. While in 4FSK, set it equal to 1.6 × symbol rate.
DISCRIMINATOR_BW
DO
T_P
RO
DU
CT
POST_DEMOD_BWRx_
INVERT
IF_F
ILT
ER
_BW
ADDRESSBITS
TD
4
TD
3
RI1
RI2
DW
1
DW
2
DW
6
DW
7
DW
9
DW
10
TD
1
TD
2
DW
8
DW
3
DW
4
DW
5
DP
1
DS
3
C2(
0)
C1(
0)
C3(
1)
C4(
0)
DS
1
DS
2
TD
6
TD
5
TD
10
TD
9
TD
7
TD
8
IFB
2
IFB
1
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
DS1
01010101
DEMOD_SCHEME
2FSK LINEAR DEMODULATOR2FSK CORRELATOR DEMODULATOR3FSK DEMOD4FSK DEMODRESERVEDRESERVEDRESERVEDRESERVED
DP1
01
DOT_PRODUCT
CROSS_PRODUCTDOT_PRODUCTD
RI1
0101
Rx_INVERT
NORMALINVERT CLKINVERT DATAINVERT CLK/DATA
IFB1
0101
IF_FILTER _BW
DS2DS3
00110011
00001111
0011
RI2
IFB2
0011
DW3
00....1
DW1
10....1
POST_DEMOD_BW
12....1023
DW2
01....1
DW10
00....1
DW6
00....1
.
.
.
.
.
.
.
.
DW5
00....1
DW4
00....1
TD3
00....1
TD1
10....0
DISCRIMINATOR_BW
12....660
TD2
01....0
TD10
00....1
TD6
00....0
.
.
.
.
.
.
.
.
TD5
00....1
TD4
00....0
DEMOD_SCHEME
0724
6-03
4
9 kHz13.5 kHz18.5 kHzINVALID
Data Sheet ADF7021-N
Rev. B | Page 55 of 65
REGISTER 5—IF FILTER SETUP REGISTER
Figure 68. Register 5—IF Filter Setup Register Map
A coarse IF filter calibration is performed when the IF_CAL_COARSE bit (R5_DB4) is set. If the IF_FINE_ CAL bit (R6_DB4) has been previously set, a fine IF filter calibration is automatically performed after the coarse calibration.
Set IF_FILTER_DIVIDER such that
kHz50__
DIVIDERFILTERIF
XTAL
IF_FILTER_ADJUST allows the IF fine filter calibration result to be programmed directly on subsequent receiver power-ups, thereby saving on the need to redo a fine filter calibration in some instances. Refer to the Filter Bandwidth Calibration Readback section for information about using the IF_FILTER_ ADJUST bits.
R5_DB[20:31] are used for image rejection calibration. Refer to the Image Rejection Calibration section for details on how to program these parameters.
IR_PHASE_ADJUST_MAGIR
_PH
AS
E_
AD
JUS
T_D
IRE
CT
ION
IR G
AIN
_A
DJU
ST
_I/Q
IR_G
AIN
_A
DJU
ST
_UP
/DN
IR_GAIN_ADJUST_MAG IF_FILTER_DIVIDER
IF_C
AL
_CO
AR
SE
IF_FILTER_ADJUSTADDRESS
BITS
IFA
1
IFD
9
IFD
5
IFD
6
PM
2
PM
3
GM
1
GM
2
GM
4
GM
5
IFD
7
IFD
8
GM
3
PM
4
PD
1
IFD
4
IFD
3
C2
(0)
C1
(1)
C3
(1)
C4
(0)
IFD
1
CC
1
IFD
2
IFA
3
IFA
2
PM
1
IFA
6
IFA
4
IFA
5
GA
1
GQ
1
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
CC1 IF_CAL_COARSE
01
NO CALDO CAL
PD1
01
IR_PHASE_ADJUST_DIRECTION
ADJUST I CHADJUST Q CH
GA1
01
IR_GAIN_ADJUST_UP/DN
GAINATTENUATE
GQ1
01
IR_GAIN_ADJUST_I/Q
ADJUST I CHADJUST Q CH
IFD3
00....1
IFD1IF_FILTER_DIVIDER
10....1
12....511
IFD2
01....1
IFA2
001..1001.1
IFA6
000..011111
GM3IR_GAIN_
ADJUST_MAGGM5
000.1
IFD9
00....1
IFD6
00....1
.
.
.
.
.
.
.
.
IFD5
00....1
IFD4
00....1
010..1010.1
IFA1 IF_FILTER_ADJUST
...
...
...
...
...
...
...
...
...
...
...
0+1+2 ...+310–1–2...–31
GM4 GM2 GM1
000.1
000.1
001.1
010.1
012...31
PM2IR PHASEADJUSTPM3 PM1 PM1
000.1
000.1
001.1
010.1
012...15
IFA5
000..1000.1
0724
6-03
5
ADF7021-N Data Sheet
Rev. B | Page 56 of 65
REGISTER 6—IF FINE CAL SETUP REGISTER
Figure 69. Register 6—IF Fine Cal Setup Register Map
A fine IF filter calibration is set by enabling the IF_FINE_CAL Bit (R6_DB4). A fine calibration is then carried out only when Register 5 is written to and R5_DB4 is set.
Lower Tone Frequency (kHz) =
2VIDEER_TONE_DIIF_CAL_LOW
XTAL
Upper Tone Frequency (kHz) =
2VIDEER_TONE_DIIF_CAL_UPP
XTAL
It is recommended to place the lower tone and upper tone as outlined in Table 24.
Table 24. IF Filter Fine Calibration Tone Frequencies IF Filter Bandwidth
Lower Tone Frequency
Upper Tone Frequency
9 kHz 78.1 kHz 116.3 kHz 13.5 kHz 79.4 kHz 116.3 kHz 18.5 kHz 78.1 kHz 119 kHz
The IF tone calibration time is the amount of time that is spent at an IF calibration tone. It is dependent on the sequencer clock. For best practice, is recommended to have the IF tone calibration time be at least 800 μs.
IF Tone Calibration Time = CLKSEQ
TIMEDWELLCALIF ___
The total time for a fine IF filter calibration is
IF Tone Calibration Time × 10
R6_DB[28:30] control the internal source for the image rejection (IR) calibration. The IR_CAL_SOURCE_ DRIVE_LEVEL bits (R6_DB[28:29]) set the drive strength of the source, whereas the IR_CAL_SOURCE_÷2 bit (R6_DB30) allows the frequency of the internal signal source to be divided by 2.
IF_CAL_LOWER_TONE_DIVIDEIF_CAL_UPPER_TONE_DIVIDEIF_CAL_DWELL_TIME
IF_F
INE
_C
AL ADDRESS
BITS
CD
3
CD
2
CD
6
LT4
LT5
LT6
LT7
UT
3
UT
4
UT
6
UT
7
UT
8
CD
1
UT
5
LT8
UT
1
UT
2
LT3
LT2
CD
7
C2
(1)
C1
(0)
C3
(1)
C4
(0)
FC
1
LT1
CD
5
CD
4
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
25
DB
1
DB
0
DB
2
DB
3
FC1
01
IF_FINE_CAL
DISABLEDENABLED
UT3
000..
1
UT1
101..
1
IF_CAL_UPPER_TONE_DIVIDE
123..
UT2
011..
1
UT8
000..
0
...
...
...
...
...
...
... 127
LT3
000..
1
LT1
101..
1
123..
LT2
011..
1
LT8
000..
1
...
...
...
...
...
...
... 255
IF_CAL_LOWER_TONE_DIVIDE
CD3
000..
1
CD1
101..
1
IF_CAL_DWELL_TIME
123..
CD2
011..
1
CD7
000..
1
...
...
...
...
...
...
... 127
DB
28IR
C1
DB
29
DB
30
IRC
2
IRD
1
IR_
CA
L_
SO
UR
CE
_D
RIV
E_L
EV
EL
IR_
CA
L_
SO
UR
CE
÷2
IRC1
0101
IR_CAL_SOURCE_DRIVE_LEVELIRC2
0011
OFFLOWMEDHIGH
IRD1
01
IR_CAL_SOURCE ÷2
SOURCE ÷2 OFFSOURCE ÷2 ON
000..
1
LT7
000..
1
UT7
0724
6-03
6
Data Sheet ADF7021-N
Rev. B | Page 57 of 65
REGISTER 7—READBACK SETUP REGISTER
Figure 70. Register 7—Readback Setup Register Map
Readback of the measured RSSI value is valid only in Rx mode. Readback of the battery voltage, temperature sensor, or voltage at the external pin is not valid in Rx mode.
To read back the battery voltage, the temperature sensor, or the voltage at the external pin in Tx mode, first power up the ADC using R8_DB8 because it is turned off by default in Tx mode to save power.
For AFC readback, use the following equations (see the Readback Format section):
FREQ RB [Hz] = (AFC READBACK × DEMOD CLK)/218
VBATTERY = BATTERY VOLTAGE READBACK/21.1
VADCIN = ADCIN VOLTAGE READBACK/42.1
Temperature [°C] = 469.5 - (7.2 × TEMP_READBACK)
AD1AD2RB1RB2RB3
DB8 DB7 DB6 DB5 DB4 DB3 DB2
C2 (1) C1 (1)
CONTROLBITS
DB1 DB0
C3 (1)C4 (0)
READBACK_SELECT
ADC_MODE
AD2
0011
AD1
0101
ADC_MODE
MEASURE RSSIBATTERY VOLTAGETEMP SENSORTO EXTERNAL PIN
RB2
0011
RB1
0101
READBACK MODE
AFC WORDADC OUTPUTFILTER CALSILICON REV
RB3
01
READBACK_SELECT
DISABLEDENABLED
0724
6-03
7
ADF7021-N Data Sheet
Rev. B | Page 58 of 65
REGISTER 8—POWER-DOWN TEST REGISTER
Figure 71. Register 8—Power-Down Test Register Map
It is not necessary to write to this register under normal operating conditions.
For a combined LNA/PA matching network, always set R8_DB11 to 0, which enables the internal Tx/Rx switch. This is the power-up default condition.
PD1PD3PD4PD5
DB8 DB7 DB6 DB5 DB4 DB3 DB2
C2 (0) C1 (0)
CONTROLBITS
DB1 DB0
C3 (0)C4 (1)
LO
G_A
MP
_E
NA
BL
E
SY
NT
H_
EN
AB
LE
RE
SE
RV
ED
LN
A/M
IXE
R_
EN
AB
LE
FIL
TE
R_
EN
AB
LE
AD
C_
EN
AB
LE
DE
MO
D_
EN
AB
LE
Tx/
Rx_
SW
ITC
H_
EN
AB
LE
PA
_EN
AB
LE
_R
x_M
OD
E
CO
UN
TE
R_
RE
SE
T
Rx_RESET
CR1
DB15 DB14 DB13 DB12 DB11
LE1 PD6
DB10 DB9
SW1PD7
PD7
01
PA (Rx MODE)
PA OFFPA ON
CR1
01
COUNTER_RESET
NORMALRESET
DEMODRESET
CDRRESET
SW1
01
Tx/Rx SWITCH
DEFAULT (ON)OFF
PD6
01
DEMOD_ENABLE
DEMOD OFFDEMOD ON
PD5
01
ADC_ENABLE
ADC OFFADC ON
LE1
01
LOG_AMP_ENABLE
LOG AMP OFFLOG AMP ON
PD4
01
FILTER_ENABLE
FILTER OFFFILTER ON
PD3
01
LNA/MIXER_ENABLE
LNA/MIXER OFFLNA/MIXER ON
PD1
01
SYNTH_ENABLE
SYNTH OFFSYNTH ON
0724
6-03
8
Data Sheet ADF7021-N
Rev. B | Page 59 of 65
REGISTER 9—AGC REGISTER
Figure 72. Register 9—AGC Register Map
• It is necessary to program this register only if AGC settings, other than the defaults, are required.
• In receive mode, AGC is set to automatic AGC by default on power-up. The default thresholds are AGC_ LOW_ THRESHOLD = 30 and AGC_HIGH_ THRESHOLD = 70. See the RSSI/AGC section for details.
• AGC high and low settings must be more than 30 apart to ensure correct operation.
• An LNA gain of 30 is available only if LNA_MODE (R9_DB25) is set to 0.
AGC_HIGH_THRESHOLDLNA_GAIN
AGC_MODE
FILTER_GAINLN
A_
BIA
S
FILT
ER_
CU
RR
ENT
MIX
ER_
LIN
EAR
ITY
LNA_
MO
DE
AGC_LOW_THRESHOLDADDRESS
BITS
FG2
FG1
GL5
GL6
GL7
GH
1
GH
5
GH
6
GM
1
GM
2
LG1
LG2
GH
7
GH
2
GH
3
GH
4
GL4
GL3
C2
(0)
C1
(1)
C3
(0)
C4
(1)
GL1
GL2FI1
LG1
ML1 LI1
LI2
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
FI101
FILTER_CURRENTLOWHIGH
0123
AGC_MODEAUTO AGCMANUAL AGCFREEZE AGCRESERVED
FG20011
FG1 FILTER_GAIN0101
82472INVALID
LG20011
LG10101
LNA_GAIN31030INVALID
GL30001...111
GL11010...101
AGC_LOW_THRESHOLD1234...616263
GL20110...011
GL70000...111
GL60000...111
GL50000...111
GL40000...111
GH30001...110
GH11010...010
AGC_HIGH_THRESHOLD1234...787980
GH20110...110
GH70000...111
GH60000...000
GH50000...001
GH40000...110
LI20
LI10
LNA_BIAS800µA (DEFAULT)
LG101
LNA_MODEDEFAULTREDUCED GAIN
ML101
MIXER_LINEARITYDEFAULTHIGH
0724
6-03
9
ADF7021-N Data Sheet
Rev. B | Page 60 of 65
REGISTER 10—AFC REGISTER
Figure 73. Register 10—AFC Register Map
The AFC_SCALING_FACTOR can be expressed as
XTALRoundFACTORSCALINGAFC 5002__
24
The settings for KI and KP affect the AFC settling time and
AFC accuracy. The allowable range of each parameter is KI > 6 and KP < 7.
The recommended settings to give optimal AFC performance are KI = 11 and KP = 4. To trade off between AFC settling time and AFC accuracy, the KI and KP parameters can be adjusted from the recommended settings (staying within the allowable range) such that
AFC Correction Range = MAX_AFC_RANGE × 500 Hz
When the RF_DIVIDE_BY_2 (R1_DB18) is enabled, the programmed AFC correction range is halved. The user accounts for this halving by doubling the programmed MAX_AFC_RANGE value. For example, for a desired correction range of ±5 kHz, with RF_DIVIDE_BY_2 enabled, set MAX_AFC_RANGE (R10_DB[24:31]) equal to 20.
Signals that are within the AFC pull-in range but outside the IF filter bandwidth are attenuated by the IF filter. As a result, the signal can be below the sensitivity point of the receiver and, therefore, not detectable by the AFC.
KIKP AFC_SCALING_FACTORMAX_AFC_RANGE
AF
C_E
N
ADDRESSBITS
KP
3
KP
2
MA
3
M4
M5
M6
M7
M11
M12
KI2
KI3
KI4
KP
1
KI1
M8
M9
M10
M3
M2
MA
4
MA
5
C2
(1)
C1
(0)
C3
(0)
C4
(1)
AE
1
M1
MA
2
MA
6
MA
7
MA
8
MA
1
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
KP1
01.1
2^02^1...2^7
AE1
01
AFC_EN
OFFAFC ON
MA3
0001...111
MA1
1010...101
MAX_AFC_RANGE
1234...253254255
MA2
0110...011
MA8
0000...111
...
...
...
...
...
...
...
...
...
...
...
00.1
00.1
KP2KP3 KP KI1
01.1
2^02^1...2^15
00.1
00.1
KI2KI3 KIKI4
00.1
M3
0001...111
M1
1010...101
AFC_SCALING_FACTOR
1234...409340944095
M2
0110...011
M12
0000...111
...
...
...
...
...
...
...
...
...
...
...
0724
6-04
0
Data Sheet ADF7021-N
Rev. B | Page 61 of 65
REGISTER 11—SYNC WORD DETECT REGISTER
Figure 74. Register 11—Sync Word Detect Register Map
REGISTER 12—SWD/THRESHOLD SETUP REGISTER
Figure 75. Register 12—SWD/Threshold Setup Register Map
• Lock threshold locks the threshold of the envelope detector. This has the effect of locking the slicer in linear demodulation and locking the AFC and AGC loops when using linear or correlator demodulation.
PL20011
PL10101
SYNC_BYTE_LENGTH12 BITS16 BITS20 BITS24 BITS
MT20011
MT10101
MATCHING_TOLERANCEACCEPT 0 ERRORSACCEPT 1 ERRORACCEPT 2 ERRORSACCEPT 3 ERRORS
SYNC_BYTE_SEQUENCECONTROL
BITS
SYN
C_B
YTE_
LEN
GTH
MA
TCH
ING
_TO
LER
AN
CE
MT2
SB1
SB2
SB3
SB4
SB5
SB6
SB7
SB8
SB9
SB10
SB11
SB12
SB13
SB14
SB15
SB16
SB17
SB18
SB19
SB20
SB21
SB22
SB23
SB24
MT1
C2
(1)
C1
(1)
C3
(0)
C4
(1)
PL1
PL2
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
0724
6-04
1
DATA_PACKET_LENGTHCONTROL
BITSLOC
K_
THR
ESH
OLD
_M
OD
E
SWD
_MO
DE
IL2
IL1
C2
(0)
C1
(0)
C3
(1)
C4
(1)
LM1
LM2
DB
15
DB
14
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DP8
DP7
DP6
DP5
DP4
DP3
DP2
DP1
DB
7
DB
6
DB
5
DB
4
DB
1
DB
0
DB
2
DB
3
LOCK_THRESHOLD_MODE0 THRESHOLD FREE RUNNING1 LOCK THRESHOLD AFTER NEXT SYNCWORD2 LOCK THRESHOLD AFTER NEXT SYNCWORD FOR DATA PACKET LENGTH NUMBER OF BYTES3 LOCK THRESHOLD
DATA_PACKET_LENGTH0 INVALID1 1 BYTE... ...255 255 BYTES
SWD_MODE0 SWD PIN LOW1 SWD PIN HIGH AFTER NEXT SYNCWORD2 SWD PIN HIGH AFTER NEXT SYNCWORD FOR DATA PACKET LENGTH NUMBER OF BYTES3 INTERRUPT PIN HIGH
0724
6-04
2
ADF7021-N Data Sheet
Rev. B | Page 62 of 65
REGISTER 13—3FSK/4FSK DEMOD REGISTER Refer to the Receiver Setup section for information about programming these settings.
Figure 76. Register 13—3FSK/4FSK Demod Register Map
3FSK_CDR_THRESHOLD VITE
RB
I_PA
TH_
MEM
OR
Y
3FSK/4FSK_SLICER_THRESHOLD
CONTROLBITS
3FSK
_VIT
ERB
I_D
ETEC
TOR
PHA
SE_
CO
RR
ECTI
ON
ST4
ST5
ST6
ST7
VD1
PC1
VM1
VM2
VT1
VT2
VT3
VT4
VT5
VT6
VT7
ST3
C2
(0)
C1
(1)
C3
(1)
C4
(1)
ST1
ST2
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
1
DB
0
DB
2
DB
3
3FSK_PREAMBLE_TIME_VALIDATE
PTV1
PTV2
PTV3
PTV4
DB
24
DB
23
DB
22
DB
25
4 BITS0011
VM2VITERBI_PATH _MEMORYVM1
0101
6 BITS8 BITS32 BITS
PHASE_CORRECTION
0 DISABLED1 ENABLED
PC1
3FSK_VITERBI_DETECTOR
0 DISABLED1 ENABLED
VD1
ST3
000..1
ST1
101..1
SLICERTHRESHOLD
123..
ST2
011..1
ST7
000..1
...
...
...
...
...
...
... 127
0 00 ... 0 OFF
VT3
000..1
VT1
101..1
3FSK_CDR_THRESHOLD
123..
VT2
011..1
VT7
000..1
...
...
...
...
...
...
... 127
0 00 ... 0 OFF
PTV3
000..1
PTV1
101..1
3FSK_PREMABLE_TIME_VALIDATE
123..
PTV2
011..1
PTV4
000..1 15
0 00 0 0
0724
6-04
3
Data Sheet ADF7021-N
Rev. B | Page 63 of 65
REGISTER 14—TEST DAC REGISTER
Figure 77. Register 14—Test DAC Register Map
The demodulator tuning parameters, PULSE_EXTENSION, ED_LEAK_FACTOR, and ED_PEAK_RESPONSE, can be enabled only by setting R15_DB[4:7] to 0x9.
Using the Test DAC to Implement Analog FM DEMOD and Measuring SNR
For detailed information about using the test DAC, see Application Note AN-852.
The test DAC allows the post demodulator filter out for both linear and correlator demodulators to be viewed externally. The test DAC also takes the 16-bit filter output and converts it to a high frequency, single-bit output using a second-order, error feedback Σ-Δ converter. The output can be viewed on the SWD pin. This signal, when filtered appropriately, can then be used to do the following:
Monitor the signals at the FSK post demodulator filter output. This allows the demodulator output SNR to be measured. Eye diagrams of the received bit stream can also be constructed to measure the received signal quality.
Provide analog FM demodulation.
While the correlators and filters are clocked by DEMOD CLK, CDR CLK clocks the test DAC. Note that although the test DAC functions in regular user mode, the best performance is achieved when the CDR CLK is increased to or above the frequency of DEMOD CLK. The CDR block does not function when this condition exists.
Programming Register 14 enables the test DAC. Both the linear and correlator/demodulator outputs can be multiplexed into the DAC.
Register 14 allows a fixed offset term to be removed from the signal (to remove the IF component in the ddt case). It also has a signal gain term to allow the usage of the maximum dynamic range of the DAC.
TEST_DAC_GAIN TEST_DAC_OFFSET
TE
ST
_T
DA
C_E
N
ED
_PE
AK
_R
ES
PO
NS
E
ED
_LE
AK
_F
AC
TO
R
PU
LS
E_
EX
TE
NS
ION
ADDRESSBITS
TG
3
TG
2
ER
2
TO
4
TO
5
TO
6
TO
7
TO
11
TO
12
TO
14
TO
15
TO
16
TG
1
TO
13
TO
8
TO
9
TO
10
TO
3
TO
2
EF
1
EF
2
C2
(1)
C1
(0)
C3
(1)
C4
(1)
TE
1
TO
1
ER
1
EF
3
PE
1
PE
2
TG
4
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
ED_PEAK_RESPONSE
0123
FULL RESPONSE TO PEAK0.5 RESPONSE TO PEAK0.25 RESPONSE TO PEAK0.125 RESPONSE TO PEAK
TEST_DAC_GAIN
01...15
NO GAIN× 2^1...× 2^15
ED_LEAK_FACTOR
01234567
LEAKAGE =2^–82^–92^–102^–112^–122^–132^–142^–15
0724
6-04
4
PULSE_EXTENSION
0123
NO PULSE EXTENSIONEXTENDED BY 1EXTENDED BY 2EXTENDED BY 3
ADF7021-N Data Sheet
Rev. B | Page 64 of 65
REGISTER 15—TEST MODE REGISTER
Figure 78. Register 15—Test Mode Register Map
• Analog RSSI can be viewed on the Test_A pin by setting ANALOG_TEST_MODES to 11.
• Tx_TEST_MODES can be used to enable test modulation.
• The CDR block can be bypassed by setting Rx_TEST_ MODES to 4, 5, or 6, depending on the demodulator used.
Rx_TEST_MODES
Tx_TEST_MODES
Σ-Δ_TEST_MODES
PFD
/CP_
TEST
_M
OD
ES
PLL_TEST_MODES
ANALOG_TEST_MODES CLK_MUX
FOR
CE_
LDH
IGH
REG
1_P
D
CA
L_O
VER
RID
EADDRESS
BITS
PM4
PM3
AM
3
TM1
TM2
TM3
SD1
PC2
PC3
CM
2
CM
3
PM1
PM2
CM
1
SD2
SD3
PC1
RT4
RT3
AM
4
FH1
RD
1
CO
2
CO
1
C2
(1)
C1
(1)
C3
(1)
C4
(1)
RT1
RT2
AM
2
AM
1
DB
16
DB
15
DB
14
DB
17
DB
20
DB
19
DB
18
DB
21
DB
13
DB
12
DB
11
DB
10
DB
9
DB
8
DB
7
DB
6
DB
5
DB
4
DB
22
DB
23
DB
24
DB
26
DB
27
DB
28
DB
25
DB
1
DB
0
DB
2
DB
3
DB
29
DB
30
DB
31
ANALOG_TEST_MODES0 BAND GAP VOLTGE1 40µA CURRENT FROM REG42 FILTER I CHANNEL: STAGE 13 FILTER I CHANNEL: STAGE 24 FILTER I CHANNEL: STAGE 15 FILTER Q CHANNEL: STAGE 16 FILTER Q CHANNEL: STAGE 27 FILTER Q CHANNEL: STAGE 18 ADC REFERENCE VOLTAGE9 BIAS CURRENT FROM RSSI 5µA10 FILTER COARSE CAL OSCILLATOR O/P11 ANALOG RSSI I CHANNEL12 OSET LOOP +VE FBACK V (I CH)13 SUMMED O/P OF RSSI RECTIFIER+14 SUMMED O/P OF RSSI RECTIFIER–15 BIAS CURRENT FROM BB FILTER
Rx_TEST_MODES0 NORMAL1 SCLK, SDATA -> I, Q2 REVERSE I,Q3
LINEAR SLICER ON RXDATA
4CORRELATOR SLICER ON TxRxDATA
ADDITIONAL FILTERING ON I, Q
ENVELOPE DETECTOR WATCHDOG DISABLEDRESERVED
ENABLE REG 14 DEMOD PARAMETERS
PROHIBIT CALACTIVE
ENABLE DEMOD DURING CALFORCE CALACTIVE
56
I,Q TO TxRxCLK, TxRxDATA
789101112
POWER DOWN DDT AND ED IN T/4 MODE
131415
Tx_TEST_MODES0
Tx CARRIER ONLY1Tx +VE TONE ONLY2Tx –VE TONE ONLY3Tx "1010" PATTERN4Tx PN9 DATA, AT PROGRAMED RATE5Tx SYNC BYTE REPEATEDLY6
Σ-Δ_TEST_MODES0 DEFAULT, 3RD ORDER SD, NO DITHER1 1ST ORDER SD2 2ND ORDER SD3 DITHER TO FIRST STAGE4 DITHER TO SECOND STAGE5 DITHER TO THIRD STAGE6 DITHER × 87 DITHER × 32
0 NORMAPLL_TEST_MODES
L OPERATION1 R DIV2 N DIV3 RCNTR/2 ON MUXOUT4 NCNTR/2 ON MUXOUT5 ACNTR TO MUXOUT6 PFD PUMP UP TO MUXOUT7 PFD PUMP DN TO MUXOUT8 SDATA TO MUXOUT (OR SREAD?)9 ANALOG LOCK DETECT ON MUXOUT10 END OF COARSE CAL ON MUXOUT11 END OF FINE CAL ON MUXOUT12
13 TEST MUX SELECTS DATA14 LOCK DETECT PERCISION15 RESERVED
PFD/CP_TEST_MODES0 DEFAULT, NO BLEED1 (+VE) CONSTANT BLEED2 (–VE) CONSTANT BLEED3 (–VE) PULSED BLEED4 (–VE) PULSE BLD, DELAY UP?5 CP PUMP UP6 CP TRI-STATE7 CP PUMP DN
CLK MUXES ON CLKOUT PIN01234567
CAL_OVERRIDE0 AUTO CAL1 OVERRIDE GAIN2 OVERRIDE BW3 OVERRIDE BW AND GAIN
FORCE_LD_HIGH0 NORMAL1 FORCE
REG1_PD0 NORMAL1 PWR DWN
SDATA TO CDR
FORCE NEW PRESCALER CONFIG.FOR ALL N
NORMAL OPERATION
NORMAL, NO OUTPUTDEMOD CLKCDR CLKSEQ CLKBB OFFSET CLKSIGMA DELTA CLKADC CLKTxRxCLK
3FSK SLICER ON TxRxDATA
0724
6-04
5
Data Sheet ADF7021-N
Rev. B | Page 65 of 65
OUTLINE DIMENSIONS
Figure 79. 48-Lead Lead Frame Chip Scale Package [LFCSP]
7 mm × 7 mm Body and 0.75 mm Package Height (CP-48-5)
Dimensions shown in millimeters
ORDERING GUIDE Model1 Temperature Range Package Description Package Option ADF7021-NBCPZ −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP] CP-48-5 ADF7021-NBCPZ-RL −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP] CP-48-5 ADF7021-NBCPZ-RL7 −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP] CP-48-5 EVAL-ADF70XXMBZ2 Evaluation Platform Mother Board EVAL-ADF7021-NDB9Z 169 MHz Daughter Board EVAL-ADF7021-NDBIZ 421 MHz to 458 MHz Daughter Board EVAL-ADF7021-NDBEZ 421 MHz to 440 MHz Daughter Board EVAL-ADF7021-NDBZ2 860 MHz to 870 MHz Daughter Board EVAL-ADF7021-NDBZ5 Matching Unpopulated Daughter Board 1 Z = RoHS Compliant Part.
FOR PROPER CONNECTION OFTHE EXPOSED PAD, REFER TOTHE PIN CONFIGURATION ANDFUNCTION DESCRIPTIONSSECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-WKKD.
1
0.50BSC
BOTTOM VIEWTOP VIEW
PIN 1INDICATOR
7.00BSC SQ
48
132425
3637
12
EXPOSEDPAD
PIN 1INDICATOR
4.254.10 SQ3.95
0.450.400.35
SEATINGPLANE
0.800.750.70 0.05 MAX
0.02 NOM
0.20 MIN
0.20 REF
COPLANARITY0.08
0.300.230.18
08-1
6-20
10-B
©2008–2016 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D07246-0-3/16(B)