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IEEE JOURNAL OF SELECTED TOPICS IN QUANTUM ELECTRONICS, VOL. 13, NO. 1, JANUARY/FEBRUARY 2007 79

High-Speed Control of Lightwave Amplitude, Phase,and Frequency by Use of Electrooptic Effect

Tetsuya Kawanishi, Senior Member, IEEE, Takahide Sakamoto, Member, IEEE, and Masayuki Izutsu, Fellow, IEEE

(Invited Paper)

Abstract—High-speed control of lightwave using electrooptic(EO) effect is investigated in this paper. Agile optical frequency shiftcan be achieved by optical single-sideband (SSB) and frequency-shift-keying (FSK) modulators, where high-speed optical phase-shift-keying (PSK) signals can also be generated by using FSK/SSBmodulators. We also describe ultrahigh extinction ratio optical in-tensity modulation (IM) technique for two-tone lightwave signalswith high spurious suppression, which is useful for photonic mi-crowave and millimeter-wave generation. In addition, we investi-gated high-order optical sideband generation techniques: quadru-ple dual-sideband suppressed carrier (QDSB-SC) modulation andreciprocating optical modulation (ROM). Sub-tetrahertz signalscan be obtained from lightwaves with high-order sidebands.

Index Terms—Amplitude, frequency, microwave, millimeterwave, optical modulation, phase, sideband.

I. INTRODUCTION

O PTICAL modulators using electrooptic (EO) or electro-absorption (EA) effect play important roles in microwave

photonic applications as well as in high-speed optical com-munication systems. Intensity modulation (IM) and on–offkeying (OOK) are commonly used in commercial systems.However, recently, various types of modulation techniques,for example, differential phase-shift-keying (DPSK) [1],differential quadrature phase-shift-keying (DQPSK) [2]–[5],amplitude- and phase-shift-keying (APSK) [6], [7], frequency-shift-keying (FSK) [8]–[12], single-sideband (SSB) modulationtechniques [13]–[15], etc., were investigated to obtain enhancedspectral efficiency or receiver sensitivity in optical transmissionsystems. Orthogonal modulation techniques with OOK andFSK or OOK and DPSK are also attractive for optical labeling inoptical systems [10], [16], [17]. In this paper, we investigate EOmodulators, which can control the phase and frequency of light-wave, as well as the intensity. In a material having the EO effect,the refractive index (RI) depends on the voltage applied on thematerial, so that an optical phase modulator can be constructedwith an optical waveguide and an electrode on EO material.Various types of modulators, such as intensity modulator, FSK

Manuscript received August 1, 2006; revised September 5, 2006. This workwas supported in part by the New Energy and Industrial Technology De-velopment Organization of Japan under the Industrial Technology ResearchGrant Program in 2004 and in part by the Ministry of Education, Culture,Sports, Science and Technology, Japan, under Grant-in-Aid 17686032 for YoungScientists (A).

The authors are with the National Institute of Information and Commu-nications Technology, Tokyo 184-8795, Japan (e-mail: [email protected];[email protected]; [email protected]).

Digital Object Identifier 10.1109/JSTQE.2006.889044

modulator, can be built with optical phase modulators. A com-bination of an optical Mach–Zehnder (MZ) intensity modulatorand an optical phase modulator can generate an optical DQPSKsignal [2]; however, the time delay between the modulatorsshould be precisely controlled for stable operation. Integratedoptical devices for QPSK modulation are indispensable forhigh-performance and cost-effective DQPSK systems.

EO modulation is a key technology in various micro/millimeter-wave systems, such as pico-cellular mobile networks[18], photonic local oscillators (LOs) for millimeter-wave radioastronomy [19], etc. A two-tone lightwave signal consisting oftwo phase-locked optical spectral components can be used forphotonic LOs, where the two-tone signal is converted into aradio frequency (RF) signal by using a high-speed photodetec-tor. Optical phase-locked loop can be used for this technique;however, complicated feedback systems are needed to obtainstable operation. However, EO modulation can easily generatephase-locked spectral components, where stable upper sideband(USB) and lower sideband (LSB) components are obtained byfeeding sinusoidal RF signals. Double-sideband suppressed car-rier (DSB-SC) optical modulation is one of the ideal techniquesfor two-tone lightwave generation. The frequency separationbetween the two spectral components is precisely equal to dou-ble the modulating signal frequency. However, the modulationfrequency is limited by the frequency response of the modulator.A typical 3-dB bandwidth of a modulator is 30 GHz, so that thefrequency upper limit of the two-tone signal generated by DSB-SC modulation cannot be greater than 100 GHz. Recently, weproposed novel optical frequency multiplication schemes forsub-tetrahertz signal generation: quadruple DSB-SC (QDSB-SC) modulation [19], [20] and reciprocating optical modulation(ROM) [21]–[26]. QDSB-SC modulation whose setup consistsof two optical intensity modulators can achieve quadruple fre-quency multiplication with large spurious harmonic suppres-sion. An ROM, consisting of a pair of optical filters and anoptical phase modulator, can generate high-order sideband com-ponents stably and effectively, where one of the optical filters isplaced at the optical input port (input filter), and the other is atthe output port (output filter) [21], [22]. In ROMs, some of thesideband components are fed to the optical modulator again forgeneration of specific sideband components, where the desiredsideband components are taken out from the output filter. Thispaper is organized as follows. In Section II, the principle of op-tical phase modulation using EO effect is described briefly. InSection III, mathematical expressions for intensity modulatorsusing MZ structures are given, where ultrahigh extinction ratio

1077-260X/$25.00 © 2007 IEEE

80 IEEE JOURNAL OF SELECTED TOPICS IN QUANTUM ELECTRONICS, VOL. 13, NO. 1, JANUARY/FEBRUARY 2007

Fig. 1. Optical phase modulator using EO effect, where an electrode is placedalong an optical waveguide.

IM technique is also described. Optical FSK and SSB modula-tors can control optical frequency and phase. Optical frequencyshift, FSK and QPSK are described in Section IV. High-speedoptical FSK and QPSK modulations were demonstrated by us-ing integrated lithium niobate (LiNbO3) modulators. Section Vgives novel techniques for high-order sideband generation,where lightwaves with sub-tetrahertz signals can be obtained.

II. PHASE MODULATOR

Electrical control of optical phase can be achieved by usingEO materials, such as lithium niobate (LN), lithium tantalate(LT), gallium arsenide (GaAs), etc. Fig. 1 shows a schematicof an optical modulator using EO effect. An electrode is placedalong with an optical waveguide. By applying electric voltageon the electrode, we can change the RI in the waveguide, so thatthe phase of the lightwave can be controlled. For simplicity, theinput lightwave is assumed to be monochromatic, and can bedescribed by e2πif0t. The output lightwave R can be expressedby

R = ALWe2πif0t+if(t) (1)

f(t) ≡ KV (t) (2)

where V (t) and K, respectively, denote the electric voltage onthe electrode and the coupling coefficient between the electricsignal and the lightwave signal. ALW is the optical transmittancein the waveguide. When V (t) has high-frequency components,the frequency response of K should be taken into account. Thecoefficient K can be assumed to be a constant if the wavelengthsof the high-frequency components on the electrode are muchlonger than the length of the electrode. The response of EOmaterials is much faster than the frequency response of K,which is dominated by that of the electrode and by the phasemismatch between the electric and lightwave signals [27]–[29].The halfwave voltage defined by

Vπ ≡ π

K(3)

is the voltage for π (180◦) optical phase shift.Here, we consider that f(t) is a sinusoidal signal described

by ARF sin 2πfmt. The optical output can be expressed by

R = ALW exp i[2πf0t + ARF sin 2πfmt]

= ALW∞∑

n=−∞Jn(ARF) exp 2πi[f0t + nfmt] (4)

where Jn(A) is the first-kind nth-order Bessel function. ARF

is an index for optical phase deviation induced by the elec-tric signal, so that it is called induced phase. When f(t) =

Fig. 2. The first-kind Bessel functions. Amplitude of the nth order sidebandis proportional to the first-kind nth-order function. Solid, dashed, dotted, andone-dot-chain lines are for J0, J1, J2, and J3, respectively.

Fig. 3. Schematic of an MZ structure consisting of two optical phase modula-tors (A and B). Each optical phase modulator has an electrode for optical phasecontrol.

ARF cos 2πfmt, the output is expressed by

R = ALW exp i[2πf0t + ARF cos 2πfmt]

= ALW∞∑

n=−∞inJn(ARF) exp 2πi[f0t + nfmt]. (5)

The spectral component of e2πi(f0+nfm )t, the nth-order side-band, can be expressed by the nth-order Bessel function, whichare shown in Fig. 2. The first-order function J1(A) has themaximum point that can be defined by

J ′1(Am) = 0 (6)

J ′1(A) ≡ dJ1(A)

dA, 0 ≤ A ≤ π (7)

where J1(Am) = 0.583, Am = 1.841. In most applications ofoptical modulation techniques, the desired component is thefirst-order sidebands, so that the induced phase A should be lessthan Am to achieve effective modulation without generation ofundesired high-order sideband components.

III. MZ INTENSITY MODULATOR

A. Basic Principle

MZ structures are commonly used to achieve IM by EO ef-fects. MZ structures for IM are called MZ modulators (MZMs).As shown in Fig. 3, the MZM consists of two optical phasemodulators. We can control the output optical intensity by ap-plying a pair of electric signals to the two phase modulators.Balanced push–pull scheme is ideal for IM, where the induced

KAWANISHI et al.: HIGH-SPEED CONTROL OF LIGHTWAVE AMPLITUDE, PHASE, AND FREQUENCY BY USE OF EO EFFECT 81

Fig. 4. Cross sections of MZ structures for push–pull operation. (a) x-cut MZmodulator. (b) z-cut MZ modulator. (c) z-cut dual-electrode MZ modulator.

Fig. 5. Principle of IM. When the modulator is in “off” state, the lightwave isconverted into radiative modes at the junction of the output.

phase of modulator A is g(t)/2, and that of modulator B is−g(t)/2, where g(t) describes the optical phase difference be-tween the two arms of the MZ structure. An LN substrate haslarge EO effect along the c-axis of the LN crystal, where c-axisis directed at x- and z-axes in x- and z-cut LN substrates, re-spectively. The balanced push–pull operation can be obtained byusing an x-cut LN substrate and a coplanar waveguide (CPW)for electric signal guiding, as shown in Fig. 4(a), where an elec-tric signal is fed to the CPW. A z-cut LN MZ structure witha CPW shown in Fig. 4(b) can also be used for IM. The z-cutstructures can have large modulation coefficient K; however,the cross section is asymmetric, so that the induced phases atphase modulators A and B are different [30]. Thus, the mod-ulator having two CPWs (dual-electrode MZM), as shown inFig. 4(c), is needed to achieve the balanced push–pull oper-ation, where the electric signals with 180◦ phase differenceare applied on the electrodes. Fig. 5 shows the principle ofoperation. When the lightwaves in the two arms are in 221.68(D-0.0),)-0an


Fig. 7. High extinction ratio intensity modulator using active trimmers. Imbal-ance in the modulator can be compensated by changing voltages on electrodesA and B.

Fig. 8. Time domain profile of on–off keyed signal measured by zero spanmode of an optical spectrum analyzer. Solid and dashed lines denote the profileswith and without the compensation technique, respectively.

of electrode C was 15 GHz. Insertion loss of the device was6.9 dB. We can maximize the extinction ratio of the IM by thefollowing steps: 1) adjusting the bias voltages on A, B, and C(bias A, B, and C, henceforth) to maximize the output opticalpower; 2) adjusting bias C to minimize the output power; 3)stirring bias A or B, where slight change of one (bias A or B)causes reduction of the output power, but the other increases it;4) minimization of the output by using the bias that decreasesthe power in the previous step, and bias C. In the last step, wecan obtain optimal bias voltages A and B for high extinctionratio IM. In our experiment, we used bias B in step 4. It meansthat the optical intensity in the lower arm of the fabricated mainMZ is larger than in the upper arm, before compensation. Anoptimal bias voltage on electrode B for the compensation was0.21 (normalized by Vπ), which corresponds to 5% amplitudereduction in the lower arm. A half of the amplitude differencebetween the arms would be coupled into the output port of themodulator. The residual output in the “off” state without com-pensation should be−32.0 dB (= 20 log 0.05/2), which agreesvery well with the measured result. Thus, we conclude that theimbalance between the arms is dominant in the residual outputin the “off” state of fabricated MZ modulators. As shown inFig. 8, we used zero span operation of an optical spectrum an-alyzer to measure high extinction ratio on–off keying, but timedomain response of the measurement setup was dominant in therise time. The input power and wavelength were 4.8 dBm and1550 nm, respectively. The extinction ratios with and withoutcompensation were, respectively, 71.4 dB and 31.9 dB.

C. IM With RF Signals

Lightwaves with RF signals can be obtained by optical IMwith sinusoidal RF signals, where the MZ modulator generatesUSB and LSB components. When g(t) is a sinusoidal signaldescribed by 2ARF sin 2πfmt + φB, the optical output can beexpressed by

R =12ALWe2πif0t

[ei(ARF sin 2πfm t+φB/2)

+ e−i(ARF sin 2πfm t+φB/2)]

(10)

=12ALWe2πif0t

∞∑n=−∞

Jn(ARF)e2πi[f0t+nfm t]

× [eiφB/2 + (−1)ne−iφB/2] (11)

= ALWe2πif0t

[cos

φB

2

∞∑n=−∞

J2n(ARF)e2πi[f0t+2nfm t]

+ i sinφB

2

∞∑n=−∞

J2n+1(ARF)e2πi[f0t+(2n+1)fm t]

].

(12)

The output optical intensity |R|2, which can be detected by ahigh-speed photodetector, is expressed by

|R|2 � |ALW|2[J2

0 (ARF) cos2φB

2+ 2J2

1 (ARF) sin2 φB

2

− 4J0(ARF)J1(ARF) sinφB

2cos

φB

2sin 2πfmt

+ 2{

2J0(ARF)J2(ARF) cos2φB

2

− J21 (ARF) sin2 φB

2

}cos (2 · 2πfmt)

](13)

where we assumed that ARF � 1, and the high-order com-ponents are neglected. By using Taylor’s expansion of Besselfunction, we get a simple equation

|R|2|ALW|2 =

12

+1 − |ARF|2

2cos φB

− ARF sin φB sin 2πfmt

+12|ARF|2 cos φB cos (2 · 2πfmt) (14)

which is useful to discuss performance of optical systems for RFsignal transmission where the photodetector generates RF sig-nals. The intensities of the fundamental component sin(2πfmt)and the second-order harmonic cos (2 · 2πfmt) can be con-trolled by the dc-bias φB. The fundamental and second-ordercomponents are proportional to sin φB and cos φB, respec-tively. The ratio between the average power and the RF signalcomponent dominates conversion efficiency from lightwaves toRF signals at the photodetector. The ratios for the fundamentaland second-order components are, respectively,


Fig. 9. Optical spectra of DSB-SC signals with high carrier suppression usingan MZM with intensity trimmers. Solid and dashed lines denote the DSB-SCsignals with and without the imbalance compensation, respectively.

D1 =∣∣∣∣ 2ARF sin φB

1 + (1 − |ARF|2) cos φB

∣∣∣∣ (15)

D2 =∣∣∣∣ |ARF|2 cos φB

1 + (1 − |ARF|2) cos φB

∣∣∣∣ . (16)

In the case of φB = π, the even order components in the outputR, including the carrier component e2πif0t, become zero and theaverage power |R|2 goes to a minimum of |ARF|2/2, where thedominant components are the first-order USBs and LSBs. Thismodulation scheme, called DSB-SC modulation, is suitable forRF signal transmission, because undesired dispersion effects inoptical fibers are suppressed, and the ratio between the averagepower and the second-order RF signal D2 becomes a maxi-mum. In addition, D2 does not depend on ARF, while D1 goesto zero when ARF → 0. Thus, DSB-SC modulation can provideeffective RF signal generation even when the modulation signalamplitude (ARF) is very small. However, there are some un-desired spectral components due to imperfection of modulatorsor electric circuits for feeding. In order to achieve small sig-nal modulation, we need to suppress the undesired components.The carrier suppression ratio in DSB-SC depends on the extinc-tion ratio of the MZM, so that the carrier can also be suppressedlargely by using the high extinction ratio IM with trimmers [31].We demonstrated DSB-SC modulation to generate a two-tonelightwave signal by using the high extinction ratio modulationdescribed in the previous section. We applied sinusoidal a mi-crowave signal of 20 GHz to electrode C, where the signal powerwas 13.6 dBm. Bias C was set to be in a minimum dc transmis-sion point. Fig. 9 shows the optical spectra of DSB-SC signalswith and without using the trimming technique. Carrier suppres-sion ratios using the compensation technique with respect to thefirst-order components was 40.4 dB. We also measured a beatsignal of the two spectral components, by feeding the opticalsignal to a high-speed photodetector. Figs. 10 and 11 show widespan (dc—50 GHz) spectra of the beat signals with and withoutthe compensation technique, respectively. A millimeter-wavesignal of 2fm was successfully generated with high suppres-sion of undesired components. Spurious suppression ratio wasgreater than 62.1 dB without using any optical or electrical filtersfor suppression of undesired components, while that of conven-

Fig. 10. Spectrum of millimeter-wave signal generated from the DSB-SCoptical signal with imbalance compensation. The source modulating signalfrequency was 20 GHz. The second-order harmonic at 40 GHz was effectivelygenerated with high spurious suppression.

Fig. 11. Spectrum of millimeter-wave signal generated from the DSB-SCoptical signal without imbalance compensation. The optical output had theresidual source modulation signal at 20 GHz.

Fig. 12. Noise figure of millimeter-wave signal generated from the DSB-SCoptical signal.

tional technique was 21.0 dB, as shown in Fig. 11. The linewidthof 40.0 GHz signal was less than 1 Hz. Fig. 12 shows the noisefigure of the second-order harmonic generation, which is definedby SSB phase noise ratio between the source modulating sig-nal (20.0 GHz) and the generated millimeter-wave signal (40.0GHz), where the phase noise of the 40 GHz millimeter-wavesignal was −100.5 dBc/Hz at 10 kHz offset. An average noisefigure in an offset frequency region from 10 Hz to 1 MHz was5.7 dB and close to the theoretical lower limit 20 log 2 = 6dB.Thus, we deduce that phase noise due to the high extinction ratio


Fig. 15. Principle of optical SSB modulation. One of the sideband components(USB or LSB) is eliminated at the junction of the output, while the inputlightwave component is vanished in the sub-MZ structures. In this illustration,USB is selected at the junction; however, LSB can be generated instead of USBby changing the dc-bias voltage on electrode DCC or RFC.

Fig. 16. Conversion efficiency and SNR of optical frequency shift using SSBmodulation, where the SNR was defined by the ratio between the desired side-band (USB or LSB) and undesired spurious components.

can be switched quickly by feeding a high-speed non-return-to-zero (NRZ) signal to electrode RFC. The amplitudes of USBand LSB are, respectively, described by [1 + i exp (iφFSK)]/2and [−1 + i exp (iφFSK)]/2, where φFSK is the induced phasedifference at RFC, and φFSK = −90◦ corresponds to an op-timal condition for USB generation (U = 1). Thus, by feed-ing an NRZ signal, whose zero and mark levels correspond toφFSK = −90◦,+90◦, to RFC, we can generate an optical FSKsignal, without any parasitic IM. The bandwidth for the FSKsignal should be smaller than the RF frequency fm, when theFSK signal is demodulated by optical filters.

The signal-to-noise-ratio (SNR) and conversion efficiencyof the frequency shift by the modulator are given byJ1(ARF)/J3(ARF) and J1(ARF), respectively (see Fig. 16).The conversion efficiency has a maximum of 0.582 (−5.36 dB)when ARF = Am. As shown in (24), the output lightwave hasundesired high-order sideband components whose orders are−3,+5,−7, . . ., for U = +1, and 3,−5,+7, · · ·, for U = −1.Thus, the frequency difference between the generated spectralcomponents equals 4fm. We note that the dominant high-ordercomponent (J3) can be suppressed by using a predistortiontechnique [14].

B. 40-Gb/s FSK Modulation and 40-GHzOptical Frequency Shift

By using the z-cut dual-electrode configuration shown inFig. 4(c), we fabricated a high-speed FSK/SSB modulator de-

Fig. 17. z-Cut dual-electrode FSK/SSB modulator, consisting of six opticalphase modulators with traveling-wave electrodes.

Fig. 18. Frequency response of the six phase modulators with the traveling-wave electrodes in the z-cut FSK/SSB modulator, measured by a networkanalyzer.

signed for 40-Gb/s FSK modulation and 40-GHz optical fre-quency shift [32]. The modulator has six electrodes (A1, A2,B1, B2, C1, and C2), as shown in Fig. 17. For FSK modulation,a high-speed data signal should be fed to the main MZ structure,to change the output optical frequency according to the data sig-nal. In order to reduce Vπ for the high-speed data signal, the FSKmodulator should have long traveling-wave electrodes for themain MZ structures, so that the length of the sub-MZ structurewas limited by the wafer size, where Vπ of the sub-MZ structureswas greater than 10 V at 40 GHz. Frequency response of thephase modulators in the fabricated FSK modulator are shown inFig. 18. The 3-dB bandwidths were about 30 GHz and the 6-dBbandwidths were greater than 40 GHz. The main and sub-MZstructures were successfully integrated onto a single-chip usingz-cut LN integration platform. The length of the electrodes inthe two sub-MZ structures (A1, A2, B1, and B2) was 16 mm,while that of the electrodes in the main MZ structure (C1 andC2) was 32 mm. The Vπ of the sub-MZ structures (MZA andMZB) and the main MZ structure (MZC) were, respectively,4.9 V and 2.5 V in push–pull operation at low frequency, wherethe insertion loss of the modulator was 4.2 dB.

Fig. 19 shows the experimental setup for FSK modulation.Four sinusoidal electric signals having 90◦ phase differenceswere applied to electrodes A1, A2, B1, and B2, for generationof sideband components. The phase differences were controlledby using tunable delay lines. Fig. 20 shows the spectrum of theFSK modulator output, where we applied dc voltage on the mainMZ structure. As described in [14], the modulator can suppressthe input component (carrier) and one of the sidebands (USB orLSB), so that output can be a single mode signal consisting ofUSB or LSB. This scheme is called single-sideband suppressedcarrier (SSB-SC) modulation. By changing the dc voltage φC,


Fig. 19. Experimental setup for FSK modulation using the z-cut FSK/SSBmodulator. FSK signal was demodulated by an AWG.

Fig. 20. Optical spectra of frequency shifted signals generated by the z-cutFSK/SSB modulator, where the modulation frequency was 40 GHz.

we can select the output optical frequency (USB or LSB), asshown in Fig. 20. The extinction ratio of undesired componentswas 17 dB, where ARF � 1.7 rad. We deduce that the even or-der spurious sideband components in Fig. 20 were mainly dueto misalignment in the RF circuits shown in Fig. 19. The imbal-ance in the modulator can also affect the spurious suppressionratio, where the zeroth-order component can be largely sup-pressed by using the high extinction ratio modulation techniquedescribed in the previous section. The signal frequency fm was40 GHz, so that the optical frequency deviation was 80 GHz. Bysweeping the frequency of the sinusoidal electric signals as de-scribed in [33], we can construct an optical frequency sweeperof ± 40 GHz tunable range.

As shown in Fig. 19, an optical 40 Gb/s FSK signal was gen-erated by feeding a non-return-to-zero (NRZ) 223 − 1 pseudo-random-bit-sequence (PRBS) 40 Gb/s data signal to the mainMZ structure (electrode C1). Fig. 21 shows an optical spectrumof the 40 Gb/s FSK signal. The optical FSK signal was demod-ulated into an OOK signal, by an arrayed-waveguide (AWG).One of the sideband components (USB or LSB) can be taken outfrom an optical output port of the AWG whose channel separa-tion was 50 GHz, as shown in Figs. 22 and 23. The results showthat the eyes are clearly opened both in USB and LSB. Slight

Fig. 21. Spectrum of the 40 Gb/s optical FSK signal consisting of USB andLSB, where frequency deviation was 80 GHz.

Fig. 22. Spectra of demodulated optical FSK signal. The frequency deviationof the AWG for demodulation was 100 GHz, while that of the FSK signal was80 GHz.

Fig. 23. Eye diagrams of demodulated optical FSK signal. USB (left). LSB(right).

intensity difference between USB and LSB in Fig. 22 would bedue to some misalignment in the AWG. The frequency devia-tion was greater than the bit rate in this experiment, so that theoutput was a wide-band FSK signal, which can be demodulatedincoherently by using a conventional optical filter. However, theFSK modulator can also be applied to narrow-band FSK for-mats, such as continuous-phase FSK, including minimum shiftkeying, by using synchronization between data for sideband se-lection and sinusoidal signal for sideband generation [11], andinitial phase control technique [12].

C. 80-Gb/s QPSK Modulation

By using the SSB modulator designed for optical frequencyshift, we can control both phase and amplitude of the outputlightwave simultaneously [4]. We apply a pair of basebandsignals g1(t) and g2(t) to electrodes RFA and RFB, respec-tively, where the dc-bias condition is identical to that of SSB


Fig. 24. Experimental setup for DQPSK modulation using the z-cut FSK/SSBmodulator. DQPSK signal was demodulated by a 1-b delay interferometer.

modulation. The output can be expressed by

R =ALWeiπ/4

2e2πif0t [ cos [g1(t)/2] + i cos [g2(t)/2]] (28)

=ALWeiπ/4

2

√cos2[g1(t)/2] + cos2[g2(t)/2]

× exp i[2πf0t + tan−1( cos [g2(t)/2]/ cos [g1(t)/2])

](29)

where U is assumed to be −1(φFSK = π/2). By using foursymbols of (g1, g2) = (0, 0), (2π, 0), (0, 2π), (2π, 2π), we cangenerate a QPSK signal, where the phases of the output,tan−1(cos [g2(t)/2]/ cos [g1(t)/2]), are 0, π/2, π, 3π/2. Sim-ilarly, binary PSK can be achieved by an MZM with symbols ofg = 0, 2π.

For QPSK modulation, a pair of data signals are appliedto the two sub-MZ structures, to achieve control of in-phaseand quadrature components [32]. Thus, the modulator shouldhave long electrodes in the sub-MZ structures, to reduce Vπ

of the sub-MZ structures. We fabricated an FSK/SSB modula-tor whose electrode lengths of the main and sub-MZ structureswere, respectively, 16 and 32 mm. The Vπ of the main and sub-MZ structures were, respectively, 4.9 and 2.5 V in push–pulloperation at low frequency. Optical 3 dB bandwidth of eachelectrode was greater than 27 GHz. The insertion loss of themodulator was 5.1 dB. Fig. 24 shows the experimental setup forDQPSK modulation. Each of the sub-MZ structures was in anull-bias point, where optical phase difference between the twosub-MZ structures was adjusted to π/2 by using electrode C1or C2. A pair of NRZ data streams at 40 Gb/s were obtainedfrom a 4:1 multiplexer that combines four 10-Gb/s subchannelsof 27 − 1 PRBS. As shown in Fig. 24, one of the streams wasfed to MZA for I component modulation and the other was fedto MZb for Q component, where the delay between the twostreams was adjusted to be 115 bit. The amplitude of I and Qsignals at the input ports of the modulator was 6.5 V (peak-to-peak), corresponding to 2Vπ at 40 Gb/s, to generate an 80 Gb/soptical DQPSK signal at the output port of the modulator. Asshown in Fig. 25, we measured an optical spectrum of a DQPSKsignal at the output port of the modulator, without using any op-tical filters, where full spectral width measured 20 dB belowthe maximum of the central wavelength peak was 60 GHz. Atthe DQPSK demodulator shown in Fig. 24, the DQPSK sig-

Fig. 25. Spectrum of 80 Gb/s optical DQPSK signal.

Fig. 26. Eye diagrams of tributary channels of ∆φ = +π/4 (left) and ∆φ =−π/4 (right) demodulated from 80-Gb/s DQPSK signal.

Fig. 27. BER curves of two tributaries of the 80-Gb/s DQPSK signal.

nal generated at the modulator was decoded by a one-bit delayinterferometer whose constructive and destructive ports wereconnected to a balanced photodetector. However, no precoderwas employed for our experiment, and hence, there was a de-terministic mapping of data from input to output. In order toallow bit-error-ratio (BER) measurements, the error detectorwas programmed with the expected data sequence. We used asingle receiver to decode each 40-Gb/s tributary by adjustingthe differential optical phase in the one-bit delay interferometer(∆φ) at π/4 or −π/4. Fig. 26 shows eye diagrams measuredat the electric output of the balanced photodetector. In back-to-back transmission, clear eye openings were observed for thetwo tributaries whose symbol rate was 40 Gbaud. We measureda back-to-back BER curve of a subchannel extracted from eachtributaries by a 1 : 4 demultiplexer, as shown in Fig. 27, wherethe receiver sensitivity at the BER of 10−9 was −20 dBm.


Fig. 28. Principle of quadruple DSB-SC modulation, where the second-orderUSB and LSB are generated with suppression of the other sideband components.

V. HIGH-ORDER SIDEBAND GENERATION TECHNIQUES

A. Quadruple DSB-SC Modulation

Fig. 28 shows the principle of QDSB-SC modulation. Thesecond-order USB and LSB components can be obtained byusing two MZMs that are connected in series [19], [20]. Thefirst-order USB and LSB components are generated at the firstMZM, where the carrier component is highly suppressed. Theoptical frequency of the USB is f0 + fm, and that of the LSBis f0 − fm, where f0 is the optical frequency of the input light-wave. In this section, a spectral component whose frequencyis f0 + Nfm is, henceforth, called f0 + Nfm component. Atthe second modulator, the f0 + fm component (the USB at thefirst modulator) is converted into f0 and f0 + 2fm components.At the same time, f0 and f0 − 2fm components are generatedfrom the f0 − fm component (the LSB at the first modulator).Thus, there are three spectrum components: f0, f0 + 2fm, andf0 − 2fm in the output of the second modulator. However, wecan control the intensity of the f0 component by changing thephase difference between the modulating signal at the first andthe second modulator. When the phase difference is equal to90◦, the f0 component from the f0 + fm component and thatfrom the f0 − fm component interfere destructively, so that thecarrier f0 component can be suppressed at the output. As shownin Fig. 28, the output has f0 + 2fm and f0 − 2fm components,where other components are highly suppressed. Thus, we canobtain a millimeter-wave signal, whose frequency is 4fm, byfeeding the output lightwave signal to a high-speed photodetec-tor. In this paper, we call this technique for the fourth-order har-monic component (4fm), quadruple frequency DSB-SC modu-lation. We demonstrated generation of 42-GHz millimeter-wavefrom 10.5-GHz microwave by using the quadruple frequencyDSB-SC modulation technique. Fig. 29 shows the output light-wave spectra, where the phase difference between the modulat-ing signals at the first and second modulators are 0◦ and 90◦.In the case of 90◦ phase difference, the carrier component islargely suppressed, while that in the case of 0◦ phase differ-ence, it is greater than the f0 + 2fm and f0 − 2fm components.The phase difference can be adjusted by an optical or electricaldelay. In our experiment, we used an electrical delay, wherethe carrier suppression ratio with respect to the f0 + 2fm andf0 − 2fm components was 45.8 dB. The suppression ratio of theother components was 34 dB, where f0 + 3fm and f0 − 3fm

Fig. 29. Optical spectrum of the quadruple DSB-SC signal. The modulatingfrequency was 10.5 GHz. The frequency deviation between the second-orderUSB and LSB was 42.0 GHz.

Fig. 30. Electric spectrum of the quadruple DSB-SC signal. Spurious suppres-sion ratio with respect to the fourth-order harmonic at 42.0 GHz was 41.8 dBm.

components were most significant. The third-order componentsare due to nonlinearity of optical modulation. The input opticalpower was 12.7 dBm. The modulating signal frequency fm is10.5 GHz, while the signal powers were 20.9 dBm at the firstmodulator and 21.0 dBm at the second modulator. Both themodulators had active trimmers, and the imbalances in the armswere compensated by using the technique described in Fig. 7.We used two polarization controllers at the input ports of themodulators. The insertion losses of the polarization controllerwere 1.1 dB at the first modulator and 0.4 dB at the secondmodulator. By feeding the optical output to a high-speed pho-todetector, we generated a millimeter-wave signal, as shown inFig. 30, where the spurious suppression ratio with respect to4fm component was 41.8 dB.

B. ROM

ROM, whose setup consists of a pair of optical filters placedat the optical input and output ports of an optical modulator,can generate a lightwave modulated by an RF signal whose fre-quency is an integer multiple of that of an electric RF signalapplied to the modulator [21]–[24]. Some of the optical side-band components are fed to the optical modulator again, wheredesired sideband components are taken out from the optical filterplaced at the output port, and are not fed to the modulator again,in order to effectively generate specific sideband componentswithout spreading the optical power over undesired sideband


Fig. 31. Schematic of ROM consisting of an optical phase modulator and apair of FBGs.

Fig. 32. Reflectivities of input and output filters using uniform FBGs.

components. This is in contrast to mode-lock lasers [34], andoptical Fabry–Perot modulators [35], [36], where all generatedsideband components are recycled into the modulators regard-less of the optical frequency. Lightwave reciprocates severaltimes in the modulator, but lightwave oscillation is not usedin ROM. As a result, we can effectively obtain stable outputwithout any complicated feedback systems that are necessary toconventional techniques for optical generation of RF signals.

Fig. 31 shows a schematic of an integrated ROM, consistingof an LN phase modulator and two uniform FBGs, where theFBGs were fixed on SiO2 substrates and directly attached to thephase modulator chip. The uniform FBGs have narrow pass-bands near the edges of the main reflection bands, as shown inFig. 32. The bandwidth of the main reflection band was slightlynarrower than 320 GHz, where the passbands were at 1550.9and 1553.5 nm. Designed frequency of modulating signal fedto the phase modulator was 40 GHz. When an input lightwavewavelength is close to 1550.9 nm, the eighth-order sidebandwhose wavelength would be close to 1553.5 nm can be takenout from the output filter. The optical phase modulator had atraveling-wave electrode to achieve high-speed operation, whereVπ halfwave voltage of the modulator was 4.1 V at dc and 7.8 Vat 40 GHz. The modulator had a pair of RF input ports, RF inputs1 and 2 in Fig. 31, in order to obtain bi-directional modulation.In ROM sideband generation process, sideband components inthe reflection band reciprocate between the FBGs, and pass themodulator several times, so that the bidirectional modulation isindispensable for effective high-order sideband generation.

Fig. 33 shows an optical output spectrum, where the opticalinput power and wavelength were 5.7 dBm and 1550.965 nm,respectively. The RF powers for forward and backward prop-agating lightwaves were, respectively, 21.4 and 20.6 dBm, sothat the total RF signal power fed to the two RF input ports was

Fig. 33. Output lightwave spectrum, where the RF signal frequency was39.06 GHz.

24.0 dBm (5.0 V zero-to-peak amplitude). The modulator wasdesigned for 40 GHz RF input; however, high-order sidebandgeneration efficiency of a fabricated modulator became maxi-mum when the RF signal frequency was 39.06 GHz. We deducethat this is due to the difference between designed and actualgroup delays in the FBGs. As shown in Fig. 32, the input light-wave was in the narrow passband near the shorter wavelengthband edge of the main reflection band. The eighth-order LSBcomponent was in the passband near the longer wavelength bandedge. Thus, the LSB components lower than the eighth-orderwere reflected by the input and output filters, to reciprocateseveral times in the ROM. The frequency deviation betweenthe input lightwave and the eighth-order LSB was 312.48 GHz,where spectral components generated by ROMs are stationaryphase-locked to each other without using feedback control. Theconversion efficiency from the input lightwave to the eighth-order sideband component was −30.0 dB, while that of theconventional optical phase modulation with an RF signal of24.0 dBm was −95.9 dB, so that the enhancement factor in theeighth-order sideband generation was 65.9 dB.

VI. CONCLUSION

We investigated control of lightwave amplitude, phase, andfrequency using high-speed EO modulation techniques. OpticalMZ intensity and FSK/SSB modulators based on phase mod-ulation can provide precise and agile control of lightwaves.Ultrahigh extinction ratio IM can be achieved by MZ intensitymodulators with active trimming technique. FSK/SSB modula-tors can also generate optical QPSK signals as well as opticalFSK signals, where 40 Gb/s FSK and 80 Gb/s DQPSK modu-lation were achieved. For sub-tetrahertz signal generation, wedescribed high-order sideband generation techniques: QDSB-SC and ROM. 312-GHz photonic LO signal generation wasachieved by using an ROM with uniform FBGs.

ACKNOWLEDGMENT

The authors would like to express their appreciation toDr. M. Tsuchiya and Dr. S. Shinada of the National Instituteof Information and Communications Technology for their use-ful discussion. They also wish to acknowledge the support ofJ. Ichikawa, S. Oikawa, K. Higuma, and T. Fujita of SumitomoOsaka Cement and K. Yoshiara of Mitsubishi Electric.


Takahide Sakamoto (S’98–M’03) was born inHyogo, Japan, on May 23, 1975. He received theB.S., M.S., and Ph.D. degrees in electronic engineer-ing from the University of Tokyo, Tokyo, Japan, in1998, 2000, and 2003, respectively.

In 2003, he joined the Communications ResearchLaboratory (currently National Institute of Informa-tion and Communications Technology), Tokyo. Hehas been engaged in all-optical signal processingbased on nonlinear optics. His current research inter-ests include electrooptic devices, such as LiNbO3

modulators, and their applications to photonic communication systems.Dr. Sakamoto is a member of the IEEE Lasers and Electro-Optics Society

(LEOS) and the Institute of Electronics, Information and Communication En-gineers (IEICE) of Japan.

Masayuki Izutsu (S’70–M’75–SM’90–F’04) re-ceived the B.E., M.E., and D.Eng. degrees in electri-cal engineering from Osaka University, Osaka, Japan,in 1970, 1972, and 1975, respectively.

In 1975, he joined the Department of ElectricalEngineering, Faculty of Engineering Science, OsakaUniversity, where he was engaged in the field ofguided-wave optoelectronics. From 1983 to 1984, hewas a Senior Visiting Research Fellow, Departmentof Electronics and Electrical Engineering, Universityof Glasgow, Glasgow, U.K. In 1996, he joined the

Communications Research Laboratory, Ministry of Posts and Telecommuni-cations (currently the National Institute of Information and CommunicationsTechnology), Tokyo, Japan, where he is currently a Distinguished Researcher,and is in-charge of its New Generation Network Research Center. He is also anExpert Researcher at the Research Center for Science Systems, Japan Societyfor Promotion of Science (JSPS).

Dr. Izutsu is a Corporate Member of the Science Council of Japan and aFellow of the Institute of Electronics, Information and Communication Engi-neers (IEICE). He is the recipient of the Best Paper Award and the Award forSignificant Achievement in 1981 and 1988, respectively, from IEICE.

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