linear technology linear technology...linear technology magazine • august 1995 3 lt1510 battery...

AUGUST 1995 VOLUME V NUMBER 3

The LT1304: MicropowerDC/DC Converterwith IndependentLow-Battery Detector

by Steve Pietkiewicz

IntroductionIn the expanding world of low power

portable electronics, a 2- or 3-cellbattery remains a popular powersource. Designers have many optionsfor converting the 2V-4V battery volt-age to 5V, 3.3V, and other requiredsystem voltages using low voltageDC/DC converter ICs. The LT1304offers users a micropower step-upDC/DC converter featuring BurstModeTM operation and a low-batterydetector that stays alive when theconverter is shut down. The deviceconsumes only 125µA when active,yet can deliver 5V at up to 200mAfrom a 2V input. High frequencyoperation up to 300kHz allows theuse of tiny surface mount inductorsand capacitors. When the device isshut down, the low-battery detectordraws only 10µA. An efficient internalpower NPN switch handles 1A switchcurrent with a drop of 500mV. Up to85% efficiency is obtainable in2-cell-to-5V converter applications.The fixed-output LT1304-5 andLT1304-3.3 versions have internalresistor dividers that set the outputvoltage to 5V or 3.3V, respectively.

OperationThe LT1304’s operation can best

be understood by examining theblock diagram in Figure 1 (page 17).

Comparator A1 monitors the outputvoltage via resistor-divider string R3/R4 at the FB pin. When VFB is higherthan the 1.24V reference, A2 and thetimers are turned off. Only the refer-ence, A1, and A3 consume current,typically 120µA. As VFB drops below1.24V plus A1’s hysteresis (about6mV), A1 enables the rest of the cir-cuit. Power switch Q1 is then cycledon for 6µs, or until current compara-tor A2 turns of f the on-timer,whichever comes first. Off-time is fixedat approximately 1.5µs. Q1’s switch-ing causes current to alternately buildup in inductor L1 and discharge intooutput capacitor C2 via D1, increas-ing the output voltage. As VFBincreases enough to overcome C1’shysteresis, switching action ceases.C2 is left to supply current to theload until VOUT decreases enough toforce A1’s output high, and the entirecycle repeats.

If switch current reaches 1A, caus-ing A2 to trip, switch on-time isreduced. This allows continuous-mode operation during bursts. A2monitors the voltage across 7.2Ωresistor R1, which is directly relatedto the switch current. Q2’s collectorcurrent is set by the emitter-arearatio to 0.5% of Q1’s collector

IN THIS ISSUE . . .

COVER ARTICLEThe LT®1304: Micropower DC/DCConverter with IndependentLow-Battery Detector ................ 1Steve Pietkiewicz

Editor's Page ............................2Richard Markell

LTC in the News ....................... 2

DESIGN FEATURESLT1510 Battery Charger ICSupports All Battery Types,Including Lithium-Ion ............... 3Chia Wei Liao

LT1118 TerminatesActive SCSI and More ................ 5Gary Maulding

LT1580 Low Dropout RegulatorUses New Approach toAchieve High Performance ........ 7Craig Varga

Power Factor Correction —Part Three: Discontinuity ......... 9Dale Eagar

The New LT1508/LT1509 CombinesPower Factor Correction and aPWM in a Single Package ........ 11Kurk Mathews

LTC®1329 Micropower,8-Bit, Current Output DAC ...... 16K.S. Yap

DESIGN INFORMATIONRS232 Transceiver Protected to±15kV IEC-801-2 ESD Standards...............................................20Gary Maulding

V.35 Transceivers Allow3-chip V.35 Port Solution ....... 22Y.K. Sim

DESIGN IDEAS ................. 24–37(complete list on page 24)

New Device Cameos ................ 38

continued on page 17

, LTC and LT are registered trademarks of Linear Technology Corporation.Burst Mode and Super Burst are trademarks of Linear Technology Corporation.

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

2 Linear Technology Magazine • August 1995

EDITOR'S PAGE

Eggs on the Wall? by Richard Markell

I was in the lunch room severalweeks ago calmly eating my lunch,when suddenly a large “boom”sounded behind me. (No, it wasn’t anunderrated tantalum capacitor.) Iturned around (slowly) to find thatone of our physicists had tried tohard boil an egg in the microwavewithout first putting a vent hole inthe eggshell. My first reaction was toexclaim that any physicist shouldknow that when most things get hot,they expand (go boom!). Sure, wecould plot the size of the vent holeversus the “time to go boom.” Theconservative engineer might do this.Both the analytical approach and try-ing things to see what happens areuseful, but maybe we need more“booms.” It seems that, in today’sbusiness atmosphere, there is toomuch conservatism and too fewpeople just trying things to see whathappens. At LTC we try to encourageboth approaches.

The featured article for this issuehighlights the LT1304 micropowerDC/DC converter, which includes alow-battery detector that stays alivewhen the converter is shut down. TheLT1304 is a micropower step-upDC/DC converter tailored for opera-tion from two or three cells andfeaturing Burst Mode operation.The device consumes only 125µ Awhen operating and can deliver 5V atup to 200mA from a 2V input. Thedevice features 300kHz operation,allowing the use of tiny surface mountinductors.

A second article features theLT1118 family of fixed voltage regula-tors that can both source and sinkcurrent. These devices are ideal foruse as active SCSI terminators. TheLT1118 family of regulators featuresexcellent stability and load transientsettling with most capacitive loads.The devices can source up to 800mAand sink up to 400mA. Quiescentcurrent is only 600µA unloaded.

The LT1510 is a battery charger ICthat allows fast charging of all typesof batteries, including Li-Ion. The

LT1510 allows users to precisely setbattery current and voltage, and tochoose the best charging algorithmfor a given application. For manysituations, such as Li-Ion, no exter-nal monitoring of the battery isrequired. The LT1510 is a currentmode PWM battery charger thatallows constant-current and/or con-stant-voltage charging. The internalswitch is capable of delivering 1.5ADC current (2A peak current).

Another featured product is theLT1508/LT1509 “combo” powerfactor correction circuit and PWMcontroller in a single package. TheLT1508/LT1509 integrates thefunctionality of an LT1248 powerfactor controller with a 50% maxi-mum duty cycle PWM in a single20-pin DIP or SOIC.

“Power Factor Correction: PartThree,” continues the series that be-gan two issues ago. This installmentdiscusses discontinuity, and the griefit causes for those trying to under-stand the operation of the PFC.

The LTC1329, a micropower, 8-bit,current output DAC, is the subject ofa short article. The LTC1329 is de-signed to digitally control a powersupply’s output voltage or to serve asa trimmer pot replacement. This is aversatile part with a 1, 2 or 3 wireinterface. Additional information onapplying LTC’s DAC product line willbe found in a feature on 12-bit voltagemode DACs. This article highlightscircuits ranging from a computer-controlled 4-to-20mA current loop toan opto-isolated serial interface.

In our “Design Information” sec-tion, we present information onelectrostatic discharge (ESD) and theIEC-801-2 specification. Our popularRS232 transceiver, the LT1137A, hasbeen upgraded to include on-chipESD protection compliant with levelfour of the IEC-801-2 standard. Thisupgrade allows full ESD protectionwith no energy absorbing componentsexternal to the LT1137A.

Our “Design Ideas” section is chockfull of useful circuits. These include

circuits for driving diode-ring mixers,a fully differential 8-channel, 12-bitA/D system, and a circuit fortouchscreen interface.

LTC in the News...Linear Technology reports record

annual and quarter sales!!!Thanks to all of you for your hard

work and constant drive to make LinearTechnology the best analog companyworld wide, the company’s annualsales once again increased to a record$265,023,000 for fiscal year end July2, 1995, an increase of 32% over netsales from the previous year. Thecompany also reported record netincome for the year of $84,696,000 or$2.22 per share, an increase of 49%over the $56,827,000, or $1.51 pershare, reported for fiscal 1994.

Linear Technology also announcedthat the board of directors approvedboth an increase in the quarterlydividend to $0.08 from $0.07 pershare and a two-for-one stocksplit of its common stock. A cashdividend will be paid on August 23,1995 to shareholders of record onAugust 4, 1995. Shareholders ofrecord on August 11, 1995 will beissued certificates reflecting the stocksplit; these shares will be distributedon September 1, 1995.

Chief Executive Magazine’s May,1995 issue features the prestigious‘Charting The Chief Executive 100Index.’ Robert Swanson Jr. is amongthe top ten Chief Executives for a thirdyear in a row. In order to make theCE100, the following criteria had tobe met. Each CEO had to have been intheir position throughout 1992-1994,the company had to be public for thattime frame and the market value of thefirms public stock had to exceed $500million at the end of 1994.

Financial World’s July 18, 1995issue included Linear TechnologyCorporation on its list of “America’s 50Best Mid-Cap Companies.” LTC wasranked number 5 up from 21 theprevious year.

Investor’s Business Daily June28,1995 issue featured an interviewwith Paul Coghlan, titled ANALOGLIFE, Linear Technology Keeps PulseOf Light, Heat, Sound, Speed. Mr.Coghlan attributes the company’ssuccess to being in the right market atthe right time with a good strategy.

Linear Technology Magazine • August 1995 3

LT1510 Battery ChargerIC Supports All Battery Types,Including Lithium-Ion by Chia Wei Liao

The LT1510 can charge batteriesranging from 1V to 20V; ground-sensing of current is not required.The battery negative terminal can betied directly to ground. A saturatingswitch running at 200kHz gives highcharging efficiency and small induc-tor size. A blocking diode betweenthe chip and the battery is not re-quired because the chip goes intosleep mode and drains only 3µA whenthe wall-adapter is unplugged. Soft-start and shutdown features are alsoprovided. The LT1510S8 comes in an8-pin SOIC and the LT1510S16 comesin a 16-pin, fused-lead, power SOpackage with a thermal resistance of45°C/W.

Circuit OperationThe LT1510 is a current mode PWM

step-down (buck) switcher (see Fig-ure 1). The battery DC chargingcurrent is programmed by a resistorRPROG (or a DAC output current) atthe PROG pin. Amplifier CA1 con-verts the charging current throughRS1 to a much lower current (IPROG =500µA/A), which is then fed into thePROG pin. Amplifier CA2 comparesthe output of CA1 with the pro-grammed current and drives the PWMloop to force them to be equal. HighDC accuracy is achieved with averag-ing capacitor CPROG. Note that IPROGhas both AC and DC components.IPROG flows through R1 and generatesa ramp signal that is fed to the PWM

Figure 1. LT1510 block diagram

IntroductionA sure way to start an argument

among electronic engineers is to bringup the subject of battery charging.This relatively simple problem hasspawned more solutions than theproverbial mouse trap. This situationseems to be the result of misinforma-tion about the basic electrochemistryof the batteries themselves, the diffi-culty of quantifying results, and thefertile imaginations of engineers wholike to create. At times, battery charg-ing seems to have as much to do withreligion as with science.

The LT1510 was created with thissituation in mind. It is a basic build-ing block for transferring energyefficiently and accurately from onesource to another. Battery currentand voltage are precisely defined bythe user, who can decide which charg-ing algorithm is best and set up theLT1510 in whatever programmedcurrent or voltage combination isrequired. For many situations, espe-cially lithium-ion, no extra externalbattery monitoring is needed.

The LT1510 current mode PWMbattery charger is the simplest, mostefficient solution to fast charge mod-ern rechargeable batteries, includinglithium-ion, nickel-metal-hydride(NiMH), and nickel-cadmium (NiCd),that require constant-current and/or constant-voltage charging. The in-ternal switch is capable of delivering1.5A DC current (2A peak current).The onboard current sense resistor(0.1Ω) makes programming the charg-ing current very simple. One resistor(or a programming current from aDAC) is required to set the full charg-ing current (1.5A max.) to within 5%or the trickle-charge current (150mA)to 10% accuracy. With 1% reference-voltage accuracy, the LT1510S16meets the critical constant-voltagecharging requirement for lithium cells.

DESIGN FEATURES

–

+

–

+

–

+

–

+

– +

VSW

0.7V

1.5V

VBAT

VREF

VC

GND

SLOPE COMPENSATION

R2

R3

C1

PWMB1

CA2

–

+

–

+

CA1

VA

+

+

VREF 2.465V

SHUTDOWN

200kHz OSCILLATOR

S

RR

R1 1k

RS1IBAT

IPROG

IPROG

VCC

VCC

BOOST

SW

SENSE

BAT

0VP

1510 BD

PROG

RPROGCPROG

60k

IPROG IBAT

= 500µA/A

QSW

gm = 0.64Ω

CHARGING CURRENT IBAT = (IPROG)(2000)

= 2.465V RPROG

(2000)( )


control comparator C1 through bufferB1 and level-shift resistors R2 andR3, forming the current mode innerloop. The BOOST pin drives the NPNswitch (QSW) into saturation and re-duces power loss. For batteries suchas lithium, which require both con-stant-current and constant-voltagecharging, the 1% 2.5V reference andthe amplifier VA reduce the chargingcurrent when battery voltage reachesthe preset level. For NiMH and NiCd,VA can be used for overvoltage pro-tection. When an input voltage is notpresent, the charger goes into lowcurrent (3µA typical) sleep mode asinput drops to 0.7V below the batteryvoltage. To shut down the charger,pull the VC pin low with a transistor.

Typical Charging AlgorithmsThe following algorithms are

representative of current techniques:Lithium-Ion — charge at constant

voltage with current limiting set toprotect components and to avoid over-loading the charging source. Whenthe battery voltage reaches the pro-grammed voltage limit, current willautomatically decay to float levels.The accuracy of the float voltage iscritical for long battery life. Be awarethat lithium ion batteries in seriessuffer from “walk away,” because ofthe required constant-float-voltagecharging technique. “Walk away” is acondition where the batteries in theseries string wind up in different statesof the charge/discharge cycle. Theymay need to be balanced by redistrib-

Figure 3. Charging Li-Ion batteries

DESIGN FEATURES

1510_2. eps

+

+ +

SW

BOOST

10µF

R1 50K

1µF

0.1µF

IBAT 2V TO 20V

IBAT = 1A IBAT = 0.1A

1K300Ω

R2 5.6K

GND

SENSE

VCC

C1 0.22µF

WALL ADAPTOR

D1 1N914

1N5819

PROG

VC Q1 VN2222

BATON:

OFF:

22µF

LT1510S8

30µH

–

Figure 2. Charging NiCd or NiMH batteries

1510_3. eps

+

++

SW

BOOST

10µF

3.83kΩ0.1µF

4.2V

1K

R3 59K R4

25K

OVP

GND

SENSE

**OPTIONAL

VCC

C1 0.22µF

11V TO 25V DC WALL ADAPTOR

D1 1N914

1N5819

PROG

VC

Q3** VN2222

BAT

22µF

LT1510S16

30µH

–+

4.2V–

+–

1µF

300Ω

uting charge from one battery to an-other. This phenomenon is minimizedby carefully matching the batterieswithin a pack.

Nickel-Cadmium — charge at aconstant current, determined by thepower available or by a maximumspecified by the manufacturer. Moni-tor battery charge state using voltagechange with time (dV/dt), second de-rivative of voltage (d2V/dt), batterypressure, or some combination ofthese parameters. When the batteryapproaches full charge, reduce thecharging current to a value (top-off)that can be maintained for a long timewithout harming the battery. Afterthe top-off period, usually set by asimple time out, reduce the currentfurther to a trickle level that can bemaintained indefinitely, typically 1/10to 1/20 of the battery capacity.

Nickel-Metal-Hydride — same asNiCd, except that some NiMH batter-ies cannot tolerate a continuous lowlevel trickle charge. Instead, theyrequire a pulsed current of moderate

value, with a low duty cycle, so thatthe average current represents atrickle level. A typical scenario wouldbe one second “on” and thirty sec-onds “off,” with the current set tothirty times desired trickle level.

Recharging NiMHor NiCd Batteries

The circuit in Figure 2 will chargebattery cells with voltages up to 20Vat a full charge current of 1A (whenQ1 is on) and a trickle charge currentof 100mA (when Q1 is off). If the thirdcharging level is needed, simply adda resistor and a switch. The basicformula for charging current is:

2 4651 2

2000 1

2 4651

2000 1

.

.

R Rwhen Q on

Rwhen Q off

× ( )

× ( )For NiMH batteries, a pulsed trickle

charge can be easily implementedwith a switch in series with R1; switchQ1 at the desired rate and duty cycle.If a micro-controller is used to controlthe charging, connect the DAC cur-rent sink output to the PROG pin.

Recharging Lithium-IonBatteries

The circuit in Figure 3 will chargelithium ion batteries at a constantcurrent of 1.5A until battery voltagereaches 8.4V, set by R3 and R4. Itthen goes into constant-voltage charg-ing and the current slowly tapers offto zero. Q3 can be added to discon-nect R3 and R4 so they will not drainthe battery when the wall-adapter isunplugged.


LT1118 Terminates Active SCSIand More

The new LT1118 family of positive,fixed voltage regulators is unique inmaintaining output voltage regula-tion while both sinking and sourcingcurrent. Conventional positive volt-age regulators lose regulation if theload circuit attempts to return cur-rent to the regulator output. Capableof sourcing up to 800mA and sinking400mA, LT1118 family regulatorsfeature excellent stability and load-transient settling with any capacitiveload greater than 0.22µF. Their in-sensitivity to capacitive loading makesthe regulators easy to use in anyapplication within their power level.Quiescent current is only 600µA whenunloaded. 5V, 2.5V, and 2.85V ver-sions of the LT1118 are available in3-pin SOT-223 or fused-lead SO-8packages. The SO-8-packaged devicesinclude an enable pin that allowsthe device to be shut down with theoutput at high impedance. Quiescentcurrent drops to zero when the enablepin is low.

The ability to maintain regulationwhile both sourcing and sinking cur-rent is valuable in several regulatorapplications. Appropriate applica-tions are data-bus termination,power-supply splitting, and any ap-plication where fast settling of largeload transients is needed. The 2.85Vversion is ideal for use as an activetermination for SCSI cables. The 2.5Vversion makes a convenient low powersupply splitter in 5V systems. The 5Vversion is ideal where fast settling of

load transients is required. The shut-down pin on the SO-8-packageddevices is valuable for power manage-ment and supply sequencing.

Active SCSI TerminatorThe SCSI parallel bus requires cable

terminations at each end of the cable.The termination serves the dual pur-poses of impedance matching andproviding the pull-up to the 2.8Vnegated logic level. The open-drain oropen-collector drivers on the bus es-tablish the low level and makemultiplexing of equipment simple,due to the “wired-OR” nature of thebus signals.

Passive terminators consisting of aresistor divider from the termpowersupply (Figure 1) were commonly usedon early SCSI networks. Passive ter-minators have mostly disappearedfrom use in recent systems due to thehigh power dissipation and lack ofpower supply rejection of the resistordivider. The Boulay or active termina-tion has replaced the passiveterminations in today’s equipment. ABoulay active termination uses a2.85V voltage regulator to establishthe negated logic level and 110Ω re-sistors to provide the impedancematch to each data and control line(Figure 2). Power consumption whenall lines are in the high (negated)state is dominated by the low quies-cent current of the regulator, a 0.225Wsaving over resistor terminators in a27-line wide SCSI bus.

SCSI-2 introduced the use ofactive-negation drivers in SCSI bussystems. Active-negation drivers havethree states: active (low), negated(high), and high impedance. By rais-ing the data line above the 2.85Vtermination voltage when negated,the effects of cable reflections andcrosstalk between lines are overcome,improving the noise margin in highspeed data busses. Active negationmay be used on all but three SCSIdata and control lines (three lines,BSY, SEL, and RST, must be wire-OR’d and cannot use active negationdrivers). Each negated driver cansource up to 7mA when in the highstate. Most voltage regulators willlose regulation if any current is re-turned to their outputs, causing thetermination voltage to rise.

The LT1118-2.85, with currentsinking, is the perfect solution forthis application. The 800mA sourcingcapability and 400mA current-sinkcapability provide ample margin toterminate a 27-line wide SCSI busin the fully asserted or negated state.Even though SCSI standards restrictthe active negation drivers to a 3.24Vand 7mA negated signal level, manydriver circuits in use exceed the al-lowed SCSI drive levels. If themaximum of 24 lines are negated in aWide-SCSI bus, 168mA of current isforced back into the terminator. The400mA sink capability of the LT1118gives more than twice the required

by Gary Maulding

Figure 1. SCSI passive terminator Figure 2. Boulay active terminator

DESIGN FEATURES

1118_1.eps

5VTERMPOWER

220Ω

ONE PER BUS LINE

330Ω

1118_2.eps

5VTERMPOWER 18 OR

27 LINES

1µF0.1µF2.85V

REGULATOR

VIN VOUT

Authors can be contactedat (408) 432-1900


DESIGN FEATURES

degrade the noise margin of the bussignals. The capacitance inevitablycomes from the pin-to-pin capaci-tance of the IC package, the ESDprotection devices on the chip, andthe distributed capacitance of the dif-fused resistors on the chip. Thereduced noise margin of the SCSI busincreases error rates and limits thelength of the bus and the number ofnodes that can reliably be connected.The external surface mount resistorsused with the LT1118-2.85 reducethe capacitance to the stray PC boardcapacitance.

Power consumption further limitsthe capabilities of SCSI terminatorswith on-chip resistors. The worst-case power consumption for a SCSIterminator occurs when all data andcontrol lines are active (low). For a27-line wide SCSI bus, the regulatordissipates 1.2W and the resistors dis-sipate 2.0W. The surface mount ICpackages used for SCSI terminatorscannot dissipate this total power of3.2W. This fundamental limit restrictsthe devices with on-chip resistors tonine lines. Three terminator chipswith integral resistors must be used,versus one LT1118-2.85 with inex-pensive surface mount resistors. Thecost of three SCSI terminator ICswith integral resistors is more thanthree times the cost of an LT1118-2.85 and the external 110Ω resistors.

Power Supply SplitterOften a system has only a single 5V

power supply, but has analog sec-tions that require positive and negativesupplies. The current source and sinkcapabilities of the LT1118-2.5 areideal for generating a virtual ground

current handling without damage orloss of regulation.

The fast settling time and stabilityof the LT1118 allow the regulatoroutput capacitor to be reduced from atypical 10µF to a surface mount 1µFceramic capacitor saving boardspace and cost. The low, 600µAquiescent current of the regulatorallows the terminator to be connectedto the SCSI term power line when notin use.

The maximum voltage spikes forFigure 3’s circuit are less than 0.3Vand settle out in 5µs. These tests wereunder maximum load transient con-ditions: 800mA sourcing to 400mAsinking. This transient is far moresevere than would be seen in a SCSIbus termination, yet the voltageoutput is always within acceptableSCSI voltage levels. The amplitudeof the output voltage spikes maybe reduced further by use of largeroutput capacitors.

Several competitors offer activeSCSI terminators that use on-chipresistors. These terminators, whileoffering some convenience, are lim-ited in performance and are not aseconomical a solution as the LT1118-2.85. Capacitance on the output pinsof the terminators cause impedancemismatches at high frequencies,which increase cable reflections and

Figure 4. Supply splitter

Figure 5. Power sequencing using the LT1118

Figure 3. LT1118 typical application

LT1118-2.85

GND

1118_3.eps

IN

2.2µF

5V

1µF CERAMIC

ILOAD

OUT

+

to allow the use of analog compo-nents without a negative supply.Figure 4 shows the LT1118-2.5 usedas a 5V supply splitter. The LT1118can absorb large DC or transientground currents that would force highpower consumption if a resistive sup-ply splitter were used. The regulator’sfast settling time allows the use ofsmall capacitors, and its stability withany large capacitive load makes iteasy to apply to almost any splitterapplication. Competing supply split-ters are known to suffer from stabilityproblems with capacitive loading.

Power ManagementThe enable control on the SO-8

packaged LT1118 provides easy powermanagement solutions. A logic lowon the enable pin causes theregulator’s output to go high imped-ance and its quiescent current todrop to zero. A logic high brings theregulator into normal regulation.Intelligent power controllers canmake use of this capability to selec-tively power subcircuits as needed(see Figure 5). The small output ca-pacitor requirements of the LT1118minimize turn-on time and the sup-ply current spikes that result fromcharging large filter capacitors.

1118_4.eps

5V

2.5V VIRTUAL GROUND

0.1µF 1µF

GND

LT1118-2.5

VIN VOUT


1118_5.eps

+V

VOUT 1 5V (FIRST ON/LAST OFF)

VOUT 2 5V (LAST ON FIRST OFF)

0.1µF

0.1µF 0.22µF

GND

LT1118-5

IN

EN

EN

OUT

0.1µF

0.01µF

0.22µF

GND

LT1118-5

IN OUT

10k

10kOFF

ON


LT1580 Low Dropout RegulatorUses New Approach toAchieve High PerformanceIntroduction

Low dropout regulators have be-come more common in desktopcomputer systems as microprocessormanufacturers have moved away from5V-only CPUs. A wide range of supplyrequirements exist today, with newvoltages just over the horizon. In manycases the input-output differential isvery small, effectively disqualifyingmany of the low dropout regulatorson the market today. Several manu-facturers have chosen to achieve lowerdropout by using PNP-based regula-tors. The drawbacks of this approachinclude much larger die size, inferiorline rejection, and poor transientresponse.

Enter the LT1580The new LT1580 NPN regulator is

designed to make use of the highersupply voltages already present inmost systems. The higher voltagesource is used to provide power forthe control circuitry and supply thedrive current to the NPN output tran-sistor. This allows the NPN to be driveninto saturation, thereby reducing thedropout voltage by a VBE compared toa conventional design. Applicationsfor the LT1580 include 3.3V to 2.5Vconversion with a 5V control supply,5V to 4.2V conversion with a 12Vcontrol supply, or 5V to 3.6V conver-sion with a 12V control supply. It iseasy to obtain dropout voltages aslow as 0.4V at 4A, along with excel-lent static and dynamic specifications.

The LT1580 is capable of 7A maxi-mum, with approximately 0.8Vinput-to-output differential. Thecurrent requirement for the controlvoltage source is approximately 1/50of the output load current, or about140mA for a 7A load. The LT1580presents no supply-sequencing is-sues. If the control voltage comes up

first, the regulator will not try to sup-ply the full load demand from thissource. The control voltage must beat least 1V greater than the output toobtain optimum performance. Foradjustable regulators the adjust-pincurrent is approximately 60µA andvaries directly with absolute tempera-ture. In fixed regulators the groundpin current is about 10mA and staysessentially constant as a function ofload. Transient response performanceis similar to that of the LT1584fast-transient-response regulator.Maximum input voltage from themain power source is 7V, and theabsolute maximum control voltage is14V. The part is fully protected fromovercurrent and over-temperatureconditions. Both fixed voltage andadjustable versions are available. Theadjustables are packaged in 5-pinTO-220s, whereas the fixed voltageparts are 7-pin TO-220s.

Figure 1. LT1580 delivers 2.5V from 3.3V at up to 6A

The LT1580 BringsMany New Features

Why so many pins? The LT1580includes several innovative featuresthat require additional pins. Both thefixed and adjustable versions haveremote-sense pins, permitting veryaccurate regulation of output voltageat the load, where it counts, ratherthan at the regulator. As a result, thetypical load regulation over a range of100mA to 7A with a 2.5V output isapproximately 1mV. The sense pinand the control-voltage pin, plus theconventional three pins of an LDOregulator, give a pin count of five forthe adjustable design. The fixed-voltage part adds a ground pin forthe bottom of the internal feedbackdivider, bringing the pin count to six.The seventh pin is a no-connect.

Note that the adjust pin is broughtout even on the fixed-voltage parts.This allows the user to greatlyimprove the dynamic response of theregulator by bypassing the feedback

DESIGN FEATURES

1580_1.eps

+

+

+

+ C2 220µF 10V

VIN

1

3

2VCC

VSS

4VCONT53.3V

5V

RTN

SENSE

VOUTADJ

R1 110Ω 1%

VOUT = 2.5V

U1 LT1580

C3 22µF 25V

C4 0.33µF

R2 110Ω 1%

C1 100µF

10V

100µF 10V 2X

1µF 25V 10X

MICROPROCESSOR SOCKET

by Craig Varga


temperature, guaranteed, whileoperating with an input/output dif-ferential of well under 1V.

In some cases a higher supply volt-age for the control voltage will not beavailable. If the control pin is tied tothe main supply, the regulator willstill function as a conventional LDOand offer a dropout specificationapproximately 70mV better thanconventional NPN-based LDOs. Thisis the result of eliminating the voltagedrop of the on-die connection to thecontrol circuit that exists in olderdesigns. This connection is now madeexternally, on the PC board, usingmuch larger conductors than arepossible on the die.

Circuit ExamplesFigure 1 shows a circuit designed

to deliver 2.5V from a 3.3V sourcewith 5V available for the control volt-age. Figure 2 shows the response to aload step of 200mA to 4.0A. The cir-cuit is configured with a 0.33µFadjust-pin bypass capacitor. The per-formance without this capacitor isshown in Figure 3. This difference inperformance is the reason for provid-ing the adjust pin on the fixed-voltage

Figure 4. Small FET adds shutdown capability to LT1580 circuit

divider with a capacitor. In the past,using a fixed regulator meant suffer-ing a loss of performance due to lackof such a bypass. A capacitor value of0.1µF to approximately 1µF will gen-erally provide optimum transientresponse. The value chosen dependson the amount of output capacitancein the system.

In addition to the enhancementsalready mentioned, the referenceaccuracy has been improved by afactor of two, with a guaranteed 0.5%tolerance. Temperature drift is alsovery well controlled. When combinedwith ratiometrically accurate inter-nal divider resistors, the part caneasily hold 1% output accuracy over

devices. A substantial savings inexpensive output decoupling capaci-tance may be realized by adding asmall “1206-case” ceramic capacitorat this pin.

Figure 4 shows an example of acircuit with shutdown capability. Byswitching the control voltage ratherthan the main supply, the transistorproviding the switch function needsonly a small fraction of the currenthandling ability that it would need ifit was switching the main supply.Also, in most applications, it is notnecessary to hold the voltage dropacross the controlling switch to a verylow level to maintain low dropoutperformance.

DESIGN FEATURES

1580_4.eps

+

+

+ C2 220µF 10V

VIN

1

3

2

4VCONT53.3V

5V

SHUTDOWN

RTN

SENSE

VOUTADJ

R1 110Ω 1%

VOUT = 2.5V

U1 LT1580

Q1 Si9407DY

C3 22µF 25V

C4 0.33µF

R2 110Ω 1%

R3 10k

LOAD

C1 100µF

10V

50µs/DIV

2A/DIV

50mV/DIV

2A/DIV

50mV/DIV

50µs/DIV

Figure 3. Transient response without adjust-pin bypass capacitor. Otherwise, conditionsare the same as in Figure 2

Figure 2. Transient response of Figure 1’scircuit with adjust-pin bypass capacitor. Loadstep is from 200mA to 4A


Power Factor Correction —Part Three: Discontinuity

In Part One of this series (in LinearTechnology V:1, pp. 17–18, 21), weinvestigated power factor correction(PFC) by looking at its line-frequencyvoltage, current, and power wave-forms. We developed the concept ofa DC variac and demonstrated itsutility for performing PFC.

In Part Two (LT V:2, pp. 3–7), wedeveloped the boost converter andinvestigated its properties when op-erating in the continuous-mode. Weshowed that a continuous-mode boostconverter does a pretty good job ofemulating the DC variac.

Part Three continues our investi-gation of the boost converter byfocusing on its properties when oper-ating in the discontinuous-mode. Wewill see that a discontinuous-modeboost converter doesn’t look anythinglike a DC variac.

There is elegance and simplicity inthe concept of the DC variac. There isbeauty in its application to PFC, abeauty that is manifest in the por-tions of line cycle where the boostconverter is operating in the continu-ous-mode.

InsubordinationWreaks Havoc

In analog engineering, we like toview things as continuous andsmooth, with all nonlinearities welldefined. We look at an AC sine waveand see the voltage go from positive,through zero, to negative with nodiscontinuity. This explains a goodportion of the genuine perplexity ex-perienced by the author when thebank practices alchemy on perfectlylegitimate checks—changing thepaper into rubber. One would think

the bank would know that it is onlytwo days until payday and the checkbook is still half full.

Research into the character ofbanking employees reveals the star-tling truth that the local bank isstaffed by the genetic precursor of thedigital engineer, a human so vile, sobase, as to actually enjoy insertingdiscontinuities into an otherwisecontinuous system. In the boost con-verter it is the output diode that actsas the banker.

The boost converter shown in Fig-ure 1a is the boost converter fromheaven. Its glory is directly attributedto its having a switch, rather than avile diode, in its output. The switchallows current flow in both direc-tions, and thus inserts no additionaldiscontinuities. The boost converterfrom heaven always operates in thecontinuous-mode. Back here on earthwe must live with discontinuities.

In a boost converter operating incontinuous-mode, the volt-seconds,and thus the current in the inductor,is entirely controlled by:

1. The input voltage2. The output voltage3. The duty factor.

The boost converter of Figure 1b isthe mortal version. The inductor cur-rent (I1) in the boost converter, drivenby the three factors above, goes tozero when the input line voltageswings through zero. Figure 2 showsthat the input line current is actuallythe average of the inductor currentover a full switching-frequency cycle.Figure 2a is the inductor current whenthe input line voltage has reached itspeak with the average value repre-senting the line current at the plug.(The ripple is filtered out by the inputEMI filter.) Similarly, Figure 2b showsthe inductor current and input cur-rent when the instantaneous inputline voltage is at 40% of its peakvalue. Finally, Figure 2c shows the

inductor current and input line cur-rent as the instantaneous input linevoltage is at 10% of its peak value.Notice that on each switching-frequency cycle, the inductor currentgoes to zero and then tries to reverse.When the inductor current attemptsto reverse, the diode commits thedisgraceful act of insubordination,destroying the beauty, simplicity, andcontinuity of the control function ofthe continuous-mode boost converter.Figure 2c also shows the area gaineddue to the clipping of the negativeportion of the inductor current. Thenet effect is that the average currentis higher than the current that thecontinuous-mode’s duty factor con-trol of this DC variac would haverequested. The very act of preventingthe inductor current from reversing,a hideous act indeed, necessitates anew set of models to describe thediscontinuous-mode operation of theboost converter. Further, a new con-trol law is required to control thediscontinuous-mode boost converter.

The Boundaryis a Moving Target

The boost converter could belikened to a fish tank, with the sur-face of the water representing theboundary of continuity, above whichthe boost converter operates in thecontinuous-mode and below which itoperates in the discontinuous-mode.Folks living outside the fish tank tendto obtain their oxygen from the air,whereas folks living inside the fishtank tend to get their oxygen from thewater. Air folk and water folk havegreat trouble communicating, as

by Dale Eagar

Figure 1a. The boost converter from heaven Figure 1b. Mortal boost converter

DESIGN FEATURES

pfc3_1a.eps

LOADI1

L1 SW1 OFF

ONE

pfc3_1b.eps

LOAD

I1

L1

E SW1

D1


neither is equipped to speak, let alonebreathe, in the other’s medium. Thislack of communication has had asignificant effect on the way the twogroups think.

When air folk look at controllingtheir boost converters, they speak ofduty-factor, the all important con-trolling entity. Water folk control theirboost converters with on-time. We,looking at the whole system from theoutside, know that the switching-frequency is constant, and thus on-time and duty-factor are describingthe same thing.

When air folk plot duty-factor, theysee a smooth curve extending down tothe surface of the water, and thenbeing refracted at the surface to adifferent slope as it goes further downto the bottom of the tank. The air folkmathematicians have developedequations that predict the duty-factor plot, equations that exactlydescribe everything it does in the air.The air folk equations are incapableof predicting what happens when theplot is below the water. The water folkmathematicians have the exact sameproblem in reverse. We the onlookers

can inspect the equations, and whenwe do, we see that the respectiveequations have no hope of beingconsolidated into one. The air folkequation states that the duty-factoris a function of instantaneous inputand output voltage and the rate ofchange in the current flow, but isaltogether independent of the steady-state value of the current flow,the operating frequency, and theinductor’s inductance value.

The water folk equation states thatthe on-time is a function of the steady-state current flow, the inductance ofthe inductor, the switching frequency,and the input voltage, and is inde-pendent of the output voltage and therate of change in current flow. Fortu-nately for all of us, both equationsconverge on the same point at thewater’s surface.

Air folks use volt-seconds to de-scribe the stretch in their inductors,and water folk use joules to describethe energy stored in their inductors.As onlookers, we can convert betweenJoules and volt-seconds by using thefollowing equations:

To further complicate matters, theindex of refraction of the waterchanges and is directly related to thewater level. The water level changesdue to input voltage, output voltage,switching frequency, output current,and inductance of the inductor.

Memory LostIn continuous-mode, the choke

current, and thus power interceptedfrom the input line, depends on whatthe choke current was during theprevious cycle. Evaluation of con-tinuous-mode boost is best performedby looking at the overall action of anintegral number of switching cyclesof a running converter.

In discontinuous-mode, the chokecurrent goes to zero during each

J = 12L

(V • S)2

2LJV • S =

J = Energy in Joules L = Inductance in Henries V • S = Stretch in Volt Seconds


DESIGN FEATURES

SWITCH ON

SWITCH OFF

ACTIVE OFF

TIME

AVE

I1

0 0 0

AVE

INACTIVE OFF

TIME

AVE

DIODE COMMUTATES AREA GAINED

pfc3_2.eps(2a) (2b) (2c)

Figure 2. Choke-current waveforms


The New LT1508/LT1509 CombinesPower Factor Correction and a PWMin a Single PackageIntroduction

When a supply requires power fac-tor correction (PFC), designerscommonly add a PFC controller, suchas the LT1248, to the front end of atraditional off-line DC/DC converter.Although this is a viable solution,additional circuitry is usually re-quired to ensure that the twocontrollers work well together. TheLT1508/LT1509 eliminate the needfor the additional circuitry and PWMby integrating the functionality of aLT1248 power factor controller and a50% maximum-duty-cycle PWM in asingle 20-pin DIP or SOIC. The LT1508retains all the pins of the LT1248except the EN/SYNC, and adds quiet-ground, soft-start, control-voltage,current-limit and gate-driver pinsto form a voltage-mode PWM (seeFigure 1). In the LT1509, the currentlimit becomes the ramp pin (PWMcurrent sense), forming a current-mode PWM (see Figure 2).

Full Featured PFCAll features of the LT1248 remain,

requiring no compromises whenimplementing PFC with the LT1508/LT1509. Continuous-boost averagecurrent mode reduces magnetics sizewhile operating at a fixed frequency,without the need for slope compensa-tion. Built-in overvoltage protectionreduces the external parts count.Current amplifier offset voltage is dealtwith internally, minimizing crossoverdistortion. Full differential input cur-rent sense is available, as are 1.5Agate drivers for both the PFC andPWM stages. In addition to thesefamiliar performance features thereare a few new ones incorporated inthe PWM section.

Hand-HoldingWhether the application involves

voltage mode or current mode con-trol, smooth interaction between thePFC and PWM stages requires somehand-holding. The first thing toconsider is start-up. The PFC pre-regulator turns on with soft start asusual. During this period the PWM’ssoft-start pin is held low, preventingthe second stage converter from turn-ing on until the intermediate busvoltage reaches its preset value (typi-cally 380VDC to 400VDC for auniversal input). In addition to delay-ing the PWM soft start during start-up,the same comparator employs hys-teresis to shut down the PWM stage inthe event the intermediate bus volt-age drops below 73% of the presetvoltage (typically 280VDC). Once en-abled, the PWM runs somewhatindependently of the PFC controller,with a couple of exceptions. The oscil-lator frequency is set by the values ofthe capacitor and resistor to groundon the CSET and RSET pins. Both thePFC and PWM stages are synchro-nized to this oscillator and operate atthe same switching frequency. Syn-chronization is accomplished bydelaying the turn-on of the PWM stagethrough 50% of the switching period.In addition to making oscilloscopemeasurements easier, these featuresensure that both switches don’t turnon simultaneously, causing bus volt-age spikes, noise and problems withcurrent mode in the PWM stage.Single frequency and synchroniza-tion also ensure the absence of beatfrequencies in the conducted emis-sions. A preset duty cycle limit of50% and leading-edge blanking fur-ther ease the design of the PWMsection. The PWM stage is quicklydisabled by pulling the controlvoltage low. Finally, the LT1509’s

current mode comparator has a 1.2VDC offset to accommodate opto-feed-back, and both the LT1508 andLT1509 contain independent over-current comparators.

Typical ApplicationFigure 3 shows a 24VDC, 300W

power-factor-corrected, universal-input supply. The continuous,current mode boost PFC preregulatorminimizes the differential-mode in-put filter size required to meetEuropean low-frequency conductedemission standards while providinga high power factor. The 2-transistorforward converter offers many ben-efits including low peak currents, anon-dissipative snubber, 500VDCswitches and automatic core resetguaranteed by the LT1509’s 50%maximum duty-cycle limitation.An LT1431 and inexpensive opto-isolation is used to close the loopconservatively at 3kHz with excessmargin (see Figure 4). Figure 5 showsthe output voltage’s response to a 2Ato almost 10A current step. Regula-tion is maintained to within 0.5V.Efficiency curves for output powersof 30, 150, and 300W are shownin Figure 6. The PFC preregulatoralone has efficiency numbers of be-tween about 87% and 97% over lineand load.

Start up of the circuit begins withthe LT1509’s VCC bypass capacitorstrickle charging through 91kΩto 16VDC, overcoming the chip’s.25mA typical start-up current (VCC ≤lockout voltage). PFC soft start isthen released, bringing up the382VDC bus with minimal overshoot.As the bus voltage reaches its finalvalue, the forward converter comesup, powering the LT1431, andclosing the feedback loop. A 3-turn

by Kurk Mathews

DESIGN FEATURES


Figure 1. Block diagram: LT1508

DESIGN FEATURES

+ –– +

– +– +

– + + –

– +

+ – I M =

I A2 I

B

200µ

A2

11 16

13

+ –

– +

415

19

1815

08_1

.eps

V SEN

SE

7.5V

7.9V

V CC

16V

TO 1

0V

12µA

12µA

7µA

7.0V

4.

7V

EACA

CL

2.2V

M1

I AI M

I B25

k

1V

14

VAOU

T

10

V REF

7.5V

V R

EF

12

MOU

T

8

I SEN

SE 7

CAOU

T

6

C SET

R SET

I LIM

IT

V C

50µA

0.7V

RUN

RUN

R 2R

R RQ

S

OSC

55%

DE

LAY

150n

s BL

ANKI

NG

PKLI

M

5GN

D1 3

V CC

17

GTDR

1

16V

16V

1

GND2 2

GTDR

220

I AC 9 OVP

SS1

SS2

PWM

OK

+ –

+ –R S

Q

R



Figure 2. Block diagram: LT1509

DESIGN FEATURES

+ –– +

– +– +

– +

+ –

– +

+ – I M =

I A2 I

B

200µ

A2

11 16

13

+ –

– +

415

1918

1508

_2.e

ps

V SEN

SE

7.5V

7.9V

V CC

16V

TO 1

0V

12µA

12µA

7µA

7.0V

4.

7V

EACA

CL

2.2V

M1

I AI M

I B25

k

1V

1.2V

14

VAOU

T

10

V REF

7.5V

V R

EF

12

MOU

T

8

I SEN

SE 7

CAOU

T

6

C SET

R SET

RAM

PV C

50µA

0.7V

RUN

RUN

R RQ

S

OSC

55%

DE

LAY

150n

s BL

ANKI

NG

PKLI

M

5GN

D1 3

V CC

17

GTDR

1

16V

16V

1

GND2 2

GTDR

220

I AC 9 OVP

SS1

SS2

PWM

0K

+ –

+ –

+–

R SQ

R


Figure 3. Schematic diagram of 300W, 24V DC output, power-factor corrected, universal-input supply

DESIGN FEATURES

+–

+–

–+

–+

1508

_3.e

ps

V SEN

SE

VAOU

TV R

EFPK

LIM

MOU

TI S

ENSE

V REF

V REF

V CC

V IN

OVP

IAC

9

2

1114

1012

58

7

CAOU

TGT

DR1

GTDR

2

1.2V

LT14

31

61

20 19

20k

1%

4.02

k 1%

4.02

k 1%

499k

1%

499k

1%

499k

1%

382V

BUS 20k

1%

15k

1%

1k

220Ω

20k

2N29

072.

2k

330Ω

20k

20Ω

IRF8

40

2N22

22A

2N22

22A

220Ω

10Ω

15V,

1N

965

(× 2

)20

k

IRF8

40

MUR

450

MUR

450

499k

1%

0.00

47µF

0.1

“X”

0.00

1µF

0.04

7µF

0.01

µF

0.47

µF33

0k

GI

KBU4

J0.

15, 3

W

FUKU

SHIM

A M

PC71

10k

1.8k

470µ

20k

2.2k

10Ω

2W

1k

13

1N58

19

1N58

19

GND2

0.00

1µF

0.1µ

F

4700

pF

“Y”

4700

pF

“Y”

1000

pF

100p

F

1µF

20Ω

20Ω

0.00

1µF

3

GND1

LT15

09

4

C SET

1µF

13

SS2

0.04

7µF

0.02

2µF

COLL

62

5

GNDF

GNDS

COM

P3.

4k

1%

1µF

63V

FILM

V+4

24V O

UT,

12.5

AM

PS

8 7

RTOP

30.1

k 1%

10Ω

2W

2000

pF

REF

RMD

OUTP

UT C

OM

100p

F

100p

F

RAM

P

CNY1

7-3

200µ

F

16

15V

SS1

17

V C

1815

R SET

1µF

2.2µ

F 50

V33

0µF

35V

2.2µ

F 50

V

10k

1%24.9

k 1%

V REF

470µ

F, 5

0V

NICH

ICON

PL

12.5

X25

(× 3

)

67µH

39

T 1

2AW

G T1

50-5

2

T2 7

TUR

NS

0.9"

× 0

.005

" CU

ETD4

4-P

LPRI

= 3

.1m

H

NOTE

: UNL

ESS

OTHE

RWIS

E SP

ECIF

IED

1. A

LL R

ESIS

TORS

1/4

W, 5

%

2. A

LL C

APAC

ITAN

CE V

ALUE

S IN

MIC

ROFA

RADS

GI

FEP

30DP

1M

1/2W

90 T

O 26

4 VA

C0.

1 “X

”

4700

pF

“Y”

KETE

MA

SG57

SU

RGE-

GARD

4700

pF

“Y”

6A

FAST

0.1

“X”

20k

1µF

400V

EL

ECTR

ONIC

CO

NCEP

TS

5MP1

2J10

5K

180µ

F 45

0V

91k

2W

V IN

382V

BUS

T1

COIL

TRON

ICS

CTX0

2-12

378-

2

1000

pF

“Y”

ERA8

2-00

4

FUJI

ER

A82-

004

0.6A

MP/

40VR

10 :

15

TURN

S 28

AWG

846T

500-

3E2A

T2

35 T

URNS

29

AWG

× 4 T2

35

TUR

NS

29A

WG

× 4

0.51

, 2W

RG

ALL

EN

RFS2

(×

2)

MUR

H860

CT15

V

ERA8

2-00

4

T1

IRF8

40

+

+

+

+

+

+


100µs/DIV

DESIGN FEATURES

FREQUENCY (Hz)

–40

0

–20

40

20

80

60

LOOP

GAI

N (d

B)

0

30

15

60

45

90

75

PHAS

E M

ARGI

N (D

EGRE

ES)

1508_4.eps

10 100 1K 100K10K

100µs/DIV

5A/DIV

0.5V/DIV

VIN (AC)

70

75

80

85

90

EFFI

CIEN

CY (%

)

1508_6.eps

100 150 200 300250

300W

150W

30W

1118, continued from page 6

The enable pin provides many otheruseful capabilities to the resourcefuldesigner. Figure 5 shows two LT1118-5 regulators configured to providesequential turn-on and turn-off fromlogic control. A logic high level turnson the first regulator; the second regu-

lator turns on only after the firstoutput voltage is stable. Turn-off is inthe reverse order, due to the rapiddischarge of capacitor C3 throughthe diode.

These and other applications willbenefit from the LT1118 family’s

unique performance attributes. Anyapplication that requires fast tran-sient load response, unconditionalstability under capacitive loading,or logic control of the power outputis a potential application for theLT1118 family.

Figure 4. Bode plot of the circuit shown inFigure 3

Figure 5. Figure 3’s response to a 2A toapproximately 10A load step

Figure 6. Efficiency curves for Figure 3’scircuit

secondary added to the 70-turn pri-mary of T1, bootstraps VCC to about15VDC, supplying the chip’s 13mArequirement as well as about 25mA tocover the gate current of the threeFETs and high-side transformerlosses. A 0.15Ω sense resistor sensesinput current and compares it to areference current (IM) created by theouter voltage loop and multiplier. Thusthe input current follows the inputline voltage and changes, as neces-sary, in order to maintain a constant

bank voltage. The forward convertersees a voltage input of 382VDC un-less the line voltage drops out, inwhich case the 180µF main capacitordischarges to 250VDC before the PWMstage is shut down. Compared to atypical off-line converter, the effectiveinput voltage range of the forwardconverter is smaller, simplifying thedesign. Additionally, the higher busvoltage provides greater hold-up timesfor a given capacitor size. The high-side transformer effectively delays the

turn-on spike to the end of the built-in blanking time, necessitating theexternal blanking transistor.

Conclusion The LT1508 and LT1509 offer

complete solutions for low-to-highpower, isolated, off-line power-factor-corrected supplies. The manybuilt-in features ensure a simplified,low-parts-count design for a varietyof applications.

PFC, continued from page 10

off-time. This allows us to model thediscontinuous-mode boost converteras a sum of discrete events. Eachevent consists of an on-time and anoff-time, with the off-time being longerthan the time it takes to empty allstored energy from the inductor.

Figure 2c shows that the off-timeconsists of two separate times:

the time when the diode isconducting, the active off-time.

the time after the diode hascommitted, yet before the nexton-time, the inactive-off-time.

The Energy BucketWe look at a discontinuous-mode

boost converter as an energy bucketthat gets filled and dumped with eachcycle. During each cycle the buckettakes a certain amount of energy (J).Over a given number of cycles thebucket will intercept many buckets-ful of energy and deliver them to theload. A boost converter operating at aswitching frequency of f and inter-cepting an amount of energy of Jduring each cycle, will be interceptingf × J watts of power from the input

power line. Because this converteris very efficient, essentially all ofthe intercepted power is delivered tothe load.

This concludes part three of ourseries introducing the power factorcorrection conditioner.


LTC1329 Micropower,8-Bit, Current Output DAC by K.S. Yap

DESIGN FEATURES

The LTC1329 is a micropower,8-bit, current output DAC thatprovides precision current output overa wide supply range. This part is idealfor portable and battery-poweredapplications.

DescriptionThe LTC1329 can communicate

with external circuitry by means ofone of three interface modes: stan-dard, 3-wire serial-data mode, 1-wirepulse mode, and 2-wire pulse mode.In serial-data mode, the system mi-croprocessor can serially transfer the8-bit data to and from the LTC1329.In pulse mode, the upper 6 bits ofDAC output can be programmed forincrement-only (1-wire interface) orincrement/decrement (2-wire inter-face) operation. In increment-onlymode, when the count increases be-yond full scale, the counter rolls overand sets the DAC to zero. In incre-ment/decrement mode, the counterstops incrementing at full scale andstops decrementing at zero, and willnot roll over.

The LTC1329 can operate over awide supply range from 2.5V to 6.5V,with a very low quiescent current ofonly 50µA, which drops to only 0.2µAin shutdown mode. The DAC currentoutput can be biased from −20V to+2.5V. The full-scale DAC currentoutput is 10µA ±2.5% and 50µA±2.5% for the −10 and −50 versionsrespectively. On power-up, the DACcurrent output is automatically setat mid-range.

The LTC1329’s micropower opera-tion makes it ideal for portable andbattery-powered applications likeLCD contrast control, backlightbrightness control, power supplyvoltage adjustment, battery chargervoltage/current adjustment, and trimpot replacement. The LTC1329 isavailable in 8-pin surface mount andDIP packages. Figure 2. LTC1329 used to null op amp’s offset voltage

+

–

IOUT

VOUT

VCC

VCC = 3.3V

SHDN

CLK

P1.0

1

2

3

4

DOUT

DIN

VIN

GND

CS

LTC1329-50

LT1006

V–

RI

R1 =

TRIM RANGE = ±(0.5) (IFULL SCALE) (R2)(0.5) (IFULL SCALE)

V–

1329_2.eps

R2 100Ω

RF

R1 600k

MPU (e.g. 8051)

8

7

6

5

Power SupplyVoltage Adjustment

Figure 1 is a schematic of a digi-tally controlled power supply voltageadjustment circuit using a 2-wireinterface. The LT1107 is configuredas a step-up DC/DC converter, withthe output voltage (VOUT) determinedby the values of the feedback resis-tors. The LTC1329’s DAC currentoutput is connected to the feedbacknode of these resistors, and an 8051microprocessor is used to interface tothe LTC1329. By simply clocking theLTC1329, the DAC current output isdecreased or increased (decreased ifDIN = 0, increased if DIN = 1), causingVOUT to change accordingly.

Trimmer Pot ReplacementFigure 2 is a schematic of a digitally

controlled offset-voltage adjustmentcircuit using a 1-wire interface. Bysimply clocking the LTC1329, the DACcurrent output is increased, causingVR2 to increase accordingly. When theDAC current output reaches full scale,it will roll over to zero, causing VR2 tochange from the maximum offset trimvoltage to the minimum offset trimvoltage.

Figure 1. LTC1329 digitally controls the output voltage of a power supply

ILIM VIN

GND SW2

SW1

VIN 2V TO 3V

VOUT 4V TO 6V 20mA

IOUT

VCC

VCC = 3.3V

SHDN

CLK

P1.0

P1.1

1

2

3

4

DOUT

DIN

GND

CS

FB

LT1107

LTC1329-50

100µF

L1* 100µH 1N5817

47Ω

*L1 = COILTRONICS CTX100-4

1329_1.eps

39k

10k

+

47µF+ MPU

(e.g. 8051)

8

7

6

5


DESIGN FEATURES

sag, possibly tripping the low-batterydetector. Output voltage reaches 5Vin approximately 1ms. The additionof R1 and C1 to Figure 2’s circuitlimits inrush current at start-up, pro-viding for a smoother turn-on, asindicated in Figure 6.

A 4-Cell-to-5V ConverterA 4-cell-to-5V converter is more

complex than a simple boost con-verter because the input voltage canbe either above or below the outputvoltage. The single-ended primary in-ductance converter (SEPIC) shown inFigure 7 accomplishes this task, withthe additional benefit of output isola-tion. In shutdown conditions, theconverter’s output will go to zero,unlike the simple boost converter,where a DC path from input to outputthrough the inductor and diode re-mains. In this circuit, peak current islimited to approximately 500mA bythe addition of 22k resistor R1. Thisallows very small, low-profile compo-nents to be used. The 100µFcapacitors are D-case size, with aheight of 2.9mm, and the inductorsare 3.2mm high. The circuit can de-liver 5V at up to 100mA. Efficiency isrelatively flat across the 1mA-to-100mA load range.

Super BurstTM Mode Operation:5V/100mA DC/DC with 15µAQuiescent Current

The LT1304’s low-battery detectorcan be used to control the DC/DCconverter. The result is a reduction inquiescent current by almost an orderof magnitude. Figure 9 details thisSuper Burst circuit. VOUT is moni-tored by the LT1304’s LBI pin viaresistor divider R1/R2. When LBI isabove 1.2V, LBO is high, forcing theLT1304 into shutdown mode and re-ducing current drain from the batteryto 10µA. When VOUT decreases enoughto overcome the low battery detector’shysteresis (about 35mV), LBO goeslow. Q1 turns on, pulling SHDN highand turning on the rest of the IC. R3limits peak current to 500mA; it canbe removed for higher output power.Efficiency is illustrated in Figure 10.

current. When R1’s voltage drop ex-ceeds 36mV, corresponding to 1Aswitch current, A2’s output goes high,truncating the on-time part of theswitch cycle. The 1A peak current canbe reduced by tying a resistor be-tween the ILIM pin and ground, causinga voltage drop to appear across R2.The drop offsets some of the 36mVreference voltage, lowering peakcurrent. A 22k resistor limits currentto approximately 550mA. A capacitorconnected between ILIM and groundprovides soft start. Shutdown isaccomplished by grounding theSHDN pin.

The low-battery detector A3 has itsown 1.2V reference and is always on.The open collector output device cansink up to 500µA. Approximately35mV of hysteresis is built into A3 toreduce “buzzing” as the battery volt-age reaches the trip level.

A 2-Cell-to-5V ConverterA compact 2-cell-to-5V converter

can be constructed using the circuitof Figure 2. Using the LT1304-5 fixed-

output device eliminates the need forexternal voltage-setting resistors, low-ering component count. As the batteryvoltage drops, the circuit continuesto function until the LT1304’s under-voltage lockout disables the part atapproximately VIN = 1.5V. 200mA isavailable at a battery voltage of 2.0V.As the battery voltage decreases be-low 2V, cell impedance starts toquickly increase. End-of-life is usu-ally assumed to be around 1.8V, or0.9V per cell. Efficiency is detailed inFigure 3. Burst Mode micropoweroperation keeps efficiency above 70%,even for load current below 1mA.Efficiency reaches 85% for a 3.3Vinput. Load transient response is il-lustrated in Figure 4. Since theLT1304 uses a hysteretic comparatorin place of the traditional linear feed-back loop, the circuit respondsimmediately to changes in load cur-rent. Figure 5 details start-up behaviorwithout soft-start circuitry (R1 andC1 in Figure 2). Input current rises to1A as the device is turned on, whichcan cause the input supply voltage to

LT1304, continued from page 1

–

+

1304_1.eps

–

+

–

+

1.2V

R4

36mV

A3

OFF

LBI

ENABLE

A2

LB0

C1 L1

SW

GNDILIMSHDN

FBA1

8

6 5

4VIN

VOUT

VIN

3

7

1

1.5V UNDERVOLTAGE

LOCKOUT

Q3

Q2 1x

Q1 200x

R2 1k

R3

R1 7.2Ω

1k

DRIVER

BIAS ~1V

2

1.24V VREF SHUTDOWN

+

C2

D1

+

TIMERS 6µs ON

1.5µs OFF

Figure 1. LT1304 Block diagram. Independent low-battery detector A3 remains alive whendevice is in shutdown


Figure 2. 2-cell-to-5V/200mA boost converter takes four external parts. Components withdashed lines are for soft-start (optional)

Figure 3. 2-cell-to-5V converter efficiency

Figure 6. Start-up response with 1µF/1MΩcomponents in Figure 2 added. Input currentis more controlled. VOUT reaches 5V in 6ms.Output drives 20mA load

Figure 7. 4-cell-to-5V step-up/step-down converter, also know as SEPIC (single-ended primaryinductance converter). Low profile components are used throughout

DESIGN FEATURES

VIN SW

GND IL

SENSELBI

SHDNLB0

LT1304-5

100µF

C1 1µF

100µF2 CELLS

22µH* MBRS130L

R1 1M

5V 200mA

SHUTDOWN

*SUMIDA CD54-220 (708) 956-0666

1304_2.eps

+

+

+

LOAD CURRENT (mA)

40

50

60

70

80

90

EFFI

CIEN

CY (%

)

1304_3.eps

0.1 1 10 500100

VIN = 1.8V

VIN = 3.3V

VIN = 2.5V

VIN SW

IL GND

SENSELBI

SHDNLB0

LT1304-5

100µF

47µF4 CELLS

3.5V TO 6.5V 22µH* 100µF

22µH*

MBR0530

R1 22k

5V 100mA

SHUTDOWN

*SUMIDA CD43-220 (708) 956-0666

1304_7.eps

+

+

+

LOAD CURRENT (mA)

50

85

80

75

70

65

60

55

EFFI

CIEN

CY (%

)

100

1304_8.eps

1 10

VIN = 5VVIN = 6V

VIN = 3V

VIN = 4V

Figure 8. Efficiency plot of SEPIC convertershown in Figure 7

VOUT 100mV/DIVAC COUPLED

ILOAD

100µs/DIV

200mA0

Figure 4. Boost converter load-transientresponse with VIN = 2.2V

1ms/DIV

VOUT 2V/DIV

IIN500mA/DIV

VSHDN10V/DIV

VOUT 2V/DIV

1ms/DIV

IIN500mA/DIV

VSHDN10V/DIV

An output capacitor charging cycleor “burst” is shown in Figure 11, withthe circuit driving a 50mA load. Theslow response of the low-battery de-tector results in the high number ofindividual switch cycles or “hits”within the burst.

Figure 12 depicts output voltage atthe modest load of 100µA. The burst

The converter is approximately 70%efficient at a 100µA load, 20 pointshigher than the circuit of Figure 2.Even at a 10µA load, efficiency is inthe 40%-50% range, equivalent to100µW–120µW total power drain fromthe battery. In contrast, Figure 2’scircuit consumes approximately300µW–400µW unloaded.

repetition rate is around 4Hz. Withthe load removed, the repetition ratedrops to approximately 0.2Hz, or oneburst every 5 seconds. Systems thatspend a high percentage of operatingtime in sleep mode can benefit fromthe greatly reduced quiescent powerdrain of Figure 9’s circuit.

Figure 5. Start-up response. Input currentrises quickly to 1A. VOUT reaches 5V inapproximately 1ms. Output drives 20mA load


DESIGN FEATURES

VIN SW

SHDN GND

LBI

Q1 2N3906

LB02

2

1

6

3 4

7 5

ILFB

LT1304

100µF2 CELLS

IQ ≈ 15µA 33µH* MBR0530

47k

5V 80mA

1304_9.eps

22k

R2 1.21M

R1 3.83M

47k

0.01µF

330µF

200k

*SUMIDA CD54-330 (708) 956-0666

+

+

Figure 10. DC/DC converter efficiency ofFigure 9

LOAD CURRENT (mA)

40

50

70

60

80

90

EFFI

CIEN

CY (%

)

100

1304_10.eps

0.01 0.1 1.0 10

VIN = 3V

VIN = 2V

VSW 1V/DIV

5ns/DIV

50ms/DIV


50ms/DIV


50µs/DIV

IL1A/DIV


VSW5V/DIV

Figure 11. Super Burst Mode Operationin action. VIN = 2.5V, ILOAD = 50mA

Figure 12. Circuit using Super BurstMode Operation, 100µA load. Burstoccurs approximately once every 240ms

Figure 15. Switch fall time. Lower slope insecond and third graticules shows effect oflead and bond wire inductance

LayoutThe LT1304 switch turns on and

off very quickly. For best performancewe suggest the component placementin Figure 13. Improper layouts willresult in poor load regulation, espe-cially at heavy loads. Parasitic leadinductance must be kept low forproper operation. Switch turn-off isdetailed in Figure 141. A close look atthe rise time (5ns) will confirm theneed for good PC board layout. The200MHz ringing of the switch voltage

is attributable to lead inductance,switch and diode capacitance, anddiode turn-on time. Switch turn-on isshown in Figure 15. Transition timeis similar to that of Figure 14. Adher-ence to the layout suggestions willresult in working DC/DC converterswith a minimum of trouble.

1. Instrumentation for oscillographs of Figures 14and 15 include Tektronix P6032 active probe,Type 1S1 sampling unit and type 547 mainframe.

1304_13.eps

8

7

6

54

3

2

1

LT1304

COUT

SHUTDOWN

VOUT

VIN

+

CIN+

GND (BATTERY AND LOAD RETURN)

Figure 13. Suggested layout for best performance. Input capacitor placement as shown ishighly recommended. Switch trace (pin 4) copper area is minimized

Figure 9. 2-cell-to-5V DC/DC converter using Super Burst Mode Operation draws only15µA unloaded. Two AA Alkaline cells will last for years

Figure 14. LT1304 Switch rise time is in the5ns range. These types of edges emphasizethe need for proper PC board layout


0.7ns < TRISE < 1.0ns

TIME (ns)

0

8

16

24

32

AMPS

ESD_2.eps

–20 20 60 100 140 180 220

IEC-801-2 HUMAN BODY MODEL

RS232 Transceiver Protected to±15kV IEC-801-2 ESD Standards

by Gary Maulding

Linear Technology’s popularLT1137A RS232 transceiver hasbeen upgraded to include on-chipESD protection compliant withIEC-801-2 level 4 standards. Devicepins connected to the RS232 line canabsorb up to ±15kV air-gap or ±8kVcontact-mode transients using theIEC-801-2 ESD model, with nodamage or interruption of operation.The 28-lead SOIC or SSOP packagedcircuit and five inexpensive ceramiccapacitors are the only componentsrequired to implement a 5V poweredAT serial port that is fully compliantwith RS232 or V.28 electricalspecifications and IEC-801-2 static-discharge standards.

ESD Effects onElectronic Equipment

Electrostatic discharge (ESD) is awell known killer of semiconductordevices. Unless protective measuresare taken, ESD will result in variousforms of device degradation. Dis-charge damage ranges from increasedleakage currents on device inputsand outputs, to increased offset orshifted threshold voltages, to failureof the affected circuit.

To minimize the monetary loss dueto damaged IC’s and to improve thereliability of equipment, the electron-ics industry has adopted manystandard safeguard proceduresagainst ESD. These safeguards in-clude the use of grounded worksurfaces, equipment, and personnel;

conductive plastic shipping materi-als, on-chip protection circuits at theinput and output pins, and testing ofthe ESD sensitivity of circuits.

The “Human Body Model,” definedby Mil-Std-883B Method 3015, is acommonly used ESD test standardfor rating integrated circuits for ESD-damage immunity. This test methodis aimed directly at reducing the inci-dence of failed devices installed onequipment circuit boards. The testdischarges a charged 100pF capaci-tor through a 1.5kΩ resistance directlyto the IC’s pins. Both positive andnegative voltages are tested. Rarely isthe device powered when testing isconducted, and only permanent dam-age to the circuit is considered.

The widespread adoption of envi-ronmental static control and thedesign of integrated circuits withon-chip protection devices has dra-matically reduced the incidence ofcircuit failures due to ESD damageduring circuit manufacturing andhandling. Most semiconductor de-vices enter a much safer environmentonce installed in a PC board. Installa-tion of the PC board into a chassisfurther shields the IC from the envi-ronment. No longer are most ICs’input and output pins exposed todirect ESD strikes, but ESD is still anoperational and reliability hazard forelectronic equipment. Everyone hasexperienced the shock caused bywalking across a nylon carpet on adry winter day and touching a piece of

electrical equipment. Thisstatic discharge can dam-age internal circuitry ofthe equipment if themanufacturer does notprovide a safe dischargepath for the energy. In-put/output ports onpersonal computers areespecially vulnerable. The

ports are available so that users canattach cabling to printers, mice, ornetworks. Portable computers areparticularly vulnerable since the portconnections are made and brokenmuch more often than for desktopmachines. Most consumers would notlike to wear static control groundstraps when using their personal com-puters. Hence, it is important thatESD protection be built in to the enduser equipment.

What is IEC-801-2?The International Electrotechnical

Commission (IEC) has defined a testand rating standard for electronicequipment exposed to ESD transients.IEC-801-2 defines two alternate testmethodologies. The preferred methodis a contact-mode test, but the alter-nate air-gap test is more easilyimplemented and is more commonlyused. Equipment certified for IEC-801-2 testing consists of a chargingcircuit to apply the desired test volt-age to a 150pF capacitor. A dischargeswitch discharges the energy into theequipment under test through a 330Ωresistance (see Figure 1). IEC-801-2certified test equipment must be ableto demonstrate that the applied test

150pF

ESD_1.eps

DEVICE UNDER TEST

HIGH VOLTAGE SUPPLY

330Ω

Figure 1. IEC-801-2 ESD test system Figure 2. ESD Transient current waveforms

DESIGN INFORMATION


voltage and current waveform meetthe prescribed rise and decay wave-shape shown in Figure 2. LinearTechnology uses a Keytek MiniZapfor IEC-801-2 ESD testing.

The test results on equipment areclassified by the voltage level suc-cessfully applied to the equipmentand the affect of test discharges onthe equipment. The IEC-801-2standard defines the following per-formance criteria:

1. Normal performance withinspecification limits

2. Temporary degradation or loss offunction or performance that isself-recoverable

3. Temporary degradation or loss offunction or performance thatrequires operator intervention orsystem reset

4. Degradation or loss of functionthat is not recoverable, due todamage of equipment (compo-nents) or software, or loss of data

Table 1 details the rated test levelsfor both contact and air-gap tests.The contact-mode test is thepreferred methodology due to itssuperior repeatability. The air-gaptest may be influenced by the testenvironment’s relative humidity, aswell as by the approach speed of thetester tip to the device under test.

LT1137A ESDTest Performance

The LT1137A is an RS232 inter-face transceiver with an integralcharge pump for single 5V operation.The complement of three drivers andfive receivers matches the require-ments for AT serial ports on personalcomputers. The eight RS232 lines aredirectly connected to the socket onthe computer’s case, and are there-fore directly exposed to possible ESDtransients when cables are installedor removed from the machine. Theinclusion of rugged ESD protectionon the chip eliminates the need forthe user to install costly and space-consuming transient suppressiondiodes on the RS232 lines.

The new, upgraded LT1137A hasbeen tested to level 4 IEC-801-2 test-ing. In both the ±8kV contact-modetest and the ±15kV air-gap test, thetransceiver suffered no damage andno interruption of normal operation.

Linear Technology provides thisenhanced ESD protection on theLT1137A at no extra cost to the user.The improved level of protection re-quires no increase in die area over theolder ±10kV protected devices. Theuser must only follow the data sheetand use low ESR capacitors (ceramicchip caps are perfect) located close tothe device pins to obtain the benefitsof the on-chip protection (see Figure3). The capacitors and the length of

PC board traces are important, sincethe on-chip protection devices shuntthe ESD transient energy throughthe capacitors and away from activecircuitry. The user should take careto provide a short ground return pathfor the energy in order not to causesecondary damage to other PC boardcomponents caused by the high-speed, high-current transients thatwill flow during an ESD discharge.

Is ESD Protection Effective?In 1992 Linear Technology intro-

duced the first integrated circuitscapable of withstanding ±10kV ESDtransients as tested per Mil-Std.-883BMethod 3015 (Human Body Model).Since their introduction, millions ofunits have been installed in equip-ment, but not one single unit hasbeen returned as a field failure due toESD damage. Prior to the on-chip10kV ESD protection, interface cir-cuits would routinely be found tohave suffered ESD damage when theywere used in systems that did not usea TransZorb for ESD protection. Theon-chip ESD protection has proven tobe a great enhancement to equip-ment reliability and a cost saving tocomputer manufacturers, who nolonger need to add protective devicesto their I/O ports. The enhanced pro-tection achieved by meeting theIEC-801-2 test standards will pro-long this perfect record.

Figure 3. LT1137A ESD test circuit

DESIGN INFORMATION

Table 1. IEC-801-2 severity levels

Test Voltage Test VoltageLevel Contact Discharge (kV) Air Discharge (kV)

1 2 22 4 43 6 84 8 15

TransZorb is a registered trademark of General Instruments, GSI.

ESD-3.eps

5V VCC

0.1µF

0.2µF

0.1µF

RS232 LINE PINS

PROTECTED TO ±15kV

LT1137A1

2

3

4

5

6

7

8

9

10

11

12

13

14

DRIVER 1 OUT

RX1 IN

DRIVER 2 OUT

RX2 IN

RX3 IN

RX4 IN

DRIVER 3 OUT

RX5 IN

ON/OFF

28

27

26

25

24

23

22

21

20

19

18

17

16

15

0.2µF

DRIVER 1 IN

RX1 OUT

DRIVER 2 IN

RX2 OUT

RX3 OUT

RX4 OUT

DRIVER 3 IN

RX5 OUT

+ +

+

V–V+

++0.1µF

GND

DRIVER DISABLE



V.35 Transceivers Allow3-chip V.35 Port Solution by Y.K. Sim

receiver with 100Ω as load termina-tion. The ground potential betweenthe source and load is allowed to varyup to ±4V, and the common-moderesistance at both source and load is150Ω. The specification calls for dif-ferential transmitter signals of 0.55V±20% and no more than ±0.6Vcomon mode, exclusive of ground-potential differences. Early V.35implementations used constant-volt-age-output drivers for the differentialsignals, and resistors in series withthe outputs to provide the required±0.55V output swing. This techniquesuffered from a strong dependenceon power supply voltage and driveroutput impedance to control the out-put voltage. Further, common-modevoltages present at the driver outputhad a strong effect on output swing,significantly limiting the driver-out-put common mode range.

The LTC1345/LTC1346 transceiv-ers avoid these problems by using aconstant-current, high impedanceoutput driver. Designed to be usedwith matching external resistor net-works to meet the Appendix IIspecifications, the transmitter outputstructure consists of two complemen-tary switched current sources thatmaintain a 0.55V signal magnitudeover ±4V common mode range. Thesecurrent sources are trimmed to sourceand sink 11mA at the complemen-tary outputs. The differential outputvoltage of the transmitter is set by theexternal source and load terminationresistor networks. The typical differ-ential source/load resistance of 100Ωgives an output voltage of 0.55V (11mA× 100Ω/2). The common mode volt-age at the transmitter is typicallyless than 0.2V with zero ground po-tential difference. The LTC1345 andLTC1346 require 5% tolerance in thetermination resistors to meet the20% differential voltage variationspecification. A termination resistor

network such as BI Technologies partnumber 627T500/1250 (SOIC) or899TR50/125 (DIP) meets this speci-fication and is recommended forproper cable termination. Discrete5% resistors can also be used to ter-minate the drivers and receivers;either a Y (wye, Figure 1A) or ∆ (delta,Figure 1B) connection can be used togive the required 100Ω differentialand 150Ω common mode impedances.

The LTC1345/LTC1346 differen-tial input receivers have 30kΩ inputimpedance and a typical hysteresis of40mV. This allows the receivers tomeet the V.35 specification with thespecified termination resistor net-work. Additionally, they can be usedto detect RS422 signals if the 125ΩV.35 common mode resistor is re-moved. Both the LTC1345 andLTC1346 transceivers feature fourdigitally-selectable operating modes:DTE, with two drivers and three re-ceivers active; DCE, with three driversand two receivers active; all threedrivers and receivers active, and shut-down. In shutdown, supply currentdrops to 1µA typically. The differen-tial transceivers are capable ofoperating at data rates above10MBaud in non-return-to-zero(NRZ) format.

The RS232 handshaking lines canbe implemented with standard RS232transceivers. The LT1134A providesfour RS232 drivers and four receiv-ers, enough to implement the extended8-line handshaking protocol speci-fied in V.35. The LT1134A alsoincludes an onboard charge pump togenerate the higher voltagesrequired by RS232 from a single 5Vsupply, making it an ideal companionto the LTC1345. These two chips, to-gether with the BI Technologiestermination resistor network chip,provide a complete, surface mount-able, 5V-only V.35 data port. Systemsthat have multiple power supply

Two new LTC interface devices, theLTC1345 and the LTC1346, providethe differential drivers and receiversneeded to implement a V.35 inter-face. When used in conjunction withan RS232 transceiver like theLT1134A, they allow a complete V.35interface to be implemented with justtwo transceiver chips and one resis-tor-termination chip. The signals ina typical V.35 interface are dividedinto two groups, with five high-speeddifferential signals for clock and datatransfer and five or eight slower,single-ended V.28 (RS232) signals forhandshaking. The LTC1345 andLTC1346 provide the three differen-tial drivers and receivers necessary toimplement the high-speed path, andthe LT1134A provides the four RS232drivers and receivers required for thehandshaking interface. Both theLTC1345 and the LT1134A provideonboard charge pump power sup-plies, allowing a complete V.35interface to be powered from a single5V supply. For systems where ±5Vsupplies are present, the LTC1346 isoffered without charge pumps, repre-senting a 30% power savings.

Appendix II of the CCITT V.35specification describes the differen-tial signal requirements for thehigh-speed interface lines. The speci-fication stipulates a differentialtransmitter of 100Ω source termina-tion impedance driving a 100Ωtwisted-pair cable and a differential

Figure 1. Y and ∆ termination topologies

DESIGN INFORMATION

50Ω

50Ω

125Ω

A

B1345_2.eps

300Ω

300Ω

120Ω


DESIGN INFORMATION

voltages available and use only thesimpler 5-signal V.35 handshakingprotocol can use the LTC1346 withthe LT1135A or LT1039 RS232 trans-

ceivers; this combination provides acomplete port while saving boardspace and complexity. Figure 2 showsa typical LTC1345/LTC1134A V.35

Figure 2. Typical V.35 implementation using LTC1345 and LTC1346

implementation with five differentialsignals and five basic handshakingsignals, with an option for three addi-tional handshaking signals.

4

1µF

VCC1 5V 2 1

1µF

1µF

28

273

61

2

26

25

12P

S

U

W

Y

X

V

T

R

1011

16

15

1µF

DX

LTC1345 LTC1346

BI 627T500/

1250 (SOIC)

BI 627T500/

1250 (SOIC)

VCC2 5V

0.1µF

21

0.1µF

VEE2

RXT

TXD (103)

SCTE (113)

TXC (114)

RXC (115)

RXD (104)

GND (102)

CABLE SHIELD

T

73

AA

P

S

U

W

Y

X

V

T

R

B B

A A

AA

4

24

23

1011

9

14

13DX RXT T

1114

13

20

19

14

2

24

23T T

1212

11

18

17

10 14 89

35

4

22

21T T

H H

8

1310

9

16

15

75 37

56

6

20

19T T

RX

RX

RX

VCC1

8 127

VCC2

DTR (108)

DX

DX

DX

4

0.2µF

3 22

0.2µF 0.2µF 0.2µF

0.1µF

23

241

2

21

0.1µF LT1134A LT1134A

4 3 22

0.1µF

23

241

0.1µF

19

13 13

1345_1.eps

50Ω

=125Ω

T

50Ω

DX

DX

17

OPTI

ONAL

SIG

NALS

DX

15DX

20RX

18RX

16RX

14

5

7

9

11

6

8

10

12

20

18

2

16

14

21

19

17

15

6

8

10

12

5

7

9

11RX

RX

RX

RX

RX

DX

DX

DX

DX

C CRTS (105)

E EDSR (107)

D DCTS (106)

F FDCD (109)

NN NNTM (142)

N NRDL (140)

L LLLB (141)

ISO 2593 34-PIN DTE/DCE

INTERFACE CONNECTOR

ISO 2593 34-PIN DTE/DCE

INTERFACE CONNECTOR

+ +

+ +

++


dI1377_1.eps

LT1377

CTX20-2P*

2k

1.24k

5V 1A

*CTX20-2P, COILTRONICS 20µH **OSCON, SANYO VIDEO COMPONENTS

47nF

MBRS130

150µF 6.3V OSCON**

100nF

D1 1N5818

4.7nF

C1 2.2µF

SHDN

VSW

NFBN.C.

8V TO 30V INPUT

SG

3.57k 10Ω1N41484

32

6 1 7

5 8

PGVC

V+

PFB

+

+

+100µF

1MHz Step-Down Converter Ends455kHz IF Woes

There can be no doubt that switch-ing power supplies and radio IFs don’tmix. One-chip converters typicallyoperate in the range of 20kHz–100kHz,placing troublesome harmonics rightin the middle of the 455kHz band.This contributes to adverse effectssuch as “de-sensing” and outrightblocking of the intended signals.A new class of switching convertermakes it possible to mix high-efficiency power supply techniquesand 455kHz radio IFs without fear ofinterference.

The circuit shown in Figure 1 usesan LT1377 boost converter, operatingat 1MHz, to implement a high-efficiency buck-topology switchingregulator. The switch is internallygrounded, calling for the floating

supply arrangement shown (D1 andC1). The circuit converts inputs of8V–30V to a 5V/1A output.

The chip’s internal oscillator oper-ates at 1MHz for load currents ofgreater than 50mA, with a guaranteedtolerance of 12% over temperature.Even wideband 455kHz IFs areunaffected, as the converter’s operat-ing frequency is well over one octavedistant.

Figure 2 shows the efficiency ofFigure 1’s circuit. You can expect80%–90% efficiency over an 8V–16Vinput range with loads of 200mA ormore. This makes the circuit suitablefor 12V battery inputs (that’s how I’musing it), but no special consider-ations are necessary with adapterinputs of up to 30V.

Figure 1. Schematic diagram: 1MHz LT1377-based boost converter

DESIGN IDEAS

by Mitchell Lee

IOUT (mA)

50

60

70

80

90

100

EFFI

CIEN

CY (%

)

1000

dI1377_2.eps

200 400 800600

VIN = 8V

VO = 5V

12V

16V

Figure 2. Efficiency graph of the circuitshown in Figure 1

DESIGN IDEAS

1MHz Step-Down Converter Ends455kHz IF Woes ...................... 24Mitchell Lee

A Circuit That Smoothly SwitchesBetween 3.3V and 5V .............. 25Doug La Porte

12-Bit Voltage-Mode DACs PackVersatility into 8-Pin SOs ....... 26Kevin R. Hoskins

Wideband RMS Noise Meter ..... 28Mitchell Lee

LTC1477: 70mW ProtectedHigh-Side Switch Eliminates“Hot Swap” Glitching ............. 32Tim Skovmand

Driving a High-Level Diode-RingMixer with an OperationalAmplifier ................................33Mitchell Lee

Fully Differential, 8-Channel,12-Bit A/D System Using theLTC1390 and LTC1410 .......... 36Kevin R. Hoskins


A Circuit That Smoothly SwitchesBetween 3.3V and 5V

Many subsystems require supplyswitching between 3.3V and 5V tosupport both low-power and high-speed modes. This back-and-forthvoltage switching can cause havoc tothe main 3.3V and 5V supply buses.If done improperly, switching the sub-system from 5V to 3.3V can cause amomentary jump on the 3.3V bus,damaging other 3.3V devices. Whenswitching the subsystem from 3.3Vto 5V, the 5V supply bus can bepulled down while charging thesubsystem’s capacitors, and may in-advertently cause a reset.

The circuit shown in Figure 1 al-lows smooth voltage switchingbetween 3.3V and 5V, with addedprotection features to ensure safeoperation. IC1 is an LTC1470 switch-matrix device. This part has on-chipcharge pumps running from the 5Vsupply to fully enhance the internalN-channel MOSFETs. The LTC1472also has guaranteed break-before-make switching to prevent cross-conduction between buses. It alsofeatures current limiting andthermal shutdown.

When switching the subsystemfrom 5V to 3.3V, the holding capaci-tor and the load capacitance areinitially charged up to 5V. Connect-ing these capacitors directly to themain 3.3V bus causes a momentarystep to 5V. This transient is so fastthat the power supply cannot react intime. Switching power supplies havea particularly difficult time copingwith this jump. Switching suppliessource current to raise the supplyvoltage and require the load to sinkcurrent to lower the voltage. A switch-ing supply will be unable to react tocounter the large positive voltage step.This jump will cause damage to manylow-voltage devices.

The circuit in Figure 1 employs acomparator (IC2) and utilizes the highimpedance state of the LTC1470 toallow switching with minimal effecton the supply. When the 3.3V outputis selected, IC1’s output will go into ahigh impedance state until its outputfalls below the 3.3V bus. The outputcapacitors will slowly discharge to3.3V, with the rate of discharge de-pending on the current being pulledby the subsystem and the size of the

Figure 1. Schematic diagram of 3.3V and 5V switchover circuit

holding capacitor. The example shownin Figure 1 is for a 250mA subsystem.The discharge time constant shouldbe about 4ms. Once the subsystemsupply has dropped below the 3.3Vsupply, the comparator will trip, turn-ing on the 3.3V switch. Thecomparator has some hysteresis toavoid instabilities. The subsystemsupply will reach a low point of about3V before the 3.3V switch is fullyenhanced.

When switching from 3.3V to 5V,IC1’s current limiting prevents themain 5V bus from being dragged downwhile charging the holding capacitorand the subsystem’s capacitance.Without current limiting, the inrushcurrent to charge these capacitorscould cause a droop in the main 5Vsupply.

If done improperly, supply voltageswitching leads to disastrous systemconsequences. The voltage switchshould monitor the output voltageand have current limiting to preventmain supply transient problems. Acorrectly designed supply voltageswitch avoids the pitfalls and resultsin a safe, reliable system.

by Doug La Porte

DESIGN IDEAS

dI1470_1.eps

34

8

74

3

2 6

8

1

7

21k

IC1 LTC1470

5V0.1µF

220µF TANTALUM HOLDING CAPACITOR

TO SUBSCRIBER

EN0

EN1

VOUT

VOUT

3.3V

0 = 5V 1 = 3.3V

51k

5–

+

3.3V1µF

5V

3VIN 3VIN5VIN

1µF

IC2 LT1011

+

+ +

100µs/DIV

5V

3.3V

0V

5V5V/DIV

500mV/DIV

2ms/DIV

Figure 2. Oscillograph of the switchoverwaveform showing smooth transitions


12-Bit Voltage-Mode DACs PackVersatility into 8-Pin SOsIntroduction

Let’s face it, microprocessors andmicrocontrollers cannot control me-chanical potentiometers. They haveno fingers, just pins. And they haveneither eyes with which to set a wiper’sposition nor the intuition to knowwhen it’s set correctly. They need theorgans that provide these functionstransplanted into their “operationalenvironment.” The “organs” that fillthese roles are analog-to-digital con-verters (ADCs) and the sensorsconnected to their inputs, and digi-tal-to-analog converters (DACs) andthe manipulative circuits or mechan-ics they drive.

This article concentrates on the“fingers” and “hands” capabilities thatDACs bestow upon microprocessorsand microcontrollers. Linear Tech-nology has recently introduced 12-bit,serially interfaced, voltage-outputDACs packaged in small 8-pin SOs.These parts include the LTC1257,LTC1451, LTC1452, and LTC1453.All of these DACs are micropower,have 12-bit monotonic transfer func-tions, and operate on 4.75V to 15.75V(LTC1257), 5V (LTC1451), or 3V to 5V(LTC1452 and LTC1453) supply volt-ages. Table 1 captures the importantspecifications of the LTC1257,LTC1451, LTC1452, and LTC1453.

Figures 1 and 2 show the blockdiagrams of the LTC1257 and theLTC1451, LTC1452, and LTC1453.These devices share many commoncircuit elements. With a logic lowapplied to the CS input, serial data isapplied to a 12-bit serial shift regis-ter. Each bit is latched on the rising

edge of the CLK input signal. Theshift register’s contents are trans-ferred to the DAC register on therising edge of CS, which in turn up-dates the output voltage’s magnitude.Each of these DACs also has a dataoutput, DOUT, making it easy todaisy-chain multiple DACs on a singleserial data line. The LTC1257 andLTC1451 have an on-chip 2.048Vbandgap reference. The LTC1453 hasa 1.22V bandgap reference. TheLTC1257 has an internal 5V regula-tor that powers all onboard digitalcircuitry.

System AutorangingSystem auto-ranging, adjusting an

ADC’s full-scale range, is an applica-tion area for which these DACs areappropriate. Auto-ranging is espe-cially useful when using an ADC withmultiplexed inputs. Without auto-ranging, only two reference valuesare used: one to set the full-scalemagnitude and another to set the

zero-scale magnitude. Since it is com-mon to have input signals withdifferent zero-scale and full-scalemagnitude requirements, fixed refer-ence voltages present a problem.Although the ranges selected for someof the inputs may take advantage ofthe full range of ADC output codes,inputs that do not span the samerange will not generate all codes, re-ducing the ADC’s effective resolution.One possible solution is to match thereference voltage span to the need ofeach multiplexer input.

The circuit shown in Figure 3 usestwo LTC1257s to set the full-scaleand zero operating points of theLTC1296 12-bit, 8-channel ADC. TheADC shares its serial interface withthe DACs. To further simplify busconnections, the DACs’ data is daisy-chained. Two chip selects are used,one to select the LTC1296 when pro-gramming its multiplexer, and theother to select the DACs when settingtheir output voltages.

Figure 1. LTC1257 block diagram

Supply Differential Integral LinearityPart Number Voltage Range Supply Current (Typ) Nonlinearity (Max) Error (Max)LTC1257 4.75V to 15.75V 350µA at 5V Supply ±0.5 LSB ±3.5 LSBLTC1451 4.5V to 5.5V 400µA at 4.5V ≤ VCC ≤ 5.5V ±0.5 LSB ±3.5 LSBLTC1452 2.7V to 5.5V 225µA at 2.7V ≤ VCC ≤ 5.5V ±0.5 LSB ±3.5 LSBLTC1453 2.7V to 5.5V 250µA at 2.7V ≤ VCC ≤ 5.5V ±0.5 LSB ±3.5 LSB

Table 1. Key specifications of the LTC1257, LTC1451, LTC1452, and LTC1453 8-pin, SO packaged, voltage-out DACs

DESIGN IDEAS

–

+DAC

5V REGULATOR

12-BIT SHIFT REGISTER

12-BIT LATCH

2.048V REFERENCE

VCC

DOUT

GND

VOUT

DIN

LOGIC SUPPLY

CLK

LOAD

REF 12

12

dIDAC_1.eps

by Kevin R. Hoskins


The circuit in Figure 4 is a com-puter controlled 4mA-to-20mAcurrent loop. It is designed to operateon a single supply over a range of 3.3Vto 30V. The circuit’s zero output ref-erence signal, 4mA, is set by R1 andcalibrated using R2, and its full-scaleoutput current is set by R3 and cali-brated using R4. The zero- andfull-scale output currents are set asfollows: with a zero input code appliedto the LTC1453, the output current,IOUT, is set to 4 mA by adjusting R2;next, with a full-scale code applied tothe DAC, the full-scale output cur-rent is set to 20mA by adjusting R4.

The circuit is self regulating, forc-ing the output current to remainstable for a fixed DAC output voltage.This self regulation works as follows:starting at t = 0, the LTC1453’s fixedoutput (in this example, 2.5V) isapplied to the left side of R3; instan-taneously, the voltage applied to theLT1077’s input is 1.25V; this turnson Q1 and the voltage across RS startsincreasing beginning from 1.25V; asthe voltage across RS increases, itlifts the LTC1453’s GND pin above0V; the voltage across RS continues toincrease until it equals the DAC’soutput voltage.

Once the circuit reaches this stablecondition, the constant DAC outputvoltage sets a constant currentthrough R3 + R4 and R5. This con-stant current fixes a constant voltageacross R5 that is also applied to theLT1077’s noninverting input. Feed-back from the top of RS is applied tothe inverting input. As the op ampforces its inputs to the same voltage,it will fix the voltage at the top of RS.This in turn fixes the output currentto a constant value.

Opto-Isolated Serial InterfaceThe serial interface of the LTC1451

family and the LTC1257 makeopto-isolated interfaces very easy andcost effective. Only three opto-isola-tors are needed for serial datacommunications. Since the inputs ofthe LTC1451, LTC1452, and LTC1453have generous hysteresis, the switch-ing speed of the opto-isolators is not

Figure 2. Block diagram of the LTC1451, LTC1452, and LTC1453

During the conversion process, U2and U3 receive the full- and zero-scale codes, respectively, thatcorrespond to a selected multiplexerchannel. For example, let channel 2’sspan begin at 2V and end at 4.5V.When a host processor wants a con-version of channel 2’s input signal, itfirst sends code that sets the outputof U2 to 2V and U3 to 4.5V, fixing thespan to 2.5V. The processor sendsdata to the LTC1296, selecting chan-nel 2. The processor next reads datathat was generated during the con-version of the 3.5VP–P signal applied

to channel 2 from the LTC1296. Asother multiplexer channels are se-lected, the DAC outputs are changedto match their spans.

Computer Controlled4mA-to-20mA Current Loop

A common and useful circuit is the4mA-to-20mA current loop. It is usedto transmit information over long dis-tances using varying current levels.The advantage of using current overvoltage is the absence of IR losses andthe transmissions errors and signallosses they can create.


DESIGN IDEAS

DAC REGISTER

LD

+

–

REFERENCE LTC1451: 2.048V LTC1453: 1.22V

12-BIT SHIFT

REGISTER

POWER-ON RESET

dIDAC_2.eps

CLK 1

DIN 2

DOUT 4

VOUT7

REF6

GND5

VCC8

3CS/LD

12-BIT DAC

5V

0.1µF VCC

VREF

VOUT

DIN

CLK

LOAD

DOUT

U2 LTC1257

VCC

VREFGND

VOUT

DIN

CLK

LOAD

DOUT

U3 LTC1257

dIDAC_3.eps

0.1µF

0.1µF

100Ω

100Ω

µP8 ANALOG

INPUT CHANNELS U1 LTC1296

CS

DOUT

CLK

DIN

CH0

• • •

CH7 COM REF+ REF– SSO

22µF

5V

50k 50k

VCC

74HC04

GND

+

Figure 3. Using two LTC1257 12-bit voltage-output DACs to set the input span of the 12-bit,8-channel LTC1296


Wideband RMS Noise Meterby Mitchell Lee

Recently, I needed to measure andoptimize the wideband RMS noise ofa power supply over about a 40MHzbandwidth. A quick calculationshowed that the 12nV–15nV/√Hznoise floor of my spectrum analyzerwould come up short—my circuit waspredicted to exhibit a spot noise ofperhaps 8nV–10nV/√Hz. In fact,I didn’t have a single instrument inmy lab that would measure 50µV–60µV RMS.

For the 40MHz bandwidth, theHP3403C RMS voltmeter is a goodchoice, but its most sensitive range is100mV, about 66dB shy of my re-quirement. This obsolete instrumenttoday carries a hefty price on the usedmarket. The fact that here in theSilicon Valley HP3403Cs are a com-mon sight at flea markets is of littleconsolation to most customers wish-ing to reproduce my measurements.We have several of these meters in theLTC design lab, but they are in con-stant use and closely guarded by “TheKeepers of the Secret RMS Knowl-edge.” I resolved to build my own

meter using an LT1088 thermal RMSconverter.

Full scale on the LT1088 is 4.25VRMS. To measure 50µV full-scale, I’dneed an amplifier with a gain of100,000. At 40MHz bandwidth, thisdidn’t sound like it would have a goodchance of working first-time—builtby hand and without benefit of acustom casting.

Rather than build a circuit with40MHz bandwidth and a gain of100dB, I decided to use just enoughgain to put my desired noise perfor-mance around twice minimum scale.Aside from gain, this amplifier wouldalso need less than 5nV/√Hz inputnoise, and the output stage wouldhave to drive the 50 ohm load pre-sented by the LT1088.

It wasn’t hard to find an appropri-ate output stage. The LT1206 (seeFigure 1) can easily drive the required120mA peak current into the LT1088converter, and there’s plenty left overfor handling noise spikes. To pre-serve 40MHz bandwidth, the LT1206was set to run at a gain of 2.

Figure 1. Noise meter gain stage

DESIGN IDEAS

–

+–

+

–

+

LT1192

–5V

10nF

10nF

10nF

100nF

5V

35

7

4

6

1200Ω

300Ω

2

dI1226_1.eps

LT1226

–5V

10nF

10nF

10nF

100nF

+5V

3 7

4

6

1200Ω

1000Ω 100Ω 100Ω 1000Ω

SIGNAL INPUT

TO LT1206

100nF

2LT1226

–5V

10nF

10nF

10nF

100nF

5V

–5V

100nF

5V

3 7

4

6

1200Ω

100Ω 100Ω 1000Ω

2

The front end was harder to solve.I needed a low noise, high speedamplifier that could give me plenty ofgain. Here I selected the LT1226. Thisis a 1GHz GBW op amp with only2.6nV/√Hz input noise. It has a mini-mum stable gain of 25, but in thiscircuit, high gain is an advantage.

Cascading two LT1226s on thefront end gives a gain of 625, a littleshy of the 5,000 to 10,000 required.Another gain of 5, plus the gain of 2 inthe LT1206 adds up to a gain of6,250—just about right.

There are several ways to get 40MHzbandwidth at a gain of 5, includingthe LT1223 and LT1227 current-feed-back amplifiers, but I settled on theLT1192 voltage amplifier because itis the lowest cost solution. This bringsthe gain up to 6250, for a minimumscale sensitivity of 34µV RMS, and afull-scale sensitivity of 680µV RMS.

My advice-filled coworkers assuredme that there was no way I couldbuild a wideband amplifier with again of 6250, and make it stable.Nevertheless, I built my amplifier on a



Figure 3. LT1088 RMS detector section

Figure 2. LT1206 buffer/driver section

1.5" × 6" copperclad board, takingcare to maintain a linear layout. Thefinished circuit was stable, providedthat a coaxial connection was madeto the input. The amplifier was flat,with 3dB points at 4kHz and 43MHz,and some peaking at high frequencies.

The performance of the amplifierand thermal converter can be opti-mized by adjusting the value of thefeedback and gain setting resistorsaround the LT1206. Slightly morebandwidth can be achieved at theexpense of higher peaking by reduc-ing the resistor values 10%. Reducingthe resistor values will decreasepeaking effects at the expense ofbandwidth. A good compromise valueis 680Ω.

I’ve shown the LT1226 amplifiersoperating from ±5 supplies, whichputs their bandwidth on the edge at

DESIGN IDEAS

–

+ U2 1/4

LT1014

10nF

15V

15V

12

14

10k

10k

10k

OUTPUT1322nF 12k

33k

500Ω

1k 1%

1k 1%

1k

9.09M 1%

9.09M 1%

27.4k 1%

27.4k 1%

5V

100nF

5V

100nF

–5V

+

– U1 1/4

LT1014

6

7

5

–

+ U4 1/4

LT1014

3

1

2

–

+U3 1/4

LT1014

8

6131 7 8

5122 9

30k

4

15V

–15V

LT1088

SUB

2N3904

2N2219

FROM LT1206

9

10

11

SHUTDOWN CONTROL

14

SUB

10nF

3.3nF

dI1226_3.eps

–

+

dI1226_2.eps

LT1206

–15V

100nF

10nF

10nF

10nF

100nF

15V

2FROM LT1192

4

6

3

5

7

24kΩ

620Ω

1000ΩTO LT1088

SHUTDOWN CONTROL

620Ω

100nF

10nF1


Coaxial MeasurementsWhen measuring low-level sig-

nals, it is difficult to get a clean,accurate result. Scope probes havetwo problems. 10× probes attenuatethe already small signal, and both1× and 10× probes suffer from cir-cuitous grounds. Coaxial adaptersare a partial solution, but these areexpensive. They make for a lot ofwear and tear on the probes, and,without a little forethought, theycan be a bear to attach to the circuitunder test. My favorite way to getclean measurements of small sig-nals is to directly attach a shortlength of coaxial cable as shown inFigure 4.

I use the good part of a damagedBNC cable, cutting away the shorterportion to leave at least 18" of RG-58/U and one good connector. Atthe cut, or as I call it, “real world”end of the cable, I unbraid, twist,and tin a very small amount of outerconductor to form a stub 1/4" to3/8" long. Next, I cut away the di-electric, exposing a similar length ofcenter conductor, which I also tin.Now the probe is ready for use. It canbe soldered directly to a circuit orbreadboard, eliminating any leadlength that might otherwise pick upstray noise, or worse, act as an an-tenna in a sensitive, high gain circuit.

Small signals aren’t the only ben-eficiaries of this technique. This worksgreat for looking at ripple on the out-puts of switching supplies. Ripplemeasurements are simplified becausethe large voltage swings associatedwith the switch node are completelyisolated and no loop is formed wheredi/dt could inject magneticallycoupled noise.

In some instances, I’ve found itimportant to retain the 50Ω termina-tion impedance on the cable, but it israrely possible to place a terminatorat the “BNC” end of the cable, sincethis creates a DC path directly acrossthe circuit under test. There is, how-ever, another way as shown in Figure5. Here, a technique known as back-termination is used. No terminationis used at the far end of the cable, buta 51Ω resistor is connected in serieswith the measurement end. Signalssent down the cable reach the BNCconnector without attenuation, andfast edges that bounce off theunterminated end are absorbed backinto the 51Ω source resistor. I’ve foundthis especially useful for measuringfast switch signals, or, when measur-ing the RMS value of small signals,for ensuring that the amplifier inputsees a properly terminated source.The resistor trick does not work if the

node under test is high impedance;an FET probe is a better choice forhigh-impedance measurements.

While I’m at it, I might as wellgive away my only other secret. We’veall encountered ground loop prob-lems, giving rise to 60Hz (50Hz formy friends overseas) injection intosensitive circuits. Every lab is re-plete with isolation transformers and“controlled substance” line cordswith missing ground prongs for bat-tling ground loops. There are similarproblems at high frequencies, butthe victim is wave fidelity, not ACpick up. To determine whether ornot high frequency grounding,ground loops, or common-mode re-jection is a problem for youroscilloscope, simply clamp a smallferrite E core around the probe leadwhile observing any effects on thewaveform (see Figure 6). Sometimesthe news is bad; the waveform reallyis messed up and there is some workto be done on the circuit. But occa-sionally the circuit is exonerated,the unexplainable aberrations dis-appear, proving that high frequencygremlins are at work. If necessary,several passes of the probe cablecan be made through the E-Core,and it can be taped together for aslong as needed.

Figure 4. BNC cable used as a “probe” Figure 5. BNC cable probe back terminated

40MHz. Their bandwidth can be im-proved by operating at ±15V.

Because the LT1206 operates on15V rails, it is possible to overdrivethe LT1088 and possibly cause per-manent damage. One section of the

LT1014 (U3) is used to sense an over-drive condition on the LT1088 andshut down the LT1206. Sensing thefeedback heater instead of the inputheater allows the LT1088 to accom-modate high-crest-factor waveforms,

DESIGN IDEAS

shutting down only when the averageinput exceeds maximum ratings.

By the way, my power supply noisemeasured 200µV; filtering brought itdown to less than 60µV.

dicoax_1.eps dicoax_2.eps

51Ω


12-Bit DACs, continued from page 27

critical. Further, because each of theseDACs can be daisy-chained to others,only three opto-isolators are required.

Control RightWhere It’s Needed

Two major benefits of these DACsare their small SO-8 size and serialinterface. These features make it veryeasy to place an LTC1257 or anLTC1451, LTC1452, or LTC1453 on aPCB exactly where needed, next to itsassociated circuitry. Unlike the com-promises that exist with placing amulti-DAC package and the difficultyin optimizing analog-signal-tracelength and routing, the placementflexibility of the LTC1257 andLTC1451, LTC1452, or LTC1453 re-duces analog-signal-path lengths toa minimum.

ConclusionThe LTC1257 and the LTC1451,

LTC1452, and LTC1453 serially pro-grammed 12-bit DACs are idealsolutions for applications that

require 12-bit functionality in smallSO-8 packages and the ease andeconomy of a serial interface.

DESIGN IDEAS

Figure 4. The LTC1453 forms the heart of this isolated 4mA-to-20mA current loop

dicoax_3.eps

Figure 6. An E-Core serves to choke off high frequency common mode currents from a ’scope probe

dIDAC_4.eps

R5 3k

10k

1k

R3 45k

R4 5k

R1 90k

R2 5k

Q1 2N3440

RS 10Ω

VLOOP 3.3V TO 30V

IOUT

OUTIN

CLK

DIN

CS/LD

CLK DIN CS/LDCLK

DIN CS/LD

VCC

VOUT

1µF

LTC1453

4N28

OPTO-ISOLATORS

3.3V

500Ω

LT1121-3.3

FROM OPTO-

ISOLATED INPUTS

VREF

+

–LT1077


LTC1477: 70mΩ ProtectedHigh-Side Switch Eliminates“Hot Swap” Glitching

When a printed circuit board is“hot swapped” into a live 5V socket, anumber of bad things can happen.

First the instantaneous connec-tion of a large, discharged,supply-bypass capacitor may causea glitch to appear on the power bus.The current flowing into the capaci-tor is limited only by the socketresistance, the card-trace resistance,and the equivalent series resistance(ESR) of the supply bypass capacitor.This supply glitch can create realhavoc if the other boards in the sys-tem have power-on RESET circuitrywith thresholds set at 4.65V.

Second, the card itself may be dam-aged due to the large inrush of currentinto the card. This current is some-times inadvertently diverted tosensitive (and expensive) integratedcircuits that cannot tolerate eitherover-voltage or over-current condi-tions even for short periods of time.

Third, if the card is removed andthen reinserted in a few milliseconds,the glitching of the supply may “con-fuse” the microprocessor or peripheralICs on the card, generating errone-ous data in memory or forcing thecard into an inappropriate state.

Fourth, a card may be shorted,and insertion may either grossly glitchthe 5V supply or cause severe physi-cal damage to the card.

Figure 1 is a schematic diagramshowing how an LTC1477 protectedhigh-side switch and an LTC699power-on RESET circuit reduce thechance of glitching or damaging thesocket or card during “hot swapping.”

The LTC1477 protected high-sideswitch provides extremely low RDS(ON)switching (typically 0.07Ω) withbuilt-in 2A current limiting andthermal shutdown, all in an 8-pinSO package.

As the card is inserted, the LTC699power-on RESET circuit holds theenable pin of the LTC1477 low forapproximately 200ms. When the en-able pin is asserted high, the outputis ramped on in approximately 1ms.Even if a very large supply bypasscapacitor (for example, over 100µF) isused, the LTC1477 will limit the in-rush current to 2A and ramp thecapacitor at an even slower rate. Fur-ther, the board is protected againstshort-circuit conditions by limitingthe switch current to 2A.

The 5V card supply can be dis-abled via Q1. The only current flowingis the standby quiescent currentof the LTC1477, which drops below1µA, the 600µA quiescent current ofthe LTC699, and the 10µA consumedby R1.

Figure 1. “Hot swap” circuit featuring LTC1477 and LTC699

by Tim Skovmand

DESIGN IDEAS

+

dI1477_1.eps

LTC699CS86

3 4 5 8

7

21

Q1 2N7002

DISABLE

LTC1477CS8R1 510k

C1 1µF

CLOAD 100µF

VOUT, ISC = 2A

VIN1

VIN2

EN

VOUT

VIN2

VIN3

GND

VOUT

NC

5V



Driving a High-Level Diode-RingMixer with an Operational Amplifier

by Mitchell Lee

One of the most popular RF build-ing blocks is the diode-ring mixer.Consisting of a diode ring and twocoupling transformers, this simpledevice is a favorite with RF designersanywhere a quick multiplication isrequired, as in frequency conversion,frequency synthesis, or phase detec-tion. In many applications thesemixers are driven from an oscillator.Rarely does anyone try building anoscillator capable of delivering +7dBmfor a “minimum geometry” mixer, letalone one of higher level. One or morestages of amplification are added toachieve the drive level required by themixer. The new LT1206 high-speedamplifier makes it possible to amplifyan oscillator to +27dBm in one stage.

Figure 1 shows the complete cir-cuit diagram for a crystal oscillator,LT1206 op amp/buffer, and diode-ring mixer. Most of the components

Figure 1. Oscillator buffer drives +17 to +27dBm double balanced mixers

DESIGN IDEAS

–

+

dI1206_1.eps

LT120610nF

10nF 100nF

–15V

BUFFER/AMPLIFIEROSCILLATOR (TYPICAL) DIODE RING MIXER

–15V

10nF 100nF

15V15V

620Ω

51Ω, 2W OPTIONAL DOUBLE-

TERMINATION RESISTOR LO IFRF2 4

6

3

5

7

24kSHUTDOWN

OPERATE

10nF

100nF62pF

10MHz 110nF

1kΩ

120k

5k

PN4416

10k

1

Figure 2. Spectrum plot of Figure 1’s circuit driving +30dBm into a 50Ω load (singletermination)

dI1206_2.eps

–

+LT1206

620Ω

50Ω

30dBm


dI1206_3.eps

–

+LT1206

620Ω

50Ω

27dBm

are used in the oscillator itself, whichis of the Colpitts class. Borrowingfrom a technique I first discovered inHewlett Packard’s Unit Oscillator, thecurrent of the crystal is amplified,rather than the voltage. There areseveral advantages to this method,the most important of which is lowdistortion. Although the voltagespresent in this circuit have poor waveshape and are sensitive to loading,the crystal current represents essen-tially a filtered version of the voltagewaveform and is relatively tolerant ofloading effects.

DESIGN IDEAS

The impedance, and therefore thevoltage at the bottom of the crystal iskept low by injecting the current intothe summing node of an LT1206 cur-rent-feedback amplifier. Loop gainreduces the input impedance to wellunder one ohm. Oscillator bias isadjustable, allowing control of themixer drive. This also provides a con-venient point for closing an outputpower servo loop.

One of the most popularRF building blocks is the

diode-ring mixer.Consisting of a diodering and two coupling

transformers, this simpledevice is a favorite withRF designers anywhere aquick multiplication is

required, as in frequencyconversion, frequencysynthesis, or phase

detection

Figure 3. Spectrum plot of Figure 1’s circuit driving +27dBm into a 50Ω load (single termina-tion)

Figure 4. Spectrum plot of Figure 1’s circuit driving +23dBm into a 50Ω load (singletermination)

dI1206_4.eps

–

+LT1206

620Ω

50Ω

23dBm


Operating from ±15V supplies, theLT1206 can deliver +32dBm to a 50Ωload, and, with a little extra head-room (the absolute maximum supplyvoltage is ±18), it can reach 2W out-put power into 50Ω. Peak guaranteedoutput current is 250mA.

Shown in Figures 2–6 are spectralplots for various combinations ofsingle and double termination atpower levels ranging from +17 to+27dBm—not bad for an inductor-less circuit. Double termination maybe used to present a 50Ω source im-pedance to the mixer, or to isolate twoor more mixers driven simultaneouslyfrom one LT1206 amplifier.

Although a 10MHz example hasbeen presented here, the LT1206’s65MHz bandwidth makes it useful incircuits up to 30MHz. In addition, theshutdown feature can be used to gatedrive to the mixer. When the LT1206is shut down, the oscillator will likelystop, since the crystal then sees aseries impedance of 620Ω and themixer itself. Upon re-enabling theLT1206 there will be some time delaybefore the oscillator returns to fullpower. The circuit works equally wellwith an LC version of the oscillator.

Note that the current feedback to-pology is inherently tolerant of straycapacitive effects at the summingnode, making it ideal for this applica-tion. Another nice feature is theLT1206’s ability to drive heavy ca-pacitive loads while remaining stableand free of spurious oscillations.

For mixers below +17dBm, theLT1227 is a lower cost alternative,featuring 140MHz bandwidth in com-bination with the shutdown featureof the LT1206.

DESIGN IDEAS

Figure 5. Spectrum plot of Figure 1’s circuit driving +27dBm into a 100Ω load (doubletermination)

Figure 6. Spectrum plot of Figure 1’s circuit driving +17dBm into a 100Ω load (doubletermination)

dI1206_5.eps

–

+LT1206

620Ω

50Ω

50Ω

27dBm

dI1206_6.eps

–

+LT1206

620Ω

50Ω

50Ω

17dBm


Fully Differential, 8-Channel, 12-BitA/D System Using the LTC1390and LTC1410

Figure 1. Fully differential 8-channel data acquisition system achieves 625ksps throughput

The LTC1410’s fast 1.25Msps con-version rate and differential ±2.5Vinput range make it ideal for applica-tions that require multichannelacquisition of fast, wide-bandwidthsignals. These applications includemultitransducer vibration analysis,race-vehicle telemetry data acquisi-

tion, and multichannel telecommun-ications. The LTC1410 canbe combined with the LTC13908-channel serial-interfaced analogmultiplexer to create a differentialA/D system with conversion through-put rates up to 625ksps. This rateapplies to situations where the

selected channel changes with eachconversion. The conversion rate in-creases to 1.25Msps if the samechannel is used for consecutive con-versions.

Figure 1 shows the complete dif-ferential, 8-channel A/D circuit. TwoLTC1390s, U1 and U2, are used

by Kevin R. Hoskins

DESIGN IDEAS

++

+

0.1µF

0.1µF

0.1µF

dI1390_1.eps

+AIN

–AIN

VREF

REFCOMP

AGND

D11

D10

D9

D8

D7

D6

D5

D4

DGND

AVDD

DVDD

VSS

BUSY

CS

CONVST

RD

SHDN

NAP/SLP

OGND

D0

D1

D2

D3

U3 LTC1410

S0

S1

S2

S3

S4

S5

S6

S7

V+

D

V–

DATA2

DATA1

CS

CLK

GND

U2 LTC1390

S0

S1

S2

S3

S4

S5

S6

S7

+CH0

+CH1

+CH2

+CH3

+CH4

+CH5

+CH6

+CH7

–CH0

–CH1

–CH2

–CH3

–CH4

–CH5

–CH6

–CH7

SERIAL DATA

CHIP SELECT MUX

SERIAL CLK

V+

D

V–

DATA2

DATA1

CS

CLK

GND

U1 LTC1390

0.1µF

+5V –5V

10µF 0.1µF

10µF10µF 0.1µF

µP CONTROL LINES

12-BIT PARALLEL

BUS

0.1µF


DESIGN IDEAS

Figure 2. Timing diagram of circuit in Figure 1

as noninverting and inverting inputmultiplexers. The outputs of the non-inverting and inverting multiplexersare applied to the LTC1410’s +AINand −AIN inputs, respectively. TheLTC1390 share the CHIP SELECTMUX, SERIAL DATA and SERIALCLOCK control signals. This arrange-ment simultaneously selects the samechannel on each multiplexer: S0 forboth +CH0 and −CH0, S1 for both+CH1 and −CH1, and so on.

As shown in the timing diagram(Figure 2), MUX channel selectionand A/D conversion are pipelined tomaximize the converter’s throughput.The conversion process begins withselecting the desired multiplexerchannel-pair. With a logic high ap-plied to the LTC1390’s CS input, thechannel-pair data is clocked into eachDATA 1 input on the rising edge of the5MHz clock signal. CHIP SELECTMUX is then pulled low, latching the

channel-pair selection data. The sig-nals on the selected MUX inputs arethen applied to the LTC1410’s differ-ential inputs. CHIP SELECT MUX ispulled low 700ns before the LTC1410’sconversion-start input, CONVST, ispulled low. This accounts for the maxi-mum time needed by the LTC1390’sMUX switches to fully turn-on. Thisensures that the input signals arefully settled before the LTC1410’s S/Hcaptures its sample.

The LTC1410’s S/H acquires theinput signal and begins conversionon CONVST’s falling edge. During theconversion, the LTC1390’s CS inputis pulled high and the data for thenext channel-pair is clocked intoDATA 1. This pipelined operationcontinues until a conversion sequenceis completed. When a new channel-pair is selected for each conversion,the sampling rate of each channel is78ksps, allowing an input-signal

bandwidth of 39kHz for each channelof the LTC1390/LTC1410 system.

To maximize the throughput rate,the LTC1410’s CS input is pulled lowat the beginning of a series of conver-sions. The LTC1410’s data outputdrivers are controlled by the signalapplied to RD. The conversion’s re-sults are available 20ns before therising edge of BUSY. The rising edgeof the BUSY output signal can beused to notify a processor that a con-version has ended and data is readyto read.

This circuit takes advantage of theLTC1410’s very high 1.25Msps con-version rate and differential inputsand the LTC1390’s ease of program-ming to create an A/D systemthat maintains wide input-signalbandwidth while sampling multipleinput signals.

DATA 0 DATA 1 DATA 2

SERIAL CLOCK LTC1390

5MHz, 200ns

A LOGIC LOW IS APPLIED TO THE LTC1410S CS INPUT DURING A CONVERSION SEQUENCE.

CHIP SELECT MUX LTC1390

DATA LTC1390

EN B2 B1 B0

700ns

EN B2 B1 B0 EN

CONVERSION 0 CONVERSION 1 CONVERSION 2

B2 B1 B0 EN B2 B1 B0

CONVST LTC1410

BUSY LTC1410

RD LTC1410

OUTPUT DATA LTC1410

dI1390_2.eps


NEW DEVICE CAMEOS

New Device CameosThe DAC provides simple “bits-

to-lamp-current control,” andcommunicates externally in twointerface modes: standard SPI modeand pulse mode. On power-up, theDAC counter is reset to half-scaleand the chip is configured to SPI orpulse mode, depending on the CSsignal level. In SPI mode, the systemmicroprocessor serially transfers thepresent 8-bit data and reads back theprevious 8-bit data from the DAC. Inpulse mode, the upper six bits of theDAC implement increment-only(1-wire interface) or increment/dec-rement (2-wire interface) operation.The operational mode (increment-onlyor increment/decrement) is selectedby the DIN signal level.

The LT1186 operates from a logicsupply voltage of 3.3V or 5V. The ICalso has a battery supply voltage pinthat operates from 4.5V to 30V. TheLT1186 draws 6mA typical quiescentcurrent. An active low shutdown pintypically reduces total supply currentto 35µA for standby operation whilethe DAC retains its last setting. A200kHz switching frequency mini-mizes the size of required magneticcomponents and the use of current-mode switching techniques gives highreliability and simple loop-frequencycompensation. The LT1186 is avail-able in a 16-pin narrow body SO.

LTC1477/LTC1478: Singleand Dual 70mΩ, 2A,Protected High-Side Switchesin SO Packaging

The LTC1477/LTC1478 single anddual protected high-side switchesprovide extremely low RDS(ON) switch-ing with built in protection againstshort-circuit and thermal-overloadconditions. The LTC1477 single isavailable in 8-lead SO packaging andthe LTC1478 dual is available in16-lead SO packaging. Both partsoperate from 2.7V to 5.5V.

The LTC1477/LTC1478 providetwo levels of protection. The first level

LTC1266/LTC1266-3.3/LTC1266-5 SynchronousControllers Feature PrecisionOutput Regulation

The LTC1266, LTC1266-3.3, andLTC1266-5 synchronous controllersfor N- or P-channel MOSFETs providea precision 1% internal reference fortighter regulation of output voltage.This precision reference, combinedwith the LTC1266’s tight load regula-tion and ability to drive two N-channelMOSFETs for high currents, providesthe accuracy and current capabilitynecessary to meet the stringentdemands of many Pentium micro-processor applications.

The LTC1266 is a synchronous,current mode, switching regulatorcontroller with automatic Burst Modeoperation (Burst Mode operation canbe disabled), available in the 16-leadnarrow SO package. Two user-pro-grammable pins make it possible todrive an N-channel MOSFET for highcurrent or step-up operation, or a P-channel MOSFET for low dropoutoperation. With an N-channel top-side MOSFET for higher efficiency athigh currents and Burst Mode opera-tion enabled for high efficiencies atlow currents, the LTC1266 can pro-vide 90% or better efficiency for loadsfrom 10mA to 10A.

Other features of the LTC1266 in-clude a shutdown mode, whichreduces supply current to 40µA, andan on-chip comparator for use as a“power-good” monitor, which staysactive in shutdown. The maximumoperating frequency of 400kHz al-lows the use of small, surface mountinductors and capacitors.

LT1368 and LT1369Precision Rail-to-Rail OpAmps with Excellent High-Frequency Supply Rejection

The LT1368 dual and LT1369 quadare the latest members of the LT1366rail-to-rail op amp family. Designed

to operate with a 0.1µF output-com-pensation capacitor, these devicesfeature improved high frequency sup-ply rejection and lower high frequencyoutput impedance. The output ca-pacitance acts as a filter, reducingnoise pickup and ripple from the sup-ply. This additional filtering is helpfulin mixed analog/digital systems withcommon supplies, or in systems em-ploying switching supplies.

Like the LT1366, the LT1368 andLT1369 combine rail-to-rail input andoutput operation with precision speci-fications. These devices maintain theircharacteristics over a supply range of1.8V to 36V. Operation is specifiedwith +3V, +5V, and ±15V supplies.The input offset voltage is typically150µV, with a minimum open loopgain (AVOL) of 1 million while driving a2k load. The typical input-bias andoffset currents are 10nA and 1nArespectively. The supply current peramplifier is typically 375µA.

Common mode rejection is typi-cally 90dB over the full rail-to-railinput range, and supply rejection is110dB (DC). The filter formed withthe amplifier’s output impedance andthe external compensating capacitorprovides approximately 40dB of sup-ply rejection above 30kHz.

The LT1368 dual is available inplastic 8-pin DIP and 8-lead SOpackages. The LT1369 quad is avail-able in a plastic 16-lead SO package.

LT1186 DAC-ProgrammableCCFL Switching Regulator(Bits-to-NitsTM)

The LT1186 is a fixed frequency,current mode switching regulator thatprovides the control function for cold-cathode fluorescent lighting. TheLT1186 supports grounded-lamp andfloating-lamp configurations. The ICincludes an efficient, high-currentswitch, an oscillator, output drivelogic, control circuitry, protection cir-cuitry, and a micropower 8-bit, 50µA,full-scale current output DAC.

Bits-to-Nits is a trademark of Linear Technology Corporation.Pentium is a registered trademark of Intel Corporation.


For further information on theabove or any of the other devicesmentioned in this issue of LinearTechnology, use the reader servicecard or call the LTC literature ser-vice number: 1-800-4-LINEAR. Askfor the pertinent data sheets.

NEW DEVICE CAMEOS

is short-circuit current limiting, whichis set at 2A. The short-circuit currentcan be reduced to as low as 0.85A bydisconnecting portions of the powerdevice. The second level of protectionis thermal-overload protection, whichlimits the die temperature to approxi-mately 130°C. A built in charge pumpproduces gate drive from the 2.7V-to-5.5V power supply input forcontrolled ramping of the internalultra-low RDS(ON) NMOS switch. Nohigher gate voltage power supply isrequired. Rise time is set internally at1ms, to reduce inrush currents tonegligible levels; therefore, extremelylarge load capacitors can be rampedwithout glitching the system powersupply. The switch resistance is typi-cally 0.07Ω at 5V and only rises to0.08Ω at 3.3V. Output leakage cur-rent drops to 0.01µA when the outputis switched off.

The NMOS switch has no parasiticbody diode; therefore, no currentflows through the switch when it isturned off and the output is forcedabove the input supply voltage. (DMOSswitches have parasitic body diodesthat become forward biased underthese conditions.)

All operating modes are micropowerand the quiescent current automati-cally drops below 1µA when theswitch is turned off.

The combination of micropoweroperation and full protection makethese devices ideal for battery pow-ered applications such as palmtopcomputers, notebook computers, per-sonal digital assistants, instruments,handi-terminals, and bar-codereaders.

LTC1550/LTC1551: Low-Noise, Switched-Capacitor,Regulated Voltage Inverters

The LTC1550 and LTC1551 areswitched-capacitor voltage inverterswith onboard linear post regulators,designed to generate a low noise, regu-lated negative supply for use as biasvoltage for GaAs transmitters FETs inportable-RF and cellular-telephoneapplications. They operate from asingle 4.5V to 7V supply and providea fixed −4.1V or adjustable output

voltage, with low output ripple (1mVtypical) and a typical quiescent cur-rent of 5mA at VCC = 5V. An on-chiposcillator sets the charge pump fre-quency at 900kHz, away fromsensitive 400kHz–600kHz IF bands.The LTC1550-4.1 and the LTC1551-4.1 provide fixed −4.1V outputs,whereas the LTC1550 offers adjust-able output voltage. Both devicesinclude a TTL-compatible shutdownpin that drops supply current to 0.2µAtypical. The LTC1550 shutdown pinis active low (SD), and the LTC1551shutdown pin is active high (SD).

Only three external componentsare required for the fixed-voltageparts: two 0.1µF charge pump ca-pacitors and a 10µF filter capacitor atthe linear regulator output. Each ver-sion of the LTC1550/LTC1551 willsupply up to 20mA output currentwith guaranteed output regulation of±5%. All except the 8-pin versions ofthe LTC1550/LTC1551 include anopen-drain REG output, which pullslow to indicate that the output iswithin 5% of the set value. This al-lows external circuitry to monitor theoutput without additional compo-nents. The adjustable LTC1550requires two additional resistors toset the output voltage.

The LTC1550-4.1 and LTC1551-4.1 are available in 8-pin SO, 14-pinSO, and 16-pin SSOP packages. TheLTC1550 adjustable-output voltageversion is available in 14-pin SO and16-pin SSOP packages.

Low-Cost, High-Accuracy, 12-Bit Multiplying DAC Family

Linear Technology has introducedthe LTC8043, LTC8143, LTC7543,and LTC7541A, four 12-bit, currentoutput, 4-quadrant multiplying digi-tal-to-analog converters (DACs). Theyare superior pin-compatible replace-ments for the industry standardDAC-8043, DAC-8143, AD7543, andAD7541A. Low cost, tight accuracyspecs, low power, small size, and out-standing flexibility make these DACsexcel in process control, remote orisolated applications, and software-programmable gain, attenuation, andfiltering. Linear Technology’s parts

have improved accuracy and stabil-ity, tighter timing specs, reducedsensitivity to external amplifier VOS,lower output capacitance, and re-duced cost.

The LTC8043 comes in a small SO-8 or PDIP package and has aneasy-to-use serial interface. TheLTC8143 and LTC7543 come in 16-Pin SO or PDIP packages and featurea more flexible serial interface thatincludes an asynchronous Clear pin.The LTC8143 also has a serial dataoutput so users can daisy-chainmultiple DACs on a 3-wire bus. TheLTC7541A has a 12-bit wide parallelinterface and comes in 18-pin SO andPDIP packages.

These DACs boast unbeatable ac-curacy and stability, with INL andDNL of 1/2LSB MAX, 1/8LSB TYPover temperature. Gain Error is 1LSBMAX, eliminating adjustmentsin most applications. Gain andnonlinearity temperature coefficientsare typically better than 1 ppm/°Cand 0.1 ppm/°C respectively.

All four; parts are offered in theextended industrial temperaturerange and the LTC7541A is also avail-able in commercial grades.

Burst Mode is a trademark of LinearTechnology Corporation. , LTC and LTare registered trademarks used only toidentify products of Linear Technology Corp.Other product names may be trademarksof the companies that manufacture theproducts.

Information furnished by LinearTechnology Corporation is believed to beaccurate and reliable. However, LinearTechnology makes no representation thatthe circuits described herein will not infringeon existing patent rights.


AppleTalk is a registered trademark of Apple Computer, Inc.

© 1995 Linear Technology Corporation/ Printed in U.S.A./27K

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DESIGN TOOLSApplications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user to calculate circuit noiseusing LTC op amps, determine the best LTC op amp for a low noise application,display the noise data for LTC op amps, calculate resistor noise, and calculatenoise using specs for any op amp. Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be used with any version ofSPICE for general analog circuit simulations. The diskette also containsworking circuit examples using the models, and a demonstration copy ofPSPICETM by MicroSim. Available at no charge.

Technical Books1990 Linear Databook, Volume I — This 1440 page collection of data sheetscovers op amps, voltage regulators, references, comparators, filters, PWMs,data conversion and interface products (bipolar and CMOS), in both commer-cial and military grades. The catalog features well over 300 devices. $10.00

1992 Linear Databook Supplement — This 1248 page supplement to the1990 Linear Databook is a collection of all products introduced since then.The catalog contains full data sheets for over 140 devices. The 1992 LinearDatabook Supplement is a companion to the 1990 Linear Databook, whichshould not be discarded. $10.00

1994 Linear Databook, Volume III — This 1826 page supplement to the 1990Linear Databook and 1992 Linear Databook Supplement is a collection ofall products introduced since 1992. A total of 152 product data sheets areincluded with updated selection guides. The 1994 Linear Databook Volume IIIis a supplement to the 1990 and 1992 Databooks, which should not bediscarded. $10.00

Linear Applications Handbook • Volume I — 928 pages full of applicationideas covered in depth by 40 Application Notes and 33 Design Notes.This catalog covers a broad range of “real world” linear circuitry. In addition todetailed, systems-oriented circuits, this handbook contains broad tutorialcontent together with liberal use of schematics and scope photography.A special feature in this edition includes a 22 page section on SPICEmacromodels. $20.00

1993 Linear Applications Handbook • Volume II — Continues the streamof “real world” linear circuitry initiated by the 1990 Handbook. Similar in scopeto the 1990 edition, the new book covers Application Notes 41 through 54 andDesign Notes 33 through 69. Additionally, references and articles from non-LTC publications that we have found useful are also included. $20.00

Interface Product Handbook — This 424 page handbook features LTC’scomplete line of line driver and receiver products for RS232, RS485,RS423, RS422, V.35 and AppleTalk applications. Linear’s particularexpertise in this area involves low power consumption, high numbers ofdrivers and receivers in one package, mixed RS232 and RS485 devices, 10kVESD protection of RS232 devices and surface mount packages.

Available at no charge.

SwitcherCAD Handbook — This 144 page manual, including disk, guidesthe user through SwitcherCAD—a powerful PC software tool which aids in thedesign and optimization of switching regulators. The program can cut days offthe design cycle by selecting topologies, calculating operating points andspecifying component values and manufacturer's part numbers. $20.00

1995 Power Solutions Brochure, First Edition — This 64 page collectionof circuits contains real-life solutions for common power supply designproblems. There are over 45 circuits, including descriptions, graphs andperformance specifications. Topics covered include PCMCIA power manage-ment, microprocessor power supplies, portable equipment power supplies,micropower DC/DC, step-up and step-down switching regulators, off-lineswitching regulators, linear regulators and switched capacitor conversion.

Available at no charge.

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