linear technology linear technology · plague the dual rechargeable battery systems found in most...

DECEMBER 1996 VOLUME VI NUMBER 4

IN THIS ISSUE . . .COVER ARTICLELTC®1409/LTC1415 High Speed,Low Power 12-Bit ADCs ............ 1Kevin R. Hoskins

Issue Highlights ...................... 2LTC in the News ....................... 2DESIGN FEATURESLTC1553 Synchronous RegulatorController Powers Pentium® Proand Other Big Processors......... 3Y.L. Teo, S.H. Lim and Craig Varga

The LT®1575/LT1577 UltraFast™Linear Regulator ControllersEliminate Bulk Output Capacitors................................................ 9Anthony Bonte

LTC1479 PowerPath™ ControllerSimplifies Portable PowerManagement Design............... 14Tim Skovmand

New Rail-to-Rail Amplifiers:Precision Performance fromMicropower to High Speed.............................................. 18William Jett and Danh Tran

High Efficiency, Low DropoutLithium-Ion Battery ChargerCharges Up to Five Cells at4 Amps or More ...................... 23Fran Hoffart

LTC1069-X: a New Family of 8thOrder Monolithic Filters ........ 27Nello Sevastopoulos and Philip Karantzalis

DESIGN IDEASMicropower ADC and DAC in SO-8Give PC 12-Bit Analog Interface.............................................. 30Kevin R. Hoskins

High Efficiency, Low Power,3-Output DC/DC Converter ...... 33John Seago

Synchronizing LTC1430sfor Reduced Ripple ................ 34Craig Varga

DESIGN INFORMATIONNew Voltage References AreSmaller and More Precise ...... 35John Wright

New Device Cameos ................ 38Design Tools .......................... 39Sales Offices .......................... 40

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,LinearView, Micropower SwitcherCAD, PowerPath and UltraFast are trademarks of Linear Technology Corporation.Pentium is a registered trademark of Intel Corp. Other product names may be trademarks of the companies thatmanufacture the products.

continued on page 20

LTC1409/LTC1415High Speed, Low Power12-Bit ADCsHigh Speed, Less PowerExpanding the family of high speed,low power dissipation 12-bit ADCsthat began with the LTC1410, LTCrecently introduced the LTC1409 andthe LTC1415, increasing the numberof high speed, low power ADCs avail-able to designers of high speed dataacquisition systems. With these newchoices, a designer can pick an ADCthat is optimized for his or her par-ticular application. The LTC1409 andLTC1415 are ideal solutions forapplications such as ADSL, HDSL,modems, direct downconversion,CCD imaging, DSP-based vibra-tion analysis, waveform digitizers andmultiplexed systems.

The new LTC1409 and LTC1415have much lower power dissipationand higher performance than other12-bit ADCs currently on the mar-ket. The LTC1409 and LTC1415dissipate just 80mW and 60mW, re-spectively. Package size is also a majoradvantage of the LTC1409 andLTC1415, since they are available in

9041CTL 5141CTL

tuphguorhTnoisrevnoC spsk008 spsM2.1

noitapissiDrewoPwoL (Wm08 ± )ylppusV5 )seilppusV5+(Wm06

sedoMPEELSdnaPAN

—egakcaPllamSPOSSnip-82

Table 1. LTC1409/LTC1415 key features

by Kevin R. Hoskins

28-pin SO and SSOP packages. Someof the other key features of these newdevices are shown in Table 1.

PerformanceEnhancing Features

Differential S/Hand Wideband CMRRLike the LTC1410, the LTC1409 andLTC1415 have fully differential inputs.The differential inputs have a verygood CMRR: 60dB or better over a 0-to-10MHz bandwidth. The bandwidthof the input sample-and-hold is typi-cally 30MHz. All of these featurescombine to create improved solutionsfor present and future data- or signal-acquisition systems.

Figure 1 is a block diagram of theLTC1409 and LTC1415. The out-standing conversion speed andaccuracy of these parts is the result ofthe high performance differentialsample-and-hold and the extremely

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

2 Linear Technology Magazine • December 1996

EDITOR'S PAGE

Issue HighlightsLTC in the News...This issue of Linear Technology is

loaded with new products. Our coverarticle introduces a pair of new highspeed, low power 12-bit ADCs, theLTC1409 and LTC1415. These newparts offer conversion rates of 800kspsand 1.2Msps respectively and havemuch lower power dissipation thanother ADCs currently on the market.This makes them ideal for applica-tions such as ADSL, HDSL, modems,CCD imaging and the like.

Power products feature promi-nently in this issue: the LTC1553, asynchronous switching regulator con-troller, is designed to convert 5V/12V“silver box” supplies to the low volt-ages required by the Intel Pentiumprocessor and other big CPUs. Anonboard 5-bit DAC conforms to therequirements of the Intel Pentium Proprocessor power supply specification,making the LTC1553 a perfect fit inthese applications. The LTC1553, likethe LTC1430, provides current-limitand short-circuit protection withoutthe use of an external sense resistor.

Also featured in this issue is thenew LT1575/LT1577 family of con-troller ICs. These new, easy-to-usedevices drive discrete N-channelMOSFETs as source followers to pro-duce extremely low dropout, UltraFasttransient response regulators. Thesecircuits completely eliminate expen-sive tantalum or bulk electrolyticcapacitors in the most demandingmicroprocessor applications. For ex-ample, a 200MHz Pentium processorcan operate with only the twenty-four1µF ceramic capacitors that Intel al-ready requires for the microprocessor.

Another power control device forcomputer applications is introducedin this issue: The LTC1479PowerPath™ controller eliminatespower-management nightmares thatplague the dual rechargeable batterysystems found in most notebook com-puters and other portable equipment.

Finally, we have the LT1620, an ICdesigned to be used with a currentmode PWM controller (such as theLTC1435) to increase the output volt-

age range and optimize the circuit forbattery charging applications. In thisarticle, the LT1620 and LTC1435 arefeatured in a high current, highperformance constant-voltage/con-stant-current battery charger forlithium-ion and other battery types.

Several new additions to LTC’s fam-ily of rail-to-rail op amps are presentedherein. These include the LT1498/LT1499 C-Load™ op amps, whichfeature a 10MHz gain-bandwidthproduct, 4V/µs slew rate and theability to drive 10,000pF; the lowcurrent LT1466–69, with quiescentcurrent of only 50µA; and the highprecision LT1218A/LT1219A, featur-ing VOS trimmed to 100µV max.

Filters are represented in this issue

by the LTC1069-X, a family ofsemicustom filters that can integrateany single 8th order or dual 4th orderclassical filter approximation, or anyapplication-specific filter response, inan SO-8 package.

This issue includes a modest col-lection of Design Ideas: a 12-bit analoginterface for the PC, based on theLTC1298 ADC and LTC1446 DAC,with sample code; a low power, 3-output DC/DC converter built aroundthe LTC1435; and a technique forsynchronizing two LTC1430 buckregulators for reduced output ripple.We conclude with Design Informationon the LT1460 and LT1236 voltagereferences, and a page of New DeviceCameos.

Linear Technology Corp.Reports Steady Sales andProfits for Q1 1997 in a TightSemiconductor EnvironmentLinear Technology Corporation’s netsales for its first quarter, endedSeptember 29, 1996, were$90,063,000, an increase of 4% overthe first quarter of the previousyear.

The Company also reported netincome for the quarter of$31,358,000 or $0.40 per share, anincrease of 3% over $30,520,000 or$0.39 per share, reported for thefirst quarter of last year. A cashdividend of $0.05 per share will bepaid on November 13, 1996 to share-holders of record on October 25,1996.

According to Robert H. Swanson,president and CEO, “Despite a slug-gish semiconductor environment,we were able to maintain our netsales and profitability. Our returnon sales of approximately 35% con-tinues to lead the industry. Wegenerated approximately $10 mil-lion in cash even after payingapproximately $16 million to re-purchase shares of our own stock.However, in the short term we con-

tinue to be in an unpredictable en-vironment whereby reduced backlogand shorter lead times cause thebusiness to be very dependent onorders that are received and shippedin the same quarter.”

A detailed look at the best-per-forming stock index of them all—theNasdaq 100—reveals that LinearTechnology Corp. is ranked highagain this year among the best-run,most effective and highest-valuedcompanies in the nation. Coming inwell above such familiar names asAltera (70th), Maxim (71st), Micron(83rd) and Cirrus Logic (84th), Lin-ear Technology Corp. is listed 52ndon the Nasdaq 100, with a marketcapitalization of more than $2.2billion as of press time.

The financial performance of LTChas been so good that one major-fund manager who prefers bonds tostocks says that he would make anexception for LTC. Quoted byKathleen Gallagher in her syndi-cated column “Street Smart,”Thomas M. Wargin, president of Lib-erty LaSalle Financial Group, Inc.,says that he “likes Linear Technol-ogy, a specialty chip maker whosestock price dropped (this summer).He expects its earnings to grow 21%annually for the next five years.”

Linear Technology Magazine • December 1996 3

DESIGN FEATURES

LTC1553 Synchronous RegulatorController Powers Pentium® Proand Other Big Processors

by Y.L. Teo, S.H. Lim and Craig VargaIntroductionOver the past few years, the operatingvoltages of Pentium class and othermodern microprocessors havedropped from 5V to 3.3V and below,while operating currents have steadilyincreased. Voltage regulation require-ments have also tightened as the safetymargin between proper operation andchip destruction has decreased alongwith feature size. To complicate mat-ters further, the newest Pentium Proprocessors from Intel® require a digi-tally adjustable power supply, so thatthe processor itself can determine thepower supply voltage. The “silver box”power supplies provide only 5V/12V,with the exception of a few suppliesalso capable of delivering 3.3V. Dueto the extreme accuracy and excep-tionally fast load transient responserequired by today’s processor sup-plies, the supply has been forced ontothe computer’s motherboard. To fitthis niche, Linear Technology intro-duces the LTC1553, a synchronousswitching regulator controller de-signed to convert the 5V/12V “silverbox” rails to the lower 1.80V–3.5Vrequired by the CPU. An onboard 5-bit DAC conforms to the requirementsof the Intel Pentium Pro processorpower supply specification, makingthe LTC1553 a perfect fit in theseapplications.

The LTC1553 is the newest mem-ber of the LTC switching regulatorfamily. It shares many performancefeatures with the popular LTC1430,including excellent (±1%) output regu-lation over temperature, line voltageand load current variations. TheLTC1553, like the LTC1430, providescurrent-limit and short-circuit pro-tection without the use of an externalsense resistor. This is accomplishedby measuring the voltage drop acrossthe external high-side MOSFET dur-

ing its on-time. To compliment themain voltage-feedback loop, theLTC1553 includes two additional feed-back loops to provide good large-signaltransient response. The LTC1553

adds additional internal circuits toconform to the Intel Pentium Pro pro-cessor power converter requirementswhile minimizing the number of ex-ternal components. An on-chip 5-bit

4DIV 3DIV 2DIV 1DIV 0DIV )CDV(0 1 1 1 1 *0 1 1 1 0 *0 1 1 0 1 *0 1 1 0 0 *0 1 0 1 1 *0 1 0 1 0 *0 1 0 0 1 *0 1 0 0 0 *0 0 1 1 1 *0 0 1 1 0 *0 0 1 0 1 08.10 0 1 0 0 58.10 0 0 1 1 09.10 0 0 1 0 59.10 0 0 0 1 00.20 0 0 0 0 50.21 1 1 1 1 UPCoN1 1 1 1 0 1.21 1 1 0 1 2.21 1 1 0 0 3.21 1 0 1 1 4.21 1 0 1 0 5.21 1 0 0 1 6.21 1 0 0 0 7.21 0 1 1 1 8.21 0 1 1 0 9.21 0 1 0 1 0.31 0 1 0 0 1.31 0 0 1 1 2.31 0 0 1 0 3.31 0 0 0 1 4.31 0 0 0 0 5.3

noisnapxeerutufrofdevreseR*

Table 1. Output voltage vs VIDn code


DESIGN FEATURES

digital-to-analog converter (DAC) pro-vides output voltages conforming toIntel’s specifications. This allows theLTC1553 to read the code sent by theprocessor and provide it with the re-quested voltage. The LTC1553 alsoprovides a power-good indication(PWRGD) to the system. There is alsoan on-chip overvoltage protection cir-cuit that latches the regulator in anoff state if the output voltage everrises 15% or more abovethe DAC-requested voltage. Over-temperature protection is availablewith only two external components (aresistor and a thermistor) connectedto the OUTEN pin.

In applications with other proces-sors, the four DAC inputs can berouted to a jumper block, zero ohmresistors or a DIP switch, or hardwired, to set the desired output volt-age. This allows the output voltage tobe programmed easily in steps whileeliminating the need to stock an as-sortment of precision resistors. Thisflexibility in output voltage setting ischeap insurance against last-minutepower supply voltage changes by mi-croprocessor manufacturers.

LTC1553 OverviewThe LTC1553 runs at a fixed switch-ing frequency, nominally 300kHz,without any external oscillatorcomponents. The on-chip, 5-bit digi-tal-to-analog converter (DAC) allowsthe output voltage to be adjustedfrom 1.80V to 3.5V, as shown in Table1. Voltage mode control eliminatesthe need for a current sense resistor.

Current limiting is maintained bysensing the voltage drop across theRDS(ON) of the high-side MOSFET. Theoutput enable pin employs a multi-level voltage threshold scheme thatpermits overtemperature sensing aswell as providing the normal enablefunction.

The LTC1553 is designed to beused with an all-N-channel MOSFET,synchronous buck regulator topol-ogy. The gate drive is able tosignificantly exceed the main powersupply voltage. This allows the high-side N-channel MOSFET(s) to be fullyenhanced, ensuring low RDS(ON) andmaximum efficiency. The driver out-puts can source and sink enoughcurrent to drive multiple, paralleledpower MOSFETs if desired, in orderto obtain very low conduction losses.Low loss operation eliminates the needfor a heat sink in most applicationsand results in very high efficiency. Asoft-start circuit is included on thechip, permitting the rate of rise of theoutput voltage at turn-on to becontrolled.

Internal DAC andOutput Voltage AccuracyAn on-chip 5-bit DAC is used to setthe output voltage. No external feed-back resistors are required. Five TTLinputs (VID0, VID1, VID2, VID3 andVID4) program the internal DAC (seeTable 1). Each of the VIDn pins has aninternal pull-up resistor to ensure ahigh state if it is not connected. Whenall the VIDn pins are in the high state,the LTC1553 shuts down, forcing theoutput voltage to zero and droppingthe quiescent current to approxi-mately 150µA. The ten lowest voltagecodes are disabled at this time, allow-ing for future expansion. The DACaccuracy, initial reference voltage tol-erance and internal feedback resistortolerances result in a maximum ini-tial output voltage error of ±1% of theselected output voltage. The line andload regulation plus temperature driftover the 0°C to 70°C temperaturerange will contribute another ±1% tothe output error budget. This gives atotal static operating error of lessthan ±2%, providing sufficient head-

room (3%) for the dynamic responseto remain within a ±5% output voltagetolerance, while still requiring areasonable amount of outputcapacitance.

External MOSFET DriversThe on-chip output drivers are pow-ered from the PVCC pin, which isspecified with a 20V maximum sup-ply voltage. The PVCC supply shouldbe at least 5V higher than the mainVIN supply to allow the G1 outputdriver to fully enhance a high-sideN-channel MOSFET. A major advan-tage of the LTC1553 over somecompeting devices results from itsCMOS implementation: full rail-to-rail gate drive provides as much as 2Vmore drive than would a bipolar de-sign. This results in a 20%–25% lowerRDS(ON) for a given MOSFET. Figure 1ashows a typical LTC1553 circuit withPVCC powered from a 12V supply andthe main VIN powered from a highpower 5V supply. This provides 7V ofenhancement for the high-side switch.The amount of current required of the12V supply varies with the amount ofMOSFET gate capacitance and willtypically be less than 50mA. If a 12Vsupply is not available, the gate drivevoltage can be generated with a simplecharge-pump. Figure 1b shows a dou-bler charge-pump used to generatethe PVCC voltage in a 5V-only system.A tripler charge-pump (Figure 1c) maybe used in circuits where a highervoltage is required to fully enhancethe external MOSFETs.

1552_01a.eps

LTC1553

CVCC CPVCC

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1N5248B* 18V

*OPTIONAL FOR VIN > 5V

VIN

1N5817

CIN

CO

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0.1µF

LO

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Figure 1a. Typical LTC1553 circuit with PVCCpowered from a 12V supply and main VINpowered from a high power 5V supply

Figure 1b. A doubler charge pump generatesthe PVCC voltage for a system powered from asingle 5V supply.


DESIGN FEATURES

Note that if VIN is 10V or higher, astandard doubling charge-pump willcause the PVCC pin to exceed its 20Vlimit. Such circuits should use analternate 17V charge-pump circuit(see Figure 1d). In any circuit wheretransient spikes at the PVCC pin mayapproach the 20V maximum rating,an 18V Zener diode is recommendedfrom PVCC to GND (Figures 1b–1d).

Multiple Feedback LoopsImprove Transient ResponseThe LTC1553 uses a standard voltagefeedback loop to control the outputvoltage (see Figure 2). The erroramplifier, ERR, compares the resis-tor-divided output voltage at FB tothe internal reference voltage, VREF.This reference is controlled by theinternal DAC. The resulting error volt-age is amplified by the error amplifier.A pulse width modulated (PWM) sig-nal is generated by comparing theerror signal with a sawtooth wave-form from the internal oscillator. ThisPWM and its complement signal drivethe gates of power switches Q1 andQ2, respectively. Feedback loop com-pensation is set with an externalcompensation network at the COMPpin. Voltage mode control eliminatesthe need for a high loss, high cost,external sense resistor required by atypical current mode design.

In addition to the main feedbackloop, the LTC1553 also includes two

additional “safety belt” comparators(MIN and MAX in Figure 2). In gen-eral, a control loop’s bandwidth, andhence its slew rate, is limited bystability considerations. In someinstances, it may be desirable to havethe ability to respond to events fasterthan the error amplifier is capable ofslewing. The MIN/MAX comparatorsprovide this capability. These twocomparators help to prevent extremeoutput perturbations with fast loadcurrent transients, while allowing themain feedback loop to be optimallycompensated for stability. The MAXloop responds when the outputexceeds the set point by more than5%, forcing the duty cycle to 0% andholding Q2 on until the output dropsback into the acceptable range. Simi-larly, if the output voltage sags 5%below the set point, the MIN loopkicks in, forcing the Q1 to 85% dutycycle until the output recovers. The95% maximum duty cycle ensuresthat the gate drive charge-pump (ifused) is refreshed every cycle. Theresponse times of the MIN and MAXcomparators are controlled to preventthem from triggering on noise spikes.

Soft-Start and Current LimitJust one external capacitor is neededat the SS pin to set the soft-start time.During start-up, Q1 and Q2 areswitched off until the input voltagehas risen to the threshold of an inter-

nal undervoltage lockout circuit. Thesoft-start capacitor is then chargedby an internal 10µA current source.The soft-start function overrides theerror amplifier and takes control ofthe pulse width modulator. As the SSpin voltage rises, the LTC1553’s G1duty cycle increases slowly until theoutput voltage is in regulation, atwhich point control is transferred tothe voltage feedback loop.

A significant advantage of employ-ing voltage mode control in theLTC1553 is the elimination of thecurrent sense resistor found in mostother designs. With this resistor goesthe simple means of providing over-current protection. Instead, theLTC1553 sets the output current limitby monitoring the voltage drop acrossthe RDS(ON) of the high-side MOSFET,Q1, during its ON state (see Figure 3).The current limit is controlled by set-ting the maximum voltage dropallowed across Q1 with a single exter-nal resistor, RIMAX, at the IMAX pin. Aninternal 180µA current sink forces avoltage across RIMAX. This voltage iscompared to the voltage drop acrossQ1 during its on-time. The IFB pinconnects to the source of Q1 andKelvin senses the voltage drop acrossQ1. For VIN = 12V, a 15V Zener diode(1N5245B or equivalent) will preventvoltage spikes at IFB from exceedingthe maximum voltage rating. The cur-rent limit is designed to engage slowly

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Figure 1c. A tripler charge pump provides more gate drive for theexternal MOSFETs.

Figure 1d. An alternate 17V charge pump circuit prevents PVCC fromexceeding its 20V limit.


DESIGN FEATURES

under mild current overloads,allowing the LTC1553 circuit to with-stand momentary overloads withoutseriously af fecting the outputregulation.

When an overcurrent condition issensed, the output current compara-tor, CC, starts pulling current out ofthe external soft-start capacitor. Thisstarts lowering the duty cycle andregulating the output current. Undersevere overloads or output short-cir-cuit conditions, a “hard” current-limitcircuit is activated. An internal switch(MHCL in Figure 2) pulls down the SSpin immediately, stopping all switch-ing and preventing damage to theoutput components. After a short timedelay, the SS pin is released and theLTC1553 reruns a soft-start cycle,

attempting to restart. If the over-current condition is still present, thecycle repeats until the fault is re-moved. If desired, current limit can bedisabled by floating IMAX and tying IFBto VCC.

Power-Good andOvervoltage SignalsThe LTC1553 provides a power goodsignal (PWRGD) to the host system toindicate that the output voltage iswithin ±5% of the set voltage. PWRGDis valid whenever both the MIN andMAX comparators are inactive. Aninternal time delay is designed to pre-vent noise at the sense pin fromtoggling PWRGD unnecessarily. Afterthe output has settled to within ±5%of the rated output for more than

300µs, PWRGD is set to a logic high.Similarly, PWRGD will go low onlyafter the output is out of regulationfor more than 100µs.

Severe overvoltage faults at theoutput will trigger the FAULT flag. Ifthe output voltage rises 15% higherthan the programmed voltage, G1 andG2 will be disabled and the FAULTpin will go low. FAULT can be used tofire an external crowbar SCR or MOS-FET to pull down the errant supplyand protect the CPU from damage.When the LTC1553 detects an over-voltage, the fault flag’s low state islatched, holding the regulator off. Torestore normal operation, the OUTENpin must be toggled or the input powerremoved and then restored. The mostlikely cause of an overvoltage fault is

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115% VREF

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10%

FC

Figure 2. LTC1553 block diagram


DESIGN FEATURES

a shorted high-side MOSFET. There-fore, activating some form of externalclamp is preferable to depending onthe LTC1553 to shut the supply down,as the controller will be unable toturn off a shorted FET.

Overtemperature ProtectionThe OUTEN pin is an active-high digi-tal input that enables the G1 and G2MOSFET driver outputs. WhenOUTEN is pulled to a TTL low level,both G1 and G2 pull to ground, shut-ting off the external MOSFETs andleaving the output in a high imped-ance state. OUTEN is designed withmultiple thresholds to allow it to alsobe used for overtemperature protec-tion. The power MOSFET operatingtemperature can be monitored withan external negative temperaturecoefficient (NTC) thermistor mountednext to the external MOSFET that isexpected to run the hottest—usuallythe high-side device, Q1. Electrically,the thermistor should form a voltagedivider with another resistor (R1)connected to VCC. Their midpoint isconnected to OUTEN (see Figure 4).As the thermistor temperature

increases, the OUTEN pin voltage isreduced. Under normal operatingconditions, the OUTEN pin shouldstay above 2V. All circuits will functionnormally and the OT (overtem-perature) pin will remain in a highstate. If the temperature gets abnor-mally high, the OUTEN pin voltagewill eventually drop below 2V. OT willswitch to a logic low, providing anovertemperature warning to the sys-tem. As OUTEN drops below 1.7V, theLTC1553 disables both FET drivers.This shuts the driver supply down,preventing any further heating. If theOUTEN pin is pulled below 1.2V, theLTC1553 will enter shutdown mode.All internal switching stops, theCOMP, SS, OT and PWRGD pins pullto ground and the quiescent currentis reduced to 150µA. This residualquiescent current keeps the ther-mistor sensing circuit at OUTEN aliveto allow the circuit to recover once itcools down. If the overtemperatureprotection circuit is not required, theOUTEN pin can be connected directlyto a TTL compatible signal.

Oscillator SynchronizationThe LTC1553’s internal oscillator canbe synchronized to an external clockfrequency higher than 300kHz. Thisis accomplished by connecting a TTL-level external clock signal to OUTEN.Note that if OUTEN is used for syn-chronization, it cannot be used fortemperature monitoring. Also, if theoperating frequency is forced sub-stantially higher than 300kHz, the

gain of the main feedback loop willincrease and the compensation net-work may have to be readjusted foroptimum performance.

Typical ApplicationA typical application for LTC1553 isconverting 5V to 1.8V–3.5V in a Pen-tium Pro processor based personalcomputer. The supply may be in theform of a voltage regulator module(VRM) or may be implemented di-rectly on the motherboard. The outputis used to power the Pentium Proprocessor and the input is taken fromthe system’s 5V supply. The circuitshown in Figure 6 provides 1.80V–3.5V at 14A while maintaining outputregulation within ±1%. The outputvoltage is determined by connectingthe five DAC inputs to the VID pins ofthe processor. The power MOSFETsare sized to minimize board spaceand allow operation without the needof a heat sink. With proper airflow,ambient temperature conditions ofup to 50˚ Celsius are acceptable. Typi-cal efficiency is above 90% from 1A to10A at 3.3V out. (see Figure 7). Achiev-ing higher output currents fromLTC1553 based designs is simply amatter of selecting appropriate MOS-FETs and passive components.

It pays to look at the regulatordesign from two perspectives: electri-cal and thermal. Most processorapplications operate at average cur-rents that are approximately 80% orless of the specified peak current. Assuch, the thermal design can be based

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NORM

AL O

PERA

TION

0V

Figure 3. Current-limit setting forthe LTC1553 Figure 4. Using the OUTEN pin for overtemperature protection

Figure 5. The OUTEN pin provides threethreshold levels for temperature monitoring.


DESIGN FEATURES

on the lower current level. Highercurrents, while present, are typicallynot of sufficient duration to signifi-cantly heat the power devices. Thedesign does, however, need to becapable of delivering the peak currentwithout entering current limit orresulting in device failures. Keep inmind that the power dissipation in aresistive element, such as a MOSFET,varies as the square of load current.As such, raising the load current from80% to 100% translates to approxi-mately 56% more power dissipation(1/0.82). Designing for this higherthermal load results in a huge, andmost likely unnecessary, design mar-gin. A good understanding of yoursystem requirements can result insubstantial savings in the size andcost for the power supply.

RIMAX sets current limit to thedesired level. Add one-half of the in-ductor ripple current to the maximumload current to determine the peakswitch current. Multiply this currentby the maximum on-resistance of theselected MOSFET switch to deter-mine the minimum current limitthreshold voltage. It’s a good idea toadd at least a 10% margin to thislimit. Also, be sure to use the hot on-resistance of the MOSFET. A multiplierof about 1.4 times the room tempera-ture RDS(ON) should be used to

determine the hot resistance.In the case o f two paral le lMTD20N03HDLs (Q1A and Q1B), thecold resistance is approximately0.035Ω each; therefore, assume thehot resistance to be approximately0.050Ω. Divide this by two becausethe FETs are in parallel. The thresh-old voltage is programmed bymultiplying the IMAX pin’s sink cur-rent by the value of RIMAX. Since wenow can determine the requiredthreshold, we need to calculate thevalue of RIMAX. Use the specified mini-mum sink current, 150µA, to calculatethe resistor value.

The soft-start time is programmedby the 0.01µF cap connected to theSS pin. The larger the value of thiscapacitor, the slower the turn-onramp.

Inductor LO is sized to handle thefull load current, up to the onset ofcurrent limit, without saturating. Avalue of between 2µH and 3µH isadequate for most processor supplydesigns. Be careful not to overspecifythe inductor. The inductor need notretain its no-load inductance up tothe current-limit threshold. If theinductor still retains on the order of25% to 30% of its initial inductanceunder worst-case short-circuit cur-rent conditions, the circuit shouldprove reliable. However, you do want

to ensure that approximately 60% to75% of the initial inductance isretained at nominal full load. Exces-sive inductance roll-off will result inhigher than expected output ripplevoltage at high loads, along with in-creased dissipation in the power FETsand the inductor itself.

Proper loop compensation is criti-cal for obtaining optimum transientresponse while ensuring good stabil-ity margins. The compensationnetwork shown here gives goodresponse when used with the induc-tor and the output capacitors valuesshown in Figure 6. Several low ESRcapacitors are placed in parallel toreduce the total output ESR, result-ing in lower output ripple andimproved transient performance.

G1

IFB

G2

1552_06.eps

Pentium Pro Processor SYSTEM LTC1553

COMP

Q1A, Q1B, Q2: MOTOROLA MTD20N03HDL

LO 2.0µH/18A

VCC

VIN = 5V

VOUT

PWRGD

Q1A, Q1B (2 IN PARALLEL)

Q2

DALE NTHS-1206N02

CONNECTING VID0-VID4

TO DIP SWITCH TO SET VOUT

SS GND PGND SENSE

IMAX PVCC

5.6k5.6k RIMAX5.6k

1.8k

5V C0 2310µF

7 × 330µF

0.1µF

0.1µF

1N581710µF CIN 990µF 3 × 330µF

VID0–VID4

OUTEN

C1 100pF

CSS 0.01µF

0.1µFCC 0.01µF

RC 20k

FAULTOT

+

Figure 6. Typical 5V to 2.1V–3.5V/14A LTC1553 application

LOAD CURRENT (A)

0

30

40

50

60

10

20

70

80

90

100

EFFI

CIEN

CY (%

)

100

1552_07.eps

0.1 1 10

Figure 7. Efficiency plot for Figure 6’s circuit



DESIGN FEATURES

The LT1575/LT1577 UltraFast LinearRegulator Controllers Eliminate BulkTantalum/Electrolytic Output Capacitors

by Anthony BonteIntroductionThe current generation of micropro-cessors places stringent demands onthe input supply that powers the pro-cessor core. These microprocessorscycle load current from near zero toapproximately 5 amps in tens of nano-seconds. Output voltage tolerancesas low as ±100mV include transientresponse as part of the specification.Some microprocessors require only asingle input supply voltage to operateboth the core and the I/O circuitry.Other higher performance processorsrequire separate power supply volt-ages for the processor core and theI/O circuitry. These requirementsdemand very accurate, very high speedregulator circuits.

Solutions employed previouslyinclude monolithic 3-terminal linearregulators, PNP transistors driven bylow cost control circuits and simplebuck converter switching regulators.The 3-terminal regulator achieves ahigh level of integration, the PNP-driven regulator achieves low dropoutvoltage performance and the switch-ing regulator achieves high electricalefficiency.

However, the common trait mani-fested by these solutions is a transientresponse time measured in microsec-onds. This translates into an expensive

regulator output decoupling capaci-tor scheme. This scheme requiresseveral hundred microfarads of verylow ESR bulk capacitance compris-ing multiple capacitors surroundingthe CPU. This bulk capacitance is inaddition to the ceramic decouplingcapacitor network that handles thetransient load response during thefirst few hundred nanoseconds. Theceramic capacitors also act as a highfrequency decoupling network to mini-mize noise associated with fast, highcurrent spikes. The cost of the outputdecoupling capacitors is a significantpercentage of the total power supplycost and the bulk tantalum/electro-lytic capacitors comprise the majorityof the capacitor cost.

New LTCRegulator ControllersThe LT1575/LT1577 family of single/dual controller ICs are new, easy-to-use devices that drive discreteN-channel MOSFETs as source fol-lowers to produce extremely lowdropout, UltraFast™ transient re-sponse regulators. These circuitsachieve superior regulator bandwidthand transient load performance, andcompletely eliminate expensive tan-talum or bulk electrolytic capacitors

in the most demanding micropro-cessor applications. For example, a200MHz Pentium® processor can op-erate with only the twenty-four 1µFceramic capacitors that Intel alreadyrequires for the microprocessor. Us-ers realize significant savings becauseall additional bulk capacitance is re-moved. The additional savings ofinsertion cost, inventory cost andboard space are readily apparent.

Precision-trimmed adjustable andfixed-output voltage versions accom-modate any required microprocessorpower supply voltage. Dropout volt-age can be user defined via selectionof the N-channel MOSFET RDS(ON).The only output capacitors requiredare the high frequency ceramicdecoupling capacitors. The regulatorresponds to transient load changes ina few hundred nanoseconds—a greatimprovement over regulators thatrespond in many microseconds. Theceramic capacitor network generallyconsists of ten to twenty-four 1µFcapacitors, depending on individualmicroprocessor requirements. TheLT1575/LT1577 family also incorpo-rates current limiting at no additionalsystem cost, provides on/off controland can provide overvoltage protec-tion or thermal shutdown with the

1

2

3

4

8

7

6

5

SHDN

VIN

GND

OUT

IPOS

INEG

GATE

COMP

C2 1µF

C5 330µF

5V

GND1575/77 TA01

VOUT 3.5V 5A

R2 5Ω

R1 7.5k

12V LT1575-3.5

C4 1000pF

FOR T > 45°C: C6 = 24 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS. PLACE C6 IN THE MICROPROCESSOR SOCKET CAVITY

FOR T ≤ 45°C: C6 = 24 × 1µF Y5V CERAMIC SURFACE MOUNT CAPACITORS.

*Q1 IRFZ24

+

C3 10pF

C6* 24µF

Figure 1. UltraFast transient response 5V to 3.5V, low dropout regulator

2A/DIV

50mV/DIV

0

200µs/DIVI = 0.2A to 5A

Figure 2. Transient response for 0.2A–5Aoutput load step


DESIGN FEATURES

addition of a few simple external com-ponents. The LT1575 is available in8-pin SO or PDIP and the LT1577 isavailable in 16-pin narrow-body SO.

Figure 1 shows the basic regulatorcontrol circuit. The input voltage is astandard 5V “silver box” and the out-put voltage is set to 3.5V, the PentiumP54 VRE microprocessor supply volt-age. The typical maximum outputcurrent is about 5A in most Pentiummicroprocessor applications. The out-put capacitor network consists of onlytwenty-four inexpensive 1µF ceramic,surface mount capacitors. Proper lay-out of this decoupling network iscritical to proper operation of thiscircuit. Consult Linear TechnologyApplication Note 69: Using the LT1575Linear Regulator Controller, for detailson board layout.

The photo in Figure 2 shows thetransient response performance foran output load current step of 0.2A to5A. The main loop compensation inFigure 1’s regulator circuit is pro-vided by R1 and C4 at the COMP pin.Capacitor C3 introduces a high fre-quency pole and provides adequategain margin beyond the unity-gaincrossover frequency of 1MHz. Thiscompensation network limits over-shoot/undershoot to 50mV underworst-case load transient conditions.With a 1% specified worst-case out-put voltage tolerance, the 100mVoutput voltage error budget for a P54VRE microprocessor is easily met withproduction margin to spare. All bulktantalum/electrolytic capacitors arecompletely eliminated.

The discrete N-channel MOSFETchosen is a low cost InternationalRectifier IRFZ24 or equivalent. Theinput capacitance is approximately1000pF with VDS = 1V. The specifiedon-resistance is 0.1Ω at room tem-perature and about 0.15Ω at 125˚C.At 7A output current, the dropoutvoltage is only 1.05V. This eases therestriction on local input decouplingcapacitor requirements because sig-nificant droop in the typical 5V inputsupply voltage is permitted beforedropout voltage operation is reached.(Note that 5V supply tolerance

1

2

3

4

8

7

6

5

SHDN

VIN

GND

OUT

IPOS

INEG

GATE

COMP

C2 1µF

C5 330µF

5V

GND

1575/77 TA12

VOUT 3.3V 5A

R2 5Ω

R3* 0.007Ω

R1 3.9k

12V LT1575-3.3

C4 1500pF

C1 1µFRESET

R3 IS MADE FROM “FREE” PC BOARD TRACE C6 = 12 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS. PLACE C6 IN THE MICROPROCESSOR SOCKET CAVITY

*

**

Q2 VN2222L

Q1 IRFZ24

+

C3 10pF

C6** 12µF

restrictions are typically limited by a±5% tolerance so that 5V logic sys-tems will operate correctly.) However,a simple LC input filter can eliminatethe need for large input bulk capaci-tance at the regulator 5V supply foradditional system cost savings.

Figure 3 shows a more completesystem configuration that incorpo-rates current limiting and currentlimit time-out with latch-off. Currentlimit is incorporated for no additionalsystem cost by manufacturing thecurrent limit resistor from a Kelvin-sensed section of pc board trace. Inthis example, current limit is set to7A. A capacitor from the SHDN pin toground sets a fault condition time-out period that latches off the drive tothe external MOSFET if the time-outperiod is exceeded. The regulator isreset by pulling the SHDN pin low.The output voltage in this applicationis set to 3.3V. The ±5% tolerancepermitted in 3.3V systems translatesto a ±165mV output-voltage toler-ance. This permits a 50% reductionin the number of ceramic capacitorsrequired from twenty-four to twelve.Loop compensation is adjustedaccordingly.

Figure 4 shows an application cir-cuit using the LT1577, a dualregulator. All functions for each regu-lator are identical to those of theLT1575. One section is configured fora 3.3V output and the other section isconfigured for a 2.8V output. Thiscircuit provides all the power require-ments for a split-plane system: 3.3Vfor the logic supply and 2.8V for the

processor-core supply. Both regula-tor sections use the resistorlesscurrent limit technique. This tech-nique is discussed in detail below.Note that both SHDN pins are tied toa common time-out capacitor. If eitheror both regulators encounter a faultcondition, both regulator sections arelatched off after the time-out period isexceeded.

Block DiagramFunctional DescriptionFigure 5 is a block diagram of thefixed voltage LT1575. The primaryblock diagram elements comprise asimple feedback control loop and thesecondary block diagram elementscomprise multiple protection func-tions. A start-up circuit providescontrolled start-up for the IC, includ-ing the precision-trimmed bandgapreference, and establishes all inter-nal current and voltage biasing.

Precision Reference/OutputVoltage PerformanceReference voltage accuracy for theadjustable version and output volt-age accuracy for the fixed-voltageversions are specified as ±0.6% atroom temperature and as ±1% overthe full operating temperature range.This places the LT1575/LT1577 fam-ily among a select group of regulatorswith very tightly specified output volt-age tolerances. The accurate 1.21Vreference is tied to the noninvertinginput of the main error amplifier inthe feedback control loop.

Figure 3. 5V to 3.3V regulator


DESIGN FEATURES

Wide-Bandwidth Error AmplifierPermits Fast Loop ResponseThe error amplifier consists of a singlehigh gain gm stage with a transcon-ductance equal to 15 millimhos. Theinverting terminal is brought out asthe FB pin in the adjustable voltageversion and as the OUT pin in fixedvoltage versions. The gm stage pro-vides differential-to-single-endedconversion at the COMP pin. The out-put impedance of the gm stage isabout 1M; therefore, 84dB of typicalDC error amplifier open-loop gain(AVOL = gm RO) is realized along with atypical 75MHz uncompensated unity-gain crossover frequency. Externalaccess to the high impedance gainnode of the error amplifier permitstypical loop compensation to beaccomplished with an RC network toground.

A high speed, high current outputstage buffers the COMP node anddrives up to 5000pF of effectiveMOSFET gate capacitance with al-most no change in load transientperformance. The output stage deliv-ers up to 50mA peak when slewingthe MOSFET gate in response to loadcurrent transients. The outputimpedance of the GATE pin is typi-cally 2Ω. This pushes the pole causedby the error-amplifier output im-pedance and the MOSFET input

capacitance well beyond the loopcrossover frequency. If the capaci-tance of the MOSFET used is lessthan 1500pF, it may be necessary toadd a small value series gate resistorof 2Ω–10Ω. This gate resistor helpsdamp the LC resonance created bythe MOSFET gate’s lead inductanceand input capacitance. In addition,the high frequency pole formed by thegate resistor and the MOSFET inputcapacitance can be fine tuned.

Because the MOSFET pass tran-sistor is connected as a sourcefollower, the power path gain is muchmore predictable than that of designsemploying a discrete PNP transistoras the pass device. This is because ofthe significant production variationsencountered with PNP beta. MOSFETsare also very high speed devices, andhence are more suitable than PNPsfor creating a stable, wide-bandwidthcontrol loop. An additional advantageof the follower topology is inherentlygood line rejection. Input supply dis-turbances do not propagate throughto the output. The feedback loop for aregulator circuit is completed by pro-viding an error signal to the FB pin inthe adjustable voltage version and tothe OUT pin in the fixed voltage ver-sion. In both cases, a resistor dividernetwork senses the output voltageand sets the regulated DC bias point.

In general, the LT1575 regulator feed-back loop permits a loop crossoverfrequency on the order of 1MHz whilemaintaining good phase and gainmargins. This unity-gain frequencyis a factor of twenty to thirty times thebandwidth of currently implementedregulator solutions for microproces-sor power supplies. This significantperformance benefit is what permitsthe elimination of all bulk outputcapacitance.

High-Side SenseCurrent LimitingSeveral other unique features in-cluded in the design increase itsfunctionality and robustness. Thesefunctions comprise the remainder ofthe block diagram.

A high-side sense, current-limitamplifier provides active currentlimiting for the regulator. The cur-rent-limit amplifier uses an externallow value shunt resistor connected inseries with the external MOSFET’sdrain. This resistor can be a discreteshunt resistor or can be manufac-tured from a Kelvin-sensed section offree PC board trace. All load currentflows through the MOSFET drain,and thus, through the sense resistor.The advantage of using high-side cur-rent sensing in this topology is thatthe MOSFET’s gain and the main

+ C2 330µF 6.3V

C3 0.33µF

C6 1500pF

R2 3.9k

R3 1.21k

R4 1.21k

R7 2.1k

R1 3.9Ω

R5 3.9Ω

R8 1.6k

C9 TO C20* 1µF

Q1 IRFZ24

VI/O 3.3V

C5 10pF C8

1000pF

R6 7.5kC7

10pF

AN69 F06

FAULT RESET

C4 0.1µF

12V

C1 330µF

6.3V

INPUT 5V

+

Q2 IRFZ24

*X7R CERAMIC 0805 CASE

C21 TO C44* 1µF

VCORE 2.8V

1

2

3

4

16

15

14

13

IPOS

INEG

GATE

COMP

SHDN

VIN

GND

FB

1/2 LT15775

6

7

8

12

11

10

9

IPOS

INEG

GATE

COMP

SHDN

VIN

GND

FB

1/2 LT1577

Figure 4. LT1577 dual regulator for split-plane systems


DESIGN FEATURES

feedback loop’s gain remain unaf-fected. The sense resistor develops avoltage equal to IOUT × RSENSE. Thecurrent-limit amplifier’s 50mV thresh-old voltage is a good compromisebetween power dissipation in thesense resistor, dropout voltage andnoise immunity. Current limit is acti-vated when the sense resistor voltageequals the 50mV threshold.

Two events occur when currentlimit is activated: first, the current-limit amplifier drives Q2 in the blockdiagram and clamps the positive swingof the COMP node in the main erroramplifier to a voltage that provides anoutput load current of 50mV/RSENSE(this action continues as long as theoutput current overload persists);second, a timer circuit is activated atthe SHDN pin. This pin is normallyheld low by a 5µA active pull-downthat saturates to approximately100mV above ground. When current

limit is activated, the 5µA pull-downturns off and a 15µA pull-up currentsource turns on. Placing a capacitorin series with the SHDN pin to groundgenerates a programmable-time rampvoltage.

The SHDN pin is also the positiveinput of comparator COMP1. Thenegative input is tied to the internal1.21V reference. When the SHDN pinramps above VREF, the comparatordrives Q4 and Q5. This action pullsthe COMP and GATE pins low andlatches the external MOSFET driveoff. This condition reduces theMOSFET power dissipation to zero.The time period until the latched-offcondition occurs typically equals(CSHDN × 1.11V)/15µA. For example,a 1000pF capacitor on the SHDN pinyields a 74µs ramp time. In short,this unique circuit block performs acurrent-limit time-out function thatlatches off the regulator drive after a

predefined time period. The time-outperiod selected is a function of systemrequirements including start-up andsafe-operating area. The SHDN pin isinternally clamped to 1.85V (typical)by Q6 and R2. The comparator tied tothe SHDN pin has typical hysteresisof 100mV to provide noise immunity.The hysteresis is especially usefulwhen using the SHDN pin for thermalshutdown.

Normal operation can be restoredafter the load-current fault is clearedin one of two ways: recycle the nomi-nal 12V LT1575 supply voltage (aslong as an external bleed path for theshutdown pin capacitor is provided)or provide an active reset circuit thatpulls the SHDN pin below VREF. Pull-ing the SHDN pin below VREF turns offthe 15µA pull-up current source andreactivates the 5µA pull-down. If theSHDN pin is held below VREF during afault condition, the regulator contin-

SW2NORMALLY CLOSED

I2 5µA

–

+ERROR AMP

COMP

1575/77 BD2

–

+COMP1

Q6

SHDN

VIN

GND

OUT

R2 5k

SW1NORMALLY OPEN

100mV HYSTERESIS

I1 15µA

I3 100µA

–

+ILIM AMP VTH1

50mV

+–VTH3

500mV+–

VTH2 1V

+–

D1

IPOS

INEG

GATE

D2 –

+

–

+COMP2

COMP4

–

+

COMP3

OR2

START-UP VREF1.21V

R1 50k

OR1

Q4

Q3Q2Q1

Q5

R3*

*VOUT = (1 + R3/R4)VREF

R4*

Q7

Figure 5. LT1575 block diagram


DESIGN FEATURES

ues to operate in current limit into ashort. This action requires the abilityto sink 15µA from the SHDN pin atless than 1V. The 5µA pull-down cur-rent source and the 15µA pull-upcurrent source are designed to be lowenough in value that an externalresistor divider network can drive theSHDN pin to provide overvoltageprotection or to provide thermal shut-down with the use of a thermistor inthe divider network. It is simple todiode-OR these functions together andobtain multiple functions from onepin.

Senseless Current LimitingIf the current-limit amplifier is notused, two choices are available. Thesimplest choice is to tie the INEG pindirectly to the IPOS pin. This actiondefeats current limit and provides themost basic circuit. An example of anapplication in which the current limitamplifier is not used is one in whichan extremely low dropout voltage mustbe achieved and the 50mV thresholdvoltage cannot be tolerated.

A second available choice permitsa user to provide short circuit protec-tion with no external sensing. Thistechnique is activated by groundingthe INEG pin. This action disables thecurrent limit amplifier becauseSchottky diode D1 clamps theamplifier’s output and prevents Q2from pulling down the COMP node. Inaddition, Schottky diode D2 turns offpull-down transistor Q1. Q1 isnormally on and holds internal com-parator COMP3’s output low. Thiscomparator circuit, which is nowenabled, monitors the GATE pin anddetects saturation at the positive rail.When it detects a saturated condi-tion, COMP3 activates the shutdowntimer. Once the time-out periodoccurs, the output is shut down andlatched off. The operation of resettingthe latch remains as described above.

Note that this technique does notlimit the FET current during the time-out period. The output current is onlylimited by the input power supplyand the input/output impedance.Output currents in the range of 50A–

100A are possible. Setting the timerto a short period in this mode ofoperation keeps the external MOSFETwithin its SOA (safe operating area)boundary and keeps the MOSFET’stemperature rise under control.

No Power SupplySequencing ProblemsThe issue of power supply sequencingis important because the typicalLT1575 application has inputs fromtwo separate power supply voltages.A unique circuit design incorporatedinto the LT1575 alleviates all con-cerns about power supply sequencing.If the typical 12V VIN supply voltage isslow in ramping up, insufficient MOS-FET gate drive is present, andtherefore the output voltage does notcome up. If the VIN supply voltage ispresent but the typical 5V supplyvoltage tied to the IPOS pin has not yetstarted, the feedback loop wants todrive the GATE pin to the positive VINrail. This will result in a very largeMOSFET current as soon as the 5Vsupply starts to ramp up. However,undervoltage lockout circuit COMP2,which monitors the IPOS supply volt-age, holds Q3 on and pulls the COMPpin low until the IPOS voltage risesabove the internal 1.21 reference volt-age. The undervoltage lockout circuitthen smoothly releases the COMP pinand allows the output voltage to comeup in dropout from the input supplyvoltage. An additional benefit derivedfrom the speed of the LT1575 feed-back loop is that turn-on overshoot isvirtually nonexistent in a properlycompensated system.

An additional circuit feature is builtin to the LT1575 fixed-voltage ver-sions. When the regulator circuitstarts, it must charge the outputcapacitors. The output voltage typi-cally tracks the input voltage supplyas it ramps up with the input/outputvoltage difference defined by the drop-out voltage. Until the feedback loopcomes into regulation, the circuitoperation results in the GATE pinbeing at the positive VIN rail; thisstarts the timer if the current limit

amplifier is not used. However, inter-nal comparator COMP4 monitors theinput-output voltage differential. Thiscomparator does not permit the shut-down timer to start until thedifferential voltage is greater than500mV. This permits normal start-up.

Fixed Voltage VersionsEliminate Precision DiscreteResistor Divider NetworksOne final benefit results from using afixed voltage version of the LT1575.Today’s highest performance micro-processors dictate that precisionresistors must be used with currentlyavailable adjustable-voltage regula-tors to meet the initial set pointtolerance. The LT1575 fixed voltageversions incorporate the precisionresistor divider network into the ICand still maintain a 1% output volt-age tolerance over temperature. TheLT1575 offers fixed voltage options of1.5V (GTL+ termination), 2.8V(Pentium P55C), 3.3V, 3.5V (PentiumP54 VRE) and 5.0V.

ConclusionThe unique design of the new LT1575/LT1577 family combines the benefitsof low dropout voltage, high functionalintegration, precision performanceand ultrafast transient response, aswell as providing significant cost sav-ings on the output capacitance neededin fast load-transient applications.As lower input/output differentialvoltage applications become increas-ingly prevalent, an LT1575-basedsolution achieves efficiency perfor-mance comparable to that of aswitching regulator at appreciable costsavings.

The new LT1575/LT1577 family oflow dropout regulator controller ICssteps to the next level of performancerequired by system designers for thelatest generation of motherboards andmicroprocessors. The simple versa-tility and benefits derived from thesecircuits meets the power supply needsof today’s high performance micro-processors with ease.


DESIGN FEATURES

LTC1479 PowerPath ControllerSimplifies Portable PowerManagement Design by Tim Skovmand

As the computing power of por-table equipment rises, increasingdemands are placed on the portablepower management system. Theenergy stored in the Li-Ion or NiMHbattery packs must be transferred assmoothly and efficiently as possibleto the input of the DC/DC switchingregulator. These demands, coupledwith the increased need for higheroperating and charging currents, con-spire to complicate the front end ofthe power management system—theso-called “power path.”

This is where the battery packs,the AC wall adapter, the batterycharger and the standby system con-verge to create a power-managementnightmare. The real world problemsassociated with the switching of poweramong these sources are often quitesubtle and daunting, which may ex-plain why the prevailing solutions arealmost as varied as the portable equip-ment in which they reside. Manysolutions require a large amount of

printed circuit board space and aconsiderable number of discrete com-ponents to implement—it is notuncommon to find an eclectic mix-ture of regulators, comparators,references, glue logic, MOSFETswitches and drivers in the powerpath area of the circuit board.

Fortunately, some commonalityhas emerged among power-pathswitching schemes and the solutionsto these real world problems havebeen integrated into a new family ofpower-management controllers thatsimplify the monitoring and switch-ing of the batteries, the AC adapter,the battery charger and the standbysystem.

The LTC1479 PowerPath™ control-ler drives low loss N-channel MOSFETswitches to direct power in the mainpower path of a dual rechargeablebattery system, the type found inmost notebook computers and otherportable equipment.

Figure 1 is a conceptual block dia-gram that illustrates the main featuresof an LTC1479 dual-battery powermanagement system, starting withthe three main power sources andending at the input of the DC/DCswitching regulator.

Switches SWA/B, SWC/D andSWE/F direct power from either theAC adapter (DCIN) or one of the twobattery packs (BAT1 and BAT2) to theinput of the DC/DC switching regula-tor. Switches SWG and SWH connectthe desired battery pack to the batterycharger. These five switches are intel-ligently controlled by the LTC1479,which interfaces directly with thepower management microprocessor.

Back-to-Back SwitchesEach of the simple SPST switchesshown in Figure 1 actually consists oftwo back-to-back N-channel MOSFETswitches. Figure 2 is a simplified sche-matic diagram, which shows only thethree main power-path switches for

BAT1

BAT2

BATTERY CHARGER (LT1510)

SWA/B

SWC/D

SWE/F

SWG SWH

+

HIGH EFFICIENCY DC/DC SWITCHING

REGULATOR (LTC1435, ETC.)CIN

DCIN

LTC1479 PowerPath™ CONTROLLER POWER

MANAGEMENT µP

LTC1479 - BLK1

BACK-UP REGULATOR

AC ADAPTER

5VINRUSH CURRENT LIMITING

Figure 1. Dual PowerPath controller conceptual block diagram


DESIGN FEATURES

descriptive purposes. The low loss,N-channel switch pairs are housed in8-pin SO packaging and are readilyavailable from a number of MOSFETmanufacturers.

The back-to-back topology elimi-nates the problems associated withthe inherent body diodes in powerMOSFET switches and allows eachswitch pair to block current flow inboth directions when the two halvesare turned off.

Inrush Current LimitingThe back-to-back topology also allowsindependent control of each half ofthe switch pair, and thus, the use ofbidirectional inrush current limiting.

The voltage across a single lowvalue resistor, RSENSE, is measured todetermine the instantaneous currentflowing through the three main switchpairs, SWA/B, SWC/D, and SWE/F.The inrush current is then controlledby the gate drivers until the transi-tion from one power source to anotherhas been completed. The current flow-ing in and out of the three main power

sources and the DC/DC converterinput capacitor is dramaticallyreduced.

Tantalum CapacitorsIn many applications, inrush currentlimiting makes it feasible to use lowprofile tantalum surface mountcapacitors in place of bulkier electro-lytic capacitors at the input of theDC/DC converter.

Built-In Step-Up RegulatorThe gate drive for all five low lossN-channel switches is supplied by amicropower step-up regulator, whichcontinuously generates 37V. The VGGsupply provides sufficient headroomabove the maximum 30V operatingvoltage of the three main powersources to ensure that the logic-levelMOSFET switches are fully enhancedby the gate drivers, which supply aregulated 5.7V gate-to-source volt-age, VGS, when turned on.

The power for the micropower boostregulator is taken from three internal

diodes connected to each of the threemain power sources—DCIN, BAT1 andBAT2. The highest voltage potential isdirected to the top of an inexpensive1mH surface mount inductor, L1.

A fourth internal diode directs thecurrent from L1 to the VGG outputcapacitor, C2, further reducing theexternal parts count. In fact, onlythree external components arerequired by the VGG regulator: L1, C1and C2.

Typical Application CircuitA typical dual Li-Ion battery powermanagement system is illustrated inFigure 3. If “good” power is availableat the DCIN input (from the ACadapter), both MOSFETs in switchpair SWA/B are on—providing a lowloss path for current flow to the inputof the LTC1538-AUX DC/DC con-verter. Switch pairs SWC/D andSWE/F are turned off to block cur-rent from flowing back into the twobattery packs from the DC input.

BAT1

BAT2

SWA

SWC

+HIGH

EFFICIENCY DC/DC

SWITCHING REGULATOR

(LTC1435, ETC)

5V

3.3V

12V

CIN

DCIN

LTC1479 PowerPath™ CONTROLLER

POWER MANAGEMENT

µP

SWB

SWD RSENSE

V+

SW

VGG

L1 1mH

C1 1µF 50V

C2 1µF 50V

+

STEP-UP SWITCHING REGULATOR

+

SWE SWF

GATE DRIVER

GATE DRIVER

GATE DRIVER

INRUSH CURRENT SENSING

AND LIMITING

Figure 2. Dual-battery PowerPath™ controller: VGG regulator, inrush limiting and switch gate drivers


DESIGN FEATURES

Battery ChargingThe LTC1479 works equally well withboth Li-Ion and NiMH batteries andchargers. In this application, anLT1510 constant-voltage, constant-current (CC/CV) battery chargercircuit is used to alternately chargetwo Li-Ion battery packs.

The power management micropro-cessor decides which battery is inneed of recharging by either queryinga smart battery pack directly or bymore indirect means. After the deter-mination is made, switch pair SWG orSWH is turned on by the LTC1479 topass charger current to one of thebatteries. Simultaneously, the se-lected battery voltage is returned tothe voltage feedback input of theLT1510 CV/CC battery charger via abuilt-in switch in the LTC1479.

After the first battery is charged, itis disconnected from the charger cir-cuit. The second battery is thenconnected through the other switch

pair and the second battery is charged.(The LTC1479 works equally well withthe LT1511 3A CC/CV BatteryCharger and LTC1435/LT1620 4ACC/CV Battery Charger.)

Running on BatteriesWhen the AC adapter is removed, theLTC1479 instantly informs the powermanagement microprocessor that theDC input is no longer “good” and thedesired battery pack is connected tothe input of the LTC1538-AUX highefficiency switching regulator througheither switch pair SWC/D or SWE/F.

Back-Up Powerand System RecoveryBackup power is provided by astandby switching regulator, which istypically powered from a smallrechargeable battery and ensures thatthe DC/DC input voltage does notdrop below a predetermined level (forexample, 6V).

The “Three Diode Mode”When the system is powered by thebackup regulator, the LTC1479 en-ters a unique operating state calledthe “three diode mode,” as illustratedin Figure 4. Under normal operatingconditions, both halves of each switchpair are turned on and off simulta-neously. For example, when the inputpower source is switched from a goodDC input (AC adapter) to a good bat-tery pack, BAT1, both gates of switchpair SWA/B are turned off and bothgates of switch pair SWC/D are turnedon. The back-to-back body diodes inswitch pair SWA/B block current flowin or out of the DC input connector.

In the three diode mode, only thefirst half of each power path switchpair, that is, SWA, SWC and SWE, isturned on; and the second half , thatis, SWB, SWD and SWF, is turned off.These three switch pairs now act asthree diodes connected to the threemain input power sources. The power

DCIN

DCIN

BAT1

BAT2

+ +

V+ SW VGG

1µF 50V

1mH * 1µF 50V

VCC

+

RSENSE0.033Ω

SWA SWB

SWC SWD

SWE SWF

GA GB GC GD GE GFSAB SCD SEF SENSE+ SENSE-

+

VCCP

2.2µF 16V 0.1µF

VBATPOWER

MANAGEMENT µP

BDIV

VBKUP BACK-UP REGULATOR

LTC1538-AUX TRIPLE, HIGH EFFICIENCY, SWITCHING REGULATOR

DCDIV

LTC1479 PowerPath™ CONTROLLER

LT1510 Li-Ion BATTERY

CHARGER

GG SG GH SH

CHGMON

3.3V

5.0V

SWG SWH

DCIN

Li-Ion BATTERY PACK #1

Li-Ion BATTERY PACK #2

LTC1479 - FIG03

BACKUP BATTERY

* 1812LS-105 XKBC, COILCRAFT (708) 639-1469

12V AUX

RB2

RB1

RDC2

RDC1

MBRS140T3

Figure 3. Dual Li-Ion battery power-management system (simplified schematic)


DESIGN FEATURES

path diode with the highest inputvoltage passes current through to theinput of the DC/DC converter to en-sure that the system cannot lock upregardless of how power is initiallyapplied.

After “good” power is reconnectedto one of the three main inputs, theLTC1479 drives the appropriateswitch pair on fully as the other twoare turned off, restoring normaloperation.

Interfacing to the PowerManagement MicroprocessorThe LTC1479 takes logic levelcommands directly from the micro-processor and makes changes at highcurrent and high voltage levels in thepower path. Further, it providesinformation directly to the micropro-cessor on the status of the AC adapter,the batteries and the charging system.

BAT1

BAT2

SWA

SWC

SWE+

HIGH EFFICIENCY

DC/DC SWITCHING REGULATOR

5V

3.3V

12V

CIN

DCIN

LTC1479POWER

MANAGEMENT µP

LTC1479 - FIG04

SWB

SWF

SWDON OFF

ON OFF

ON OFF

RSENSE

The LTC1479 logic inputs and out-puts are TTL level compatible andtherefore interface directly withstandard power management micro-processor. Because of the directinterface via five logic inputs and twologic outputs, there is virtually nolatency (time delay) between themicroprocessor and the LTC1479. Inthis way, time-critical decisions canbe made by the microprocessor with-out the inherent delays associatedwith bus protocols and the like. Thesedelays are acceptable in certain por-tions of the power managementsystem, but it is vital that the powerpath switching control be madethrough a direct connection to thepower management microprocessor.The remainder of the power manage-ment system can be easily interfacedto the microprocessor through eitherparallel or serial interfaces.

The Power ManagementMicroprocessorThe power management microproces-sor provides intelligence for the overallpower system, and is easily pro-grammed to accommodate the customrequirements of each system and toallow performance updates withoutresorting to costly hardware changes.Many inexpensive microprocessorsare available that can easily fulfillthese requirements.

ConclusionThe LTC1479 is the “heart” of a totalpower management solution forsingle- and dual-battery notebookcomputers and other portable equip-ment. It works in concert with otherLinear Technology power manage-ment products, such as the LTC1435family of high efficiency DC/DC con-verters and the LT1510 family ofbattery chargers to end your power-management nightmares. TheLTC1479 is available in 36-lead SSOPpackaging.

Figure 4. LTC1479 PowerPath™ controller in “three diode mode”


DESIGN FEATURES

New Rail-to-Rail Amplifiers:Precision Performance fromMicropower to High SpeedIntroductionLinear Technology’s latest offeringsexpand the range of rail-to-rail ampli-fiers with precision specifications.Rail-to-rail amplifiers present anattractive solution for signal condi-tioning in many applications. Forbattery-powered or other low voltagecircuitry, the entire supply voltagecan be used by both input and outputsignals, maximizing the system’sdynamic range. Circuits that requiresignal sensing near the positive sup-ply are straightforward using arai l-to-rai l amplifier. LinearTechnology’s family of rail-to-railamplifiers satisfies the need for rail-to-rail outputs and provides precisioninput-offset specifications. Because,for rail-to-rail applications, input off-set is important across the entirecommon mode range, LTC’s family ofamplifiers uses a proprietary trimscheme that minimizes the input off-set at two common mode voltages,equal to the positive and negativesupplies. To make design using thedevices straightforward, the offsetvoltage under these two conditions isclearly defined on the data sheet.

Rail-to-Rail Op Amp FamilyThe latest additions to LTC’s family ofprecision rail-to-rail op amps spanthe range of applications frommicropower to high speed. All mem-bers of the family have both rail-to-railinput and output capability. The fast-est members of the family, theLT1498/LT1499 C-Load™ op ampsfeature a 10MHz gain-bandwidthproduct, a slew rate of 4V/µs and theability to drive 10,000pF. For lowcurrent applications, the LT1466–69series supplies precision performancewith a quiescent current of only 50µAper amplifier. Finally, the most accu-rate members of the family, theLT1218/LT1219, feature VOS trimmedto less than 100µV, the tightest VOSspec among LTC’s rail-to-rail amps.

These devices combine precisionwith low voltage operation, operatingwith supplies as low as 2.2V, and arefully specified for 3V operation. Theinput offset voltage is less than 475µVfor the LT1466 and LT1498 and lessthan 100µV for LT1218. In keepingwith the precision nature of the parts,the open loop gain AVOL is one millionor greater driving a 10k load. The rail-

to-rail operation is fully specified andtested over the entire supply range.The input offset voltage and inputbias currents are specified at com-mon mode voltages equal to both VCCand VEE.

The LT1466–69 and the LT1218/LT1219 are offered in two versions,which differ in their frequency com-pensation. The LT1466 dual andLT1467 quad and the LT1218 singlehave conventional compensation. Foruse in low frequency or DC applica-tions, the LT1468 dual, LT1469 quad,and LT1219 single are C-Load ampli-fiers, compensated for use with a0.1µF capacitor at the output. In anoisy environment, the large outputcapacitor clean ups the output signalby improving the supply rejection andproviding a low output impedance athigh frequencies.

by William Jettand Danh Tran

VBIAS

R2R_01.eps

Q12

Q13

Q9

Q11

I1

Q10

Q7Q2Q1

D1

Q3 Q4

Q6

V–

IN–

IN+

V+

Q8

Q5

OUT

V–

C1

C2

CC

BUFFER AND OUTPUT BIAS

Figure 1. Rail-to-rail amplifier simplified schematic

Figure 2. LT1498 small signal response

Figure 3. LT1498 large signal response

CL = 10nF

CL = 500pF

CL = 0pF

CL = 0pF

CL = 500pF

CL = 10nF


DESIGN FEATURES

The devices are available in a vari-ety of package options. The LT1466,LT1468 and LT1498 are dual ampli-fiers, available in either 8-pin SO or8-pin miniDIP packages. The LT1467,LT1469 and the LT1499 are quadamplifiers available in the 14-pin SOonly. The LT1218 and LT1219 aresingle amplifiers with a shutdownfunction, available in 8-pin SO and 8-pin miniDIP packages.

The Rail-to-Rail ArchitectureThough the new rail-to-rail amplifiersdescribed differ in detail, they share acommon approach to the input andoutput stages. Figure 1 shows a sim-plified schematic. The input stageconsists of two differential amplifiers,a PNP stage Q1–Q2 and an NPN stageQ3–Q4, which are active over differ-ent portions of the input commonmode range. Each input stage istrimmed for offset voltage. A comp-lementary output configuration(Q12–Q13) is employed to create anoutput stage with rail-to-rail swing.The devices are fabricated on LinearTechnology’s proprietary complemen-tary bipolar process, which ensuresvery similar DC and AC characteris-tics for the output devices Q12 andQ13.

First, looking at the input stage,Q5 switches the current from currentsource I1 between the two inputstages. When the input common modevoltage VCM is near the negative sup-ply, Q5 is reverse biased, so thecurrent from I1 becomes the tail cur-rent for the PNP differential pairQ1–Q2. At the other extreme, whenVCM is near the positive supply, PNPsQ1–Q2 are biased off. The currentfrom I1 then flows through Q5 to thecurrent mirror D1–Q6, furnishing thetail current for the NPN differentialpair Q3–Q4. The switchover pointbetween stages occurs when VCM isequal to the base voltage of Q5, whichis biased approximately 1.3V belowthe positive supply.

The collector currents of the twoinput pairs are combined in the sec-ond stage, consisting of Q7–Q11. Mostof the voltage gain in the amplifier is

contained in this stage. The output ofthe second stage is then buffered andapplied to the output devices Q12and Q13. Capacitors C1 and C2 formlocal feedback loops around the out-put devices, lowering the outputimpedance at high frequencies.Capacitor CC sets the amplifierbandwidth.

PerformanceTable 1 summarizes the performanceof the newest rail-to-rail amplifiers.As mentioned earlier, input offset volt-age and bias currents are tested withthe input common mode voltages atboth VEE and VCC.

The LT1498/LT1499—10MHz Bandwidth andC-Load PerformanceThe LT1498/LT1499 are the highestfrequency members of the LTC’s pre-cision rail-to-rail amplifier family,featuring a 10MHz gain-bandwidthand a 4V/µs slew rate. Designed forease of use, the LT1498/LT1499 areC-Load amplifiers, stable at unity gainwhen loaded with up to 10nF. Boththe small signal and large signal tran-sient response with a capacitive loadare well behaved. Figures 2 and 3illustrate the stability of the devicesfor small signal and large signalconditions.

ApplicationsThe ability to accommodate any inputor output signal that falls within theamplifier supply range makes theseamplifiers very easy to use. The fol-lowing applications demonstrate theversatility of the family of amplifiers.continued on page 37

–

+–

+

R2R_04.eps

100pF

1/2 LT1498

6.81k

330pF

VS — 2

VIN11.3k6.81k 47pF

1/2 LT1498 VOUT

5.23k

1000pF

10.2k5.23k

Figure 4. 100kHz 4th order Butterworth filter

retemaraP lauD6641TL elgniS8121TL lauD8941TL

reifilpmAreptnerruCylppuS 05 µA 004 µA01 µ nwodtuhsA Am7.1

tcudorPhtdiwdnaBniaG zHk001 zHk004 zHM01etaRwelS /V30.0 µs /V11.0 µs /V4 µs

V,egatloVtesffO MC V= EE V, CC 574< µV 001< µV 574< µVR(niaGpooLnepO L )k01= /V1 µV /V1 µV /V2 µV

V,tnerruCsaiBtupnI MC V= EE V, CC An5 An52 An052,tnerruCtesffOtupnI

V MC V= EE V, CCAn1 An6 An02

noitarutaStuptuOegatloV

daoLoNIO Am5.2=

Vm03Vm572

Vm6Vm002

Vm51Vm002

tnerruCtiucriCtrohS Am02 Am51 Am02egnaRegatloVylppuSgnitarepO V21otV2.2 V21otV2.2 otV0.2 ± V51

segatloVylppuSdeificepS ,V5,V3 ± V5 ,V5,V3 ± V5 ,V5,V3 ± V51

Table 1. Amplifier performance: VS = 5V, 25˚C


DESIGN FEATURES

FREQUENCY (Hz)

0

12

11

10

9

8

74

68

62

56

50

7

6

5

4

3

2

1

ENO

Bs

SIN

AD

10M0 10k 100k 1M

Figure 2. This curve shows that the dynamicperformance of the LTC1409 remains robustout to an input frequency of 1MHz, where itmaintains 11-bit performance.

FREQUENCY (Hz)

–120

–60

–80

–100

0

–20

–40

AMPL

ITU

DE

(dB)

200k 250k 300k 350k 400k0 50k 100k 150k

Figure 3. FFT of the LTC1415’s conversion of a full-scale 100kHz sine wave shows excellentresponse with a low noise floor while sampling at 800ksps.

SAMPLE/ HOLD

CIRCUIT

PRECISION 12-BIT DAC

LTC1409 AND

LTC1415

SAR

2kΩ

+AIN

–AIN

VREF (2.5V)

REFCOMP (4.1V)

OUTPUT BUFFER

12 12

LOW DRIFT VOLTAGE

REFERENCE

CLOCK

NAP/SLP SHDN RD CONVST CS

CONTROL LOGIC BUSY

COMPARATOR

Figure 1. LTC1409 and LTC1415 block diagram

fast successive-approximation ADC.Supporting these high speed circuitsis an onboard voltage reference,power-saving shutdown circuitry andan easy to use parallel digital inter-face. This interface easily connects toFIFOs, microprocessors and DSPs.

To increase interface flexibility,both devices have a digital VCC thatpowers only the output-logic circuitry,easing connection to a 3V host pro-cessor. This allows the analogconversion to operate on 5V, maxi-mizing the dynamic range and SINAD,while the conversion data’s logic lev-els are compatible with the hostprocessor’s 3V logic.

DC and AC PerformanceThe DC specifications include a±0.8LSB maximum differential lin-earity error and ±0.5LSB maximumintegral linearity error guaranteedover temperature. The ADCs’ gain isheld constant over temperature withan on-chip 10ppm/°C curvature-cor-rected bandgap reference.

The sample-and-hold used in theLTC1409 and LTC1415 determinestheir dynamic performance. TheseADCs have a wide bandwidth andvery low distortion differential sampleand hold. Dynamic performancespecifications include THD of –84dBfor a 625kHz input and a sample-and-hold input bandwidth of 30MHz.

Figure 2 shows the widebandintegrity of the LTC1409’s conver-sion. This curve shows that theeffective number bits (ENOB) is 11.8and remains flat out to an input fre-quency of 200kHz. Even at 1MHz,the LTC1409 maintains 11-bitperformance.

An FFT of the performance of theLTC1415 operating at full conversionrate of 800ksps is shown in Figure 3.The input signal is a full-scale 100kHzsine wave. The curve shows excellentresponse with a low noise floor whilesampling at 800ksps.

The LTC1409 and the LTC1415have vanishingly low bit-error rates.The error rates are so low that mea-suring them is very difficult. To beginto uncover the error’s characteristics,

measurements were made at anelevated temperature of 150°C (bit-error rates increase with increasingtemperature). Even at this high tem-perature, the bit-error rate was foundto be less than one in 100 billion(10–11). At room temperature, the bit-error rate is projected to be less thanone in 2,000,000 billion (2 × 10–15). Tounderstand the magnitude of thiserror, consider that the LTC1409 orthe LTC1415, operating at full con-version rate, would be free of bit errorsfor at least 50 or 78 years, respectively.

Flexible ReferenceFigure 4 shows a simplified circuitdiagram of the reference circuitryfound in the LTC1409 andLTC1415.The reference’s temperaturecoefficient is 10ppm/°C, making itsuitable as a system reference. Thisallows other circuits using this refer-ence to track each other over

LTC1409/LTC1415, continued from page 1


DESIGN FEATURES

–

+

INTERNAL CAPACITOR

DAC

LTC1409/LTC1415

0.625R10µF

VREF (2.5V) 3

REFCOMP 4

AGND 5R

BANDGAP REFERENCE

REF AMP

VCC

2kΩ

Figure 4. Flexible reference allows use of internal or external reference source

temperature. When this internal ref-erence is used to set the full-scalerange (±2.5V for the LTC1409 and 0Vto 5V for the LTC1415), pin 3 is leftunconnected and a 10µF tantalumelectrolytic is connected betweenREFCOMP and AGND. The circuit inFigure 5 can be used to trim the full-scale (gain) error.

When an external reference is used,it can be connected in one of twoways, depending on its magnitude. Ifthe external reference voltage isbetween 2.5V and 3.0V, the VREF pincan be used. The REFCOMP pin canaccept reference voltages in the rangeof 2.5V to 5.0V. The VREF pin is appro-priate for applications that have afixed reference or, if the referencevoltage is changed, can tolerate slowsettling times. For applications suchas scanners, which require fastresponse to changing reference volt-ages, the REFCOMP pin should beused as the reference input. Becausethe internal reference amplifier is by-passed, the settling time for changingreference voltages is limited by thesize of the external bypass capacitor.When the REFCOMP pin is driven byan op amp, the filter capacitor can beeliminated. In this case, the settlingtime is a function of the op amp.

Differential InputsCancel WidebandCommon Mode NoiseUnwanted noise is a problem for mostcircuits. The problem of noise becomesincreasingly acute when dealing withhigh speed, high resolution ADCs. AnLSB’s magnitude decreases with in-creasing resolution. At a resolution of12 bits, the LSB is just 1.22mV for a5V full-scale input range. Noise fromfluorescent lights, electrical motorsor digital circuits can quickly com-bine to create unwanted signals whosemagnitude approaches a level thatcreates conversion errors. Althoughfiltering and shielding signal lineswill reduce the effects of noise, thesetechniques are sometimes inadequateor may limit bandwidth. The LTC1409and the LTC1415 offer anotherweapon in the fight against the debili-

tating effects caused by noise: differ-ential inputs.

Figure 6 shows a typical single-ended sampling system whose inputsignal is contaminated by groundnoise. This noise may be a combina-tion of 60Hz, digital clock noise and/orEMI. This ground noise adds directlyto the input signal. When the single-ended system samples this signal,the final conversion result is a combi-nation of the actual desired signaland the unwanted noise.

The single-ended system cannotdifferentiate between the desired sig-nal and the noise. The resultingconversion result is not correct.

Figure 7 shows the differentialinputs of the LTC1409 and theLTC1415. These differential inputsovercome this problem of groundnoise. Since the differential inputsare connected across the signalsource, ground noise is common

mode. The ADC’s excellent commonmode rejection ratio (CMRR) rejectsthe common mode ground noise andthe perturbations it creates in theconversion result. The CMRR isconstant over the entire Nyquist band-width (fS/2) and drops 3dB at 5MHz.This ability to reject high frequencycommon mode signals is very helpfulin sampling systems where noise of-ten has high frequency componentscaused by switching transients. Ad-ditionally, the differential inputs rejectin-band common mode noise, some-thing that is much more difficult toachieve with filters, and preserve theLTC1409’s and LTC1415’s wide band-width capabilities.

The LTC1409 and LTC1415 areespecially appropriate for applicationsthat require low power and high con-version rates. To conserve additionalpower, the typical power dissipationof 80mW (LTC1409) or 60mW

–

+

1409_3.eps

INTERNAL CAPACITOR

DAC

LTC1409/LTC1415

0.625R10µF

VREF 3

24kΩ

50kΩ47kΩ

REF COMP 4

AGND 5R

BANDGAP REFERENCE

REF AMP

VCC

2kΩ

Figure 5. This simple circuitry facilitates full-scale trim and adds nothing to the signal path.


DESIGN FEATURES

MEASURED SIGNAL

GROUND NOISE

SINGLE INPUT ADC

AGND

AINMEASURED

SIGNAL

GROUND NOISE

LTC1409 OR

LTC1415

AGND–AIN

+AIN

Figure 6. A single-ended ADC is not able todifferentiate the desired signal from theerror-inducing ground noise, losing signalintegrity.

Figure 7. The LTC1409’s and LTC1415’sexcellent broadband CMRR cancels commonmode noise, preserving the input signal’sintegrity.

(LTC1415) can be significantlyreduced during periods between con-versions using the two shutdownmodes. The NAP/SLP pin is used toselect one of the two power reductionshutdown modes. The Nap mode saves95% of the typical power dissipation.In Nap mode, the LTC1409 and the

LTC1415 reduce the bias current ofall internal circuitry, except the refer-ence, to zero. This mode allows thepart to return from shutdown asquickly as possible by leaving thereference circuitry operational. TheSleep mode shuts down the bias cur-rent to all internal circuitry, including

the reference. Sleep mode saves thegreatest amount of power, butrequires more time to awaken thanthe Nap mode.

In the Nap mode, all data outputcontrols are functional: data from thelast conversion prior to starting Napmode is still available and can beread using RD and CS.

ConclusionThe newest members of LTC’s highspeed 12-bit family offer effectivesolutions to many dynamic samplingapplications. These include high-speed telephony, compressed videoand high frequency data acquisition.The LTC1409 and the LTC1415 offeran unbeatable combination of highspeed and low power dissipation.

Generally speaking, low ESR, highvalue output capacitors should bechosen to optimize the use of boardspace. However, if the ESR value istoo low for a given capacitor value,loop stability problems can occur.The feedback loop depends on thefrequency of the ESR “zero” being wellbelow the loop crossover frequency.(You remember poles and zeros fromyour bumpy rides on the S-Plane,don’t you?) There is 45˚ of positivephase shift at the frequency wherethe capacitive reactance equals theESR of the capacitor. Without thisphase shift, the loop would be impos-sible to stabilize. Low ESR, AVXTPS-series tantalum capacitors are avery good compromise between ESR,capacitance value and physical size.

Input capacitors are included tosuppress the input switching noiseand to keep the input 5V supply varia-tion to a minimum during the Q1 ON/OFF cycle. Excessive conducted emis-sions are usually traced back toinadequate input capacitance or poorlayout of the power-path traces. Thecrucial parameter for the input ca-pacitors is ripple current rating. Areasonable rule of thumb says that

the input capacitor ripple current isgoing to be approximately 50% of theload current. Therefore, in a typicalPentium Pro processor application,the input capacitors should be ratedfor close to 7ARMS. An excellent choicefor the input capacitors are SanyoOS-CONs or the equivalent. They haveextremely high ripple current ratingsfor their size and have demonstratedexcellent reliability in this type ofapplication. Low ESR aluminium elec-trolytic capacitors are a viable optionfrom both input and output. Althoughlower in cost than OS-CONs or tanta-lum capacitors, their long-termreliability is not as good. Using 105˚Ccapacitors and keeping operating tem-peratures low will help to obtainreasonable capacitor life.

The combination of the Dale NTHS-1206N02 thermistor and the 1.8kresistor are for overtemperature moni-toring. The OT flag trips if the ambienttemperature at Q1 reaches about90°C; at 100°C the G1 and G2 driversstop operating. If the system moni-tors the OT flag, there should beample time to take precautions, savingdata and system configuration infor-mation prior to an overtemperature

shutdown. Alternatively, CPU activ-ity could be reduced, lowering powersupply current and allowing the sup-ply to cool down.

The PWRGD pin gives the CPU rail-voltage OK indication. If, for anyreason, the output regulation fallsout of the ±5% limit (including anovertemperature shutdown), PWRGDwill provide a logic low signal to thesystem monitor.

ConclusionThe LTC1553 is designed to be usedin all N-channel, synchronous buckswitching regulators for Pentium Proand other high performance micro-processors. A high level of integrationholds external parts count to a mini-mum, while providing a flexiblesolution. High efficiency can beachieved, eliminating the need forheat sinks in applications with cur-rents as high as 14 amps steadystate. All required system monitoringfunctions are supplied on chip. Com-ponent count and cost are reduced byeliminating the need for low resis-tance, high power, current senseresistors.

LTC1553, continued from page 8


DESIGN FEATURES

High Efficiency, Low DropoutLithium-Ion Battery Charger ChargesUp to Five Cells at 4 Amps or More

by Fran HoffartIntroductionRechargeable lithium batteries fea-ture higher energy density per volume,higher energy density per weight andhigher voltage per cell than any of thecompeting battery chemistries. Forthese reasons, manufacturers of por-table equipment are adopting thelithium-ion rechargeable battery asthe battery of choice for high perfor-mance portable equipment. Lighterweight and increased operating timebetween charges are important fea-tures that customers want and needfrom portable products.

Laptop computers are one of manyareas where rechargeable lithiumbatteries are rapidly replacing otherbattery types. Today’s laptops featurefaster microprocessors, more memory,larger back-lit liquid crystal displays,larger hard disks and more built-infeatures than ever before. At the sametime, laptop manufacturers are striv-ing to reduce the size and weight of

their products. These advances placeincreased energy demands on thebattery.

These increased demands haveforced manufacturers to use multiplecells in a combination of series andparallel configurations. Parallelingcells increases the amount of currentthat can drawn from the battery and/or increases the operating timebetween charges, but it also increasesthe current requirements of thecharger.

Linear TechnologyBattery Charger ProductsLinear Technology offers several dedi-cated switch-mode battery chargerICs with charging capabilities up to3A. For step-down applications, theLT1510, operating at 200kHz, canprovide up to 1.5A of charging cur-rent in a constant-current and/orconstant-voltage mode. For currents

up to 3A, the LT1511 is available. Inaddition to a constant-current/con-stant-voltage charge, the LT1511 alsofeatures a programmable current limiton the input side of the charger. Thisallows the input power source (ACadapter) to power a load, such as alaptop computer, and charge a batterysimultaneously, without overloadingthe input power source.

Two 500kHz chargers designed forSEPIC (single-ended primary induc-tance converter) topology are alsoavailable. The SEPIC design allowsthe input voltage to be less than,equal to or greater than the batteryvoltage. The LT1512 (1.5A) andLT1513 (3A) can be used in a constant-current/constant-voltage mode.

Higher Charge CurrentsParalleling cells, regardless of cellchemistry, requires relatively highcharge currents to bring the battery

+

+

AVG8

R3 (SEE TEXT)

1620_01.eps

SGND

SENSE+

VOSENS

PGND

BG

BOOST

INT VCC

SW

TG

5 4

6

10

VCC

IN–LT1620CM58

6

4

IN+ 5

SENSE

PROG

RPROG = 21k FOR 4A

GND

IOUT1

7

IPROG

3

2 11

15

12D2 D1

C6, 0.33µF

C7, 4.7µF

C5, 0.1µF

Q2

R2 1M

0.1%

RSENSE 0.02Ω

IBATT

C3 22µF

C8 100pF

Q1

L1 27µH

+VIN

C1, C2 22µF ×2 35V TANT.

C4 0.1µF

C10 100pF

C12 0.1µF

C14 1000pF

R5 1k

C13 0.033

C18 0.1µF

C15 0.1µF

R6 4k

C16 0.33µF

R7 1.5M

SHUTDOWN INPUT (SD = 0V)

14

16

13VIN

LTC1435CG

SFB

SENSE–

ITH

COSC

RUN/SS

8

7

3C11 56pf

1

2

L1 =CTX27-4, COILTRONICS Q1, Q2 =Si4412DY, SILICONIX D1, D2 =CMDSH-3, CENTRAL

C1, C2 =22 µF,35V, AVX TPS SERIES C3 =22 µF, 25V, AVX TPS SERIES

C17, 0.01µF

C9, 100pF

Figure 1. Complete schematic of high efficiency, 4A constant-voltage/constant-currentcharger using all surface mount components, with a circuit board area of 1.5 in2


DESIGN FEATURES

up to full charge in a short period oftime. When charging needs exceedthe 3A maximum rating of the LT1511or LT1513, the circuit shown in Fig-ure 1 can provide much higher currentsolutions, and very high efficiency.This circuit uses the LTC1435 andLT1620 in a charger that delivers 4A ormore with exceptional efficiency andlow dropout voltage (Figures 2 and 3).

The LT1435 SwitchingRegulator ControllerThe LTC1435 is a step-down currentmode switching regulator controllerdesigned to drive two external N-chan-nel power MOSFETs. Operating frominput voltages between 3.5V and 36V,this device includes a programmableswitching frequency, synchronous

rectification, Burst Mode™ operationand a 99% maximum duty cycle forlow dropout voltage. Additional fea-tures include a 1% tolerance outputvoltage (adjustable between 1.2V and9V), programmable soft start, logic-controlled micropower shutdown anda secondary feedback control pin.Because external MOSFET switchesare used, the maximum output loadcurrent is determined by the currentcapabilities of the selected FETs.

The LTC1435as a Battery ChargerThe low dropout voltage, high currentcapability and high efficiency of theLTC1435 switching regulator wouldseem to make it an appropriate choicefor high current battery chargers, butit has several limitations. The abso-lute maximum output voltage of 10volts allows only two series-connectedlithium cells to be charged and theoutput current is not readilyprogrammable.

Introducing the LT1620The LT1620 is an IC designed to beused with a current mode PWM con-troller (such as the LTC1435 andsimilar products) to increase the out-put voltage range and optimize thecircuit for battery charging applica-

tions. Used together, these two prod-ucts overcome the voltage and currentprogramming limitations previouslymentioned, to produce a high current,high performance constant-voltage/constant-current battery charger forlithium-ion and other battery types.

How They Work TogetherTo understand how the two partswork together, a brief review of theLTC1435 operation is necessary. SeeFigure 4. During each cycle of opera-tion, the series MOSFET switch Q1 isturned on by the LTC1435 oscillator(Q2 is off). This causes a current tobegin ramping up in inductor L1.When the current in L1 reaches apeak level determined by the voltageat the ITH pin, Q1 is turned off and thesynchronous MOSFET Q2 is turnedon, causing the current in L1 to rampdown to the level at which it started.Thus, a sawtooth of inductor ripplecurrent is generated, with a peaklevel set by the voltage on the ITH pin.This inductor current is sensed via anexternal, low value sense resistor inseries with the inductor and is usedto drive the LTC1435 internal currentsense amplifier for the current modefeedback signal. This current senseamplifier has a maximum commonmode voltage limit of 10V, which limitsthe maximum output voltage to 10V.

Enter the LT1620. The LT1620 alsocontains a current sense amplifier,which has a common mode rangethat extends up to 28V. This amplifieris used to level shift the differentialsense voltage, which is riding on thebattery voltage, and reference it to the

CHARGE CURRENT (A)

80

90

85

95

100EF

FICI

ENCY

(%)

5

1620_02.eps

0 1 2 3 4

VBATT = 16.8V

VBATT = 12.8V

VIN = 24V

CHARGE CURRENT (A)

0

1.0

0.5

1.5

2.0

DROP

OUT

VOLT

AGE

(V)

5

1620_03.eps

0 1 2 3 4

V BATT =16.8V I PROG =200 µA CONSTANT CURRENT PROGRAMMED FOR 4A

++

1620_04.eps

INT VCC

SENSEITH

INVCC

VOSENS

BG

TG

LT1620PROG

IOUT

SENSE

AVG

Q2

IBATT

Q1

L1

R2

R3

R6

PROGRAM CONSTANT CURRENT

RPROG

RSENSE

+VIN

CSS

SHUTDOWN INPUTVCC

LTC1435

GND

C15

COSC

RUN/SS

PROGRAM CONSTANT VOLTAGE

Figure 2. Charger efficiency for 3- and 4-cellapplications

Figure 3. Charger dropout voltage vs chargecurrent

Figure 4. Simplified diagram of constant-voltage/constant-current charger


DESIGN FEATURES

internal 5V VCC voltage generated bythe LTC1435. This level-shifted sig-nal is used to drive the LTC1435current sense pins, thus providingcurrent mode feedback for the con-stant-voltage feedback loop. Thissignal is also used to control theconstant output current feedbackloop, as explained below.

Constant Charge CurrentThe LT1620 also provides a simplemethod of accurately programmingthe constant-current output. Sink-ing an adjustable current from thePROG pin to ground controls thecharge current from zero current tomaximum current. This program cur-rent can be derived from a variety ofsources, such as a single resistor toground or the output of a DAC.

The constant-current feedback loopoperates as follows. With a dischargedbattery connected to the charger, andassuming that the battery voltage isless than the float voltage programmedby R2 and R3, the error amplifier inthe LTC1435 begins pulling up on theITH pin. This increases the peak in-ductor current in an effort to force thebattery voltage to be equal to theprogrammed voltage. By limiting thevoltage on the ITH pin, the peak induc-tor current and the average outputcurrent can be controlled. The ITH pinhas an internal 2.4V clamp that setsthe peak inductor to its maximum

level. This 2.4V clamp provides somedegree of current regulation, but theaverage battery current will vary con-siderably as a result of dependenceon inductor ripple current andLTC1435 parameter variations. Byadding the LT1620 to the circuit, theconstant charging current controlperformance is considerably im-proved. As shown in Figure 5, thesignal from the current sense amplifierin the LT1620 is amplified by 10,averaged by CAVG (C15 in Figure 1)and compared to the voltage dropacross R6. This voltage is developedby a current, IPROG, flowing throughR6. When the voltage at the LT1620AVG pin approaches the voltage onthe PROG pin, IOUT begins to pull theITH pin of the LTC1435 down, limitingthe peak inductor current and com-pleting the constant-current feedbackloop.

Complete Charger CircuitThe circuit shown in Figure 1 cancharge up to five series-connectedlithium-ion cells at currents up to 4A.Using low RDS(ON) MOSFET switchesfor the switch and synchronous recti-fier results in efficiency exceeding95% and allows all surface mountcomponents to be used, resulting in adesign that occupies less than 1.5 in2

of board space. This circuit operatesat a switching frequency of 200kHzand is capable of up to 99% duty

cycle; it can operate over a very wideinput voltage range, from a minimuminput of only 600mV greater than thebattery charging voltage to a maxi-mum of 28V (limited by the MOSFETs).

Constant-voltage charging withbetter than 1.2% accuracy and con-stant-current charging with 7.5%accuracy provides almost ideall i th ium-ion bat tery charg ingconditions.

In battery charger designs, animportant issue is reverse batterydrain current caused by the chargerwhen the input power is removed orthe charger is shut down, or both. Ifthe battery will remain connected tothe charger for extended periods oftime, it is important to minimize thisreverse drain current to prevent dis-charging the battery. The charger canbe shut down by using the RUN/SSpin on the LTC1435. This stops thecharging current and results in areverse battery drain current in thetens of microamps.

The LTC1435 and LT1620 havebeen configured so that the batterycan remain connected to the chargerwhen the input power is removed, butbecause of the inherent body diode inthe Q1 MOSFET, current can flowfrom the battery, through the Q1 bodydiode, to the LTC1435’s VIN pin, keep-ing it powered up. In this situation,because the charger is effectively pow-ered by the battery, the reverse batterydrain can be several mA, which coulddischarge the battery over an extendedperiod. Figure 6 contains circuitrythat automatically shuts down theLTC1435 when the input power isremoved and puts it into a low quies-cent current condition. Because theLT1620 is powered from the LTC1435INT VCC pin, it is also turned off.

When input power is applied, thecharger can still be shut down withan external signal to the RUN/SS pin.Shutdown occurs by pulling this pinlow; releasing it allows the capacitorto charge up via the internal 3µAcurrent source, producing a soft start.

By substituting higher currentMOSFETs and changing some com-ponent values, much higher chargingcurrents can be obtained.

RPROG

PROG

800mV

2.5kX10

AVG PIN

CAVG

40mV OFFSETIOUT VCC

+5V

+5V

gm

SENSE

LT1620

IN–

IBATT

CURRENT SENSE AMPLIFIER

IN+

RSENSE

80mV CHARGE CURRENT

INDUCTOR CURRENT

EXTERNAL AVERAGING CAPACITOR, C15

X1

ITH INT VCC +SENSE

LTC1435

–SENSE

4.2V

R6

IPROG

≈800mV

1620_05.eps

Figure 5. Simplified diagram of constant-current control loop


DESIGN FEATURES

Selecting Battery VoltageProgramming ResistorsThe charging voltage of lithium-ioncells is either 4.1 or 4.2 volts per cell,depending on the battery chemistry.Contact the battery manufacturer forthe recommended charge voltage. Toprogram battery charging voltage (floatvoltage) use the following equation(for best accuracy and stability, use0.1% resistors).

VBATT = VREF

VREF = 1.19V; USE APPROXIMATELY 100k Ω FOR R3

R2 R3

1 + ( )

R2 = R3VBATT VREF )) – 1

Selecting RSENSERSENSE is an external, low value resis-tor that is placed in the inductorcurrent path to develop a signal rep-resentative of the inductor or chargecurrent (IBATT). This signal is used asfeedback to control the switchingregulator constant-voltage and con-stant-current loops. To minimizeoverall dropout voltage and powerdissipation in the sense resistor, asense voltage of 80mV was chosen torepresent maximum charging cur-rent. Use the following equation toselect current sense resistor RSENSE.The maximum battery charge current(MAX IBATT) must be known.

RSENSE = 0.08V MAX IBATT

Selecting IPROGIPROG is a current from the PROG pinto ground that is used to program themaximum charging current. IPROG canbe derived from a resistor to ground,from the output of a DAC or by othermethods. This program current isgenerated using resistors and the 5VVCC available from the LTC1435.

Refer to the simplified diagram ofthe constant-current control loopshown in Figure 5. The DC voltageacross CAVG is proportional to theaverage charge current. This voltage

drives one input of a transcon-ductance (gm) amplifier. A programvoltage (relative to the 5V VCC line)proportional to the desired, or pro-grammed charge current is applied tothe other input of the transcon-ductance amplifier. This voltageshould be selected to be ten times theaverage voltage dropped across RSENSEwhen the charger is in a constant-current mode.

If the voltage across CAVG increasesto a level equal to the voltage at thePROG pin, the transconductanceamplifier begins pulling down on theITH pin of the LTC1435, thereby limit-ing the peak inductor current, andthus the average charge current.

The program voltage needed on theprogram pin can easily be generatedby two resistors, as shown in Figure5. A current (IPROG) is generated bythese resistors and the 5V VCC volt-age. This IPROG develops a voltageacross R6, which is used to set themaximum constant charge currentlevel. The circuit is designed for anapproximate PROG voltage of 800mV(don’t exceed the maximum spec of1.25V), referenced to the LT1620 VCCpin. Because of the gain-of-10 ampli-fier, this corresponds to a typicalvoltage across RSENSE of 80mV (with amaximum of 125mV).

The recommended range of resis-tor values for R6 is approximately2kΩ to 10kΩ. With 0.8V across R6,this will result in program currents(IPROG) between 400µA and 80µA.

The LT1620 was designed to reducethe charging current to zero under allconditions when the IPROG is set tozero. To ensure that the charging

current will always go to zero, anoffset was designed into the trans-conductance amplifier. In theequations for R6 and RPROGRAM, thisoffset is represented by using 840mVrather than 800mV.

Example:

GIVEN: MAXIMUM IBATT = 4A I PROG = 200µA (FOR MAXIMUM IBATT)

RSENSE = 0.08V MAX IBATT

0.08V 4A

= 0.02Ω=

R6 = 0.84V IPROG

0.84V 200µA

= 4.2kΩ=

5V – 0.84V IPROG

5V – 0.84V 200µA

= 20.8kΩ=RPROG =

Once R1 and R6 are known, the fol-lowing equations can be used todetermine RPROG and IPROG for lowerIBATT currents:

RPROG =

IPROG =

R6 [5 – 10(IBATT)(R1)] 0.04 + 10(IBATT)(R1)

10(IBATT)(R1) + 0.04 R6

PC Board LayoutAs with any high frequency switchingregulator, layout is important. Switch-ing current paths and heat producingthermal paths should be identifiedand the printed circuit board designedusing good layout practices.

Even with efficiency numbers inthe mid 90s, under some chargingconditions power losses can be as

+

1620_06.eps

TG Q1L1

D3

1M

D4SHUTDOWN

47k CSS

+VIN

Q3 2N3906

VCC

LTC1435

RUN/SS

D3, D4 = 1N4148

Figure 6. Circuitry that shuts down the charger when input power is removed, minimizingreverse battery current drain



DESIGN FEATURES

LTC1069-X: a New Family of 8th OrderMonolithic Filters in SO-8 Packages

by Nello Sevastopoulos and Philip Karantzalis

The LTC1069 switched capacitor fil-ter technology offers economicalsolutions for a wide range of filterrequirements. The filter response, thesampling rate and the power con-sumption can be easily reprogrammedby means of a single processing layer.The LTC1069 semicustom filter tech-nology can integrate any single 8thorder or dual 4th order classical filterapproximation (Butterworth, Bessel,Chebyshev or elliptic) or any applica-tion specific filter response, in an8-pin SO package. Three new devicesillustrate this technology’s versatility.

LTC1069-1: Low PowerElliptic Filter Works fromSingle 3.3V to ±5V SuppliesThe LTC1069-1 has a frequencyresponse that provides a flat pass-band in combination with steepattenuation in the vicinity of the cut-off frequency. The cutoff frequency ofthe filter is clock tuned with a clock-to-cutoff-frequency ratio of 100:1. Thepassband is “flat” up to 95% of thecutoff frequency; the typical ripple is±0.1dB. Attenuations of 20dB at 1.2× fCUTOFF and 52dB at 1.4 × fCUTOFF areobtained. Beyond these frequenciesthe transition band keeps rolling offinstead of “bouncing back” the waytextbook elliptics do. Figure 1 illus-trates this “progressive” roll-off. The

attenuation floor reaches 82.5dB at2.3 × fCUTOFF with a ±5V supply and75dB with a single 3.3V supply. TheLTC1069-1 is designed for low power,single- or dual-supply, low frequencyantialiasing filter applications. Fig-ures 2 and 3 show typical connectionsfor single 5V and ±5V supplies. Theanalog ground of the device is inter-nally biased to one half the total powersupply voltage, thus eliminating theneed for additional external compo-nents and power waste. The low powerconsumption and speed obtainablefrom the part are shown in Table 1.

Despite the low power consump-tion of the device, extreme care wasapplied to maintain wide dynamicrange. The best dynamic range isobtained with ±5V supplies. Figure 4shows a S/(Noise+THD) ratio of bet-ter than 70dB for almost a decade ofinput signal range. The frequency ofthe input signal was set to 1kHz.

LTC1069-7 Linear-Phase,Raised-Cosine FilterThe LTC1069-7 lowpass filter usesexactly the same technology as theLTC1069-1, yet it is designed to per-form an entirely different task. First,a filter transfer function was synthe-sized to approximate a raised-cosineamplitude response with an alphafactor of one and linear phase in thepassband (Figure 5). A simplified dis-cussion of raised-cosine filters can befound in the boxed section on page

28. Digital communication systemsin need of pulse shaping and channelband limiting require the propertiesof raised-cosine filters. Note that theraised-cosine response is not relatedto the classical filter responses likeButterworth, Bessel or elliptic; itsrealization with classic RC active tech-niques requires many precisionpassive components and tuningadjustments. When compared to aconventional Bessel filter of the sameorder (for example, the LTC1064-3),the LTC1069-7 provides steeper roll-off at the vicinity of its cutoff frequency(see Figure 5). The impulse responseof the LTC1069-7 is also fully sym-metrical (see Figure 6) and is designedto shape an incoming pulse for opti-mum reception.

The LTC1069-7 trades power con-sumption and, to some degree,sampling rate, for speed. The maxi-mum cutoff frequency is 200kHz, thesampling-rate to cutoff-frequencyratio is 50:1 and the clock-to-cutofff requency rat io is 25:1. The

FREQUENCY (kHz)

–90

–40

–50

–60

–70

–80

10

0

–10

–20

–30

GAIN

(dB)

205.02.5 7.5 10 12.5 15 17.5

VS = 3.3V

VS = ±5V

GND

V+LTC1069-1

OR LTC1069-6

NC

VINVIN fCLK = 400kHz

8

7

6

5

1

0.47µF

0.1µF5V

2

3

4

VOUT VOUT

V –

NC

CLK

GND

V+

LTC1069-1

NC

VINVIN fCLK = 400kHz

8

7

6

5

1

0.1µF 0.1µF5V –5V

2

3

4

VOUT VOUT

V –

NC

CLK

Figure 1. LTC1069-1 elliptic filter frequencyresponse (fCUTOFF = 5kHz, fCLK = 500kHz)

Figure 2. Single 5V supply 4kHz ellipticlowpass filter

Figure 3. ±5V supply 4kHz elliptic lowpassfilter

VS I YLPPUS )pyT( f FFOTTUC )xaM(

V3.3 Am5.1 zHk5

V5 Am5.2 zHk8

± V5 Am2.3 zHk21

Table 1. LTC1069-1 power consumptionand speed


DESIGN FEATURES

TDECISION INSTANTS

OVERALL WAVEFORM

TIME

0100110

VOLTAGE

VT

fSAMPLING = 1 T

FREQUENCY

–50

–20

–30

–40

10

0

–10

GAIN

(dB)

20 0.25 0.50 0.75 1 1.25 1.50 1.75

ALPHA = 1

ALPHA = 0.5

Figure A. A sequence of pulses with minimum intersymbol interference at the output of a raised-cosine filter

Figure B. Gain versus frequency for raised-cosine filters

LTC1069-7 is pin compatible with theLTC1069-1. Table 2 illustrates themaximum cutoff frequency of the de-vice for different power supplies.

LTC1069-6: Single-Supply,Very Low Power, ProgressiveElliptic Lowpass FilterThe LTC1069-6 is designed for mini-mal power consumption and for single3V and 5V supply operation. Theamazingly low supply current of 1mA(3V supply) and 1.2mA (5V supply)includes the current spent to inter-nally bias the AGND pin for optimumvoltage swing when operating the parton a single supply. To save power, theclock-to-cutoff frequency ratio is low-ered to 50:1 but the sampling rate isstill 100:1. With a 50:1 clock-to-cut-off frequency ratio and for clocks equalto or less than 1MHz, cutoff frequen-cies within the audio band can beobtained.

The amplitude response of the fil-ter is also progressive elliptic, asillustrated in Figure 7, where the clockis set to 600kHz for a 12kHz cutofffrequency. Figure 7 also shows theresponse of the LTC1069-1 with aclock of 1.2MHz and a 12kHz cutofffrequency. When compared to theLTC1069-1, the LTC1069-6 has asharper roll-off in the vicinity of its

Raised-Cosine FiltersIn digital communications, infor-mation is transmitted and receivedas sequences of signal units of veryshort duration. Each signal (referredto as a symbol) generates a transientresponse as it is processed througha communications link. The detec-tion of valid symbols requires precisesampling every T seconds for a sig-naling rate of 1/T. During sampling,the decaying oscillations of a previ-ous symbol can add errors in thedetection process. When an incor-rect decision is made about thesymbol originally transmitted,intersymbol interference (ISI) is saidto occur. In order to minimize ISIwith as small a transmission band-width as possible, raised-cosine

filters are used in data transmissionto shape signals. When a pulsed signalis transmitted through a raised-cosinefilter, it will be received with mini-mum ISI, assuming the channel isproperly designed. Figure A illustrateshow two consecutive pulses can beshaped and properly detected withminimum ISI. Raised-cosine filtersare defined by the type of roll-off inthe stop band (often referred to as the“alpha” of the filter—Figure B).

One way to create a communica-tions channel with a flat frequencyresponse and low distortion is tomatch the receiver filter to the trans-mitter filter; if both filters are designedwith a root-raised-cosine response,the overall channel is raised-cosine.

For root-raised-cosine designs usingthe LTC1069 family of filters, pleasecontact LTC Marketing.

INPUT VOLTAGE (VRMS)0.1

–85

THD

+ NO

ISE

(dB)

–80

–75

–70

–65

–45

1 3

–60

–55

–50VS = ±5V fIN = 1kHz

Figure 4. LTC1069-1 THD + noise versusamplitude with ±5V supply

VS I YLPPUS f FFOTTUC

± V5 Am81 zHk002

V5 Am21 zHk041

Table 2. LTC1069-7 maximumcutoff frequencies


DESIGN FEATURES

Despite the LTC1069-6’s very lowpower consumption, its noise anddistortion performance are quite im-pressive. The single supply connectionof Figure 2 was used to operate theLTC1069-6 with a single 5V supply,600kHz clock and 12kHz cutoff fre-quency. A 2VP–P signal was sweptfrom 1kHz to 12kHz. The output ofthe filter was buffered to drive thecable and the input stage of the signalanalyzer. With 2VP–P input, the sig-nal-to-noise ratio measured at 1kHzwas 74dB; the –67dB worst-case har-monic distortion occurred for inputfrequencies of 4kHz, that is, at 1/3 ofthe filter cutoff frequency. Figure 8illustrates the above data. Table 3shows the maximum cutoff frequencythe device can reach and the corre-sponding power consumption. The

FREQUENCY (kHz)10

–70

GAIN

(dB)

–60

–50

–40

–30

10

100 1000

–20

–10

0

8TH ORDER BESSEL

LTC1069-7

VS = SINGLE 5V

FREQUENCY (kHz)

–80

0

–10

–20

–30

–40

–50

–60

–70

10

GAIN

(dB)

411 5 9 13 17 21 25 29 33 37

a: LTC1069-6 f CLK = 600KHz b: LTC1069-1 fCLK = 1.2MHz V IN = 2VP–P

a b

a b

FREQUENCY (kHz)

– 90

–65

–70

–75

–80

–85

–40

–45

–50

–55

–60

201 10

V S = 5V V IN = 2V P–P f CUTOFF = 12kHz

20 lo

g

(dB)

THD+

NOIS

EV I

N(

)

Figure 5. LTC1069-7 amplitude responsecompared to 8th order Bessel

Figure 6. The pulse response of theLTC1069-7 is fully symmetrical.

Figure 7. Amplitude response comparison:LTC1069-6 versus LTC1069-1

cutoff frequency; the key feature is42dB–50dB attenuation for frequen-cies between 1.27 × fCUTOFF and 1.31 ×fCUTOFF. Conversely, the LTC1069-1has better stop band attenuation anda flatter passband and is thereforemore selective right at the cutofffrequency.

LTC1069-6 can reach higher cutofffrequencies than the LTC1069-1 forless power. This is mainly due to thelower clock-to-cutoff frequencyratio.

Figure 8. LTC1069-6 signal to (noise + THD)ratio versus frequency

Table 3. LTC1069-6 maximum cutofffrequencies and resulting power consumption

VS

I YLPPUS)pyT(

f FFOTTUC)xaM(

f KCOLC)xaM(

V3 Am0.1 zHk41 zHk007

V5 Am2.1 zH02 zHM1

The Linear Technology WorldWide Web Site is Open

by Kevin R. Hoskins

Linear Technology Corporation’scustomers can now quickly and con-veniently find and retrieve the latesttechnical information covering thecompany’s products on LTC’s newinternet web site. Located atwww.linear.com or www.linear -tech.com, this site allows anyone withinternet access and a web browser tosearch through all of LTC’s technicalpublications, including data sheets,application notes, Linear Technologymagazine issues and other LTC pub-lications, to find information on LTCparts and applications circuits. Otherareas within the site include help,news and information about Linear

Technology and its sales offices. Thesite includes a map that easesnavigation through the different in-formation areas.

Other web sites usually require thevisitor to download large documentfiles to see if they contain the desiredinformation. This is cumbersome andinconvenient. To save you time andensure that you receive the correctinformation the first time, the firstpage of each data sheet, applicationnote, and LT magazine is recreated ina fast, download-friendly format. Thisallows you to determine whether thedocument is what you need, beforedownloading the entire file.

The site is searchable. Among thepossible search criteria are partnumbers, function, topics and appli-cations. The search is performed on a

user-defined combination of datasheets, application notes, design notesand Linear Technology magazine ar-ticles. Any data sheet, applicationnote, design note or magazine articlecan be downloaded or Faxed back.(Files are downloaded in Adobe Acro-bat PDF format; you will need a copyof Acrobat Reader to view or printthem. The site includes a link fromwhich you can download this pro-gram.)

The site also makes it possible toorder the LinearView CD-ROM,databooks, application handbooks,and other paper-based publicationsand make sample requests. Designtools, such as SwitcherCAD, Micro-PowerSwitcherCAD and FilterCAD,can be downloaded.

0.2V

/DIV

5µs/DIV


DESIGN IDEAS

Micropower ADC and DAC in SO-8Give PC 12-Bit Analog Interface

by Kevin R. HoskinsNeeding to add two channels of

simple, inexpensive, low powered,compact analog input/output to a PCcomputer, I chose the LTC1298 ADCand LTC1446 DAC. The LTC1298 andthe LTC1446 are the first SO-8 pack-aged 2-channel devices of their kind.The LTC1298 draws just 340µA. Abuilt-in auto shutdown feature fur-ther reduces power dissipation atreduced sampling rates (to 30µA at1ksps). Operating on a 5V supply, theLTC1446 draws just 1mA (typ). Al-though the application shown is forPC data acquisition, these two con-verters provide the smallest, lowest

power solutions for many other ana-log I/O applications.

The circuit shown in Figure 1 con-nects to a PC’s serial interface usingfour interface lines: DTR, RTS, CTSand TX. DTR is used to transmit theserial clock signal, RTS is used totransfer data to the DAC and ADC,CTS is used to receive conversionresults from the LTC1298 and thesignal on TX selects either theLTC1446 or the LTC1298 to receiveinput data. The LTC1298’s andLTC1446’s low power dissipationallows the circuit to be powered fromthe serial port. The TX and RTS lines

DI1466_01.eps

VOUTB

VCC

GND

AOUT2

AOUT1

VOUTA

CLK

DIN

CS/LD

LTC1446

DOUT

8

7

6

5

1

2

3

4

VCC

INPUT 1

INPUT 2

CLK

DOUT

DIN

CS

CH0 0.1µF

CH1

LTC1298

510Ω

510Ω

510Ω

510Ω

4 x 1N914

GND

8

7

6

5

1

2

3

4

Q

CLR

Q

D

PR

CK

1/2 74HC74

LT1021-5

47µFC4

150µF

5V

5

1

6

2

4

3

Q

CLR

Q

D

PR

CK

1/2 74HC745

1

6

10

5

13

1 51kTX

RTS

DTR

CTS

SELECT

DIN

SCLK

DOUT

51k

51k

11

6

12

6 2

4

9

3

8

4

D3 1N914

D4 1N914

2

4

3

7

0.1µF

14

5V

0.1µF

+

+

2

charge capacitor C4 through diodesD3 and D4. An LT1021-5 regulatesthe voltage to 5V. Returning the TXand RTS lines to a logic high aftersending data to the DAC or comple-tion of an ADC conversion providesconstant power to the LT1021-5.

Using a 486-33 PC, the through-put was 3.3ksps for the LTC1298 and2.2ksps for the LTC1446. Your mile-age may vary.

Listing 1 is C code that promptsthe user to either read a conversionresult from the ADC’s CH0 or write adata word to both DAC channels.Thecode is available on disk from LTC.

Figure 1. Communicating over the serial port, the LTC1298 and LTC1446 in SO-8 create a simple,low power, 2-channel analog interface for PCs.

Listing 1. Configure analog interface with this C code

#define port 0x3FC /* Control register, RS232 */

#define inprt 0x3FE /* Status reg. RS232 */

#define LCR 0x3FB /* Line Control Register */

#define high 1

#define low 0

#define Clock 0x01 /* pin 4, DTR */

#define Din 0x02 /* pin 7, RTS */

#define Dout 0x10 /* pin 8, CTS input */

#include<stdio.h>

#include<dos.h>


DESIGN IDEAS

#include<conio.h>

/* Function module sets bit to high or low */

void set_control(int Port,char bitnum,int flag)

char temp;

temp = inportb(Port);

if (flag==high)

temp |= bitnum; /* set output bit to high */

else

temp &= ~bitnum; /* set output bit to low */

outportb(Port,temp);

/* This function brings CS high or low (consult the schematic) */

void CS_Control(direction)

if (direction)

set_control(port,Clock,low); /* set clock high for Din to be read */

set_control(port,Din,low); /* set Din low */set_control(port,Din,low); /* set Din high to make CS goes high */

else

outportb(port, 0x01); /* set Din & clock low */

Delay(10);

outportb(port, 0x03); /* Din goes high to make CS go low */

/* This function outputs a 24-bit (2x12) digital code to LTC1446L */

void Din_(long code,int clock)

int x;

for(x = 0; x<clock; ++x)

code <<= 1; /* align the Din bit */

if (code & 0x1000000)

set_control(port,Clock,high); /* set Clock low */

set_control(port,Din,high); /* set Din bit high */

else

set_control(port,Clock,high); /* set Clock low */

set_control(port,Din,low); /* set Din low */

set_control(port,Clock,low); /* set Clock high for DAC to latch */

/* Read bit from ADC to PC */

Dout_()

int temp, x, volt =0;


DESIGN IDEAS

for(x = 0; x<13; ++x)

set_control(port,Clock,high);

set_control(port,Clock,low);

temp = inportb(inprt); /* read status reg. */

volt <<= 1; /* shift left one bit for serial transmission */

if(temp & Dout)

volt += 1; /* add 1 if input bit is high */

return(volt & 0xfff);

/* menu for the mode selection */

char menu()

printf(“Please select one of the following:\na: ADC\nd: DAC\nq: quit\n\n”);

return (getchar());

void main()

long code;

char mode_select;

int temp,volt=0;

/* Chip select for DAC & ADC is controlled by RS232 pin 3 TX line. When LCR’s bit 6 is set, theDAC is selected and the reverse is true for the ADC. */

outportb(LCR,0x0); /* initialize DAC */

outportb(LCR,0x64); /* initialize ADC */

while((mode_select = menu()) != ‘q’)

switch(mode_select)

case ‘a’:

outportb(LCR,0x0); /* selecting ADC */

CS_Control(low); /* enabling the ADC CS */

Din_(0x680000, 0x5); /* channel selection */

volt = Dout_();

outportb(LCR,0x64); /* bring CS high */

set_control(port,Din,high); /* bring Din signal high */

printf(“\ncode: %d\n”,volt);

break;

case ‘d’:

printf(“Enter DAC input code (0 – 4095):\n”);

scanf(“%d”, &temp);

code = temp;

code += (long)temp << 12; /* converting 12-bit to 24-bit word */

outportb(LCR,0x64); /* selecting DAC */

CS_Control(low); /* CS enable */



DESIGN IDEAS

High Efficiency, Low Power,3-Output DC/DC Converter

by John Seago

The recent proliferation of batterypowered products has created a lot ofinterest in low power, high efficiencyDC/DC converter designs. Theseproducts are small, lightweight andportable, so space for bulky batteriesis limited. Often, operating timebetween charges is a major sellingfeature, making the efficient use ofbattery power very important. Sincemany products cannot function witha single regulated voltage, multiple-output DC/DC converters arerequired.

Although developed for somewhathigher power levels, the single outputLTC1435 can be used in applicationsrequiring a very efficient, very small,low power, multiple-output DC/DCconverter (see Figure 1). This isaccomplished through the use of anoverwound buck inductor. With addi-tional windings, the inductor canprovide additional outputs, requiringonly a diode and filter capacitor for

each output. As with the less efficientflyback topology, the additional out-puts are not as well regulated as theprimary output, but the regulation issuitable for most applications.

The circuit of Figure 1 provides3.3V at 0.5A, 5V at 0.1A and –5V at0.05A, and has greater than 93%efficiency for test loads between 1.25Wand 2.4W with a 6V input. Load andline regulation of the positive outputsare quite good. Each output voltagewas measured with all output cur-rents varied independently between

20% and 100% of their full load range,while the input voltage was variedfrom 6V to 20V. Table 1 shows theworst-case output voltages measured.

The buck regulator with an over-wound inductor is a good solution forthose applications that do not havelarge load current or line voltage varia-tions. The smaller the load and linevariations, the smaller the voltagevariations on the overwound outputs.As a general rule, output voltage regu-lation is suitable for most applicationsif the switch duty cycle is kept between15% and 50% and minimum loadcurrent is kept above 20% of maxi-mum. Since load variation and linevariation have an additive effect onoutput voltage, applications withrelatively constant load currentrequirements can have a larger inputvoltage range and vice versa. For zerooutput current requirements, a smallpreload resistor can be used.

+

+

+

+

+

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

TG

BOOST

SW

VIN

INT VCC

BG

PGND

EXT VCC

U1 LTC1435CS

C6 0.001µF

C3 330pFC4, 47pF

C2, 0.1µF

C5 100pF

C1 68pF

6V TO 20V

R1, 10k

R3, 100Ω

C8 0.1µF

C7 22µF 35V

Q1 1/2 Si9936

C11 220pF

C12 100µF 10V

C13 100µF 10V

C14 100µF 10V

–5V/0.05A

3.3V/0.5A

5V/0.1A

GND

D4 MBR0540

D3 MBR0540

T1

T1

R5 0.1Ω

R6 35.7k

R7 20k

Q2 1/2 Si9936

D1, MBR0530

D2 MBRS130L

R4 10

C9, 0.1µF

C10, 4.7µF

R2, 100Ω

COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE–

SENSE+

DI1435_01.eps

SPECIAL PARTS R5 = IRC, LR2010-01-R100-J C7 = AVX, TPSE226M035R0300 C12, 13, 14 = AVX, TPSD107M010R0100 T1 = COILTRINOCS, CTX02-13299 Q1/Q2 = SILICONIX, Si9936DY U1 = LINEAR TECHNOLOGY, LTC1435CS

30µH

Figure 1. High efficiency, 3-output DC/DC converter

tuptuO muminiM mumixaM

V3.3 V703.3 V513.3

V5 V30.5 V42.5

V5– V89.4– V15.5–

Table 1. Worst-case output voltages


DESIGN IDEAS

Synchronizing LTC1430sfor Reduced Ripple by Craig Varga

The recent move to split-planemicroprocessors by several CPUmakers has led to the inclusion ofmultiple switching regulators on manymotherboard designs. These regula-tors typically provide 3.3V for systemlogic and a separate supply for theprocessor core. Current requirementsof 5A–10A or more per supply are notunusual. The LTC1430 synchronousbuck regulator is commonly used toprovide these tightly regulated sup-plies. By nature, the input currentwaveform in the buck topology is dis-continuous, resulting in large inputripple current. By synchronizing apair of supplies out of phase, it ispossible to achieve a degree of ripplecurrent cancellation. This results inless stress on the input capacitors(the number of input capacitors could

be reduced) and lower EMI. The rippleis easier to filter since the frequencyis effectively doubled and the peak-to-peak current is reduced.

It is extremely simple to synchro-nize a pair of LTC1430s in anappropriate phase relationship.Simply connect a resistor divider fromthe low gate drive of a “master” regu-lator to the sync pin of a “slave”regulator. The resistors should dividethe gate-drive voltage down to some-thing slightly less than the VCC supplyof the slave regulator, typically from12V down to approximately 4.5V. Totaldivider resistance of 20k to 30k isadequate. Also, the slave regulatormust be set up to free run slower thanthe master regulator. If, for example,the master is configured to run atapproximately 300kHz (a 130k resis-

tor from FSET to ground) the slavecan be left to run at its natural fre-quency of 200kHz. The slave frequencywill be forced up to that of the master.

The sync function on the LTC1430works as follows: when the shutdownpin is pulled low, the high-side switchturns off; normal duty factor controldetermines when the high-side switchwill turn back on. As long as theshutdown pin is held low for less thanapproximately 40µs, the chip will notshut down.

The simplified schematic (Figure1) shows the synchronization cir-cuitry. For a detailed description ofLTC1430-based regulator designs, seethe LTC1430 data sheet. The scopephoto (Figure 2) shows the voltage atthe common connection of the twoFETs of each regulator.

+

+

+

+

15

14

11

12

8

10

9

4

2

1

13

16

3

5

7

6

PVCC1

G1

IFB

G2

PGND

–SENS

+SENS

FB

U1 LTC1430

12V 5V

C3

C2

Q1

Q2

L1

R1 130k

PVCC2

Vcc

FSET

IMAX

SD

COMP

SS

SGND

R2 15k

R3 10k

15

14

11

12

8

10

9

4

2

1

13

16

3

5

7

6

PVCC1

G1

IFB

G2

PGND

–SENS

+SENS

FB

U2 LTC1430

12V 5V

OUT 2

OUT 1

C4

C1

Q3

Q4

L2

PVCC2

Vcc

FSET

IMAX

SD

COMP

SS

SGND

DI1430_01.eps

Figure 1. Simplified schematic diagram of synchronization circuitry

Figure 2. Phase relations looking at the switching nodes of bothregulators

Authors can be contacted at (408) 432-1900

MASTER

SLAVE


DESIGN INFORMATION

New Voltage References AreSmaller and More Precise by John Wright

IntroductionTwo new series voltage referencesbridge the gap between small pack-age size and high precision. Advancesin design, process and packaging havemade the introduction of these newvoltage references possible. The lowpower LT1460 is designed for mini-mum space and is available in allpopular voltages, including 2.5V, 5Vand 10V. By contrast, the LT1236 isdesigned for use in 12-bit systemsand combines 0.05% accuracy, lownoise and low drift. These new seriesreferences provide supply current andpower dissipation advantages overolder shunt references that must idlethe entire load current to operate.Furthermore, series references areseeing more duty as mini-regulatorsin data conversion and low powerelectronics.

The Small FryThe LT1460 is a tiny, low power,bandgap series reference with bigperformance. It uses curvature com-pensation to obtain low drift, andtrimmed precision thin-film resistorsto achieve high output accuracy. Itsperformance does not degrade ineither the SO-8 or MSOP packages.Surface mount packages are veryspace efficient, but the use of a largeoutput capacitor to stabilize the ref-erence defeats the purpose of thesmall package. The LT1460 is stablewith load capacitors, or with no ca-pacitor at all (see Figure 1). This canbe helpful when fast settling is a goal,

since load capacitors that change theirvoltages must recover before the ref-erence value is accurate.

The LT1460 uses so little PC boardspace that there is a temptation to

use the device as a “local” regulator.This is exactly what the LT1460 wasdesigned to do: deliver substantialload current while maintaining refer-ence-like features. At IOUT = 30mA,

Figure 1. LT1460 transient response forIOUT = 10mA, and CL = 0, 0.1µF, 1µF, 10µF

Table 2. Key specifications of the LT1236-10 voltage reference, SO-8 package

retemaraP snoitidnoC eulaV

ecnareloTegatloVtuptuOA0641TLB0641TL

75%0.0%01.0

tneiciffeoCerutarepmeTA0641TLB0641TL

C˚/mpp01C˚/mpp02

noitalugeReniL V03otV5.4 V/mpp01

gnicruoSnoitalugeRdaoL I TUO = Am03 Am/mpp07

egatloVtuoporD I TUO Am0= V7.0

tnerruCylppuS I TUO Am0= 001 µA

egakaeLesreveR I TUO V,Am0= NI V02–= 02 µA

egatloVesioNtuptuO zH1.0 ≤ f ≤ zH01 52 µV P–P

segatloVtuptuO V01,V0.5,V5.2

0˚C ≤ TJ ≤ +85˚C

Table 1. Key specifications of the LT1460 voltage reference, SO-8 package

retemaraP snoitidnoC eulaV

ecnareloTegatloVtuptuOA6321TLB6321TLC6321TL

%50.0%1.0%51.0

arepmeT tneiciffeoCerutA6321TLB6321TLC6321TL

C˚04– ≤ TJ ≤ C˚58+C˚/mpp5C˚/mpp01C˚/mpp51

noitalugeReniL V5.11 ≤ V NI ≤ V04 V/mpp4

gnicruoSnoitalugeRdaoL Am0 ≤ I TUO ≤ Am01 Am/mpp52

esioNtuptuO zH01 ≤ f ≤ zHk1 0.6 µV SMR

tnerruCylppuS I TUO Am0= Am7.1

10µF

1µF

0.1µF

0µF


DESIGN INFORMATION

TIME (DAYS)

LONG TERM STABILITY

–10

0

10

20

30

40

50

60OU

TPUT

VOL

TAGE

CHA

NGE

(PPM

)

35 42 49 56 63 70 77 84 910 7 14 2821

LT1236-5 SO-8

the load regulation is 70ppm/mA, yetthe LT1460 can withstand a short toground without being destroyed.

Additionally, if the power suppliesare reversed, the reverse batteryprotection keeps the referencefrom conducting current and beingdamaged (see Table 1).

Figure 2. LT1236 long-term stability

Higher Performance,Industrial TemperatureRange and Surface MountThe LT1236 is a precision referencethat combines ultralow drift and noisewith excellent long-term stability andhigh output accuracy. To addresssmall package requirements, the newreference is available in the SO-8

package and guarantees critical ref-erence parameters from −40˚C to 85˚C(see Table 2). The LT1236 output willboth source and sink 10mA, is almosttotally immune to input voltage varia-tions and, like the LT1460, is stablewith any load capacitor. Two outputvoltages are available: 5V and 10V.The 10V version can be used as ashunt regulator (2-terminal Zener)with the same precision characteris-tics as the 3-terminal connection.Special care has been taken to mini-mize thermal regulation effects andtemperature-induced hysteresis. TheLT1236 combines superior accuracyand temperature-coefficient specifi-cations, without using high power,on-chip heaters. The LT1236 refer-ences are based on a buried Zenerdiode structure that eliminates noiseand stability problems that plaguesurface-breakdown devices.

high as 4 watts. These losses areprimarily in the two MOSFETs, theinductor and the current sensingresistor. Since these are surfacemount components, the major ther-mal paths are through the pc boardcopper to the surrounding air. Maxi-mizing copper area around the heatproducing components, increasingboard area and using double-sidedboard with feedthrough vias all con-tribute to heat dissipation. Remember,the pc board is the heat sink.

One exception to the maximumcopper area rule is the switch nodeconsisting of Q1’s source, Q2’s drainand the left side of L1. This nodeswitches between ground and VIN at a200kHz rate. To minimize radiationfrom this node, it should be short anddirect. Other copper traces related toinput and output capacitors andMOSFET connections should also beas short as practical. See the LTC1435data sheet for information on goodlayout practices and additional appli-cations information.

Loads For TestingWhen testing the charger, several dif-ferent types of loads can be used. If

the battery to be charged is available,use it for the load. Alternatively, abattery can be simulated by adjust-ing a power supply to the nominalbattery voltage, preloading it with asmany amps as the charger is capableof delivering, (using power resistors)and then connecting it to the chargeroutput. A resistor can also be used fora load, but the charger may requireadditional output capacitance toprevent oscillations. This addedcapacitance is not needed when abattery is connected to the chargeroutput.

Noncharger ApplicationsThe LTC1435 and LT1620 can also beused for an adjustable current limit,high efficiency, low dropout powersupply in situations where higheroutput voltages (greater than the 10Vmaximum limit of the LTC1435 byitself) are needed.

When used for nonbattery applica-tions, more low ESR capacitance isneeded on the output and the com-pensation component values (R5, C13,and C14) may require changes. Pulseloading the output and checking thetransient response is a simple meansof checking loop stability.

Other Versionsof the LT1620The 16-pin version of the LT1620includes an additional, gain-of-20amplifier and a comparator with anopen-collector output. This outputcan be used for charge termination ina Li-Ion battery charger or as a signalto switch from a charge voltage to afloat voltage in a lead-acid batterycharger system.

Lithium-ion and lead-acid batter-ies are normally charged with acurrent limited, constant-voltagesource. As the battery charges, theinternal battery voltage rises and thecharging current begins to taper off.When the charging current drops to1/20 of the maximum programmedcharging current, the comparatorpulls low. Also available in a 16-pinDIP is a LT1621. This part containstwo independent current control loopsfor dual-loop control. The second con-trol loop can be used for input currentlimiting from the power source.

Refer to the LTC1435 and LT1620data sheets for addi t ionalinformation.

LT1620, continued from page 26


Din_(code,24); /* loading digital data to DAC */

outportb(LCR,0x0); /* bring CS high */

outportb(LCR,0x64); /* disabling ADC */

set_control(port,Din,high); /* bring Din signal high */

break;

–

+

–

+

R2R_08.eps

74HC04

VIN1

VOUT

5V

S/D

S/DLT1218

LT1218

5V

VIN2

INPUT SELECT

Figure 8. MUX amplifier

Figure 9. MUX amplifier waveforms

Rail-to-Rail, continued from page 19

PC Analog Interface, continued from page 32

100kHz 4th Order ButterworthFilter for 3V OperationThe filter shown in Figure 4 uses thelow voltage operation and wide band-width of the LT1498. Operating in theinverting mode for lowest distortion,the output swings rail-to-rail. Thegraphs in Figures 5–7 display themeasured lowpass and distortioncharacteristics with a 3V power sup-ply. As seen from the graphs, thedistortion with a 2.7VP–P output isunder 0.03% for frequencies up to thecutoff frequency of 100kHz. The stopband attenuation of the filter is greaterthan 90dB at 10MHz.

MultiplexerA buffered MUX with good offset char-acteristics can be constructed usingthe shutdown feature of the LT1218.In shutdown, the output of the LT1218assumes a high impedance, so theoutputs of two devices can be tiedtogether (wired OR, as they say in thedigital world). As shown in Figure 8,

the shutdown pins of each LT1218are driven by a 74HC04 buffer. TheLT1218 is active with the shutdownpin high. The photo in Figure 9 showsthe switching characteristics with a1kHz sine wave applied to one inputand the other input tied to ground. Asshown, each amplifier is connectedfor unity gain, but either amplifier orboth could be configured for gain.

ConclusionThe latest members of LTC’s family ofrail-to-rail amplifiers expand the ver-satility of rail-to-rail operation tomicropower and high speed applica-tions. The devices maintain precisionVOS specifications over the entire rail-to-rail input range and have openloop gains of one million or more.These characteristics, combined withlow voltage operation, makes for trulyversatile amplifiers.

FREQUENCY (Hz)

–110

–50

–60

–70

–80

–90

–100

0

–10

–20

–30

–40

10

GAIN

(dB)

10M

R2R_05.eps

100 100k 1M1k 10k

V S =3V V IN =2.7V P-P

AMPLITUDE (VP-P)

0.001

0.01

0.1

1

10

THD

+ NO

ISE

(%)

10

R2R_06.eps

0.01 0.1 1

V S =3V ƒ =20kHz

FREQUENCY (Hz)

0.001

0.01

0.1

1

10

THD

(%)

100k

R2R_07.eps

100 1k 10k

V S =3V V O =2.7V P-P

Figure 5. Filter frequency response Figure 6. Filter distortion vs amplitude Figure 7. Filter distortion vs frequency

VIN1

VOUT

INPUTSELECT

CONTINUATIONS


NEW DEVICE CAMEOS

LTC1473 PowerPath SwitchThe LTC1473 is a power managementIC designed to drive and protect twosets of back-to-back N-channelMOSFET switches in battery-oper-ated equipment such as notebookcomputers. The LTC1473 has a widesupply range (5V to 30V) to work withmost available battery packs and DCadapters. Within this supply range,the supply current is a mere 100µA.When V+ drops below 3.2V, theLTC1473 shuts down and draws just5µA of supply current.

An internal boost regulator pro-vides the voltage to fully enhance thelogic level N-channel MOSFETswitches. N-channel MOSFETs areideal for battery operated equipmentbecause of their extremely low RDS(ON)and small package size. Connectedback-to-back (actually, source tosource), two N-channel MOSFETs ina single package block current inboth directions when they are off. Aunique start-up mode allows the sys-tem to start regardless of which of thetwo sets of MOSFET switches receivespower first.

The LTC1473 uses a current senseloop to limit current flow through thebatteries and system supply capaci-tor during switchover transitions orduring a fault condition. When anN-channel MOSFET switch enters thecurrent limit condition, a fault timerwill start timing. If this conditionexceeds a user-programmable delaytime, the MOSFET switch latches off.Deselecting the MOSFET switch resetsthe latch.

The LTC1473 is available in a16-pin narrow SSOP package.

LT2078/LT2079and LT2178/LT2179Single Supply, Micropower,Precision Amplifiers inSurface Mount PackagesWith their outstanding DC precisionand low supply current (only 55µAand 21µA per op amp, respectively),the LT1078/LT1079 and LT1178/LT1179 have become true industrystandards. However, these deviceshave much better offset voltage and

New Device Cameos

For further information on anyof the devices mentioned in thisissue of Linear Technology, usethe reader service card or callthe LTC literature servicenumber:

1-800-4-LINEAR

Ask for the pertinent data sheetsand Application Notes.

Authors can be contacted at (408) 432-1900

offset voltage drift specifications inthe dual inline package (DIP) than inthe small surface mount package (SO).This is because the plastic surfacemount packages, in cooling, exertstress on the top and sides of the die,causing changes in the offset voltage.

In response to this problem, LTChas created the new LT2078/LT2079and LT2178/LT2179 single supply,surface mount op amps. These de-vices use a new, thin (approximately50 microns), jelly-like coating, ap-plied via a new dispensing system, toreduce stress on the top of the die.The result is that these parts havesignificantly superior VOS and VOSdrift specs than their predecessors.The circuit design and process usedfor the LT2078/LT2079 and LT2178/LT2179 is the same as that for theLT1078/LT1079 and LT1178/LT1179, resulting in identical ACperformance.

These new parts are available in 8-and 14-lead SO packages, for opera-tion over the commercial, industrialand extended temperature ranges.


Applications on DiskNoise Disk — This IBM-PC (or compatible) programallows the user to calculate circuit noise using LTC opamps, determine the best LTC op amp for a low noiseapplication, display the noise data for LTC op amps,calculate resistor noise and calculate noise using specsfor any op amp. Available at no charge.

SPICE Macromodel Disk — This IBM-PC (or compat-ible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be usedwith any version of SPICE for general analog circuitsimulations. The diskette also contains working circuitexamples using the models and a demonstration copyof PSPICE™ by MicroSim. Available at no charge.

SwitcherCAD — SwitcherCAD is a powerful PC softwaretool that aids in the design and optimization of switch-ing regulators. The program can cut days off the designcycle by selecting topologies, calculating operatingpoints and specifying component values andmanufacturer's part numbers. 144 page manual in-cluded. $20.00

SwitcherCAD supports the following parts: LT1070series: LT1070, LT1071, LT1072, LT1074 and LT1076.LT1082. LT1170 series: LT1170, LT1171, LT1172 andLT1176. It also supports: LT1268, LT1269 and LT1507.LT1270 series: LT1270 and LT1271. LT1371 series:LT1371, LT1372, LT1373, LT1375, LT1376 andLT1377.

Micropower SwitcherCAD — MicropowerSCAD is apowerful tool for designing DC/DC converters basedon Linear Technology’s micropower switching regula-tor ICs. Given basic design parameters,MicropowerSCAD selects a circuit topology and offersyou a selection of appropriate Linear Technology switch-ing regulator ICs. MicropowerSCAD also performscircuit simulations to select the other componentswhich surround the DC/DC converter. In the case of abattery supply, MicropowerSCAD can perform a bat-tery life simulation. 44 page manual included. $20.00

MicropowerSCAD supports the following LTC micro-power DC/DC converters: LT1073, LT1107, LT1108,LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,LT1300, LT1301 and LT1303.

Technical Books1990 Linear Databook, Vol I —This 1440 page collec-tion of data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conver-sion and interface products (bipolar and CMOS), inboth commercial and military grades. The catalogfeatures well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 pagesupplement to the 1990 Linear Databook is a collectionof all products introduced in 1991 and 1992. Thecatalog contains full data sheets for over 140 devices.The 1992 Linear Databook, Vol II is a companion to the1990 Linear Databook, which should not be discarded.

$10.00

DESIGN TOOLS

1994 Linear Databook, Vol III —This 1826 pagesupplement to the 1990 and 1992 Linear Databooks isa collection of all products introduced since 1992. Atotal of 152 product data sheets are included withupdated selection guides. The 1994 Linear DatabookVol III is a companion to the 1990 and 1992 LinearDatabooks, which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 pagesupplement to the 1990, 1992 and 1994 LinearDatabooks is a collection of all products introducedsince 1994. A total of 80 product data sheets areincluded with updated selection guides. The 1995Linear Databook Vol IV is a companion to the 1990,1992 and 1994Linear Databooks, which should not bediscarded. $10.00

1996 Linear Databook, Vol V —This 1152 page supple-ment to the 1990, 1992, 1994 and 1995 LinearDatabooks is a collection of all products introducedsince 1995. A total of 65 product data sheets areincluded with updated selection guides. The 1996Linear Databook Vol V is a companion to the 1990,1992, 1994 and 1995Linear Databooks, which shouldnot be discarded. $10.00

1990 Linear Applications Handbook, Volume I —928 pages full of application ideas covered in depth by40 Application Notes and 33 Design Notes. This cata-log covers a broad range of “real world” linear circuitry.In addition to detailed, systems-oriented circuits, thishandbook contains broad tutorial content togetherwith liberal use of schematics and scope photography.A special feature in this edition includes a 22-pagesection on SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —Continues the stream of “real world” linear circuitryinitiated by the 1990 Handbook. Similar in scope to the1990 edition, the new book covers Application Notes40 through 54 and Design Notes 33 through 69.Additionally, references and articles from non-LTCpublications that we have found useful are also in-cluded. $20.00

Interface Product Handbook — This 424 page hand-book features LTC’s complete line of line driver andreceiver products for RS232, RS485, RS423, RS422,V.35 and AppleTalk® applications. Linear’s particularexpertise in this area involves low power consumption,high numbers of drivers and receivers in one package,mixed RS232 and RS485 devices, 10kV ESD protec-tion of RS232 devices and surface mount packages.

Available at no charge

Power Solutions Brochure — This 80 page collectionof circuits contains real-life solutions for commonpower supply design problems. There are over 79circuits, including descriptions, graphs and perfor-mance specifications. Topics covered include batterychargers, PCMCIA power management, microproces-sor power supplies, portable equipment power supplies,micropower DC/DC, step-up and step-down switching

regulators, off-line switching regulators, linear regula-tors and switched capacitor conversion.

Available at no charge.

High Speed Amplifier Solutions Brochure —This 72 page collection of circuits contains real-lifesolutions for problems that require high speedamplifiers. There are 82 circuits including descrip-tions, graphs and performance specifications. Topicscovered include basic amplifiers, video-related appli-cations circuits, instrumentation, DAC and photodiodeamplifiers, filters, variable gain, oscillators and currentsources and other unusual application circuits. Avail-able at no charge

Data Conversion Solutions Brochure — This 52 pagecollection of data conversion circuits, products andselection guides serves as excellent reference for thedata acquisition system designer. Over 60 productsare showcased, solving problems in low power, smallsize and high performance data conversion applica-tions—with performance graphs and specifications.Topics covered include ADCs, DACs, voltage refer-ences and analog multiplexers. A complete glossarydefines data conversion specifications; a list of se-lected Application and Design Notes is also included.

Available at no charge

DESIGN TOOLS

AppleTalk is a registered trademark of Apple Computer,Inc.

Information furnished by Linear Technology Corporationis believed to be accurate and reliable. However, LinearTechnology makes no representation that the circuitsdescribed herein will not infringe on existing patent rights.

CD-ROMLinearView — LinearView™ is Linear Technology’sinteractive PC-based CD-ROM. LinearView allows youto instantly access thousands of pages of product andapplications information, covering Linear Technology’scomplete line of high performance analog products,with easy-to-use search tools.

The LinearView CD-ROM includes the complete prod-uct specifications from Linear Technology’s Databooklibrary (Volumes I–IV) and the complete ApplicationsHandbook collection (Volumes I and II). Our extensivecollection of Design Notes and the complete collectionof Linear Technology magazine are also included.

A powerful search engine built into the LinearView CD-ROM enables you to select parts by various criteria,such as device parameters, keywords or part numbers.All product categories are represented: data conver-sion, references, amplifiers, power products, filtersand interface circuits. Up-to-date versions of LinearTechnology’s software design tools, SwitcherCAD,FilterCAD, Noise Disk and Spice Macromodel library,are also included. Everything you need to know aboutLinear Technology’s products and applications is readilyaccessible via LinearView. Available at no charge.


InternationalSales OfficesFRANCELinear Technology S.A.R.L.Immeuble “Le Quartz”58 Chemin de la Justice92290 Chatenay MalabryFrancePhone: 33-1-41079555FAX: 33-1-46314613

GERMANYLinear Technology GmbHOskar-Messter-Str. 2485737 IsmaningGermanyPhone: 49-89-962455-0FAX: 49-89-963147

JAPANLinear Technology KK5F NAO Bldg.1-14 Shin-Ogawa-cho Shinjuku-kuTokyo, 162 JapanPhone: 81-3-3267-7891FAX: 81-3-3267-8510

KOREALinear Technology Korea Co., LtdNamsong Building, #403Itaewon-Dong 260-199Yongsan-Ku, Seoul 140-200KoreaPhone: 82-2-792-1617FAX: 82-2-792-1619

SINGAPORELinear Technology Pte. Ltd.507 Yishun Industrial Park ASingapore 2776Phone: 65-753-2692FAX: 65-754-4113

SWEDENLinear Technology ABSollentunavägen 63S-191 40 SollentunaSwedenPhone: 46-8-623-1600FAX: 46-8-623-1650

TAIWANLinear Technology CorporationRm. 602, No. 46, Sec. 2Chung Shan N. Rd.Taipei, Taiwan, R.O.C.Phone: 886-2-521-7575FAX: 886-2-562-2285

UNITED KINGDOMLinear Technology (UK) Ltd.The Coliseum, Riverside WayCamberley, Surrey GU15 3YLUnited KingdomPhone: 44-1276-677676FAX: 44-1276-64851

World Headquarters

U.S. AreaSales OfficesNORTHEAST REGIONLinear Technology Corporation3220 Tillman Drive, Suite 120Bensalem, PA 19020Phone: (215) 638-9667FAX: (215) 638-9764

Linear Technology Corporation266 Lowell St., Suite B-8Wilmington, MA 01887Phone: (508) 658-3881FAX: (508) 658-2701

NORTHWEST REGIONLinear Technology Corporation1900 McCarthy Blvd., Suite 205Milpitas, CA 95035Phone: (408) 428-2050FAX: (408) 432-6331

SOUTHEAST REGIONLinear Technology Corporation17000 Dallas ParkwaySuite 219Dallas, TX 75248Phone: (214) 733-3071FAX: (214) 380-5138

Linear Technology Corporation5510 Six Forks RoadSuite 102Raleigh, NC 27609Phone: (919) 870-5106FAX: (919) 870-8831

CENTRAL REGIONLinear Technology CorporationChesapeake Square229 Mitchell Court, Suite A-25Addison, IL 60101Phone: (708) 620-6910FAX: (708) 620-6977

SOUTHWEST REGIONLinear Technology Corporation21243 Ventura Blvd.Suite 227Woodland Hills, CA 91364Phone: (818) 703-0835FAX: (818) 703-0517

Linear Technology Corporation15375 Barranca ParkwaySuite A-211Irvine, CA 92718Phone: (714) 453-4650FAX: (714) 453-4765

Linear Technology Corporation1630 McCarthy BoulevardMilpitas, CA 95035-7417Phone: (408) 432-1900FAX: (408) 434-0507

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7417(408) 432-1900 FAX (408) 434-0507WWW.LINEAR.COMFor Literature Only: 1-800-4-LINEAR

© 1996 Linear Technology Corporation/ Printed in U.S.A./37.5K

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