planar antennas zÜrich presented by ing. él. dipl. epfl

Research Collection

Doctoral Thesis

Planar antennas for integrated front-ends

Author(s): Fries, Matthias

Publication Date: 2005

Permanent Link: https://doi.org/10.3929/ethz-a-004997970

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DISS. ETH No. 15880

PLANAR ANTENNAS

FOR

INTEGRATED FRONT-ENDS

A dissertation submitted to the

SWISS FEDERAL INSTITUTE OF TECHNOLOGY

ZÜRICH

for the degree of

Doctor of Sciences

presented byMATTHIAS FRIES

Ing. él. dipl. EPFL

École Polytechnique Fédérale de Lausanne EPFL, Switzerland

born October 22, 1973

citizen of Triengen (LU)

accepted on the recommendation of

Prof. Dr. R. Vahldieck, examiner

Prof. Dr. F. Gardiol, co-examiner

2004

Abstract

The success of wireless communication systems drastically increases the demand

for data transmission capacity. Because the available frequency spectrum is limited

the spectral efficiency of communication systems has to be increased. This can be

done by optimizing the radiation characteristics of the receiving and transmitting

antenna with regard to the transmission environment. At the same time there is

a strong pressure on costs of modern communications systems going mainly in the

commodity consumer market.

By integrating the antenna together with the front end electronics using printedcircuit board technology the system costs can be lowered and at the same time

active elements can be implemented in the antenna, allowing to actively control the

radiation properties. The two main printed antenna architectures are microstrip

patch antennas and slot antennas. Slot antennas show the advantage that they

allow the use of cheap dielectric substrates without suffering from substantial losses

and are better suited for the integration of active elements due to their uniplanar

architecture.

The focus of this thesis is to introduce new uniplanar antennas, which are based

on a low cost architecture and exhibit circular or switchable polarization. In a first

step the circularly polarized annular slot antenna will be investigated and several

low-cost designs will be introduced. Then two different techniques used to achieve

single-sided radiation will be introduced. The performance of the single-sided radi¬

ating and the double-sided radiating annular slot antenna will be compared. Finally

two different architectures allowing to switch the polarization of the annular slot

antenna are introduced. Simulation and measurement results of various polariza¬

tion switchable antennas will be shown. The antennas presented in this thesis are

interesting candidates for modern wireless communication systems due to the low

cost design and the suitability of integration with the front end electronics.

Kurzfassung

Der Erfolg drahtloser Kommunikationssysteme führt zu einem drastisch erhöhten

Bedarf an Übertragungskapazität. Da aber die verfügbare Frequenzbandbreite limi¬

tiert ist, muss die spektrale Übertragungsemzienz erhöht werden. Dies kann er¬

reicht werden, indem die Abstrahleigenschaften der Sende- bzw. Empfangsantenne

fortlaufend an die Umgebung angepasst werden. Gleichzeitig stehen drahtlose Kom¬

munikationssysteme unter einem enormen Preisdruck, da sie zu einem Konsumgut

geworden sind. Durch die Integration von Antenne und Front-End-Elektronik

auf einer einzigen gedruckten Schaltung, können sowohl die Systemkosten gesenkt

wie auch die Funktionalität der Antenne erhöht werden, indem die Abstrahleigen¬

schaften der Antenne mit Hilfe von aktiven Bauelementen gesteuert werden. Anten¬

nen, welche auf gedruckten Leiterplatten realisiert werden, können in zwei Haupt¬

gruppen unterteilt werden: Schlitzantennen und Mikrostreifenantennen. Kosten-

massig bieten Schlitzantennen gewichtige Vorteile gegenüber Mikrostreifenanten¬

nen. Sie benötigen nur eine einzige Metallschicht (uniplanar), lassen sich auf bil¬

ligen Dielektrika herstellen ohne bedeutende Verluste aufzuweisen und sind besser

geeignet für die Integration von aktiven Elementen.

Diese Dissertation präsentiert verschiedene Antennen, welche sich durch tiefe

Herstellkosten und zirkuläre oder umschaltbare Polarisation auszeichnen. Die An¬

tennen basieren auf der klassischen Schlitzringantenne (annular slot antenna). In

einem ersten Schritt werden eine Studie der zirkulär polarisierten Schlitzringan¬tenne gezeigt und verschiedene Billigvarianten dieser Antenne vorgestellt. Dann

werden zwei Techniken untersucht, die es erlauben eine einseitige Abstrahlcharak¬

teristik zu erzielen. Die Eigenschaften der einseitig abstrahlenden und der zweiseitig

abstrahlenden Schlitzringantenne werden miteinander verglichen. In einem letzten

Schritt werden zwei Antennenarchitekturen vorgestellt, die es erlauben, zwischen

verschiedenen Polarisationen umzuschalten. Die Tauglichkeit dieser Architekturen

wird an verschieden Beispielen demonstriert. Die Antennen, welche in dieser Dis¬

sertation präsentiert werden, eignen sich für die Integration in moderne Kommu¬

nikationssysteme.

Seite Leer /

Blank leaf

Contents

1 Introduction 1

1.1 Motivation and Objective 1

1.2 Advantages of Uniplanar Circuits 3

1.3 Focus of this Thesis 3

1.4 Outline 4

2 Uniplanar Antennas 5

2.1 Printed Circuits 5

2.2 Uniplanar Transmission Lines 6

2.2.1 Slotline 6

2.2.2 Coplanar Waveguide cpw 7

2.2.3 Coplanar Strip Line cps 8

2.2.4 Comparison 8

2.3 Classification of Uniplanar Antennas 9

2.3.1 Introduction 9

2.3.2 Rectangular Slot Antenna 11

2.3.3 Annular Slot Antenna (ASA) 11

2.3.4 Further Slot Antenna Shapes 12

2.4 Unidirectional Radiation 13

2.4.1 Reflector Backing 13

2.4.2 Electromagnetic Band-Gap Structures Reflector 14

2.4.3 Cavity Backing 15

2.4.4 Slot Coupled Patch Antenna 15

2.4.5 Dielectric Lens 15

3 Annular Slot Antenna 17

3.1 Introduction 17

3.2 Antenna Fields and Radiation Pattern 17

3.2.1 Aperture Fields 17

3.2.2 Radiation Pattern 19

3.2.3 Radiation Resistance 19

3.3 Numerical Simulation 20

3.4 Feeding Architectures 21

3.4.1 Slotline Transitions 22

3.5 Excitation of Circular Polarization 24

vi Contents

3.5.1 Introduction 24

3.5.2 Antenna Topology 25

3.5.3 Input Impedance 26

3.5.4 Radiation Characteristics of Circularly Polarized

ASA's 28

3.5.5 Summary 29

3.6 Parameter Study 29

3.6.1 Objectives 29

3.6.2 Antenna Size 30

3.6.3 Axial Ratio Bandwidth (ARB) 33

3.6.4 Input Impedance 35

3.6.5 Summary of Parameter Study 37

3.7 Examples 38

3.7.1 Objectives 38

3.7.2 Microstrip-Fed Circularly Polarized ASA 39

3.7.3 CPW-Fed Circularly Polarized ASA 39

3.7.4 Coaxial-Line Fed Circularly Polarized ASA 41

3.8 Chapter Conclusion 44

4 Unidirectional Radiation 45

4.1 Introduction 45

4.2 Cavity Backing 45

4.2.1 Previous Work 45

4.2.2 Objectives 46

4.2.3 Antenna Topology 46

4.2.4 Results of Linearly Polarized ASA 47

4.2.5 Results of Circularly Polarized ASA 50

4.2.6 Summary 54

4.3 Annular Slot Coupled Patch Antenna 55

4.3.1 Objectives 55

4.3.2 Study on Annular Slot Coupled Circular Patch

Antenna 56

4.3.3 Study on Annular Slot Coupled Annular Patch

Antenna 60

4.3.4 Example 62

4.3.5 Summary 65

5 Reconfigurable Slot Antennas 69

5.1 Introduction 69

5.1.1 Reconfigurable Antennas 69

5.1.2 Previous Work 70

5.1.3 Low-Cost Architecture 71

5.1.4 Chapter Outline 71

5.2 Uniplanar Switching Architecture 71

5.2.1 Switching Principle 71

Contents vil

5.2.2 PIN Diode 72

5.2.3 Bias Architecture 73

5.2.4 Antenna With Polarization Switching Between Linear And

Circular Polarization 74

5.2.5 Antenna With Polarization Switching Between RHCP And

LHCP 77

5.2.6 Slot-Coupled Annular Patch Antenna With Polarization Switch¬

ing Between Linear and Circular Polarization 81

5.2.7 Summary 84

5.3 Microstrip Switching Architecture 84

5.3.1 Switching Principle 84

5.3.2 Biasing Architecture 85

5.3.3 ASA With Switchable Polarization Between Linear Polariza¬

tion, LHCP and RHCP 87

5.3.4 Annular Slot-Coupled Circular Patch Antenna With

Switchable Polarization Between Linear Polarization, LHCP

and RHCP 94

5.3.5 Summary of Results 97


6 Conclusion 101

7 Outlook 105

Glossary 113

Acknowledgements 115

Curriculum Vitae 116

List of Publications 119

Chapter 1

Introduction

1.1 Motivation and Objective

In 1888 Heinrich Hertz was the first person who experimentally verified the existence

of electromagnetic waves. Only 13 years later in 1901 Guglielmo Marconi success¬

fully transmitted radio signals over the Atlantic sea between Poldhu, Cornwall, and

St. John's, Newfoundland, over a distance of 3400 Km. In 100 years between then

and today wireless communications experienced a stunningly fast evolution. Un¬

til the 1980's the research in wireless systems was driven by military applications

but since the last 20 years commercial wireless market drives the innovation. The

enormous success lead to constantly increasing demand of bandwidth and therefore

drove up the operating frequencies to the microwave band.

Wireless transmission systems use electromagnetic waves to carry information

through the ether. On the transmitting side of the wireless communication system

the so called baseband signal is modulated on a microwave carrier and radiated with

an antenna. On the receiving side the signal is received with an antenna and down-

converted to the base band. The unit, which performs the transformation from

the baseband signal to the radiated electromagnetic wave and vice versa is called

front end. It includes mixers, amplifiers and filters for the frequency conversion

of the signal and an antenna for radiation or reception. In classical front end

architectures antenna and signal processing electronics are separated parts. The

tendency goes however more and more to front ends with integrated antennas,

which means, that antenna and signal processing electronics are realized with the

same technology. With the apparition of microstrip patch and slot antennas it is

possible to integrate electronic rf-circuits and antennas on the same printed circuit

board. Besides the cost reduction, this approach yields several further advantages.

The antenna input impedance need not to be matched necessarily to 50 £1 but can

be adapted to the input impedance of the signal processing electronics, which allows

more design freedom. Active elements can be integrated directly on the antenna

to make its properties, such as polarization, radiation pattern or input impedancetuneable. Antennas whose properties can be tuned with active circuits are called

reconfigurable antennas.

2 Introduction

Frequency spectrum is a scarce good. Therefore various technologies were de¬

veloped in order to increase the capacity of the used frequency bands. Modulation

techniques such as TDMA (Time Division Multiple Access) and CDMA (CodeDivision Multiple Access) allow a flexible bandwidth management and therefore

an increased capacity. Besides the optimisation of the modulation technique the

capacity of a given frequency band can be increased by exploiting the radiation

characteristics of the wireless communication system. Cellular technique limits the

radiation to a specified area called cell. Outside this cell the same frequency band

can be reused. This technique is widely used for commercial mobile phone systems.

A further capacity increase can be achieved with diversity techniques. Hereby

the antenna radiation characteristics are continuously adapted to the environmental

conditions using an active circuit in order to maximize the signal quality. This

can be done by pointing the antenna beam in the maximum signal direction or

by creating a null in the direction or arrival of an unwanted signal. Besides the

radiation direction the polarization of the radiated fields can be exploited as well. In

a propagation environment having few reflections two orthogonal polarizations can

be used in order to double the channel capacity. In a highly reflective propagation

environment suffering from fading, e. g. in buildings, the signal strength can

be optimized using an antenna which can switch between different polarizations.

Considerable differences in the fading level can occur for different polarizations. In

low-power systems, polarization diversity can also be used as a modulation scheme.

With the introduction of the different diversity techniques the requirements on

the antenna performances have been increased. Flexible bandwidth and polarization

control and beam or null steering capabilities are needed. The antenna, which used

to be a simple radiator, is now additionally in charge of certain 'signal processing'

tasks. This lead to the development of reconfigurable antennas, which are capable

of varying their radiation characteristics.

Besides the performance, the production-cost is a key factor for the commercial

success of antennas and entire front ends. Not the technically superior structure is

required but the technically sufficient structure. Low cost front end architectures

allowing to integrate antenna and signal processing electronics are required. In this

thesis novel passive and reconfigurable antenna architectures based on a uniplanarand quasi-uniplanar printed circuit architecture will be introduced and systemati¬

cally investigated. Uniplanar printed circuits consist of a substrate layer whose one

side is metalized. In contrary to microstrip printed circuits the opposite side of the

substrate is completely non metalized. The microwave circuit is realized on this sin¬

gle metalization layer which makes it easy and cheap to produce. Quasi-uniplanarcircuits still comprise a second metalization-layer but the major part of the circuit

is realized in uniplanar technology.

1.2 Advantages of Uniplanar Circuits 3

1.2 Advantages of Uniplanar Circuits

In recent years printed circuit technology has been established in most microwave

areas and in particular also for antennas. The advantages of printed circuits are

their suitability for low-cost production, their low profile and the ease of mounting

of active elements. Due to advances in dielectric materials a big variety of low loss

substrate materials are available today. The most widely used printed transmission

line is the microstrip line. However uniplanar transmission lines which are coplanar

waveguide (cpw), slotline or coplanar strip line (cps) gain more importance [1,

2]. Only one metalization layer is needed, which simplifies the production. In

contrary to the microstrip line the major part of the electromagnetic field resides

in the air and not in the substrate. This allows to use lossy and therefore cheap

substrate materials without introducing significant losses but drastically reducing

the production costs.

During the last decade the cpw became popular for state-of-the-art microwave

monolithic integrated circuit's (mrnic). The integration of lumped elements and

active devices is more convenient than in microstrip technology because series and

shunt mounting can be realized without the need of via holes. At millimeter-wave

frequencies microstrip technology becomes problematic because of dispersion and

unwanted surface wave excitation. This problem is less severe in uniplanar circuits.

Investigations on uniplanar antennas have shown, that compared to microstrip

antennas they exhibit lower mutual coupling, lower losses at millimeter-wave fre¬

quencies and that they require a smaller area [3].

1.3 Focus of this Thesis

Uniplanar and quasi-uniplanar technology is well suited for antennas and integratedfront ends of modern commercial wireless communication systems. This technology

allows the integration of electronics into the antenna, leads to low production costs

due to the compatibility with cheap substrate materials, and offers easy integration

of lumped elements. Few works, mostly on linearly polarized structures, have been

published on uniplanar antennas so far. However, modern communication systems

require also circularly polarized and polarization switchable radiation elements.

The focus of this thesis is to introduce new uniplanar antennas to realize circular

and switchable polarization. The antennas are based on the classical annular slot

antenna (also called slot ring antenna). A rigorous study on circularly polarizedannular slot antennas will be presented. Different feeding solutions will be intro¬

duced. Classical slot antennas exhibit a double-sided radiation pattern. In this

thesis annular slot architectures showing single sided radiation will be introduced

and compared to double sided radiating structures. Finally different active polariza¬

tion switchable slot ring architectures will be shown. These architectures are very

interesting because the uniplanar architecture allows an easy mounting of active

elements and does not require expensive microwave substrates. The circular shapeof the annular slot radiator does not enforce a specific polarization and is therefore

4 Introduction

an ideal candidate for polarization switchable antennas. Single and double sided

radiation patterns can be achieved. The presented architectures are interesting can¬

didates for modern wireless communication systems. The results published in this

thesis have been picked up already by international research groups such as [4].

1.4 Outline

This thesis is structured in 5 chapters. In chapter 2 the work of this thesis is

put into context with the international literature on this topic. Uniplanar circuits,

their advantages and drawbacks will be introduced and compared to other printed

circuits. A brief overview of the state of the art of uniplanar antennas will be given.

In chapter 3 the annular slot antenna will be introduced. A rigorous study

of different circularly polarized annular slot antennas will be presented. Different

feeding solutions will be introduced. Several examples of novel broadband low-cost

circularly polarized annular slot ring antennas will be presented.The antennas presented in chapter 3 exhibit double-sided radiation patterns.

Chapter 4 presents two different architectures allowing to achieve single-sided radi¬

ation. One architecture consists of coupling the slot circuit feed to a patch located

on top of the slot ring in order to achieve single sided radiation, whereas the other

method consist of backing the slot ring with a metal cavity. Merits and drawbacks

of the two architectures will be compared.In Chapter 5 different reconfigurable annular slot antennas will be presented.

By introducing active elements the polarization of the presented antennas can be

switched between different states. Furthermore, this architecture allows the tuningof the operation frequency of the antenna. Reconfigurable antennas with single

sided as well as with double sided radiation patterns will be demonstrated. The

presented architecture is cost effective because it requires few active and passive

lumped elements and does not require the use of expensive rf-substrates and does

not suffer from noticeable losses.

Finally, the conclusions of this work will be drawn.

Chapter 2

Uniplanar Antennas

2.1 Printed Circuits

Printed circuits consist of a thin sheet of low-loss insulating material, called the

dielectric substrate. Both sides of the substrate can be covered with a thin con¬

ducting metal layer. By using a chemical etching process parts of this metal layer

can be removed. In such a way all imaginable shapes of two dimensional printed

circuits can be produced. There exist different printed transmission lines allowing

to guide electromagnetic waves in a tightly bound mode. The most commonly used

printed waveguide is the microstrip line. It consists of a dielectric whose one side is

completely metalized (groundplane) and whose other side contains a metal strip of

a constant width. The width (w) of the strip as well as the thickness (h) and the

permittivity (er) of the substrate determine the impedance of the microstrip line.

The electromagnetic fields of the microstrip mode are mainly concentrated between

the strip and the groundplane but are also partially located in the air. Figure 2.1

shows a two-dimensional and a three-dimensional view of a microstrip line as well as

the transverse electrical fields of the microstrip mode. Besides simple transmission

lines for interconnect purposes various different circuit elements such as resonators,

filters, circulators, power splitters etc. can be realized and are widely used.

Microstrip antennas consist usually of microstrip resonators. There exist hun¬

dreds of resonator shapes but the most commonly used are either rectangular or

circular patches, so called patch antennas [5, 6]. In the case of patch antennas

the resonator field is concentrated between patch and groundplane. The radiation

takes place at the edges of the the patch where the resonator field is exposed to

free space. The radiation surface (two edges) is small compared to the total size of

the resonator which leads to a small radiation resistance and therefore to a small

impedance bandwidth (typically below 1%) of these antennas. The bandwidth can

be increased, however, by replacing the substrate with air and/or by exciting sev¬

eral resonances. Various modified patch antenna structures exhibiting impedance

bandwidths of more than 50% have been proposed [7]. Due to the groundplane,

microstrip patch antennas exhibit a single sided radiation pattern with a high front

to back ratio.

6 Uniplanar Antennas

Electric field

Magnetic field

a) b)

Figure 2.1. Microstrip line; a) Cross section view showing the transverse electric and

magnetic fields of the microstrip mode; b) 3-dimensional view

2.2 Uniplanar Transmission Lines

Uniplanar transmission lines consist of printed circuits where only one side of the

substrate is metalized. The three most commonly used uniplanar transmission lines

are the coplanar waveguide (cpw), the slotline and the coplanar stripline (cps).

2.2.1 Slotline

The slotline consists of a slot in a conducting plane [8, 9], usually realized on top

of a substrate by using chemical etching. The slotline configuration and the trans¬

verse electric and magnetic fields of the dominant mode are shown in Figure 2.2.

Both, electric and magnetic field, exhibit transversal and longitudinal components.

Therefore the mode is hybrid. However the longitudinal component of the electric

field is weak. For a waveguiding structure to be useful as a transmission line or a

circuit element it is necessary to confine the fields near the structure. A field analy¬

sis shows, that the higher the permittivity of the dielectric the faster the transversal

fields decay normal to the slot, which means the mode is closely confined to the

slot area [10, 11], Practically substrates having a permittivity higher than er = 10

have to be used. Slotlines are not efficient waveguiding structures because they

are dispersive and tend to radiate easily in case of perturbations. Furthermore very

narrow slot widths are required to achieve a characteristic impedance of 50 Ü, which

leads to high losses. Therefore they are impractical for interconnection purposes in

printed circuits. Several slotline circuits such as baluns, filters, branch line couplers,

magic-T circuits and various active circuits [12] have been reported but they are

seldomly used. Slotlines have gained some interest in hybrid circuits where they are

located in the groundplane of a microstrip circuit [13].For antenna applications however, slotlines are interesting candidates, because

they radiate easily. Various linearly polarized but also a few circularly polarized

2.2 Uniplanar Transmission Lines

Electric field

Magnetic field

a) b)

Figure 2.2. Slotline; a) Cross section view showing the transverse electric and magnetic

fields of the slotline mode; b) 3-dimensional view showing the electric and magnetic

fields of the slotline mode

slotline antenna designs have been published [5]. Compared to microstrip lines,

slotlines have the advantage that they allow an easy integration of active elements

in both, series and shunt configuration, no via holes are needed and the uniplanar

architecture makes production easier.

2.2.2 Coplanar Waveguide cpw

The cpw consists of two parallel slots (also called gap) in a metalized plane. It was

first proposed by Wen [14]. The impedance of the cpw is determined by the slot

width w and the width of the central conductor s, the thickness h of the substrate

and the permittivity (eT) of the substrate. The cpw supports two fundamental

modes, which are called even mode and odd mode. Figure 2.3 shows the fields of

the usually used even mode. The electrical field is symmetric to the center plane

Electric field

Magnetic field

w

<*

>

\\

/

v' J ,'„ V BZjiji

b)

Figure 2.3. Coplanar waveguide cpw; a) Cross section view showing the transverse electric

and magnetic fields of the even mode; b) 3-dimensional view.


of the line, which means that the electric fields across the slots show opposite

directions. Consequently the two outer conductors, which are the signal ground,

are on the same electrical potential. In contrary to this the electric fields of the

odd mode have the same direction across the slot and the potential of the two outer

conductors are therefore unequal. Actually the cpw odd mode is very similar to the

slotline mode. It is dispersive, radiates easily and is therefore unwanted. It can be

suppressed by shorting the two outer conductors with a so called air bridge, forcing

the two outer conductors to the same electrical potential. Throughout this thesis

it will be assumed that the cpw is operated only with the even mode.

The even mode of the cpw is quasi-TEM ,its fields are well bound to the slot area

and radiate less easily than the slotline mode. Various coplanar circuit elements

exist such as filters, couplers, antennas, etc.

2.2.3 Coplanar Strip Line cps

The coplanar strip line consist of two printed metallic strips located on the same

side of the substrate. It is the complementary structure to the cpw. The transversal

electric and magnetic fields correspond to the magnetic and electric fields of the cpw,

respectively. The cps can also be seen as a printed two-wire line. This transmission

line is seldomly used due to its balanced nature except for transitions between

different lines or for feeding of balanced devices (e.g. printed dipole antennas).

2.2.4 Comparison

Comparing cpw and slotline it turns out, that depending on the application, one or

the other is more advantageous. The cpw even mode is well suited for waveguiding,

filter applications, branchline couplers. In all these applications the field has to be

confined to the strip-slot area. It requires however a more complicated production

process because airbridges are needed to short the unwanted odd mode. The slotline

is not well suited for interconnects because the field is less confined to the slot

area, especially for permittivities below eT —10. Furthermore the slotline radiates

easily if the mode is disturbed by a perturbation. Thus slotlines are interesting as

radiators. Especially when a low permittivity substrate is used slotline resonators

radiate easily. Slotline resonator antennas are more broadband than microstrip

patch antennas due to the larger radiating surface. The entire slot resonator surface

is radiating whereas in the case of the patch antenna the radiation takes place only

at the edges of the resonator.

The losses of printed transmission lines have been published in [15] for a sub¬

strate with a permittivity of er~10 and a thickness h = 0.635 mm. For a given

frequency the losses increase with a decreasing slot width in the case of cpw, cps

and slotline or a decreasing strip-width in the case of a microstrip line, respec¬

tively. For a given slot or strip width the slotline shows the lowest losses of all lines

followed by the cps, the microstrip line and the cpw. Usually it is however more

meaningful to compare the conductor losses of these transmission lines having the

same characteristic impedance instead of having the same width. Comparing these


lines having a characteristic impedance of 75 0, it turns out, that the slotline still

exhibits the lowest losses followed by the cpw, the microstrip line and the cps. In

order to achieve a characteristic impedance of 50 f2 with a slotline the slot width

becomes very narrow. For the above mentioned substrate values, e. g., the slot

width for a 50 S~2 slotline would be about 60 ^m, which leads to conductor losses

that are more than double of any other line with the same impedance. To conclude

the loss discussion it can be stated that the slotline losses are high if it is used as

a 50 tt transmission line, but they are low if the slotline width is in the millimeter

range.

2.3 Classification of Uniplanar Antennas

2.3.1 Introduction

Uniplanar antennas are usually realized on a dielectric material using one single

metalization layer. Two basic elements exist to build printed uniplanar anten¬

nas: printed strips and slots. Two typical printed strip uniplanar antennas are the

printed dipole antenna and the printed loop antenna. Both can be fed with a cps.

Typical uniplanar slot antennas are the cpw-fed slot dipole antenna or the cpw-

fed slot loop antenna. This thesis treats exclusively slot based antennas, because

they are better suited for the integration of lumped and active elements where a

groundplane is required.

Printed Resonator Antennas

The purpose of an antenna is to radiate guided waves into free space or vice versa.

Therefore, the antenna has to transform the guided mode of the feed into a free

space wave. This can be accomplished with a resonator whose fields are not totally

enclosed in a metal structure but are at least partially open to free space in order

to be able to radiate. Before steady state is reached the incoming power from the

feed is partially radiated and partially stored in the resonating mode at resonance,

but not reflected back into the feed line (in case of a matched antenna). The higherthe resonator fields the higher will be the radiated power. The steady state is

reached when the magnitude of the resonator mode increased to the point where

the radiated power is equal to the incoming power from the feed. The radiation

quality factor Qr is defined as a ratio between the total energy in the resonator Wt

and the radiated power Pr [5]:

Qr =^ (2.1)

The quality factor associated with the conductor losses Qc, the dielectric losses Qdand the surface wave losses Qaurf are defined in a similar way [5]. The total qualityfactor QT of the antenna can be defined in terms of the quality factors associated

with the different losses:

11111M,N

—— 1 1- — -| (2.2)

Qt Qr Qd Qc Qaurf


If low loss materials are used the radiation loss dominates and the other losses

can be neglected.The impedance bandwidth of an antenna is inverse proportional to the total

quality factor. For a feed line Voltage Standing Wave Ratio being smaller than a

given parameter S (VSWR < S) the bandwidth BW is given by [16]:

q_

1

BW ==—~

Q^/S

The two main types of printed resonator antennas, the microstrip patch antenna and

the slot antenna, show fundamental differences in terms of radiation characteristics,

impedance bandwidth and dielectric losses, which is due to the different kinds of

modes. In the case of microstrip patch antennas the resonator fields are enclosed

between the patch and the groundplane. Radiation takes place only at the open

edges of the resonator, where the fields are exposed to free space. The size of the

radiating apertures are determined by the width of the patch, which corresponds to

about one half of the guided wavelength and the height of the substrate. Because

only a small part of the resonator field contributes to the radiation the ratio between

the stored reactive energy and the radiated energy hence the quality factor is high.

This leads to low impedance bandwidths. For a regular substrate, e. g. er = 2.2

and h —0.635 mm, the 10 dB-impedance bandwidth is below 1%.

In the case of the rectangular slot antenna, the resonating fields consist of a

slotline standing wave. In contrary to the patch antenna the entire resonating

fields are open to free space and contribute to the radiation. Therefore for a given

radiated power less reactive energy is stored in the resonator than in the case of

a patch antenna, which leads to a lower quality factor and therefore to a higher

bandwidth. The radiating aperture can be simply increased by enlarging the slot

width whereas in the case of the patch antenna is is determined by the thickness

of the substrate. Typical 10 dB-impedance bandwidths of slot antennas using the

above mentioned substrate are between 10% and 15%.

Losses in Printed Resonator Antennas

If good conductors (copper, silver) and low loss microwave substrates (tan<5 <

0.001) are used the conductor and substrate losses are negligible compared to the

radiation. For lossy substrates however, e. g. FR4 with tan 5 — 0.02, the losses

become significant and have to be considered. The dielectric power loss P<i can be

calculated with the following formula [5]:

Pa =ueoertenö JJJ ]Ef dV = „tmSWr (2.4)

antenna volume

where Wr corresponds to the total energy in the antenna. From this formula it can

be concluded, that for a given substrate the dielectric losses are directly proportionalto the stored energy and therefore to the quality factor of the antenna. Consequently

slot antennas exhibit lower losses, than patch antennas, due to the lower quality

factor. In the above mentioned comparison between a rectangular patch antenna

(2.3)


and a rectangular slot antennas, e. g., the dielectric losses of the slot antenna are

about 10 times lower than the losses of the patch antenna.

Traveling Wave Antennas

There exist also non-resonating antennas, so called traveling wave antennas. They

can be seen as transmission lines, which transform the feed line impedance to the free

space impedance. Typical examples are the vivaldi antenna or the axial mode helix

antenna. Traveling wave antennas exhibit usually a larger bandwidth. However

the size of them is in the order of several wavelengths whereas the size of resonator

antennas is about half a wavelength. In this thesis the work is concentrated on

slot-based resonator antennas.

2.3.2 Rectangular Slot Antenna

The simplest slot antenna shape is a rectangular slot. Figure 2.4 shows the geometry

of a rectangular printed slot antenna. It has a length I and a width w and resonates

for a length of approximately half a wavelength. The electric field is polarized

across the slot. It is zero at the two ends of the resonator due to the shorting

and maximum in the center of the slot. The rectangular slot antenna is linearly

polarized and shows a very low cross polarization (<-35dB) [17]. Different feeding

techniques can be used. The first microstrip slot antennas have been excited by

strip lines [18, 19]. However a strip line feed shows the disadvantage of exciting

parallel plate modes beside the wanted slot antenna mode. Various microstrip fed

slot antennas have been published and matching techniques have been proposed

[20, 17].For uniplanar structures a microstrip feed is not possible but cpw or slotline

feed have to be used. Figure 2.4 shows different feed lines. Figure 2.4 a) shows a

top view and exploded view of a microstrip fed rectangular slot antenna. In b) a

slotline-feed and in c) a cpw feed are shown, respectively. Note that in the case

of the cpw-feed the length of the antenna is about one wavelength [21]. Actually

it consists of two half wavelength resonators. The typical impedance bandwidth

of the standard printed slot dipole is about 12%-20% which is much larger than

the typical impedance bandwidth of a microstrip antenna. By adding a second

resonator to the cpw-fed slot dipole an impedance bandwidth of 49% has been

achieved [22].

2.3.3 Annular Slot Antenna (ASA)

The annular slot antenna (ASA) is also called slot ring antenna. It consists of

an annular slot in a metal groundplane with radius r and a slot width w. The

first resonant mode occurs for a circumference equal to one guided wavelength.The first resonant mode, the radiation pattern and the input impedance have been

rigourously investigated in [23, 24]. The ASA supports two orthogonal degeneratedmodes whose field maxima are located 90 degrees apart of each other.


ca. X/2

a) Microstrip fed rectangular slot antenna

ca. X/2

b) Slotline feed c) CPW-feed

Figure 2.4. Rectangular slot antenna; a) microstrip feed; b) slotline feed; cpw-feed

The ASA is a very interesting radiation element for several reasons. It supports

two orthogonal degenerated modes, which allows to produce any wanted polariza¬tion by properly adjusting the relative magnitude and phase of the two modes.

The polarization direction is not imposed by the shape of the resonator but by the

feed. Due to the rotational symmetry the resonant mode can be polarized in any

direction. The uniplanar architecture is well suited for the integration of lumpedelements. This has been demonstrated in [23] for a quasi optical mixer integratedin an ASA.

2.3.4 Further Slot Antenna Shapes

Besides the rectangular and the annular slot antenna various other slot antenna

shapes exist. In this paragraph some interesting ones will be mentioned for the


sake of completeness. An often used shape is the rectangular slot antenna or the

folded slot dipole antenna. The radiation characteristics of the rectangular slot

loop antenna are similar to the circular slot loop antenna whereas the radiation

characteristics of the folded slot antenna are similar to the rectangular slot antenna.

In [25] a rigorous study of the folded slot antenna was published. A method to

control the input impedance is proposed. An analysis of the cpw-fed rectangular slot

loop antenna and a circuit containing an oscillating rectangular slot loop antenna

have been published in [26].A well known technique to increase the impedance bandwidth of antennas is to

excite several resonant modes. In [27] a so called cpw-fed cusp antenna yielding an

impedance bandwidth of 34% is presented. The geometry was based on a circular

slot-loop where the slot width is continuously widened along the line. Two different

resonances are excited in the slot. A cpw-fed bow-tie antenna exhibiting an impe¬

dance bandwidth of 34% is presented in [28]. Widening of the radiating slot element

allows a larger bandwidth but shows also several drawbacks. The diffraction effects

become more important and the radiation pattern shows more ripples. The control

of the polarization is more difficult especially to achieve a good polarization purity

over the entire impedance bandwidth. Wide slots do not have a uniform phase dis¬

tribution of the electrical fields which leads to tilted far field pattern. Furthermore,

the wider the slot the less the wave is bound and the easier the antenna is perturbed

by nearby objects. In [29] a cpw-fed slot antenna with a wide radiation aperture

exhibiting a single sided radiation pattern and an impedance bandwidth of 50%

was presented. The cross-polarization was reduced by the use of metal strips short

circuiting the unwanted field components.

Various traveling wave slot antennas can be found in the literature, e. g. [30, 31].

2.4 Unidirectional Radiation

A slot antenna is open to two sides and exhibits therefore a bidirectional radiation

pattern. For various applications, however, a single sided radiation pattern may

be required. Several techniques for achieving unidirectional radiation have been

published in literature. In this section an overview of the mostly used techniques

will be given. Some of these techniques have been investigated in this thesis.

2.4.1 Reflector Backing

The easiest approach is to place a conducting plane parallel to the slot plane in a

distance of a quarter wavelength. In this way the back side radiation is reflected back

towards the slot. The reflected wave undergoes a total phase rotation of 360 degrees.180 degrees are due to the propagation distance (twice a quarter wavelength) and

180 degrees are due to the reflection on the conducting plane. However, not all the

reflected energy will be radiated. Part of the energy couples into a parallel plate

mode propagating between the slot plane and the reflector. A study of a microstrip

line fed rectangular slot antenna with conductor backing showed, that a spacing


of a quarter wavelength between slot and reflector was ideal for the impedance

matching [32]. However no data about front side gain increase due to the reflector

was reported. A second study on a reflector backed rectangular slot antenna is

published in [33]. Without the reflector the antenna radiates 50% of the energy to

the front side and the back side, respectively. Using an infinitely extended backside

reflector at a distance of ^ from the slot plane the backside radiation is completely

suppressed and the front side radiation increases to 56%, which is rather poor. The

study showed that most of the backside radiation is not directed to the front side

but couples into a parallel plate mode between the slot plane and the reflector.

However, by using two parallel slots at a specific distance the parallel plate mode

excitation could be reduced and the front radiation could be increased up to 90%.

For a reflector, which is not infinitely extended the reflector efficiency increases.

In an own study [34] done on the reflector (finite size) backed folded slot antenna

reflector efficiencies of 80% were achieved.

The reflector backing may be a valid solution for specific cases but it is not ap¬

propriate as a general technique to achieve single-sided radiation because depending

on the shape and the arrangement of the slots parallel plate modes are excited. Es¬

pecially in array application this can lead to unwanted coupling and thus severe

degradation of the array performance. Therefore, cavity backing is preferred be¬

cause it encloses completely the backside fields, which limits the backside radiation

and prevents coupling between different antennas.

2.4.2 Electromagnetic Band-Gap Structures Reflector

Considerable work on electromagnetic band-gap structures (EBG) has been pub¬

lished over the last five years [35]. EBG's exhibit interesting properties. For example

they allow to suppress parallel plate modes at specific frequencies [36]. When illumi¬

nated by a plane wave with normal incidence the reflection factor is close to 1, which

is contrary to a perfect electric conductor (PEC) which yields a reflection factor of

-1. When used as a back-reflector for a slot antenna the distance between the EBG

and the reflector can be fairly small compared to a PEC-reflector for which the ideal

distance is close to a quater wavelength. This allows the design of single-sided radi¬

ating slot antennas with low profile [37]. EGB's are periodic structures built from a

basic cell. A minimum number of cells is required (usually about 10 x 10) in order

to produce the above mentioned EBG characteristics. Different EBG-geometriesexist. However, most of them are electrically large, which limits their usefulness

as a reflector for a single slot antenna because the size of the reflector has to be

one to several wavelengths large. For array applications, however, where the an¬

tenna size itself is large as well, they become interesting. Recently more compact

structures called reactive impedance surfaces (RIS) have been published [38]. They

require high permittivity substrates and complicated production processes and are

therefore not well suited for mass fabrication.

For these reasons EBG's are not investigated in this thesis.


2.4.3 Cavity Backing

Cavity backing consists of shielding the backside of the slot antenna with a metal

cavity. The walls of the cavity are connected to the groundplane of the slot, which

allows to suppresses any radiation to the backside or any excitation of parallel

plate modes. Therefore they are better suited as a general technique to achieve

unidirectional radiation with slot antennas, than a simple reflector.

Theoretical [39, 40] and experimental [41] work has been published on cavity

backed rectangular slot antennas and on cavity backed annular slot antennas [42,

43, 44]. In this thesis the influence of cavity backing on annular slot antennas

will be investigated in terms of radiation pattern, impedance bandwidth, circular

polarization bandwidth, gain and cavity size.

2.4.4 Slot Coupled Patch Antenna

Instead of returning the backside radiation with a reflector the radiation can be

attracted to the front side of the antenna by a structure located on the front side.

By using a circular or annular patch on the front side of an annular slot, the slot

fields couple into a resonating patch antenna mode. There have been investigations

on different cpw-fed slot coupled patch antennas [45, 46]. In this thesis annular slot

coupled circular and annular patch antennas will be presented. The particularity

of the presented structure is that the polarization of the antenna is not determined

by the shape of the patch but by the shape of the coupling slot, which distinguishes

this architecture from previously published ones. This property of the antenna

is important for reconfigurable active antenna designs as they will be presented

in this thesis. In this solution the electrical slot shape is changed by the use of

switching diodes in order to switch between different polarizations. In contrary to

other polarization switchable patch designs [47] our approach can be realized with

uniplanar technology. All active elements are on one single layer and no dielectric is

needed between the patch and the feed layer. Compared to the cavity backing this

approach shows the disadvantage, that there can be considerable radiation to the

back side, due to the coupling slot. A study of the backside radiation as a function

of the coupling slot size will be presented.

2.4.5 Dielectric Lens

A further possibility to direct the radiation of a slot antenna to one side is the use

of a high permittivity substrate. If the medium of one side of the slot is filled with

a dielectric of relative permittivity er (dielectric halfspace), the percentage of the

power radiated to the dielectric side is significantly higher. Typically for er — 10

the power radiated to the dielectric side is about 80% of the total radiated power.

In reality a dielectric lens with a radius of about 20 wavelengths is used providing

a single sided radiation and high gain [48], Due to the large electrical size of the

lens this technique is primarily interesting for millimeter-wave applications where

the wavelength is short. The uniplanar circuit is directly realized on the metalized


backside of the lens. This avoids the excitation of surface waves as it would be the

case for a regular planar dielectric circuit.

Chapter 3

Annular Slot Antenna

3.1 Introduction

This chapter contains different aspects of annular slot ring antennas. First an

introduction to the electromagnetic modes of annular slot ring antennas will be

given. The calculation of the radiation pattern will be explained. Different feeding

methods will be presented. Then different techniques to achieve circularly polarized

radiation are introduced. A rigorous study and comparison of these techniques,

published by the author [49], will be presented.

3.2 Antenna Fields and Radiation Pattern

3.2.1 Aperture Fields

For the analysis of the electromagnetic fields of the annular slot antenna (ASA) the

geometry shown in figure 3.1 is considered. It consists of an annular slot with an

Figure 3.1. Geometry of ASA lying in a infinite groundplane.

inner radius r% and an outer radius r0 lying in an infinitely large perfectly conducting

18 Annular Slot Antenna

groundplane. The average radius rav of the ASA is defined as

rav = ^p (3.1)

The antenna is assumed to be fed by a current or voltage source, which does not

disturb the antenna mode.

The resonance frequency can be determined approximately with a transmission

line model [23] with an accuracy of about 10% - 15%. The resonance occurs for an

average circumference of one guided wavelength:

27rra„ = Xg (3.2)

The fields in the slot must satisfy the Helmholtz equation. For cylindrical coordi¬

nates this leads to a time harmonic variation in the (^-direction shown in Figure 3.2

whereas the variation in radial direction is described by a sum of Hankel-functions

[24]. However for a narrow slot r0 — ri « A the radial dépendance can be estimated

as shown in equation 3.3 [23].

1

EJr) = - for Ti < r < r0r

Er(r) — 0 otherwise (3.3)

£0(r) = 0

For the angular dépendance at resonance a eJ'n^ variation is assumed, where n is the

mode number. The annular slot exhibits two degenerated modes for each resonance.

The fields of these modes are orthogonal in space which corresponds to a rotation

in the groundplane of <f> — ~. Figure 3.2 shows the electrical field distribution of

the first three modes of the ASA.

first resonance second resonance third resonance

Figure 3.2. E-field distribution of first three resonance modes of ASA.

3.2 Antenna Fields and Radiation Pattern 19

3.2.2 Radiation Pattern

Following the procedure given in [23] the far field can be calculated. Using the

standard spherical coordinates r, 9, and (ft the far field equations are

Ee(r,e,ci>) = ~ko J—-~ E0(k0sm9)) (3.4)r 2 l J

p-jk0r jn+lejn<j> _

__

,

Et(r,6,<t>) = +hr

J

2 [Ee(k0 sin 0)j (3.5)

where k0 = lj^/JIoEq and the linear combinations of the Hankel-transformed esti¬

mates are used

É0(k0sin9) = Ë(+)k0 sin 9 — i?(_)/c0 sin 0 (3.6)

Êe(k0sin9) = Ë(+)ko sin 9 + Ë(-)k0 sin 9 (3.7)

The (n ± l)th-order Hankel transforms are defined by

r0

%>(<*)= [jn±i(w)dr (3.8)

rt

where Jn(ctr) is the nth-order Bessel function of the first kind, a is the Hankel-

transform variable, and r; and r0 are the inner and outer ring radii, respectively, n

is the order of resonance and u — ujq is the resonance frequency.In order to derive the above presented far field formulas some assumptions on

the slot fields had to be made (equation 3.3), which are not necessarily fulfilled.

Therefore, nowadays it has become more convenient to use numerical field simu¬

lators to calculate antenna performance. They are fast, accurate and can handle

different kind of geometries without making any assumptions on the field distribu¬

tion. Figure 3.3 shows a typical radiation pattern of an ASA calculated with HFSS.

3.2.3 Radiation Resistance

Assuming a certain field distribution as described by equation 3.3 the input impe¬

dance can be calculated. At resonance the reactive part of the input impedance is

zero and the resistive part is equal to the radiation resistance because the feed is

located at field maximum. Therefore the input impedance can be expressed as

U2Zin = ~r— (3-9)

* radiated

,where

U

r0

= [ Er(r)dr = In- (3.10)J ri

rt


E-plane main polarization

H-plane main polarization

E-plane cross polarization

180°

Figure 3.3. Far field of linearly polarized ASA in the first resonance.

and

radiated —

j ivWp±EZdsJJ sphare rj (3.11)

where n corresponds to the free space wave impedance and the factor 2 arises

from the fact that the field equations 3.4 and 3.5 comprise the far field of the two

degenerate modes, but with one feed only one mode is excited. Therefore only half

the calculated power has to be taken into account. A more detailed calculation of

the input impedance of one and multiport-ASA's can be found in [24].

3.3 Numerical Simulation

The formulas presented in the previous section are a good approximation of the

annular slot antenna behaviour. However, they can only be used for generic an¬

tenna geometries. For more complex geometries including irregular shapes, finite

grounplanes, eg., it is very cumbersome or often impossible to derive analytical

solutions.

Modern numerical full wave field solvers have evolved to a point where they yield

results with very good accuracy and there are nearly no restrictions on the struc¬

ture under investigation except the amount of its computational effort. Therefore

commercial numerical field solvers were used to calculate the antennas investigated

in this thesis. The results showed good agreements with measurements for both,

the impedance and the radiation of the different antennas. In particular effects of

mode-splitting perturbations were modelled accurately.

For this thesis mainly three different numerical codes werde used: ENSEMBLE

from Ansoft, HFSS from Agilent and HFSS from Ansoft. HFSS (from Ansoft and

Agilent) is a three-dimensional full wave solver and is based on the Finite Element


Method (FEM). ENSEMBLE is a 2.5-D solver allowing to calculate layered struc¬

tures including vertical vias of cavities with vertical walls. ENSEMBLE can not

handle finite groundplanes or finite size dielectric substrates. In the numerical sim¬

ulation slot antennas having a finite groundplane and substrate size can be approx¬

imated with a simulation model using an infinite groundplane and substrate size.

This approximation is valid if the groundplane dimensions are about the double (or

higher) of the slot resonator dimensions. Hereby the general antenna behaviour is

well modelled but certain discrepancies appear. Typically, the resonance frequency

is shifted by about l%-5%. Furthermore differences in the radiation pattern can be

seen. The simulated radiation pattern assuming an infinite groundplane is broader

than the measured one. For cavity backed slot antennas the simulation with EN¬

SEMBLE shows no backside radiation at all whereas in reality backside radiation

occurs because of the diffraction at the groundplane edges. During this thesis var¬

ious antennas have been simulated and measured. In the beginning ENSEMBLE

was used for the simulation. In the course of the thesis however HFSS 8.5 from

Ansoft was preferred, because it did not exhibit the drawbacks of ENSEMBLE.

Excellent agreement was found between HFSS simulations and measurements.

3.4 Feeding Architectures

Figure 3.4 shows different feeding solutions proposed for the annular slot [50]. On

the right side a microstrip feed located on the opposite side of the groundplaneis shown. The fields of the microstrip mode couple into the annular slot. The

length of the microstrip line stub can be chosen for matching. The microstrip feed

is often used but is not compatible with uniplanar technology. On the left and on

CPWfeed m^mmmm Hm^^^^^^V"".^H|^B microstrip feed

slotline feed

Figure 3.4. Different ASA-feed solutions proposed in [50]

the bottom a cpw-feed and a slotline feed are shown, respectively. The cpw-feed


is capacitively coupled to the slot ring whereas the slotline is inductively coupled.

These two feeding methods are not well suited for an antenna feed but rather for

coupling into a high-Q resonator because the coupling is very weak. Therefore

two alternative feeds were utilized. Figure 3.5 a) shows a suitable slotline feed

for the ASA and figure 3.5 b) shows a suitable cpw-feed for the ASA with the

corresponding resonance modes. Note that for the latter the slot ring is shorted at

the feed point, which is necessary in order not to short the even cpw-mode. The

shorting of the slot ring leads to a different resonance mode with a circumference of

about §AS as shown in Figure 3.5. Furthermore, no degenerate mode exists because

the rotational symmetry has been destroyed. This makes it impossible to produce

circular polarization with this kind of antenna. For a given resonance frequency the

cpw-fed antenna yields a larger size due to larger circumference.

In the case of the slotline feed the polarization of the antenna is perpendicular

to the feed line whereas in case of the cpw-feed the polarization is parallel to the

feed.

a) slotline feed b) cpw feed

Figure 3.5. Uniplanar feeds for ASA

3.4.1 Slotline Transitions

The slotline is the appropriate feed line for uniplanar circularly polarized ASA's.

In contrast to the cpw-feed it allows the existence of two degenerate modes in the

slot ring resonator, which are necessary to produce circular polarization. Most of

the antennas investigated for this thesis are fed by a slotline.

On the other hand as discussed in section 2.2.1 the slotline is not well suited

to guide an electromagnetic signal because it is dispersive, tends to radiate and

does not confine the field very well. Moreover it is a balanced line and therefore

inappropriate for active devices which need a defined ground. Therefore, to achieve


a transition from an unbalanced line to a slotline a balun is employed and only

a short slotline section is used to feed the ASA. Three different balun transitions

were used: the cpw-to-slotline transition, the microstrip-to-slotline transition and

the coaxial line-to-slotline transition. Figure 3.6 shows the setup of the different

transitions. Various cpw-slotline transitions have been proposed [51, 52], Figure

a) cpw-to-slotline b) coaxial line-to-slotline c) microstrip-to-slotline

Figure 3.6. Slotline transitions to cpw, coaxial line and microstrip line.

3.6 a) shows one of the used cpw-to-slotline transitions. The left slot of the cpw

continues straight into the slotline whereas the right slot is bent into an extra loop

introducing a 180 phase shift to the slot field in order to match the field distribution

to the slotline mode [53]. An airbridge is needed to short the unwanted odd mode

of the cpw. Figure 3.6 b) shows the coaxial-line-to-slotline transition. This works

similar to the cpw-to-slotline transition. The coaxial line signal is coupled into

a even cpw-mode by connecting ground conductor to the ground of the slotline

and the center conductor to the opposite side of the slotline. The two ends of the

slotline form a cpw whose even mode is excited. Using the principle of the extra

phase shifting loop for one slot of the cpw, the transition to a slotline can be made

as previously explained. This transition does not need an airbridge, because the

unwanted odd mode is shorted through the groundplane at the location where the

coaxial line is connected to the slotline. Figure 3.6 c) shows a microstrip slotline

transition introduced in [54]. Note, that the microstrip line (depicted with the

dashed line) is located on the opposite side of the substrate as the slotline or in

other words the slotline is etched into the groundplane of the microstrip line. With

the length of the slotline stub as well as the microstrip stub two degrees of freedom

are available for matching. In comparison to the cpw-to-slotline transition transition

the microstrip-to-slotline transition shows several advantages: It exhibits a smaller

size, allows to match a larger impedance range and it is easier to fabricate and to

handle during experimentation, because it does not require air bridges, which are

usually realized with thin bondwires. However, this transition is not uniplanar. It

can be used in quasi-uniplanar circuits for testing of uniplanar elements, where it

is finally replaced by a cpw-to-slotline transition.


3.5 Excitation of Circular Polarization

3.5.1 Introduction

Circular polarization results when two orthogonal linearly polarized modes with

equal amplitude and 90 degrees phase shift are excited. The ASA supports two or¬

thogonal linear polarized modes and is therefore well suited as a circularly polarized

radiating element. The purity of circular polarization is expressed with the axial

ratio, which is the ratio of the magnitude of the two excited modes. The circular

polarization is ideal for an axial ratio of 1 (OdB) and deteriorates for higher val¬

ues. The axial ratio changes for different radiation directions as well as for different

frequencies. Usually the antenna is designed for having an ideal axial ratio in the

direction of main radiation. In the case of the ASA this is the front side direction

(also called broadside direction) or back side direction. Both are perpendicular to

the substrate plane. The requirements for the axial ratio depend on the application

but generally for circularly polarized applications an axial ratio lower than 3 dB

is required. The axial ratio bandwidth (ARB) is defined as the frequency band in

which the axial ratio is lower or equal to the required axial ratio. It can be expressed

in Hertz or in %.

Two basic techniques exist to excite the two modes with the proper amplitude

and phase. The first technique consists in using a dual feed in combination with

a phase shifting network. The signal fed to the antenna is split into two equal

parts having 90 degrees phase shift between each other, which are then fed at two

different points to the ASA in order to excite the two linear orthogonal modes in¬

dependently. The second technique consists in using one single feed in combination

with a perturbation in the ASA provoking mode splitting. The perturbation is

placed asymmetrically with respect to the feed and it's shape is optimized in order

to achieve a good axial ratio. The first technique using an external power split¬

ter, phase shifter and two separate feeds exhibits a larger axial ratio bandwidth

(ARB) but shows a smaller impedance bandwidth. The required phase shifting and

power splitting network, usually a branch line coupler, is cumbersome and increases

the antenna size considerably. Furthermore this technique is difficult to realize in

uniplanar technology because it would require a coplanar branch line coupler with

various air bridges. An alternative and more compact solution would be to use a

T-junction splitting the feed signal into two parts followed by two feed lines with a

^-length difference. However, this solution exhibits a smaller axial ratio bandwidth.

Little work has been published on circularly polarized ASA's. Qing and Chia

presented a dual microstrip-fed circularly polarized slot-ring antenna using a branch

line coupler [55]. An 3dB-ARB of 22% was achieved. In terms of free space wave¬

length (A0) the size of the antenna together with the feed network was approximately

0.3A0 x 1A0. The same authors presented a single microstrip-fed rectangular slot an¬

tenna exhibiting an ARB of 11% [56]. The feed consists of a microstrip line whose

groundplane comprises the annular slot. The feeding microstrip line crosses the

radiating slot at two different locations in order to feed it at two different points.


The most interesting papers treating uniplanar circularly polarized slot anten¬

nas are introduced subsequently. A CPW-fed circularly polarized rectangular slot

operating at 25 GHz achieving an ARB of 9.9% was presented by Soliman et al.

[27]. The feed consists of a cpw-to-slotline T-junction splitting the incoming signal

from the cpw into two equal slotline signals. The phase difference of 90 degreesbetween the two signals is achieved by a length difference of ^. The overall size of

the antenna including the feed network was about 0.31 Ao x .47A0. Matsuzawa and

Ito presented a CPW-fed circularly polarized antenna consisting of two circularly

polarized radiating elements. The achieved ARB was only 3% and the overall size

of the antenna was about 0.4A0 x .52A0 [57]. In contrast to the previous designs they

used an asymmetry in the radiating slot structure to achieve circular polarization.

In this section the principle of single-fed circularly polarized ASA's will be pre¬

sented. A comparison between the impedance behaviour of linearly polarized ASA's

without perturbations and circularly polarized ASA's with one or two perturbations

will be shown. The radiation characteristics of the circularly polarized ASA will be

analyzed.

3.5.2 Antenna Topology

In order to illustrate and to compare the characteristics of linearly and circularly

polarized ASA's three different examples, shown in figure 3.7, are investigated.

They are built on a 0.635 mm-thick Duroid6010 substrate having a permittivity of

er = 10.2. The inner slot ring radius is 13 mm and the outer radius 15 mm. The

a) no perturbation b) single perturbation c) two perturbations

linear polarization circular polarization circular polarization

Figure 3.7. Linear and circularly polarized microstrip-fed ASA.

microstrip feed line is located on the opposite side of the slot structure and has a

characteristic impedance of 50 il. The ASA's are fed with a short slotline through

a microstrip-to-slotline transition introduced in 3.6 c). Antenna a) without any

perturbation is linearly polarized whereas antennas b) and c) having one and two

perturbations, respectively, are circularly polarized. The perturbation consists of


a square slot widening to the outer side of the slot ring. Other types of pertur¬

bations can be used and will be investigated in the following section. If only one

perturbation is used the ideal position shows either an angle of 45 degrees or -135

degrees with respect to the feed. Although one perturbation is enough to excite

circular polarization it turned out that two perturbations located at 45 degrees and

-135 degrees, respectively, are more advantageous. They yield a larger ARB and a

more symmetric radiation pattern. Note that if a single perturbation is used it's

size is larger than if two perturbations are used. Simulations showed that the ARB

of an antenna using two perturbations is about 5% to 25% higher than the ARB

of an antenna employing only one perturbation. Furthermore the main beam was

tilted off broadside for an antenna employing only one perturbation. Therefore, all

realized antenna designs comprise two perturbations.

3.5.3 Input Impedance

The classical ASA without perturbation exhibits one single resonance which trans¬

lates into one resonant circle in the smith chart or one minimum of the sn-reflection

parameter. The introduction of one or two perturbations causes the additional ex¬

citation of the second degenerate mode. In fact the perturbation splits the energy

into the two modes. Due to the created asymmetry the resonance frequency of one

mode is slightly increasing whereas the resonance frequency of the other mode is

slightly decreasing [58]. Therefore two different resonance circles in the smith chart

and consequently two minima of the sn-reflection parameter can be observed. At

resonance the phase of the electromagnetic field of the excited mode undergoes a

zero crossing. At frequencies between the two resonance frequencies the phase of

the lower resonance mode and the higher resonance mode are clearly distinct. At

a specific frequency between the two resonance frequencies a phase difference of

exactly 90 degrees is achieved and therefore the phase condition for ideal circular

polarization is fulfilled. An equal amplitude of the two modes at that frequency can

be obtained by tuning of the perturbations. Figure 3.8 shows the input impedance

as well as the return loss (normalized to 50 Q) of the investigated antennas. The

linearly polarized antenna (Figure 3.7 a)) shows one single minimum in the return

loss which translates into one resonant circle in the smith chart. Adding one or two

perturbations (Figure 3.7 b) and c)) leads to a second resonance and therefore to an

additional minimum in the return loss, which translates also to a second impedance

circle in the smith chart. The 10 dB-impedance bandwidth of the three antennas are

6%, 14% and 15%, respectively. It can therefore be concluded, that the impedance

bandwidth of the single fed circularly polarized ASA doubles the bandwidth of the

linearly polarized antenna.

The lower graph of Figure 3.8 shows the real and imaginary part of the input

impedance of the three investigated antennas (including the microstrip feed line).At resonance the input impedance of a classical resonator shows a maximum for

the real part and a zero crossing for the imaginary part, which corresponds to

one circle in the smith chart. The input impedance of the three antennas from

Figure 3.7, however, is not directly determined at the feed point of the resonator


but includes the microstrip-to-slotline transition and the microstrip feed. This

results in a transformation of the resonator input impedance along the microstrip

line. Observing the input impedance including the microstrip feed, the number of

resonances can be identified by the number of slope reversals (not zero crossings)of the curve showing the imaginary part of the impedance or by the number of

maxima of the real part of the impedance. The real part of the input impedance of

the linearly polarized antenna a) shows one maximum and the corresponding slope

reversal of the imaginary part with a zero crossing. It can be clearly seen, that

only one resonance is excited. The real parts of the input impedances of antennas

b) and c) show two maxima and the corresponding imaginary parts show two slope

reversals, which indicates that two resonances are excited. Note that there is no

fundamental difference in the impedance behaviour of antenna b) and c), because

for both antennas the same kind of resonances are excited.

On the right side of Figure 3.8 the impedance of the three antennas is depicted

in the smith chart. The linearly polarized Antenna a) exhibits one resonance, which

2 2.1 2.2 2.3 2-4 2.5 2.6

Frequency [GHz]

Figure 3.8. Input impedance of the investigated ASA's.


corresponds to one circle in the smith chart whereas the two circularly polarized

antennas b) and c) exhibit two resonances, which corresponds to two circles in the

smith chart.

3.5.4 Radiation Characteristics of Circularly Polarized

ASA's

Figure 3.9 shows the simulated radiation pattern (ENSEMBLE), the gain and the

axial ratio of antennas b) and c) of figure 3.7. Both antennas radiate with right

handed circular polarization (RHCP) to the front side and left handed circular

polarization (LHCP) to the back side. If the location of the perturbation(s) would be

rotated 90 degrees around the center of the slot ring in clockwise or counterclockwise

direction with respect to the current position, the rotation sense of the polarization

would be inverse. The opposite polarization sense between front side radiation and

Antenna b) RHCP Antenna b) gain

— Antenna b) LHCP Antenna c) gain

-— Antenna c) RHCP Antenna c) axial ratio

Antenna c) LHCP Antenna b) axial ratio

Figure 3.9. Far field characteristics of circularly polarized ASA's.

back side radiation is due to the fact, that the rotation sense of the radiating slot

fields is the same, but the direction of propagation (front side and back side) and

therefore also the sense of circular polarization are opposite. In the case of linear

polarization however the polarization of the fields radiated to the front side and the

back side are the same.

The left graph of figure 3.9 shows the axial ratio of the two circularly polar¬

ized antennas. The minimum does not occur at the same frequency but differs byabout 40 MHz (corresponding to about 1.8% difference). The 3dB-ARB of an-


tenna c)(Figure 3.7), having two perturbations, is 3.2% and the ARB of antenna

b) is 3.05%, which is about 5% less. Further investigations showed, that using two

perturbations instead of one leads to higher ARB's (5% to 25% higher). The 3 dB

ARB is typically about 5 times smaller than the 10 dB-impedance bandwidth. The

broadside gain of antennas b) and c) is about 2.5 dBi, which is typical for electrically

small antennas radiating on two sides.

3.5.5 Summary

In order to illustrate the characteristics of single fed circularly polarized ASA's an

investigation on two circularly polarized ASA's using one and two perturbations,

respectively, and a linearly polarized ASA serving as a reference structure has been

made. All antennas have the same slot radius and slot width. The perturbations

are designed to produce circular polarization in broadside direction.

The 10 dB-impedance bandwidth of the circularly polarized antennas b) and

c) is more than the double of the impedance bandwidth of the linearly polarized

antenna a). This is due to the excitation of the second degenerate mode, which has

a different resonance frequency and therefore enlarges the impedance bandwidth.

Antennas b) and c) show a similar impedance behaviour.

The 3 dB-ARB is about 5 times smaller than the 10 dB-impedance bandwidth.

The 3 dB-ARB of antenna c) is about 5% larger than the one of antenna b). In

general it can be seen, that the ARB of an circularly polarized ASA using two

perturbations is up to 25% larger than an ASA using only one perturbation. The

radiation patterns as well as the gain of antennas b) and c) are alike. The main

beam of antenna b) is slightly tilted off the broadside direction. For ASA's with

a larger slot width and only one perturbation the main beam can be significantly

tilted, which is usually unwanted.

The advantage of the perturbation technique over the dual feed technique are

the easier feeding, the smaller size and the broader impedance bandwidth. However,

the disadvantage of this solution is its inherent narrower ARB. This is due to the

rapid phase change of the two exited modes around the operating frequency which

allows to fulfill the relative phase criterion only over a relatively narrow bandwidth

around the operating frequency. The lower the quality factor of the two resonances

the slower is the phase change around the resonance frequency resulting in a broader

ARB.

3.6 Parameter Study

3.6.1 Objectives

The characteristics of a circularly polarized ASA, such as antenna size, ARB, in¬

put impedance and radiation pattern, are influenced by the antenna parameters

such as slot width, type of perturbation, permittivity and the thickness of the used

dielectric. In order to investigate the influence of the different parameters a numer-


ical simulation was performed. Five different antenna types (type A - E) shown in

Figure 3.10 were compared. Antennna A is a linearly polarized antenna without

perturbations, which serves as a reference structure. Antennas B-E exhibit two per¬

turbations and produce circular polarization. For each circularly polarized antenna

type (B-E) a different perturbation type is used.

Figure 3.10. Geometry of investigated ASA's.

The first and second perturbation type (Fig. 3.10 type B and C) consist of

wider slotline sections. In one case widened towards the outer side of the slot ring

(type B) and in the other case widened towards the inner side of the slot ring (type

C). The perturbation of antenna D consists of a narrowed ring section and the

perturbation of type E consists of a narrowed slot ring section protruding inside

the ring. The antennas are excited with a short slotline section as shown in figure

3.10. The simulations are performed with ENSEMBLE from Ansoft, assuming an

infinite groundplane.

Every antenna type was simulated with slot widths ranging from 0.5 mm to 5 mm

and for two different substrates with a thickness of 0.635 mm and a permittivity of

£V = 2.2 (referred as diell) and eT — 10.2 (referred as diel2), respectively. The slot

ring diameter was designed in order to achieve resonance around 2.5 GHz. For each

antenna type with a given slot width and substrate the perturbation was adjusted

(dimensions indicated by arrows in figure 3.10) in order to achieve a maximum

ARB.

3.6.2 Antenna Size

The electrical size of the different investigated antennas is defined as the largest

dimension of the slot ring structure normalized to the free space wavelength, which

does not include the groundplane size, and will be referred to as antenna size. The

antenna is normalized to the free space wavelength (A0) at the operating frequency.

The groundplane size is not considered because it has only a minor influence on

the resonance frequency. For the simulation it was assumed infinitely large. In

reality groundplane sizes of about 1 Ao x 1 Ao or larger are used. Compared to the

infinite groundplane the use of a finite groundplane size produces a narrower far

field beam and a frequency shift of the axial ratio minimum (typically l%-3%),


but does not alter the 10 dB-impedance bandwidth or the 3 dB-ARB. Therefore an

infinite groundplane can be used for the relative comparison of different antenna

types in terms of impedance bandwidth, ARB and antenna size.

Figure 3.11 shows the antenna size of the investigated antennas as function of

the slot width. The linearly polarized ASA resonates at a frequency where the

mean circumference of the slot ring is equal to one guided wavelength. The guided

wavelength of the slotline is a function of the slotline width, the permittivity of the

substrate and the ratio of the substrate thickness to the slot width. The larger the

slot line width, the longer is the guided wavelength. The higher the permittivity

and/or the thicker the substrate the shorter is the guided wavelength, respectively

[10].

-*- antenna. A diell

-- antenna A diel2

-A- antenna B diell

-*- antenna B diel2

-&~ antenna C dtell

-#- antenna D diell

-A- antenna D diel2

-Ä- antenna E diell

— antenna S diel2

Slot Width [mm]

Figure 3.11. Comparison of the electrical antenna size versus the slot ring width of the

investigated antennas.

In a first step the influence of the slot ring width and the substrate permittivity

on the size of the linearly polarized antennas (type A) will be discussed. Then the

results of the circularly polarized antennas (type B-D) will be compared relative to

the linearly polarized ones.

Linearly Polarized Antennas (type A)Observing the antenna size of the linearly polarized ASA's (type A) it can be seen,

that for a given substrate the antenna size increases with an increasing slotline

width of the slot ring due to the increasing guided wavelength. The influence of

the permittivity can be seen comparing the size of two antennas having the same

slot width but a different substrate permittivity. For a slot width of 0.5 mm the

antenna size using diel2 (er = 10.2) is 33% smaller than using diell [er = 2.2).This means that the antenna built on the high permittivity substrate is about 66%


of the size of the same antenna built on the lower permittivity substrate. For a slot

width of 5 mm the size of the antenna using diel2 is 20% smaller than the size of

the antenna built on diell. Hence, keeping the same substrate thickness the size

difference becomes smaller with an increasing slot width. This is due to the fact

that with an increasing slot width the effective permittivity, which depends on the

substrate permittivity and the ratio of the slot width and the substrate thickness

[10], decreases faster for diel2 than for diell.

A narrow slot width and/or a high substrate permittivity reduce the antenna

size. On the other hand they decrease the radiation surface of the antenna, which

increases the antenna quality factor. As explained in section 2.3 this results in

smaller impedance bandwidths and higher losses. Hence an antenna size reduction

reduces the impedance bandwidth and increases the losses.

Circularly Polarized Antennas(type B-E)The introduction of perturbations causes the excitation of two modes. The antenna

size is defined relative to the free space wavelength A0 of the operating frequency,

which is the frequency where the minimum axial ratio occurs. In the circularly

polarized case the antenna size not only depends on the slot width, the substrate

permittivity and the substrate thickness but also on the chosen perturbation type.

Comparing the different perturbation types relative to the linearly polarized

antenna for a given substrate it can be observed that type B (widening to the

outside) leads to a larger antenna size whereas perturbation types C,D and E lead

to a smaller antenna size than in the linearly polarized case. Figure 3.12 shows the

current distribution of antennas B and C, respectively. The currents are indicated

by arrows where the size of the arrow is proportional to the magnitude of the

current. In case of antenna B the length of the outside current path is prolongated

due to the perturbation, which causes larger antenna size. In contrary, in case of

antenna C the inside current path is prolongated causing a smaller antenna size.

antenna B antenna C

Figure 3.12. Current distribution of antenna B and antenna C.


In case of antenna D the narrowing of the two slot line sections leads to a

shorter guided wavelength in these sections and consequently reduces the total size

of the antenna. The smallest antenna size is achieved with antenna E where not

only a narrowed section is used but this section is protruding inside the ring, thus

increasing the electrical length of the structure.

There exist various types of perturbations allowing to excite circular polariza¬

tion. The comparison of the different antenna types (A-E) shows that the antenna

size depends significantly on the chosen perturbation. Compared to the linearly po¬

larized antenna A, antenna B shows a larger size and antennas C-E show a smaller

size. The drawback of a smaller antenna size is the reduced ARB as will be shown

in the next section.

3.6.3 Axial Ratio Bandwidth (ARB)

The axial ratio is a measure for the purity of a circular polarization. Usually an axial

ratio below 3 dB is required, which corresponds to a cross polarization level being

lower than -15 dB. The frequency bandwidth for which the axial ratio ranges below

a specified value is called axial ratio bandwidth (ARB). It is a crucial parameter,

because determines the useable frequency range of a circularly polarized antenna

and should ideally be as large as possible.In the case of single fed circularly polarized antennas two orthogonal modes

with a slightly different resonance frequency are excited as shown in section 3.5.

The circular polarization occurs at a frequency between the two resonances where

they show equal amplitude and 90 degrees phase shift. This phase and amplitude

condition is ideal for one frequency point, but deteriorates the farther the operating

frequency is from this ideal frequency. The rapidness with which the axial ratio

degrades with a frequency change depends on the quality factor of the two excited

resonances. The higher the quality factor, the faster is the phase change (with

frequency) of each resonance and therefore the smaller is the ARB. In the case of

antennas the quality factor of the radiating resonance modes is increasing if the

antenna size decreases as explained in section 2.3. This is due to the reduction of

the radiating surface. Therefore it can be stated that for a given antenna type the

ARB is increasing with the antenna size. Different perturbation types, however,

might exhibit different ARB's for the same antenna size.

Figure 3.13 shows the ARB of the different investigated circularly polarized

ASA's in function of the antenna size. On the top graph the results for antennas

B,C,D and E realized on diell (er = 2.2) are shown. The bottom graph shows the

results of antennas B,D and E realized on diel2 (er — 10.2).Different interesting findings can be taken out of these results. It can be seen,

that for each antenna type the ARB increases with the antenna size. The compar¬

ison of the different antenna types shows that antenna B yields the largest ARB

going up to 11%. This is due to the fact that the size of antenna B is significantly

larger than the size of antennas using other perturbation types as it was explainedin section 3.6.2. For the antenna types D and E it is difficult to increase the ARB


-a

3"B

«

a

öS

a

"fi

12

11

10—

« 9--

8

4 —

7

5

—

1- 1

1

1

+

1

^_ antenna B diell

-B- antenna C diell

-*- antenna D diell

-*- antenna E diell

0.25 0.3 0.35 0.4 0.45 0.5

Antenna Size [X]

0.55 0.6

tit 0—

js 5--

3a

ca

a

a

'8

0.25 0.3 0.35

Antenna Size [X]

Figure 3.13. Comparison of the 3dB-axial ratio bandwidth as function of the antenna

size.

above 8% by further widening the slot size, because the shape of the perturbations

would become unrealizable (e. g. the width of the narrowed section of antenna D

would be in the micrometer range). Antenna C exhibits the smallest ARB (5.8%).The concentration of the currents in the center of the antenna, as it is shown in

figure 3.12, limits the achievable ARB.


A next interesting point is to compare ARB relative to the antenna size. It can

be seen that for a given ARB below 8% antenna E shows the smallest size whereas

antenna B exhibits the largest size.

The influence of the substrate permittivity can be seen by comparing the two

graphs of figure 3.13. Comparing two antennas of the same type and having the same

slot width but being realized on the different substrates (diell and diel2) shows that

in case of diel2 not only the size of the antenna is reduced (compared to diell) but

also the ARB. The size reduction leads to a reduction of the radiating surface and

therefore to an increased antenna quality factor, which reduces the ARB. Antenna

B realized on diell and having a slot width of 1 mm exhibits a similar size and ARB

as if it is realized on diel2 with a slot width of 5 mm. The conclusion of this result is

that for a given antenna type the ARB is proportional to the antenna size. Latter

can be influenced either by the substrate thickness and permittivity or by the slot

width. This is also confirmed by the results for antennas D and E.

3.6.4 Input Impedance

Figure 3.14 shows the real part of the input impedance, also called input resistance,

calculated at the beginning of the short slotline feed (shown in figure 3.10). The

values range from 10 Ohm up to 300 Ohm. On the top graph it is represented for all

antennas as function of the slot width. On the bottom graph the input resistance

of the circularly polarized antennas (type B-E) is represented as a function of the

ARB.

The input resistance is proportional to the radiation resistance. For a given

antenna type (A-E) the radiation resistance is proportional to the antenna size. In

figure 3.14 a) it can be seen that the input resistance increases with the slot width

for all antenna types, because with an increasing slot width the radiating surface

and therefore the radiation resistance are increasing. Furthermore it can be seen,

that for a given permittivity and slot width the real part of the input impedance

is higher for the circularly polarized antennas (type B-E) than for the linearly

polarized antenna A. This is due to the fact that in the circularly polarized case two

radiating modes exist, which reflects in a higher radiation resistance. Comparing

the real part of the input impedance with respect to the substrate permittivity

shows, that for an antenna of given type and slot width a higher permittivity yields

a lower input impedance. This is mainly due to the fact that a higher permittivity

causes a smaller antenna size and therefore a smaller radiation resistance. For the

high permittivity dielectric (diel2) it is lower than for the low permittivity dielectric

(diell).The representation of input resistance as a function of the ARB (figure 3.14

b)) gives further interesting insights. It can be seen that for a given dielectric and

ARB the real part of the input impedance is very similar for antennas B, D and E.

Antenna C however behaves differently due to the modified current paths on the

inside of the ASA as shown in figure 3.12. Furthermore it can be seen that for a

given ARB the antennas realized on the high permittivity substrate (diel2) exhibit


a)350

cf 300

& 3

Slot Width [mm[

-*- antenna A diell

-*- antenna A diel2

-*- antenna B diell

-m- antenna B dielB

-9- antenna C diell

*- antenna D diell

-*- antenna D diel2

-W- antenna E diell

— antenna E diel2

aa.e

b)

35(h

2. 300

250

| 200

150

—D— antenna S diell

- anienna S dielÊ

-X-- antenna C diell

- antenna D diell

- antenna D diel2

- antenna E diell

-X-- antenna E diel2

10 12

Axial Ratio Bandwidth [%]

Figure 3.14. Real part of input impedance versus slot width (top) and versus axial ratio

bandwidth (bottom).

a higher real part of the input impedance than the antennas realized on a the low

permittivity substrate (diell)

Figure 3.15 shows the imaginary part of the input impedance (called input

reactance) of the circularly polarized antennas versus the slot width Antenna B

shows a strongly negative input reactance showing values up to -j 200 Q Antennas


IT3V

I

50'

o-

-50-

S

I"o1 -100-

? -150-

t&

Ë -200-

•250' 1- -t-

2 S

-a- antenna B diell

-x- antenna B diel2

-e- antenna C diell

-9- antenna D diell

-a- antenna D diel2

-*- antenna E diell

— antenna E diel2

Figure 3.15. Imaginary part of input impedance versus slot width.

D and E exhibit a slightly negative input reactance (up to -j 40 ft) for diell and a

slightly positive input reactance for diel2 (up to +j40fi). The input reactance of

antenna C is close to zero. The input reactance depends on the slot width and on

the perturbation type.

3.6.5 Summary of Parameter Study-

In this parametric study four different circularly polarized and a linearly polarized

ASA's were investigated in terms of size, impedance and ARB. The ring slot widths

were varied from 0.5 mm to 5 mm.

The electrical size of an antenna depends on the slot ring width, the used sub¬

strate and the perturbation type. It increases with an increasing slot width and

a decreasing substrate permittivity. Compared to the linearly polarized antenna

A, antenna B shows a larger electrical size whereas the other circularly polarized

antennas show a smaller electrical size.

The ARB is proportional to the antenna size. Antenna B exhibits the largest

ARB (11%) of all antennas. For a given ARB however antennas D and E exhibit

a smaller electrical size. Antenna C shows clearly the lowest ARB and is therefore

not interesting.

Observing the input impedance of the different antennas it can be seen that the

input resistance depends on the antenna size and the substrate permittivity. The

larger the size and permittivity the higher is the input reactance. The achieved

values range from 10 £7 up to 300 ft. The imaginary part of the input impedance,

the input reactance, depends mainly on the perturbation type.

Antenna B is the most interesting type because it achieves the highest ARB.


However it exhibits the largest size (for a given ARB) and a high input reactance.

The high input resistance and reactance have to be transformed to a commonly

used 50 ft input impedance with the help of a proper feeding circuit.

3.7 Examples

3.7.1 Objectives

The parameter study of the previous section showed the general behaviour of the

different circularly polarized ASA's. In this section some practical examples with

different feeding solutions will be shown. They have all been fabricated, measured

and compared with numerical simulations.

The input impedance of the slotline-fed antennas of the previous section range

from 10 ft to 300 ft showing large imaginary parts. A suitable feed structure must be

found to match this impedance to a 50 f2-input signal coming from an unbalanced

feed. Three different transitions allowing to feed a slot line by an unbalanced line

(namely microstrip line, cpw or coaxial line) have been introduced in section 3.4.1.

Among the three transitions shown in figure 3.6 the microstrip-to-slotline transition

(figure 3.6 c)) is the most flexible one for matching purposes. The two stubs (slotlineand microstrip line stub) allow to compensate for the imaginary part of the input

impedance and to transform the real part to 50 ft. However this solution is not

uniplanar. The results of four different circularly polarized ASA's using this feed

will be presented. For uniplanar circuits the cpw-to-slotline transition is the most

convenient feeding solution. A variety of cpw-to-slotline transitions exist [10]. A

drawback is that these circuits require more space and are less flexible for matching

purposes than the microstrip-to-slotline transition. Furthermore, in order to realize

a 50 f2-cpw line the permittivity er of the used substrate needs to be higher than

4. For lower permittivities the ratio of the central conductor width to the slot

width of the cpw becomes too large and therefore impossible to fabricate. The

electromagnetic fields are not as strongly concentrated in the substrate as in the

case of a microstrip line. Therefore the requirement on the losses of the dielectric

substrate are less severe and allow the use of cheaper materials such as FR4. This

makes the uniplanar architecture suitable for low-cost applications. An example of

a cpw-fed circularly polarized ASA built on FR4-substrate will be presented. The

used cpw-to-slotline transition is shown in figure 3.6 a) and was introduced in [53].The third feed-type, which will be presented in this chapter uses a coaxial line feed

together with a coaxial line-to-slotline transition shown in figure 3.6 c) and was

introduced in chapter 3.4. This transition does not need any substrate and can

be realized on a simple metal plane, which makes it very interesting for low-cost

applications.

3.7 Examples 39

3.7.2 Microstrip-Fed Circularly Polarized ASA

Four different microstrip-fed circularly polarized ASA's have been investigated nu¬

merically and experimentally in order to confirm the findings of the previous section.

Their geometry are type B and D (figure 3.10), respectively. Each antenna type has

been built on two different substrates, namely the RT/Duroid 5880 with a thickness

of h — 0.508 mm and a permittivity of eT — 2.2 and the RT/Duroid 3010 with a

thickness of h — 0.635 mm and a permittivity of er — 10.2. Figure 3.16 shows the

geometry of the investigated antennas.

Figure 3.16. Geometry of investigated antennas. Dimensions are given in table 3.1.

The microstrip feed allows a good matching flexibility and does not interfere

with the radiated fields. In order to confirm the findings of the previous section

four electrical specifications of the analysed antennas were determined, namely the

10 dB-impedance bandwidth, the 3 dB axial ratio bandwidth, the size in terms of

wavelengths and the frequency of the axial ratio minimum. The dimensions as well

as the electrical specs of all four antennas are summarized in table 3.1.

The results confirm that the impedance bandwidth depends mainly on the elec¬

trical size of the ASA. The perturbation type or substrate permittivity or thickness

play a minor role. The 3dB-ARB is about 3.3 times smaller than the impedance

bandwidth in case of the low permittivity substrate and about 4.6 times smaller

in case of the high permittivity substrate. This is typical for single fed circularly

polarized antennas and confirms that in terms of operating bandwidth the ARB is

the limiting factor.

The microstrip-to-slotline transition proofs to be suitable in order to match the

ASA input impedance to a 50 il-feed line.

3.7.3 CPW-Fed Circularly Polarized ASA

This section shows an example of a cpw-fed circularly polarized ASA on a 0.8 mm-

thick low-cost FR4 material (et — 4.4, tanô = 0.025). The relatively high permit¬

tivity of FR4 is necessary in order to realize the cpw as explained in section 3.7.1.


Antenna B Antenna D

er = 2.2 er = 10.2 eT = 2.2 £r = 10.2

h = 0.508 h = 0.635 h = 0.508 h = 0.635

'in15.2 13 17 12

^out 17.2 15 20 15

'stub26.2 18 19 13

d 6 5

size[\o[ 0.42 0.27 0.33 0.24

impedance bandw. [%] 24.9 14.6 18.8 12

center freq. of ARB [GHz] 2.43 2.23 2.46 2.42

3dB axial ratio bandw. [%] 7.4 3.18 5.76 2.49

Table 3.1. Dimensions and electrical specs of the investigated microstrip-fed ASA shown

in figure 3.16. All dimensions arc in mm.

Although FR4 is a lossy substrate, it introduces only small losses, compared to a

patch antenna using the same substrate. This is due to the lower antenna quality

factor (as explained in chapter 2.3.1), which is due to the fact the ASA exhibits a

large radiating aperture and only a small part of the antenna fields resides in the

substrate. The geometry of the antenna is shown in figure 3.17a). The antenna is

too

100

100

120

a) cpw-fed cicularly polarized ASA b) coaxial line-fed circularly polarized ASA

Figure 3.17. Geometry of uniplanar circularly polarized ASA's. a) cpw-feed b) coaxial line

feed, (all dimensions in mm)

3.7 Examples 41

fed through a cpw-to-slotline transition introduced in section 3.4.1. One slot of the

cpw is directly connected to the feeding slot line whereas the other slot of the cpw

is bent into a phase shifting loop to achieve a 180 degrees phase shift before being

connected to the slotline. The slot ring width is 4 mm and the perturbation is of

type B. In section 3.5 it was found that the wider the slot width the larger the ARB

but also the higher the input impedance. Therefore, the largest possible slot width

was chosen, which could still be matched with the cpw-to-slotline transition. The

transition transforms the slotline input impedance of 200 ft — j 120 SI (at 2.4 GHz)to the 50 ft of the feeding cpw. The total size of the slot structure (not including

the groundplane) is 45 mm x 45 mm corresponding to 0.4 A x 0.4 A.

The antenna was simulated with ENSEMBLE from Ansoft. Figure 3.18 shows

both the simulated and the measured results of the return loss, the axial ratio and

the far-field radiation pattern(2.4GHz). An ARB of 9.3% was achieved in simula¬

tion but measurements have shown only 5.9%. The ripple occurring in the measure¬

ment of the axial ratio indicates a measurement error. The 10 dB-impedance band¬

width was 40 %, with simulation and measurement in good agreement. With over

40 % impedance bandwidth this antenna is a low-cost broadband antenna solution.

For the axial ratio a fairly good agreement between measurement and simulation is

found. A minimum axial ratio of 0.8 dB is achieved and the frequency of the min¬

imum axial ratio differs by only 0.8 % between measurement and simulation. The

radiation pattern is shown for the frequency of the minimum axial ratio at 2.4 GHz.

The antenna radiates RHCP to the front side and LHCP to the back side. For the

main radiation direction at broadside the cross polarization lies 20 dB below the

main polarization. A good agreement is achieved between measurement and sim¬

ulation. The maximum gain is 2.5 dBi in simulation and 2.3 dBi in measurement.

The simulated efficiency of the antenna is better than 90 %.

This antenna demonstrates clearly the various advantages of uniplanar circularly

polarized ASA's. It is realized with only one metalization layer using a low-cost sub¬

strate, which makes it ideal for low-cost applications. A large ARB (covering easily

the ISM-band) as well as a large impedance bandwidth are achieved. Although a

relatively lossy material is used the efficiency of the antenna is high.

3.7.4 Coaxial-Line Fed Circularly Polarized ASA

For numerous antenna applications the signal is fed by a coaxial line. To demon¬

strate the compatibility of the ASA with coaxial feeding an example for such an

antenna was realized. The geometry of the antenna is shown in figure 3.17 b). The

size of the slot structure (not including the groundplane) is 50 mm x 50 mm cor¬

responding to 0.45 Ao x 0.45 A0. The transition from the feeding coaxial line to

the ASA is realized with the coaxial-line-to-slotline transition introduced in section

3.4.1. For this antenna a 0.127 mm-thick Duroid 5880 substrate with a permittivity

of eT — 2.2 was used. The main purpose of the dielectric is to serve as a mechanical

support of the conducting parts. From the electrical point of view the effect of the

substrate is negligible because it is very thin and the electromagnetic field pene¬

trates very little into the substrate. Therefore it can be replaced by a lossier and


2.2 2.4 2.6

Frequency [GHz]

2.8

§

«1

a

05

Simulation

Measurement

2 3

Frequency [GHz]

18(f

aim. RHCP

LHCP

........ ffieas. LHCP

meas. RHCP

Figure 3.18. Simulation result of cpw-fed circularly polarized ASA. The radiation pattern

is measured at 2.4 GHz and shows the horizontal plane of the antenna.

cheaper supporting material such as a simple plastic foil. This together with the

fact that no airbridges are needed makes the design extremely cheap and suitable

for low-cost mass applications.The input impedance of the antenna as well as the ARB increases with the ring

slot width. The maximum slot width, which could still be matched to the 50 ft-

coaxial feed line was chosen in order to achieve the largest possible ARB. This leads

to a slot width of 6 mm yielding an ARB of 15% and an impedance bandwidth of

more than 60 %. The simulated and measured results for the axial ratio, the return

loss and the far field are shown in figure 3.19.

A good agreement between measurement and simulation can be seen for the

axial ratio and the return loss. However, the measured axial ratio shows significant

ripples which are due to measurement effects. The far field at 2.45 GHz is shown on

the bottom of figure 3.19. The front side radiation is now LHCP and the backside

radiation is RHCP. Note that the location of the perturbations with respect to the

3.7 Examples 43

l\/ !

5 ^

Frequency [GHz]

sim. LHCP

RHCP

meas. RHCP

-'—- meas. LHCP

istr

Figure 3.19. Simulation result of cpw-fed circularly polarized ASA. The radiation pattern

is measured at 2.4 GHz and shows the horizontal plane of the antenna.

feed are opposite to the previous antenna. The agreement between measurement

and simulation is not as good as for the cpw-fed antenna of the previous section

but it still is reasonable. The measurement shows more back side ripples which

are due to groundplane edge diffraction. The simulated gain is 2.6 dBi whereas the

measurement yields 2.1 dBi.

The presented antenna shows various advantages. It is uniplanar, does not need

an expensive rf-substrate, shows a large impedance bandwidth and ARB and does

not need any airbridge. If a lumped circuit is connected to the antenna instead of a

feed line the latter could be connected at the same location where the coaxial line

is connected to the slotline circuit.


3.8 Chapter Conclusion

This chapter has treated different types of annular slot antennas. The basic modes

of the ASA were introduced. It was shown how to excite circular polarization. The

differences in the input impedance behaviour between linearly polarized and circu¬

larly polarized ASA's was illustrated. Suitable feed transitions for microstrip line,

cpw and coaxial line were presented. A parameter study has revealed the mutual

dependencies between electrical size, axial ratio and input impedance. Finally, these

findings were confirmed by building several circularly polarized ASA's employing

different feeds and dielectric substrates. The potential for low-cost applications of

the uniplanar circularly polarized ASA's was demonstrated.

Chapter 4

Unidirectional Radiation

4.1 Introduction

Slot antennas are inherently double-sided radiators due to the opening to both

sides of the groundplane. For various applications, however, single sided radiation

is required. Besides the advantage that radiating the energy to only one side leads

to higher gain one-sided radiation reduces also the problem of coupling to nearby

objects. If the antenna is designed for use in free space, but is mounted close to some

objects, the antenna characteristics such as impedance matching, radiation pattern

and polarization can deteriorate significantly. In such a situation patch antennas

have advantages since they radiate inherently in only one direction. Nevertheless the

effort to make inherently double-sided radiating slot antennas single-sided radiating

is justified if the advantages of the slot antennas can be preserved. Additionally the

production advantages due to the easy mounting of active elements and the cost

advantages due to the possibility of using low cost dielectrics are preserved as well.

A brief overview of the most commonly used techniques to achieve single sided

radiation were introduced in chapter 2.4. For this thesis two of these techniques,

namely the cavity-backing and the coupling to a patch, have been adapted to the

ASA and systematically investigated. These investigations will be presented in this

chapter.

4.2 Cavity Backing

4.2.1 Previous Work

The first work on cavity-backed slot antennas was done on rectangular slot anten¬

nas. Theoretical investigations were made in [39, 40] and experimental work was

published in [41]. Similar work was done for the cavity backed annular slot an¬

tenna [42, 43, 44]. Recently some work has been published on broadband inverted

microstrip-fed cavity-backed slot antennas [59, 60, 61]. In those papers the authors

made use of asymmetrical slot structures in order to excite several modes. This

allows to broaden the impedance bandwidth but at the same time degrades the

46 Unidirectional Radiation

symmetry of the radiation pattern and the purity of the polarization.

4.2.2 Objectives

Backing the linearly and circularly polarized ASA's presented in chapter 3 alters

antenna specs such as impedance bandwidth, ARB and antenna gain. Up to the

authors knowledge there is no work published comparing cavity backed ASA's with

non cavity backed ASA's in order to quantify the influence of the cavity backing.

In this section a systematic study will be presented comparing cavity backed

and non cavity backed linearly and circularly polarized ASA's allowing to quantify

the influence of the cavity backing on the different antenna parameters.

4.2.3 Antenna Topology

The antenna topology of the investigated linearly and circularly polarized antennas

is shown in figure 4.1. The ASA is built on a 0.8 mm-thick FR4 substrate having

a permittivity of er = 4.4 and a loss tangent of tanÔ = 0.025. The lossy FR4

Figure 4.1. Cavity backed ASA; Exploded view (left side) and top view (right side). In

the linearly polarized case no perturbation is present (r., = r0).

substrate was chosen in order to demonstrate that with the presented slot antenna

architecture cheap and lossy substrates can be used without suffering from substan¬

tial losses. The antenna is fed with a microstrip line through a microstrip-to-slotlinetransition. The printed circuit is used in an inverted configuration, where the mi¬

crostrip feed is located on the top and the groundplane-slot-layer on the bottom

of the circuit, respectively. The cavity is located below the printed circuit and is


size depth relative dimensions reson. fr.1

x x y z

cavity 1 86.9 x 86.9 30.7 0.72 A x 0.72 A x 0.26 A 2.44 GHz

cavity 2 86.9 x 86.9 6.1 0.72 A x 0.72 A x 0.06 A 2.44 GHz

cavity 3 50x45 30.7 0.42 A x 0.38 A x 0.26 A 4.48 GHz

cavity 4 50x45 6.1 0.42 A x 0.38 A x 0.06 A 4.48 GHz

1resonance frequency of TMi=ii!/=i]Î=o cavity mode

Table 4.1. Absolute and relative dimensions of investigated cavities, (all dimensions in

mm)

soldered to the antenna groundplane. In this way the back radiation is trapped in

the cavity and redirected to the front side.

Four different cavity sizes were used in order to study the influence of the cavity

size on the antenna specs. Their dimensions are summarized in table 4.1. Cavity 1

and cavity 2 have the same size (length and width) but different depths. Similarly,

cavity 3 and 4 have the same size (different from 1 and 2), but different depths. The

size of cavities 3 and 4 is significantly smaller than that of cavities 1 and 2. The

size of cavity 1 and 2 was chosen in order to have the first resonance (TMno mode)at 2.44 GHz. 1 The size of cavities 3 and 4 were chosen to be the smallest possible,which means slightly larger than the slot antenna size. The two base sizes of the

cavities were chosen in order to study the influence of the size on the impedanceand axial ratio bandwidth as well as on the slot antenna size. The choice of the

two depths was motivated by similar considerations. For reflecting back planes a

distance ofj was seen to be ideal (chapter 2.4.1). For the practical reasons the

smallest possible depth is favoured. Therefore two different cavity depths were

used. Cavity 1 and 3 have a depth of about £ and cavities 2 and 4 have a depth of

about ^j. All dimensions of the cavities as well as their first resonance frequencies

are summarized in table 4.1.

4.2.4 Results of Linearly Polarized ASA

Three different linearly polarized antennas are investigated. Antenna A does not

have a cavity backing and serves as a reference structure. Antenna B and C are

backed by cavity 1 and 2, respectively. The most important antenna dimensions

are summarized in table 4.2. All antennas have the same slot width and nearlythe same diameter. Figure 4.2 shows the simulated impedance and return loss of

antennas A, B and C.

1Note that the calculation of the resonance frequency is based on an entirely closed metallic

cavity neglecting the slot of the antenna.


antenna A antenna B antenna C

mean radius [mm] 17.2 17.7 17.7

slot width [mm] 4 4 4

circumference [Ao] 0.88 0.91 1.06

cavity size [Ao] - 0.72 x 0.72 0.72 x 0.72

cavity depth [A0] 0.25 0.06

impedance bandwidth [%r 24.8 / 17.6 9.1 /7.1 13.7 / -

gain1 [dBi] 2.3 / 3.7 5.1 / 5.9 5.43 / -

both, simulation and measurement result are shown (sim./meas.)

Table 4.2. Major dimensions and electrical key figures of linearly polarized cavity backed

ASA's.

Frequency [GHz]

Figure 4.2. Impedance of antennas A,B and C.

Antenna Size

The resonance occurs at 2.44 GHz for antenna A, at 2.49 GHz for antenna B and at

2.87 GHz for antenna C. Relative to antenna A this corresponds to an increase of

2% for antenna B and 17% for antenna C. The normalized circumferences of the slot

ring are 0.88 Ao for antenna A, 0.91 Ao for antenna B and 1.06 Ao for antenna C. In

comparison to antenna A the normalized slot-ring circumference increases by 3% in

case of antenna B and by 20% in case of antenna C. It can be concluded that cavity 1,

with a depth of |, does not significantly alter the resonance frequency compared to

the same antenna without any cavity. For a shallow cavity, however, the resonance

frequency and consequently the electrical slot ring size (meaning the size in terms of

the free space wavelength at the resonance frequency) increases significantly. Table

4.2 summarizes the most important dimensions and performances.


Input impedanceThe input impedance of the antenna seen on the feeding slotline varies strongly from

antenna A without any cavity to antennas B and C with a cavity. The microstrip-

to-slotline transition introduced in section 3.4 allows to match these different input

impedances to the 50 J7-microstrip feed lines. The 10 dB-impedance bandwidth of

the three antennas are 24.8% (antenna A), 9.1% (antenna B) and 13.7% (antenna

C). In comparison to antenna A this corresponds to a bandwidth decrease of 63%

for antenna B and 45% for antenna C. As explained in chapter 2.3.1 the impe¬

dance bandwidth is inversely proportional to the quality factor of the antenna. The

cavity backing reduces the radiation resistance because the backside radiation is

suppressed and therefore increases the total quality factor. The roughly halving of

the bandwidth in case of antenna B is due to suppression of the backside radiation.

Antenna C exhibits a higher impedance bandwidth than antenna B because the slot

ring structure and therefore the radiating aperture is larger.

Far field characteristics

Figure 4.3 shows the simulated and measured radiation pattern of antennas A and

B. Antenna C has not been measured and is therefore not shown. The simulated

and measured radiation pattern show significant differences due to the assumption

of the infinite groundplane in the simulation. In reality the groundplane has a size

of 100 mm x 130 mm which corresponds to an electrical length of 0.8 Ao x 1A0. The

Simulation Measurement

antenna B

Figure 4.3. Simulated (with ENSEMBLE) and measured radiation pattern of antennas

A and B.

finite groundplane size alters the radiation pattern in several ways compared to the

simulation, where an infinite groundplane is assumed. The pattern nulls seen in


simulation at the angles of ± 90 degrees. In the case of antenna B, the simulation

shows no back side radiation but measurements reveal a backside radiation of 11 dB

below the front side radiation. Hence the backside radiation is clearly lowered but

not completely suppressed. The remaining backside radiation is not due to cavity

leaking, but to diffraction effects due to the finite size groundplane. The simulated

gain of antenna A is 2.3 dBi whereas the measurement shows 3.7 dBi. The higher

gain in the measurement is due to the narrower beam in the H-plane of the antenna

for a finite-size groundplane. Antenna B shows a simulated gain of 5.1 dBi, and a

measured gain of 5.9dBi and antenna C which was only simulated shows a gain of

5.4 dBi. The simulated increase of gain of antennas B and C compared to the non-

cavity-backed antenna A is 2.8 dB for antenna B and 3.1 dB antenna C, respectively.The gain increase of more than 3 dB in the latter case is possible, because with the

cavity backing the physical size of the radiating slot is increased compared to the

non cavity backed antenna. Compared to the gain of antenna A the measurement

yields an increase of gain of 2.2 dB for antenna B. Hence a clear gain increase

can be achieved by the cavity backing. In case of antenna B the lower measured

gain increase compared to the simulated one is due to backside radiation in the

measurement, which is not properly modelled in the simulation due to the infinite

groundplane.

To conclude this section it can be stated that the cavity backing allows to direct

the radiation to one side. The back side radiation is efficiently reduced but not

completely suppressed due to diffraction on the finite size groundplane. Compared

to a non cavity backed double-sided radiating ASA the gain increase achieved with

the cavity backing ranges between 2 and 3 dB.

In case of antenna B using cavity 1 with a depth of | the electrical size of the slot

ring structure increases only negligibly compared to a non cavity backed antenna

A. Using cavity 2 (antenna C) with a depth being smaller than ^ the slot ring

size increase by 20%. The impedance bandwidth of antenna B is roughly halved

in comparison to antenna A due to the halving of the radiation space2. The use of

cavity 2 yields a larger impedance bandwidth than cavity 1, which comes from the

increased electrical size of the slot ring structure.

4.2.5 Results of Circularly Polarized ASA

In this section four different antennas are compared in order to study the influence

of the cavity backing on circularly polarized ASA's. Antenna D is the reference

antenna without cavity backing, antennas E, F, and G are backed by cavities 1, 3

and 4, respectively. The dimensions of the cavities are given in table 4.1. All four

antennas present the same slot width. The mean radius of the slot ring is the same

2With the suppression of the backside radiation the radiated power generated from a givenfield distribution in the slot ring aperture is halved. This corresponds to a halving of the radiation

resistance. Furthermore the antenna stores additional reactive energy in the backside cavity.Both effects lead to an increase of the quality factor and therefore to a decrease of the impedance

bandwidth.


for the antennas E,F and G but slightly smaller for antenna D. The antenna setup

is shown in figure 4.1 and the most important antenna dimensions are summarized

in table 4.3.

Antenna Size

Figure 4.4 shows the simulated impedance and the return loss of the four antennas.

The return loss shows the two characteristic minima of the single-fed circularly

polarized ASA. In order to compare the electrical sizes of the different antennas

their circumference was normalized to the free space wavelength A0 at the center

frequency. The results (summarized in table 4.3) show that compared to antenna

D the electrical size of the slot ring (circumference) increases by 13% in case of

antenna E, by 17% in the case of antenna F and by 47% in the case of antenna G.

The smaller the cavity the larger becomes the electrical slot ring size.

Impedance Bandwidth

The simulated and measured 10 dB-impedance bandwidth of the four antennas are

summarized in table 4.3 as well. Antenna D without a cavity backing shows a

broad impedance bandwidth of 34%. Antennas E, F, and G exhibit an impedance

bandwidth of 19%, 23% and 22%, respectively. As it was seen in the linearly

polarized case the cavity backing with cavity 1 (antenna E) leads to a roughly

halving of the bandwidth in comparison to antenna E. Using smaller cavities in

case of antennas F and G increases the bandwidth compared to antenna E by 20%

and 15%, respectively. This is due to the larger slot ring size.

The impedance bandwidth is inverse proportional to the quality factor (section

2.1). By the suppression of the backside radiation, the radiation losses are halved

and therefore the quality factor doubles. This explains the halving of the impe¬

dance bandwidth in case of antenna E. Antennas F and G exhibit a electrically

antenna D antenna E antenna F antenna G

mean radius [mm] 14.7 16.5 16.8 16.8

slot width [mm] 3 3 3 3

stub length rs [mm] 25.1 22.6 24 26

stub width bs [mm] 6 6 6 6

circumference [Ao] 0.75 0.85 0.88 1.1

cavity type - cav 1 cav 3 cav 4

impedance bandw.1 [%] 34.6/34.6 19/17 23.3/14.4 22.0/21.23 dB-ARB1 [%] 7.1/8 3.7/3.4 3.4/4.1 0/0gain1 [dBi] 1.5/4 4.7/6.2 4.8/5.5 4.6/5.3

both, simulation and measurement result are shown (sim./meas.)

Table 4.3. Main dimensions and electrical specifications of circularly polarized cavity

backed ASA's.


*

> /

antenna D

antenna E

antenna F

antenna G

_i —j i—, 1 1

3 3.2 3.4 3.6 3.8 4

Frequency [GHz]

Figure 4.4 Simulated input impedance of circularly polarized antennas D, E, F and G.

larger radiation aperture (slot ring) and therefore also present a larger impedance

bandwidth.

Far Field Characteristics

Figure 4.5 shows the simulated and measured radiation patterns of antennas D and

E. Antenna D radiates to both sides whereas antenna E is backed by a cavity and

radiates therefore only to one side. The radiation pattern of antennas F and G

are similar to the one of antenna E. Comparing simulation and measurement two

major discrepancies can be seen. Due to the infinite groundplane assumed in the

simulation, the simulated beam width is wider than the measured one. Furthermore,

the simulated backside radiation is zero for antenna E,F and G whereas in the

I

o

as

u

10

N

20» '• A» /i -. t \ f

.» i ;* i » ii *. \ ; ••

i t \ »V

In/ii W

1

30

40 H,

2.2 2.4 2.6 2.8


measurement the backside radiation is existent although small. A front to back ratio

of -18 dB is achieved for the cavity backed antennas. Table 4.3 shows the simulated


antenna D RHCP — • antenna E RHCP

antenna D LHCP antenna E LHCP

Figure 4.5. Simulated and measured radiation pattern of antennas D and E.

and measured gains of antennas D,E,F and G as well as their ARB's. Note that in

the simulation the gain increase of antennas E, F and G is slightly higher than 3 dB,which comes from the fact that the electrical size of the slot increases slightly. For

the same antenna the measured gain is up to 2 dB higher than the simulated one.

This results from the finite groundplane producing a narrower beam and therefore a

higher gain. The measured gain of antenna E,F and G are 2.2 dB, 1.5 dB and 1.3 dB

higher than the gain of antenna D. There are three main reasons that the gain does

not completely double. The radiation pattern of the cavity backed antenna shows

a wider beam (therefore a lower directivity) than the non-cavity backed antenna

(see fig. 4.5), the back side radiation is not completely eliminated and the cavity

produces losses. The losses are especially pronounced in the case of antenna E and

F, which exhibit approximately the same beam width and back side radiation as

antenna E but at significantly lower gain.

Besides the gain an investigation of the influence of the cavity backing on the

ARB was done. For antennas D, E and F an axial ratio minimum below 0.5 dB

could be achieved in the simulation. For antenna G using cavity 4 however the best

achievable axial ratio was only 6 dB. If the cavity dimensions are too small the two

slot ring modes can not be properly excited. Figure 4.6 shows the simulated and

measured axial ratios. A good agreement is achieved for antennas E, F and G. In



Frequency [GHz] Frequency [GHz]

antenna D antenna F

antenna E antenna G

Figure 4.6. Simulated and measured broadside axial ratio of circularly polarized antennas.

the measurement Antenna D exhibits the same 3-dB axial ratio bandwidth as in

the simulation. In contrary to the latter, however it contains important ripples and

shows a higher axial ratio minimum (2 dB instead of 0.5 dB in simulation). This is

due to diffraction effects on the finite groundplane, which do not occur in simulation

(ENSEMBLE).The 3-dB ARB of antennas E and F (3.4% and 4.1%, respectively) are roughly

half of antenna D (8%). The ARB is inversely proportional to the the quality factor,

which explains the halving of the ARB. The 3dB-ARB is about five times smaller

than the 10-dB impedance bandwidth. Hence the ARB is the limiting factor of the

operating frequency bandwidth of the antenna. A fairly well agreement between

measurement and simulation can be seen for antennas E, F and G. For antenna

D the measured minimum of the axial ratio is significantly higher than predicted.

The 3-dB ARB in measurement and simulation however are nearly the same. For

all antennas a slights frequency upshift of the measured curve compared to the

simulated one can be seen.

4.2.6 Summary

Cavity backing is an interesting solution to achieve single-sided radiation for an

ASA. It allows to redirect the backside radiation to the front side without ma¬

jor changes of the slot ring structure. This results in a higher gain and a good

electromagnetic isolation to the back side. However, the backside radiation is not

completely eliminated due to diffraction on the groundplane edges. With a slot ring

size of 0.4 Ao x 0.4 Ao and a groundplane size of 1 Ao x 0.8 Ao a front to back ratio

of 20 dB was achieved.

Compared to the double-sided radiating ASA most electrical properties are al-


tered by the cavity backing. The impedance bandwidth and the ARB are roughlyhalved due to the increase of the quality factor. The electrical size of the investi¬

gated slot ring structures increases up to 17% for a |-deep cavity backing and up

to 40% for a ^j-deep cavity backing. The gain increases but is not exactly doubled

due to the persisting backside radiation and the losses.

4.3 Annular Slot Coupled Patch Antenna

4.3.1 Objectives

In the previous section it was shown how to achieve unidirectional radiation by

redirecting the backside radiation using a cavity. In this section it will be shown

how to achieve unidirectional radiation with the use of a patch resonator located on

the front side of the antenna. This approach presents several advantages compared

to the cavity backing. It can be produced as a layered structure and does neither

need vertical metallic walls nor soldering to the groundplane. This makes the design

more suitable for mass production by reducing the costs.

The difference between this architecture and the previously introduced ASA

architectures resides in the fact that the radiation does not arise from the slot ring

but from the patch resonator. The electromagnetic field in the slot ring couples

into a patch resonator mode whose fields reside between patch and groundplane.The field is then radiated like in a traditional patch antenna. Figure 4.7 shows

an exploded view of two circularly polarized annular-slot-coupled patch antennas.

The main difference compared to the double sided radiating ASA is in the excited

circular patch annular patch

Figure 4.7. Exploded view of annular slot coupled circular and annular patch antenna.

electromagnetic mode. In the case of the ASA a slot ring resonance is excited and

the circumference of the ring corresponds to one guided wavelength. In the slot

coupled patch antenna a patch resonance is excited and the circumference of the


coupling slot is smaller than one guided wavelength. The major radiation arises

from the patch mode and is directed to the front side. However, due to its opening

to the backside, the coupling slot produces a small but non-negligible radiation to

the backside of the antenna. The level of the backside radiation depends on the

electrical size of the coupling slot and the electrical size of the patch.

Several single-fed slot coupled patch antenna designs showing similar electrical

properties and lower back side radiation levels are known from the literature [5,6, 20, 7]. In these designs the circular polarization is excited with the help of an

asymmetric patch shape. The particularity of the here introduced architecture,

however lies in the fact, that the polarization is determined by the coupling slot

shape and not by the patch shape. The used patch shape is either a circular or

an annular patch not enforcing or favouring any specific polarization. By changing

the shape of the coupling slot different polarizations can be obtained keeping the

same patch shape. Therefore this architecture is well suited for active switching of

the polarization and the operating frequency. The switching circuitry can be easily

integrated in the uniplanar slot circuit. In the next chapter methods to switch

polarization by electrically changing the geometry of the slot ring structure will be

introduced. Using the antenna architecture presented here all switching elements

and biasing circuits can be included in the groundplane and are therefore well suited

for mass production. If the perturbation exciting the circular polarization would

be integrated in the shape of the patch, the complexity and costs to make this

perturbation switchable with active devices would be much higher. Furthermore,

using this architecture we can use the same switching architecture for double-sided

ASA's as well as for single-sided radiating annular slot coupled patch antennas.

In this chapter different annular slot coupled patch antennas will be introduced.

An investigation on the influence of geometrical parameters on the main electrical

specifications will be presented. Finally, a uniplanar cpw-fed slot coupled annular

patch antenna will be presented and investigated in detail both numerically as well

as experimentally.

4.3.2 Study on Annular Slot Coupled Circular Patch

Antenna

Antenna SetupIn this section a numerical study on an annular slot-coupled circular patch antenna

operating in the range of 1.9 GHz to 2.5 GHz will be presented. Figure 4.8 shows the

antenna setup consisting of the microstrip feed line, the coupling slot and a circular

patch. The groundplane of the antenna is located on top of a 0.79 mm-thick Duroid

5880 dielectric with a permittivity of ey = 2.2. The size of the groundplane is

100 mm x 100 mm. The microstrip feed is located on the same substrate but on the

opposite side of the coupling slot. The circular patch is located 4 mm above the

groundplane. Patch and groundplane are separated by a foam material (Rohacell)with electrical properties close to air. The inner and outer radius of the annular

slot are 12.65 mm and 15.65 mm, respectively. Two square slot perturbations are


exploded view

top view

antenna setup

patch

foam (Rohacell)

h = 4inm

e„ = 1.05

ground plane

dielectric (duroid)

h = 0.79mm

Er = 2.2

microstrip feed

patch size

fPatch electr. size

20mm 0.33 X0

23mm 0.35 X0

25mm 0.37 X0

27mm 0.38 \

29mm 0.39 X0

Figure 4.8. Antenna setup of microstrip fed circularly polarized annular slot coupled

circular patch antenna.

connected through a narrow slotline to the slot ring to excite circular polarization.

Investigation of Electrical Specifications

For a chosen patch size of an annular slot coupled patch antenna different coupling

slot sizes can be used. Simulations showed that for the chosen antenna setup the


annular coupling slot can be reduced down to a circumference of 0.59 Ao3. With a

reduction below this size it is not possible anymore to excite a circular polarization.

The maximum coupling slot size can be increased up to 0.72A0. Increasing the slot

size more causes a double-sided radiation and is therefore not meaningful. The ratio

between the coupling slot size and the patch size influences electrical specificationsof the antenna such as resonance frequency, ARB, gain and front to back ratio. In

order to analyze these dependencies five different antennas having the same coupling

slot width and circumference but different patch sizes were simulated4. The radius of

the patch was varied from 20 mm up to 29 mm. This leads to resonance frequenciesbetween 2.0 GHz and 2.44 GHz. The normalized patch diameter ranges from 0.33 Ao

up to 0.39 A0 and the normalized coupling slot circumference ranges from 0.59 Ao

up to 0.72 A0.

In the case of a classical circular patch antenna, without coupling slot, the

resonance frequency is directly proportional to the patch diameter. For a given

dielectric the ratio between the patch diameter and the free space wavelength A0

(at resonance) is approximately constant. This is not the case for the annular slot

coupled circular patch antenna. The resonance frequency depends mainly on the

patch diameter but also on the relative size of the coupling slot. On the bottom

of figure 4.8 the electrical patch size (patch diameter divided by the free space

wavelength) is given for the five antennas. It can be seen that the electrical patchsize increases if the physical size of the coupling slot decreases relative to the patch

size. The reason for that is that in the case of a large coupling slot the electrical

Regular Circular Patch Annular Slot Coupled Circular Patch

annular coupling slot (ground plane)

Figure 4.9. Schematic illustration of current distribution of the linearly polarized mode on

circular patch antenna without and with slot coupling. Note that the annular couplingslot is located in the groundplane.

currents on the patch are forced to flow along the outer side increasing the current

3If a dielectric with a higher permittivity is used lower values can be achieved. For a permittivity

of er = 4.4 e. g. the minimum slot circumference decreased to 0.33 Ao-4For each patch size the perturbation elements of the coupling slot were adjusted in order to

achieve a good circular polarization (minimum axial ratio below 0.7dB). The size of the pertur¬bations is in the order of 3 mm x 4 mm.


path length. The larger the coupling slot compared to the patch size the more the

electrical currents are bent and therefore the smaller becomes the electrical size

of the patch. Figure 4.9 shows schematically the patch current distribution of the

linearly polarized mode for a regular circular patch (without slot coupling) and for

two annular slot coupled circular patch antennas. It can be seen that the presence

of the coupling slot(in the groundplane) suppresses the current flow through the

center of the antenna and lengthens therefore the current path, which results in

a smaller electrical patch size. The same findings apply for circular polarization,

which is simply a superposition of two linear polarized modes.

The 3 dB-ARB and the front to back ratio depend on the electrical size of the

coupling slot. Figure 4.10 shows the ARB and the front to back ratio in function of

the normalized coupling slot circumference. It can be clearly seen, that the ARB

increases with an increasing slot size but at the same time the front to back ratio

decreases. With the chosen antenna setup ARB's ranging from 2.4% to 3.9% and

front to back ratios ranging from 7.2 dB to 3.4dB have been achieved. The larger

^

e

1 r'

! —'—i r—'

I

<

,

— < - • axial ratio bandwidth

—• front to back ratio

>' -

*

s.'

s

'\»-t

*... .Jt'

y*

X. ..--*'

..-*-""""""••-,

+""

iii.ii

3.8

3.6

3.4

3.2

3

2.8

2.6

2.4

2.2

2

0.58 0.60 0.62 0.64 0.66 0.68 0.70 0.72

Slot circumference [X0]

5

- 5

$

Figure 4.10. Front to back ratio and axial ratio in function of electrical antenna size.

the chosen coupling slot size the more power is lost to the back side because the

radiating aperture is increased. Besides the lowering of the front to back ratio this

lowers the antenna quality factor, which results in an increase of the ARB. The

antenna gain ranges from 8 dBi (in case of the smallest coupling slot) to 6 dBi (incase of the largest coupling slot).


Conclusions

Unidirectional radiation can be achieved by coupling the slot fields to a patch

whereby the polarization is exclusively determined by the shape of the coupling

slot. The simulation yields an antenna gain between 6 dBi and 8 dBi, which demon¬

strates clearly the single sided radiation. Contrary to the double sided radiating

structure, the circumference of the slot is smaller than one guided wavelength. It

can be chosen in between a certain range, depending on the used antenna setup.

The larger the coupling slot the higher is the achieved ARB, but the lower is the

front to back ratio. In comparison to the cavity backed ASA the annular slot cou¬

pled circular patch antenna shows a higher gain, similar ARB (depending on the

coupling slot size) but a significantly reduced front to back ration, between 3 dB

and 8 dB.

4.3.3 Study on Annular Slot Coupled Annular Patch

Antenna

Antenna Setup

For many applications a compact antenna size is required. Furthermore for array

applications a small patch size may be desirable in order to minimize mutual cou¬

pling between elements. Various patch antenna size reduction techniques have been

published in the literature [62, 63]. Most of these techniques are not appropriate

for the antenna architectures presented here because they use an asymmetric patch

shape. This makes it impossible to to apply polarization switching as it is pre¬

sented in chapter 5. For any size reduction technique the patch shape has to be

axially symmetric in order not to favour any specific polarization. The annular-ring

microstrip antenna [64] fulfills this criterion because it exhibits rotation symmetry

and provides a smaller electrical size than the circular patch antenna. In contrary

to the circular patch antenna the current can not flow in the center of the patchbut is forced to flow on the outer side. The lengthening of the patch currents leads

to an antenna size reduction exactly in the same way as illustrated in figure 4.9.

Therefore an annular slot coupled annular patch architecture, shown in figure

4.11, was systematically investigated. The annular patch is positioned above the

ring slot in the same manner as the circular patch (figure 4.7). The slot fields couple

into the annular patch resonance mode. For a given patch size different coupling slot

sizes can be used. The minimum possible slot circumference was found to be 0.54 Aq.

In order to investigate the dépendance between these geometrical parameters and

the electrical parameters three different antennas having the same slot size but

different patch sizes have been simulated and compared. The antenna setup is

shown in figure 4.11. The coupling slot is printed on a 1 mm-thick FR4 dielectric.

A simple slotline feed was used for this simulation. Patch and ground are separated

by a 4 mm-thick foam material. The slot ring inner radius is 8.5 mm and the outer

radius 11.5 mm. The perturbation slots used to excite circular polarization are about

4 mm x 5 mm. For each patch size the perturbation size is adapted to achieve a goodcircular polarization. The annular patch inner radius is 11.5 mm and outer radius


exploded view

top view

' Patch

antenna setup

patch

foam (Rohacell)

h = 4rnm

tr=1.05

ground plane

dielectric (FR4)

h = lmm

z=4-3

patch size

rPatch electr. size

17mm 0.29 X„

25mm 0.36 \

30mm 0.39 Xa

slot line feed

Figure 4.11. Antenna setup of circularly polarized annular slot coupled annular patch

antenna.

is 17, 25 and 30 mm, respectively.

Investigation of Electrical Specifications

The resonance frequencies of these antennas are 2.51, 2.14 and 1.92 GHz. The

electrical patch size (diameter) is 0.29, 0.36 and 0.39 A0, respectively. It increases

if the patch size increases relative to the slot size. Figure 4.12 shows the ARB

and the front to back ratio of the three investigated antennas as a function of the


normalized slot circumference. The antennas exhibit an ARB 4.3%, 2.8% and 2.6%

5 |1 1

r— , , r ,, \12

4.5

I

i 3.5

1

1 i '

— 4 - • axial ratio bandwidth

-H»- •— front to back ratio

*"».

X

'• ê

*

v

«1——*""

i i i i i i i i

10

§

s %i.

-yu

s

fi

I

2.5 i 1 1_ 1 1 1 i—i 1 1 2

0.54 0.56 0.58 0.6 0.62 0.64 0.66 0.68 0.7 0.72

Slot circumference [X^

Figure 4.12. Front to back ratio and axial ratio in function of electrical antenna size

and a front to back ratio of 3.5dB, 8 dB and 10.5 dB, respectively. As seen in the

previous study an increasing coupling slot size (relative to the patch size) causes a

higher ARB, a lower front to back ration and a smaller patch size. The simulated

gain of the antennas varied between 6 dB and 7.5 dB.

Conclusions

The comparison of the annular patch with the circular patch shows that in terms

of the electrical patch size there is no significant difference. This is due to the fact

that the coupling slot enforces the bending of the patch currents (as shown in figure

4.9 independently of the metalization in the center of the patch. Therefore both

antenna types show a very similar current distribution. Well understood, without

the annular coupling slot the annular patch would exhibit a significantly reduced

electrical size compared to the circular patch as reported in [12].Comparing the ARB and front to back ratio of the two antennas it can be seen,

that the results are similar. The annular patch presents slightly higher ARB and

front to back ratio.

4.3.4 Example

Slotlines are seldomly used as feeding lines because they are balanced lines and

tend to radiate easily as explained in chapter 2.2.1. The two mainly used printed

transmission lines are the microstrip line and the cpw. Both are unbalanced lines


and radiate less easily than slotlines. In [65] two examples of microstrip-fed annular

slot coupled circular patch and annular patch antennas were presented. In this

section a cpw-fed slot coupled circularly polarized annular patch antenna will be

presented.

Antenna SetupThe antenna is based on the setup shown in figure 4.11. Additionally it comprises a

cpw-feed line and a cpw-to-slotline transition. The complete antenna setup and the

most important dimensions are shown in figure 4.13. The groundplane slot circuit

is realized on FR4. Besides the patch only one metalization layer is needed due

the the uniplanar feed circuit. The circular polarization is excited in the slot ring

having two perturbations.The transition between the 50 ft cpw-feed line and the slotline is done using

a cpw-to-slotline transition published in [52]. It includes a slotline stub (square)and an airbridge. The slotline stub contains a metal island in order to reduce the

radiation of the stub. The impedance matching is done by adjusting the width of

the coupling slot as well as the stub of the transition.

Far Field Characteristics

Figure 4.14 shows the far field pattern at 2.4 GHz (where the minimum of the axial

ratio occurs) and the broadside gain of the two polarizations (bottom) as function

of the frequency. The far field pattern is calculated with HFSS and is shown in two

planes, x-plane and y-plane. The structure shows clearly unidirectional radiation

with similar patterns for both planes. The front to back ratio is 8 dB. The main

polarization is RHCP. The unwanted back side radiation is LHCP. Although some

back side radiation exists, the front to back ratio is above 9 dB, which is acceptablefor many applications. The bottom of figure 4.14 shows the simulated and the mea¬

sured broadside gain for the two circular polarizations. A good agreement between

measurement and simulation was achieved. The HFSS simulation shows clearly

better results, than simulations performed with ENSEMBLE, which is mainly due

to the inclusion of the finite groundplane. The maximum antenna gain is 6 dBi.

This confirms that a good unidirectional radiation is achieved and that the losses

are small even though a lossy substrate is used. The simulated and measured axial

ratio is shown in figure 4.15. Also here a good agreement between measurement

and simulation has been obtained. An ARB of 3.3% and 2.7%, respectively, was

achieved in simulation and measurement.

Input ImpedanceThe width of the annular coupling slot and the size of the rectangular stub of the

cpw-to slotline transition provide two degrees of freedom for the impedance match¬

ing. This is less convenient than in the case of the microstrip-to-slotline transition

(shown in figure 3.6), where the transition itself offers two degrees of freedom, but is

sufficient for the here presented antenna. Figure 4.16 shows the simulated and mea¬

sured input return loss of the antenna structure of figure 4.13. The 10 dB-impedance


antenna setup

patch

foam (Rohacell)

h = 4mm.

er=1.05

dielectric (FR4)

h = lmm

s=4-3

ground plane

0.62

Figure 4.13. Setup and dimensions of cpw-fed slot coupled circularly polarized annular

patch antenna.

bandwidth is 15.4 %. Although this is in principle confirmed by the measurement,

the latter shows a frequency downshift of 5%. The achieved impedance bandwidth

confirms the findings of the previous chapter, that the impedance bandwidth is usu¬

ally about 5 times larger than the ARB and therefore is never critical to achieve.


-180°

RHCP x~plan&

LHCP x-plane

—- RHCP y-plane

LHCP y-plane

18(f

1U i ' 'i' "' —

'i

5

0

I"Ï"

•. \ '•

\ \ ./.•

-. \ / '

'• \ /'•'

-15 -. \ l :

: \ i Ï

-20 \ J'

-25

v.-:

2.2 2.3 2.4 2.5

Frequency [GHz]

2.6

RHCP simulation RHCP measurement

LHCP simulation LHCP measurement

Figure 4.14. Far field characteristics; top: far field at 2.4 GHz; Bottom: broadside gain.

4.3.5 Summary

The aim of this chapter was to show, that the ASA-architecture can provide uni¬

directional radiation by placing a patch above the slot ring. It was shown, that a


Ol 1 i 1 1 —-J

2.3 2.3S 2.4 2.45 2.5 2.55

Frequency [GHz]

Figure 4.15. Broadside axial ratio (simulation and measurement).

Figure 4.16. Impedance matching (simulation and measurement).

patch mode is excited, which is radiating mainly to the front side of the antenna.

The polarization, however is exclusively determined by shape of the coupling slot.

This distinguishes this architecture clearly from previously published single-fed cir¬

cularly polarized antennas (e. g. [5]) and provides it with the necessary flexibilityneeded to design reconfigurable architectures.

Comparing the electrical properties of the slot coupled circular or annular patchantenna to the double-sided radiating ASA it can be seen, that the gain increases

but the impedance bandwidth and the ARB decrease. The explanation for this is

that the suppression of the backside radiation reduces the radiation resistance and

therefore increases the antenna quality factor.

Although the main part of the energy is radiating to the front side a non negli-


gible backside radiation remains. The front-to-back ratio and the ARB depend on

the size of the coupling slot relativ to the size of the patch. Increasing the coupling

slot size increases the ARB but decreases the front-to-back ratio. Depending on the

application a trade off has to be found. A trade off has to be found between these

two figures.

Two different patch shapes, the circular patch and the annular patch were used.

These shapes are ideally suited for the presented architecture, because they do not

impose any polarization direction. The type and direction of the polarization is

exclusively determined by the location of the slot perturbations and the feedingslot line. Finally a slot coupled annular patch antenna with a completely uniplanarfeed circuit has been presented including numerical and experimental results. A

very good agreement is found between simulation (HFSS) and measurement.

Seite Leer /

Blank leaf

Chapter 5

Reconfigurable Slot Antennas

5.1 Introduction

5.1.1 Reconfigurable Antennas

Reconfigurable antennas are characterized by the fact that their electrical properties

can be changed with the use of active elements, such as transistors, pin-diodes or

varactor diodes. They allow to optimize the radiation properties in an environment

with changing propagation characteristics or moving signal sources. This is typi¬

cally the case for mobile phones, wireless communication on a mobile platform but

also in indoor wireless communications. By adapting electrical antenna character¬

istics, such as the radiation pattern, the polarization or the frequency bandwidth,

a maximum signal strength and therefore a larger transmission capacity can be

achieved.

There exist alternative techniques to enhance the wireless transmission capac¬

ity by modifying the transmitted signal. The most popular techniques are Smart

Antennas and MIMO (multiple input multiple output) [66]. A smart antenna con¬

sists of an array of antenna elements connected to a digital signal processor. Such

a configuration enhances the capacity of a wireless link through a combination of

diversity gain, array gain, and interference suppression. MIMO refers to radio links

with multiple antennas at the transmitter and the receiver side. Given multiple

antennas, the spatial dimension can be exploited to improve the performance of

the wireless link. Both techniques require antenna arrays and a complex and costly

signal processing device. In contrary the here presented reconfigurable techniques

can be applied on a single antenna element and act directly on the rf-signal not

requiring digital signal processing.In wireless local area networks (WLAN) e.g., polarization diversity is used to

avoid the detrimental fading loss, caused by multipath effects [67]. In microwave

tagging systems it is used as a modulation scheme such as the Circular Polarization

Modulation [68] where a logical zero is transmitted with one circular polarizationand a logical one with the orthogonal circular polarization.

Radiation pattern diversity [69] consists in altering the radiation pattern. A

specific direction of incidence is favoured and/or certain directions are suppressed

70 Rcconfigurable Slot Antennas

in order to maximize the signal strength. Often the ideal radiation pattern is not

known upfront, but can be determined by switching through the available radiation

patterns, while monitoring the signal strength.Besides polarization and radiation pattern diversity, reconfigurable antennas

can also be designed to allow adaptation of their input impedance bandwidth and

operating bandwidth. This is useful in cases where objects in the proximity of

the antenna detune the input impedance, which leads to severe mismatching losses.

Especially electrically small antennas react sensitive to changes in their environment

because their near field couples to the environment [70]. Typical examples are

mobile handset antennas or antennas for wireless-LAN access cards in personal

computers. In all these cases the antenna induces currents on nearby objects (e.

g. on the hand of a mobile phone user) which become part of the radiating device.

Hence the input impedance of the antenna can not be determined without knowing

its environment. By active impedance tuning, however, the impedance can be

constantly adapted and therefore a good matching can be achieved for a variety of

applications.

5.1.2 Previous Work

Several antenna architectures, offering polarization diversity, have been published.

Here the most relevant architectures are briefly discussed. In [71] a patch antenna,

which allows switching between two linear and two circular polarizations has been

presented. The switching was achieved by using several pin-diodes mounted between

the patch and the groundplane at different locations. By switching one or several of

these diodes using a dc-bias circuit a specific polarization can be excited in the patch

antenna. However, this solution requires a relatively complex biasing network and,

in addition, needs mounting of the diodes between patch and ground plane, which is

inconvenient for fabrication. In [72] a probe-fed patch antenna was presented whose

polarization can be switched between right hand circular polarization (RHCP) and

left hand circular polarization (LHCP). The circular polarization is excited by a slot

which is cut in the patch. The switching is achieved by alternatively shorting this

slot at two different locations with a pin-diode mounted across the slot. However

the design of the dc-bias network is delicate because the pin-diodes are mounted

on top of the antenna patch. The dc-connection between patch and groundplanemust not interfere with the rf-signals. The positioning of the switching circuit and

the dc-feed on patch plane requires an antenna setup, where the patch plane is

printed on an expensive rf-substrate. In contrary the annular slot coupled patch

architecture presented in chapter 4 does not require an rf-dielectric because neither

active devices, nor biasing circuitry are placed on the patch plane. In a similar

design proposed by the same group [47] not the polarization but the operating

frequency band can be switched by changing the electrical length of the patch.In [73] a circularly polarized microstrip-fed aperture-coupled patch antenna was

introduced. Here the polarization was switchable between LHCP and RHCP by

changing the length of the coupling slot in the groundplane of the antenna utilizing

pin-diodes.


5.1.3 Low-Cost Architecture

In the last decade commercial wireless applications have seen an enormous growth.

They replaced the military application as driving force for rf-innovation. As a re¬

sult of this antenna designers have to pay more attention to cost. The production

costs of printed circuits are mainly dependent on the price of the substrate, the

rf-architecture (uniplanar versus microstrip) and the number of metalization lay¬

ers used. Currently used printed circuit materials, such a FR4, exhibit important

losses (tan<5 > 0.02) at microwave frequencies. This is especially severe in the case

of resonant microstrip circuits, where the electromagnetic fields are mainly concen¬

trated in the substrate between strip and groundplane. Therefore various low loss

rf-suitable dielectric materials have been developed for microwave applications in

the last 20 years. Their costs, however, are about five times higher than that of

FR4.

In the case of uniplanar circuits (cpw and slotline) only a small part of the

electromagnetic field resides in the substrate (if the thickness of the substrate is in

the order of the gap width). This allows to use relatively lossy FR4 material without

introducing substantial losses. Besides lower material costs, uniplanar circuits allow

easier and hence cheaper realization of active circuits. Both, series and shunt mount

of lumped elements can be utilized without the need for (expensive) via holes.

5.1.4 Chapter Outline

In this chapter two novel reconfigurable ASA architectures are introduced. Polar¬

ization switching and switching of the operating bandwidth will be demonstrated.

The switching is realized with pin-diodes and a dc-bias circuit. The first architec¬

ture allows switching between two different polarization states utilizing a simple

uniplanar biasing circuit. The second architecture employs a more complex mi¬

crostrip biasing circuit allowing to switch all diodes independently. This allows

switching between three different polarization states and additionally switching of

the operating frequency bandwidth. Using the same switching architecture both,

single and double sided radiation characteristics can be achieved. For both archi¬

tectures, simulation and measurement results of various antennas operating in the

frequency range from 2 GHz to 2.6 GHz will be shown. The architectures presentedin the following are promising candidates for reconfigurable antennas for commercial

wireless applications due to their flexibility and their low production costs.

5.2 Uniplanar Switching Architecture

5.2.1 Switching Principle

A simple ASA fed by a slot line as shown in figure 5.1 a) exhibits x-polarized linear

polarization. With the use of two perturbations placed at -45 degrees and 135

degrees with respect to the feed point as shown in figure 5.1 b) circular polarization

72 Reconfigurable Slot Antennas

a) linear polarization b) circular polarizaion c) switchable polarization

Figure 5.1. Principle of switching between linear and circular polarization.

is excited (chapter 3.5). The polarization switching is realized by mounting a pin-

diode across the perturbation as shown in figure 5.1 c). In the forward biased

state the diode behaves almost as a short circuit and suppresses the effect of the

perturbation. Consequently, the antenna acts like the structure in figure 5.1 a) and

exhibits linear polarization. In the reverse biased state the diode behaves almost as

an open circuit and the antenna acts like the structure in figure 5.1 b), exhibiting

circular polarization. The diode state is controlled with a dc-bias circuit.

5.2.2 PIN Diode

The equivalent circuit of the pin diode is shown in figure 5.2. Rf is the the resis-

£„

w-

R,

R,Mb

T^MKP—i

-W-~

Figure 5.2. Equivalent circuit of pin diode.

tance in the forward biased state and R? and C3 are the resistance and junction

capacitance in the reverse biased state. Lp and Cp are parasitic inductance and

capacitance of the diode. LB is the series inductance due to the connecting wires

and C2 is the extra capacitance due to the resulting structure consisting of diode

and slotline section over which the diode is mounted. For short connections Ls is

negligibly small. C2 must not be considered because its effect is already included


in the field simulation. Cj and Rf vary with the biasing current whereas the other

parameters are constant.

A nearly ideal pin diode (HPND-4050 from Hewlett Packard) with a beam lead

packaging was chosen for the first architecture. In the used bias configuration, the

resistance value in the forward biased state is Rf = 2.2 ft and the capacitance in the

reversed biased state is equal to C,=0.18pF (which corresponds to an impedance

of -j362f2 at 2.44 GHz). Due to the very small packaging size (0.7 x 0.3 mm) the

parasitic capacitance and inductances are negligibly small. In the HFSS simula¬

tion, the diode is modeled as lumped element having the values Rf —2.2 ft and

Q = 0.18pF.

5.2.3 Bias Architecture

In order to operate the pin-diode a dc-bias voltage has to be applied between the

two diode terminals. An elegant solution for this is to split the groundplane into

two separate parts as shown in figure 5.3. Two thin slots running from the end

Figure 5.3. Bias circuit

of the slot perturbations to the edge of the groundplane accomplish the complete

separation of the groundplane into two parts. The terminals of the pin diodes bridgethe two different parts of the groundplane thus allowing the application of the dc-

bias across the gap. Rf-wise the groundplane is not separated if shorting capacitors

are applied as shown in figure 5.3. The dc-bias voltage is provided by a 1.5 Volt

coin cell battery mounted at the edge of the antenna groundplane. The use of a

battery instead of an external dc-voltage source prevents any unwanted coupling

between the radiated signals and the dc feed cables. The diode current is controlled

with a series resistance Rs of 180 ft.


5.2.4 Antenna With Polarization Switching Between Linear

And Circular Polarization

In this section simulated and experimental results are presented for an antenna

which can be switched between linear and circular polarization. Figure 5.4 shows

the complete antenna setup. The biasing circuit is setup in the same way as shown

in figure 5.3. The antenna is build on a 0.8 mm-thick FR4 substrate and fed by a

100

Figure 5.4. Antenna setup

microstrip line via a microstrip-to-slot line transition. Instead of the microstrip feed

a cpw feed could be employed as well, as explained in section 4.3.4. The microstrip

feed is used here because of its convenience in experimenting. The microstrip-to-

slotline transition provides more flexibility in matching. Furthermore in the case of

the cpw feed the necessary airbridges make experimental handling of the antenna

delicate. The slot ring exhibits a diameter of 30.8 mm and the dimensions of the

groundplane are 100 mmx 100 mm. The capacitance value of the shorting capacitors

is Ca= 47 pF. The series resistance, which is used to control the diode current, has

a value of Rs = 180 ft.

In order to achieve a good linear polarization purity the resistance of the diode

in the forward biased state has to be sufficiently small. Otherwise it would cause

a local loading of the slot ring and would therefore have the same effect as a slot

perturbation resulting in an excitation of the unwanted orthogonal mode. In prac¬

tice both, the shorting capacitor Ca and the diode show a small (1-2Î7) impedance,which is small enough to excite proper linear polarization. In the simulation, how¬

ever, it can not be approximated with a short circuit but has to be modeled with

its real value in order to achieve correct results. In the reverse biased state the

diode should ideally be an open circuit exhibiting an infinite impedance. In realitythe diode shows a value of 0.18 pF, which is equivalent to a reactance of -j 362 SI

at 2.44 GHz. For the reverse biased state the non ideal (non-infinite) impedance

value is acceptable because it simply results in an additional capacitive loading.

The capacitive loading introduced by the pin diode is compensated by reducing the


size of the perturbation slot1. It is important to include the non-ideal behaviour of

lumped elements in the full wave simulation.

Radiation Characteristics

Figure 5.5 shows the radiation pattern in both polarization states. In the circularly

circular polarization linear polarization

0° o°

LHCP

RHCP

Figure 5.5. Radiation pattern at 2.39 GHz for both polarization states (x-plane).

polarized state the antenna exhibits RHCP on the front side (main radiation direc¬

tion) and LHCP on the back side. A cross polarization level (between RHCP and

LHCP) below -20 dB is achieved for broadside direction. In the linearly polarizedstate the antenna exhibits horizontal (x-direction) polarization. The cross polariza¬

tion (y-polarization) level is 17 dB below the main polarization. Both polarization

states yield good polarization purity and smooth radiation pattern.

In the first measurement series an external dc-voltage source was connected to

the groundplane with a two-wire line. This created visible ripples in the radiation

pattern due to coupling of the radiated fields to this two-wire line. Therefore,

to avoid ripples external dc lines should not be used or an arrangement should be

found which does not couple to the radiated field. For all antennas with polarization

switching, batteries were used for providing the dc-bias.

The measured gain is 3.2 dBi for the linear polarization and 4 dBi for the circu¬

lar polarization, respectively. Figure 5.6 shows the simulated and measured axial

1The effect of the pin diode can be seen by comparing the perturbation size of the antenna

shown in figure 5.4 with and without pin diode. Without pin diode a stub size of 6 mmx 3.6 mm

is required to excite a proper circular polarization. With the pin diode the stub size reduces to

4.4mmx 3.2mm. Investigations on the stub shape showed that the effect is proportional to the

stub surface independently of the ratio between length and width.

x-pol. linear

y-pol. linear


ratio. The agreement is reasonable. A 3 dB-ARB of 4.2% and 3.4% are achieved

2.2

QSimulation

\ Measurement

t M I

fi \\ / *

0 \\ / /

Ü / /

* \ * f /y, \ \ / -'

•fi / /

« S //S J''

n i i i i i

2.3 24 2.5

Frequency [GHz]

2.6

Figure 5.6. Measured and simulated axial ratio for the circularly polarized state.

in measurement and simulation, respectively, which is sufficient to cover the whole

ISM-band at 2.4 GHz (Industrial Scientific Medical).

Input ImpedanceThe input impedance of the two polarization states differs considerably due to the

different nature of the excited modes. Figure 5.7 shows the simulated and mea¬

sured return loss and input impedance for both polarization states. In the circular

§10

t 20

I

30

^

i

\ V\ ;'. ."

\;V

-;.:::.ÜT

2 2.2 2.4 2.6 2.8

Frequency [GHz]

••• linear pol simulation

— circular pol. simulation

linear pot measurement

circular pol. measurement

Figure 5.7. Measured and simulated input impedance of the two polarization states.


polarized state two resonances are excited, which translates into two resonant cir¬

cles in the smith chart. In the linear polarized state only one resonance is excited.

At the operating frequency around 2.4 GHz it is not possible to achieve a good

matching (better 10 dB) for both polarization states at the same time. The an¬

tenna was optimized for a good matching in the circularly polarized state for which

a broad impedance bandwidth of more than 20% was achieved. In the linearly

polarized state the return loss does not exceed 8 dB. However the 7 dB impedance

bandwidth, which is acceptable for many applications, is wider than 30%.

Comparing the simulation with the measurement results shows a good agree¬

ment, especially around 2.4 GHz. HFSS uses a so called 'fast frequency sweep'

technique. This means, that only a few discrete frequency points are calculated.

An extrapolation from these results allows to generate the solution for any frequencyin the considered band. Furthermore the simulation program uses one single finite

element mesh for all frequencies. This mesh is optimized at one frequency, which

is usually chosen close to the operating frequency. The program refines the finite

element mesh at locations which have a higher field strength. Using one mesh for all

frequencies can be problematic in the case where multiple resonances exist whose

fields reside in different parts of the structure. In this case the mesh is optimizedfor the resonance at the meshing frequency, but is not well suited for the calculation

of resonances at other frequencies. Therefore the best match between simulation

and measurement is usually achieved around the meshing frequency which can also

be noted in figure 5.7, where the meshing frequency was 2.4 GHz.

5.2.5 Antenna With Polarization Switching Between RHCP

And LHCP

The concept of the demonstrated polarization switching between linear and circular

polarization can easily be extended to polarization switching between the two cir¬

cular polarizations. Figure 5.8 shows the corresponding antenna setup, which was

derived from the previously presented antenna.

This time four perturbations, each being equipped with a switching diode, are

employed. In figure 5.8 the perturbations and the according diodes are numbered 1

to 4. While diodes 1 and 3 are in the forward biased state (shorted) diodes 2 and 4

are in the reversed biased state (open) and vice versa. Shorting diodes 1 and 3 (andconsequently leaving diodes 2 and 4 open) produces LHCP (front side). Shortingdiodes 2 and 4 and leaving open diodes 1 and 3 (by switching the applied dc bias

voltage) switches the polarization of the antenna to RHCP.

The required biasing architecture in this case becomes more complex. Four sep¬

arating slots running from the end of the perturbation to the edge of the ground-

plane are incorporated, which splits the groundplane in four different parts. The

diodes are mounted in a circular arrangement. Note that in contrary to the pre¬

vious mounting the pin diodes of the oppositely located perturbations are oriented

in opposite direction. Figure 5.9 shows the equivalent dc bias circuit. The same

dc-voltage has to be applied to the groundplane parts which lie opposite to each



other. This is realized by connecting them with a dc-connection line running alongthe edge of the groundplane as shown in figure 5.8.

Between the dc-connection line and the groundplane unwanted resonances could

be excited. To avoid coupling between the radiating fields and these unwanted

resonances the dc-line is shorted rf-wise to the groundplane by mounting large

shorting capacitors between the line and the groundplane. First measurements

without shorting capacitors showed visible ripples in the far field and intolerable

additional resonances in the impedance behaviour.

Vn

rits

•ii 4d_ c.L± 4_b

V0=1.5V, Cs=47pF, RS=180C1 Diode: C-O.lSpF, Rs=2.S£l

Figure 5.9. Equivalent biasing circuit. (Diodes are numbered 1 to 4 as shown in figure

5.8)


Figure 5.10 shows the radiation pattern of the x-plane in the two polarization states.

The y-plane radiation pattern is very similar. The left graph shows the case where

diodes 1 and 3 are open and diodes 2 and 4 are shorted, resulting in RHCP radi¬

ation to the front side. The right side shows the opposite polarization state when

diodes 1 and 3 are shorted and 2 and 4 are open. Switching the stubs corresponds

geometrically to an antenna rotation of 180 degrees around the y-axis. Looking


RHCP

-9(f

4?y/ \J5°

i À 0&/ 1 V / X

^-2MB\ \c\-10dB \

1 1 '\ \ / \ J v' t

1 \ ' \ \ x /\ \ * \ \ y y\ \ ' V x J

\ •\ /

Àuiiiiiiy^ i

X »,

"^"""^X /X /

'

\ l/ x. x^ \l

f

*

18F

18CP RHCP

LHCP

Figure 5.10. X-planc radiation pattern of the two circularly polarized states at 2.4 GHz.

at the radiation patterns it can be seen that the patterns of one polarization state

corresponds to the rotated (180 degrees) pattern of the other polarization state.

The measured antenna gain is 4 dBi, which is the same value as for the previously

presented antenna.

Figure 5.11 shows the measured and simulated broadside axial ratios for both

polarization states. A good agreement between measurement and simulation is

-

»;.' —i 1 —r-

////'\ '»» 1 / 1

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X*7\y'

RHCP sim.

RHCP meas.

LHCP sim.

LHCP meas.

2.2 2.3 2.4 2.5

Frequency [GHz]

2.6

Figure 5.11. Simulated and measured broadside axial ratio for the two polarization states,

found. The frequency of the minimum axial ratio and the value of the minimum


axial ratio predicted by the simulation for the two polarization states are slightly

different. This is due to the asymmetry introduced by the radiation of the slotline

feed and the slotline stub.

For the RHCP a 3 dB-ARB of 4.2% is simulated and 4.3% are obtained in the

measurement. For the LHCP 4.1% are predicted and 3.4% are measured.

Input Impedance

For the antenna with switching between linear and circular polarization (presentedin section 5.2.4) it was not possible to achieve 10 dB matching for both polarizationstates simultaneously, due to the different nature of the excited modes. For the

antenna presented in this section an excellent matching can be achieved for both

polarization states because the latter exhibit the same impedance behaviour due

the same nature of the excited modes. In both polarization states two orthogonal

linear modes are excited. The switching affects the phase relation between the

two orthogonal linear modes but not the input impedance. Figure 5.12 shows the

simulated and measured results for both polarization states. An excellent agreement

between measurement and simulation can be noted. In all cases a 10 dB-impedance

bandwidth of more than 35% is achieved.

RHCP simulation LHCP simulation

RHCP measurement LHCP measurement



5.2.6 Slot-Coupled Annular Patch Antenna With Polariza¬

tion Switching Between Linear and Circular Polariza¬

tion

Antenna SetupFor various applications single sided radiation is preferred over double sided radi¬

ation. The slot coupled annular patch antenna, which was introduced in chapter

4.3, is a suitable solution for this. The annular slot is no longer the radiatingelement but determines the polarization of the radiated fields. The switching cir¬

cuit can therefore be integrated in the slotline structure (located on the antenna

groundplane), which leads to a polarization switching architecture for single-sided

radiators. Figure 5.13 shows the antenna setup including the main dimensions of

a slot-coupled annular patch antenna with switchable polarization between linear

polarization and LHCP. The groundplane of the antenna is equivalent to the dou-

Figurc 5.13. Antenna setup

ble sided radiating structure presented in section 5.2.4 including the same biasingcircuit. The annular patch is located 4 mm above the groundplane. Between the

patch and the groundplane Rohacell was used. For mass production however an alu¬

minum patch produced with a cheap stamping technique and mounted with plastic

distance holders could be used. The main antenna dimensions are indicated.

Radiation Pattern

Figure 5.14 shows the radiation pattern for both polarization states. In the linearly

polarized state the main polarization is x-polarized. The cross polarization (y-

polarization) is about 15 dB below the main polarization. A front to back ratio of

6 dB was achieved. A gain of 5.5 dBi was measured for the linear polarized state,

which corresponds to a 2 dB gain increase compared to the double sided radiatingantenna presented in section 5.2.4. Increasing the size of the patch relativ to the

coupling slot would increase the gain but reduce the ARB (chapter 4).


linear polarization circular polarization

0°

180° 180°

x-polarized

y-polarized

RHCP

LHCP

Figure 5.14. X-plane radiation pattern of the two polarization states at 2.37 GHz.

On the right side of figure 5.14 the radiation pattern in the circularly polarized

state is shown. The antenna radiates LHCP to the front side and RHCP to the

back side. A gain of 6.1 dBi and a front to back ratio of 4 dB is measured. As in

the linearly polarized case, the antenna radiates most of its energy to the front side

but in terms of isolation is not well shielded to its back side. A gain increase of

2 dB compared to the double sided radiating antenna is achieved.

Figure 5.15 shows the simulated and measured broadside axial ratio in the cir¬

cularly polarized state. The simulation predicts an 3 dB-ARB of 2.9% whereby the

measurement yields 3.8%. The measured ARB is only 10% smaller than the one

achieved for the double sided radiating structure presented in section 5.2.4. The

back side radiation could be strongly reduced by using a larger patch or a cavity

backing .This however would reduce the ARB significantly. As explained in section

4.3 a trade off exists between back side radiation level and ARB. In this examplethe focus was put on a large ARB. Using larger patch dimensions a front to back

ratio of 17 dB and an ARB of 2.7% was obtained.

Input ImpedanceThe simulated and measured input impedance are shown in figure 5.16. A good

agreement between measurement and simulation is found. In order to achieve a good

matching for both polarization states the matching circuit was designed such that in

the smith chart the impedance curves for both polarization states are approximately

equidistant to the smith chart center. Around the frequency of the minimum axial

ratio (at 2.37 GHz) the return loss for the circularly polarized state is 13 dB and for


a•fi

/ *

g t

\ * Simulation /t

\ VMeasurement

itft

11

\>\l

\li*

\% #>

\l //

/*

/'ft

\/ *

K / *

A. £ *

A. / *

*\ / /* \ / *

\ ^-~>>* /

1 1 i*i i i

2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55

Frequency [GHz]

Figure 5.15. Measured and simulated axial ratio for the circularly polarized state.

2.2 2.4 2.6 2.

Frequency [GHz]

linear pol. simulation

circular pol. simulation

linear pol. measurement

circular pol. measurement


the linearly polarized state 8 dB. In the circularly polarized state a 10 dB-impedancebandwidth of 17% has been achieved. In the linearly polarized state the antenna

yields a 10 dB-impedance bandwidth of 9%, which is however shifted above the

operating frequency of 2.37 GHz.


5.2.7 Summary

An antenna architecture allowing polarization switching between linear and circular

polarization or between the two circular polarization states has been introduced.

Double and single sided radiation can be produced. Although a low-cost and lossy

FR4 substrate is used an antenna gain of 4 dBi in the double sided radiating case and

a gain of 6.1 dBi in the single sided radiating case were achieved. The biasing circuit

for the pin diodes is directly integrated on the slot layer using few elements. The

achieved ARB is sufficient to cover the ISM band. Considering the broad impedance

and axial ratio bandwidth and the potential for low piece cost the proposed antennas

are interesting candidates for commercial wireless communication systems requiring

polarization diversity.

5.3 Microstrip Switching Architecture

5.3.1 Switching Principle

The attractiveness of the previously presented uniplanar switching architecture lies

in its simplicity. It's potential, however, is limited by the fact that it is difficult to

make the diodes individually switchable, which limits the diversity to two polariza¬tion states. In order to enable individual switching of each diode two separating

slots would be needed for each single diode, which would rapidly increase the com¬

plexity of the biasing circuit.

A second drawback of the previously presented design is the fact that the antenna

can not be well matched for both, the linearly and the circularly polarized state due

to the different nature of the excited modes. This can be overcome by introducing

a tuning option into the matching circuit in order to adapt the input impedance for

the different polarization states. The impedance tuning is realized by the mounting

of a pin diode across the slot line stub of the microstrip-to-slot line transition,

allowing to vary the length of the slot line stub.

Based on this feature a new antenna architecture enabling the individual switch¬

ing of each diode was developed. Figure 5.17 shows the antenna slot including the

placement of the switching diodes. For a better overview, the diodes are numbered

from 1 to 5. Diode 1 is used to tune the input impedance for the different polariza¬tion states (linear and circular). With diodes 2 to 5 the four slot ring perturbations

can be switched individually. Out of the 32 possible diode state configurations,

only four are of practical interest. The 4 diode states, the corresponding far field

polarization and resonance frequency are shown in table 5.1.

Leaving all diodes open or shorted, the antenna exhibits linear polarization (x-

polarization) having a resonance frequency at 2.44 GHz and at 2.1 GHz, respectively.

Shorting diode 2 and 4 produces RHCP whereas shorting diodes 3 and 5 producesLHCP.



Diode state1 Polarization Resonance

Dl D2 D3 D4 D5 frequency

s s s s s linear x-pol. 2.44 GHz

0 o o o o linear x-pol. 2.10 GHz

0 o S 0 s LHCP to front side 2.44 GHz

0 SOS 0 RHCP to front side 2.44 GHz

o : open; s : shorted

Table 5.1. Diode states and corresponding far field polarization and resonance frequencies.

5.3.2 Biasing Architecture

In order to switch 5 diodes individually the dc bias can not be applied directly

through the groundplane because too many groundplane separating slot would be

required. Each diode would require a dc wise isolated groundplane part, thus two

separating slots per diode would be needed, which means at least 10 separating slots

in total. Because this is not pratical a different biasing circuit is used including five

microstrip lines located on the opposite side of the substrate. For each diode the

dc-bias is fed individually between the antenna groundplane and the correspond¬

ing microstrip line. In this case the diode is mounted across the narrow slot line

connecting the antenna stub to the annular slot. On one side of the slot line a

small island of of groundplane is dc-wise separated from the rest of the groundplaneas shown in figure 5.18. This island is connected by a via to the dc-carrying mi¬

crostrip line on the opposite side of the dielectric. Rf-wise the island is shorted to


Exploded view Top view

shorting capacitor switching diode

Figure 5.18. Dc-bias architecture.

ground using a large shorting capacitor. This architecture works well for multiple

individually switchable diodes.

The biggest problem of this architecture is the excitation of unwanted parasitic

modes. The antenna field can couple into parasitic modes running between the dc-

feed line and the groundplane. These parasitic modes deteriorate the polarization

purity and affect the input impedance. They can be suppressed by shorting the dc

bias lines rf-wise with large shorting capacitors. A numerical study using HFSS has

shown that the shorting capacitors must be spaced no more than | apart in order to

efficiently short circuit any possible parasitic resonance [74]. Consequently the prize

to pay for the achieved switching flexibility compared to the previous architecture

is the increased complexity of the biasing network.

It has been tried to use (narrow) high impedance dc-feed lines to avoid coupling.It turned out however, that the deterioration was still present with the 0.4 mm-thick

(120 ft) dc feed line.

An often used method to suppress rf signals on dc feed lines are microstrip

stepped impedance filters [75]. However, for the biasing circuit presented in this

section they are not practical because they would require too much space. The

available space for the dc lines is not large enough to incorporate five stepped

impedance filters.


5.3.3 ASA With Switchable Polarization Between Linear

Polarization, LHCP and RHCP

Antenna setup

Figure 5.19 shows the antenna setup. For this antenna a duroid 5880 substrate

with a thickness of h — 0.79 mm (er = 2.2, tanÔ — 0.0002) was used. One side

of the dielectric contains the groundplane including the slot circuit, the switching

diodes and the shorting capacitors for the dc islands. The opposite side contains the

microstrip rf-feed, the microstrip dc-feed lines as well and the shorting capacitors

for the dc-lines (not shown in the figure).The dc-voltage is supplied by a 1.5 Volt coin-cell battery, which is directly

mounted on the edge of the antenna. The negative pole of the battery is connected to

the antenna groundplane. A five element dip-switch is mounted between the positive

pole of the battery and the dc-feed lines allowing to switch each line individually.

On each dc-feed line a series SMD resistor of 180 ft is mounted in order to limit the

diode current.

For this antenna a new pin diode was used, namely the BAR89-02L silicon pin

diode from Infineon Technologies. This diode exhibits similar electrical specifica¬

tions as the diode used for the antennas presented in section 5.2 with lower costs.

It is packaged in a TSLP-2-1 housing having dimensions of 1 x 0.6 x 0.4 mm. The

most important electrical diode characteristics are summarized in table 5.2. Special

BAR89-02L Diode Parameter Symbol Value

Diode capacitance reverse bias V^, = 0 V

Diode resistance forward bias Ip=4.4mA

Series inductance

Cr 0.18 pF

R; 1.2 ft

Ls 0.4 nH

Table 5.2. Pin diode characteristics of BAR89-02L.

care has to be taken for the shorting capacitors. Capacitors with large values in the

nano-farad range exhibit important losses in the GHz-range. In the first designs

shorting capacitors with a value of 4.7nF were used. They turned out to be im¬

practical because they suffer from important losses. Significant differences between

the simulation (assuming no losses) and the measurement were found. The use

of 100 pF-capacitors, exhibiting lower losses, showed a better agreement between

simulation and measurement2. Measurements at 2.5 GHz yielded an insertion loss

of 0.15 dB in the case of the 100 pF-capacitor and 0.8 dB in the case of the lnF-

capacitor.

2Even if the capacitor losses can be included in the simulation they are not acceptable, because

they produce a loading effect exciting unwanted modes. It is important to produce low-loss short

circuits in order to excite the proper antenna modes.


antenna setup

slot circuit

including lumped

elements

vias

duroid 5880

h = 0.79 mm

er = 2.2

dc feed circuit

and

rf microstrip feed

Figure 5.19. Exploded view and top view of the antenna. (Shorting capacitors of dc-lines

are not shown)



Figure 5.20 shows the simulated and measured far field characteristics at 2.44 GHz

for the different polarization states. Figure 5.20 a) and b) show the x-plane radiation

patterns of the two circularly polarized states. A good agreement can be seen

between measurement and simulation for the RHCP state. For the LHCP state

two major discrepancies occur. The simulation shows the same gain for front side

(LHCP) and back side (RHCP) whereas the measurement yields a 1.5 dB higher

back side gain. Furthermore, the cross polarized radiation to the front and the

back side is significantly higher than predicted by the simulation, especially in the

sector between 0 and -50 degrees and 180 and -135 degrees, respectively. This

phenomenon arises in both circular polarization states but is more pronounced in

the LHCP case.

Figure 5.20 d) shows the simulated and measured broadside axial ratios. The

simulation yields the same axial ratio behaviour for both circularly polarized states

with an ARB of 4.4% whereas the measurement yields 4.5% for the RHCP case and

1.5% for the LHCP case, respectively. For the RHCP state the agreement between

measurement and simulation is good in terras of ARB and the axial ratio minimum

(below ldB). A slight frequency downshift of 0.7% is observed, however, in the

measurement. For the LHCP case the measurement shows significant differences

compared to the simulation. The measured axial ratio minimum is 2.3 dB (in simu¬

lation 0.4dB), the 3dB-ARB is 1.5% (in simulation 4.4%) and the frequency of the

axial ratio minimum shows a downshift of 1% with respect to the simulation.

An antenna gain of 5.0 dBi is achieved for both circularly polarized states in

the simulation. The measurement yields 4.2 dBi for the RHCP state and 3.2 dBi

for the LHCP state. The gain difference between measurement and simulation

can be explained with the higher backside radiation level seen in the measurement

(lowering the maximum gain) and the non modelled losses of the shorting capacitors,

which are small but non negligible. The gain difference between the two circularly

polarized states however is difficult to explain, because the impedance match is goodfor both cases and the same switching circuitry is involved. One possible source for

this problem could be measurement error or the slightly asymmetrically placed feed

line (with respect to the groundplane), which could act as a parasitic radiator.

The measured and simulated linearly polarized radiation pattern at 2.44 GHz

for the case where all diodes are shorted is shown in figure 5.20 c). The main po¬

larization is the x-coordinate and the cross polarization is the y-coordinate. The

cross polarization level is -18 dB. A fairly good agreement between simulation and

measurement is seen. However, the same discrepancies with respect to the simula¬

tion seen for the LHCP case appear, namely a ldB lower front side gain than back

side gain and a higher cross polarization level at angles between 0 degrees and -50

degrees and 180 degrees and -135 degrees, respectively. The simulation predicts an

antenna gain of 5.0 dBi. The measurement yields 3.5 dBi.

Leaving all diodes open instead of shorting them produces a linearly polarizedradiation with a resonance around 2.1 GHz instead of 2.44 GHz, with basically the

same radiation properties. The downshift of the resonance frequency is due to the


increase of the electrical antenna size. In fact, the perturbations act as a loading

of the antenna, which increases its electrical size compared to the same antenna

without perturbations.

a) RHCP b) LHCP

180°

RHCP simulation

LHCP simulation

180°

RHCP measurement

LHCP measurement

c) linear polarization d) Axial ratio

n," 1

f i j\ i /tS t fjI ." / * '

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CQ "' *\ //•'a. 6 \*\ ' >/-'Q

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fci/ f*

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0

*•*

*<4

! /i /i /

180°

2.3 2.4 2.5 2.6

Frequency [GHz]

x-polarized simulation

y-polarized simulation

x-polarized measurement

y-polarized measurement

LHCP simulation

RHCP simulation

LHCP measurement

RHCP measurement



Input ImpedanceFigure 5.21 shows the simulated and measured input impedance and return loss

for the four different diode state constellations shown in table 5.1. Figure 5.21

a) shows the input impedance for the two circularly polarized states and figure

5.21 b) shows the input impedance for the two linearly polarized states (all diodes

shorted and all diodes open). For all cases the agreement between measurement

and simulation is fairly good. The most important disagreement can be seen for the

linearly polarized case, where all diodes are shorted. Compared to the simulation

the measured resonance frequency (2.4 GHz) is downshifted by 2% compared the

the simulated one (2.44GHz).Due to the tuning switch (diode 1), the antenna can be well matched for linear

and circular polarization at the same operating frequency of 2.44 GHz. A return

loss better than 15 dB is achieved for the three different polarization states and

the 10 dB-impedance bandwidth for the two circularly polarized states is better

than 30%, which is much higher than the ARB. The impedance bandwidth for the

linearly polarized case is 13% for the case where all diodes are shorted and 14% for

the case when all diodes are open. Together the two linearly polarized states3 cover

an 10 dB impedance bandwidth of 25%.

Analysis of discrepancies between simulation and measurement

Different discrepancies between measurement and simulation were observed. For

the RHCP-state the best agreement of all polarization states between measurement

and simulation could be observed. The most important differences observed in the

measurement were a 0.8 dB lower gain and a 0.7% downshift of the frequency of the

axial ratio minimum. For the LHCP state more important differences were observed

in the measurement. A 1.5 dB lower gain, a higher cross polarization level, a higher

axial ratio and a front side gain being 1.5 dB lower than back side gain. The higher

cross polarization level and the lower front side gain were observed as well for the

linearly polarized state.

The identification of the origin for the difference between simulation and mea¬

surement is not obvious. To exclude substrate imperfections the high-quality (low

loss, stable permittivity) duroid 5880 substrate was chosen. Another error source

could be the pin diodes. To find it out the antenna (figure 5.19) was built up with

higher quality beam lead pin diodes (used in section 5.2.4). However the same

discrepancies between measurement and simulation were found, which excludes the

pin diodes as error source. A further error source could be losses in the short¬

ing capacitors. The first fabricated antenna (figure 5.19) was equipped with InF

shorting capacitors. As mentioned earlier in this section, the InF-capacitors ex¬

hibit significantly higher losses then the finally used lOOpF-capacitors. Using the

InF capacitors produced a downshift of the frequency of the minimum axial ratio

and in an increase of the axial ratio itself. Furthermore a significant disagreement

3Note that both linear polarized states exhibit x-polarization but have different resonance

frequencies. It is not possible to produce y-polarization with the chosen feeding method.


o>) Circular Polarization

as

10

20

30-

4/.'X•w

V ffl

\\ * If \V *. * /'

w\7».'

It /

\*y

2.2 2.4 2.6

Frequency [GHz]

2.8

RHCP simulation

LHCP simulation

RHCP measurement

LHCP measurement

b) Linear Polarization

CO

I

10

20

30

^*"—*"r"

>Nv -'..••'

\ ^vS -*t

\ iï /

\ * /

\ //

\ : I11 1*1

M1 ll /*

1 /- '1/.Is1V

AS 2.4 2.6 2.8

Frequency [GHz]

diodes shorted simulation diodes shorted measurement

diodes open simulation diodes open measurement

Figure 5.21. Measured and simulated input impedance of the linear and circular polar¬

ization states.

between simulation and measurement could be observed for the return loss. The

replacement with the lOOpF-capacitors clearly improved the measured results.


A closer look at the return loss of the linearly polarized states (figure 5.21 b) )shows that the agreement between measurement and simulation is clearly better

for the case where all diodes are open. In the case where all diodes are shorted

(simulated resonance at 2.44 GHz) the measured resonance frequency (2.4 GHz) is

downshifted by 2% compared to the simulation. This indicates, that the losses

produced in the shorting capacitors, bridging the groundplane islands with the rest

of the groundplane, affect the antenna behaviour. In the shorted state the current

flowing through these capacitors is higher than in the open state and consequently

the capacitor losses affect the antenna behaviour more clearly, which can be seen

from the return loss curves.

The losses in the shorting capacitor do not explain the differences observed

between the RHCP state and the LHCP state. In both states, the same number

of diodes are switched on and therefore the losses and radiation properties (gain,

ARB) should be similar. Although the entire antenna structure including dc-feed

lines, shorting capacitors and the battery holding structure were modeled in the

simulation, the differences seen in the measurement could not be reproduced. One

possible source of error could be the feeding line, which is used in the measurement,

but not modeled in the simulation.


5.3.4 Annular Slot-Coupled Circular Patch Antenna With

Switchable Polarization Between Linear Polarization,

LHCP and RHCP

Antenna SetupUnidirectional radiation can be achieved by coupling the annular slot fields to a

circular patch located above the slot. This was explained in chapter 4.3. The po¬

larization switching architecture can still be integrated in the slot circuit although

the radiation arises from the microstrip patch mode. Figure 5.22 shows the an¬

tenna setup of the slot coupled patch antenna allowing to switch between linear

polarization, LHCP and RHCP. Exactly the same antenna setup as for the double

sided radiating ASA is used except an additional circular patch and a 4 mm-thick

Rohacell substrate separating the patch and groundplane. The antenna is designed

for the operating frequency (mimimum axial ratio) of 2.44 GHz. The circumference

of the slot is reduced to 0.87 A instead of 1A in the case of the double sided radiating

structure. For a chosen slot size, different patch sizes can be used as explained in

chapter 4.3. A small patch size was chosen here, resulting in a large ARB but a

poor front to back ratio.

From a production point of view the single sided radiating ASA and the double

sided radiating slot coupled circular patch antennas are very similar. The produc¬

tion difference consists in the mounting of the patch, which is a minor additional

production step.


Figure 5.23 shows the simulated and measured radiation characteristics of the three

different polarization states. Figure 5.23 a) + b) show the radiation patterns of the

two circularly polarized states and figure 5.23 c) shows the radiation pattern of the

linearly polarized state. A good agreement between measurement and simulation

was achieved for the main lobe of the radiation pattern. The measured cross po¬

larization level is significantly higher than predicted by simulation. The antenna

exhibits a good polarization purity for the RHCP and the linearly polarized state

(cross polarization 18 dB down). In the LHCP case a cross polarization level of only

15 dB (corresponding to an axial ratio of 3 dB) was observed in the measurement.

The simulation yields a gain of 7 dBi and a front to back ratio of 3.5 dB for all

polarization states. In the measurement a gain of 6.1 dBi is measured for the RHCP

and a gain of 5.0 dBi for the LHCP and the linear polarization, respectively. Hence

a gain increase of roughly 2 dB is achieved over the double-sided radiating antenna

introduced in section 5.3.3. The measured gain difference between the RHCP and

the LHCP as well as the linear polarization are not predicted by the simulation and

must therefore be due to some non-ideal behaviour of the lumped elements. The

same discrepancies were observed for the double-sided radiating structure of section

5.3.3.

The simulated and measured broadside axial ratio of the two circularly polarized

states is shown in figure 5.23 d). The simulation shows an 3 dB-ARB of 3.8% for the


antenna setup

circular patch

slot circuit

including lumped

elements

duroid 5880

h = 0.79 mm

e. = 2.2

dc feed circuit

and

rf microstrip feed

100

series resistances


RHCP state and 3.4% for the LHCP state, respectively. The measurement yields

3.5% for the RHCP and 0% for the LHCP state. For the LHCP state, the minimum


axial ratio is slightly above 3 dB.

Compared to the double sided radiating structure (introduced in section 5.3.3)

the annular slot coupled patch antenna, presented in this section shows a 2 dB

higher gain and a 25% reduction of the 3 ARB (simulation).

-Si

a) RHCP

o°

'4JT/

^>S0jIe\

\I5°

1 1 \ -"/^v

'/M'\-10dB\

\ \ !/ >A*

-jjK^x^.^ 135°

180°

OdB

90° SO"

b) LHCP

RHCP simulahon

LHCP simulation

180°

RHCP measurement

LHCP measurement

1 "/ \/"^ \-20ßB N! ! Y,

( AwSjy\ /KlOdB \

\ \ 1; ' \y V#\\/ ji x A /

-13^\^^ .t.j>£$f^l3F

'dB

90°

c) linear polarization

o°

-4sy

180°

x-polanzed simulation

y-polanzed simulation

x-polanzed measurement

y-polarized measurement

d) Axial ratio

a

"S

2.4 2.5

Frequency [GHz]

LHCP simulation

RHCP simulation

RHCP measurement

LHCP measurement

2.6



Input Impedance

Figure 5.24 shows the measured and simulated input impedance and return loss

of the two circularly polarized and linearly polarized states. For the circularly

polarized states shown in figure 5.24 a) a very good agreement between measurement

and simulation was found. The return loss of the linearly polarized states (shown in

figure 5.24) shows similar disagreements as in the case of the double sided radiating

structure (section 5.3.3). The return loss measurement for the state where all diodes

are shorted shows a downshift of the measured resonance frequency of 2% compared

to the simulation.

The antenna yields a 10 dB-impedance bandwidth of 21% for both circularly

polarized states and a 10 dB-impedance bandwidth of more than 12% for the two

linearly polarized states. Together they cover a band going from 2.1 GHz to 2.6 GHz.

A broad impedance band can be covered for all polarization states.

Analysis of discrepancies between simulation and measurement

The same discrepancies between measurement and simulation are observed as in

section 5.3.3. A good agreement is found for RHCP whereas LHCP and linear

polarization show an increased cross polarization level an reduced gain (comparedto RHCP).

5.3.5 Summary of Results

Table 5.3 summarizes the simulation and measurement results of the most important

antenna key figures for the four different polarization states. Several conclusions

can be drawn.

The antenna architecture allows switching between 3 different polarization states.

A good polarization purity can be produced in the simulation. The measurement

shows a good purity for the linear polarization and the RHCP. A broad impedance

matching bandwidth can be achieved for all polarization states due to the switchable

stub in the feed line. By coupling the slot fields to a patch a single sided radiation

with switchable polarization can be produced using the same switching architecture

as for the double sided radiating structure. This demonstrates the flexibility of this

architecture.

The HFSS simulation shows a very good agreement with the measurement for

the input impedance and the radiation characteristics of the RHCP state. For

the LHCP state and the linearly polarized state major discrepancies occur. The

front side gain is about 1 dB lower than predicted and the measurement shows a

significantly higher cross polarization level for the LHCP and the linearly polarized

state. The cause of this discrepancy could not be identified. One cause could be

measurement errors.

The annular slot -coupled circular patch antenna shows a 2 dB higher gain than

the double sided radiating ASA. The ARB and the impedance bandwidth however

are reduced by 23% and 35%, respectively. The 3 dB-ARB of the double sided

radiating ASA covers the entire ISM-band. In the case of the slot-coupled patch

antenna an axial ratio lower than 4 dB can be assured over the entire ISM-band.


a) Circular Polarization

u

10*NI )t>

O

J

20

30

\t/

V A*V*"*. /i-»

^YV'f'iCH

40

\ V.i ii i\ i *.'

S.S S.^ 2.6

Frequency [GHz]

RHCP simulation

• LHCP simulation

RHCP measurement

• LHCP measurement

b) Linear Polarization

'.2 2.4 2.6 2,

Frequency [GHz]

diodes shorted simulation diodes shorted measurement

diodes open simulation diodes open measurement

Figure 5.24. Measured and simulated input impedance of the four polarization states.

This is sufficient for most applications. Due to the bandwidth, the flexibility and

the simplicity this antenna architecture is very interesting for commercial wireless

communications. Additional switching diodes can be used without increasing the

complexity of the architecture.


5.4 Chapter Conclusion

In this chapter two different reconfigurable antenna architectures based on an an¬

nular slot antenna have been introduced. Slot perturbations can be switched on

and off with the use of a pin diode in order to change the antenna polarization.

The annular slot can be either used as a resonant radiator yielding a double sided

radiation pattern or it can be used in conjunction with a resonant patch yielding a

single sided radiation pattern. In both cases the polarization is determined exclu¬

sively by the annular slot and the switching circuit, which is integrated on the slot

layer.

In a first architecture the dc-bias for the pin diodes is directly applied through

the groundplane, on which the slot antenna is realized. A narrow slot separates the

groundplane in two parts for the dc signal. Rf-wise the two groundplane parts are

RHCP

gainARB

impedance BW

front to back ratio

LHCP

gainARB

impedance BW

front to back ratio

ASA

sim. meas.

Circ. Patch

5.0 dBi

4.4%

32%

5.0 dBi

4.4%

32%

4.2 dBi

4.5%

31%

3.5 dBi

1.5%

31%

linear polarization (all diodes shorted)

gain 5.0 dBi 3.5 dBi

impedance BW

front to back ratio

13 % 13 %

linear polarization (all diodes open)

gain 4.8 dBi 3.3 dBi

impedance BW

front to back ratio

14 % 14 %

sim. meas.

7.0 dBi 6.1 dBi

3.8 % 3.5 %

21 % 21 %

3.5 dB 3.5 dB

7.0 dBi 5.0 dBi

3.4 % 0 %

21 % 21 %

3.5 dB 2.4 dB

7.0 dBi 5.0 dBi

12 % 14 %

3.5 dB 2.4 dB

6.8 dBi 4.9 dBi

13 % 14 %

3.5 dB 2.4 dB

Tabic 5.3. Measurement and simulation results for the different polarization states of the

two antennas.


connected to each other with a shorting capacitor. Switching between linear and

a circular polarization and switching between the two circular polarizations was

demonstrated. In the case of the switching between the two circular polarizations

a good impedance match could be achieved for both polarization states whereas in

the case of the switching between linear and circular polarization a good matching

could only be achieved for one polarization due to the different nature of the ex¬

cited modes. The biggest advantage of this first architecture is its simplicity. The

drawbacks are that the perturbations can not be switched individually but have to

be switched in pairs. Furthermore switching between more than two polarization

states is impractical to realize because the complexity of the biasing architecture

would increase dramatically.

To alleviate this problem a second architecture allowing individual switching of

pin diodes was developed. The bias voltage for each diode is fed through a mi¬

crostrip line lying on the opposite side of the substrate. The slot antenna comprises

four switchable perturbations allowing to control the radiation characteristics and

one switchable slot line stub allowing to match the input impedance for the dif¬

ferent polarization states. Due to the increased switching flexibility, three different

polarizations can be produced with the same antenna. Furthermore, a good impe¬

dance matching can be achieved for all three polarization states with the help of an

impedance tuning switch. The higher functionality of this architecture was payed

for with an increased complexity of the bias circuit. Via holes, shorting capacitors

and dc-voltage switches are required.The two architectures are interesting candidates for commercial wireless appli¬

cations. Depending on the system requirements, they offer a low-cost solution with

reduced functionality or an increased functionality requiring a more complex and

therefore more expensive architecture.

Chapter 6

Conclusion

Uniplanar circuits exhibit several advantages over microstrip circuits. They

consist of only one metalization layer. In contrary to microstrip circuits, lossy and

therefore cheap substrates can be used without suffering important losses due to

the fact, that only a small part of the electromagnetic field resides in the substrate.

Lumped elements are easily mounted in series and shunt configuration without the

need of expensive via holes. Due to the above mentioned properties uniplanar ar¬

chitectures can be produced at lower costs than microstrip architectures, which

makes them interesting for commercial applications. Using a coplanar architecture,

active circuits can be easily integrated together with the radiating structure. Po¬

larization diversity has gained more importance in recent years because it allows

to increase the capacity of wireless communication channels. Antennas for modern

commercial wireless communication systems should exhibit low production costs

and polarization diversity.

In this thesis several aspects of printed uniplanar or quasi-uniplanar annular

slot antennas were investigated systematically in order to show their potential for

commercial wireless applications. The focus was hereby on low-cost solutions and

on polarization diversity solutions. The annular slot ring was chosen as radiating

element because it exhibits two orthogonal resonances which allows to produce

different polarizations with one single feed. Furthermore its shape does not favour

any polarization direction, which is important for reconfigurable antennas.

The work done in this thesis can be classified into three major parts. In part

one a systematic study of circularly polarized single fed ASA's was done. The

influence of different perturbation shapes, slot widths and substrate materials on

the ARB, the input impedance and the antenna size was investigated. Different

feeding methods were introduced and several examples were realized and comparedwith simulation results. Furthermore it was shown, that various perturbation types

can be used in order to excite circular polarization. The low-cost potential of the

ASA was clearly demonstrated with two antennas built on cheap substrates. A

cpw-fed ASA built on FR4 substrate and a coaxial line fed ASA not requiring any

specific substrate.

102 Conclusion

Slot antennas exhibit inherently double-sided radiation. For many applications

however single sided radiation is preferred or even compulsory. In a second part of

this thesis two different architectures allowing to achieve single sided radiation with

slot antennas have been investigated.The first technique consists in backing the antenna with a metal cavity in order

to redirect the back side radiation to the front side. Different cavity-backed slot

antennas have been treated in literature so far, however the influence of cavity

backing on slot antennas compared to non cavity backed antennas has not been

studied so far. In this thesis the focus is put on this aspect. The investigation

showed that the cavity backing increases the gain by about 2 dB and reduces the

back side radiation efficiently. A front to back ratio of roughly 20 dB is achieved.

The ARB and the impedance bandwidth are reduced by roughly 50% compared to

the double sided radiating ASA.

The second investigated technique to produce single sided radiation consists in

coupling the slot fields to a patch resonator. Patch antennas have been studied

extensively in literature. The interest of the here presented approach however lies

in the fact that the polarization is not determined by the patch shape but by the

coupling slot shape. This is advantageous for the integration of active polarization

switching circuits. The active circuit can be realized in uniplanar technology and

the same circuit architecture can be used for the double sided radiating antenna

without patch and the single sided radiating antenna with patch. Dc-biasing is

much easier to apply on the groundplane than on the antenna patch.

Two different antenna types were investigated, the annular slot coupled circular

patch and the annular slot coupled annular patch antenna. A trade off has to

be found between the two figures. For a slot size where a front to back ratio of

17 dB is achieved the ARB and the impedance bandwidth is reduced by about 50%

compared to a single sided radiating ASA as it is the case for cavity backing. For

larger slot sizes the ARB and the impedance bandwidth are higher. It can be stated

that the slot coupled patch antenna is more broadband than the cavity backed ASA

but that it is not well isolated to its backside.

In the last part of this thesis two different polarization switching architectures

for ASA's were implemented. The polarization is changed with the help of switching

diodes, which are mounted across the slot ring perturbations. By shorting them or

leaving them open the polarization can be altered.

In a first very simple architecture polarization switching between linear and one

circular polarization and polarization switching between the two circular polariza¬

tions can be realized with a simple switching circuit requiring few lumped elements.

The dc-bias is directly applied through the antenna groundplane. Single and double

sided radiation characteristics can be produced using the same switching architec¬

ture. The architectures presents two major weaknesses. The polarization switching

is limited to two polarization states and in the case of the switching between linear

and circular polarization it is not possible to achieve a good impedance matching

for both polarization states.

A second more sophisticated switching architecture overcomes the deficiencies of

103

the first architecture at the price of a more complex and therefore more expensive

switching circuit. Each antenna perturbation can be switched individually and

in addition the input impedance can be matched for the linear and the circular

polarization state with the help of a switchable tuning stub. Two circular and one

linear polarization can be produced with the same antenna. Single and double sided

radiation characteristics can be produced using the same switching architecture.

Few polarization switching architectures have been published so far in literature.

The two novel solutions which are presented in this thesis are interesting candidates

for modern commercial integrated front ends. They allow to uses cheap substrates

without suffering from important losses. The electronics and the antenna can be

realized on the same substrate using either coplanar or microstrip technology. The

antennas exhibit a wide flexibility in terms of polarization diversity radiation char¬

acteristics. In the case of the second architecture allowing the individual switching

of pin-diodes many other applications than polarization switching can be imagined.

One could think for example of pattern diversity or switching between different

frequency bands.

Seite Leer /

Blank leaf

Chapter 7

Outlook

In this thesis different starting points for further work can be identified. From my

perspective I identified three major areas, which would increase the attractiveness

of the introduced antennas.

In this work the transition from the cpw to the slot antenna was realized with a

cpw-to-slotline transition as introduced in section 3.4.1. This is a quite cumbersome

way of exciting a slot antenna with a cpw. A rigorous study of coupling mecha¬

nisms between the cpw and slot antenna might provide more compact and flexible

architectures allowing to achieve this coupling.

A second interesting point meriting more research effort is the unidirectional

radiation properties. The here presented solutions show the drawback of either

exhibiting an important height (cavity backing) or a low front to back ratio. In

recent years a lot of effort has been spent in the development of high impedance

surfaces. Some examples have been shown, where by locating them on the backside

of slot antennas at a close distance, unidirectional radiation could be achieved. It

would be worth to investigate single sided radiating slot antennas employing these

high impedance surfaces. Especially in technologies like LTCC where, rather height

than complexity is expensive, they might also cost-wise add a clear benefit.

This thesis introduced reconfigurable antennas allowing to switch between differ¬

ent defined polarization states. In current and future applications it could however

be requested that the radiation properties are continuously adapted in order to

maximize data transmission. Therefore not switching between a number of defined

polarization states but continuous polarization tuning would be required. This could

be achieved by employing tuning elements rather than switching elements. The use

of varactor diodes could allow to tune the value of the antenna perturbations and

the matching stub. In this way both, the polarization and impedance match could

be continuously adapted to the environmental conditions.

Seite Leer /Blank leaf

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Diploma Thesis, ETH Zürich, Tech. Rep., 2003.

[75] D. Pozar, Microwave Engineering, 2nd ed. Wiley, 1998.

Glossary

Symbol Unit Description

ASA annular slot antenna

ARB % axial ratio bandwidth

CP circularly polarized

cps coplanar strip line

cpw coplanar waveguide

EBG electromagnetic band-gap

ISM industrial scientific medical band 2.4GHz-2.5GHz

LHCP left hand circular polarization

RHCP rigth hand circular polarization

Ao m free space wavelength

\ m guided wavelength

Acknowledgements

This work was carried out at the laboratory for field theory and microwave

electronics (IFH) of the Swiss Federal Institute of Technology (ETH) in Zürich,

Switzerland. During my work an entire team was backing me, which allowed me

to concentrate on the essential parts of my research. Here I would like to thank all

these people for their support.

First of all I would like to express my gratitude to my supervisor Prof. Dr.

Rüdiger Vahldieck. With his positive attitude, his open minded character and his

uncomplicated being he created a very agreeable work climate. He let me a lot of

freedom to go into the direction I wanted, was always available if I needed him and

encouraged me from early on to present my work on scientific conferences.

I would like to thank also Prof. Dr. Fred Gardiol, who accepted to be co-

examiner of this thesis. He taught me the basics of electromagnetics and was the

examiner of my diploma thesis.

During my research work at ETH I supervised several student projects, which

contributed to this thesis. Herein I would like to thank these persons, in particularly

Cristiano Pianezzi, Mischa Gräni, Marcus Jacob and Stéphanie Jarno.

The fabrication and measurement of radio frequency devices is a delicate task

and requires the support of skilled people. I was in the lucky situation to benefit

from the immense experience of the people at IFH. I would like to thank especially

Hansruedi Benedickter for his unconditional support in all aspects of fabrication

and measurement. Many thanks also to Martin Lanz, Claudio Maccio and Stephen

Wheeler.

Next to skilled people and good equipment, research requires an agreeable work

climate. I very much appreciated the people and the climate in our group. It was

a exciting time with fruitful discussions not only on electromagnetics but about

cultures, politics and numerous headlines of 'Bild'.

Finally I would like to thank my wife Sabrina for her comprehension and support

during this time.

Matthias Fries

Curriculum Vitae

Name: Matthias Fries

Birth: October 22, 1973 in Sursee, Switzerland

Citizenship: Switzerland (Triengen LU)

Education

1998-2004

2001-2004

1997-1998

1993-1998

1986-1993

1980-1986

PhD Student and Research Assistant

Laboratory for Electromagnetic Fields and Microwave Electronics

Swiss Federal Institute of Technology (ETH), Zürich

Postgraduate Studies in Management

Swiss Federal Institute of Technology (ETH), Zürich, Switzerland

Dipl. NDS in Betriebswissenschaften

Diploma Thesis

Setup of near-field antenna measurement range

Royal Institute of Technology (KTH), Stockholm, Sweden

Studies of Electrical Engineering

Ecole polytechnique fédérale de Lausanne EPFL

Swiss Federal Institute of Technology (EPFL), Lausanne, Switzerland

Diplôme d'ingénieur électricien

High School (Gymnasium)Kantonsschule Sursee LU, Switzerland

Matura Typus C (Scientific)

Primary School

Primarschule Triengen LU, Switzerland

List of Publications

Conference Papers

[1] M. Fries, M. Kossel, R. Vahldieck, and W. Bächtold, "Aperture coupled patch

antennas for an rfid system using circular polarization modulation", in ESA

Millenium Conference on Antennas & Propagation, Davos, Switzerland, April

9-14 2000.

[2] M.K. Fries and R. Vahldieck. "Small microstrip patch antenna using slow-

wave structure", IEEE Antennas and Propagation Society International Sym¬

posium, pages 770-773, July 2000, Salt Lake City, USA

[3] Matthias K. Fries, and Rüdiger Vahldieck, "Novel Circularly Polarized Uni¬

planar Antenna Architectures", In Proc. 2001 URSI International Symposium

on Electromagnetic Theory, pages 624-626, Victoria, Canada, May 2001. Uni¬

versity of Victoria.

[4] M.K. Fries and R. Vahldieck, "A novel concept for slot coupled circularly

polarized patch antenna", IEEE Antennas and Propagation Society Interna¬

tional Symposium, volume 3, pages 490-493, July 2001, Boston MA, USA

[5] M.K. Fries, M. Gräni, and R. Vahldieck, "Slot-antenna with switchable polar¬

ization",IEEE Antennas and Propagation Society International Symposium,

volume 2, pages 440-443, June 2002, San Antonio, USA

[6] M.K. Fries, R. Vahldieck and R. Peter," Low cost patch antenna for passive

microwave tagging system ", IEEE Antennas and Propagation Society Inter¬

national Symposium, volume 2, pages 692-695, June 2003, Columbus, USA.

Journal Papers

[7] M.K. Fries and R. Vahldieck,"

Uniplanar circularly polarized slot-ring antenna

architectures", Radio Science, 38(2), November 2002.

[8] M.K. Fries, M. Gräni, and R. Vahldieck, "A reconfigurable slot antenna with

switchable polarization.", IEEE Microwave and Wireless Components Letters,

13(11):490-492, November 2003.

planar antennas zÜrich presented by ing. él. dipl. epfl

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