the technician's radio receiver handbook

This book deals with radio receivers. We start the process of dissecting a radio receiver bylooking at the various receiver configurations and architectures and the performance parame-ters of receivers. We then look at the front-end circuit (filtering, RF amplifiers, mixers). Next,we take a look at the IF amplifier and IF filtering circuits (surface acoustic wave, crystal, andmechanical). Finally, we take a look at the demodulator circuits used in radio receivers.

After the basic receiver circuits, we look at spectrum analyzer receivers, digital signal pro-cessing, and special purpose receivers. Receiver tests and measurements are covered, as isplanning a receiver system. A chapter is provided on improving receiver performance in a highEMI environment, which is increasingly important. Finally, we deal with diversity receptiontechniques.

Joseph J. Carr Falls Church, Virginia

[email protected]

ix

Preface

Radio communication is about transmittingintelligence from one place to another, with-out the benefit of wires or fiber optics. Thebasic building blocks of the radio communi-cations system are the transmitter and the re-ceiver. Radio communication takes placewhen a transmitter sends an electromagneticwave to a receiver (Figure 1.1). The receiverdemodulates the signal and recovers the intel-ligence impressed on the transmitted signal.

The receiver that demodulates a signalis a relatively complex subsystem to the com-munications system. Although that commu-nications system includes a transmitter and

antennas, they are better left to other treat-ments. This book focuses on the receiver.

We look at various receiver configura-tions, and the performance characteristics thatpertain to them. We then look at the front-end of a typical radio receiver in Chapters 4–7(“The Front End: An Overview,” “Front-EndFiltering,” “RF Amplifiers and Preamplifiers,”and “The Mixer Stage”) and deal with the var-ious parameters that affect performance ofthe receiver.

We also look at the local oscillator andfrequency synthesizer circuits. These circuitsare necessary to convert the radio frequency(RF) signal to an intermediate frequency (IF)for further processing.

The IF amplifier and the filters that gowith it also are covered. The IF amplifier andfilter circuit provides most of the bandwidthand gain of the typical superheterodyne re-ceiver, which makes them very important tothe performance of the receiver.

Finally, we deal with the demodulatorand “the other” receiver circuits (automaticgain control, noise blankers and eliminators,squelch, etc.).

Following the concentration on specificcircuits, we turn attention to certain receiver

Chapter 1

The Role of the Receiver

1

Fig. 1.1 Communication system showing thetransmitter and receiver.

TRANSMITTER RECEIVER

technology topics: MMIC technology, spec-trum analyzers, digital signal processing,special purpose receivers, receiver tests andmeasurements improving the performanceof the receivers in a high electromagneticinterference environment, and diversity re-ception. Finally, we deal with planning a receiver system.

We begin with a discussion of receiverarchitectures and technology. First, we dis-

cuss the central problem of radio reception:the signal-to-noise ratio. The radio receiver isjudged by its ability to overcome the noise ofthe system.

The various receiver configurations in-clude the crystal video, the tuned radio fre-quency, direct conversion, and of course, thesuperheterodyne. We discuss each of thesearchitectures.

2 THE TECHNICIAN’S RADIO RECEIVER HANDBOOK

The radio receiver is designed to receive ra-dio signals. How it does that job depends onthe receiver architecture. That is what thischapter is all about. But first, before coveringreceiver architectures, I discuss a little aboutthe basic problem of radio reception: sepa-rating signals from noise.

SIGNALS, NOISE, AND RECEPTION

No matter how simple or fancy the systemmay be, the basic function of a radio re-ceiver is the same: to distinguish signalsfrom noise. The concept of “noise” coversboth human-made and natural radio fre-quency signals. Human-made signals in-clude all signals in the pass band other thanthe one being sought.

In communications systems, the signalis some form of modulated (AM, FM, PM,on-off telegraphy, etc.) periodic sine wavepropagating as an electromagnetic (i.e., ra-dio) wave. The noise, on the other hand,may be a random signal that sounds like the“hiss” heard between stations on a radio.The spectrum of such noise signals appearsto be Gaussian (“white noise”) or pseudo-

gaussian (“pink noise,” or bandwidth lim-ited noise).

In radio astronomy and satellite com-munications systems, the issue is compli-cated because the signals also are noise.The radio emissions of Jupiter and the Sunare very much like the signals that, in othercontexts, are nothing but useless noise. Infact, in the early days of radar, the galacticnoise tended to mask returns from incomingenemy aircraft, so, to the radar operators,these signals were noise of the worst kind.Yet, to a radio astronomer, those signals arethe goal. In satellite communications sys-tems, the “signals” of the radio astronomerare limitations and annoyances, at best, anddevastating, at worse. The trick is to sepa-rate out the noise you want from the noiseyou do not.

Figure 2.1A shows an amplitude vs.time plot of a typical noise signal, whileFigure 2.1B shows a type of regular radio sig-nal that could be generated by a transmitter.Note the difference between the two. Thesignal is regular and predictable. Once youknow the frequency and period you can pre-dict the amplitude at other points along thetime line. The noise signal, on the other

Chapter 2

Radio Receiver Architectures

3

hand, is unpredictable. Knowing the cycle-to-cycle amplitude and duration (there is notrue period) does not confer the ability topredict anything at all about the followingcycles.

In some receivers, especially those de-signed for pulse reception, the differenceshighlighted between A and B in Figure 2.1are used to increase the performance of thereceiver. An integrator circuit finds the timeaverage of the input signal. True Gaussiannoise integrated over a sufficiently longterm will average to zero. This occurs be-cause Gaussian noise contains all phases,amplitudes, and polarities randomly distrib-uted. Pseudogaussian noise is bandwidthlimited, so it may not integrate to zero butvery near it. The signal, on the other hand,will integrate to some nonzero value, so itwill stand out in the presence of integratednoise.

Thermal Noise

Every electronic system (even a simple resis-tor) generates thermal noise, even if nopower is flowing through it. One goal of thesystem designer is to minimize the noiseadded by the system, so that weaker signalsare not obscured. A basic form of noise seenin systems is thermal noise. Even if the am-plifiers in the receiver add no additional

noise (they will), thermal noise will be foundat the input due to the input resistance.

If you replace the antenna with a resistormatched to the system impedance and totallyshielded, noise still will be present. The noiseis produced by the random motion of elec-trons inside the resistor. At all temperaturesabove absolute zero (about −273.16ºC), theelectrons in the resistor material are in randommotion. At any given instant, a huge numberof electrons will be in motion in all directions.The reason why there is no discernible currentflow in one direction is that the motions cancelout each other, even over short time periods.The noise power present in a resistor is

PN = KTBR watts (2.1)

where

PN is the noise power in watts;T is the temperature in degrees

Kelvin (K);K is Boltzmann’s constant

(1.38 × 10 − 23 joules/K);R is the resistance in ohms (Ω);B is bandwidth in hertz (Hz).

Note that, by international agreement, T is setto 290K.

Consider a receiver with a 1 MHz band-width and an input resistance of 50 Ω. Thenoise power is (1.38 × 10–23 joules/K) × (290K)× (1,000,000 Hz) × (50 Ω) = 2 × 10–13 W.


Fig. 2.1 Amplitude vs. time: (A) noise; (B) signal.

Time

+V

-V

0V

Time

+V

-V

0V

A B

The Reception Problem

Equation 2.2 shows the basic problem of ra-dio reception, especially in cases where thesignal is very weak. The signal in Figure 2.2Ais embedded in noise of relatively high am-plitude. This signal is lower than the noiselevel, so it is very difficult (perhaps impossi-ble) to detect. The signal in Figure 2.2B iseasily detectable because the signal ampli-tude is higher than the noise amplitude.Detection becomes difficult when the signalis only slightly stronger than the averagenoise power level.

The signal-to-noise ratio (SNR) of a re-ceiver system tells us something about thedetectability of the signal. The SNR normallyis expressed in decibels (dB), which are de-fined as

(2.2)

where

SNR is the signal-to-noise ratio in deci-bels (dB);

PS is the signal power level;PN is the noise power level.

How high an SNR is required dependson a lot of subjective factors when a humanlistener is present. Skilled radio operators can

detect signals with an SNR of less than 1 dB—but the rest of us cannot even hear that sig-nal. Most radio operators can detect 3 dB SNRsignals, but for “comfortable” listening, a 10dB SNR usually is specified. For digital sys-tems, the noise performance usually is de-fined by the acceptable bit error rate (BER).

Strategies

A number of strategies can be used to im-prove the SNR of a system. First, of course, isto buy a receiver that has a low internal“noise floor” and do nothing to upset that fig-ure. High-quality receivers have very lownoise, but sometimes some creative specwriting in the advertisements uses differentbandwidths for the measurement, and onlythe most favorable value—which may not bethe bandwidth that matches your needs—isreported.

By common sense, we see that thereare two approaches to SNR improvement: ei-ther increase the signal amplitude or de-crease the noise amplitude. Most successfulsystems do both, but they must be donecarefully.

One approach to SNR improvement isto use a preamplifier ahead of the receiverantenna terminals. This approach may ormay not work and under some situationsmay make the situation worse. The problem

SNR

P

PS

N

=

10 log dB

Radio Receiver Architectures 5

Fig. 2.2 The noise problem: (A) signal imbedded in noise; (B) good signal-to-noise ratio.

Time

+V

-V

0V Time

+V

-V

0V

A B

is that the preamplifier adds noise of its ownand will amplify noise from outside (receivedthrough the antenna) and the desired signalequally. If you have an amplifier with a gainof, say, 20 dB, then the external noise andthe signal both are increased by 20 dB. Theresult is that the absolute numbers are biggerbut the SNR is the same. If the amplifier pro-duces any significant noise of its own, thenthe SNR will degrade. The key is to use avery low-noise amplifier (LNA) for the pre-amplifier. Using an LNA for the preamplifiermay actually reduce the noise figure of thereceiver system.

Another trick is to use a preselectorahead of the receiver. A preselector is ei-ther a tuned circuit or a bandpass filterplaced in the antenna transmission lineahead of the receiver antenna terminals. Apassive preselector has no amplification(uses L-C elements only), while an activepreselector has a built-in amplifier. The am-plifier should be an LNA type. The prese-lector can improve the system because itamplifies the signal by a fixed amount, butonly the noise within the passband is am-plified the same amount as the signal.Improvement comes from bandwidth limit-ing the noise but not the signal.

Yet another practical approach is to usea directional antenna. This method works es-pecially well when the unwanted noise isother human-made signal sources. An omni-directional antenna receives equally well inall directions. As a result, both natural andhuman-made external noise sources operat-ing within the receiver’s passband will bepicked up. But if the antenna is made highlydirectional, then all noise sources not in thedirection of interest are suppressed.

Highly directional antennas have gain,so the signal levels in the direction of interestare increased. Although the noise also in-creases in that direction, the rest of the noisesources (in other directions) are suppressed.The result is that the SNR is increased byboth methods.

When designing a communications sys-tem, the greatest attention usually should bepaid to the antenna, then to an LNA or low-

noise preselector, and then to the receiver.Generally speaking, money spent on the an-tenna gives more signal to noise than thesame money spent on amplifiers and otherattachments.

RADIO RECEIVER ARCHITECTURES

Radio receivers are at the heart of nearly allcommunications activities. In this chapter, wediscuss the different types of radio receivers onthe market. We learn how to interpret receiverspecifications in Chapter 3. Later, we look atspecific designs for specific applications.

Origins

The very earliest radio receivers were not re-ceivers at all in the sense we know the termtoday. Early experiments by Hertz, Marconi,and others used spark gaps and regular tele-graph instruments of the day. Range was se-verely limited because those devices have aterribly low sensitivity to radio waves. Later,around the turn of the 20th century, a devicecalled a Branly coherer was used for radio sig-nal detection. This device consisted of a glasstube filled with iron filings placed in series be-tween the antenna and the ground. Althoughconsiderably better than earlier apparatus, thecoherer was something of a dud for weak sig-nal reception. In the first decade of this cen-tury, however, Fleming invented the diodevacuum tube and Lee DeForest invented thetriode vacuum tube. The latter device madeamplification possible and detection a lotmore efficient.

A receiver must perform two basic func-tions: (1) It must respond to, detect, and de-modulate desired signals; and (2) it must notrespond to, detect, or be adversely affectedby undesired signals. If it fails in either ofthese two functions, then the design performspoorly.

Both functions are necessary. Weaknessin either function makes a receiver a poorbargain, unless there is some mitigating cir-cumstance. The receiver’s performance spec-ifications tell us how well the manufacturer


claims that its product does these two functions.

Crystal Video Receivers

Crystal video receivers (Figure 2.3) grew outof primordial crystal sets but sometimes areused in microwave bands even today. Theoriginal crystal sets (pre-1920) used a natu-rally occurring PN junction “diode” madefrom a natural lead compound, called galenacrystal, with an inductor-capacitor (L-C)tuned circuit. Later, crystal sets were madeusing germanium or silicon diodes. Whenvacuum tubes became generally available, it

was common to place an audio amplifier atthe output of the crystal set.

Modern crystal video receivers use sili-con or gallium-arsenide microwave diodes anda wideband video amplifier (rather than theaudio amplifier). Applications include somespeed radar receivers, aircraft warning re-ceivers, and some communications receivers(especially short range).

Tuned Radio Frequency Receivers

The tuned radio frequency (TRF) radio re-ceiver uses an L-C resonant circuit in thefront end, followed by one or more radio


Fig. 2.3Crystal video receiver.

ANTENNA

HIGH-GAIN AMPLIFIER

D1CRYSTAL

DETECTOR

OUTPUT

Fig. 2.4 TRF receiver.

TUNED CIRCUITHIGH-GAINAMPLIFIER

DETECTOR OUTPUT

ANTENNA

frequency amplifiers ahead of a detectorstage. Two varieties are shown in Figures 2.4and 2.5. The version in Figure 2.4 is a tunedgain-block receiver. It commonly is used inmonitoring very low frequency (VLF) signalsto detect solar flares and sudden ionosphericdisturbances (SIDs).

Later versions of the TRF concept usemultiple TRF circuits between the amplifierstages. These designs also are used in VLFsolar flare and SID monitoring. Early modelsused independently tuned L-C circuits, butthose proved very difficult to tune withoutcreating an impromptu Miller oscillator cir-cuit. Later versions mechanically linked(“ganged”) the tuned circuits to operate froma single tuning knob.

Superheterodyne Receivers

Figure 2.6 shows the block diagram of a su-perheterodyne receiver. We use this hypothet-ical receiver as the basic generic frameworkfor evaluating receiver performance. The de-

sign in Figure 2.6 represents the largest classof radio receivers; it covers the vast majority ofreceivers on the market.

The purpose of a superheterodyne re-ceiver is to convert the incoming RF fre-quency to a single frequency where most ofthe signal processing takes place. The front-end section of the receiver consists of the ra-dio frequency amplifier and any RF tuningcircuits that may be used (A, B, and C inFigure 2.6). In some cases, the RF tuning isvery narrow and basically tunes one fre-quency. In other cases, the RF front-end tun-ing is broadband. In that case, bandpassfilters are used.

The frequency translator section (D andE) also is considered part of the front end inmost textbooks, but here we label it a sepa-rate entity. The translator consists of a fre-quency mixer and a local oscillator. Thissection does the heterodyning, which is dis-cussed in more detail later. The output of thefrequency translator is called the intermedi-ate frequency.


Fig. 2.5 Multituned TRF receiver.

TUNEDCIRCUIT

TUNEDCIRCUIT

TUNEDCIRCUIT

AMPLIFIER AMPLIFIERAMPLIFIER

DETECTOROUTPUT STAGESOUTPUT

ANTENNA

The translator stage is followed by the in-termediate frequency amplifier. The IF ampli-fier (F, G, and H) basically is a radio frequencyamplifier tuned to a single frequency. The IFcan be higher or lower than the RF frequency,but it always will be a single frequency.

A sample of the IF amplifier output signalis applied to an automatic gain control (AGC)section (L and M). The purpose of this sectionis to keep the signal level in the output moreor less constant. The AGC circuit consists of arectifier and a ripple filter that produce a DCcontrol voltage. The DC control voltage is pro-portional to the input RF signal level (N). It isapplied to the IF and RF amplifiers to raise orlower the gain according to signal level. If thesignal is weak, then the gain is forced higher;and if the signal is strong, the gain is lowered.The result is to smooth out variations of theoutput signal level.

The detector stage (I) is used to recoverany modulation on the input RF signal. Thetype of detector depends on the type ofmodulation used for the incoming signal.Amplitude modulation (AM) signals generally

are handled in an envelope detector. In somecases, a special variant of the envelope de-tector, called a square law detector, is used.The difference is that the straight envelopedetector is linear, while the square law detec-tor is nonlinear. Single sideband (SSB), dou-ble sideband suppressed carrier (DSBSC),and keyed CW signals will use a product de-tector, while FM and PM need a frequency orphase sensitive detector.

The output stages (J and K) are used toamplify and deliver the recovered modula-tion to the user. If the receiver is for broad-cast use, then the output stages are audioamplifiers and loudspeakers. In some radioastronomy and instrumentation telemetry re-ceivers, the output stages consist of integra-tor circuits and DC amplifiers.

Heterodyning

The main attribute of the superheterodyne re-ceiver is that it converts the radio signal’s RFfrequency to a standard frequency for furtherprocessing. Although today the new frequency,


Fig. 2.6 Superheterodyne receiver.

RFTUNED

CIRCUITRF AMPLIFIER

RFTUNED

CIRCUITMIXER

IFTUNED

CIRCUIT

IFTUNED

CIRCUITIF AMPLIFIER

LOCALOSCILLATOR

DETECTOROUTPUTSTAGESOUTPUT

ANTENNA

RECTIFIERRIPPLEFILTER

DC CONTROL VOLTAGE

AUTOMATIC GAIN CONTROL

A

E

DCB HGF

IJ

K

LM

N

called the intermediate frequency, may be ei-ther higher or lower than the RF frequencies,early superheterodyne receivers always downconverted RF signal to a lower IF frequency(IF < RF). The reason was purely practical, forin those days higher frequencies were moredifficult to process than lower frequencies.Even today, because variable tuned circuitsstill tend to offer different performance overthe band being tuned, converting to a singleIF frequency and obtaining most of the gainand selectivity functions at the IF allows moreuniform overall performance over the entirerange being tuned.

A superheterodyne receiver works byfrequency converting (heterodyning—the su-per part is 1920s vintage advertising hype) theRF signal. This occurs by nonlinearly mixingthe incoming RF signal with a local oscillator(LO) signal. When this process is done, disre-garding noise, the output spectrum will con-tain a large variety of signals according to theformula

FO = mFRF ± nFLO (2.3)

where

FRF is the frequency of the RF signal; FLO is the frequency of the local

oscillator;m and n are either 0 or integers (0, 1,

2, 3, . . ., n)

Equation 2.3 means that a large numberof signals will be at the output of the mixer,although for the most part the only ones ofimmediate concern to understanding super-heterodyne operation are those for which mand n are either 0 or 1. Therefore, for ourpresent purpose, the output of the mixer willbe the fundamentals (FRF and FLO) and sec-ond-order products (FLO − FRF and FLO + FRF),as seen in Figure 2.7. Some mixers, notablythose described as double-balanced mixers(DBM), suppress FRF and FLO in the mixeroutput, so only the second-order sum anddifference frequencies exist with any appre-ciable amplitude. This case is simplistic andused only for the present discussion. Lateron, we look at what happens when third-order (2F1 ± F2 and 2F2 ± F1) and fifth-order(3F1 ± 2F2 and 3F2 ± 2F1) become large.

Note that the local oscillator frequencycan be either higher than the RF frequency(high-side injection) or lower than the RF fre-quency (low-side injection). Ordinarily, thereis no practical reason to prefer one over theother except that it will make a differencewhether an analog main tuning dial (if used)reads high to low or low to high.

The candidates for IF are the sum (LO + RF) and difference (LO − RF) second-order products found at the output of themixer. A high-Q tuned circuit following the


Fig. 2.7 Relationship between LO, RF, and the IFs.

FRF

FLOFRF - FLO

FRF + FLO

FREQUENCY

mixer will select which of the two are used.Consider an example. Suppose an AMbroadcast band superheterodyne radio hasan IF frequency of 455 kHz, and the tuningrange is 540 to 1700 kHz. Because the IF islower than any frequency within the tuningrange, the difference frequency will be se-lected for the IF. The local oscillator is setto be high-side injection and so will tunefrom (540 + 455) = 995 kHz to (1,700 + 455)= 2155 kHz.

Front-End Circuits

The principal task of the front-end and fre-quency translator sections of the receiver inFigure 2.6 is to select the signal and convertit to the IF frequency. But many radio re-ceivers have additional functions. In somecases (but not all), an RF amplifier will beused ahead of the mixer. Typically, these am-plifiers have a gain of 3–10 dB, with 5–6 dBbeing very common. The tuning for the RFamplifier sometimes is a broad bandpassfixed-frequency filter that admits an entireband. In other cases, it is a narrow band butvariable frequency tuned circuit.

Intermediate Frequency Amplifier

The IF amplifier is responsible for providingmost of the gain in the receiver, as well as thenarrowest bandpass filtering. It is a high gain,often multistaged, single-frequency tuned ra-dio frequency amplifier. For example, one HFshortwave receiver block diagram lists 120 dBof gain from antenna terminals to audio out-put, of which 85 dB are provided in the 8.83MHz IF amplifier chain. In the example ofFigure 2.6, the receiver is a single conver-sion design, so there is only one IF amplifiersection.

Detector

The detector demodulates the RF signal andrecovers whatever audio (or other informa-tion) is to be heard by the listener. In astraight AM receiver, the detector will be anordinary half-wave rectifier and ripple filter,

called an envelope detector. In other detec-tors—notably double sideband suppressedcarrier, single sideband suppressed carrier(SSBSC or SSB), or continuous wave (CW orMorse telegraphy)—a second local oscillator,usually called a beat frequency oscillator(BFO), operating near the IF frequency, isheterodyned with the IF signal. The resultantdifference signal is the recovered audio. Thattype of detector is called a product detector.Many AM receivers today have a sophisti-cated synchronous detector rather than thesimple envelope detector.

Audio Amplifiers

The audio amplifiers are used to finish thesignal processing. They also boost the out-put of the detector to a usable level to drivea loudspeaker or set of earphones. The au-dio amplifiers are sometimes used to pro-vide additional filtering. It is quite commonto find narrow band filters to restrict audiobandwidth, or notch filters to eliminate inter-fering signals that make it through the IF am-plifiers intact.

Double- and Triple-Conversion Receivers

Double- and triple-conversion receivers aredesigned to take advantage of two aspectsof radio design. The first is that a high IFfrequency will yield superior image perfor-mance, and the second is that it is easier toget high gain and bandwidth limiting filtercharacteristics at low frequencies. Figure 2.8shows a basic double-conversion receiver. Itfirst converts the IF to a high IF frequencyand then down converts it to a lower IFfrequency.

The particular frequencies selected forthe first and second IF depend on the ap-plication. In high-frequency shortwave re-ceivers, the first IF will be on the order of50 MHz to gain the advantages of a high IFfrequency (better image response). Thesecond IF will be 10.7 MHz, 9 MHz, 8.83MHz, or 455 kHz, depending on the design.In the VHF/UHF bands the high IF may be10.7 MHz, 50 MHz, or 70 MHz; whereas the



second IF will be 10.7 MHz or 455 kHz, de-pending on the design.

Direct-Conversion Receivers

The direct-conversion receiver (DCR) is asubset of the superheterodyne in which theintermediate frequency is equal to the base

band frequencies. It converts radio signals di-rectly to audio rather than to an IF. The localoscillator on the direct conversion receiver istuned to either the RF signal or a certain off-set that depends on the tone you wish to lis-ten to in CW reception. The result is directaudio conversion in the DCR receiver.

Fig. 2.8 Double-conversion receiver.

RF AMPLIFIER

ANTENNA

MIXERNo. 1

LO No. 1

1st IFAMPLIFIER

MIXERNo. 2

LO No. 2

2nd IFAMPLIFIER

DETECTOR

Three basic areas of receiver performancemust be considered. Although interrelated,they are sufficiently different to merit individ-ual consideration: noise, static attributes, anddynamic attributes. We look at all these areasin this chapter. But first we look at the unitsof measuring receiver performance factors.

UNITS OF MEASURE

Input Signal Voltage

Input signal level, when specified as a voltage,typically is stated in either microvolts (µV) ornanovolts (nV). The volt is simply too large aunit for practical use on radio receivers. Signalinput voltage (or sometimes power level) of-ten is used as part of the sensitivity specifica-tion or as a test condition for measuringcertain other performance parameters.

Two forms of signal voltage are used forinput voltage specification: source voltageand potential difference, as illustrated inFigure 3.1. The source voltage (VEMF) is theopen terminal (no load) voltage of the signalgenerator or source, while the potential dif-ference (VPD) is the voltage that appears

across the receiver antenna terminals with theload connected (the load is the receiver an-tenna input impedance, Rin). When Rs = Rin,the preferred “matched impedances” case inradio receiver systems, the value of VPD is onehalf that of VEMF. This can be seen in Figure3.1 by noting that Rs and Rin form a voltage di-vider network driven by VEMF, with VPD as theoutput:

(3.1)

Special Units

DBM

These units refer to decibels relative to onemilliwatt (1 mW) dissipated in a 50 Ω resis-tive impedance (defined as the 0 dBm refer-ence level), calculated from

(3.2)

or

(3.3) dBm = 10 log ( )PMW

dBm Watts=

100 001

log.

P

V

V R

RPDEMF in

s

=

Chapter 3

Receiver Performance Factors

13

In the noise voltage case calculated inChapter 2, 0.028 µV in 50 Ω, the power isV2/50, or 5.6 × 10–10 W, which is 5.6 × 10–7

mW. In dBm notation, this value is 10 log(5.6 × 10–7), or −62.5 dBm.

DBMV This unit is used in television receiver sys-tems in which the system impedance is 75 Ω,rather than the 50 Ω normally used in otherRF systems. It refers to the signal voltage,measured in decibels, with respect to a signallevel of one millivolt (1 mV) across a 75 Ω re-sistance (0 dBmV). In many TV specs, 1 mVis the full quieting signal that produces no“snow” (i.e., noise) in the displayed picture.Note: 1 mV = 1000 µV.

DBµV This unit refers to a signal voltage, measuredin decibels, relative to one microvolt (1 µV)developed across a 50 Ω resistive impedance(0 dBµV). For the case of our noise signalvoltage, the level is 0.028 µV, which is thesame as −31.1 dBµV. The voltage used for thismeasurement usually is the VEMF, so to findVPD divide it by 2 after converting dBµV to µV.

Rule of Thumb: To convert dBµV to dBm,subtract 113 dB; that is:

100 dBµV = (100 dBµV − 113 dB) = −13 dBm.

It requires only a little algebra to convertsignal levels from one unit of measure to an-other. This job sometimes is necessary whena receiver manufacturer mixes methods in thesame specifications sheet. In the case of dBmand dBµV, 0 dBµV is 1 µV VEMF, or a VPD of0.5 µV, applied across 50 Ω, so the power dis-sipated is 5 × 10–15 W, or −113 dBm.

NOISE

A radio receiver must detect signals in thepresence of noise. The signal-to-noise ratio isthe key here because a signal must be abovethe noise level before it can be successfullydetected and used.

Noise comes in a number of differentguises, but for sake of this discussion, we di-vide them into two classes: sources externalto the receiver and sources internal to the re-ceiver. One can do little about the externalnoise sources, for they consist of natural andhuman-made electromagnetic signals that fallwithin the passband of the receiver. Figure3.2A shows an approximation of the externalnoise situation from the middle of the AMbroadcast band to the low end of the VHF re-gion. A somewhat different view, which cap-tures the severe noise situation seen byreceivers, is shown in Figure 3.2B. One mustselect a receiver that can cope with externalnoise sources, especially if the noise sourcesare strong.

Some natural external noise sources areextraterrestrial. These signals form the basisof radio astronomy. For example, if you aim abeam antenna at the eastern horizon prior tosunrise, a distinct rise of noise level occurs asthe sun slips above the horizon, especially inthe VHF region (the 150–152 MHz band isused to measure solar flux). The reverse oc-curs in the west at sunset but is less dramatic,probably because atmospheric ionization de-cays much slower than it is generated. DuringWorld War II, British radar operators noted anincrease in received noise level any time theMilky Way was above the horizon, decreasingthe range at which they could detect in-


Fig. 3.1Receiver input voltages.

Rs

VEMF

Rin RECEIVERVPD

Receiver Performance Factors 15

Fig. 3.2 Noise: (A) in the HF band; (B) sources throughout the bands.

10 100 1000 10,000

FREQUENCY (MHz)

-10

0

10

20

30

40

50

60

dBn

dBn is noise figureabove kToB

NO

ISE

FIG

UR

E A

BO

VE

kT

oB (

dB)

SOLAR(QUIET DAY)

STANDARDOR TYPICALRECEIVER

ATMOSPHERIC

HUMAN-MADE(URBAN)

(SUBURBAN)

GALACTIC

HUMAN-MADE

-20

-15

-10

-5

0

+5

+10

+15

+20

dBµV

FREQUENCY (MHz)

1MHz

10MHz

30MHz

F

A

B

bound German bombers. Also, some well-known, easily observed noise emanates fromthe planet Jupiter in the 18–30 MHz range.

The receiver’s internal noise sources aredetermined by the design of the receiver. Idealreceivers produce no noise of their own, sothe output signal from the ideal receiver wouldcontain only the noise present at the inputalong with the radio signal. But real receivercircuits produce a certain level of internalnoise of their own. Even a simple fixed-valueresistor is noisy. Figure 3.3A shows the equiva-lent circuit for an ideal, noise-free resistor;while Figure 3.3B shows a practical real-worldresistor. The noise in the real-world resistor isrepresented by a noise voltage source, VN, inseries with the ideal, noise-free resistance, R1.At any temperature above absolute zero (0K orabout −273oC), electrons in any material are inconstant random motion. Because of the in-herent randomness of that motion, however,there is no detectable current in any one direc-tion. In other words, electron drift in any sin-gle direction is cancelled over even short timeperiods by equal drift in the opposite direc-tion. Electron motions therefore are statisticallydecorrelated. However, a continuous series ofrandom current pulses is generated in the ma-terial, and those pulses are seen by the outsideworld as noise signals.

If a perfectly shielded 50 Ω resistor isconnected across the antenna input terminalsof a radio receiver, the noise level at the re-ceiver output will increase by a predictableamount over the short-circuit noise level.Noise signals of this type are called by sev-eral names: thermal agitation noise, thermalnoise, or Johnson noise. This type of noisealso is called white noise, because it has a

very broadband (nearly Gaussian) spectraldensity. The thermal noise spectrum is domi-nated by mid-frequencies (104–105 Hz) andessentially is flat. The term white noise is ametaphor developed from white light, whichis composed of all visible color frequencies.The expression for such noise is

(3.4)

where

VN is the noise potential in volts (V);K is Boltzmann’s constant

(1.38 × 10–23 J/K);T is the temperature in degrees Kelvin

(K), normally set to 290 or 300Kby convention;

R is the resistance in ohms (Ω);B is the bandwidth in hertz (Hz).

Table 3.1 and Figure 3.4 show noisevalues for a 50 Ω resistor at various band-widths out to 5 kHz and 10 kHz, respec-tively. Because different bandwidths are usedfor different reception modes, it is common

V KTBRN = 4


Fig. 3.3 Resistors: (A) The ideal resistor is noisefree; (B) real resistors have noise.

R1

A

R1

B

VN

Table 3.1 Bandwidth vs. Noise Voltage

BW (Hz) Noise Volts × E-08

500 1.0

1000 1.41

1500 1.73

2000 2.0

2500 2.24

3000 2.45

3500 2.65

4000 2.83

4500 3.0

5000 3.16

5500 3.32

6000 3.46

6500 3.61

7000 3.74

7500 3.87

8000 4.0

8500 4.12

9000 4.24

9500 4.36

10,000 4.47

practice to delete the bandwidth factor inequation 3.4 and write it in the form

(3.5)

With equation 3.5, one can find thenoise voltage for any particular bandwidthby taking its square root and multiplying itby the equation. This equation essentially isthe solution of the previous equation normal-ized for a 1 Hz bandwidth.

Signal-to-Noise Ratio

Receivers are evaluated for quality on the ba-sis of signal-to-noise ratio (S/N, SNR), some-times denoted SN. The goal of the designer isto enhance the SNR as much as possible.Ultimately, the minimum signal level de-tectable at the output of an amplifier or radioreceiver is that level which appears justabove the noise floor level. Therefore, thelower the system noise floor, the smaller isthe minimum allowable signal.

Noise Factor, Noise Figure, and Noise Temperature

The noise performance of a receiver or am-plifier can be defined in three different butrelated ways: noise factor, noise figure, andequivalent noise temperature. These proper-ties are definable as a simple ratio, decibelratio, or Kelvin temperature, respectively.

NOISE FACTOR

For components such as resistors, the noisefactor (FN) is the ratio of the noise producedby a real resistor to the simple thermal noiseof an ideal resistor. The noise factor of a ra-dio receiver (or any system) is the ratio ofoutput noise power (PNO) to input noisepower (PNI):

(3.6)

To make comparison easier, the noisefactor is usually measured at the standard

FP

PNNO

NI T

=

=290K

V KTR VN = 4 Hz


Fig. 3.4 Thermal noise voltage vs. bandwidth frequency.

0.00E+00

1.00E-08

2.00E-08

3.00E-08

4.00E-08

5.00E-08

6.00E-08

7.00E-08

8.00E-08

9.00E-08

1.00E-07

0 2000 4000 6000 8000 10000 12000

BANDWIDTH IN HERTZ (Hz)

TH

ER

MA

L N

OIS

E V

OLT

AG

E

temperature (To) of 290K (standardized roomtemperature); although in some countries299 or 300K commonly is used (the differ-ences are negligible). It also is possible todefine noise factor, FN, in terms of the outputand input signal-to-noise ratios:

(3.7)

where

SNI is the input signal-to-noise ratio;SNO is the output signal-to-noise ratio.

NOISE FIGURE

The noise figure (NF) frequently is used tomeasure the receiver’s “goodness”; that is, itsdeparture from the ideal. Therefore, it is afigure of merit. The noise figure is the noisefactor converted to decibel notation:

NF = 10 log (FN) (3.8)

where

NF is the noise figure in decibels (dB); FN is the noise factor; log refers to the system of base-10

logarithms.

NOISE TEMPERATURE

The noise “temperature” (Te) is a means forspecifying noise in terms of an equivalentnoise temperature; that is, the noise level thatwould be produced by a matching resistor(e.g., 50 Ω) at that temperature (expressed indegrees Kelvin). Evaluating the noise equa-

tions shows that the noise power is directlyproportional to temperature in degrees Kelvinand that noise power collapses to 0 at thetemperature of absolute zero (0K).

Note that the equivalent noise tempera-ture, Te, is not the physical temperature ofthe amplifier but rather a theoretical con-struct that is an equivalent temperature pro-ducing that amount of noise power in aresistor. The noise temperature is related tothe noise factor by

Te = (FN − 1)TO (3.9)

and to noise figure by

(3.10)

Noise temperature often is specified forreceivers and amplifiers in combination withor in lieu of the noise figure.

Noise in Cascade Amplifiers

A noise signal is seen by any amplifier fol-lowing the noise source as a valid input sig-nal. Each stage in the cascade chain (Figure3.5) amplifies both signals and noise fromprevious stages and contributes some addi-tional noise of its own. Thus, in a cascadeamplifier, the final stage sees an input signalthat consists of the original signal and noiseamplified by each successive stage plus thenoise contributed by earlier stages. The over-all noise factor for a cascade amplifier can becalculated from Friis’s noise equation:

T K T

NFe O=

−−log 1

101

F

S

SNNI

NO

=


Fig. 3.5 N stages in cascade.

GAIN = G1

NOISE FACTOR. = F 1

GAIN = G3

NOISE FACTOR. = F 3

GAIN = G2

NOISE FACTOR. = F 2

GAIN = GN

NOISE FACTOR. = F N

A3

AN

A2A1

(3.11)

where

FN is the overall noise factor of Nstages in cascade;

F1 is the noise factor of stage 1; F2 is the noise factor of stage 2; Fn is the noise factor of the nth stage; G1 is the gain of stage 1; G2 is the gain of stage 2; Gn–1 is the gain of stage (n − 1).

As you can see from Friis’s equation,the noise factor of the entire cascade chain isdominated by the noise contribution of thefirst stage or two. High-gain, multistage RFamplifiers typically use a low-noise amplifier(LNA) circuit for the first stage or two in thecascade chain. Therefore, you will find anLNA at the feedpoint of a satellite receiver’sdish antenna and possibly another one at theinput of the receiver module itself, but otheramplifiers in the chain might be more modestwithout harming system performance.

The matter of signal-to-noise ratio (S/N)sometimes is treated in different ways, eachof which attempts to crank some reality intothe process. The signal-plus-noise-to-noiseratio (S+N/N) quite often is needed. As theratios get higher, the S/N and S+N/N con-verge (only about 0.5 dB difference at ratiosas little as 10 dB). Still another variant is theSINAD (signal-plus-noise-plus-distortion-to-noise) ratio. The SINAD measurement takesinto account most of the factors that can de-teriorate reception.

Receiver Noise Floor

The noise floor of the receiver is a statementof the amount of noise produced by the re-ceiver’s internal circuitry and directly affectsthe sensitivity of the receiver. The noise floortypically is expressed in dBm. The noisefloor specification is evaluated as follows: themore negative, the better. The best receivers

have noise floor numbers of less than −130dBm, while some very good receivers offernumbers of −115 dBm to −130 dBm.

The noise floor depends directly on thebandwidth used to make the measurement.Receiver advertisements usually specify thebandwidth, but be careful to note whether ornot the bandwidth that produced the verygood performance numbers is also the band-width that you will need for the mode oftransmission you want to receive. If, forexample, you are interested only in weak 6kHz wide AM signals and the noise floor isspecified for a 250 Hz CW filter, then thenoise floor might be too high for your use.

STATIC MEASURES OF RECEIVERPERFORMANCE

The two principal static levels of performancefor radio receivers are sensitivity and selectiv-ity. Sensitivity refers to the level of input sig-nal required to produce a usable outputsignal (variously defined). Selectivity refers tothe ability of the receiver to reject adjacentchannel signals (again, variously defined). Welook at both of these factors. Keep in mind,however, that, in modern high-performanceradio receivers, the static measures of perfor-mance, although frequently cited, also maybe less relevant than the dynamic measures(especially in environments with high inter-ference levels).

Sensitivity

Sensitivity is a measure of the receiver’s abil-ity to pick up (“detect”) signals, often speci-fied in microvolts (µV). A typical specificationmight be “0.5 µV sensitivity.” The question toask is this: Relative to what? The sensitivitynumber in microvolts is meaningless unlessthe test conditions are specified. For mostcommercial receivers, the usual test conditionis the sensitivity required to produce a 10 dBsignal-plus-noise-to-noise ratio in the modeof interest. For example, if only one sensitiv-ity figure is given, you must find out whatbandwidth was used: the 5–6 kHz for AM,

F FF

G

F

G G

F

G G G

N

n

n

= + − +−

+

+ −

−

12 3

1

1

1

1

1 2

1

1 2

. . .

. . .


2.1–3 kHz for single sideband, 1.8 kHz for ra-dioteletype, or 200–500 Hz for CW?

Indeed, “creative spec writing” for com-mercial receivers is involved in advertise-ments that enthusiastically cite the sensitivityfor a narrow bandwidth mode (e.g., CW),while the other specifications are cited for amore commonly used, wider bandwidthmode (e.g., SSB). In one particularly egre-gious example, an advertisement claimed asensitivity number that was applicable to the270 Hz CW mode only, yet the 270 Hz CWfilter was an expensive option that had to beordered separately.

The amount of sensitivity improvementis seen by evaluating some simple numbers.Recall that a claim of “x µV” sensitivity refersto some standard such as “x µV to produce a10 dB signal-to-noise ratio in y Hz band-width.” Consider the case where the mainmode for a high frequency (HF) shortwavereceiver is AM (for international broadcast-ing), the sensitivity is 1.9 µV for 10 dB SNR,

and the bandwidth is 5 kHz. If the bandwidthwere reduced to 2.8 kHz for SSB, then thesensitivity would improve by the square rootof the ratio, or √(5/2.8). If the bandwidth isfurther reduced to 270 Hz (i.e., 0.27 kHz) forCW, then the sensitivity for 10 dB SNR is√(5/0.27). The 1.9 µV AM sensitivity thereforetranslates to 1.42 µV for SSB and 0.44 µV forCW. If only the CW version is given, then thereceiver might be made to look a whole lotbetter than it is, even though the typical usermay never use the CW mode (note differ-ences in Figure 3.6).

The sensitivity differences also explainwhy weak SSB signals can be heard underconditions when AM signals of similarstrength have disappeared into the noise orwhy the CW mode has as much as 20 dB ad-vantage over SSB, ceterus paribus.

In some receivers, the difference inmode (AM, SSB, RTTY, CW, etc.) conceivablycan result in sensitivity differences that aremore than the differences in the bandwidths


Fig. 3.6 Sensitivity vs. bandwidth.

0

0.2

0.4

0.6

0.8

1

1.2

1.4

1.6

1.8

2

0 1000 2000 3000 4000 5000 6000

BANDWIDTH (Hz)

SE

NS

ITIV

ITY

(µV

)

associated with the various modes. The rea-son is that sometimes a “processing gain” isassociated with the type of detector circuitused to demodulate the signal at the outputof the IF amplifier. A simple AM envelopedetector is lossy because it consists of a sim-ple diode (e.g., 1N60) and an R-C filter (apassive circuit without amplification). Otherdetectors (product detector for SSB, synchro-nous AM detectors) have their own signalgain, so they may produce better sensitivitynumbers than the bandwidth suggests.

Another indication of sensitivity is mini-mum detectable signal (MDS), usually speci-fied in dBm. This signal level is the signalpower at the antenna input terminal of the re-ceiver required to produce some standardS+N/N ratio, such as 3 dB or 10 dB (Figure3.7). In radar receivers, the MDS usually is de-scribed in terms of a single pulse return and aspecified S+N/N ratio. Also, in radar receivers,the sensitivity can be improved by integratingmultiple pulses. If N return pulses are inte-grated, then the sensitivity is improved by afactor of N if coherent detection is used, and√(N) if noncoherent detection is used.

Modulated signals represent a specialcase. For those sensitivities, it is common to

specify the conditions under which the mea-surement is made. For example, in AM re-ceivers, the sensitivity to achieve 10 dB SNRis measured with the input signal modulated30% by a 400 or 1000 Hz sinusoidal tone.

An alternate method is sometimes usedfor AM sensitivity measurements, especially inservicing consumer radio receivers (whereSNR may be a little hard to measure with theequipment normally available to technicianswho work on those radios). This is the “stan-dard output conditions” method. Some manu-als specify the audio signal power or audiosignal voltage at some critical point, when the30% modulated RF carrier is present. In oneautomobile radio receiver, the sensitivity wasspecified as “X µV to produce 400 mW across8 Ω resistive load substituted for the loud-speaker when the signal generator is modu-lated 30% with a 400 Hz audio tone.” Thecryptic note on the schematic showed an out-put sine wave across the loudspeaker withthe label “400 mW in 8 Ω (1.79 volts), @30%mod. 400 Hz, 1 µV RF.” What is missing ismention of the level of total harmonic distor-tion (THD) permitted.

The sensitivity sometimes is measuredin essentially the same way, but the signal


Fig. 3.7SNR under two circumstances.

dB

-5

0

+3

+10

NOISE FLOOR (0 dB)

10 dB SNR

3 dB SNRS

IGN

AL

LEV

EL

(dB

)

6 dB6 dB

levels specify the voltage level that will ap-pear at the top of the volume control or out-put of the detector/filter, when the standardsignal is applied. Thus, there are two waysseen for specifying AM sensitivity: 10 dB SNRand standard output conditions.

There also are two ways to specify FMreceiver sensitivity. The first is the 10 dB SNRmethod just discussed; that is, the number ofmicrovolts of signal at the input terminals re-quired to produce a 10 dB SNR when the car-rier is modulated by a standard amount. Themeasure of FM modulation is deviation ex-pressed in kilohertz. Sometimes, the full de-viation for that class of receiver is used,while for others a value that is 25–35% fulldeviation is specified.

The second way to measure FM sensitiv-ity is the level of signal required to reduce theno-signal noise level by 20 dB. This is the 20dB quieting sensitivity of the receiver. If youtune between signals on an FM receiver, youwill hear a loud “hiss” signal, especially in theVHF/UHF bands. Some of that noise is gener-ated externally, while some is generated in-ternally. When an FM signal appears in thepassband, that hiss is suppressed, even if theFM carrier is unmodulated. The quieting sen-sitivity of an FM receiver is a statement of thenumber of microvolts required to producesome standard quieting level, usually 20 dB.

Pulse receivers, such as radar and pulsecommunications units, often use the tangen-tial sensitivity as the measure of performance,which is the amplitude of pulse signal re-quired to raise the noise level by its own RMSamplitude (Figure 3.8).

Selectivity

Although no receiver specification is unim-portant, if one had to choose between sensi-tivity and selectivity, the proper choice mostof the time would be selectivity. Selectivity isthe measure of a receiver’s ability to rejectadjacent channel interference. Or, to put an-other way, it is the ability to reject interfer-ence from signals on frequencies close to thedesired signal frequency.

To understand selectivity requirements,one must first understand a little bit of thenature of radio signals. An unmodulated ra-dio carrier theoretically has an infinitesimal(near-zero) bandwidth (although all real un-modulated carriers have a very narrow, butnonzero, bandwidth, because they are mod-ulated by noise and other artifacts). As soonas the radio signal is modulated to carry in-formation, however, the bandwidth spreads.Even an on/off telegraphy (CW) or pulse sig-nal spreads out either side of the carrier fre-quency an amount that is dependent on the


Fig. 3.8Tangential sensitivity.

V1

V2

VA

RMS NOISE FLOOR

OUTPUTSIGNALLEVEL

INPUTSIGNALLEVEL

PULSE

sending speed and the shape of the keyingwaveform.

An AM signal spreads out an amountequal to twice the highest audio modulatingfrequencies. For example, a communicationsAM transmitter will have audio componentsfrom 300–3000 Hz, so the AM waveform willoccupy a spectrum equal to the carrier fre-quency (F) plus or minus the audio band-width (F ± 3000 Hz in the case cited). An FMcarrier spreads out according to the deviation.For example, a narrow-band FM landmobiletransmitter with 5 kHz deviation spreads out±5 kHz, while FM broadcast transmittersspread out ±75 kHz

An implication of the fact that radio sig-nals have bandwidth is that the receiver musthave sufficient bandwidth to recover all thesignal. Otherwise, information may be lost andthe output is distorted. On the other hand, al-lowing too much bandwidth increases thenoise picked up by the receiver and therebydeteriorates the SNR. The goal of the selectivitysystem of the receiver is to match the band-width of the receiver to that of the signal. Thatis why receivers will use 270 or 500 Hz band-width for CW, 2–3 kHz for SSB, and 4–6 kHzfor AM signals. They allow matching the re-ceiver bandwidth to the transmission type.

The selectivity of a receiver has a num-ber of aspects that must be considered: front-end bandwidth, IF bandwidth, IF shape factor,and the ultimate (distant frequency) rejection.

FRONT-END BANDWIDTH

The “front end” of a modern superheterodyneradio receiver is the circuitry between the an-tenna input terminal and the output of the firstmixer stage. The reason why front-end selec-tivity is important is to keep out-of-band sig-nals from afflicting the receiver. Transmitterslocated nearby easily can overload a poorlydesigned receiver. Even if these signals arenot heard by the operator, they can desensi-tize a receiver or create harmonics and inter-modulation products that show up as “birdies”or other types of interference on the receiver.Strong local signals can take up a lot of the re-ceiver’s dynamic range and make it harder tohear weak signals.

In some crystal video microwave re-ceivers, that front end might be wide openwith no selectivity at all, but in nearly all otherreceivers some form of frequency selectionwill be present.

Two forms of frequency selection typi-cally are found. A designer may choose touse only one in a design. Alternatively, bothmight be used in the design but separately(operator selection). Or, finally, both mightbe used together. These forms can be calledthe resonant frequency filter (Figure 3.9A)and bandpass filter (Figure 3.9B) approaches.

The resonant frequency approach usesL-C elements tuned to the desired frequencyto select which RF signals reach the mixer.In some receivers, these L-C elements aredesigned to track with the local oscillatorthat sets the operating frequency. That iswhy you see two-section (or three-section)


Fig. 3.9 Filters with (A) narrow bandwidth and(B) wider bandwidth.

A

FREQUENCY

F

A

AM

PLI

TU

DE

SIGNALS

B

FREQUENCY

F

A

AM

PLI

TU

DE

variable capacitors for AM broadcast receiverswith two different capacitance ranges for thesections. One section tunes the LO and theother section tunes the tracking RF input. Inother designs, a separate tuning knob (“pre-selector” or “antenna”) is used.

The other approach uses a suboctavebandpass filter to admit only a portion ofthe RF spectrum into the front end. Forexample, a shortwave receiver that is de-signed to take the HF spectrum in 1 MHzpieces may have an array of RF input band-pass filters each of which is 1 MHz wide(e.g., 9–10 MHz).

In addition to the reasons cited previ-ously, front-end selectivity also helps im-prove a receiver’s image rejection and 1st IFrejection capabilities.

IMAGE REJECTION

An image in a superheterodyne receiver is asignal that appears at twice the IF distancefrom the desired RF signal and is located onthe opposite side of the LO frequency from thedesired RF signal. In Figure 3.10, a super-heterodyne operates with a 455 kHz (i.e.,0.455 MHz) IF and is turned to 24.0 MHz (FRF).Because this receiver uses low-side LO injec-tion, the LO frequency, FLO, is 24.0 − 0.455, or23.545 MHz. If a signal appears at twice the IFbelow the RF (i.e., 910 kHz below FRF) andreaches the mixer, then it too has a differencefrequency of 455 kHz, so it will pass rightthrough the IF filtering as a valid signal. Theimage rejection specification tells how well thisimage frequency is suppressed. Normally, any-thing over about 70 dB is considered good.

The tactics to reduce image responsevary with the design of the receiver. The bestapproach, at design time, is to select an IF fre-quency high enough that the image frequencywill fall outside the passband of the receiverfront end. Some HF receivers use an IF of 8.83MHz, 9 MHz, 10.7 MHz, or something similar;and for image rejection, these frequencies areconsiderably better than 455 kHz receivers inthe higher HF bands. However, a common de-sign trend is to do double conversion. In mostsuch designs, the first IF frequency is consid-erably higher than the RF, being in the range35–60 MHz (50 MHz is common in HF re-ceivers, 70 MHz in microwave receivers).

The high IF makes it possible to sup-press the VHF images with a simple low-passfilter. If the 24.0 MHz signal (described previ-ously) were first up converted to 50 MHz (74MHz LO), for example, the image would be at124 MHz. The second conversion brings the IFdown to one of the frequencies just men-tioned or even 455 kHz. The lower frequen-cies are preferable to 50 MHz for bandwidthselectivity reasons because good-quality crys-tal, ceramic, or mechanical filters in the lower-frequency ranges filters are readily available.

1ST IF REJECTION

The 1st IF rejection specification refers tohow well a receiver rejects radio signals op-erating on the receiver’s first IF frequency.For example, if your receiver has a first IF of50 MHz, it must be able to reject radio signalsoperating on that frequency when the re-ceiver is tuned to a different frequency.Although the shielding of the receiver also is


Fig. 3.10RF, LO, and two IF frequencies.

FLO

FRFFIMG

23.09MHz

23.545MHz

24MHz

0.455MHz

FIF

455kHz

455kHz

910kHz

F

an issue with respect to this performance, thefront-end selectivity affects how well the re-ceiver performs against 1st IF signals.

If there is no front-end selectivity to dis-criminate against signals at the IF frequency,then these signals arrive at the input of themixer unimpeded. Depending on the designof the mixer, they then may pass directlythrough to the high-gain IF amplifiers and beheard in the receiver output.

IF BANDWIDTH

Most of the selectivity of the receiver is pro-vided by the filtering in the IF amplifier sec-tion. The filtering might use L-C filters(especially if the principal IF is a low fre-quency, like 50 kHz), a ceramic resonator, acrystal filter, or a mechanical filter. Of these,the mechanical filter usually is regarded bestfor narrow bandwidths, with the crystal filterand ceramic filters coming in next.

The IF bandwidth is expressed in kilo-hertz and measured from the points on the IFfrequency response curve where gain dropsoff −3 dB from the mid-band value (Figure3.11). This is why you sometimes see selec-tivity specified in terms such as “6 kHz be-tween −3 dB points.”

The IF bandwidth must be matched tothe bandwidth of the received signal for bestperformance. If a too-wide bandwidth is se-lected, then the received signal will be noisyand the SNR will deteriorate. If too narrow,

then you might experience difficulties recov-ering all of the information transmitted. Forexample, an AM broadcast band radio signalhas audio components out to 5 kHz, so thesignal occupies up to 10 kHz of spectrumspace (F ± 5 kHz). If a 2.8 kHz SSB IF filter isselected, then it will tend to sound “mushy”and distorted.

IF PASSBAND SHAPE FACTOR

The shape factor is a measure of the steep-ness of the receiver’s IF passband, taken bymeasuring the ratio of the bandwidth at −6dB to the bandwidth at −60 dB (Figure3.12A). The general rule is that the closerthese numbers are to each other, the betterthe receiver. Anything in the 1:1.5 to 1:1.9region can be considered high quality,while anything worse than 1:3 is not worthlooking at for “serious” receiver uses. If thenumbers are between 1:1.9 and 1:3, thenthe receiver could be regarded as middlingbut useful.

The importance of the shape factor isthat it modifies the notion of bandwidth. Thecited bandwidth (e.g., 2.8 kHz for SSB) doesnot take into account the effects of strong sig-nals just beyond those limits. Such signals caneasily “punch through” the IF selectivity if theIF passband “skirts” are not steep. After all,the steeper they are, the closer a strong signalcan be without messing up the receiver’s op-eration. The situation is illustrated in Figure3.12B. The curve inverts Figure 3.12A by plot-ting attenuation vs. frequency. Assume thatequal amplitude signals close to Fo are re-ceived (Figure 3.12C), the relative postfilter-ing amplitudes will match Figure 3.12D.Therefore, selecting a receiver with a shapefactor as close to the 1:1 ideal as possible willresult in a more usable radio.

DISTANT FREQUENCY (“ULTIMATE”) REJECTION

This specification tells something about thereceiver’s ability to reject very strong signalslocated well outside the receiver’s IF pass-band. This number is stated in negative deci-bels; and the higher is the number, the better.An excellent receiver will have values in the−60 to −90 dB range, a middling receiver will


Fig. 3.11 Definition of bandwidth.

FOFL FH

dB

0 dB

-3 dB

BWBW

see numbers in the −45 to −60 dB range, anda terrible receiver will be −44 or worse.

Stability

The stability specification measures howmuch the receiver frequency drifts as timeelapses or temperature changes. The LO driftsets the overall stability of the receiver. Thisspecification usually is given in terms ofshort-term drift and long-term drift (e.g.,from LO crystal aging). The short-term driftis important in daily operation, while thelong-term drift ultimately affects general dialcalibration.

If the receiver is variable frequency os-cillator (VFO) controlled or uses partial fre-quency synthesis (which combines VFOswith crystal oscillators), then the stability isdominated by the VFO stability. In fully syn-thesized receivers, the stability is governedby the master reference crystal oscillator. Ifeither an oven-controlled crystal oscillator(OCXO) or a temperature-compensated crys-tal oscillator (TCXO) is used for the masterreference, then stability on the order of 1 partin 108/°C is achievable.

For most users, the short-term stabilityis what is most important, especially whentuning SSB, ECSS or RTTY signals. A com-mon specification value for a good receiverwill be 50 Hz/hr after a 3 hr warm-up, or 100

Hz/hr after a 15 min warm-up. The smallerthe drift, the better is the receiver.

The foundation of good stability is de-termined at design time. The local oscillator,or VFO portion of a synthesizer, must be op-erated in a cool, temperature-stable locationwithin the equipment and must have thecorrect type of components. Capacitor tem-perature coefficients are selected to cancelout temperature-related drift in inductancevalues.


Fig. 3.12 Shape factor (SF): (A) defined; (B) attenuation vs. frequency; (C) input signal; (D) what gets through.

0 dB

-6 dB

-60 dB

BW6

BW60

60

6

BW

BWFS =

F

ATT

EN

UAT

ION

FOFO

A

B

C

D

Postdesign changes can help, but theseare less likely to be possible today than inthe past. The chief cause of drift problems isheat. In the days of vacuum tube oscillators,the internal heating filament produced lots ofheat, which in turn created drift.

A related phenomenon seen on low-cost receivers, or certain homebrew receiversof doubtful merit, is mechanical frequencyshifts. Although not seen on most modern re-ceivers (even some very cheap designs), itonce was a serious problem on less costlymodels. This problem usually is seen on VFOcontrolled receivers in which vibration to thereceiver cabinet imparts movement to eitherthe inductor (L) or capacitor (C) element inan L-C VFO. Mechanically stabilizing thesecomponents will work wonders.

AGC Range and Threshold

Modern communications receivers must beable to handle signal strengths over a dynamicrange of about 1,000,000:1. Tuning across aband occupied by signals of wildly varyingstrengths is hard on the ears and hard on thereceiver’s performance. As a result, mostmodern receivers have an automatic gaincontrol (AGC) circuit that smoothes thesechanges. The AGC will reduce gain forstrong signals and increase it for weak sig-nals (AGC can be turned off on most HFcommunications receivers). The AGC rangeis the change of input signal (in dBµV) fromsome reference level (e.g., 1 µVEMF) to the in-put level that produces a 2 dB change in out-put level. Ranges of 90–110 dB commonlyare seen.

The AGC threshold is the signal level atwhich the AGC begins to operate. If set toolow, then the receiver gain will respond tonoise and irritate the user. If set too high,then the user will experience irritating shiftsof output level as the band is tuned. AGCthresholds of 0.7–2.5 µV are common on de-cent receivers, with the better receivers beingin the 0.7 to 1 µV range.

Another AGC specification sometimesseen deals with the speed of the AGC.Although it may be specified in milliseconds,

it also frequently is specified in subjectiveterms like fast and slow. This specificationrefers to the speed at which the AGC re-sponds to changes in signal strength. If settoo fast, then rapidly keyed signals (e.g.,on/off CW) or noise transients will cause un-nerving large shifts in receiver gain. If settoo slow, then the receiver might as wellhave no AGC. Many receivers provide twoor more selections to accommodate differenttypes of signals.

DYNAMIC PERFORMANCE

The dynamic performance specifications of aradio receiver are those that deal with howthe receiver performs in the presence of verystrong signals from either a cochannel or ad-jacent channel. Until about the 1960s, dynamicperformance was somewhat less importantthan static performance for most users.However, today the role of dynamic perfor-mance is probably more critical than staticperformance because of the crowded bandconditions.

There are at least two reasons for thischange in outlook. First, in the 1960s, receiverdesigns evolved from tubes to solid state. Thenew solid-state amplifiers were somewhateasier to drive into nonlinearity than tube de-signs. Second, the radio frequency signals onthe air have increased considerably. Far morestations are transmitting than ever before, andthere are far more sources of electromagneticinterference (EMI—pollution of the air waves)than in prior decades. With the advent of newand expanded wireless services available toan ever widening market, the situation canonly worsen. For this reason, it now is neces-sary to pay more attention to the dynamic per-formance of receivers than in the past.

Intermodulation Products

Understanding the dynamic performance ofthe receiver requires knowledge of intermod-ulation products (IP) and how they affect receiver operation. Whenever two signals are mixed together in a nonlinear circuit, a


number of products are created according tothe mF1 ± nF2 rule, where m and n are ei-ther integers or zero (0, 1, 2, 3, 4, 5, . . .).Mixing can occur in either the mixer stage ofa receiver front end or in the RF amplifier (orany outboard preamplifiers used ahead ofthe receiver) if the RF amplifier is overdrivenby a strong signal.

Also theoretically possible is for corro-sion on antenna connections or even rustedantenna screw terminals to create IPs undercertain circumstances. In some alleged cases,a rusty downspout on a house rain guttercaused reradiated mixed signals.

The spurious IP signals are showngraphically in Figure 3.13. The order of theproduct is given by the sum (m + n). Giveninput signal frequencies of F1 and F2, themain IPs are

Second-order: F1 ± F22F12F2

Third-order: 2F1 ± F22F2 ± F1

(3.12)3F13F2

Fifth-order: 3F1 ± 2F23F2 ± 2F15F15F2

When an amplifier or receiver is over-driven, the second-order content of the out-

put signal increases as the square of the in-put signal level, while the third-order re-sponses increase as the cube of the inputsignal level.

Consider the case where two HF sig-nals, F1 = 10 MHz and F2 = 15 MHz, aremixed. The second-order IPs are 5 and 25MHz; the third-order IPs are 5, 20, 35, and 40MHz; and the fifth-order IPs are 0, 25, 60,and 65 MHz. Any of these that are inside thepassband of the receiver can cause prob-lems. One such problem is the emergence of“phantom” signals at the IP frequencies. Thiseffect often is seen when two strong signals(F1 and F2) exist and affect the front end ofthe receiver, and one of the IPs falls close toa desired signal frequency, Fd. If the receiverwere tuned to 5 MHz, for example, a spuri-ous signal would be found from the F1–F2pair just given.

Another example is seen from strongin-band, adjacent channel signals. Considerthe case where the receiver is tuned to a sta-tion at 9610 kHz and very strong signals alsoare at 9600 kHz and 9605 kHz. The near (in-band) IP products are

Third order: 9595 kHz (∆F = 15 kHz)9610 kHz (∆F = 0 kHz)

[on channel] (3.13)

Fifth order: 9590 kHz (∆F = 20 kHz)9615 kHz (∆F = 5 kHz)

Note that one third-order product is onthe same frequency as the desired signal and


Fig. 3.13 Intermodulation distortion (IMD) levels.

1MHz

2MHz

3MHz

4MHz

5MHz

6MHz

7MHz

8MHz

F

FREQUENCY (MHz)

easily could cause interference if the ampli-tude were sufficiently high. Other third- andfifth-order products may be within the rangewhere interference could occur, especiallyon receivers with wide bandwidths.

The IP orders theoretically are infinite,because no bounds are placed on either m orn. However, in practical terms, because eachsuccessively higher order IP is reduced inamplitude compared with its next lower or-der mate, only the second-order, third-order,and fifth-order products usually assume anyimportance. Indeed, only the third order nor-mally is shown in receiver specificationssheets because they fall close to the RF signalfrequency.

A large number of IMD products fromjust two signals are applied to a nonlinearmedium. But consider that the two-tone caseused for textbook discussions rarely is en-countered in actuality. A typical two-way ra-dio installation is in a signal rich environment,so when dozens of signals are present, thenumber of possible combinations climbs to anunmanageable extent.

-1 dB Compression Point

An amplifier produces an output signal thathas a higher amplitude than the input signal.The transfer function of the amplifier (indeed,any circuit with output and input) is the ratioout/in; so for the power amplification of a re-ceiver RF amplifier, it is Po /Pin (or, in terms ofvoltage, Vo /Vin). Any real amplifier will satu-rate, given a strong enough input signal (seeFigure 3-14). The dashed line represents thetheoretical output level for all values of inputsignal (the slope of the line represents thegain of the amplifier). As the amplifier satu-rates (solid line), however, the actual gain be-gins to depart from the theoretical at somelevel of input signal (Pin1). The −1 dB com-pression point is that output level at which theactual gain departs from the theoretical gainby −1 dB.

The −1 dB compression point is impor-tant when considering either the RF amplifierahead of the mixer (if any) or any outboard

preamplifiers used. The −1 dB compressionpoint is the point at which intermodulationproducts begin to emerge as a serious prob-lem. Also, harmonics are generated when anamplifier goes into compression. A sine waveis a “pure” signal because it has no harmon-ics (all other waveshapes have a fundamentalfrequency plus harmonic frequencies). Whena sine wave is distorted, however, harmonicsarise. The effect of the compression phenom-enon is to distort the signal by clipping thepeaks, thus raising the harmonics and inter-modulation distortion products.

Third-Order Intercept Point

The third-order intercept point (TOIP) couldbe the single most important specification ofa receiver’s dynamic performance because itpredicts the performance as regards inter-modulation, cross-modulation, and blockingdesensitization.

Third-order (and higher) intermodula-tion products normally are very weak anddo not exceed the receiver noise floor whenthe receiver is operating in the linear region.But, as input signal levels increase, forcingthe front end of the receiver toward the sat-urated nonlinear region, the IPs emergefrom the noise (Figure 3.15) and begin tocause problems. When this happens, new


Fig. 3.14 The −1 dB compression point.

INPUT SIGNAL LEVEL

OU

TP

UT

SIG

NA

L LE

VE

L

1 dB

PO

PIN

spurious signals appear on the band andself-generated interference begins to arise.

Figure 3.16 shows a plot of the outputsignal vs. fundamental input signal. Note theoutput compression effect that was seen ear-lier in Figure 3.13. The dashed gain line con-tinuing above the saturation region showsthe theoretical output that would be pro-duced if the gain did not clip. It is the natureof third-order products in the output signal toemerge from the noise at a certain input leveland increase as the cube of the input level.Therefore, the slope of the third-order lineincreases 3 dB for every 1 dB increase in theresponse to the fundamental signal. Althoughthe output response of the third-order linesaturates like that of the fundamental signal,the gain line can be continued to a pointwhere it intersects the gain line of the funda-mental signal. This point is the third-order in-tercept point.

Interestingly enough, one receiver fea-ture that can help reduce IP levels back un-


Fig. 3.16 Third-order intercept point.

INPUT SIGNAL LEVEL

OU

TP

UT

SIG

NA

L LE

VE

L

PO

PIN

FUNDAMENTALSIGNAL

THIRD-ORDERSIGNAL

TOIP

Fig. 3.15Above the compression point, thenoise floor goes up.

INPUT SIGNAL LEVEL

OU

TP

UT

SIG

NA

L LE

VE

L

A

B

IM PRODUCTS EMERGEFROM NOISE FLOOR

NOISE FLOOR

IM PRODUCTSBELOW NOISE

FLOOR

der the noise is the use of a front-end atten-uator (also called an input attenuator). Inthe presence of strong signals, even a fewdB of input attenuation often is enough todrop the IPs back into the noise, while af-flicting the desired signals only a smallamount.

Other effects that reduce the overloadcaused by a strong signal also help. Situationsarise where the apparent third-order perfor-mance of a receiver improves dramaticallywhen a lower-gain antenna is used. This effectcan be demonstrated easily using a spectrumanalyzer for the receiver. This instrument is aswept frequency receiver that displays an out-put on an oscilloscope screen showing ampli-tude vs. frequency, so a single signal shows asa spike. In one test, a strong, local VHF bandrepeater came on the air every few seconds,and I could see the second- and third-orderIPs along with the fundamental repeater sig-nal. Other strong signals also were on the airbut just outside the band. Inserting a 6 dBbarrel attenuator in the input (“antenna”) lineeliminated the IP products, showing just theactual signals. Rotating a directional antennaaway from the direction of the interfering sig-nal also would accomplish this effect inmany cases.

Preamplifiers are popular receiver ac-cessories but can reduce rather than enhanceperformance. Two problems commonly oc-cur (assuming the preamp is a low-noise de-vice). The best known problem is that thepreamp amplifies noise as much as signals:While it makes the signal louder, it alsomakes the noise louder by the same amount.Since the signal-to-noise ratio is important,this does not improve the situation. Indeed,if the preamp is itself noisy, it will deterioratethe SNR. The other problem is less wellknown but potentially more devastating: Ifthe increased signal levels applied to the re-ceiver drive the receiver nonlinear, then IPsbegin to emerge.

When evaluating receivers, a TOIP of+5 to +20 dBm shows excellent performance,while up to +27 dBm is relatively easilyachieved, and +35 dBm has been achieved

with good design; anything greater than +50dBm is close to miraculous (but attainable).Receivers still are regarded good performersin the 0 to +5 dBm range and middling per-formers in the −10 to 0 dBm range. Anythingbelow −10 dBm usually is unacceptable. Ageneral rule is to buy the best third-order in-tercept performance that you can afford, es-pecially if there are strong signal sources inyour vicinity.

Dynamic Range

The dynamic range of a radio receiver is therange (measured in decibels) from the mini-mum discernible signal to the maximum al-lowable signal. While this simplistic definitionis conceptually easy to understand, in theconcrete it’s a little more complex. Severaldefinitions of dynamic range are used.

One definition of dynamic range is thatit is the input signal difference between thesensitivity figure (e.g., 0.5 µV for 10 dBS+N/N) and the level that drives the receiverfar enough into saturation to create a certainamount of distortion in the output. This defi-nition was common on consumer broadcastband receivers at one time (especially auto-mobile radios, where dynamic range wassomewhat more important due to mobility).A related definition takes the range as thedistance in dB from the sensitivity level andthe −1 dB compression point. Still anotherdefinition, the blocking dynamic range, is therange of signals from the sensitivity level tothe blocking level (see later).

A problem with these definitions is thatthey represent single signal cases, so they donot address the receiver’s dynamic character-istics. Two other definitions, one “loose” andone more formal, are somewhat more usefuland at least are standardized. The loose ver-sion is that dynamic range is the range of signals over which dynamic effects (e.g., in-termodulation) do not exceed the noise floorof the receiver. For HF receivers the recom-mended dynamic range is usually two thirdsthe difference between the noise floor and


the third-order intercept point in a 3 kHzbandwidth.

The alternative definition is that the dy-namic range is the difference between thefundamental response input signal level andthe third-order intercept point along thenoise floor, measured with a 3 kHz band-width. For practical reasons, this measure-ment sometimes is made not at the actualnoise floor (which may be hard to ascertain)but at 3 dB above the noise floor.

A certain measurement procedure pro-duces similar results (the same method isused for many amateur radio magazineproduct reviews). Two equal strength sig-nals are input to the receiver at the sametime. The frequency difference traditionallyhas been 20 kHz for HF and 30–50 kHz forVHF receivers (modern band crowding mayindicate a need for a specification at 5 kHzseparation on HF). The amplitudes of thesesignals are raised until the third-order dis-tortion products are raised to the noisefloor level. For 20 kHz spacing, using thetwo-signal approach, anything over 90 dBis an excellent receiver, and anything over80 dB is at least decent.

The difference between the single-signaland two-signal (dynamic) performance isnot merely an academic exercise. In addi-tion, the same receiver can show as muchas 40 dB difference between the two mea-sures (favoring the single-signal measure-ment), so the most severe effects of poordynamic range show up most in the dy-namic performance.

Blocking

The blocking specification refers to the abilityof the receiver to withstand very strong off-tune signals that are at least 20 kHz awayfrom the desired signal, although some use a100 kHz separation. When very strong signalsappear at the input terminals of a receiver,they may desensitize the receiver; that is, re-duce the apparent strength of desired signalsover what they would be if the interfering sig-nal were not present.

Figure 3.17 shows the blocking behav-ior. When a strong signal is present, it takesup more of the receiver’s resources than nor-mal, so not enough of the output power isleft to accommodate the weaker desired sig-nals. But, if the strong undesired signal isturned off, then the weaker signals receive afull measure of the unit’s power budget.

The usual way to measure blocking isto input two signals, a desired signal at 60dBµV and another signal 20 (or 100) kHzaway at a much stronger level. The strongsignal is increased to the point where block-ing desensitization causes a 3 dB drop in theoutput level of the desired signal. A good re-ceiver will show ≥90 dBµV, with many con-siderably better. An interesting note aboutmodern receivers is that the blocking perfor-mance is so good that it often is necessary tospecify the input level difference (dB) thatcauses a 1 dB drop, rather than a 3 dB drop,of the desired signal’s amplitude.

The phenomenon of blocking leads usto an effect often seen as paradoxical on firstblush. Many receivers are equipped withfront-end attenuators that permit fixed attenu-ation values of 1 dB, 3 dB, 6 dB, 12 dB, or 20dB (or some subset) to be inserted into thesignal path ahead of the active stages. Whena strong signal, capable of causing desensiti-zation, is present, adding attenuation often in-creases the level of the desired signals in theoutput, even though overall gain is reduced.This occurs because the overall signal the re-ceiver front end is asked to handle is belowthe threshold where desensitization occurs.

Cross-Modulation

Cross-modulation is an effect in which ampli-tude modulation (AM) from a strong undesiredsignal is transferred to a weaker desired signal.Testing usually is done (in HF receivers) with a20 kHz spacing between the desired and un-desired signals, a 3 kHz IF bandwidth on thereceiver, and the desired signal set to 1000µVEMF (−53 dBm). The undesired signal (20kHz away) is amplitude modulated to the 30%level. This undesired AM signal is increased in


strength until an unwanted AM output 20 dBbelow the desired signal is produced.

A cross-modulation specification of ≥100dB would be considered decent performance.This figure often is not given for modern HFreceivers, but if the receiver has a good third-order intercept point, then it is likely to alsohave good cross-modulation performance.

Cross-modulation is said to occur natu-rally, especially in transpolar and NorthAtlantic radio paths, where the effects of theaurora are strong. One (possibly apocryphal)legend tells of something called the RadioLuxembourg effect, discovered in the 1930s.Modulation from a very strong broadcaster(BBC) appeared on the Radio Luxembourgsignal received in North America. This effect was said to be an ionospheric cross-

modulation phenomenon and apparentlyoccurs when the strong station is within 175miles of the great circle path between thedesired station and the receiver site.

Reciprocal Mixing

Reciprocal mixing occurs when noise side-bands from the local oscillator (LO) signal in asuperheterodyne receiver mix with a strongundesired signal that is close to the desired sig-nal. Every oscillator signal produces noise, andthat noise tends to amplitude modulate the os-cillator’s output signal. It thus forms sidebandson either side of the LO signal. The productionof phase noise in all LOs is well known, but inmore recent designs, the digitally producedsynthesized LOs are prone to additional noise


Fig. 3.17Blocking: (A) desensitized signal;(B) nondesensitized signal.

STRONGINTERFERING

SIGNALDESIREDSIGNAL

FREQUENCY

AM

PLI

TU

DE

OVERALLAMPLITUDE

PERCEIVEDAMPLITUDE

STRONGINTERFERING

SIGNALREDUCED INAMPLITUDE

DESIREDSIGNAL

FREQUENCY

AM

PLI

TU

DE

OVERALLAMPLITUDE PERCEIVED

AMPLITUDE

A

B

elements as well. The noise is usually mea-sured in −dBc (decibels below carrier or, inthis case, dB below the LO output level).

In a superheterodyne receiver, the LObeats with the desired signal to produce anintermediate frequency (IF) equal to eitherthe sum (LO + RF) or difference (LO − RF) ofthe two. If present, a strong unwanted signalmight mix with the noise sidebands of theLO to reproduce the noise spectrum at the IFfrequency (see Figure 3.18). In the usual testscenario, the reciprocal mixing is defined asthe level of the unwanted signal (dB) at 20kHz required to produce noise sidebands 20dB down from the desired IF signal in aspecified bandwidth (usually 3 kHz on HFreceivers). Figures of −90 dBc or better areconsidered good.

The importance of the reciprocal mix-ing specification is that such mixing can seri-ously deteriorate the observed selectivity ofthe receiver yet not be detected in the nor-mal static measurements made of selectivity(it is a “dynamic selectivity” problem). Whenthe LO noise sidebands appear in the IF, thedistant frequency attenuation (>20 kHz off-center of a 3 kHz bandwidth filter) can dete-riorate 20 to 40 dB.

The reciprocal mixing performance ofreceivers can be improved by eliminating thenoise from the oscillator signal. Although thissounds simple, in practice, it often is quitedifficult. A tactic that works well is to addhigh-Q filtering between the LO output andthe mixer input. The narrow bandwidth of

the high-Q filter prevents excessive noisesidebands from getting to the mixer. Althoughthis sounds like a quite easy solution, as theysay “the devil is in the details.”

IF Notch Rejection

If two signals fall within the passband of a re-ceiver, both will compete to be heard. Theyalso will heterodyne together in the detectorstage, producing an audio tone equal to theircarrier frequency difference. For example, sup-pose we have an AM receiver with a 5 kHzbandwidth and a 455 kHz IF. If two signals ap-pear on the band such that one appears at anIF of 456 kHz and the other is at 454 kHz, thenboth are within the receiver passband andboth will be heard in the output. However, the2 kHz difference in their carrier frequency willproduce a 2 kHz heterodyne audio tone differ-ence signal in the output of the AM detector.

In some receivers, a tunable, high-Q(narrow and deep) notch filter is in the IFamplifier circuit. This tunable filter can beturned on and adjusted to attenuate the un-wanted interfering signal, reducing the irritat-ing heterodyne. Attenuation figures for goodreceivers vary from −35 to −65 dB or so (themore negative the better).

Some trade-offs are made in notch filterdesign. First, the notch filter Q is achievedmore easily at low IF frequencies (such as50–500 kHz) than at high IF frequencies (e.g.,9 MHz and up). Also, the higher is the Q, thebetter the attenuation of the undesired squeal


Fig. 3.18LO phase transfers to an IF signal.

LOSIGNAL

IF SIGNAL

LO PHASENOISETRANSFERRED

LO PHASE NOISE

but the touchier it is to tune. Some happymiddle ground between the irritating squealand the touchy tune is mandated here.

Some receivers use audio filters ratherthan IF filters to help reduce the heterodynesqueal. In the AM broadcast band, channelspacing is typically 8–10 kHz (depending onthe part of the world), and the transmitted au-dio bandwidths (hence the sidebands) are 5kHz. Designers of AM BCB receivers usuallyinsert an R-C low-pass filter with a −3 dBpoint just above 4 or 5 kHz, right after the de-tector, to suppress the audio heterodyne. ThisR-C filter is called a tweet filter in the slang ofthe electronic service and repair trade.

Another audio approach is to sharplylimit the bandpass of the audio amplifiers.For AM BCB reception, a 5 kHz bandpass issufficient, so the frequencies higher can berolled off at a fast rate to produce only asmall response an octave higher (10 kHz). Inshortwave receivers, this option is weakerbecause the station channels typically are 5kHz, and many do not bother to honor theofficial channels anyway. On the amateur ra-dio bands, frequency selection is a perpetu-ally changing ad-hocracy, at best. Although

the shortwave bands typically need only a 3kHz bandwidth for communications and 5kHz for broadcast, the tweet filter and audioroll-off might be insufficient. In receivers thatlack an effective IF notch filter, an audionotch filter can be provided.

Internal Spurii

All receivers produce a number of internalspurious signals that sometimes interfere withthe operation. Both old and modern receivershave spurious signals from assorted high-order mixer products, power supply harmon-ics, parasitic oscillations, and a host of othersources. Newer receivers with either (or both)synthesized local oscillators and digital fre-quency readouts produce noise and spurioussignals in abundance. (Note: Low-power digi-tal chips with slower rise times—CMOS,NMOS, and the like—generally are muchcleaner than high-power, fast rise time chips,like TTL devices.) With appropriate filteringand shielding, it is possible to hold down the“spurs” to −100 dB relative to the main maxi-mum signal output, or within about 3 dB orthe noise floor, whichever is lower.


The front end of the receiver is key to its dy-namic performance. Specifications such as thedynamic range, intermodulation distortion, −1dB compression point, and third-order inter-cept point demonstrate how well the frontend of the receiver performs.

FRONT-END ARCHITECTURES

Several different architectures are used in re-ceiver front-end circuits. Figure 4.1 shows

the simplest form of front-end architecture. Itconsists of a mixer stage and local oscillatorpreceded by a bandpass filter. The input sig-nal to the bandpass filter will come from theantenna. The bandpass filter can be narrowor broad depending on design.

Two main theories stand behind thistype of architecture. First, it costs less thanthe other architectures in some implementa-tions. Second, some authorities ask, “Whyamplify noise prior to mixing?” The goal is tonot use up the mixer’s head room with

Chapter 4

The Receiver Front End:An Overview

37

Fig. 4.1Bandpass filter at the input of the receiver.

MIXER

OSCILLATOR

TO IFAMPLIFIER

BANDPASSFILTER

ANTENNA

unneeded energy. This theory has somemerit, as was evident in the Squires-SandersSS-1 receiver in the 1960s.

The main attributes of the bandpass filterare good forward performance (i.e., withinthe passband) and good reverse isolation. Theisolation is needed to prevent the local oscilla-tor signal from reaching the antenna, where itcan be radiated. The bandpass filter has threeimportant duties:

1. It must limit the bandwidth of the in-put signal to minimize intermodulationdistortion.

2. It must attenuate spurious responses,mainly the image frequency and the 1/2-IF frequency problems.

3. It must suppress local oscillator energyto prevent it from reaching the antenna.

A second version of the front-end ar-chitecture is shown in Figure 4.2. This ver-sion uses an RF amplifier. The gain of theRF amplifier is low, certainly less than 20dB. Gains in excess of 20 dB may compro-mise stability, and the intercept point maynot be achieved. The purpose of the RF am-plifier is to isolate the mixer as well as givethe signal a small boost prior to mixing.The boost overcomes the losses in themixer and the bandpass filter. The principal

benefit of the RF amplifier is that it im-proves the isolation of the mixer/LO circuitfrom the antenna circuit.

A third version is shown in Figure 4.3.Like the other two architectures, this has amixer and local oscillator circuit (or a con-verter, which contains both mixer and localoscillator). The difference between this archi-tecture and the previous ones is in the addi-tion of a second bandpass filter.

This second bandpass filter may havethe same frequency as the first bandpass fil-ter but that is not the only arrangement.The second bandpass filter often is tuned tothe image frequency. This frequency is theRF frequency plus or minus twice the IF, lo-cated on the other side of the local oscilla-tor from the RF signal (Figure 4.4). Thatway, the image frequency gets the sametreatment in the mixer as the RF, so itcomes through the system as a valid signal.By having a filter tuned to notch the imagefrequency, while passing the desired fre-quency, we can limit this problem. Ofcourse, the image filter must track thebandpass filter at the input if the receiverhas multiple frequencies.

The second bandpass filter also may at-tenuate the receiver’s other spurious re-sponse and direct IF pick-up. Further, itattenuates noise originating in the RF ampli-


Fig. 4.2 RF amplifier and bandpass filter at the input of the receiver.

MIXER

OSCILLATOR

TO IFAMPLIFIER

BANDPASSFILTER

ANTENNA

RF AMPLIFIER

fier, preventing the noise from reaching themixer. Finally, it suppresses the second har-monic energy arising in the RF amplifier,thereby improving the receiver’s second-order intercept point. This filter shouldhave no return responses at high frequen-cies, because the mixer has poor responsefor odd harmonics of the receive frequencyand they therefore may ride through thesystem.

The RF bandpass filter’s attributes aredetermined by a combination of the first IFfrequency and the injection side of the localoscillator signal. If low side injection is se-lected, some of the spurious signal prod-ucts may be on the low side of the RFsignal. On high side injection, just the op-

posite occurs: All the spurious signals willbe on the high side of the RF signal. Thetrade-off between insertion loss and selec-tivity in the filter usually is made in favor ofinsertion loss in bandpass filter number 1,but it can be sacrificed in bandpass filternumber 2.

MIXER AND LOCAL OSCILLATORPERFORMANCE

The performance of the first mixer is key tothe performance of the receiver. It is a nonlin-ear device. Furthermore, it usually sees thehighest-level radio frequency signals in thesystem (the LO, largely). It therefore needs tohave a very high intercept point. Single-deviceactive mixers are cheap, but they have thepoorest performance of all the mixers (seeChapter 7). The best performance, generallyspeaking, is in the passive, double-balancedmixers. These filters generally have the high-est intercept points and better noise balancethan most mixer designs. Table 4.1 shows themixer performance parameters and the thingsthey affect.

A trade-off is made in the selection ofthe type of mixer circuit used in a receiver.Passive mixers have better intermodulationdistortion performance than active mixers.However, they provide no conversion gainand, in fact, are lossy devices. Active mixers

The Receiver Front End: An Overview 39

Fig. 4.3 Dual RF bandpass filters.

MIXER

OSCILLATOR

TO IFAMPLIFIER

BANDPASSFILTER 1

ANTENNA

RF AMPLIFIER

BANDPASSFILTER 2

Fig. 4.4 Image frequency.

RF

LO

IMAGE

FREQUENCY

require less in the way of local oscillatorpower but have no better noise performancethan passive mixers. Furthermore, at hightemperatures the high third-order interceptpoint performance of the active mixer maybe deteriorated.

A diplexer network often is positionedbetween the mixer’s IF output and the IF am-plifier. The diplexer network absorbs somefrequencies, while passing others. The di-plexer network must be nonreflective up toseveral times the LO frequency. Otherwise,those frequencies would be reflected back tothe mixer, degrading its performance.

The single-sideband phase noise per-formance of the local oscillator is importantto the receiver’s adjacent channel selectivity.The wideband noise often afflicts the re-ceiver sensitivity. Further, the LO signalmust be as pure as possible to prevent spu-rious responses in the receiver. It is not pru-dent to ignore the LO signal, because it is alarge signal that causes switching in themixer (which generates its own harmonics).Rather, the LO signal should be as pure aspossible.

The local oscillator must be able tooperate normally despite changes in tem-perature and power supply voltage. It must

also perform normally under microphonicconditions of vibration and impact to thereceiver.

NOISE PERFORMANCE OF THE SYSTEM

All radio reception is a matter of manipulat-ing the signal-to-noise ratio (SNR) of thesystem. Because of this problem, the noisegenerated by the mixer, local oscillator,bandpass filters, and RF amplifier should beminimized.

The noise figure for a passive, lossy de-vice, such as the filter or some mixer stages,is given by

(4.1)

where

F is the noise factor of the device;L is the loss of the device (1/G);T is the temperature of the device in

degrees Kelvin (K).

(Some double-balanced mixers can haveslightly higher noise figures.)

Friis’s equation for noise governs thesystem:

(4.2)

where

F is the equivalent noise factor;F1, F2, and F3 are the noise factors of

stages 1, 2, and 3;Fn is the gain of the nth stage;G1, G2, and G3 are the gains of stages

1, 2, and 3;Gn−1 is the gain of the stage before the

nth stage.

The overall noise factor of the receiveris determined by the noise performance ofthe stages within the system.

F FF

G

F

G G

F

G G Gn

n

= +−

+−

+

+−

−

12 1

1

3 1

1 2

1

1 2 1

K

K

FL T

= +−

11

290

( )


Table 4.1 Mixer Attributes

Third-order Intermodulation intercept point distortion

Second-order intercept 1/2 IF responsepoint

Noise balance Sensitivity and AM noiserejection

LO to RF isolation LO energy radiated byantenna

RF to IF isolation Direct IF pick-up

Conversion loss Sensitivity

Note: Sometimes a third bandpass filter is located atthe interface between the mixer and the local oscil-lator. This LO filter is used to attenuate widebandnoise and its harmonics around the LO frequency,which could degrade the mixers second-order inter-cept point.

SPURIOUS RESPONSES

A spurious response is one that is unintended.On a superheterodyne receiver, these spurscan be created in the mixer stage, althoughthey have their origin elsewhere. Most re-ceiver spurs are a result of the heterodyningof the receiver, according to

(4.3)

where

FIF is the IF frequency;FRF is the RF frequency;FLO is the local oscillator frequency;m and n are either integers or zero.

By solving equation 4.3 for FRF, we gettwo possible RF frequencies at which spurscan occur:

(4.4)

(4.5)

The most common spurs are as follows:

1. Image frequency (previously defined,see Figure 4.5).

2. 1/2 IF (see Figure 4.5).

3. Direct IF pick-up.

4. n × LO frequency.

5. LO spurious frequencies.

6. Second mixer spurs (dual conversionreceivers only).

Full duplex radio receivers (i.e., thoseused in conjunction with a transmitter at thesame time) have two additional responsesthat must be considered: full duplex imageand half duplex image:

(4.6)

(4.7)

where

FT is the transmitter frequency;∆f is the difference between the trans-

mitter and receiver frequencies.

INTERCEPT POINTS

The intercept points are a measure of circuitlinearity. They allow us to calculate inter-modulation distortion levels from the inputsignal levels. The intercept point representsan input amplitude (Figure 4.6) at which thedesired fundamental frequency is equal inamplitude to the undesired signal.

SECOND-ORDER INTERCEPT POINT

The second-order intercept point (SOIP)comes from the operation of the second-order products of a signal and increases ata rate of 2 dB for a 1 dB increase in the fun-damental level. The 1/2 IF response of themixer can be predicted from the second-order intercept point. The 1/2 IF point is dueto the second harmonics of the RF signaland the LO signal, both of which are inter-nally generated (2FRF ± 2FLO). The 1/2 IF re-jection is

(4.8) 1 2/ IF rejection =

− −IP S C2

2 Half Duplex Image = +F

fT

∆2

Full Duplex Image = −F fT ∆

F

nF F

mRFLO IF=

+

F

nF F

mRFLO IF=

−

F mF nFIF RF LO= ±


Fig. 4.5 Spectrum analyzer presentation of 1/2 IF.

FRF FLOIMAGE

FRF + 2 FIF

FRF + FIF/2

FIFFIF

where

IP2 is the second-order intercept point;S is the receiver sensitivity (dBm);C is the capture ratio or the cochannel

rejection in dB.

For example, suppose a receiver has asecond-order intercept point of 45 dBmand a sensitivity of –120 dBm. If the co-channel rejection is 6 dB, the 1/2 IF rejec-tion is (45 dBm – 120 dBm – 6 dB)/2 = 159/2= 79.5 dB.

THIRD-ORDER INTERCEPT POINT

The third-order intercept point (TOIP) is thepoint at which the fundamental signal andits own third-order products are equal inamplitude. The TOIP increases 3 dB foreach 1 dB increase in the fundamental signal.The TOIP predominantly is responsible forthe intermodulation distortion performanceof the receiver, defined as the difference (indB) between the receiver’s sensitivity and thesignal level that is sufficient to produce aspecified level of interference. It can be cal-culated from

(4.9)

where

IM is the intermodulation rejection ratio (dB);

IP3 is the TOIP;S is the receiver sensitivity (dBm);C is the capture ratio or cochannel re-

jection in dB.

Equation 4.9 covers the situation forone carrier. Unfortunately, real receivers seemany carriers. The number of such productsis n(n − 1), where n is the number of carrierspresent for both (2F1 – F1) and (2F1 + F2),and (for triple beats) n(n – 1)(n – 2)/2 for(F1 + F2 – F3) situations.

nTH-ORDER INTERCEPT POINT

With knowledge of the input levels of signalsapplied to the receiver, we can calculate thenth-order intercept points:

(4.10)

where

IPN is the nth order intercept point;n is the order of the intercept point;PA is receiver input signal power level;PIMN

is the power level of the IMD signal.

IPnP P

nNA IMN=

−− 1

IMIP S C

=− −2 2

33


Fig. 4.6The nth-order intercept point.

Response ofFundamental Signal

nth-Order Response

nth-Order Intercept Point

INPUT LEVEL (Pin) IN dBm

OU

TP

UT

LE

VE

L (P

O)

IN d

Bm


RF AMPLIFIER

The RF amplifier can have a deleterious ef-fect on the performance of the mixer stage;hence, the entire receiver. A number ofmethods can be used to reduce the effect.The first method is to use a high-power de-vice operating well below its maximumrange. A trade-off with noise performance,however, must be taken into consideration.Second, you can reduce the signal level tothe device. This can be done with attenuators

in some cases. Care must be taken to balancethe needs of sensitivity in this respect. Third,you can reduce the stage gain. Again, noiseand SNR considerations apply. Fourth, youcan use negative feedback in the amplifier.Fifth, you can increase the selectivity of theRF amplifier. A narrower bandpass producesless noise than wider bandwidths. A sixthway is to use push-pull amplifiers, becausethey tend to cancel even-order products(odd-order products are not affected), whichtend to take up mixer head room.

The filtering used on the front end of a ra-dio receiver determines some importantcharacteristics of the radio. Two forms of fil-tering are used: single frequency and a fre-quency band. We consider both types. But,first, let us look at the L-C inductor resonanttank circuits.

INDUCTOR-CAPACITOR RESONANTTANK CIRCUITS

When you use an inductor (L) and a capaci-tor (C) together in the same circuit, the com-bination forms an L-C resonant circuit, alsosometimes called a tank circuit or resonanttank circuit. These circuits are used to selectone frequency, while rejecting all others (asto tune a radio receiver). There are two basicforms of L-C resonant tank circuit: series(Figure 5.1A) and parallel (Figure 5.1B).These circuits have much in common, andmuch that makes them fundamentally differ-ent from each other.

The condition of resonance occurswhen the capacitive reactance (XC) and in-ductive reactance (XL) are equal in magnitude(|+XL| = |−XC|). As a result, the resonant

tank circuit shows up as purely resistive at theresonant frequency (Figure 5.1C) and as acomplex impedance at other frequencies. TheL-C resonant tank circuit operates by an oscil-latory exchange of energy between the mag-netic field of the inductor and the electrostaticfield of the capacitor, with a current betweenthem carrying the charge.

Because both reactances are frequencydependent and inverse to each other, the res-onance occurs at only one frequency (fr). Wecan calculate the standard resonance fre-quency by setting the two reactances equalto each other and solving for f:

(5.1)

Series Resonant Circuits

The series resonant circuit (Figure 5.1A), likeother series circuits, is arranged so that theterminal current (I) from the source (V) flowsin both components equally. The vector dia-grams of Figure 5.2 show the situation underthree different conditions.

In Figure 5.2A, the inductive reactanceis larger than the capacitive reactance, so the

fLC

= 1

2π

Chapter 5

Front-End Filtering

45

excitation frequency is greater than fr. Notethat the voltage drop across the inductor isgreater than that across the capacitor, so thetotal circuit looks like it contains a small in-ductive reactance. In Figure 5.2B, the situa-tion is reversed—the excitation frequency isless than the resonant frequency—so thecircuit looks slightly capacitive to the out-side world. Finally, in Figure 5.2C the excita-tion frequency is at the resonant frequency,so XC = XL and the voltage drops across thetwo components are equal but of oppositephase.

In a circuit that contains a resistance,inductive reactance, and a capacitive reac-

tance, three vectors must be considered(Figure 5.3) plus a resultant vector. As inthe other circuit, the north direction repre-sents XL, the south direction represents XC,and the east direction represents R. Usingthe parallelogram method, we first con-struct a resultant vector for R and XC, whichis shown as vector A. Next, we constructthe same kind of vector (B) for R and XC.The resultant vector C is made using theparallelogram method on A and B. Vector Crepresents the impedance of the circuit:The magnitude is represented by thelength, and the phase angle by the anglebetween C and R.


Fig. 5.1 Resonant circuits: (A) series; (B) parallel; (C) vector relationships at resonance.

XL

V XC XLV

+X

-X

+RR

+XL

-XC

0

CL XX -=+

22 )( XCXLRZ -+=

22 )0(+= RZ

RRZ == 2

XC

XL

V XC XLV

A B

C

Front-End Filtering 47

+X

+R

-X

-XC

+XL

C

B

A'

B'A

+X

+R

-X

-XC

+XL

C

B

A'

B'

A

0

Fig. 5.2 Vector relationships: (A) inductive reactance larger than capacitive reactance; (B) capacitive reac-tance larger than inductive reactance; (C) excitation frequency at resonant frequency.

+X

+R

-X

-XC

+XL

0

A B

C

Figure 5.4A shows a series resonant L-Ctank circuit, and Figure 5.4B shows the cur-rent and impedance as a function of fre-

quency. The series resonant circuits have alow impedance at their resonant frequenciesand a high impedance at all other frequencies.As a result, the line current (I) from the sourceis maximum at the resonant frequency and thevoltage across the source is minimum.

Parallel Resonant Circuits

The parallel resonant tank circuit (Figure5.5A) is the inverse of the series resonant cir-cuit. The line current (I) from the source splitsand flows in the inductor and capacitor sepa-rately. The parallel resonant circuit has itshighest impedance at the resonant frequency,and a low impedance at all other frequencies(Figure 5.5B). Therefore, the line current fromthe source is minimum at the resonant fre-quency, and the voltage across the LC tankcircuit is maximum. This is important in radiotuning circuits, as you will see.

TUNED RF TRANSFORMERS

Many of the resonant circuits used in RF cir-cuits, and especially radio receivers, actuallyare transformers that couple a signal from


Fig. 5.3 Relationships to resultant vectors.

+X

R+R

-X

-XC

+XL

C

B

A'

B'A

Fig. 5.4 Series resonant L-C tank: (A) circuit; (B) impedance and current vs. frequency.

VAC

XC

XL

I

Z I

FC

FREQUENCY

MA

GN

ITU

DE

A

B

one stage to another. Figure 5.6 shows sev-eral popular forms of tuned, or coupled,RF/IF tank circuits. In Figure 5.6A, onewinding is tuned while the other is untuned.In the configurations shown, the untunedwinding is the secondary circuit of the trans-former. This type of circuit often is used intransistor and other solid-state circuits orwhen the transformer has to drive either acrystal or mechanical bandpass filter circuit.In the reverse configuration (L1 = output, L2 = input), the same circuit is used for theantenna coupling network or as the inter-stage transformer between RF amplifiers inTRF radios.

The circuit in Figure 5.6B is a parallelresonant L-C tank circuit equipped with alow-impedance tap on the inductor. Thistype of circuit often is used to drive a crystaldetector or other low-impedance load.Another circuit for driving a low-impedanceload is shown in Figure 5.6C. This circuitsplits the capacitance that resonates the coilinto two series capacitors. As a result, wehave a capacitive voltage divider. The circuitin Figure 5.6D uses a tapped inductor formatching low-impedance sources (e.g., an-tenna circuits) and a tapped capacitive volt-

age divider for low-impedance loads. Finally,the circuit in Figure 5.6E uses a tapped pri-mary and tapped secondary winding tomatch two low-impedance loads while re-taining the sharp bandpass characteristics ofthe tank circuit.

CONSTRUCTION OF RF/IFTRANSFORMERS

The tuned RF/IF transformers built for radioreceivers typically are wound on a commoncylindrical form and surrounded by a metalshield can that prevents interaction of thefields of coils in close proximity to eachother.

Figure 5.7A shows the schematic for atypical RF/IF transformer, while the sec-tioned view (Figure 5.7B) shows one formof construction. This method of buildingtransformers was common at the beginningof World War II and continued into theearly transistor era. The methods of con-struction shown in Figures 5.7C and 5.7Dwere popular prior to World War II. The ca-pacitors in Figure 5.7B were built into the


Fig. 5.5 Parallel resonant tank: (A) circuit; (B) impedance and current vs. frequency.

VAC XC XL

I

ZI

FC

FREQUENCY

MA

GN

ITU

DE

A

B

base of the transformer, while the tuningslugs were accessed from holes in the topand bottom of the assembly. In general, ex-pect to find the secondary winding at thebottom hole, and the primary winding atthe top hole.

The term universal wound refers to across-winding system that minimizes the in-terwinding capacitance of the inductor andtherefore raises the self-resonant frequencyof the inductor (a good thing).

BANDWIDTH OF RF/IF TRANSFORMERS

Figure 5.8A shows a parallel resonant RF/IFtransformer, while Figure 5.8B shows theusual construction in which the two coils (L1and L2) are wound at distance d apart on acommon cylindrical form.

The bandwidth of the RF/IF transformeris the frequency difference between the fre-quencies where the signal voltage across theoutput winding falls off −6 dB from the value


Fig. 5.6 Various L-C filter circuits.

OUTL2

IN CL1

B

IN OUTL2L1C

A

OUTC2

IN

C1

L

C

OUTC2IN

C1L1

D

L2

OUTL2

IN

C1L1

E

L3

L4

C2


Fig. 5.7 Typical RF/IF transformer: (A) circuit diagram; (B), (C), and (D) physical implementations.

SLUG-TUNEDCOIL FORM

ACCESS TOTOP SLUG

ACCESS TOBOTTOM SLUG

SHIELD

FIXED CAPACITORSINSIDE BASE

A

B

C D

at the resonant frequency (fr), as shown inFigure 5.8C. If F1 and F2 are −6 dB (calledthe −3 dB point, when signal power is mea-sured instead of voltage) frequencies, thenthe bandwidth (BW) is F2–F1. The shape ofthe frequency response curve in Figure 5.8Cis said to represent critical coupling.

An example of a subcritical or under-coupled RF/IF transformer is shown in Figure5.9. As shown in Figures 5.9A and 5.9B, thewindings are farther apart than in the criti-cally coupled case, so the bandwidth (Figure5.9C) is much narrower than in the criticallycoupled case. The subcritically coupledRF/IF transformer often is used in shortwave

or communications receivers to allow thenarrower bandwidth to discriminate againstadjacent channel stations.

The overcritically coupled RF/IF trans-former is shown in Figure 5.10. Note inFigures 5.10A and 5.10B that the windingsare closer together, so the bandwidth (Figure5.10C) is much broader. In some radioschematics and service manuals (not to men-tion early textbooks), this form of couplingsometimes was called high-fidelity couplingbecause it allowed more of the sidebands ofthe signal (which carry the audio modula-tion) to pass with less distortion of frequencyresponse.


Fig. 5.8Parallel resonant RF/IF trans-former: (A) circuit diagram;(B) physical implementation;(C) frequency response. FR

6 dB

BW

L1

L2

d

A

B

C

The bandwidth of the resonant tank cir-cuit, or the RF/IF transformer, can be summa-rized in a figure of merit called Q. The Q ofthe circuit is the ratio of the bandwidth to theresonant frequency:

(5.2)

An overcritically coupled circuit has alow Q, while a narrow bandwidth subcriti-cally coupled circuit has a high Q.

Resistance in the L-C tank circuit willcause it to broaden; that is, to lower its Q.The resistor sometimes is called a de-Qing

resistor. The “loaded Q” (i.e., Q when a resis-tance is present, as in Figure 5.11A) always isless than the unloaded Q. In some radios, aswitched resistor (Figure 5.11B) is used to al-low the user to broaden or narrow the band-width. This switch might be labeled fidelityor tone or something similar.

PROBLEMS WITH IF AND RF TRANSFORMERS

The IF and RF transformer holds a high po-tential for intermittent problems in radio re-ceivers. Two basic forms of problem are

Q

BW

Fr

=


Fig. 5.9Subcritical or undercoupled RF/IFtransformer: (A) circuit diagram;(B) physical implementation;(C) frequency response.

L1

L2

dLarger

FR

A

B

C

found in the IF transformer: intermittent op-eration and intermittent noise. The best curefor a bad IF or RF transformer is replacement,but because old IF and RF transformers are

not always available today, we must placemore emphasis on repair of the transformer.

Figure 5.12A shows the basic circuit fora single-tuned IF transformer (others might


Fig. 5.10Overcritically coupled RF/IF trans-former: (A) circuit diagram;(B) physical implementation;(C) frequency response.

L1

L2

d

FR

A

B

C

Fig. 5.11 Two ways of broadening the response of an L-C IF transformer.

RLR S1

A

B

have a tuned secondary winding). If one ofthe very fine wires making up the coil breaks(Figure 5.12B), then operation is interrupted.The transformer usually can be diagnosed bylightly tapping on the shield can of the trans-former or by using a signal tracer or signalgenerator to find where the signal is inter-rupted. In some cases, the collector or drainvoltage of the IF amplifier is interruptedwhen the transformer opens, and this can bespotted using a DC voltmeter.

Repairing the IF transformer is a deli-cate operation. Examine the shield can to de-termine how it is assembled. If it is securedwith a screw or nut, then merely remove thescrew (or nut). If the IF transformer is sealedby metal tabs, then very carefully pry thetabs open (do not break them or bend themtoo far). Slide the base and coil assembly outof the shield. If there is enough slack on thebroken wire, then simply solder the wireback onto the terminal. The heat of the sol-dering iron tip will burn away the enamel in-sulation. If the wire has too little slack, thenadd a little length to the terminal by solder-ing a short piece of solid wire to the termi-nal, and then soldering the IF transformerwire to the added wire. In some cases, it ispossible to remove a portion of one turn ofthe transformer winding to gain extra length,but this procedure will change the tuning.

The noisy IF transformer needs replace-ment more than open types, but you still facethe lack of original or replacement compo-nents. The capacitor can be replaced onmany IF transformers, and that can lead to arepair. If the capacitor is a discrete compo-

nent soldered to the base, then it is simple toreplace it with another similar (if not identi-cal) capacitor. But if the IF transformer usesan embedded mica capacitor (Figure 5.13),then the job becomes more complex.

If the capacitor element is visible (as inFigure 5.13), clip one end of the capacitorwhere it is attached to the terminal. Thenbridge a disk ceramic or mica capacitor be-tween the two terminals. The value of the ca-pacitor must be found experimentally—it isthe value that will resonate the coil to the IFfrequency.

If the capacitor is embedded and notvisible, then it may become necessary to in-stall a new terminal and solder the added ca-pacitor to that terminal (with appropriateexternal circuit modifications).

CHOOSING COMPONENT VALUESFOR L-C RESONANT TANK CIRCUITS

Resonant L-C tank circuits are used to tune ra-dio receivers: These circuits select the stationto be received, while rejecting others. A su-perheterodyne radio receiver (the most com-mon type) is shown in simplified form inFigure 5.14. According to the superheterodyneprincipal, the radio frequency being received(FRF) is converted to another frequency, calledthe intermediate frequency (FIF), by beingmixed with a local oscillator signal (FLO) in anonlinear mixer stage. The output of the un-tamed mixer would be a collection of fre-quencies defined by

FIF = mFRF ± nFLO (5.3)


Fig. 5.12 Single-tuned IF transformer: (A) circuit diagram; (B) broken wire.

WIRE

TERMINAL

AB

where m and n are either integers or zero.For the simplified case discussed here onlythe first set of products (m = n = 1) is consid-ered, so the output spectrum will consist ofFRF, FLO, FRF − FLO (difference frequency), andFRF + FLO (sum frequency). In older radios, forpractical reasons, the difference frequencywas selected for FIF; today, either sum or dif-ference frequencies can be selected, depend-ing on the design of the radio.

Several L-C tank circuits are present inthis notional superhet radio. The antenna

tank circuit (C1/L1) is found at the input ofthe RF amplifier stage, or if no RF amplifieris used, it is at the input to the mixer stage.A second tank circuit (L2/C2), tuning thesame range as L1/C1 is found at the outputof the RF amplifier or the input of themixer. Still another LC tank circuit (L3/C3)is used to tune the local oscillator; this tankcircuit sets the frequency that the radio willreceive.

Additional tank circuits (only two shown)may be found in the IF amplifier section of


Fig. 5.13A bridging capacitor repairs the IF transformer.

TERMINAL

COIL FORM

EMBEDDEDCAPACITOR

BREAKADDED DISKCERAMIC OR

MICA CAPACITOR

Fig. 5.14 Superheterodyne receiver.

RFAMPLIFIER

MIXER

LOCALOSCILLATOR

IFAMPLIFIER

DETECTORTO AUDIO

AMPLIFIERS

ANTENNA

FRF FRF FIF

FLOC1

C2L2

L1

C3L3

the radio. These tank circuits will be fixedtuned to the IF frequency, which in commonAM broadcast band (BCB) radio receiverstypically is 450 kHz, 455 kHz, 460 kHz, or470 kHz, depending on the designer’s choice(and sometimes country of origin). Other IFfrequencies also are seen, but these are themost common. FM broadcast receivers typi-cally use a 10.7 MHz IF, while shortwave re-ceivers might use a 1.65 MHz, 8.83 MHz, 9MHz, or an IF frequency above 30 MHz.

THE TRACKING PROBLEM

On a radio that tunes the front end with asingle knob, which includes almost all re-ceivers today, the three capacitors (C1–C3 inFigure 5.14) typically are ganged; that is, thecapacitors are mounted on a single rotorshaft. These three tank circuits must trackeach other; that is, when the RF amplifier istuned to a certain radio signal frequency, theLO must produce a signal different from theRF frequency by the amount of the IF fre-quency. Perfect tracking probably is impossi-ble, but the fact that your single-knob-tunedradio works proves that the tracking is nottoo terrible.

The issue of tracking L-C tank circuitsfor the AM BCB receiver has not been a majorproblem for many years: The band limits arefixed over most of the world, and componentmanufacturers offer standard adjustable in-ductors and variable capacitors to tune the RFand LO frequencies. Indeed, some even offerthree sets of coils: antenna, mixer input/RFamp output, and LO. The reason why the an-tenna and mixer/RF coils are not the same,despite tuning the same frequency range, isthat these locations see different distributedor “stray” capacitances. In the United States, itis standard practice to use a 10–365 pF capac-itor and a 220 µH inductor for the 540–1600kHz AM BCB. In some other countries,slightly different combinations are sometimesused: 320 pF, 380 pF, 440 pF, 500 pF, and oth-ers are seen in catalogs.

Recently, however, two events coin-cided that caused me to examine the

method of selecting capacitance and induc-tance values. First, I embarked on a designproject to produce an AM DXers receiverthat had outstanding performance character-istics. Second, the AM broadcast band wasrecently extended so that the upper limit isnow 1700 kHz, rather than 1600 kHz. Thenew 540–1700 kHz band is not accommo-dated by the now obsolete “standard” valuesof inductance and capacitance. So I calcu-lated new candidate values. Shortly, we willsee the result of this effort.

THE RF AMPLIFIER/ANTENNATUNER PROBLEM

In a typical RF tank circuit, the inductance iskept fixed (except for a small adjustmentrange that is used for overcoming tolerancedeviations) and the capacitance is variedacross the range. Figure 5.15 shows a typicaltank circuit main tuning capacitor (C1), trim-mer capacitor (C2), and a fixed capacitor(C3) that is not always needed. The stray ca-pacitances (Cs) include the interwiring ca-pacitance, the wiring to chassis capacitance,and the amplifier or oscillator device inputcapacitance. The frequency changes as thesquare roots of the capacitances change. IfF1 is the minimum frequency in the rangeand F2 is the maximum frequency, then therelationship is

(5.4)

F

F

C

C

2

1= max

min


Fig. 5.15 Tuning scheme.

C1MAIN

TUNINGC2

TRIMMER

CPPADDER

CSSTRAY

L1

or, in a rearranged form that some find morecongenial,

(5.5)

In the case of the new AM receiver, Iwanted an overlap of about 15 kHz at thebottom end of the band and 10 kHz at theupper end, so I needed a resonant tank cir-cuit that would tune from 525 kHz to 1710kHz. In addition, because variable capacitorsare widely available in certain values basedon the old standards, I wanted to use a “stan-dard” AM BCB variable capacitor. A 10–380pF unit from a vendor was selected.

The minimum required capacitance,Cmin, can be calculated from

(5.6)

where

F1 is the minimum frequency tuned;F2 is the maximum frequency tuned;Cmin is the minimum required capaci-

tance at F2;∆C is the difference between Cmax

and Cmin.

Example

Find the minimum capacitance needed totune 1710 kHz when a 10–380 pF capacitor(∆C = 380 − 10 pF = 370 pF) is used and theminimum frequency is 525 kHz.

Solution

The maximum capacitance must be Cmin + ∆C, or 38.51 + 370 pF = 408.51 pF.

Because the tuning capacitor (C1 in Figure5.15) does not have exactly this range, exter-nal capacitors must be used, and because therequired value is higher than the normalvalue, additional capacitors are added to thecircuit in parallel to C1. Indeed, becausesomewhat unpredictable “stray” capacitancesalso exist in the circuit, the tuning capacitorvalues should be a little less than the re-quired values to accommodated strays plustolerances in the actual—versus published—values of the capacitors. In Figure 5.15, themain tuning capacitor is C1 (10–380 pF), C2is a small value trimmer capacitor used tocompensate for discrepancies, C3 is an op-tional capacitor that may be needed to in-crease the total capacitance, and Cs is thestray capacitance in the circuit.

The value of the stray capacitance can bequite high, especially if other capacitors in thecircuit are not directly used to select the fre-quency (e.g., in Colpitts and Clapp oscillators,the feedback capacitors affect the L-C tank cir-cuit). In the circuit I was using, however, the L-C tank circuit was not affected by other capacitors. Only the wiring strays and the in-put capacitance of the RF amplifier or mixerstage needed to be accounted for. From expe-rience, I apportioned 7 pF to Cs as a trial value.

The minimum capacitance calculatedwas 38.51, there was a nominal 7 pF of straycapacitance, and the minimum available ca-pacitance from C1 was 10 pF. Therefore, thecombined values of C2 and C3 must be 38.51pF − 10 pF − 7 pF, or 21.5 pF. Because ofconsiderable reasonable doubt about the ac-tual value of Cs and because of tolerances inthe manufacture of the main tuning variablecapacitor (C1), a wide range of capacitancefor C2 + C3 is preferred. It is noted from sev-eral catalogs that 21.5 pF is near the center ofthe range of 45 pF and 50 pF trimmer capac-itors. For example, one model lists its rangeas 6.8–50 pF and its center point is onlyslightly removed from the actual desired ca-pacitance. Therefore, a 6.8–50 pF trimmerwas selected and C3 was not used.

Selecting the inductance value for L1(Figure 5.15) is a matter of picking the fre-quency and associated required capacitance

22

1

1 710

525370

F

FC = C + C

C = C +

C C

min min

min min

min min

,

∆

2kHz

kHzpF

10.609 = + 370 pF = 38.61 pF

22

1

F

FC = C + C

min min ∆

F

F

C

C

2

1

2

= max

min


at one end of the range and calculating fromthe standard resonance equation solved for L:

The RF amplifier input L-C tank circuitand the RF amplifier output L-C tank circuit areslightly different cases because the stray ca-pacitances are somewhat different. In theexample, I assumed a JFET transistor RF ampli-fier, which has an input capacitance of only afew picofarads. The output capacitance is nota critical issue in this specific case because I in-tended to use a 1 mH RF choke to preventJFET oscillation. In the final receiver, the RFamplifier may be deleted altogether, and the L-C tank circuit just described will drive a mixerinput through a link coupling circuit.

THE LOCAL OSCILLATOR PROBLEM

The local oscillator circuit must track the RFamplifier and tune a frequency range that is

different from the RF range by the amount ofthe IF frequency (455 kHz). In keeping withcommon practice, I elected to place the LOfrequency 455 kHz above the RF frequency.Therefore, the LO must tune the range980–2165 kHz.

Three methods are used to make the lo-cal oscillator track with the RF amplifier fre-quency when single-shaft tuning is desired:the trimmer capacitor method, the padder ca-pacitor method, and the different-value cut-plate capacitor method.

Trimmer Capacitor Method

The trimmer capacitor method, shown inFigure 5.15, is the same as the RF L-C tank cir-cuit. Using exactly the same method as be-fore, but with a frequency ratio of (2165/980)to yield a capacitance ratio of (2165/980)2 =4.88:1, solves this problem. The results werea minimum capacitance of 95.36 pF and amaximum capacitance of 465.36 pF. An in-ductance of 56.7 µH is needed to resonatethese capacitances to the LO range.

A problem is associated with using thesame identical capacitor for both RF and LO:The difference between seems to be justenough difference that tracking between them

Lf C

L =

= H

H

H

µ

µ

π

π

µ

=

×

≈

−

106

2 24

10

(4)( )(525, 000 ) (4.085 10 )

224.97 225

low max

6

2 2 10


Fig. 5.16Frequency vs. shaft angle.

0 50 100 150 200

FRF

FLO(IDEAL)

FLO(ACTUAL)

SHAFT ANGLE (DEGREES)

FR

EQ

UE

NC

Y

always is a bit off. Figure 5.16 shows the idealLO frequency and the calculated LO fre-quency. The difference between these twocurves is the degree of nontracking. Thecurves overlap at the ends but are awful in themiddle. This problem has two cures. First, usea padder capacitor in series with the main tun-ing capacitor (Figure 5.17). Second, use a dif-ferent-value cut-plate capacitor.

Padder Capacitor Method

Figure 5.17 shows the use of a padder ca-pacitor (Cp) to change the range of the LOsection of the variable capacitor. Thismethod is used when both sections of thevariable capacitor are identical. Once thereduced capacitance values of the C1/Cp

combination are determined, the procedureis identical to the one just used. But, first,we have to calculate the value of the pad-der capacitor and the resultant range of theC1/Cp combination. The padder value isfound from

(5.7)

And solving for Cp, for the values of the se-lected main tuning capacitor and LO,

(5.8)

Solving for Cp by the lowest commondenominator method (crude, but it works)yields a padder capacitance of 44.52 pF.The series combination of 44.52 pF and a10–380 pF variable yields a range of 8.2 pFto 39.85 pF. An inductance of 661.85 µH isneeded for this capacitance to resonateover 980–2165 kHz.

Cut-Plate Capacitor Method

A practical solution to the tracking problemthat comes close to the ideal is to use a cut-plate capacitor. These variable capacitorshave at least two sections, one each for RFand LO tuning. The capacitor plates are spe-cially cut to a shape that permits a constantchange of frequency for every degree ofshaft rotation. With these capacitors, whenwell done, it is possible to produce three-point tracking or better.

BANDSPREADING

Bandspreading on a communications re-ceiver is the spreading out of a band on aseparate dial. Amateur bands, for example,often are bandspread on high-frequencycommunications receivers. Figure 5.18 showstwo methods for bandspreading. In Figure5.18A the bandspread capacitor (C2) is con-nected in parallel with the main tuning ca-pacitor (C1). In Figure 5.18B, however, it is

( ).

380

3804 88

10

10

pF

pF

pF

pF

C

C

C

Cp

p

p

p+= ( ) ×

+

C C

C C

F

F

C C

C Cp

p

p

p

1

1

2

1

1

1

2max

max +=

×+

min

min


Fig. 5.17 Padder capacitor tuning scheme.

C1MAIN

TUNINGC2

TRIMMER

CPPADDER

CSSTRAY

L1

Fig. 5.18 Two methods of bandspreading a receiver.

L1 C1 C2

L1

C1

C2

A B

connected to a tap on the inductor. In ei-ther case, the tuning capacitance is muchgreater than the bandspread capacitance(C1 >> C2).

SINGLE-FREQUENCY FILTERING VS.BANDPASS FILTERING

Single-frequency filtering is one of twomethods for filtering the input of a radio re-

ceiver (the other one is bandpass filtering).Although both are bandpass filtering meth-ods, the single-frequency filtering methodnarrows the bandpass to a small amountaround the frequency of operation. Thebandpass approach uses a band of at least250 kHz and, for VHF/UHF receivers, con-siderably more.

Figure 5.6 shows several examples ofsingle-frequency filtering. Figure 5.19 showsthree more examples. In this case, the filters


Fig. 5.19 Common impedance coupling methods.

INPUT

C1

C3

L1 L2

L3

C2

OUTPUT

C1 C2L1A L2A

L1B L2B

CM

C1 C2L1A L2A

L1B L2B

LM

A

B

C

are double tuned, with the two resonant cir-cuits sharing a common reactance to pro-vide coupling. In Figure 5.19A, the commonreactance is a capacitor (CM); while inFigures 5.19B and 5.19C, it is an inductor(LM in Figure 5.19B and L3 in Figure 5.19C).The difference between these two circuits istwofold. The circuit in Figure 5.19B useslink coupling, whereas the filter in Figure5.19C uses direct coupling. Second, note thecapacitor. In Figure 5.19B, it is directly inparallel with the inductance; and in Figure5.19C, it is in parallel with the inductanceplus the mutual inductance between thetwo sections.

The bandpass filter is wider bandwidththan the single-frequency type. Figure 5.20Ashows an example of a bandpass filter. This

filter can be designed with three coupledresonant circuits, each of which is resonantat a slightly different frequency (Figure5.20B), yielding a Chebyshev response.Figure 5.21 shows a different approach tobandpass filter design. This design uses twoparallel resonant and one series resonantcircuits to form the bandpass required. Adisadvantage of this type of circuit is thatthe component values tend to be large for50 Ω systems. A superior approach is shownin Figure 5.22. This approach connects theinput and output terminals of the networkto a tap on the inductor. That way, the in-ductors and capacitors used for L1/C1 andL3/C3 can be designed for the 200–1000 Ωrange, rather than 50 Ω. This makes thevalues more reasonable.


Fig. 5.20Bandpass filter: (A) withmultiple tuned circuits;(B) with overlappingbandwidths.

C1 L1 C2 L2 C3 L3

C4 C5 C6 C7

A

B

Fig. 5.21 Superior bandpass filter.

INPUT OUTPUTL1C1

L2 C2

C3 L3

VHF/UHF CIRCUITS

VHF/UHF circuits pose special problems forthe RF designer. The distributed capacitanceand inductance of the circuit become a muchlarger proportion of the total capacitance andinductance used. As a result, we can use or-dinary tuned circuits to about 500 MHz, orperhaps a little higher; but at higher frequen-cies, we have to use helical resonators orother approaches. Figure 5.23 shows a band-pass filter usable to about 450 or 500 MHz,although above about 300 MHz the coil tendsto be one or two turns, at most.

Figure 5.24 shows a coupled helical res-onator circuit. The helical coil is extremely

high-Q and roughly equivalent to a quarterwavelength transmission line. Figure 5.25shows a close-up of the helical resonator.The unloaded Q is

(5.9)

where

Qunloaded is the unloaded Q of the circuit;D is the internal shielded container

dimension;f is the frequency;k is 1.0 for cylindrical cases and 1.2 for

rectangular cases.

Assuming that d/D = 0.55 and A/D = 1.5,the pitch, p, of the filters is given by

(5.10)pD f=

2

90 6.

QDk

funloaded = 50


Fig. 5.22 Bandpass filter with impedance matching.

INPUT OUTPUTL1 C1

L2 C2

C3L3

Fig. 5.24 Helical filter.Fig. 5.25 Physical implementation of a helical filter.

d

p

D

A

D/4

D/4

Fig. 5.23 Bandpass filter usable to about 450 or500 MHz.

INPUT

L1 C1

L2

C2

OUTPUT

C3 L3

where

p is the pitch of the coil;D is the internal shielded container di-

mension in millimeters (mm);f is the frequency.

And the characteristic impedance is

(5.11)

If you wish to maintain the high-Q na-ture of these circuits, then coat the insides ofthe container and the coil itself with silver.

These circuits can be coupled, as inFigure 5.24, if a slot is cut into the shield be-tween the sections, so that the helixes can seeeach other. Alternate methods of coupling areshown in Figure 5.26. The left side of the il-lustration uses link coupling to the helical in-ductor. A link made of one to three turns ofwire (depending on frequency) is positionedclose to the helical coil. On the left side of the

picture, the coil is tapped, as in Figure 5.24,but the load resistor (R1) is connected to thebottom of the coil. A feedthrough capacitor(C3) is provided to bypass the currents flow-ing in the helical inductor to ground. In bothcases, capacitors (C1 and C2) are used to tunethe helical inductors.

Four helixes are connected in cascadein Figure 5.27. This circuit uses conventionalcoupling between the first two and last twohelixes and a series inductor between themiddle two helixes. In all cases, the helixesare resonated by capacitors. As in previouscases, these capacitors can be adjusted toslightly different frequencies to get a widerbandwidth.

UHF/MICROWAVE FREQUENCIES

At the upper end of the UHF spectrum andlower end of the microwave spectrum, re-ceivers use a somewhat different approach todesign of the front-end filtering. The designcan use helical resonators in the same man-ner as for lower frequencies, but at those fre-quencies, other methods can be used. Onesuch method is the trough line filters shownin Figure 5.28. These circuits can be designedto a single frequency by making the troughline quarter wavelength. In practical circuits,however, the circuit is cut a little shorter thannormal and tuned by a capacitive disk at oneend or on a side (Figure 5.28). The capaci-tance of the disk is given by

(5.12) C

Z A= λ

ω πo o tan ( )2

Z

fDO = 386


Fig. 5.26 Different forms of coupling a helical filter.

C1C2

L1

L2

L3

C3 R1

INPUT

OUTPUT

Fig. 5.27Multiple helical filters.

where

C is the capacitance;λ is the wavelength;A is the length of the trough line;ωo is the quantity 2πf, where f is the

frequency;

Zo is 138log10 for a coaxial line and

276log10 for a parallel line.

The differences among the circuits ofFigure 5.28 lay in the position of the capaci-tive disk and the means of coupling into andout of the trough line chamber. In Figure5.28A, the capacitive disk is on the end of thetrough line and the input/output coupling isaccomplished by taps on the trough line. InFigure 5.28B, the disk again is on the end,while the coupling in and out of the troughline is done by links (one to three turn in-ductors). In Figure 5.28C, one of the links isreplaced by a capacitive disk.

At frequencies above 300 MHz, it ispossible to use microstrip technology. Figure5.29 shows a quarter wavelength microstripwith a tuning capacitor at one end to resonate it. Note that the strip is bonded onthe ends of the printed wiring board to the ground plane underneath. The printedwiring board (PWB) itself should be a low-loss type such as PTFE, Teflon , Fluon , orglass epoxy fiberglass.

Calculating the length of the elementdepends on the dielectric constant of theboard (2.1–6). The exact value of the dielec-tric constant is given by

(5.13)

where

e is the dielectric constant;C is the capacitance of a known area of

double-sided material used tomake the PWB, in picofarads (pF);

e

Ct

A=

113

2D

d

D

d


Fig. 5.28 Strip line filters.

Fig. 5.29 Physical implementation of strip linefilters.

λ /4

A B C


t is the thickness of the material in mil-limeters (mm);

A is the area of the material in squaremillimeters (mm2).

The width of the material is given by

(5.14)

where

W is the width in millimeters (mm);t is the thickness in millimeters (mm);Z is the desired impedance.

The element length is found from:

(5.15)

where

L is the element length;LO is the length of half wavelength in

free space.

SURFACE ACOUSTIC WAVE FILTERS

At VHF through the low microwave spec-trum, it is possible to use surface acousticwave filters, like that shown in Figure 5.30.The filter is built on a piezoelectric substrate,with absorbers on either end. The input andoutput transducers consist of strip lines, andthe surface acoustic wave propagates be-tween them. A signal source is connected toone transducer, and a load resistor is con-nected to the other end.

LL

e=

× +O

0 475 0 67. ( . )

log (log . )

( . ) .

10 10 0 874

0 005 1 14

W t

Z e

= +

− × × +

Fig. 5.30 Surface acoustic wave filter.

SURFACEACOUSTIC

WAVE

INPUTTRANSDUCER

ABSORBEROUTPUT

TRANSDUCERABSORBER

SIGNALSOURCE

RL

The performance of radio receivers can beimproved by the use of either a preselectoror a preamplifier between the antenna andthe receiver. In this chapter, we look at smallsignal radio frequency (RF) amplifiers, prese-lectors, and preamplifiers. These devices canbe used to preamplify radio signals from an-tennas prior to input to a receiver, improvethe signal-to-noise ratio, and overcome sys-tems losses.

Most low-priced receivers (and somehigh-priced ones as well) suffer from perfor-mance problems that are a direct result ofthe trade-offs manufacturers made to pro-duce a low-cost model. In addition, older re-ceivers often suffer the same problems, asdo many homebrew radio receiver designs.Chief among these are difficulties in sensitiv-ity, selectivity, and image response.

Sensitivity is a measure of the receiver’sability to pick up weak signals. Part of thecause of poor sensitivity is low gain in thefront end of the radio receiver, although theIF amplifier contributes most of the gain.

Selectivity is a measure of the ability ofthe receiver to (1) separate two closelyspaced signals and (2) reject unwanted sig-

nals that are not on or very near the desiredfrequency being tuned. The selectivity pro-vided by a preselector is minimal for veryclosely spaced signals (that is the job of theIF selectivity in a receiver), but it is used forreducing the effects (e.g., input overloading)of large local signals—so it fits the secondhalf of the definition.

Image response affects only superhetero-dyne receivers, which most are. It is an inap-propriate response to a signal that is at afrequency of twice the receiver IF frequencyfrom the frequency to which the receiver istuned. A superheterodyne receiver convertsthe signal frequency (RF) to an intermediatefrequency (IF) by mixing it with a local oscilla-tor (LO) signal generated inside the receiver.The IF can be either the sum or difference be-tween the LO and RF (i.e., LO + RF or LO − RF);but in most older receivers and nearly all low-cost receivers, it is the difference (LO − RF).The problem is that two RF frequencies alwaysmeet that “difference” criteria: LO − IF and LO+ IF. Therefore, both LO + IF and LO − IF areequal to the IF frequency. If one is the desiredfrequency, then the other is the image fre-quency. If the image frequency gets through

Chapter 6

RF Amplifiers and Preamplifiers

67

the radio’s front-end tuning to the mixer, it willappear in the output as a valid signal.

A cure for all these problems is a littlecircuit called the active preselector. A prese-lector can be either active or passive. In bothcases, however, the preselector includes aresonant circuit tuned to the frequency towhich the receiver is tuned. The preselectoris connected between the antenna and thereceiver antenna input connector (Figure6.1), whereas the internal RF amplifier is con-nected in the same manner but is internal.Therefore, it adds a little more selectivity tothe front end of the radio to help discrimi-nate against unwanted signals.

The difference between the active andpassive designs is that the active design con-tains an RF amplifier stage, while the passivedesign does not. Therefore, the active prese-lector also deals with the sensitivity problemof the receiver. See Chapter 5 for informationon passive L-C forms of circuit.

The difference between a preamplifierand the amplifying variety of preselector isthat the preselector is tuned to a specific fre-quency or narrow band of frequencies. Thewideband preamplifier amplifies all signalscoming into the front end, with no discrimina-tion, and therein lies an occasional problem.

Another problem with any amplifierahead of the receiver is that the preamplifiermight deteriorate performance, rather thanmake it better. The amplifier may use up partof the receiver’s dynamic range or it may

cause moderate signals that it can handlenow to drive the receiver into compression.Along with compression come intermodul-ation distortion, possibly harmonic distortion,desensitization, and other problems—toomany problems to beat. So make sure you arenot going to drive the receiver into poor per-formance while trying to improve it.

Always use a preamplifier or preselec-tor that has a noise figure better than the re-ceiver being served. Friis’s equation for noisedemonstrates that the noise figure of the sys-tem is dominated completely by the noisefigure of the first amplifier. So, make surethat the amplifier is a low-noise amplifier(LNA) and has a noise figure a few dB lowerthan the receiver’s noise figure. This equa-tion is

(6.1)

where

FN is the overall noise factor of n stagesin cascade;

F1 is the noise factor of stage 1; F2 is the noise factor of stage 2; Fn is the noise factor of the nth stage; G1 is the gain of stage 1; G2 is the gain of stage 2; Gn-1 is the gain of stage (n − 1).

F FF

G

F

G GF

G G G

N

n

n

= + − +−

+

+ −

−

12 3

1

1

1

1

1 21

1 2

. . .

. . .


Fig. 6.1Use of a preselector orpreamplifier with a receiver.

PRESELECTOROR

PREAMPLIFIER

ANTENNA

RECEIVER

NOISE AND PRESELECTORS OR PREAMPLIFIERS

The weakest radio signal that you can de-tect on a receiver is determined mainly bythe noise level in the receiver. Some noisearrives from outside sources, while othernoise is generated inside the receiver. Inthe VHF/UHF/microwave range, the inter-nal noise is predominant, so it is commonto use a low-noise preamplifier ahead ofthe receiver. The preamplifier reduces thenoise figure for the entire receiver.

The low-noise amplifier should bemounted on the antenna if it is widebandand at the receiver if it is tunable. (Note: Theterm preselector applies only to tuned ver-sions, while preamplifier could denote ei-ther tuned or wideband models.) Of course,if your receiver is used only for one fre-quency, then it also may be mounted at theantenna. The reason for mounting the an-tenna right at the antenna is to build up thesignal and improve the signal-to-noise ratio(SNR) prior to feeding the signal into thetransmission line, where losses cause it toweaken somewhat.

AMPLIFIER CONFIGURATIONS

Although the transistor was not actually in-vented until the late 1940s, it was predicted astheoretically possible by physicists for about10 years before that time. Natural semicon-ductor diodes were known as early as pre-World War I days, when a lead-based mineralcrystal, called galena, was use as the detectorin radio crystal sets. The semiconductor PNjunction diode is a “blood relative” and pre-cursor to the transistor, but it took the metal-lurgy that grew out of World War II research tostimulate the development of reliable, manu-factured semiconductor diodes. Previously, itwas not possible to get germanium or siliconpure enough to make these devices; intensivewartime research solved that problem. Oncethat hurdle was out of the way, scientists atBell Laboratories were able to make theworld’s first working point contact transistor.

Today, transistors are made in the same vari-ety of ways as diodes.

SIMPLE BIPOLAR TRANSISTORS

The word transistor is derived from transferresistor. Transistors are amplifying devicesmade from semiconductor materials. Thesimple bipolar transistor often is likened to apair of diodes connected back to back. Thestory is much more complex, however, asmay be seen if you attempt to obtain any-thing resembling “transistor action” by soconnecting a pair of signal diodes.

The basic transistor consists of threesections of semiconductor material, as shownin Figure 6.2A. The figure shows two types,NPN and PNP (Figure 6.2B shows their re-spective circuit symbols). Both consist of acentral region of one type of semiconductormaterial sandwiched between two other re-gions of the opposite type of material. AnNPN device, for example, has two N-type re-gions sandwiching a region of P-type mater-ial. In the PNP transistor, just the oppositesituation obtains: an N-type central region issurrounded by two P-type sections.

The central region, called the base, con-trols the activities of the entire transistor. Thetwo end sections are called the emitter andthe collector. You may be tempted to think ofthese regions as being interchangeable be-cause of the apparent symmetry in the pic-ture. This was true in a very limited sense inthe early days of transistors, but today’s tran-sistors use a physical geometry that is likethis illustration electrically but quite differentphysically. The picture is only a graphicalmodel of the actual situation.

The base region typically is much thin-ner than either the emitter or collector regions. In addition, the base region semicon-ductor material is considerably less heavilydoped than either the emitter or base region.As a result, charge carriers (electrons andholes, depending on PNP or NPN) crossingthe base region have considerably less proba-bility of recombining with the alternate typeof charge carrier.

RF Amplifiers and Preamplifiers 69

The block diagram model of the bipolartransistor is expanded in Figure 6.3. Here wesee an NPN transistor, but it also serves as amodel for the PNP type, if the VEE and VCC

power supply polarities are reversed.There are two PN junctions in the bipo-

lar transistor: one formed by the base andemitter (B-E) junction and the other formedby the collector and base (C-B) junction. Innormal operation, the B-E junction is forwardbiased and the C-B junction is reverse biased.At each junction is a barrier potential.

Electrons in the emitter region are re-pelled by the negative terminal of the VEE po-tential and attracted into the relatively morepositive base region, forming emitter currentIE. Because of the thinness of the base regionand its light doping profile, few electrons re-combine with holes. Most of them overcomethe C-E junction barrier potential and continueinto the collector region. Once in the collectorregion, the negatively charged electrons comeunder the influence of the positive terminal ofthe VCC potential and are “collected” as current


Fig. 6.2 Basic transistors: (A) NPN and PNP; (B) memory aid.

P NN

C E

B

N PP

C E

B

C = COLLECTOR B = BASE E = EMITTER

C = COLLECTOR B = BASE E = EMITTER

MEMORY AIDON PNP ARROW

POINTS IN

PNPNPN

Fig. 6.3 Current flow in an NPN transistor.

+

EMITTER BASE COLLECTOR

+

IE IC

IB

N NP

VEE VCC

ICBO

IC = a IE

BASE-EMITTERJUNCTION

COLLECTOR-BASEJUNCTION

A B

IC. As current is drawn from the emitter re-gion, additional electrons are attracted into thedevice from the VEE power supply.

The case for PNP is similar to that forNPN, except that, in PNP devices, the chargecarriers are holes and the battery potentialsare reversed. Charge carrier movement is thesame as in Figure 6.3, although electron flowis reversed.

There is a definite ratio among the emit-ter current (Ie), base current (Ib), and collec-tor current (Ic). In normal operation,

(6.2)

(6.3)

Normally, 95–99% of the emitter currentflows in the collector circuit (i.e., 0.95 Ie ≤ Ic ≤0.99 Ie) and only 1–5% of the emitter currentflows in the base circuit.

The collector and emitter currents aredifferent by the amount of the base current,as indicated by equation 6.1, and their ratio(Ic/Ie) is called the alpha (α) factor. The alphafactor is one way to express transistor gain.

TRANSISTOR GAIN

There are actually several popular ways todenote transistor current gain, but only twoare of interest to us here: alpha (α) and beta(β). Alpha gain (α) can be defined as the ra-tio of collector current to emitter current:

(6.4)

where

α is the alpha gain;Ic is the collector current;Ie is the emitter current.

Alpha has a value less than unity (1),with values between 0.7 and 0.99 the typicalrange.

The other representation of transistorgain, and the one that seems more often fa-vored over the others, is the beta (β), which

is defined as the ratio of collector current tobase current:

(6.5)

where

β is the beta gain;Ic is the collector current;Ib is the base current.

Alpha (α) and beta (β) are related toeach other, and one can use the followingequations to compute one when the other isknown:

(6.6)

(6.7)

The values just given are for static DCsituations. In AC terms, you see AC alphagain (Hfb) defined as

(6.8)

and AC beta gain (Hfe) is defined as

(6.9)

In both equations, the Greek letter delta(∆) indicates a small change in the parameterit is associated with. Thus, the term ∆Ic de-notes a small change in collector current Ic.

LEAKAGE CURRENTS

Another current flowing in the transistor ofFigure 6.3 is the collector-to-base leakagecurrent. Recall from the earlier discussion ofPN junctions that the leakage current is themovement of minority charge carriers acrossthe PN junction under the influence of thebarrier potential. This leakage is designatedICBO in Figure 6.3.

H

I

Ifec

b

= ∆∆

H

I

Ifbc

e

= ∆∆

β αα

=−1

α ββ

=+1

β = I

Ic

b

α = I

Ic

e

I Ic b<<

I I Ie b c= +


A note about notation is in order at thispoint. Several specified currents in the tran-sistor are measured with one of the threeterminals (C, B, and E) open circuited. Thebasic notation is ICBE, with the letter for theopen-circuit terminal replaced with an “oh”(O). Therefore, ICBO is the collector-to-basecurrent as measured with the emitter opencircuited.

CLASSIFICATION BY COMMONELEMENT

The common element method of classifyingamplifier circuits revolves around notingwhich element (collector, base, or emitter) iscommon to both input and output circuits.Although technically incorrect, this is some-times referred to as the grounded element(i.e., grounded emitted amplifier). We tend touse the terms common and grounded inter-changeably, so bear with us if you are apurist. Figure 6.4 shows the different entriesinto this class.

Common Emitter Circuits

The circuit shown in Figure 6.4A is the com-mon emitter circuit. It gets its name becausethe emitter terminal of the transistor is com-mon to both input and output circuits. Theinput signal is applied to the transistor be-

tween the base and emitter terminals, whilethe output signal is taken across the collectorand emitter terminals; that is, the emitter iscommon to both input and output circuits.

The common emitter circuit offers highcurrent amplification—the beta rating of thetransistor. But the common emitter circuit alsooffers a substantial amount of voltage gain aswell. The transistor also is a voltage amplifier,especially when a series resistor is placed be-tween the collector terminal and the collectorDC power supply. The values of gain for cur-rent and voltage are vastly different, however.The current gain is Hfe, but the voltage gaindepends on other factors as well. Later, youwill see that voltage gain depends on theRL/RE ratio in some circuits and the product ofthat ratio and the beta in other cases.

The input impedance of the commonemitter amplifier is medium range, in the 1000Ω range. The output impedance, though, istypically high (up to 50 kΩ). Typical valuesare determined by the specific type of circuit,but some approximations can be made. Formost common emitter amplifiers, Zin is equalto the product of the emitter resistor, RE, andthe Hfe of the transistor. The output imped-ance essentially is the value of the collectorload resistor and will range from 5 kΩ toabout 50 kΩ.

The output signal in the common emit-ter circuit is 180º out of phase with the inputsignal. This means that the common emitteramplifier is an inverter circuit. The output


Fig. 6.4 Bipolar transistor amplifier configurations.

INPUTOUTPUT

INPUT

OUTPUT

INPUT OUTPUT

A B

C

signal will be negative going for a positive-going input signal and vice versa. The com-mon emitter transistor amplifier probably isthe most often used circuit configuration.

Common Collector Circuits

This configuration is shown in Figure 6.4B. Inthe common collector circuit the collector ter-minal of the transistor is common to both input and output circuits. This circuit alsosometimes is called the emitter follower cir-cuit. The common collector circuit offers littleor no voltage gain. Most of the time, the volt-age gain is actually less than unity (1), but thecurrent gain is considerably higher (≈ Hfe + 1).

There is no phase inversion betweeninput and output in the emitter follower cir-cuit. The output voltage is in phase with theinput signal voltage.

The input impedance of this circuittends to be high, sometimes greater than 100kΩ at frequencies less than 100 kHz. But theoutput impedance is very low, because it islimited to the value of the emitter resistor,which can be as low as 100 Ω. This situationleads us to one of the primary applications ofthe emitter follower: impedance transforma-tion. The circuit often is used to connect ahigh-impedance source to an amplifier with alow-impedance amplifier.

The emitter follower, or common col-lector amplifier, also frequently is used as abuffer amplifier, which is an intermediatestage used to isolate two circuits from eachother. One primary example of this applica-tion is in the output circuit of oscillator cir-cuits. Many oscillators will “pull” (change)frequency, if the load impedance changes.Yet some of the very circuits used with oscil-lators naturally provide a changing imped-ance situation. The oscillator proves a lotmore stable under these conditions if anemitter follower buffer amplifier is used be-tween its output and its load.

Common Base Circuits

Common base amplifiers use the base termi-nal of the transistor as the common element

between input and output circuits (Figure6.4C); the output is taken between the col-lector and base.

The voltage gain of the common basecircuit is high, on the order of 100 or more;however, the current gain is low, usually lessthan unity. The input impedance also is low,usually less than 1000 Ω, because it is limitedto the emitter resistance. On the other hand,the output impedance is quite high. Again,there is no phase inversion between inputand output circuits.

The principal use of the common basecircuit is in high-frequency (HF) and very-high-frequency (VHF) RF amplifiers in re-ceivers. The circuit requires no neutralizationat these frequencies, so it is superior to com-mon emitter circuits. Neutralization preventsoscillation due to interelement capacitances,which provides a feedback signal that is inphase with the input signal.

DC LOAD LINES, THE “Q” POINT,AND TRANSISTOR BIASING

The DC load line is a graphical method forexpressing transistor operating conditions.The basis for the load line is the Ic vs. VCE

family of curves (Figure 6.5A). The curve istraced for a circuit similar to Figure 6.5B atseveral different base currents, as the collec-tor voltage is swept from zero to maximum.

Figure 6.5B shows a common emitterNPN transistor circuit. The supply voltage is30 V DC; the collector current Ic can varyfrom zero to a maximum of 12 mA (0.012 A).A 2500 Ω collector load resistor (RL) is con-nected between the collector of the transistorand the VCC power supply. The collector-emitter voltage (VCE) can rise to VCC when Ic =0, or drop to zero when Ic is maximum. Thecollector current is set by the base current, Ib;hence, also the base voltage, VB. At any givenpoint, the value of VCE is VCC less the voltagedrop across the load resistor (VR). Because VR

is the product IcRL,

(6.10) V V I RCE CC c L= −


TRANSISTOR BIASING

Biasing sets the operating characteristics ofany particular transistor circuit and is usuallyset by the current conditions at the base ter-minal of the device. Several different biasnetworks commonly are seen in transistorcircuits, and these are summarized here.

Fixed Current Bias

The fixed current form of bias circuit (Figure6.6A) sets a base current in the transistor at afixed resistor between the VCC power supplyand the base terminal of the transistor. Thatcurrent, Ib, is defined by

(6.11)

The circuit of Figure 6.6A presents sev-eral problems, including a dependence on

the transistor beta gain (Hfe) and the value ofVBE. Variation in actual—versus published“typical”—Hfe is approximately 0.55–0.7 V insilicon transistors (0.2–0.3 V in germaniumdevices), and this barrier potential is a func-tion of temperature. A variant of the fixedbias circuit, shown in Figure 6.6B, helpssolve some of these problems.

The principal difference between Figures6.6A and 6.6B is the use of an emitter resistor,RE. The voltage drop across this resistor mustbe added to VBE when calculating RB and is

VE = Ie × RE = [(Ic − Ib) × RE ] (6.12)

or

(6.13)

A rule of thumb for RE is that it shouldbe approximately one tenth of RL and in no

V IIC

HE cfe

= −

IV V

RbCC BE

B

=−


Fig. 6.5 DC load line: (A) transistor load line; (B) circuit diagram.

0

1

2

3

4

5

6

7

8

9

10

11

12

0 5 10 15 20 25 30 35 40

COLLECTOR-TO-EMITTER VOLTAGE (VCE)

CO

LLE

CTO

R C

UR

RE

NT

(I C

)IB = 90 µA

IB = 80 µA

IB = 70 µA

IB = 50 µA

IB = 30 µA

IB = 0 µA

VCC

+ -

RLRB

IC

IB

IE

VR

VCE

A B

case be more than one fifth of RL; most of thetime 10 Ω ≤ RE ≤ 1000 Ω.

For the circuit of Figure 6.6B, the fol-lowing relationships obtain:

Zo = RL

Zin = RE Hfe

AI = Hfe

Av = RL Hfe/RE

Collector-to-Base Bias

In collector-to-base bias network, the resistorsupplying bias current to the base (RB) is con-

nected to the collector of the transistor ratherthan VCC (see Figure 6.7). An advantage ofthis circuit is that the quiescent (no signal)conditions are stabilized somewhat. The sta-bilization occurs because Ib is set by VCE

rather than VCC. Therefore, when Ic tries to in-crease, the voltage drop across RL increases,and because VCE = VCC − VRL, the value of VCE

decreases. This action, in turn, reduces Ib so,by Ic = Hfe IB, the collector current decreases.A similar action takes place when Ic tries todecrease. The result in both cases is that Ic

tends to stabilize around the quiescent value.


Fig. 6.6 Fixed current bias circuit: (A) schematic showing several problems; (B) solutions.

VCC

+ -

RLRB

IC

IB

IE

VR

VCE

VCB = VCC - VBE

VBE

VCC

+ -

RLRB

IC

IB

IE

VR

VCE

RE VE

Fig. 6.7 Collector-to-base bias network: (A) schematic diagram; (B) with an emitter resistor.

VCC

+ -

RL

RB

VR

VCE

IC

IB

IE

VCC

+ -

RL

RB

RE

A B

A B

It sometimes is prudent to gain furtherstability by inserting an emitter resistor, as inFigure 6.7B. For the circuit of Figure 6.7B,

Zo = RL

Zin = RE Hfe

AI = Hfe

Av = RLHfe /RE

Emitter Bias or “Self-Bias”

The emitter bias or “self-bias” of Figure 6.8 isrecognized as the most stable configurationfor transistor amplifier stages. This circuituses a resistor voltage divider (R1/R2) to seta fixed bias voltage (VB) on the transistor. Asa general rule, the best stability usually oc-curs when R1/R2 ≈ RE. Because there is asubstantial voltage drop across RE, the VCC

voltage required for Figure 6.8 is a bit higherthan for previous circuits.

FREQUENCY CHARACTERISTICS

Transistors, like most other electron devices,operate only over a certain specified fre-quency range. Three basic cutoff frequenciesmay interest us: alpha, beta, and the gain-bandwidth product (Ft).

The alpha cutoff frequency, Fab, is thefrequency at which the AC current gain, Hfb,drops to a level 3 dB below its low-frequency(usually 1000 Hz) gain. This is the frequencyat which Hfb = 0.707Hfbo, where Hfbo is the ACcurrent gain at 1000 Hz.

The beta cutoff frequency is similarlydefined as the frequency where the AC beta,Hfe, drops 3 dB relative to its 1000 Hz value.In general, this frequency is lower than the al-pha cutoff but is considered somewhat morerepresentative of a transistor’s performance.

The frequency specification that seemsto be quoted most often is the gain-band-width product, which is given the symbol Ft.This parameter usually is accepted only fortransistors operated in the common emitterconfiguration. It is defined as Ft = gain ×bandwidth:

(6.14)

where

Ft is the gain-bandwidth product;Hfe is the AC beta;F0 is the frequency at which gain is

measured.

The value of Ft quoted in specificationsheets is the frequency at which Hfe drops tounity.

If the beta cutoff frequency, Fae, isknown, then the gain-bandwidth productmay be approximated from

(6.15)

It must be recognized, however, thatthis is an approximation that may not holdup in every case. Also, you often can getaway with assuming that F0 is approximatelyequal to, but usually slightly less than, the al-pha cutoff frequency.

JFET AND MOSFET CONNECTIONS

Figure 6.9 shows the JFET and MOSFET con-figurations similar to the Figure 6.4 connec-tions for bipolar transistors. Figure 6.9Ashows a common source circuit, which is

F F Ht ae fe0= ×

F H Ft fe 0= ×


Fig. 6.8 Emitter bias or “self-bias” configurationof an NPN amplifier.

VCC

+ -

RLR1

RER2

VBEVB

VE

VCE

similar to the common emitter circuit. Figure6.9B shows the common drain circuit, whichis similar to the common collector circuit.Finally, Figure 6.9C shows the common gatecircuit, which is similar to the common basecircuit in bipolar technology.

JFET PRESELECTOR

Figure 6.10 shows the basic form of the junc-tion field effect transistor (JFET) preselector.This circuit will work into the low-VHF re-gion. This circuit is in the common sourceconfiguration, so the input signal is appliedto the gate and the output signal is takenfrom the drain. Source bias is supplied by thevoltage drop across resistor R2 and the drainload by a series combination of a resistor(R3) and a radio frequency choke (RFC1).The RFC should be 1000 µH (1 mH) at theAM broadcast band and HF (shortwave), and100 µH in the low-VHF region (>30 MHz). AtVLF frequencies below the broadcast band,use 2.5 mH for RFC1 and increase all 0.01 µF


Fig. 6.9 JFET amplifier configurations:(A) common source circuit; (B) common drain circuit; (C) common gate circuit.

INPUT OUTPUT

INPUTOUTPUT

INPUT

OUTPUT

A

B

C

Fig. 6.10JFET common emitter RF amplifier.

J1INPUT

Q1MPF-102

L1L2

C1365 pF

C2100 pF

R1220K

R2100

C40.01 µF

C30.001 µF

RFC11 mH

C50.01 µF

C60.01 µF

R3270

+12 VDC

capacitors to 0.1 µF. All capacitors are eitherdisk ceramic, or one of the newer dielectriccapacitors (if rated for VHF service—be care-ful, not all are).

The input circuit is tuned to the RF fre-quency, but the output circuit is untuned. Thereason for the lack of output tuning is thattuning both input and output permits theJFET to oscillate at the RF frequency—and wedo not want that. Other possible causes of os-cillation include layout or a self-resonancefrequency of the RFC that is too near the RFfrequency (select another choke).

The input circuit consists of an RF trans-former that has a tuned secondary winding(L2/C1). The variable capacitor (C1) is thetuning control. Although the value shown isthe standard 365 pF “AM broadcast variable,”any form of variable can be used if the in-ductor is tailored to it. These components arerelated by

(6.16)

where

f is the frequency in hertz;L is the inductance in henrys;C is the capacitance in farads.

Be sure to convert inductances from microhenrys to henrys, and picofarads tofarads. Allow approximately 10 pF to accountfor stray capacitances, although keep in mindthat this number is a guess that may have tobe adjusted (it is a function of your layout,among other things). We also can solveequation 6.16 for either L or C:

(6.17)

Space does not warrant making a samplecalculation, but we can report results for youto check for yourself. In a sample calculation,I wanted to know how much inductance is re-quired to resonate 100 pF (90 pF capacitorplus 10 pF stray) to 10 MHz WWV. The solu-tion, when all numbers are converted to hertzand farads, results in 0.00000253 H or 2.53

µH. Keep in mind that the calculated numbersare close but nonetheless approximate—andthe circuit may need tweaking on the bench.

Be careful when making JFET or MOS-FET RF amplifiers in which both input andoutput are tuned. If the circuit is a commonsource circuit (i.e., the input signal is acrossthe gate and source) and the output signal isbetween the drain and source, there is thepossibility of accidentally turning the circuitinto a dandy little oscillator. Sometimes, thisproblem is alleviated by tuning the input andoutput LC tank circuits to slightly differentfrequencies. In other cases, it is necessary toneutralize the stage. It is a common practiceto make at least one end of the amplifier,usually the output, untuned to overcome thisproblem (although at the cost of some gain).

Figure 6.11 shows two methods for tun-ing both the input and output circuits of theJFET transistor. In both cases, the JFET iswired in the common gate configuration, sothe signal is applied to the source and outputis taken from the drain. The dashed line indi-cates that the output and input tuning capac-itors are ganged to the same shaft.

The source circuit of the JFET is low im-pedance, so some means must be providedto match the circuit to the tuned circuit. InFigure 6.11A, a link inductor is used for L1for the lower impedance (50 Ω typically) ofthe source. In Figure 6.11B, a similar butslightly different configuration is used. In thisexample the circuit has a bias resistor, and itis bypassed by C2. This keeps the potentialfor DC but sets the AC impedance to ground.

VHF RECEIVER PRESELECTOR

The circuit in Figure 6.12 is a VHF preampli-fier that uses two JFET devices connected incascode; that is, the input device (Q1) is incommon source and directly coupled to thecommon gate output device (Q2). To pre-vent self-oscillation of the circuit a neutral-ization capacitor (NEUT) is provided. Thiscapacitor is adjusted to keep the circuit fromoscillating at any frequency within the bandof operation. In general, this circuit is tuned

L

f C= 1

39 5 2.

fLC

= 1

2π



Fig. 6.11 Two methods of tuning a JFET common base RF amplifier.

J1IN L1

L2

C1A C1B

Q1C2

0.001 µF

R1150

J2OUT

RFC11 mH

C30.001 µF

C40.01 µF

C50.01 µF

R2270

L3

+12 VDC

Q1

L1

J1IN

C1

R1150

C20.1 µFA

B

Fig. 6.12 Cascode pair in an RF amplifier.

G1

G2

S

D

SUBSTRATE

J1INPUT L1 L2 C1

Q1

Q2

C40.001

µFR1100

C3NEUTRALIZATION

C70.001 µF

R2100K

C2L3

L4

J2OUTPUT

C60.001 µF

R3270

+12VDC

to a single channel by the action of L2/C1and L3/C2.

MOSFET PRESELECTOR

The 40673 dual-gate MOSFET (Figure 6.13)used in the following preselector circuit is lowcost and readily available. It is a dual-gateMOSFET, so one gate can be used for amplifi-cation and the other for DC-based gain con-trol. The signal is applied to gate G1, whilegate G2 is either biased to a fixed positive volt-age or connected to a variable DC voltage thatserves as a gain control signal. The DC net-work is similar to that of the previous (JFET)circuits, with the exception that a resistor volt-age divider (R3/R4) is needed to bias gate G2.

This preselector project has three tunedcircuits, so it will produce a large amount ofselectivity improvement and image rejection.The gain of the device also will provide addi-tional sensitivity. All three tuning capacitors(C1A, C1B, and C1C) are ganged to the sameshaft for “single-knob tuning.” The trimmer

capacitors (C2, C3, and C4) are used to adjustthe tracking of the three tuned circuits (i.e.,ensure that all are tuned to the same fre-quency at any given setting of C1A–C).

The inductors are of the same sort asdescribed previously. It is permissible to putL1/L2 and L3 in close proximity to eachother, but these should be separated from L4to prevent unwanted oscillation due to feed-back arising from coil coupling.

VOLTAGE TUNED RECEIVERPRESELECTOR

The circuit in Figure 6.14 is a little different. Inaddition to using only input tuning (whichlessens the potential for oscillation), it alsouses voltage tuning. The hard-to-find variablecapacitors are replaced with varactor diodes,also called voltage variable capacitancediodes (D1). These PN junction diodes exhibita capacitance that is a function of the appliedreverse bias potential, VT. Although the origi-nal circuit was built and tested for the AM


Fig. 6.13 MOSFET RF amplifier.

G1

G2

S

D

J1IN L1

L2 C1A C1B

C1C

C2

C52.7 pF

L3 C3

C6100 pF

R1220K

R2100 C7

0.01 µF

Q240673

C80.001 µF

L4 C4

J2OUT

RFC11 mH

R5270 C9

0.01 µFC11

0.01 µF

R456K

R310K

C101000 pF

F.T.+12VDC

broadcast band (540–1700 kHz), it can bechanged to any band by correct selection ofthe inductor values. The designated varactoroffers a capacitance range of 440 pF down to15 pF over the voltage range 0 to +18 V DC.

The inductors may be either “store-bought” types or wound over toroidal cores.I used a toroid for L1/L2 (forming a fixed in-ductance for L2) and “store-bought” ad-justable inductors for L3 and L4. There is noreason, however, why these same inductorscannot be used for all three purposes.Unfortunately, not all values are available inthe form that has a low-impedance primarywinding to permit antenna coupling.

In both MOSFET circuits, the fixed biasnetwork used to place gate G2 at a positiveDC potential can be replaced with a variablevoltage circuit. The potentiometer in Figure6.15 can be used as an RF gain control to re-duce gain on strong signals and increase it

on weak signals. This feature allows the ac-tive preselector to be custom set to preventoverload from strong signals.


Fig. 6.14 Voltage tuned MOSFET RF amplifier.

J1IN L1

C45 pF

G1

G2

S

DC6100 pF

R4220K

R5100 C12

0.01 µF

Q240673

C110.001 µF

RFC11 mH

R5270 C10

0.01 µFC7

R456K

R310K

+12VDC

J2OUT

L2L3 L4

C81000 pF

F.T.

C131000 pF

F.T.

VT

C10.01 µF

C20.01 µF

C30.01 µF

D1 D2 D3

R1150K

R2150K

R3150K

C55 pF

C90.01 µF

0.01 µF

Fig. 6.15 Active preselector tuning scheme.

G1

G2

S

D

+12VDC

R110K

R250K

GAIN

R36.8K

C10.1 µF

Q1

BROADBAND RF PREAMPLIFIER FORVLF, LF, AND AM BCB

A broadband RF amplifier is needed in manysituations. Typical applications include boost-ing the output of RF signal generators(which tend to be normally at a quite lowlevel), antenna preamplification, loop an-tenna amplification, and in the front end ofreceivers. A number of different circuits havebeen published, including some by me, butone failing that I have noted on most ofthem is that they lack response at the lowend of the frequency range. Many designsoffer −3 dB frequency response limits of3–30 MHz, or 1–30 MHz, but rarely is thevery-low-frequency (VLF), low-frequency(LF), or even the entire AM broadcast band(540–1700 kHz) covered.

Originally, I needed an amplifier toboost AM BCB signals. Many otherwise finecommunications or entertainment grade“general coverage” receivers operate from100 kHz to 30 MHz or so, and that range ini-tially sounds real good to the VLF throughAM BCB owner. But closer examinationfound that the receiver lacked sensitivity onthe bands below either 2 or 3 MHz, so itfailed somewhat in the lower end of thespectrum. While most listening on the AMBCB is to powerful local stations (where re-ceivers with no RF amplifier and a loopstickantenna will work nicely), those interested inDXing are not well served. In addition to thereceiver, I wanted to boost my signal genera-tor 50 Ω output to make it easier to developsome AM and VLF projects I am working onand provide a preamplifier for a square loopantenna that tunes the AM BCB.

Several requirements were developedfor the RF amplifier. First, it had to retain the50 Ω input and output impedances standardin RF systems. Second, it had to have a highdynamic range and third-order intercept pointto cope with the bone crunching signal levelson the AM BCB. One of the problems of theAM BCB is that those sought-after distant sta-tions tend to be buried under multikilowattlocal stations on adjacent channels. That iswhy high dynamic range, high intercept

point, and loop antennas tend to be requiredin these applications. I also wanted the ampli-fier to cover at least two octaves (4:1 fre-quency ratio) and, in fact, achieved a decade(10:1) response (250–2500 kHz).

Furthermore, the amplifier circuit hadto be easily modifiable to cover other fre-quency ranges up to 30 MHz. This last re-quirement would make the amplifier usefulto a large number of readers, as well as ex-tending its usefulness to me.

A number of issues must be consideredwhen designing an RF amplifier for the frontend of a receiver. The dynamic range and in-tercept point requirements were mentionedpreviously. Another issue is the amount ofdistortion products (related to the third-orderintercept point) generated in the amplifier. Itdoes no good to have high capability on thepreamplifier, only to overload the receiverwith a lot of extraneous RF energy it cannothandle—energy that was generated by thepreamplifier, not from the stations being re-ceived. These considerations point to the useof a push-pull RF amplifier design.

PUSH-PULL RF AMPLIFIERS

The basic concept of a push-pull amplifier isdemonstrated in Figure 6.16. This type of cir-cuit consists of two identical amplifiers, eachof which processes half the input sine wavesignal. In the circuit shown, this job is accom-plished by using a center-tapped transformerat the input to split the signal and another atthe output to recombine the signals from thetwo transistors. The transformer splits the sig-nal because its center tap is grounded and soserves as the common for the signals appliedto the two transistors. Because of normaltransformer action, the signal polarity at endA will be opposite that at end B when thecenter tap (CT) is grounded. Thus, the twoamplifiers are driven 180º out of phase witheach other: one turning on as the other turnsoff and vice versa.

The push-pull amplifier circuit is bal-anced. As a result, it has a very interestingproperty: Even order harmonics are cancelled


in the output, so the amplifier output signalwill be cleaner than for a single-ended ampli-fier using the same active amplifier devices.

There are two general categories ofpush-pull RF amplifiers: tuned amplifiers andwideband amplifiers. Tuned amplifiers havethe inductance of the input and output trans-formers resonated to some specific frequency.In some circuits, the nontapped winding maybe tuned, but in others a configuration such asFigure 6.17 might be used. In this circuit, bothhalves of the tapped side of the transformerare individually tuned to the desired resonantfrequency. Where variable tuning is desired, asplit-stator capacitor might be used to supplyboth capacitances.

A broadband circuit is shown in Figure6.18A. In this type of circuit, a special trans-former usually is needed. The transformermust be a broadband RF transformer, which

means that it must be wound on a suitablecore so that the windings are bifilar or trifi-lar. The particular transformer in Figure6.18A has three windings, of which one ismuch smaller than the others. These must betrifilar wound for part of the way and bifilarthe rest of the way. This means that all threewindings are kept parallel until no moreturns are required of the coupling link. Thenthe remaining two windings are kept paralleluntil they are completed. Figure 6.18Bshows an example for the case where thecore of the transformer is a ferrite or pow-dered iron toroid.

The actual RF circuit is shown in Figure6.19. The active amplifier devices are junctionfield effect transistors (JFET) intended for ser-vice from DC to VHF. The device selected canbe the ever-popular MPF-102, or some similardevice. Also useful is the 2N4416 device. Theparticular device I used was the NTE-451JFET transistor. This device offers a transcon-ductance of 4000 µSiemens (1 µSiemen = 1µMho), a drain current of 4–10 mA, and apower dissipation of 310 mW, with a noisefigure of 4 dB maximum.

The JFET devices are connected to a pairof similar transformers, T1 and T2. The sourcebias resistor (R1) for the JFETs and its associ-ated bypass capacitor (C1) are connected tothe center tap on the secondary winding oftransformer T1. Similarly, the +9 V DC power


Fig. 6.16 Push-pull amplifier in a block diagram form.

Vdc

+-

A1

A2

T1 T2

INPUT OUTPUT

A

B

CT

Fig. 6.17 Tuned amplifier circuit.

supply voltage is applied through a limitingresistor (R2) to the center tap on the primarywinding of transformer T2.

Take special note of those two trans-formers. These transformers, generally knownas wideband transmission line transformers,can be wound on either toroid or binocularferrite or powdered iron cores. For this proj-ect, because of the low frequencies involved, Iselected a type BN-43-202 binocular core. Thetype 43 material used in this core is a good se-lection for the frequency range involved. Each

transformer has three windings. In each case,the B and C windings are 12 turns of #30AWG enameled wire wound in a bifilar man-ner. The coupling link in each is winding A.The A winding on transformer T1 consists offour turns of #36 AWG enameled wire; whileon T2, it consists of two turns of the samewire. The reason for the difference is that thenumber of turns in each is determined by theimpedance matching job it must do (T1 has a1:9 primary/secondary ratio, while T2 has a36:1 primary/secondary ratio). Neither the


Fig. 6.18 Broadband push-pull RF amplifier: (A) output circuit; (B) physical implementation.

A1

A2

T2

a2

a1b1

b2c2

c1

a2

a1

b1b2

c1c2

Fig. 6.19Circuit diagram for thepush-pull amplifier.

+

+9VDC

C310 µF

C40.82 µF

R227

Q2

J2OUTPUT

T2

Q1

C20.82 µF

C10.82 µF

R127T1

J1INPUT

A B

source nor drain impedances of this circuit is50 Ω (the system impedance), requiring animpedance transformation function. If the twoamplifiers in the circuit were of the sort thathad 50 Ω input and output impedances, suchas the Mini-Circuits MAR-1 through MAR-8 de-vices, then winding A in both transformerswould be identical to windings B and C. Inthat case, the impedance ratio of the trans-formers would be 1:1:1.

The details for transformers T1 and T2are shown in Figure 6.20. I elected to build aheader of printed circuit perforated board forthis part; the board holes are on 0.100 in.centers. The PC type of perf board has asquare or circular printed circuit solderingpad at each hole. A section of perf board wascut with a matrix of five holes by nine holes.Vector Electronics push terminals are insertedfrom the unprinted side, and then solderedinto place. These terminals serve as anchorsfor the wires that will form the windings ofthe transformer. Two terminals are placed atone end of the header, and three at the op-posite end.

The coupling winding is connected topins 1 and 2 of the header, wound first oneach transformer. Strip the insulation from alength of #36 AWG enameled wire for about1/4 in. from one end. This can be done by

scraping with a scalpel of X-acto knife, orby burning with the tip of a soldering pen-cil. Ensure that the exposed end is tinnedwith solder, then wrap it around terminalno. 1 of the header. Pass the wire throughthe first hole of the binocular core, acrossthe barrier between the two holes, and thenthrough the second hole. This U-shapedturn counts as one turn. Make transformerT1 pass the wire through both sets of holesthree more times (to make four turns). Thewire should be back at the same end of theheader as it started. Cut the wire to allow ashort length to connect to pin no. 2. Cleanthe insulation off this free end, tin the ex-posed portion, wrap it around pin no. 2,and solder it. The primary winding of T1 isnow completed.

The two secondary windings arewound together in the bifilar manner andconsist of 12 turns each of #30 AWG enam-eled wire. The best approach seems to betwisting the two wires together. I use anelectric drill to accomplish this job. Twopieces of wire, each 30 in. long, are joinedtogether and chucked up in an electric drill.The other ends of the wire are joined to-gether and anchored in a bench vise orsome other holding mechanism. I then backoff, holding the drill in one hand, until the


Fig. 6.20Transformer winding details.

CORE

TAPEPERF

BOARD

1

2

3

4

5

ONETURN

TWOTURNS

wire is nearly taut. Turning on the drillcauses the two wires to twist together. Keeptwisting them until you obtain a pitch ofabout 8–12 twists per inch.

It is very important to use a drill thathas a variable speed control, so that the drillchuck can be made to turn very slowly. Italso is very important that you follow safetyrules, especially as regards your eyesight,when making twisted pairs of wire. Be ab-solutely sure to wear either safety glasses orgoggles while doing this operation. If thewire breaks—and that is a common prob-lem—it will whip around as the drill chuckturns. While #36 wire may not seem verysubstantial, at high speed it can severely in-jure an eye.

To start the secondary windings, scrapeall the insulation off both wires at one end ofthe twisted pair and tin the exposed endswith solder. Solder one wire to pin no. 3 ofthe header and the other to pin no. 4. Passthe wire through the hole of the core closestto pin no. 3, around the barrier, then throughthe second hole, returning to the same endof the header as you started. That constitutesone turn. Now do it 11 more times until all12 turns are wound. When the 12 turns arecomplete, cut the twisted pair wires to leaveabout 1/2 in. free. Scrape and tin the ends ofthese wires.

Connecting the free ends of the twistedwire is easy, but you will need an ohmmeteror continuity tester to see which wire goeswhere. Identify the end that is connected atits other end to pin no. 3 of the header, andconnect this wire to pin no. 4. The remainingwire should be the one that was connectedat its other end to pin no. 4; this wire shouldbe connected to pin no. 5 of the header.

Transformer T2 is made in the samemanner as transformer T1 but with only twoturns on the coupling winding rather thanfour. In this case, the coupling winding isthe secondary one, while the other twoform two halves of the primary winding.Wind the two-turn secondary winding first,as was done with the four-turn primarywinding on T1.

The amplifier can be built on the samesort of perforated board as was used to makethe headers for the transformers. Indeed, theheaders and the board can be cut from thesame stock. The size of the board will de-pend somewhat on the exact box you selectto mount it in.

BROADBAND RF AMPLIFIER (50 Ω INPUT AND OUTPUT)

This project is an RF amplifier that can beused in a variety of ways: as a preamplifierfor receivers operating in the 3–30 MHzshortwave band or as a postamplifier follow-ing filters, mixers, and other devices thathave an attenuation factor. It is common, forexample, to find that mixers and crystal fil-ters have a signal loss of 5–8 dB (called in-sertion loss). An amplifier following thesedevices will overcome that loss. The ampli-fier also can be used to boost the outputlevel of signal generator and oscillator cir-cuits. In this service, it can be used eitheralone, in its own shielded container, or aspart of another circuit containing an oscilla-tor circuit.

The circuit is shown in Figure 6.21. Thetransistor (Q1) is a 2N5179 broadband RFtransistor. It can be replaced by the NTE-316or ECG-316 devices, if the original is unavail-able. The NTE and ECG devices are intendedfor service and maintenance replacement ap-plications, so they tend to be found in localelectronic parts distributors.

This amplifier has two main features:the degenerative feedback in the emitter cir-cuit and the feedback from collector to base.Degenerative, or negative, feedback is usedin amplifiers to reduce distortion (i.e., makeit more linear) and stabilize the amplifier.

One negative feedback mechanism ofthis amplifier is seen in the emitter. The emit-ter resistance consists of two resistors, R5 is10 Ω and R6 is 100 Ω. In most amplifier cir-cuits, the emitter resistor is bypassed by a ca-pacitor to set the emitter of the transistor atground potential for RF signals, while keep-


ing it at the DC level set by the resistance. Innormal situations, the reactance of the capac-itor should be not more than one tenth theresistance of the emitter resistor. The 10 Ωportion of the total resistance is left unby-passed, forming a small amount of negativefeedback.

The collector-to-base feedback is ac-complished by two means. First, a resistor-capacitor network (R1/R3/C2) is used; second,a 1:1 broadband RF transformer (T1) is used.This transformer can be homemade. Wind 15bifilar turns of #26 enameled wire on atoroidal core such as the T-50-2 (RED) or T-50-6 (YEL); smaller cores can be used as well.

The circuit can be built on perforatedwire board that has a grid of holes on 0.100in. centers. You can use a homebrew RFtransformer made on a small toroidal core.Use the size 37 core, with #36 enameled

wire. As in the previous case, make the twowindings bifilar.

BROADBAND OR TUNED RF/IFAMPLIFIER USING THE MC-1350P

The MC-1350P is a variant of the MC-1590device, but unlike the 1590, it is available inthe popular, easy-to-use eight-pin mini-DIPpackage. It has gain sufficient to make a 30dB amplifier, although it is a bit finicky andtends to oscillate if the circuit is not built cor-rectly. Layout, in other words, can be a verycritical factor because of the gain.

If you cannot find the MC-1350P, seekout the NTE-746 or ECG-746. These devicesare MC-1350Ps but are sold in the service andmaintenance replacement lines and usuallyare available locally.


Fig. 6.21Bipolar RF amplifier.

C10.1 µF

INPUT

T1

R1560

R33.3k

R21k

R510

R6100

C50.1 µF

C20.1 µF

C40.1 µF

C30.1 µF

+12VDC

R4150

R768

C60.1 µF

OUTPUT

Q12N5179

Figure 6.22 shows the basic circuit forthe MC-1350P amplifier. The signal is appliedto the −IN input, pin no. 4, while the +IN in-put is decoupled to ground with a 0.1 µF ca-pacitor. All capacitors in this circuit, exceptC6 and C7, should be disk ceramic or one ofthe newer dielectrics competent at RF fre-quencies to 30 MHz. A capacitor in serieswith the input terminal, C1, is used to pre-vent DC riding on the signal from affectingthe internal circuitry of the MC-1350P.

The output circuitry is connected to pinno. 1 of the MC-1350P. Because this circuit isbroadband, the output impedance load is aradio frequency choke (L1). For most HF ap-plications, L1 can be a 1 mH choke; althoughfor the lower end of the shortwave region,the medium wave band, and the AM broad-cast band, use a 2.5 mH choke. The same cir-

cuit can be used for 455 kHz IF amplifier ser-vice if the coil (L1) is made 10 mH.

Pin no. 5 of the MC-1350P device isused for gain control. This terminal needs tosee a voltage of +5 to +9 V, with the maxi-mum gain being found at the +5 V end of therange (this is opposite what is seen in otherchips). The gain control pin is bypassed forRF signals.

The DC power supply is connected topins 8 and 2 simultaneously. These pins aredecoupled to ground for RF by capacitor C4.The ground for both signals and DC powerare at pins 3 and 7. The V+ is isolated some-what by a 100 Ω resistor (R3) in series withthe DC power supply line. The V+ line is de-coupled on either side of this resistor by elec-trolytic capacitors. C6 should be a 4.7–10 µFtantalum unit, while C7 should be a 68 µF


Fig. 6.22 The MC-1350 used as a wideband RF amplifier.

+12 VDC

U1MC-1350P

INPUT

C10.1 µF

C20.1 µF

C30.1 µF

C40.1 µF

C66.8 µF

C7100 µF

R3100

R21500

L1

C50.5 µF

OUTPUT

++

4

6

82

1

5 3 7

GAINCONTROLVOLTAGE

R110K

1

TOPVIEW

SHIELD


(or greater) tantalum or aluminum electrolytic capacitor.

A partial circuit with an alternate outputcircuit is shown in Figure 6.23. This circuit istuned, rather than broadband, so might beused for IF amplification or RF amplificationat specific frequencies. Capacitor C8 is con-nected in parallel with the inductance of L1,tuning L1 to a specific frequency. To keepthe circuit from oscillating, the resonant tank

circuit is “de-Qed” by connecting a 2.2 kΩ re-sistor in parallel with the tank circuit.Although considered optional in Figure 6.23,it is not optional in this circuit if you want toprevent oscillation (the MC-1350P device hasa tendency to oscillate at higher gains). Onetactic to prevent the oscillation is to use ashield between the input and output of theMC-1350P.

Fig. 6.23 Alternative output tuning scheme for the MC-1350P.

+12 VDC

U1MC-1350P

C40.1 µF

C66.8 µF

C7100 µF

R3100

R21500

L1OUTPUT

++

82

1

C8TUNE L2

A mixer is a nonlinear circuit or device thatpermits frequency conversion (“translation”)by the process of heterodyning. Mixers areused in the front end of the most commonform of radio (the superheterodyne), in cer-tain electronic instruments, and in certainmeasurement schemes (receiver dynamicrange, oscillator phase noise, etc.).

The block diagram for a basic mixersystem is shown in Figure 7.1; this diagram,although generic, represents the front endof superheterodyne radio receivers. Themixer has three ports: F1 receives a low-level signal and would correspond to theRF input from the aerial in radio receivers;F2 is a high-level signal, corresponding tothe local oscillator (LO) in superhet radios;and F3 is the resultant mixer product (cor-responding to the intermediate frequencyin superhet radios). These frequencies arerelated by

F3 = mF1 ± nF2 (7.1)

where F1, F2, and F3 are as just describedand m and n are counting numbers (0 plusintegers 1, 2, 3, . . .).

In any given circuit, m and n can be 0or any integer, but in practical circuits it iscommon to consider only the first-, sec-ond-, and third-order products. For sake ofsimplicity, we consider a first-order circuit(m = n = 1). Such a mixer would outputfour frequencies: F1, F2, F3a = F1 + F2, andF3b = F1 − F2. In terms of a radio receiver,these frequencies represent the RF inputsignal, the local oscillator signal, the sumIF, and the difference IF. In radios, it iscommon practice to select either the sumor the difference IF by filtering and reject-ing all others.

A diplexer and filter stage, shown inFigure 7.1, are used to absorb unwanted mixerproducts and pass the desired frequencies.The filter selects which frequencies are fa-vored and which are rejected. Frequency se-lective circuits are discussed later.

A postamplifier stage sometimes fol-lows the diplexer, because the insertion lossof most passive double-balanced mixers(DBMs; discussed later) is considerable (5–12dB). The purpose of the amplifier is to makeup for the loss of signal level in the mixing

Chapter 7

RF Mixer and FrequencyConverter Circuits

91

process. Some active DBMs, incidentally,have a conversion gain figure, not a loss. Forexample, the popular Signetics NE-602 de-vice offers 20 dB of conversion gain.1

DIPLEXER CIRCUITS

The RF mixer is like most RF circuits in that itwants to be terminated in its characteristicimpedance. Otherwise, a standing-wave ratio(SWR) problem will result, causing signalloss and other problems. In addition, certainpassive DBMs (on which, more later) will notwork well if improperly terminated. A num-ber of different diplexer circuits are known,

and two of the most popular are shown inFigures 7.2 and 7.3.

A diplexer has two jobs: (1) It absorbsundesired mixer output signals, so they arenot reflected back into the mixer; (2) ittransmits desired signals to the output. InFigure 7.2, these goals are met with two dif-ferent L-C networks: a high-pass filter and alow-pass filter. The assumption in this cir-cuit is that the difference IF is desired, so ahigh-pass filter with a cutoff above the dif-ference IF is used to shunt the sum IF (plusLO and RF signals that survived the DBMprocess) to a dummy load (R1). The dummyload shown in Figure 7.2 is set to 50 Ω be-cause that is the most common system


Fig. 7.1 Front-end block diagram.

RFFILTER

LOCALOSCILLATOR

DIPLEXER

IFFILTER

MIXER

RFINPUT

F1

F2

F3

213 FnFmF ±=

Fig. 7.2A simple diplexer circuit withhigh- and low-pass filters.

DOUBLE-BALANCEDMIXER

IFOUT

TO POST-AMPLIFIER

L2C4

0.01 µF

C3C2

0.1 µF

L1XL = 50 OHMS

C1XC = 50 OHMS

R150 OHMS

impedance for RF circuits (in practice, a 51Ω resistor might be used). The dummy loadresistor can be a 1/4 W unit in most low-levelcases, but regardless of power level, it mustbe a noninductive type (e.g., carbon com-position or metal film).

The inductor (L1) and capacitor (C1)values in the high-pass filter are designed tohave a 50 Ω reactance at the IF frequency.These values can be calculated from

(7.2)

(7.3)

The low-pass filter transmits the desireddifference IF frequency to the output, reject-ing everything else. Like the high-pass filter,the L and C elements of this filter are de-signed to have reactances of 50 Ω at the dif-ference IF frequency.

Another popular diplexer circuit isshown in Figure 7.3. This circuit consists of aparallel resonant 50 Ω tank circuit (C1/L1), anda series resonant 50 Ω tank circuit (C2/L2). Theseries resonant circuit passes its resonant fre-quency while rejecting all others because itsimpedance is low at resonance and high atother frequencies. Alternatively, the parallelresonant tank circuit offers a high impedanceto its resonant frequency and a low impedanceto all other frequencies. Because C1/L1 areshunted across the signal line, it will short outall but the resonant frequency.

TYPES OF MIXER

There are a number of different types ofmixer circuit but only a few different genericclasses: single ended, singly balanced (or,simply, balanced), and double balanced.Most low-cost superheterodyne radio re-ceivers use single-ended mixers, although afew of the more costly “communications re-ceivers” use singly or double-balanced mixersfor improved performance.

Single-Ended Mixers

The most commonly used form of mixer cir-cuit is the single-ended mixer. These circuitsare simple and low cost but offer only mod-erate performance. The very simplest form isthe diode mixer circuit of Figure 7.4. In thiscircuit, a PN diode is connected so that its an-ode is driven with the RF signal and the LOsignal, while the cathode is tuned to the IFfrequency. Transformer T1 is shown in Figure7.4 is a broadband transformer, but in manycases the transformer will be tuned to the RFfrequency. The LO signal must have a highlevel, on the order of +7.5 dBm, so that thediode is switched in and out of conduction bythe LO signal. The output tuning is set to theIF frequency. In the version shown, the diodeis connected to the top of the L-C tank circuit.In many practical circuits, however, a tappedinductor (see the inset to Figure 7.4) is usedto improve the impedance match to the diode(which tends to be low). This form of mixerhas a loss of several decibels.

C

F=

× × −

1

2 50π Ω 3dB

L

f=

−

50

2

Ωπ 3dB

RF Mixer and Frequency Converter Circuits 93

Fig. 7.3A simple diplexer cir-cuit with parallel andseries resonant tankcircuits.

IN OUT

50 OHMS 50 OHMS

C1 L1

C2 L2

A single-ended active mixer based onthe junction field effect transistor is shown inFigure 7.5. This circuit offers no conversionloss but rather a conversion gain of up to 10dB. The JFET mixer circuit is very similar tothe standard common-source JFET RF ampli-fier, except that the output is tuned to the IFfrequency, while the input is tuned to the RFfrequency. In addition, an LO signal is capac-itively coupled to the gate of the JFET. TheLO signal amplitude must be on the order of+7.5 dBm so that the JFET can be switched inand out of conduction on successive swingsof the LO signal.

A variation on the JFET theme is shownin Figure 7.6. This circuit (shown only partially)is identical to that of Figure 7.5 except that theLO signal is applied to the source terminal ofthe JFET. The LO signal must be 5 V p-p, orabout +18 dBm, and is capacitively coupled tothe source of the transistor. Part of the emitterresistance (R2) remains unbypassed to providea good impedance match to the 50 Ω output ofthe signal source used as the LO.

The final example of a single-ended ac-tive mixer is the MOSFET circuit shown inFigure 7.7. The active device is a 40673 dual-gate RF MOSFET. In this circuit, the RF signal is

applied to gate 1, while the LO signal is appliedto gate 2. The LO signal must have a level of+17 dBm to drive the MOSFET in and out ofconduction. As in the previous FET cases, gate1 is tuned to the RF frequency, while the drain(output) is tuned to the IF frequency.

Singly Balanced Mixers

The basic diode double-balanced mixer isshown in Figure 7.8. This mixture consists oftwo diodes cross connected to conduct onopposite halves of the applied signal cycle.A trifilar wound transformer, connected inthe 4:1 manner, is used as the interface be-tween the diodes with the RF and LO sig-nals. The IF output of the circuit is at thejunction of the two diodes. Signals from thispoint normally are fed to some kind of filteror IF tuned circuit. Like the passive diodemixture shown earlier, this circuit suffersconversion signal loss.

An active singly balanced mixer is shownin Figure 7.9A. This circuit is based on using apair of JFET devices arranged in a differentialpair. In best practice, the two JFETs would beclosely matched and share a common thermalenvironment. These requirements are best


Fig. 7.4 Simple diode mixer circuit.

RFINPUT

LOINPUT

T1

D1

C10.01 µF

C2L1 L2 L3 L4

IFOUTPUT


Fig. 7.5 JFET mixer.

Q12N5486

MPF-102Etc.

LO+7.5 dBm

T1IF

IFOUTPUT

V+

R3100

C40.01 µF

R2100

L2L1 C1RF

C2100 pF

C30.001 µF

R1100K

ALTERNATEIF OUTPUT

Fig. 7.6JFET mixer with alternate LO input scheme.

0.001 µF

100 pF

R1100K

R3100

Q12N5486

MPF-102Etc.

R250

LOINPUT

5 VOLTS P-P+18 dBm


Fig. 7.7Dual-gate MOSFET mixer.

R2220

T1

C30.001 µF

+12VDC

R41000

R315K

C20.01 µF

IFOUTPUT

Q140673

R1220

L1A

LO

RFL1B

C1C2

0.01 µF

Fig. 7.8Double-balancedmixer.

RF IN

LO IN+13.5 dBm

T1

C10.01 µF

RFC1XL > 200 OHMS

D1

D2

C20.01 µF

IFOUT

R15.6K


Fig. 7.9 Singly balanced mixers: (A) JFET; (B) bipolar.

T2IF

DCBIAS

C1

T1

V+

LO

RF

Q1 Q2

Q3

V+

RF IN

LO IN+13.5 dBm

T1

Q1 Q2

R1100 C2

0.01 µF

RFC1

C10.01 µF

T2

R2100

C30.1 µF

IF OUT

A

B

met by using a dual JFET device; that is, anintegrated circuit device that contains a pairof independent JFETs. The gates of the twoJFETs are driven out of phase with eachother by signals from the trifilar wound inputtransformer (T1). The drain outputs of theJFETs are connected to a second trifilarwound transformer (T2).

Any of the active mixer circuits dis-cussed already has its equivalent in NPN andPNP bipolar transistor versions. In those cir-cuits, the RF is fed to the base terminal, whilethe LO is fed to either the base or emitter ter-minals. Of course, appropriate DC bias mustbe applied to the device. A differential mixerbased on NPN transistors is shown in Figure7.9B. Although this circuit is partial, it is rep-resentative of a wide range of actual devices.Perhaps, the one most commonly found isbased on the CA-3028A integrated circuit.The CA-3028A contains all the transistorsneeded, along with a bias network. Undermost circumstances, only a pair of bias resis-tors to serve the Q1/Q3 common base supplyis needed.

Double-Balanced Mixers

One advantage of the double-balanced mixer(DBM) over the other forms of mixer is that itsuppresses the F1 and F2 components of theoutput signal, passing only the sum and dif-ference signals. In a radio receiver using aDBM, the IF filtering and amplifier wouldhave to contend with only the sum and dif-ference IF frequencies and not bother withthe LO and RF signals. This effect is seen inDBM specifications as the port-to-port isola-tion figure, which can reach 30–60 dB, de-pending on the DBM model.

JFET AND MOSFET DOUBLE-BALANCED MIXER CIRCUITS

Junction field effect transistors and metal oxidesemiconductor transistors (MOSFETs) can bearranged in a ring circuit that provides gooddouble-balanced mixer operation. Figure 7.10shows a circuit based on JFET devices.Although discrete JFETs, such as the MPF-102


Fig. 7.10 JFET double-balanced mixer.

LPF

LPF

IFOUT

0.01 µF

R1

V+

T3

T4

Q1 Q2

Q3 Q4

0.01 µF

0.1 µF

T1 T2

or its equivalent, can be used in this circuitsuccessfully, performance generally is better ifthe devices are matched or are part of a singleintegrated circuit (IC) device (e.g., the U350IC). With due attention to layout and input/output balance, the circuit is capable of betterthan 30 dB port-to-port isolation over an oc-tave (2:1) frequency change.

The input and output of this circuit arebased on broadband RF transformers. Thesetransformers are bifilar and trifilar wound ontoroidal cores. The input circuit consists oftwo bifilar wound impedance transformers(T1 and T2). The LO and output circuits aretrifilar wound RF transformers. Part of the out-put circuit includes a pair of low-pass filtersthat also transform the 1.5–2 KΩ impedanceof the JFET devices to 50 Ω. As a result of theneeded impedance transformation, the filtersmust be designed with different Rin and Rout

characteristics, and that complicates the use ofloop-up tables (which would be permitted ifthe input and output resistances were equal).

Figure 7.11 shows an integrated circuitDBM based on MOSFET transistors. This de-

vice was first introduced as the SiliconixSi8901, but that company no longer makes it.Today, the device is made by CalogicCorporation (237 Whitney Place, Fremont, CA94539; phones 570-656-2900 [voice] and 570-651-3026 [fax]), as part number SD8901. TheSD8901 comes in a seven-pin metal can pack-age. The specifications data sheet for theSD8901 claims it provides as much as 10 dBmimprovement in the third-order interceptpoint over the U350 JFET design or the diodering DBM (discussed later).

A basic circuit for the SD8901 is shown inFigure 7.12. The input and output terminals areconnected to center-tapped, 4:1 impedance ra-tio RF transformers. Although these transform-ers can be homemade (using toroidal cores),the Mini-Circuits type T4-1 transformers wereused successfully in an amateur constructionproject. As in other DBM circuits, a diplexer isused at the output of the SD8901 circuit.

The local oscillator inputs are driven inpush-pull by fast rise time square waves. Thisrequirement can be met by generating a pairof complementary square waves from the


Fig. 7.11Commercial 8901 IC mixer.

5 7 3

1264

Q1

Q2

Q3

Q4

LO1 IF2 RF2

CASESUBSTRATE

LO2 RF1 IF1

same source. The circuit can use a pair ofhigh speed TTL J-K flip-flops connected withtheir clock inputs in parallel, driven from avariable frequency oscillator that operates attwice the required LO frequency. The com-plementarity requirement can be met by us-ing the Q-output of one J-K flip-flop and theNOT-Q output of its parallel twin.

The SD8901 device is capable of verygood performance, especially in the achiev-able dynamic range. In one design, a +35dBm third-order intercept point was achieved,along with a +16 dBm, 1 dB output compres-sion point and a 1-dB output blocking desen-sitization of +15 dBm. Insertion loss wasmeasured at 7 dB.

One problem with the SD8901 device isits general unavailability to amateur and hob-

byist builders. Although low in cost, Calogichas a minimum order quantity of 100, whichmakes it a little too rich for most hobbyists toconsider. A compromise that works well isMOS electronic switch IC devices, on themarket from several companies, includingCalogic. Figure 7.13 shows a typical MOSswitch, and it is easy to see how it can bewired into a circuit as in Figure 7.12. At leastone top-of-the-line radio transceiver uses aquad MOS switch for the DBM in the receiver.It would be interesting to see how well low-cost MOS switches, such as the CMOS 4066device, would work. I have seen that chipwork well as a double-balanced phase sensi-tive detector in medical blood pressure ampli-fiers, and those circuits are closely related tothe DBM.


Fig. 7.128901 IC mixer circuit.

OUT TODIPLEXER

U2J-K Flip-Flop(Divide by 2)

L.O. INPUT(F = 2 x LO)

RF

5 8

U18901Mixer

2

3

4

7

1

T1 T2

Fig. 7.13Compromise MOSFETswitch.

D1

IN OUT

SUBSTRATE

DOUBLE-BALANCED DIODE MIXER CIRCUITS

One of the most easily realized double-bal-anced circuits, whether homebrew or commer-cial, is the circuit of Figure 7.14. This circuituses a diode ring mixer and balanced input,output, and LO ports. It is capable of 30–60 dBof port-to-port isolation, yet is reasonably wellbehaved in practical circuits. DBMs such asFigure 7.14 have been used in a wide varietyof projects from direct conversion receivers, tosingle-sideband transmitters, to high-perfor-mance shortwave receivers. With proper de-sign, a single DBM can be made to operateover an extremely wide frequency range:Several models claim operation of 1–500 MHz,with IF outputs from DC to 500 MHz.

The diodes (D1–D4) can be ordinarysilicon VHF/UHF diodes such as 1N914 or1N4148. However, superior performance isexpected when Schottky hot carrier diodes,such as 1N5820 through 1N5822, are used

instead. Whatever diode is selected, all fourdevices should be matched. The best match-ing of silicon diodes is achieved by compar-ison on a curve tracer, but failing that thereshould at least be a matched forward/re-verse resistance reading. Schottky hot carrierdiodes can be matched by ensuring that theselected diodes have the same forward volt-age drop when biased to a forward currentof 5–10 mA.

Figure 7.15 shows the internal circuitryfor a very popular commercial diode DBMdevice, the Mini-Circuits SRA-1 and SBL-1 se-ries; a typical SRA/SBL package is shown inFigure 7.16. These devices offer good perfor-mance and are widely available. Some partshouses sell them at retail, as does Mini-Circuits (P.O. Box 166, Brooklyn, NY 11235,phone: 714-934-4500). I do not know theamount of the firm’s minimum order, butMini-Circuits has responded to $25 orders onseveral occasions, which certainly is morereasonable than other companies.


Fig. 7.14Double-balanced mixerwith diode ring mixerand balanced input, out-put, and LO ports.

LO

IF

RFT1 T2

D1D4

D3 D2

Fig. 7.15Mini-Circuits LabsDBM circuit.

LO

IF

RFT1

T2

D1D4

D3 D2

IF

The packages for SRA and SBL devicesare similar, being on the order of 20 mm longwith 5 mm pin spacing. The principal differ-ence between the packages for SRA and SBLdevices is in the height. In these packages,pin no. 1 is denoted by a blue bead insulatoraround the pin. Other pins are connected tothe case or have a green (or other color) beadinsulator. Also, the MCL logo on the top canbe used to locate pin no. 1: The M of the logois directly over pin no. 1. Table 7.1 shows the

characteristics of several DBMs in the SRAand SBL series, while Table 7.2 shows the pinassignments for the same devices.

In the standard series of devices, the RFinput can accommodate signals up to +1 dBm(1.26 mW), while the LO input must see a +7dBm (5 mW) signal level for proper opera-tion. Given the 50 Ω input impedance of allports of the SRA/SBL devices, the RF signallevel must be kept below 700 mV p-p, whilethe LO wants to see 1400 mV p-p. It is essen-tial that the LO level be maintained across the


Fig. 7.16Mini-Circuits Labs SRA/SBL package.

BLUE DOTDENOTES PIN NO. 1

1

2

3

4

5

6

7

8

5.08 mm0.2 in.

5.08 mm0.2 in.

BOTTOM VIEW

TOP VIEW

MCL

20.32 mm0.8 in.

Table 7.1 Characteristics of SRA/SBL Mixers

LO/RF IF Mid-Band Type No. (MHz) (MHz) Loss (dB)

SRA-1 0.5–500 DC–500 5.5–7.0

SRA-1TX 0.5–500 DC–500 5.5–7.0

SRA-1W 1–750 DC–750 5.5–7.5

SRA-1-1 0.1–500 DC–500 5.5–7.5

SRA-2 1–1000 0.5–500 5.5–7.5

SBL-1 1–500 DC–500 5.5–7.0

SBL-1X 7–1000 5–500 6.0–7.5

SBL-1Z 7–1000 DC–500 6.5–7.5

SBL-1-1 0.1–400 DC–400 5.5–7.0

SBL-3 0.025–200 DC–200 5.5–7.5

Table 7.2 SRA/SBL dBm Pin-Outs

Pin/Function

Type No. LO RF IF GND Case GND

SRA-1 8 1 3,4 2,5,6,7 2

SRA-1TX 8 1 3,4 2,5,6,7 2

SRA-1-1 8 1 3,4 2,5,6,7 2

SRA-1W 8 1 3,4 2,5,6,7 2,5,6,7

SRA-2 8 3,4 1 2,5,6,7 2,5,6,7

SRA-3 8 1 3,4 2,5,6,7 2

SBL-1 8 1 3,4 2,5,6,7 __

SBL-1-1 8 1 3,4 2,5,6,7 __

SBL-3 8 1 3,4 2,5,6,7 __

band of interest or else mixing operation suf-fers. Although the device works down to +5dBm, a great increase in spurious output andless port-to-port isolation is found. Spectrumanalyzer plots of the output signal at low LOdrive levels show considerable second- andthird-order distortion products.

Note the IF output of the SRA/SBL de-vices. Although some models in the series usea single IF output pin, most devices use twopins (3 and 4), and these must be connectedtogether externally for the device to work.

As with most DBMs and all diode ringDBM circuits, the SRA/SBL devices are sensi-tive to the load impedance at the IF output.Good mixing and freedom from the LO/RFfeedthrough problem occur when the mixerlooks into a low VSWR load. For this reason,a good diplexer circuit is required at the out-put. In experiments, I found that untermi-nated SBL-1-1 mixers produce nearly linearmixing when not properly terminated, andthis is not desired in a frequency converter.

Figure 7.17 shows a typical SRA/SBL cir-cuit: The RF drive (≤ +1 dBm) is applied topin no. 1 and the +7 dBm LO signal is appliedto pin no. 8. The IF signal output is throughpins 3 and 4, which are strapped together. Allother pins (2, 5, 6, and 7) are grounded.

The diplexer circuit consists of a high-pass filter (C1/L1) terminated in a 50 Ω dummyload for the unwanted frequencies and a low-pass filter (L2/C3) for the desired frequencies.

All capacitors and inductors are selected tohave a reactance of 50 Ω at the IF frequency.

Sometimes 1 dB resistor π-pad attenua-tors are used at the inputs and the IF outputof the DBM. In some cases, the input attenu-ators are needed to prevent overload of theDBM (overload causes generation of spuri-ous product frequencies and may destroy thedevice). In other cases, the circuit designerattempts to “swamp out” the effects of sourceor load impedance variations. Although thismethod works, it is better to design the cir-cuit to be insensitive to such fluctuationsthan to use a swamping attenuator. The rea-son is that the resistive attenuator causes asignal loss and adds to the noise generated inthe circuit (no resistor can be totally noisefree). A good alternative is to use a stableamplifier with 50 Ω input and output imped-ances, which is not itself sensitive to imped-ance variation, to isolate the DBM.

Mini-Circuits devices related to theSRA-1 and SBL-1 incorporate MAR-x seriesMMIC amplifiers internal to the DBM. Oneseries of devices places the amplifier in theLO circuit, so that much lower levels of LOsignal will provide proper mixing. Anotherseries places the amplifier in the IF outputport. This amplifier accomplishes two things:It makes up for the inherent loss of themixer and it provides greater freedom fromload variations that can affect the regularSRA/SBL devices.


Fig. 7.17 Mini-Circuits Labs SRA/SBL circuit.

DBMSRA-1SBL-1

TO POSTAMPLIFIER

L2C4

0.01 µF

C3C6

0.1 µF

L1XL = 50 OHMS

C1XC = 50 OHMS

R150 OHMS

LOINPUT

(+7 dBm)

RFINPUT

(<1 dBm)

C30.01 µF

C20.01 µF

3

4

87652

1

BIPOLAR TRANSCONDUCTANCECELL DBMs

Active mixers made from bipolar silicontransistors formed into Gilbert transconduc-tance cell circuits also are readily available.

Perhaps, the two most common devices arethe Signetics NE-602 device and the LM-1496device.

The LM-1496 device is shown in Figure7.18. Figure 7.18A shows the internal cir-cuitry, while Figures 7.18B and 7.18C show


Fig. 7.18 LM-1496: (A) circuit; (B) pin-outs for DIP; (C) pin-outs for 10-pin metal case.

V-

BIAS

-IN +IN

GAINADJUST

CARRIERINPUT

+OUT-OUT

1

7 8

14

LM-1496

+IN

-IN

GAINADJUST

+OUT

-OUT

N.C.

BIAS

CARRIERINPUT

N.C.

N.C.

V-

LM-1496

10

1

2

3

4

8

9

56

7

-IN

V-

BIAS

GAINADJUST

CARRIERINPUT

+OUT

+IN-OUT

A

B

C

the DIP and metal can packages, respec-tively. Pins 7 and 8 form the local oscillator(or “carrier” in communications terminol-ogy) input, while pins 1 and 4 form the RFinput. These push-pull inputs sometimes arelabeled high-level signal (7 and 8) and low-level signal (1 and 4) inputs. DC bias (pinno. 5) and gain adjustment (pins 2 and 3) areprovided as well.

Figure 7.19 shows the basic LM-1496mixer circuit in which the RF and carrier in-puts are connected in the single-ended con-figuration. The respective signals are appliedto the input pins through DC blocking capac-itors C1 and C2; the alternate pin inputs inboth cases are bypassed to ground throughcapacitors (C3 and C4).

The output network consists of a 9:1broadband RF transformer that combinesthe two outputs and reduces their imped-ance to 50 Ω. The primary of the trans-

former is resonated to the IF frequency bycapacitor C5.

Figure 7.20 shows a circuit that usesthe LM-1496 device to generate double side-band suppressed carrier signals. When fol-lowed by a 2.5–3 kHz bandpass filter, whichis offset from the IF frequency, this circuitalso will generate single sideband (SSB) sig-nals. In common practice, a crystal oscillatorwill generate the carrier signal (Vc), whilethe while audio stages produce the modulat-ing signal (Vm) from an audio oscillator ormicrophone input stage. I once saw a circuitthat is very similar to this one in a signalgenerator/test set used to service both ama-teur radio and marine HF-SSB radio trans-ceivers. It was the signal source to test thereceiver sections of the transceivers. The car-rier was set to 9 MHz, and both lower side-band (LSB) and upper sideband (USB) KVGcrystal filters were used to select the desired


Fig. 7.19 Single-ended LM-1496 mixer circuit.

U1LM-1496

R1510

R21K

R3100

C20.1 µF

C10.1 µF

RF

LOIF

V+

R4820

R5510

R61K

R710KC3

1 µF

C41 µF

R8100

C5

T19:1

BALUN

1

2 310

4

8

7 5

6

9

sideband. An alternate, and cheaper, schemeuses a single 9 MHz crystal filter but two dif-ferent crystals at frequencies either side ofthe crystal passband. One crystal would gen-erate the USB signal, while the other wouldgenerate the LSB signal.

For a single sideband model to be use-ful it has to be demodulated to recover theaudio modulation. The circuit of Figure 7.21does that job nicely using an NE-602 DBM asa product detector. The circuit operates at 455kHz but is easily adopted to other frequenciesby changing the L-C tank circuits. This type ofdetector works on CW, SSB, and DSB signals(all require a local oscillator injection signal)and produces the audio resultant from het-erodyning the local carrier signal against theSSB IF signal in the receiver. All SSB receiversuse some form of product detector at the endof the IF chain, and many use the LM-1496device in a circuit similar to Figure 7.21.

PREAMPLIFIERS AND POSTAMPLIFIERS

There often is justification for using ampli-fiers with DBM circuits. The inputs can bemade more sensitive with preamplifiers.When an amplifier is used following the out-put of a passive DBM (postamplifier), itmakes up for the 5–8 dB loss typical of pas-sive DBMs. In either case, the amplifier pro-vides a certain amount of isolation of theinput or output port of the DBM, which freesthe DBM from the effects of source or loadimpedance fluctuations. In these cases, theamplifier is said to act as a buffer amplifier.

Figure 7.22 shows two popular ampli-fiers. They can be used as either preampli-fiers or postamplifiers, because each has 50Ω input and output impedances. The circuitin Figure 7.22A, based on the 2N5109 RFtransistor, provides close to 20 dB of gainthroughout the HF portion of the spectrum.


Fig. 7.20 LM-1496 circuit with adjustments.

U1LM-1496

S1

10

1

4

14 5

12

2

3

6

8

R110K

R220K

NULLR310K

R451

R551

C31 µF

C10.1 µF

C20.1 µF

CARRIER

MODULATINGSIGNAL

R66.8K

R751

R851

R91K

C41 µF

R113.9K

R123.9K

R131K

OUTPUT

V+

V-

R101K

A small amount of stabilizing degenerativefeedback is provided by leaving part of theemitter resistance (R3) unbypassed whileproperly bypassing the remaining portion ofthe emitter resistance. Additional feedback

occurs because of the 4:1 impedance trans-former in the collector circuit of the transis-tor. This transformer can be homemade usingan FT-44-43 or FT-50-43 toroidal ferrite corebifilar wound with 7–10 turns of #26 AWG


Fig. 7.21 NE-602 circuit.

U1NE-602

1

2

T2455 kHz

T1455 kHz

3

7

6

455 kHz IFINPUT

4

5

C1 C2

C3

C1, C2, C3, C4, C7: 0.047 µF

C4 C7

C50.22 µF

C60.22 µF

R12200 Ohms

R22200 Ohms

AUDIOOUTPUT

Fig. 7.22 Popular amplifiers: (A) NPN transistor amplifier; (B) MAR-6 amplifier.

C10.1 µF

INPUT

T1

R1560

R23.3K

R31K

R410

R5100

C20.1 µF

C30.1 µF

C40.1 µF

C50.1 µF

+12VDC

R6150

R768

C60.1 µF

OUTPUT

Q12N5179

+5VDC

R191

RFC1(see text)

C1

C2

INPUT OUTPUT

U1MAR-6

C30.1 µF

1 3

24

A B

enameled wire (or its equivalent in othercountries).

The circuit in Figure 7.22B is based onthe Mini-Circuits MAR-x series of MMIC de-vices. These chips provide 13–26 dB gain, atgood noise figures, for frequencies from nearDC to 1000 MHz (or more in some models;e.g., 1500 or 2000 MHz). The MAR-1 deviceshown in the circuit diagram is capable of 15dB performance to 1000 MHz. The input andoutput capacitors can be disk ceramic typesup to about 100 MHz, but above that fre-quency “chip” capacitors should be used.Values of 0.01 µF should be used in the lowHF region (<10 MHz), 0.001 µF can be used

up to 100 MHz, and 100 pF above 100 MHz.The RF choke (RFC1) should be 2.5 mH in thelow HF region, 1 mH from about 10 MHz to30 MHz, 100 µH from 30 MHz to 100 MHz,and 10 µH above 100 MHz. These values arenot critical and given only as guidelines.While it might be a bit tricky to get a 1 mHRFC to operate well at 100 MHz, there is noreally hard boundary in these spectrum bands.

NOTE

1. Joseph J. Carr, “NE-602 Primer,” ElektorElectronics (January 1992).


The local oscillator is used in superhetero-dyne receivers to convert the incoming RFsignal to the IF. Similarly, in product detec-tors, local oscillators are used to convert theIF signal to audio. They come in three fla-vors: L-C variable frequency oscillators,crystal oscillators, and crystal frequencysynthesizers. In this chapter we will look atall three types.

The local oscillator must be at leastspectrally pure or the mixing will suffer. Ifthe receiver is of variable frequency, then italso must be agile (able to switch frequenciesin as little as microseconds) and the incre-ments of the frequency change must besmall. The frequency resolution usually is1–100 Hz below 30 MHz, with a few as lowas 0.001 Hz. Generally, at VHF/UHF frequen-cies the resolution can be 1000 Hz.

L-C VARIABLE FREQUENCYOSCILLATORS

Variable frequency oscillators (VFOs) are ra-dio frequency signal generators that can be

continuously tuned using an inductor con-nected to either an air variable capacitor,mica “trimmer” capacitor, or a voltage-tunedvariable capacitance diode (varactor). VFOsdiffer from crystal oscillators in that the fre-quency can be varied in the VFOs, while inthe crystal oscillators it is either fixed or vari-able over only a tiny region.

VFO circuits can be used as signalgenerators in test equipment, to controltransmitters, as the local oscillator in eithersuperheterodyne or direct conversion re-ceiver projects, or in any other applicationswhere a continuously variable source of RFenergy is needed.

RF OSCILLATOR BASICS

Both VFOs and crystal oscillators are part ofa class of circuits called feedback oscillators.Figure 8.1 shows the basic configuration ofthis type of circuit; it consists of an amplifierwith open-loop gain Avol and a feedback net-work (which usually is frequency selective)with a “gain” of β.

Chapter 8

Local Oscillator and FrequencySynthesizer Circuits

109

BARKHAUSEN’S CRITERIA

If two conditions, called Barkhausen’s crite-ria, are met, then the circuit will oscillate: (1)The loop-gain is unity or greater and (2) thefeedback signal arriving back at the input isphase shifted 360º (which is the same as 0º).For most practical circuits, with the 180º pro-vided by an inverting amplifier, an additional180º of phase shift must be provided by thefeedback network. By its nature, an inductor-capacitor (L-C) tuned circuit can provide this180º phase shift at only one frequency.

BASIC CATEGORIES OF VFO CIRCUIT

We consider three general categories offeedback RF oscillator: Armstrong oscillators,Hartley oscillators, and Colpitts oscillators(also, a subset of the Colpitts oscillators arethe Clapp oscillators). These basic configura-tions are shown in Figure 8.2. The circuit inFigure 8.2A is the Armstrong oscillator. It isidentified by the feedback link positioned asa secondary winding on the tuning coil.

The circuit in Figure 8.2B is the Hartleyoscillator. It is identified by the resonantfeedback network that contains a tapped in-ductor (which effectively forms an inductivevoltage divider).

The Colpitts and Clapp circuits (Figure8.2C) are identified by the feedback net-work that contains a tapped capacitancevoltage divider. In both cases, the feedbackvoltage divider is part of the resonant L-Ctuning network.

POWER SUPPLIES FOR VFO CIRCUITS

It always is a good idea to use only regulatedDC power supplies with VFO (or any oscilla-tor) circuits. In fact, most experts agree that asingle regulator serving the oscillator is best,because it is not affected by load changes inother circuits on the same side of the regula-tor. The reason for using only regulated DCis that most oscillators shift frequency a slightamount when the power supply voltageschange. Although not all of the oscillator cir-cuits in this chapter show the regulator, it is agood idea to use one anyway.

SOME BASIC CIRCUITCONFIGURATIONS

The basic circuits just discussed can be con-figured in any of several different ways. Inthis section we use one of three active de-


Fig. 8.1Feedback oscillator block diagram.

A1AVOL

FEEDBACKNETWORK

b

S

VF

VI

GAIN

FREQUENCY DETERMININGCOMPONENTS

vices as the amplifier portion: junction fieldeffect transistor (JFET), metal oxide field ef-fect transistor (MOSFET), or the NE-602/NE-612 RF balanced mixer integrated circuit. Thebasic JFET is the MPF-102 device, while theMOSFET is the 40673 device.

Input-Side Circuit Configurations

Figure 8.3 shows two input configurations,one each for MPF-102 and 40673. Figure 8.3Ashows the circuit for the JFET device. It con-sists of a gate resistor of 100 kΩ to ground anda diode (1N914, 1N4148, or equivalent). Inmany cases, you need a capacitor in the gatecircuit (C1), especially if a DC source orground is directly in the circuit. To preventloading the tuned circuit, it is customary tomake the coupling capacitor small comparedto the tuning capacitor: Typically values of2–10 pF are used for HF and MW VFO circuits.

The same circuit can be used on theMOSFET, but a DC bias circuit is needed for

the second gate, G2. The bias networkshown in Figure 8.3B (R2/R3) is set to biasG2 to about one third the V+ voltage. A by-pass/decoupling capacitor (C2) is used to setG2 to a low impedance for RF, while keepingit at the bias voltage for DC.

The diode in the input circuit per-plexes some people when they first see it inoscillator circuits. The function of the diodeis to clean the signal and make it closer to a low-harmonic sine wave (all non-sinewaves or distorted sine waves have har-monic content; by definition, a “pure” sinewave has no harmonics).

Figure 8.4A shows the sine wave outputsignal when the diode is connected. Notethat the waveform is a reasonably good sinewave. The waveform in Figure 8.4B is a dis-torted sine wave and has a higher amplitudethan in Figure 8.4A.

The circuit can be either parallel tuned(Figure 8.5A), in the case of Colpitts orHartley oscillators, or series tuned (Figure

Local Oscillator and Frequency Synthesizer Circuits 111

Fig. 8.2Oscillators: (A) Armstrong; (B) Hartley; (C) Colpitts.

C1A C1B

L1

A1

L1 C1L2

A1

L1

C1

A1

A

B

C

8.5B) in the case of Clapp oscillators. In theHartley oscillators, the inductor (L1) istapped. In any L-C resonant circuit, reso-nance is that point where the inductive reac-tance (XL) and capacitive reactance (XC) areequal. Because these elements cancel out,the impedance of such a circuit is resistive.

Generally, the C/L ratio should behigh, so it is common practice to select arelatively small inductance and match itwith a higher capacitance. Many of the os-

cillators in this section are designed for themiddle of the 1–10 MHz band and use in-ductance values on the order of 3.3–7 µH.These inductance values are relatively easyto obtain when using either “solenoid”(cylindrical) or “toroidal” coil forms. Bothfixed value and variable inductors can beused for these circuits.

The capacitance can be made up frommore than one capacitor, generally the bestpractice. The change of frequency is propor-


Fig. 8.3 Oscillator circuit with a diode to forward bias the transistor: (A) JFET; (B) MOSFET.

C1100 pF

TUNEDCIRCUIT

D11N4148

R1100K

GD

S

Q1MPF-102(or equiv.)

C1100 pF

TUNEDCIRCUIT

D11N4148

R1100K

G1

D

S

Q140673

(or equiv.)

G2

R233K

R3100K

+12VDC

C20.01 µF

A B

Fig. 8.4 Oscillator waveform: (A) with diode; (B) without diode.

A B

tional to the square root of the ratio of thecapacitance change:

(8.1)

That is why a variable capacitor for theAM broadcast band consists of a 25–365 pF airvariable capacitor shunted by a 25 pF (or so)trimmer capacitor (usually set to around 15pF). The 3.08:1 ratio of maximum to minimumcapacitance is more than sufficient to coverthe 3.02:1 ratio of the maximum and mini-mum frequencies of the AM broadcast band.

Selecting values of L and C is a some-what tedious and iterative affair. One is ad-vised to sit down with a calculator and makea few trials. Part of the problem occurs be-cause both fixed and variable capacitorscome in standard values that may or may notbe exactly what is needed. Juggle the induc-tance (which is easy to wind to a customvalue) to provide the desired frequencychange with the available variable capacitors.

It is common practice to make the totalof the fixed capacitors plus the maximumvalues of the variable (main tuning and trim-mer) capacitors somewhat larger than the to-

tal required to resonate at the very lowestfrequency in the desired range. When thetrimmer is set to a value less than maximum,the total capacitance will be close to the de-sired value.

Hartley JFET VFO

The Hartley oscillator, you may recall, is iden-tified by a feedback path that includes atapped inductor; the inductor also is part of theresonant tuning circuit of the oscillator. Figure8.6 shows a simple Hartley oscillator based onthe MPF-102 JFET. The output signal is takenthrough a small-value capacitor (to limit load-ing) connected to the emitter of the transistor.

The frequency of oscillation is set bythe combined effect of L1, C1, and C2:

(8.2)

To resonate at 5 MHz with a 5 µH in-ductor, a total capacitance of about 200 pFis required. Because of stray capacitancesand errors in the values of actual capaci-tors, it is common practice to use more totalcapacitance than needed and use variable

F =

L C +CHz

1

2 1 1 2π ( )( )

max

min

max

min

FF

= CC


Fig. 8.5 Circuits: (A) parallel tuned; (B) series tuned.

)321(12

1

CCCLFHz

++=

p

ASSUMES C4 << (C1+C2+C3)

L1C1

MAINC2

TRIM C3

C4

L1

C1MAIN

C2TRIM C3

C4

A B


Fig. 8.6Hartley oscillator circuit.

+12VDC

C1MAIN

L1

C310 pF

R2100K

C40.001 µF

C50.01 µF

OUTPUT

Q1MPF-102

C2TRIM

C60.1 µF

R11801-WD2

9.1V400-mW

D11N4148

Fig. 8.7 Voltage-tuned Hartley oscillator.

+12VDC

C60.1 µF

C710 µF

+

R4220

C50.01 µF

Q1MPF-102

C80.001 µF

OUT B

C40.01 µF

R3150

D11N4148

R2100KC3

10 pF

OUT A

C980 pF

TRIMMERL1

VT0-12V

C10.001 µF

D2NTE-618

(etc.)

R110K

C20.01 µF

5 t.

14 t. 8 t.

capacitors to trim. For example, we coulduse a 140 pF variable capacitor for the maintuning and an 80 pF trimmer to set the max-imum to the required value.

Figure 8.7 shows a 5 MHz Hartley VFOcircuit based on the MPF-102 JFET. It is verysimilar to the previous circuit in basic con-cept, but there are some differences. Themost significant difference is the use of avariable capacitance diode (varactor) insteadof the main tuning capacitor. The diodeshown here is a 14–440 pF varactor used totune the AM broadcast band in radio re-ceivers. Another significant difference is thatthe output signal is taken from a secondarywinding on the tuning inductor. This windingconsists of fewer turns than the lower por-tion of the main tuning inductor. Care mustbe taken to not load down this circuit byconnecting a varying load resistance acrossthe output winding.

The circuit of Figure 8.7 can be made tosweep the entire frequency range by applyinga sawtooth waveform that rises from 0 V to+12 V. This type of circuit makes it relativelyeasy to build a sweep generator circuit or aswept-tuned radio receiver. In general, thesweep rate should be around 40 Hz if the de-tector has a narrowband filter. Other sweepfrequencies are usable as well, but care mustbe taken not to “ring” any following resonantcircuits or filters by a too-fast sweep rate.

The tuning properties of a varactor L-Ccircuit are nonlinear because of the nature ofvaractor diodes. A graph of tuning voltageversus frequency is recommended and thatonly the linear portion of the curve be usedfor sweep purposes.

The circuit in Figure 8.8A is capable ofproducing as much as several volts (that isvolts). The actual tuning range depends onthe particular components used. The heartof the oscillator is an MPF-102 JFET (Q1).Two different tuning schemes are provided.For a limited range, the main tuning is pro-vided by a 365 pF AM broadcast band tuningcapacitor and the tuning range is 5000–5500kHz. In that case, the varactor diode (D2) isdisconnected and not used. The alternatescheme deletes C1 and uses either a 365 pF

variable or the varactor circuit (shown) atpoint A. This version has a rather wider tun-ing range for the same capacitance change.In the previous circuit, the tuning range wasreduced by the capacitor divider action ofC4 and C5.

The circuit of Figure 8.8A can be modi-fied by adding or subtracting capacitors fromthe tuning network. As shown, C1, C2, C3, C4,and C5 are all part of the tuning circuit. The126 pF fixed capacitor (needed for 5–5.5 MHzoperation) is made from parallel 82 and 47 pFcapacitors. A version of the circuit using theNTE-618 or ECG-618, 440 pF diode was builtwithout the mica trimmer capacitor (C2). Itproduced a voltage (Vt) versus frequencycharacteristic shown in Figure 8.8B (this curveis for a circuit with C2 removed). The varactoris comfortable over a range of 0 to +12 V (al-though my DC power supply goes down toonly +1.26 V, which is why the offset at 2.2MHz). The tuning range is 2.2–4.1 MHz, butthis can be lowered by increasing any of thecapacitors (e.g., C3) or by either eliminating orreducing the capacitors (C1–C5).

The main inductor (L1) consists of 35turns of #28 enameled wire on an AmidonT-50-2 (RED) toroidal core. The tap isplaced at eight turns from the ground end.The tap is formed by winding two separatebut contiguous windings: one of 8 turns andone of 27 turns. These two windings areconnected together electrically at the pointwhere they come together. The wire ends ofthe two coils are soldered and used as a sin-gle wire to connect to the source (S) of theJFET oscillator.

The drain (D) of the JFET is kept at aground potential for RF signals by the bypasscapacitor, C8. The output signal is taken fromthe source (S) terminal through a 10 pF capac-itor. This signal is fed to Gate 1 (G1) of the40673 dual-gate MOSFET used as an outputbuffer amplifier stage. This circuit produces a44 dB gain and is responsible for making thesignal voltage so large. The output transformer(T1) consists of an Amidon Associates FT-50-43 core wound with 20 turns of #28 enameledwire for the primary winding and 6 turns ofthe same wire for the secondary winding.


The output signal of Figure 8.8A ishuge as RF oscillators go, so it may have tobe attenuated in some cases. This job can bedone by connecting an attenuator resistorpad in series with the output signal or re-ducing the DC bias voltage on the MOSFETGate 2. This latter job can be done by reduc-ing the value of R4 until the desired output

signal is achieved (do not reduce it belowabout 1 KΩ).

Clapp VFO Circuit

The Colpitts and Clapp oscillators are verysimilar to each other in that both depend ona capacitor voltage divider (C4 and C5 in


Fig. 8.8 Hartley VFO: (A) circuit; (B) tuning voltage curve.

12

1.26

2.2 4.2F

FREQUENCY (MHz)

V

TU

NIN

G V

OLT

AG

E (

VT)

Q1MPF-102

U178L05

A

Q240673

C1365 pF

C582 pF

C234 pF

C3129 pF

C4100 pF

C63 pF

R1100K

D11N4148

C80.01 µF

C710 pF

R3100K

F.B.

R7100

C110.01 µF

C90.1 µF

R2100

R468K

R5100KC10

0.01 µF

R6100

C120.1 µF

C1310 µF

C140.01 µF

+

RF OUTPUT

T1

20 t.6 t.

+12 VDC

0.01 µF

D2440 pF

10K

VTA

B

Figure 8.9) for feedback. The difference inthe two oscillator circuits is that a Colpitts os-cillator uses a parallel resonant L-C circuit,while a Clapp oscillator uses a series reso-nant L-C circuit. The circuit in Figure 8.9 is aClapp oscillator because of the series tunedL-C network. This circuit can be used from0.5 to 7 MHz, even higher if C4 and C5 arereduced to about 100 pF.

The output circuit from this circuit istaken from the source (S) of the JFET oscilla-tor transistor. The output circuit includes anRF choke (RFC1) that builds the output am-plitude. Bias voltage for the JFET is providedby R2 and the source-drain current flowingthrough it.

NE-602 or NE-612 VFO Circuits

The NE-602AN and NE-612AN devices aredouble-balanced modulators and oscillatorintegrated circuits. Normally, they are usedas the RF front ends of radio receivers, but ifthe DBM is unbalanced by placing a 10 KΩresistor from the RF input (pin 1) to ground,it will function as an oscillator that producesabout 500 mV output signal.

Figure 8.10 shows an NE-602AN/NE-612AN Colpitts oscillator circuit. Three ca-pacitors (C1, C2, and C3) are used in thiscircuit, rather than two, because of a need forDC blocking. These capacitors should beequal to each other and have a value on theorder of 2400 pF/FMHz. The inductor shouldhave an approximate value of 7 µH/FMHz. Thetuning capacitor, C4, should have a valuethat will resonate with the selected inductor:

(8.3)

For example, a 5000 kHz (5 MHz) oscil-lator should have network capacitors of 2400pF/5 MHz, or 480 pF (use 470 pF standard-value capacitors). The inductor should be 1.4µH (17 turns on an Amidon T-50-2 (RED)core). To resonate with the 1.4 µH inductorrequires 723 pF. But 470 pF/2, or 236 pF, al-ready are in the circuit because of the seriesnetwork C2/C3; hence, a variable capacitor(C4) of 723 pF − 236 pF, or 487 pF.

An NE-602AN Hartley oscillator isshown in Figure 8.11. This circuit is identifiedby the tapped coil in the L-C network. The

C =f L

41

4 1π2 2


Fig. 8.9Series-tuned Clapp oscillator.

R1100K

C95 pF

C125 pF

C210 pF

C310 pF

D11N4148

C40.001 µF

C50.001 µF

Q1MPF-102

RFC11 mH

R2100

C60.01 µF

C733 pF

RFOUT

+9 VDC

C80.1 µF

value of the inductor is about 10 µH/FMHz,and is tapped from one fourth to one thirdthe way from the ground end. The capacitorneeds to resonate at the desired frequency.For our 5 MHz example, an inductor of 2 µH

is used, which means 20 turns of wire on aT-50-2 (RED) core.

A voltage-tuned Clapp NE-602AN oscilla-tor is shown in Figure 8.12. This circuit uses avaractor diode to set the operating frequency.


Fig. 8.10NE-602AN/NE-612ANColpitts oscillator.

R1100

IC1NE-602

+5VDC

C70.01 µF

R210K

C50.01 µF

OUT 1

OUT 2

1

2

8

3 6

7

4

5

L1 C4

C1

C2

C3

Fig. 8.11NE-602AN Hartley oscillator.

R1100

IC1NE-602

+5VDC

C60.01 µF

R210K

C40.01 µF

OUT 1

OUT 2

1

2

8

3 6

7

4

5

L1C1

C20.01 µF

C30.01 µF

With the 100 pF capacitors shown, this circuithas oscillated from about 6 MHz to about 15MHz, using an NTE-614 (33 pF) diode.

CRYSTAL OSCILLATORS

Radio frequency oscillators can be built usinga number of different types of frequency se-lective resonator. Common types include in-ductor-capacitor (L-C) networks and quartzcrystal resonators. The crystal resonator has,by far, the best accuracy and stability, and itmakes single-frequency receivers possible.

Piezoelectric Crystals

Certain naturally occurring and human-madematerials exhibit the property of piezoelec-tricity: Rochelle salts, quartz, and tourmalineare examples. Rochelle salts crystals are notused for RF oscillators, although at one timethey were used extensively for phonographpick-up cartridges. Tourmaline crystals canbe used for some RF applications but are not

often used due to high cost. Tourmaline isconsidered a semiprecious stone, so tourma-line crystals are more likely to wind up asgemstones in jewelry than in radio circuits.That leaves quartz as the preferred materialfor radio crystals.

Figure 8.13 shows a typical naturalquartz crystal. Actual crystals rarely have allthe planes and facets shown. The three opti-cal axes (X, Y, and Z) in the crystal are usedto establish the geometry and locations ofvarious cuts. The actual crystal segmentsused in RF circuits are sliced out of the maincrystal. Some slices are taken along the opti-cal axes, so are called Y-cut, X-cut, and Z-cutslabs. Others are taken from various sectionsand are given letter designations such as BT,BC, FT, AT, and so forth.

Piezoelectricity

All materials contain electrons and protons,but in most materials their alignment is ran-dom. This produces a net electrical potentialin any one direction of zero. But, in crystalline


Fig. 8.12Voltage-tuned NE-602ANoscillator.

R1100

IC1NE-602

+5VDC

C50.01 µF

R210K

C40.01 µF

OUT 1

OUT 2

1

2

8

3 6

7

4

5

L1R310K

C30.01 µF

C1100 pF

C2100 pF

D1

VT

materials, the atoms are lined up, so they canform electrical potentials. Piezoelectricity refersto the generation of electrical potentials due tomechanical deformation of the crystal.

Figure 8.14 shows the piezoelectric ef-fect. A zero-center voltmeter is connectedacross a crystal slab. In Figure 8.14A, the slabis at rest, so the potential across the surfacesis zero. But in Figure 8.14B, the crystal slab isdeformed in the upward direction, and apositive potential is seen across the slab.When the crystal slab is deformed in the op-posite direction, a negative voltage is noted.

If the crystal is mechanically “pinged”once, it will vibrate back and forth, produc-ing an oscillating potential across its termi-nals, at its resonant frequency. Due to losses,the oscillation will die out in short order. But,

if the crystal is repetitively pinged, then itwill generate a sustained oscillation on itsresonant frequency.

It is not terribly practical to stand therewith a tiny little hammer pinging the crystal allthe while the oscillator is running, however.Fortunately, piezoelectricity also works in thereverse mode: If an electrical potential is ap-plied across the slab it will deform. Therefore,if we amplify the output of the crystal andthen feed back some of the amplified outputto electrically “re-ping” the crystal, it will sus-tain oscillation on its resonant frequency.

EQUIVALENT CIRCUIT

Figure 8.15A shows the equivalent R-L-C cir-cuit of a crystal resonator, and Figure 8.15Bshows the impedance vs. frequency plot for


Fig. 8.13Crystal structure.

Y

Y

X

X

Z

Z

GTJT

BT

BC

FT

DT

ET

CT

AC

NT

MT

ATCUT

XCUT

YCUT

ZEROTEMPERATURECOEFFICIENT

FILTERS

HARMONICOSCILLATORS

LOWFREQUENCY

OSCILLATORS

HIGHFREQUENCY

OSCILLATORS

the crystal. The equivalent circuit has four ba-sic components: series inductance (Ls), seriesresistance (Rs), series capacitance (Cs), andparallel capacitance (Cp). Because there aretwo capacitances, there are two resonances:series and parallel. The series resonancepoint is where the impedance curve crossesthe zero line, while parallel resonance occursa bit higher on the curve.

CRYSTAL PACKAGING

Over the years a number of different pack-ages have been used for crystals. Even todaythere are different styles. Figure 8.16A showsa representation of the largest class of pack-ages. It is a hermetically sealed small metal

package, in various sizes. The actual quartzcrystal slab is mounted on support struts in-side the package (Figure 8.16B), which are inturn mounted to either a wire header or pins.

Some crystals use pins for the electricalconnections, typically mounted in sockets.The pin type of package can be soldered di-rectly to a printed circuit board, but care mustbe taken to prevent fracturing the crystal withheat. Not all pins are easily soldered, al-though it can help if the pins are scraped toreveal fresh metal before soldering. Normally,however, if the crystal is soldered into the cir-cuit, a wire-lead package is used.

Some crystals may short circuit if in-stalled on a printed circuit board with either


Fig. 8.14Crystals generate voltage when deflected.

0

0

0

A

B

C

through- (via-) holes or a ground plane onthe top side of the board. In those cases, theusual practice is to insert a thin insulator be-tween the PCB and the crystal (Figure 8.16C).

Temperature Performance

According to temperature performance, thereare three basic categories of crystal oscillator:room temperature crystal oscillators (RTXO),temperature-compensated crystal oscillators(TCXO), and oven-controlled crystal oscil-lators (OCXO). We look at each of thesegroups.

ROOM TEMPERATURE CRYSTAL OSCILLATORS

The RTXO takes no special precautionsabout frequency drift. But, with proper selec-tion of crystal cut and reasonable attention toconstruction, stability on the order of 2.5parts per million (ppm), 2.5 × 10–6, over the

temperature range 0–50°C is possible. TheRTXO is used only on economy model coun-ters used for noncritical applications.

TEMPERATURE-COMPENSATED CRYSTAL OSCILLATORS

The TCXO circuit also works over the 0–50°Ctemperature range but is designed for muchbetter stability. The temperature coefficientsof certain components of the TCXO are de-signed to counter the drift of the crystal, sothe overall stability is improved to 0.5 ppm (5 × 10–7). The cost of TCXOs has decreasedmarkedly over the years to the point whererelatively low-cost upgrades to economy re-ceivers give them a rather respectable stabil-ity specification.

OVEN-CONTROLLED CRYSTAL OSCILLATORS

The best stability is achieved from the OCXOtime base. These oscillators place the res-onating crystal inside a heated oven that


Fig. 8.15 Crystal: (A) equivalent circuit; (B) impedance diagram.

XL

XC

IND

UC

TIV

E R

EA

CTA

NC

EC

APA

CIT

IVE

RE

AC

TAN

CE

FS

FREQUENCY

ANTIRESONANCE

PARALLELRESONANCE

RANGE

ESR

LS CS

CP

RS

A

B

keeps its operating temperature constant,usually near 70 or 80°C.

Two forms of crystal oven are used inOCXO designs: on/off and proportional con-trol. The on/off type is similar to the simplefurnace control in houses. It has a snap ac-tion that turns the oven heater on when theinternal chamber temperature drops below acertain minimum point and off when it rises

to a certain maximum point. The propor-tional control type operates the heating cir-cuit continuously and supplies an amount ofheating proportional to the actual tempera-ture difference between the chamber and theset point. The on/off form of oven is capableof 0.1 ppm (10–7).

OCXOs that use a proportional controloven can reach a stability of 0.0002 ppm


Fig. 8.16 Crystal package: (A) external view; (B) internal view; (C) use of insulator on double-sidedprinted wiring board.

METALCASE

QUARTZCRYSTAL

ELECTRODE

SUPPORT STRUT

PINS

INSULATOR

SUPPORT STRUT

PCB

INSULATOR

CRYSTAL

A

B

C

(2 × 10–10) with a 20-min warm-up and 0.0001ppm (1.4 × 10–10) after 24 hours. It is commonpractice to design the counter to leave theOCTX turned on even when the counter is off.Some portable frequency counters, such asthose used in two-way radio servicing, have abattery back-up to keep the OCXO turned onwhile the counter is in transit.

This variation is referred to as the tem-perature stability of the counter time base. Wealso must consider short-term stability andlong-term stability (aging).

Short-Term Stability

The short-term stability is the random fre-quency and phase variation due to noise thatoccurs in any oscillator circuit. It sometimesis called time domain stability or fractionalfrequency deviation. In practice, the short-term stability has to be a type of rms (rootmean square) value averaged over 1 sec. Theshort-term stability measure is given asσ(∆f/f )(t). Typical values of short-term stabil-ity for the different forms of clock oscillatorare listed in Table 8.1.

Long-Term Stability

The long-term stability of the time base clockoscillator is due largely to crystal aging. Thenature of the crystal, the quality of the crys-tal, and the plane from which the particularresonator was cut from the original quartzcrystal are determining factors in defining ag-ing. This figure is usually given in terms of

frequency units per month, as shown inTable 8.2.

Oscillators

The amplifier used in an oscillator circuit canbe any of many different devices. In somecircuits it is a common-emitter bipolar tran-sistor (NPN or PNP devices). In others, it isJFET or MOSFET. In older equipment, it wasa vacuum tube. In modern circuits, the activedevice probably is either an integrated circuitoperational amplifier or some other form oflinear IC amplifier.

The amplifier most frequently is an in-verting type, so the output is out of phasewith the input by 180º. As a result, to obtainthe required 360º phase shift, an additionalphase shift of 180º must be provided in thefeedback network at the frequency of oscilla-tion only. If the network is designed to pro-duce this phase shift at only one frequency,then the oscillator produces a sine wave out-put on that frequency.

In a crystal oscillator, amplified noise inthe circuit at start-up initiates the crystal os-cillation, but the feedback voltage is used tocontinuously re-ping the crystal to keep itoscillating.

Colpitts Crystal Oscillator Circuit

Figure 8.17 shows a basic Colpitts oscillatorcircuit. The active element is an NPN bipolartransistor, although FET and IC versions willbe shown in circuits to be discussed later.


Table 8.2 Long-Term Stability for the DifferentForms of Clock Oscillator

Units per Parts per Oscillator Month Million

RTXO 3 × 10–7/month 0.3 ppm

TCXO 1 × 10–7/month 0.1 ppm

OCXO 1 × 10–7/month 0.1 ppm(on/off)

OCXO 1.5 × 10–8/month 0.015 ppm(proportional control) 5 × 10–10/day 0.0005 ppm

Table 8.1 Short-Term Stability for theDifferent Forms of Clock Oscillator

Root Mean Parts per Oscillator Square Million

RTXO 2 × 10–9 rms 0.002 ppm

TCXO 1 × 10–9 rms 0.001 ppm

OCXO 5 × 10–10 rms 0.0005 ppm(on/off)

OCXO 1 × 10–11 rms 0.00001 ppm(proportional control)

The transistor’s DC bias is derived from resis-tor R1 connected between V+ and the tran-sistor base terminal. The crystal resonator isconnected from the base to ground on thiscommon emitter circuit. This circuit is usablefrom about 1–18 MHz or so.

Because it is a Colpitts oscillator, thecircuit uses a capacitive voltage divider (C1and C2) for the feedback network. Whenthis circuit initially is turned on, current be-gins to flow from collector to emitter.When this current first flows, the crystal iselectrically pinged and begins to oscillate.Following initial start-up, a sample of thesignal at the emitter is fed to C1/C2 andfrom there to the base. Because the emitterAC signal voltage is in phase with the basesignal voltage, the Barkhausen require-ments are met.

Some experimentation will yield theoptimum values of C1, C2, and R2 to ensureproper starting and running. The generalrule for R1 is to use the lowest value that willpermit sure starting when the circuit is pow-ered up. Two approaches to finding valuesfor C1 and C2 are presented in the literature,but both assume that the total capacitive re-actance of the two capacitors in series is 300Ω. In one scheme, the initial trial values re-

quire C1 = C2, while in other recommenda-tions C2 = 3 × C1 or 4 × C1.

Miller Oscillators

Miller oscillators are analogous to the tunedinput/tuned output variable frequency oscil-lator, because they have a crystal at the inputof the active device and an L-C tuned circuitat the output. Figure 8.18 shows a basic Millercircuit built with a JFET. Any common RF de-vice can be used for Q1 (e.g., the MPF-102).DC bias is provided by R2, which places thesource terminal at a potential above grounddue to the channel current flowing in Q1. Thesource must be kept at ground potential forAC, so a bypass capacitor (C4) is provided.The reactance of this capacitor must be lessthan one tenth the value of R2 at the lowestintended frequency of operation.

The output circuit of the oscillator istuned by a parallel resonant L-C tank circuit,L1/C1. The tuned circuit must be adjusted tothe resonant frequency of the oscillator, al-though best performance usually occurs at afrequency slightly removed from the crystal


Fig. 8.17 NPN Colpitts crystal oscillator.

V+

C1

C2

R1220K

R21000

C3100 pF

C40.01 µF

OUTPUT

Q1

Y1

Fig. 8.18 JFET Miller crystal oscillator.

Y1R1

100K

Q1

V+

C30.01 µF

L1C1

C20.001 µF

C40.01 µF

R2220

frequency. If you monitor the output signallevel while adjusting either C1 or L1, you willnote a distinct difference between the highside and the low side of the crystal frequency.Best operation usually occurs at the low side.Whichever is selected, however, care must betaken that the oscillator will start reliablywhen cold started. Output signal can betaken either from capacitor C2, as shown, orthrough a link coupling winding on L1.

The Miller oscillator circuit of Figure 8.18has the advantage of being simple to build butsuffers from some problems. One problem isthat the feedback is highly variable from onetransistor to the next because it is created bythe gate-drain capacitance of Q1. Also output-level variations have been noted, as well asfrequency pulling, under output load imped-ance variations, which are not good attributesfor an oscillator. Further, JFETs of the sametype number and different crystals of the sametype number from the same manufacturer can

show a large difference in starting ability. Ialso noted problems with this circuit when ei-ther the JFET or crystal ages. I have seen JFEToscillators that worked well and then failed.When the JFET was replaced, it started work-ing again. What surprised me was that theJFET tested as good.

An improved Miller oscillator is shownin Figure 8.19. This circuit uses a dual-gateMOSFET transistor, such as the 40673 device,as the active element. It is a fundamentalmode oscillator that uses the parallel reso-nant frequency of the crystal. The crystal cir-cuit is connected to Gate 1, while Gate 2 isbiased to a DC level. This circuit can providea stability of 15–20 ppm if AT-cut or BT-cutcrystals are used.

A problem that might be seen with thiscircuit is parasitic oscillation at VHF frequen-cies. The MOSFETs used typically have sub-stantial gain at VHF, so could oscillate at anyfrequency where Barkhausen’s criteria are


Fig. 8.19 MOSFET Miller oscillator.

Y1C1

50 pFD1

1N4148R147K

R2470

C22200 pF

C3330

Q140673

R3100K

R447K

C40.01 µF

C5 C6

C7

OUTPUT

+12VDC

T1

G1

G2

S

D

RS10 to 47 W

met. There are two approaches to solvingthis problem. One approach is to insert aferrite bead on the lead of Gate 1 of theMOSFET. The ferrite bead acts like aVHF/UHF RF choke. The second approachis to insert a snubber resistor (RS in Figure8.19) between the crystal and Gate 1 of theMOSFET. Usually, some value between 10and 47 Ω will provide the necessary protec-tion. Use the highest value that permits surestarting of the oscillator.

One interesting aspect of the Miller os-cillator of Figure 8.19 is that it can be used asa frequency multiplier (not to be confusedwith an overtone oscillator), if the tuned net-work in the drain circuit of Q1 is tuned to aninteger multiple of the crystal frequency.

Pierce Oscillators

The Pierce oscillator is characterized by thecrystal being connected between the outputand input of the active device. Figure 8.20shows the basic Pierce crystal oscillator cir-cuit using a bipolar NPN transistor (e.g.,2N2222 or 2N5179). The crystal is connecteddirectly from the collector to the base of Q1.Output is taken through capacitor C2 con-

nected to the collector. This circuit is usedextensively in low-cost receiver circuits but isnot recommended.

An improved Pierce oscillator is shownin Figure 8.21. This circuit includes a capaci-tor (C1) for pulling the crystal a small amountto precisely tune the frequency. This circuit isdesigned for frequencies between 10 and 20


Fig. 8.20 Pierce oscillator.

V+

C1

Y1

R1

C2

Q1

C3

OUTPUT

Fig. 8.21Pierce oscillator for frequencies be-tween 10 and 20 MHz.

Y115 MHz

C4100 pF

OUTPUT

Q1

+5 to +12VDC

C147 pF

C2180 pF

C339 pF

R21K

R156K

C50.01 µF

MHz with the capacitance values shown. Ifthe output is lightly loaded (keep C4 small),then the oscillator will provide a reasonableoutput stability at a level of near 0 dBm.Figure 8.22 is a variation on the theme thatwill work in the 50–500 kHz region. This cir-cuit is very similar to the earlier circuit exceptfor increased capacitance values to accountfor the lower frequency. In both circuits, or-dinary NPN devices such as the 2N2222 canbe used successfully.

Butler Oscillators

The Butler oscillator looks superficially likethe Colpitts in some manifestations (Figure8.23). The difference is that the crystal is con-nected between the tap on the feedback net-work and the emitter of the transistor. Thisparticular circuit is a series mode oscillator.The value of R1 should be whatever valuebetween 100 and 1000 Ω will result in reli-able oscillation and starting while minimizingcrystal dissipation. A table of capacitancevalues for feedback network C1/C2 is pro-vided. For the 3–10 MHz range, use C1 = 47pF and C2 = 390 pF; for 10–20 MHz, selectC1 = 22 pF and C2 = 220 pF.


Fig. 8.22Pierce oscillator for fre-quencies in the 50–500kHz region.

V+

Y1

C160 pF

C2100 pFC3

3900 pF

OUTPUT

C43900pF

C50.1 µF

C60.1 µF

R110K R2

100

R33.3K

R42.2K

R5120K

Fig. 8.23 Butler NPN transistor oscillator.

V+

C2*

C1*

R25.6K

C40.01 µF

OUTPUT

Q12N5770

Y1R1330

C30.01 µF

R315K

L1

FREQ (MHz) C1 (pF) C1 (pF)

3 - 10

10 - 20

47 390

22 220

The collector circuit is tuned by thecombination of C1 and L1. This circuit maywell oscillate with the crystal shorted, andcare must be taken to ensure that the “free”oscillation and the crystal oscillation frequen-cies are the same. The crystal should takeover oscillation when it is in the circuit.

The Butler oscillator of Figure 8.23 iscapable of 10–20 ppm stability if a buffer am-plifier with good isolation is provided at theoutput. Otherwise, some frequency pullingwith load variations might be noted.

The output signal is taken from a cou-pling winding over L1. This winding typicallyis only a few turns of wire on one end of L1.Alternatively, a tap on L1 might be providedand connected to a low-value capacitor. Thisapproach might change some resonances un-less care is taken. Another alternative outputscheme is to connect a small value capacitorto the collector of Q1. Keep the value low toreduce loading and also the effects of theoutput capacitor on the resonance of L1/C1.

A somewhat more complex Butler oscil-lator is shown in Figure 8.24. This circuit issometimes called an aperiodic oscillator cir-cuit. It uses two additional transistors to pro-vide buffering and also as part of the feedback

circuit. This circuit operates in the frequencyrange from about 300 kHz to 10 MHz, al-though some care must be taken in the selec-tion of the transistor.

Many low-frequency crystals exhibit alower equivalent series resistance in one ofthe higher frequency modes of oscillationthan in the fundamental mode, so you mightfind this circuit oscillating at some frequencyin the medium wave or HF region, rather thanthe LF. The key to preventing this problem isto use a transistor with a lower gain-band-width product (e.g., 2N3565 or equivalent).

At this point it might be wise to point anaspect of “universal” replacement lines oftransistors. Because crystal oscillators may op-erate in an unwanted overtone mode (i.e., at ahigher frequency) or due to stray L-C compo-nents they may parasitically oscillate on a VHFor UHF frequency, you will want to keep thegain-bandwidth (GBW) product of the activedevice low. But many replacement lines use asingle high-frequency transistor with similargain, collector current, and power dissipationratings as a “one size fits all” replacement fortransistors with lower GBW products.

I have seen this happen with service re-placements on older equipment. The original


Fig. 8.24 Butler oscillator with an aperiodic oscillator circuit.

Y1

C60.001 µF

OUTPUT

V+

C150 pF

C215 pF

C30.01 µF

C40.1 µF

C50.1 µF

C70.01 µF

R110K

R227K

R36.8K

R422K

R51.5K

R61.5K

R712K

R812K

R93.3K

component may not be available, so a univer-sal, service shop replacement line device is se-lected. Then that parasitic oscillations andother problems are discovered because thenew replacement has a GBW of 200 MHz,where the old device was a 50 MHz transistor.This problem can show up especially severelyin RF amplifiers and low-frequency oscillatorswhere L-C components naturally exist or anycircuit where the stray and distributed L-C ele-ments provide the required phase shift onsome frequency above the unity GBW point.

The circuit of Figure 8.24 produces asine wave output but not without relativelystrong harmonic output. The second andthird harmonics are particularly evident.However, if harmonics are desired (as whenthe oscillator is used in a frequency multi-plier circuit), then strong harmonics up to 30

MHz can be generated from a 100 kHz crys-tal if R5 is reduced to about 1000 Ω.

The output of this oscillator is takenthrough an emitter follower buffer stage. Thiscircuit can be used as a general buffer for anumber of oscillator circuits. It generally is agood practice to use a buffer amplifier withany oscillator to reduce loading and smoothout load impedance variations.

Another variation on the Butler theme isshown in Figure 8.25. This circuit is similar toFigure 8.24 but a bit less sensitive to frequencypulling due to DC power supply voltage varia-tions. However, it is good engineering practiceto use a separate voltage regulator for all oscil-lator circuits to prevent such variation. Theavailability of low-cost, three-terminal inte-grated circuit voltage regulators makes preven-tion quite easy.


Fig. 8.25 Butler oscillator that is a bit less sensitive to frequency pulling.

Y1

+12VDC

OUTPUT

Q1

Q2

Q3

R13.9K

R2470

R31K

R427K

R51K

R61K

R7470

C150 pF

C233 pF

C30.1 µF

C40.1 µF

C50.001 µF

C60.01 µF

An improved Butler oscillator is shownin Figure 8.26. Again, this circuit is based onFigure 8.24. Both circuits can be used at fre-quencies from LF up to the mid-HF region(about 12–15 MHz) if appropriate values of R3and R5 are used. The improvement of Figure8.27 over Figure 8.24 is the limiting diodes(D1 and D2) provided between the two oscil-lator transistors (Q1 and Q2). These diodescan be ordinary 1N4148 small-signal diodes.The circuit of Figure 8.26 is more stable thanFigure 8.24 because crystal dissipation is lim-ited and it has more sure cold starting.

The Butler oscillators are series modecircuits but, because of the series capacitors,can use parallel mode crystals. For a strictlyseries mode circuit, eliminate the capacitorsin series with the crystal and replace it with ashort circuit.

Colpitts Oscillators

The Colpitts oscillator is characterized by afeedback network consisting of a tapped ca-pacitive voltage divider. In Figure 8.27, thefeedback is provided by C1 and C2, althoughthe situation is somewhat modified by the

gate capacitances of Q1. This circuit can beused with parallel-mode crystals from about 3to 20 MHz with proper values of C1 and C2(see table provided in Figure 8.27). Frequencytrimming of the oscillator can be done byshunting a small value trimmer capacitoracross the crystal. The trimmer also can beplaced in series with the crystal.

If the oscillator tends to oscillate parasit-ically in the VHF region, try using the snubberresistor method (R4 in Figure 8.27). This couldoccur because the JFET used at Q1 has suffi-cient gain at VHF to permit Barkhausen’s cri-teria to operate at some frequency wherestrays and distributed L-C elements producethe correct phase shift. A value between 10and 47 Ω usually eliminates the problem.Alternatively, a small ferrite bead can beslipped over the gate terminal of Q1 to act asa small-value VHF/UHF RF choke.

Figure 8.28 is very similar to Figure 8.27,except for two features. First, the active deviceis an n-channel MOSFET rather than a JFET.Any of the single-gate devices (e.g., 3N128)can be used. One must, however, be mindfulof the possibility of electrostatic damage whenthe MOSFET is used.


Fig. 8.26 Butler oscillator with limiting diodes between the two oscillator transistors.

Y1

C60.001 µF

OUTPUT

V+

C150 pF

C215 pF

C30.01 µF

C40.1 µF

C50.1 µF

C70.01 µF

R110K

R227K

R36.8K

R422K

R51.5K

R61.5K

R712K

R812K

R93.3K

C80.22 µF

D1D2

Q1

Q2

Q3

R10100


Fig. 8.28MOSFET Colpitts oscillator.

V+

C1*

C2*

R11 MEG

R21K

C30.001 µF

C4

OUTPUT

Q13N128

Y1

D11N4148

C550 pF

R3150


3 - 10 22 180

10 - 20 10 82

Fig. 8.27JFET Colpitts oscillator.

V+

C1*

C2*

R147K

R21K

C30.001 µF

C4

OUTPUT

Q1MPF-102

Y1C5

50 pF

R3150


3 - 10 27 68

10 - 20 10 27

R4(See Text)

The other difference is that a 1N4148small-signal diode is used shunting the gate-source path to provide a small amount of auto-matic gain control. When the signal appearingacross the crystal and feedback network is suf-ficiently large, the diode rectifies the signal andproduces a DC bias on the gate that countersthe source bias provided by R2. This diodehelps smooth out amplitude variations, espe-cially when more than one crystal is switchedin and out of the circuit.

Another variation on the Colpitts themeis the impedance inverting oscillator circuit ofFigure 8.29. It will provide stability of 10 ppmover a wide temperature range (0–60°C) if awise selection of components is made (C1–C3and L1 are particularly troublesome). The cir-cuit will also remain within ±0.001% over aDC power supply variation of 2:1 (provided

the crystal dissipation is not exceeded).Harmonic output of the circuit typically is low.

The oscillating frequency is set by ad-justing inductor L1. The turn counts shown inthe table in Figure 8.29 presume a 6.5-mmslug-tuned coil form designed for use in thefrequency range 3–20 MHz. Some experimen-tation is needed, depending on the particularform used. The idea is to set the resonant fre-quency of the coil and C1–C3 combined tosomething near the crystal frequency.

Sometimes, you want to add a tunedcircuit to the output circuit of oscillators. Theharmonics of the oscillator are suppressedwhen this is done. But, in this case, a transis-tor equivalent of the old-fashioned TGTP os-cillator will result, because of the action ofthe output tuned circuit and the L1/C1–C3combination. Do not do it.


Fig. 8.29 Tunable NPN Colpitts crystal oscillator.

V+

C2*

C3*

R26.8K

R1*

R315K

C547 pF

C40.01 µF

OUTPUT

Q12N5770

C60.01 µF

Y1

L1*

C1*

FREQ (MHz) C1 (pF) C2 (pF) C3 (pF)

10 - 15 100 220

15 - 20 100 100 680

220

100

R1

3 - 10 1000 270 270

680

1.5K

15 turns

10 turns

L1 2 - 4 MHz

6 - 10 MHz

4 - 6 MHz

R4560

60 turns

40 turns

25 turns

Overtone Oscillators

Thus far only the fundamental oscillatingmode has been discussed. But crystals oscil-late at more than one frequency. The oscil-lations of a crystal slab are in the form ofbulk acoustic waves and can occur at anywave frequency that produces an odd half-wavelength of the crystal’s physical dimen-sions (e.g., λ/2, 3λ/2, 5λ/2, 7λ/2, 9λ/2,where the fundamental mode is λ/2). Notethat these frequencies are not harmonics ofthe fundamental mode but actually valid os-cillation modes for the crystal slab. The fre-quencies fall close to but not directly onsome of the harmonics of the fundamentalmode (which probably accounts for the con-fusion). The overtone frequency will bemarked on the crystal rather than the funda-mental mode (it is rare to find fundamentalmode crystals above 20 MHz or so, becausetheir thinness makes them more likely tofracture at low values of power dissipation).

The problem to solve in an overtone os-cillator is encouraging oscillation on the correct

overtone while squelching oscillations at thefundamental mode and undesired overtones.Crystal manufacturers can help with correctmethods, but the oscillator designer still bearsresponsibility for solving this problem. Figure8.30 shows a third-overtone Butler oscillatorthat will operate at frequencies between 15 and65 MHz. The inductor (L1) is set to resonateclose to the crystal frequency and is used inpart to ensure overtone mode oscillation. Ifmoderate DC supply voltages are used (e.g.,9–12 V in most cases), the harmonic content islow (−40 dB) and stability is at least as good asa similar fundamental mode Butler oscillator.

Figure 8.31 is a third-overtone imped-ance inverting Colpitts-style oscillator thatwill operate over the 15–65 MHz range. As insimilar circuits, inductor L1 is tuned to theovertone and resonated with C1 (combinedwith the capacitances of C2 and C3). Valuesfor C1 through C3 and winding instructionsfor a 6.5-mm low-band VHF coil former areshown in the table.

Note the resistor across crystal Y1.This resistor tends to snub out oscillations


Fig. 8.30 Third-overtone Butler oscillator.

+12VDC

R115K

R24.7K

C130 pF

C222 pF

C3180 pF

C4100 pF

C50.001 µF

L1

RFC156 µH

C60.01 µF

R3220

Y1

OUTPUT

C70.001 µF

in modes other than the overtone, includ-ing the fundamental mode. Care must betaken to not make L1 too large, otherwise itwill resonate at a lower frequency (withC1–C3), forming an oscillator on a fre-quency not related to either the crystal’sfundamental mode or overtones. The oscil-lator may perfectly well act as a series-tuned Clapp oscillator. Operation of thecircuit of Figure 8.31 to 110-MHz, with fifth-or seventh-overtone crystals, can be accom-plished by modifying this circuit to the formshown in Figure 8.32.

TACTICS TO IMPROVE OSCILLATORACCURACY AND STABILITY

An ideal sine wave RF oscillator producesnice, clean harmonic and noise-free outputon a stable frequency. But real RF oscillators

tend to have certain problems that deterio-rate from the quality of their output signals.Load impedance variation and DC powersupply variation can cause the oscillator fre-quency to shift. Temperature changes alsocause frequency change problems. Noise thatmodulates the oscillator, generated externallyor internally, produces phase noise side-bands around the oscillator signal.

An oscillator that changes frequencywithout any help from the operator is said todrift. Frequency stability generally refers tofreedom from frequency changes over a rela-tively short period of time (e.g., few secondsto dozens of minutes). This problem is differ-ent from aging, which is frequency changeover relatively long periods of time (i.e.,Hz/yr) caused by the aging of the components(some electronic components tend to changevalue with long use). Temperature changes area large contributor to this problem in two


Fig. 8.31 Third-overtone Colpitts oscillator.

V+

C2

C3

R25.6K

R4470

C50.001 µF

OUTPUT

Q12N5770

Y1R1560

L1 C1

R315K

C4

FREQ (MHz) C1 (pF) C2 (pF) C3 (pF) C4 (pF)

15 - 25 100 100 68 33

25 - 55 100 68 47 33

50 - 65 68 33 15 22

L1 (0.25-inch form)

12 t, #30, CW

8 t, #30, CW

6 t, #22, CW

forms: warm-up drift (i.e., drift in first 15 min)and ambient temperature change drift.

If certain guidelines are followed, thenit is possible to build a very stable oscillator.For the most part, the comments that followapply to both crystal oscillators and L-Ctuned oscillators (e.g., variable frequency os-cillators), although in some cases one or theother is indicated by the text.

Temperature

Temperature variation has a tremendous ef-fect on oscillator stability. Avoid locating theoscillator circuit near any source of heatwithin the equipment it serves. In otherwords, keep it away from power transistors

or IC devices, voltage regulators, rectifiers,lamps, or other sources of heat.

THERMAL ISOLATION

One approach is to thermally isolate the os-cillator circuits. Figure 8.33 shows one suchmethod. The oscillator is built inside a metalshielded cabinet, as usual, but has styro-foam insulation applied to the sides. Onetype of poster board has a backing of styro-foam to give it substance enough for self-support. It is easy to cut using the hobby orrazor knives. Cut the pieces to size, thenglue them to the metal surface of theshielded cabinet, using contact cement orsome other form of cement that will adhereto metal and paper.


Fig. 8.32 Modification of crystal to operate at 65 to 110 MHz, with fifth- or seventh-overtone crystals.

+12VDC

Q12N5770

R110K

R22.7K R3

220

R4*

L2*

C41000pF C3

22 pF

C1

C2

Y1

C51000 pF

C60.001 µF

L1*

RFC122 µH

OUTPUT

FREQ (MHz)

65 - 85

85 - 110

C1 (pF) C2 (pF) C3 (pF) L1 L2

15 100150

6810010

7 t., #24, 3/16 in. CW 10 t. #34 over 10-ohm 1/4-W

4 t., #24, 3/16 in. 1WD 10 t. #34 over 10-ohm 1/4-W

CW = CLOSE WOUND1WD = SPACED 1 WIREDIAMETER

CRYSTAL OVENS

It is common practice to use a constant-temperature oven (Figure 8.34) for crystaloscillators, when very good temperaturestability is required. The oven keeps thecrystal at a constant temperature of 75 or80°C. There are two basic oven forms: snapaction and proportional. The snap actionoven uses a thermal switch that turns onand off at preset temperatures, much likethe thermometer in a centrally heatedhouse. The proportional oven uses a tem-

perature sensor and circuitry to provideheating as needed to keep the temperaturestable. A number of integrated circuits areavailable that serve as proportional con-trollers. Both socket-mounted and printed-circuit ovens are available (often fromcrystal manufacturers).

SELF-HEATING

Operate the oscillator at as low a power levelas can be tolerated to prevent self-heating ofthe active device and associated frequency-


Fig. 8.33Thermal treatmentof a VFO.

VARIABLECAPACITOR

MAININDUCTOR

PRINTEDCIRCUITBOARD

RFCONNECTOR

METALSHIELDED BOX

STYROFOAMPANEL

INSULATION

Fig. 8.34Crystal ovens.

CRYSTAL

XTAL SOCKETCRYSTAL

CIRCUITRY & TEMPSENSOR

PCB CRYSTALOVEN

determining components. It generally isagreed that a power level on the order of1–10 mW is sufficient. If higher power isneeded, then a buffer amplifier can be used.The buffer amplifier also isolates the oscilla-tor from load variations.

Other Criteria

USE LOW FREQUENCIES

In general, L-C-controlled VFOs should notbe operated at frequencies above about 12MHz. For higher frequencies, it is better touse a lower-frequency VFO and heterodyneit against a crystal oscillator to produce thehigher frequency. For example, one commoncombination uses a 5–5.5 MHz main VFO forall HF bands in SSB transceivers.

FEEDBACK LEVEL

Use only as much feedback in the oscillatoras is needed to ensure that the oscillatorstarts quickly when turned on, stays opera-tional, and does not “pull” in frequencywhen the load impedance changes. In somecases, a small-value capacitor is located be-tween the L-C resonant circuit and the gateor base of the active device. The small-valuecapacitor is used to prevent drift by lightlyloading the tuned circuit. The most commonmeans for doing this job is to use a 3-12 pFNP0 disk ceramic capacitor. Adjust its valueto the minimum level that ensures good start-

ing and freedom from frequency changes un-der varying load conditions.

OUTPUT ISOLATION

A buffer amplifier, even if it is a unity gainemitter follower, is highly recommended. Itpermits the oscillator signal power to buildup, if needed, without loading the oscillator.The principal use of the buffer is to isolatethe oscillator from variations in the outputload conditions.

Figures 8.35 and 8.36 show typicalbuffer amplifier circuits. The circuit in Figure8.35 is a standard emitter follower (i.e., com-mon collector) circuit. It produces near unityvoltage gain, but some power gain. Thebuffer amplifier shown in Figure 8.36 is afeedback amplifier using a 4:1 BALUN-styletoroidal core transformer in the collector cir-cuit. This circuit exhibits an input impedancenear 50 Ω, so it should be used only if the os-cillator circuit can drive 50 Ω. In some cases,a higher impedance circuit is needed.

DC POWER SUPPLY

Power supply voltage variations have a ten-dency to frequency modulate the oscillatorsignal. Because dynamic circuit conditionsoften result in a momentary transient droopin the supply voltage and line voltage varia-tions can cause both transient drops andpeaks, it is a good idea to use a voltage-regulated DC power supply on the oscillator.


Fig. 8.35 Standard postamplifier circuit.

C60.001 µF

OUTPUT

V+

C50.1 µF

C70.01 µF

R712K

R812K

R93.3K

INPUT

It also is a very good idea to use a volt-age regulator to serve the oscillator alone,even if another voltage regulator is used toregulate the voltage applied to other circuits(see Figure 8.37). Although this double-regu-lation approach may have been a cost bur-den, it is reasonable given the cost ofthree-terminal IC voltage regulators and theadvantage gained. For most low-powered os-cillators, a simple low-power L-series (e.g.,78L06) three-terminal integrated circuit volt-age regulator is sufficient (see U2 in Figure8.37). The L-series devices provide up to 100mA of current at the specified voltage,enough for most oscillator circuits.

Capacitors on both the input and out-put sides of the voltage regulator (C1 and C2in Figure 8.37) add further protection fromnoise and transients. The value of these ca-pacitors is selected according to the amount

of current drawn. The idea is to have a localsupply of stored current to temporarily han-dle sudden demand changes, allowing timefor a transient to pass or the regulator to“catch up” with the changed situation.

VIBRATION ISOLATION

The frequency setting components of the os-cillator can affect the stability performance,too. The inductor should be mounted to pre-vent vibration. While this requirement meansdifferent things to different styles of coil, itnonetheless is important. Some people preferto mount coils rigidly, while others will putthem on a vibration isolating shock absorbermaterial. I have seen a receiver that requireda very low SSB noise sideband local oscilla-tor be unable to meet specifications untilsome rubber shock absorbing material wasplaced underneath the oscillator printed


Fig. 8.36 Postamplifier circuit with 50 Ωinput/output impedances.

C10.1 µF

INPUT

T1

R1560

R23.3K

R31K

R410

R5100

C20.1 µF

C30.1 µF

C40.1 µF

C50.1 µF

+12VDC

R6150

R768

C60.1 µF

OUTPUT

Q12N5179

wiring board. Any time vibration affects crys-tals, inductors, or capacitors, high vibration-induced noise on the oscillator output signalis possible.

COIL CORE SELECTION

Air core coils generally are considered su-perior to those with either ferrite or pow-dered iron cores, because the magneticproperties of the cores are affected by tem-perature variation. Of those coils that do usecores, the slug tuned are said to be best, be-cause they can be operated with only asmall amount of the tuning core actually in-side the windings of the coil, reducing the vul-nerability to temperature effects. Still, toroidalcores have a certain endearing charm andcan be used wherever the ambient temper-ature is relatively constant. The type-SF ma-terial is said to be the best in this regard,and it is easily available. Or use type-6 ma-terial. For example, you could wind a T-50-6 core and expect relatively good frequencystability.

COIL-CORE PROCESSING

One source recommends tightly winding thecoil wire onto the toroidal core, then heat an-

nealing the assembly. This means placing itin boiling water for several minutes, then re-moving and allowing it to cool in ambientroom air while it sits on an insulated pad. Ihave not tried this, but the source reportedremarkable freedom from inductor-causedthermal drift.

For most applications, especially wherethe temperature is relatively stable, the coilwith a magnetic core can be wound fromenameled wire (#20–#32 AWG usually isspecified), but for best stability Litz wire isrecommended. Although a bit hard to get insmall quantities, this wire offers superior per-formance over relatively wide changes intemperature. Be aware that this nickel-basedwire is difficult to solder properly, so be pre-pared for a bit of frustration.

AIR CORE COILS

For air core coils, use #18 SWG or largerbare solid wire, wound on a low-tempera-ture coefficient of expansion formers. Figure8.38 shows how the coil can be mounted ina project. Stand-off insulators provided ade-quate clearance for the coil and hold it tothe chassis. The mount shown in Figure 8.38has shock absorbers to prevent movement.


Fig. 8.37 Power supply scheme using a voltage regulator.

U178L12

VIN VOUT

GND

U278L08

VIN VOUT

GND

+15 TO+18 V

0 V

+ + +C4100 µF

C12.2 µF

C21 µF

OSCILLATORCIRCUITC3

0.1 µF

OTHERCIRCUITS

C50.1 µF

CAPACITOR SELECTION

The trimmer capacitors used in the oscillatorcircuit should be air dielectric types, ratherthan ceramic or mica dielectric trimmers, be-cause of the lower temperature coefficient.

The small fixed capacitors used in theoscillator should be either NP0 disk ceramics(i.e., zero temperature coefficient), silveredmica, or polystyrene types. Some people dis-like the silvered mica types because theytend to be a bit quirky with respect to thetemperature coefficient. Even samples fromthe same batch can have widely differingtemperature coefficients on either side of 0.

Sometimes, fixed capacitors will comewith other than a zero temperature coeffi-cient in an oscillator frequency-determiningcircuit. These are done to make temperature-compensated oscillators. The temperaturecoefficients of certain critical capacitors are

selected to create a counterdrift that cancelsout the natural drift of the circuit.

The main tuning air variable capacitorshould be an old-fashioned double bearing(i.e., a bearing surface on each end plate).For best results, it should be made with ei-ther brass or iron (not aluminum) stator androtor plates. The capacitor should be rugged.

TEMPCO CIRCUIT

Temperature-compensated crystal oscillatorscan be built by using the temperature coeffi-cient of some of the capacitors in the circuitto cancel drift in the opposite direction. Thecircuit in Figure 8.39 was used in an radiotransceiver at one time. It is a standardColpitts crystal oscillator, but with the addi-tion of a temperature coefficient cancellationcircuit (C8A, C8B, C9, and C10).

The key to this circuit is C8, which is adifferential variable capacitor; that is, it con-tains two sections connected reciprocally sothat one is increasing capacitance as theother is decreasing capacitance as the shaftturns. In most cases, C9 and C10 are of equalvalue, but one is NP0 and the other N750 orN1500. If C8 is differential and C9 = C10,then the net capacitance across crystal Y1 isconstant with changes in C8 shaft rotation.But the net temperature coefficient doeschange. The idea is to crank in the amount oftemperature coefficient that exactly cancelsthe drift in the circuit’s other components.


Fig. 8.38 Air coil mounting.

Fig. 8.39 Temperature compensation of a crystal oscillator.

C1

C2

R110K

R3470

R210K

C3100 pF

C40.01 µF

OUTPUTTO

BUFFER

Q1

U178L05

C51 µF

+9 to +12VDC

Y1

C610 pF

C750 pF

C8A C8B

C9NPO

C10N750

The problem with this circuit is that dif-ferential capacitors are quite expensive. Itmay be useful, however, to connect the cir-cuit as shown and find the correct setting.Once the setting is determined, disconnectC8–C10 without losing the setting and mea-sure the capacitance of the two sections.With this information you can calculate thecapacitances required for each temperaturecoefficient and replace the network with ap-propriately selected fixed capacitors.

In general, when selecting temperaturecompensating capacitance, keep in mind thefollowing relationship for the temperaturecoefficient of frequency (TCF):

(8.4)

where

TCF is the temperature coefficient offrequency;

TCL is the temperature coefficient of theinductor;

TC1 is the temperature coefficient of C1;TC2 is the temperature coefficient of C2;CTotal is the sum of all capacitances in

parallel with L1, including C1, C2,and all strays and other capacitors.

This equation assumes that two capaci-tors, C1 and C2, are shunted across an induc-tor in a VFO circuit.

VARACTORS

Voltage variable capacitance diodes (varac-tors) often are used as a replacement forthe main tuning capacitor. If this is done,then it becomes critical to control the envi-ronmental temperature of the oscillator.Temperature variations seem to result inchanges in diode pin junction capacitance,and that contributes much to thermal drift.Varactor temperature coefficients of 450ppm are not unusual.

A good way to stabilize these circuitsis to supply a voltage regulator that has atemperature coefficient opposite in direc-tion to the varactor drift. By matching thesetwo, it is possible to cancel the drift of thevaractor.

Figure 8.40 shows a basic circuit thatwill accomplish this job. Diode D1, in serieswith DC blocking capacitor C1, is in parallelwith the inductor (L1). These componentsare connected into an oscillator circuit (notshown for sake of simplicity).

Normally, resistor R3 (10–100 KΩ, typi-cally) is used to isolate the diode from thetuning voltage source and is connected tothe wiper of a potentiometer that varies thevoltage. In this circuit, however, an MVS-460(USA type number) or ZTK33B (Europeantype number) varicap voltage stabilizer ispresent. This device is similar to a zenerdiode but has the required −2.3 mV/°C tem-perature coefficient. It operates at a currentof 5 mA (0.005 A) at a supply voltage of

TCF

TT C

C

T C

CCLC C

=− + × + ×

1 21 1

2Total Total


Fig. 8.40 Temperature-compensating IC.

U1MVS460/ATK33B

L1

C10.01 µFR3

10KR222

C24.7 µF

+C30.001 µF

R1

V+

D1VARACTOR

34–40 V DC. The value of resistor R1 isfound from

(8.5)

For example, when V+ is 40 V, then the re-sistor should be 1400 Ω (use 1.5 KΩ).

FREQUENCY SYNTHESIZERS

In modern radio receivers, the local oscillatorusually is a frequency synthesizer. In general,the synthesizer outputs stable phase-coherentsignals derived from a master oscillator. Themaster oscillator may be a high-precision crys-tal oscillator or an atomic standard (such as arubidium gas cell or cesium atomic beam).The characteristics that must be specified forthe frequency synthesizer include frequencyrange, frequency resolution, maximum fre-quency error, settling time, frequency indica-tion method, reference frequency, harmonicdistortion, SSB phase noise, discrete spuriousfrequencies (“spurs”), and wideband noise.

Figure 8.41 shows a single-loop, phase-locked loop (PLL) frequency synthesizer. Thesignal is generated in a voltage-controlled os-cillator (VCO) circuit. The output of the VCO isdirected to a divide-by-N counter circuit, aswell as being output. The output of the divide-by-N counter is applied to one input of a

phase detector circuit. The other input of thephase detector receives the reference oscillatorcircuit. This oscillator may be a crystal oscilla-tor with a frequency divider output, to pro-duce the reference signal. The output of thephase detector is an error signal proportionalto Fn − FR, where Fn is the VCO signal and FR isthe reference signal. This signal is processed ina low-pass filter, and often a DC amplifier, be-fore being applied to the control voltage inputof the VCO. Thus, the VCO frequency is pulledto the reference oscillator frequency.

Consider an example of a receiver de-signed to operate on 162.55 MHz. If the volt-age controlled oscillator works between 155and 165 MHz, a frequency division ratio inthe divide-by-N counter of 16,255 will reducethe operating frequency to 1 kHz.

Some signal sources produce a singleoutput frequency (or a discrete number offixed output frequencies). These receiversmay be used for channelized receiver sys-tems. Other signal sources produce outputover a very wide range of frequencies.

Output Signal Quality

It would be nice if all signal sources wereideal; that is, the output frequency and outputlevel were noiseless and perfectly calibrated.This never occurs, although the differences inthese specifications are between high- andlow-quality receivers.

R

V1

33

0 005=

+ −( )

.

Volts

Amps


Fig. 8.41PLL synthesizer block diagram.

VOLTAGECONTROLLEDOSCILLATOR

DIVIDE-BY-NCOUNTER

PHASEDETECTOR

REFERENCEOSCILLATOR

LOW PASSFILTER

DC AMPLIFIER

FOUT

Frequency

The important considerations regarding fre-quency are the range, resolution, accuracy,and (in certain receivers applications) theswitching speed.

RANGE

The frequency range is a specification thattells the specific frequencies covered. Insome cases, only one frequency, or somesmall number of discrete frequencies, is cov-ered. In other cases, one or more bands offrequencies are provided.

RESOLUTION

The resolution is the statement of the small-est increment of frequency that can be set.On analog receivers have no counter, theresolution is poor. The resolution may (butnot certainly) be improved by adding a digi-tal frequency counter to measure the outputfrequency. On modern synthesizers, it is pos-sible to set frequency with extremely goodresolution.

ACCURACY

Accuracy refers to how nearly the actual out-put frequency matches the set frequency. It isa function of the set frequency (and howclosely it can be set), FSet; long-term aging(τaging); and the time since the last calibration(τcal). Mathematically,

Accuracy = ±FSet × τAging × τCal (8.6)

For example, the receiver is set to 480MHz and has an aging rate of 0.155 ppm/yr.It has been six months (0.5 yr) since the lastcalibration.

Accuracy = ±FSet × τAging × τCal

= ± (480-MHz) × (0.155 ppm/yr) × (0.5 yr)

= ± 37.2 Hz.

There also may be some random varia-tion in the output frequency. Figure 8.42shows the uncertainty band around the setfrequency. The actual output frequency, Fo,will be FSet ± Accuracy. The general practiceis to calibrate a receiver on six-month or an-nual schedules, depending on the use.

SWITCHING SPEED (SETTLING TIME) The settling time is the length of time, usuallyin milliseconds or microseconds, required fora synthesized signal source to move to a newfrequency when digitally commanded tochange. It is calculated as the length of timefor the error of the frequency or output levelcommanded by the change to come intospecification range.

Output Level

The output level can be expressed in volt-age, power, or dBm notation. All are equiv-alent, although one or the other will bepreferred in most cases. The most commonmethod of describing the output level is indBm. As with frequency, some factors affectthe accuracy of the actual output vs. the setoutput.

Spectral Purity

The output signal is not always nice andclean. Although the purity of the output sig-


Fig. 8.42 Band of amplitude vs. frequency withuncertainty.

UNCERTAINTY

FREQUENCY

F

AM

PLI

TU

DE

FSET

nal is a distinguishing factor that differenti-ates lower- and higher-quality generators, allproduce signals other than the one desired.Figure 8.43A shows a typical spectrum out-put. This display is what might be seen on a

spectrum analyzer. The main signal is a CWsine wave, so ideally we would expect only asingle spike, with a height proportional tothe output level. But, a lot of other signalsare present.


Fig. 8.43 Spectral purity: (A) spurs and phase noise on an LO signal; (B) residual FM.

F nFnF/2 FOther

FREQUENCY

AM

PLI

TU

DE

~ -45 dB~ -65 dB

~ 30 dB

MAIN SIGNAL

PHASENOISE

SUBHARMONIC

HARMONIC

SPUR

+300 +3000-3000 -300

FO

RESIDUAL FM ISINTEGRATED PHASE

NOISE OVERSPECIFIED

BANDWIDTH

A

B

First, note that the main signal is spreadout by phase noise. This noise is randomvariation around the main frequency. Whenintegrated over a specified bandwidth, suchas 300–3000 Hz, the phase noise is calledresidual FM (Figure 8.43B).

Second, harmonics are present. If themain signal has a frequency of F, the har-monics have frequencies of nF, where n isan integer. For example, the second har-monic is 2F, and the third harmonic is 3F. Inmany cases, the 3F harmonic is strongerthan the 2F harmonic, although in generalhigher harmonics are weaker than lowerharmonics.

There sometimes also are subharmon-ics. These are integer quotients of the mainsignal. Again, if the F is the main signal fre-quency, nF/2 represents the subharmonics.Typically, unless something interferes withthe output signal, subharmonics are not asprominent. One thing that does make sub-harmonics prominent, however, is the use offrequency multiplier or divider stages (whichis the case in many modern generators).

Finally, miscellaneous spurious signals(spurs) are found on some generators. Thesemight be due to power supply ripple modu-lating the output signal, parasitic oscillations,digital noise from counter- or phase-lockedloop circuits, and other sources.

Harmonics and spurs usually are mea-sured in terms of decibels below the carrier(dBc), where the carrier is the amplitude ofthe main output signal. In general, the fewerare the unwanted components, the better thesignal source.

Phase noise warrants some special con-sideration. It usually is measured in terms ofdBc/Hz; that is, decibels below the carrierper hertz of bandwidth. This noise is concen-trated around the main signal frequency andnormally graphed on a log-log scale to per-mit both close-in and further-out noise com-ponents to be compressed on one graph.

Architectures

A synthesizer architecture is shown inFigure 8.44. This modern signal generator is

capable of producing very accurate, high-quality signals. Three main sections com-pose the signal source: reference section,frequency synthesizer, and output section.In Figure 8.44, each section is broken downinto further components for the sake of easyanalysis.

REFERENCE SECTION

The reference section is at the very core ofthe signal generation process. It is an accu-rate, stable fixed-frequency source such as acrystal oscillator. The frequency of the refer-ence section must be precisely adjustableover a small range so it can be compared to ahigher-order standard, such as a cesium beamoscillator or WWVB comparator receiver, forcalibration.

Because it controls the frequency syn-thesizer, the stability of the reference sectiondetermines the overall stability of the signalgenerator. The stability of ordinary crystal os-cillators is reasonably good for many pur-poses but not for use in the reference sectionof a signal source. For that purpose, eithertemperature-compensated crystal oscillatorsor oven-controlled crystal oscillators are used.The TCXO typically exhibits crystal aging ofbetter then ±2 ppm/yr and a temperature ag-ing of ±1 ppm/yr. The OCXO is capable of0.1 ppm/yr for crystal aging and 0.01 ppm/yrfor temperature aging.

The crystal oscillator usually is oper-ated at a frequency such as 5 MHz, butlower frequencies often are needed. To gen-erate these lower frequencies, a divide-by-Ndigital counter is provided. This circuit dividesthe output frequency by some integer, N, toproduce a much lower reference frequency.

Many signal generators provide REF.OUT and EXT. REF. IN capability (these con-nections often are located on the rear panelof signal generators, so they may be over-looked). The REF. OUT connector providesthe reference signal to other instruments, orit can be used for calibration. The EXT. REF.IN allows the use of an external referencesource in place of the internal reference sec-tion. This sometimes is done to lock up twosignal generators that must work together or


1/2

to substitute a much more accurate referencesuch as a cesium beam clock.

FREQUENCY SYNTHESIZER

The actual signal is produced in the frequencysynthesizer section. It is generated by a volt-age controlled oscillator, whose output iscompared to the reference signal. A voltagevariable capacitance diode (varactor) can beused for the VCO, as can surface acousticalwave oscillators (used at higher frequenciesand in the microwave bands).

The frequency of the VCO is set by aDC control voltage applied to its tuning inputline. This control voltage is generated by in-tegrating the output of a phase detector orphase comparator that receives the referencefrequency and a divide-by-N version of the

VCO frequency as inputs. When the two fre-quencies are equal, the output of the phasedetector is 0, so the VCO tuning voltage is atsome quiescent value.

But, if the frequency of the VCO drifts,the phase detector output becomes nonzero.The integrator output ramps up in the direc-tion that will cancel the frequency drift of theVCO. The VCO frequency is continuouslyheld in check by corrections from the inte-grated output of the phase detector. Thistype of circuit is a phase-locked loop.

Suppose, for example, a signal sourcehas a reference of 5 MHz and is divided by20 to produce a 250 kHz reference. If the fre-quency synthesizer divide-by-N stage is setfor, say, N = 511, then the VCO output fre-quency will be 0.25 MHz × 511 = 127.75


/2

Fig. 8.44 Frequency synthesizer block diagram.

DIVIDE-BY-N

REFERENCEOSCILLATOR

FREQUENCYSYNTHESIZER

REFERENCESECTION OUTPUT

SECTION

EXT.REF. IN

REF. OUT

f INTEGRATORVOLTAGE

CONTROLLEDOSCILLATOR

DIVIDE-BY-N

REFIN

DCCONTROL

OUTPUTPOWER

AMPLIFIERSAMPLER

ALCDRIVER

ALCMODULATOR

ALCRECTIFIER

VARIABLE OUTPUTATTENUATOR

MHz. Band switching, operating frequency,and frequency resolution are controlled bymanipulating the reference frequency divide-by-N and VCO divide-by-N settings. In somecases, the frequency is entered by keypad,and this tells the signal generator how to setthese values. Alternatively, “tunable” signalgenerators may have a digital encoder shaftconnected to a front panel control.

The noise component of the output sig-nal is composed of thermal noise and phasenoise. Of these two, the phase noise tends to dominate the performance of the signalsource. Both the reference oscillator and VCOphase noise contribute to the overall outputphase noise. There is a 20 log (N) degrada-tion of the phase noise performance of thesignal source because of the divide-by-N na-ture of the PLL. Fortunately, the bandwidth ofthe PLL tends to limit the contribution of theVCO phase noise to the overall phase noiseperformance.

OUTPUT SECTION

The output section performs three basicfunctions: It boosts power output to a speci-fied maximum level, provides precision con-trol over the actual output level, and keepsthe output level constant as the frequency ischanged.

The power amplifier is a wideband am-plifier that produces an output level of somevalue in excess of the required maximumlevel (e.g., +13 dBm). A calibrated precisionattenuator can be used to set the actual out-put level to any lower value required (e.g., −136 to +13 dBm).

The automatic level control (ALC) mod-ulator essentially is an amplitude modulatorcontrolled by a DC voltage developed byrectifying and filtering a sample of the RFoutput level. The ALC driver compares theactual output level with a preset value andadjusts the control signal to the ALC modula-tor in a direction that cancels the error.

Sweep generators (sweepers) are usedto produce a signal that changes frequencyover a specified range. They are used in re-ceivers that must sweep through a band offrequencies (e.g., spectrum analyzers or elec-

tronic warfare receivers). Although thesweeper nominally resembles an FM genera-tor, there are differences. For one thing, thesweep range usually is quite a bit larger thanthe deviation of an FM signal generator. Also,the sweeper tends to change frequency fromone limit to the other, then snaps back to thefirst limit.

There are three basic forms of sweep:linear (ramp) sweep, stepped sweep, and listsweep. All these forms use a voltage-con-trolled oscillator frequency synthesizer, butthe difference is in the waveform used forsweeping the output signal.

Figure 8.45A shows linear sawtoothwaveform sweep. At time T1, the ramp ap-plied to the VCO is at 0 and begins rampingup. The frequency begins to move upwardsfrom the 0 V value in a linear manner untiltime T2, at which point it snaps back to the0 V value.

The stepped sweep (Figure 8.45B) usesa series of discrete voltage steps to change thefrequency of the VCO. This method does notproduce an output on every frequency in theoutput voltage range from T1 to T2 but only atspecific values determined by the steps. Thesteps are produced in a circuit such as shownin Figure 8.46. A digital-to-analog converter(DAC) produces an output voltage that is pro-portional to the binary number applied to itsinputs. The maximum output voltage is set byan internal or external DC reference voltageand the applied binary number. In this case,the DAC input is driven by a binary counter,which in turn is incremented by a digital clock(square wave generator).

The maximum number of steps that canbe accommodated using any given binarycounter is 2N, where N is the bit length of thecounter. For the 16-bit counter shown inFigure 8.46, therefore, a total of 216 = 65,536different levels (including 0) can be created.The maximum output voltage is less than thereference voltage by the 1 LSB (least signifi-cant bit) voltage, or 1/2N. If a 16-bit counteris connected to a DAC with a 10.00 V refer-ence source, then the 1 LSB step is 0.00015 Vand the maximum output voltage is 10.00 −0.00015 V = 9.99985 V.



TIME

VO

LTA

GE

T

V

T1 T2

Fig. 8.45 Sweep waveforms: (A) sawtooth; (B) stepped from a DAC circuit.

Fig. 8.46 DAC circuit and binary counter to generate a stepped waveform.

DIGITAL-TO-ANALOGCONVERTER

16-BIT BINARY COUNTERCLOCK

DCREFERENCE

OUTPUT

VO

LTA

GE

T

V

T1 T2T0

A

B


List sweepers also have a DAC to con-trol the VCO but drive it with a series ofvalues stored in a digital memory similar tothose used in computers. This permits arbi-trary waveforms to be created.

Figure 8.47 shows one way sweepers arebuilt: open loop. The frequency synthesizerloop is broken and a linear sawtooth, stepped

waveform, or list generated waveform is ap-plied to its DC control voltage input.

Another approach is shown in Figure8.48. This is a closed loop approach. The fre-quency synthesizer loop is not open; rather,a binary counter or list memory is used todrive the divide-by-N counter in the synthe-sizer PLL feedback loop.

Fig. 8.47 Open loop synthesizer circuit.

f INTEGRATORVOLTAGE


DIVIDE-BY-N

REFIN

DCCONTROL

SWEEPWAVEFORMGENERATOR

Fig. 8.48 Closed loop synthesizer circuit.

f INTEGRATORVOLTAGE


DIVIDE-BY-N

REFIN

DCCONTROL

BINARYCOUNTER

CLOCK

A number of IF filters are used in radio re-ceivers today. In addition to the L-C filters,various types of crystal filter, monolithic ce-ramic filters, and mechanical filters must beconsidered. In this chapter, we look at thevarious types of IF filter, their characteristics,and their applications.

Before delving into the topic, however,we look at some general filter theory as ap-plied to IF passband filters and how thesefilters are used. Figure 9.1A shows theButterworth passband characteristic: Thepassband is relatively flat. The Chebyshevfilter is shown in Figure 9.1B. It has a rip-pled passband but steeper slopes than theButterworth design.

The bandwidth of the filter is the band-width between the –3 dB points (Figure9.1A). The Q of the filter is the ratio of centerfrequency to bandwidth:

(9.1)

where

Q is the quality factor of the filter;FO is the center frequency of the filter;B is the –3 dB bandwidth of the filter.

Note that FO and B are in the same units.

The shape factor of the filter is definedas the ratio of the –60 dB bandwidth to the–6 dB bandwidth. This is an indication ofhow well the filter will reject out-of-band in-terference. The lower the shape factor is, thebetter (shape factors of 1.2:1 are achievable).

Figure 9.1C shows a generic IF ampli-fier with the filters in place. The IF amplifierprovides most of the gain in a superhetero-dyne receiver, as well as the bulk of the se-lectivity of the receiver. Selectivity is thefunction of the IF filter. It has the narrowestbandpass of all the filters in the receiver.

The IF amplifier may or may not usetwo (or more) filters, depending upon thetype of design. Where only one filter is used,the filter usually is placed at the input of theamplifier to eliminate the mixer products thatcan affect the IF amplifier performance. Thenoise produced by the IF amplifier can besignificant, which means that an output IF fil-ter is indicated to eliminate that noise.

L-C IF FILTERS

The basic type of filter, once the most com-mon, is the L-C filter, which comes in varioustypes (Figure 9.2). The type shown in Figure

Q

F

BO=

Chapter 9

IF Filters: General Filter Theory

151

9.2A contains two parallel-tuned L-C sec-tions. Although it is not apparent here, theinput and output sides of the L-C networkneed not have the same impedance, but thatusually is the case. This type of IF amplifierfilter has largely been eclipsed by other typesexcept in certain IC amplifiers.

A more common form today is shownin Figure 9.2B. This form has a low-imped-

ance tap for transistor or IC applications,with the high impedance portions still avail-able. In Figure 9.2C, we see a common formof IF filter in which the low-impedance tapis available to both input and output sides,but one side of the high-impedance portionsof the transformer is not. In Figure 9.2D, wesee a single-tuned IF filter. It has a standardIF filter input side, but it has only a low-impedance link on the output side. The IFfiltering is performed by the tuned L-C cir-cuit, whereas the low-impedance link is forimpedance matching.


Fig. 9.1 Passband filters: (A) Butterworth; (B) Chebyshev; (C) in use in a typical superheterodyne receiver.

FOFL FH

-60 dB

-6 dB

-3 dB

0 dB

FO

IF FILTER IF FILTER TODETECTOR

IFAMPLIFIER

FROMMIXER

C

A B

Fig. 9.2 Various L-C IF filter circuits: (A) with two parallel-tuned L-C sections; (B) with a low-impedancetap for transistor or IC applications; (C) with a low-impedance tap to both the input and output sides;(D) single-tuned IF filter; (E) double-tuned IF filter.

A

B

C

D

E

In Figure 9.2E, we see a double-tuned IFfilter that is not magnetically coupled (so it isnot a transformer) but rather is coupledthrough a common impedance. In this case, asmall-value capacitor is used as the commonimpedance, but inductors can be used as well.

CRYSTAL FILTERS

The quartz piezoelectric crystal resonator isideal for IF filtering, because it offers high Q(narrow bandwidth) and behaves as an L-C

circuit. Because of these features, it can beused as a high-quality receiver as well as asingle-sideband transmitter (filter type). Figure9.3 shows a typical quartz crystal used for os-cillators and filters. Shown are the various cutsof the crystal and the ideal use for each partic-ular cut. Quartz has a dielectric constant ofabout 3.8.

The schematic symbol for a crystal isshown in Figure 9.4A, and the equivalent L-C-R circuit is shown in Figure 9.4B. Theequivalent circuit shows a series inductance(Ls) and capacitance (Cs) as well as a series

IF Filters: General Filter Theory 153

Fig. 9.3 Crystal structure.

Y

Y

X

X

Z

Z

GTJT

BT

BC

FT

DT

ET

CT

AC

NT

MT

ATCUT

XCUT

YCUT

ZEROTEMPERATURECOEFFICIENT

FILTERS

HARMONICOSCILLATORS

LOWFREQUENCY

OSCILLATORS

HIGHFREQUENCY

OSCILLATORS

resistance (Rs). The series inductance and ca-pacitance sometimes are called the motionalreactance values. There also is a parallel ca-pacitance, Cp.

The parallel capacitance resonates withthe inductor to form a parallel resonance,while the series capacitance resonates withthe inductor to form a series resonance. Thisis graphed in Figure 9.4C. The graph illus-trates reactance (capacitive and inductive)against frequency. The frequency, marked Fs,is the series resonance point. The antireso-

nance is “officially” the parallel resonance;but in practical terms, there is a range of par-allel resonance.

Figure 9.5 shows a typical crystal pack-age and its mounting on a printed circuitboard. The basic package is shown in Figure9.5A. Note that the pins might be actual pinsor they may be wires. A cut-away view,shown in Figure 9.5B, is a bit more informa-tive about how the crystal works. Figure 9.5Cshows how the crystal usually is mounted ona double-sided printed circuit board. An in-


Fig. 9.4 Crystal schematics: (A) schematic symbol; (B) L-C-R circuit; (C) impedance vs. frequency curve.

XL

XC

IND

UC

TIV

E R

EA

CTA

NC

EC

APA

CIT

IVE

RE

AC

TAN

CE

FS

FREQUENCY

ANTIRESONANCE

PARALLELRESONANCE

RANGE

ESR

LS CS

CP

RS

A B

C

sulator is placed on the board to preventshort circuiting the board tracks with thecrystal package.

Figure 9.6A shows a simple crystal filterthat has been around in one form or anothersince the 1930s. Figure 9.6B shows the atten-uation graph for this filter. A “crystal phasing”capacitor, adjustable from the front panel,cancels the parallel capacitance, which can-cels the parallel resonance, leaving the seriesresonance of the crystal.

Although the 1930s vintage filters didnot use it, this filter is built in trifilar form.This means that the windings of T1 arewound together, interlaced with each other.

Figure 9.7A shows the circuit for a half-lattice crystal filter, and Figure 9.7B showsthe attenuation curve. This type of crystal fil-ter is used in low-cost radios. Like the simplecrystal filter just described, this version uses atrifilar coil for T1 but with a second crystal in

the circuit instead of the phasing capacitor.The frequency relationship between the twocrystals is shown in Figure 9.7B. They haveoverlapping parallel and series resonancepoints so that the parallel resonance of crys-tal no. 1 is the same as the series resonanceof crystal no. 2.

We can use the half-lattice filter to builda cascade half-lattice filter (Figure 9.8) and afull lattice crystal filter (Figure 9.9). The cas-cade half-lattice filter has increased skirt se-lectivity and fewer spurious responses thanthe same passband in the half-lattice filter. Itis a back-to-back arrangement on a bifilartransformer (T1). In practice, close matchingis needed to make the cascaded half-latticefilter work properly.

The full-lattice crystal filter (Figure 9.9)uses four crystals like the cascade half-latticefilter, but the circuit is built on a different ba-sis than the latter type. It uses two tuned


Fig. 9.5 Crystal package: (A) external view; (B) internal view; (C) mounting on a printed wiring board.

PCB

INSULATOR

CRYSTAL

METALCASE

QUARTZCRYSTAL

ELECTRODE

SUPPORTSTRUT

PINS

INSULATOR

SUPPORTSTRUT

A

B

C

transformers (T1 and T2), with two pairs ofcrystals that are cross-connected across thetuned sections of the transformers. CrystalsY1 and Y3 are of one frequency, while Y2and Y4 are the other frequency in the pair.

A different sort of filter is shown inFigure 9.10A, with its asymmetrical attenua-tion curve shown in Figure 9.10B. This filterhas a more gradual fall-off on one side thanon the other (Figure 9.10B). The filter has the


Fig. 9.6Simple crystal filter: (A)schematic; (B) resonancecurve. FS

INSERTIONLOSS

-3 dB

C1PHASING

RL

Y1CRYSTAL

T1

INPUT

Fig. 9.7 Half-lattice crystal filter: (A) schematic; (B) resonance curve.

Y1CRYSTAL

No. 1

Y2CRYSTAL

No. 2

RL

T1

C1INPUT

CRYSTAL Y1

CRYSTAL Y2

FS1

FS2

FP1

FP2

A

B

A

B

advantage that the crystals Y1 and Y2 are thesame frequency. A capacitor (C1) in the cir-cuit tunes to the desired passband. Thebandwidth of this filter is only half what isexpected from the half-lattice crystal filter.

Crystal Ladder Filters

Figure 9.11 shows a crystal ladder filter. This fil-ter has several advantages over the other types:

1. All crystals are the same frequency (nomatching is required);

2. Filters may be constructed using anodd or even number of crystal;

3. Spurious responses are not harmful(especially for filters over four or moresections);

4. Insertion loss is very low.


Fig. 9.8Half-lattice crystalfilter.

INPUT

C1

Y1

Y2

T1

Y3

Y4 RLRS

Fig. 9.10 Asymmetric crystal filter: (A) schematic; (B) attenuation curve.

T1 T2Y1

Y2

C1

FREQUENCY

OU

TP

UT

(dB

)

Fig. 9.9 Full-lattice crystal filter.

T1

C1INPUT RLC2

Y1

Y2

T2

Y3

Y4

A B

Both Butterworth and the equiripple orChebyshev responses can be created usingthis design. Ideally, in the Chebyshev design,the number of positive peaks should be thesame as the number of crystals and of equalamplitude over the passband of the filter. Inreality, fewer peaks than that are found,some merging with each other. In addition,the amplitude of the ripple increases towardthe edges of the band.

The design of this filter can be simpli-fied by using a test fixture to dope out theproblem first. The value of the end capaci-tors is

(9.2)

The value of the coupling capacitors is

(9.3)

And the value of REND is

(9.4)

where

B is the bandwidth in hertz (Hz);CEND is the end capacitance in pico-

farads (pF);CJK are the shunt capacitors in pico-

farads (pF);FO is the crystal center frequency;∆f is the bandwidth measured in a test

fixture;kJK is the normalized values given in

Tables 9.1 and 9.2;q is normalized end section Q given in

Tables 9.1 and 9.2;

RS is the end termination of the filter(RS > REND);

REND is the end termination to be usedwithout matching capacitors.

A special version of the crystal ladderfilter is the Cohn (minimum loss) filter ofFigure 9.12. This filter rotates the end capac-itors and gives the shunt capacitors equalvalue. It preserves a reasonable shape factor,while minimizing loss when built with prac-tical resonators. Like the crystal ladder filter,the Cohn filter uses the same frequency crys-tals throughout. The error in frequency be-tween the crystals (∆FO) should be less than10% of the desired bandwidth of the filter.

The design procedure given by Hayward(1987) is simplified here:

1. Pick a crystal frequency (between 2and 12 MHz). Hayward used 3.579MHz color burst crystals;

2. Pick a capacitance for the filter (200 pFis a good start, a higher capacitance

R

B

q fRSEND =

−120

∆

Cf

Bk FJKJK O

=

−1326 10∆

CR F

R

RS O

SEND

END

=×

× −

−1 59 10

1 55.


Fig. 9.11Crystal ladder filter.

C12

Y1 Y2 Y3 Y4

C23 C34 CENDINPUT

RS

CEND RL

Table 9.1 Normalized k and q Values(Butterworth)

N q k12 k23 k34 k45

2 1.414 0.7071 — — —

3 1.0 0.7071 0.7071 — —

4 0.785 0.8409 0.4512 0.8409 —

5 0.618 1.000 0.5559 0.5559 1.0

Table 9.2 Normalized k and q Values(Chebyshev)

N q k12 k23 k34 k45

2 1.638 1.6362 0.7016 — —

3 1.433 0.6618 0.6618 — —

4 1.345 0.685 0.5421 0.685 —

5 0.301 0.7028 0.5355 0.5355 0.7028

yields a narrower bandwidth buthigher insertion loss);

3. Vary the end termination impedanceto obtain a ripple-free passbandwhile providing sufficient stopbandattenuation.

Table 9.3 gives various Cohn filterbandwidths, termination impedances, andcapacitor values for a three-crystal filter.

Monolithic Ceramic Crystal Filters

Figure 9.13 shows a monolithic crystal filter.These filters often are made with syntheticpiezoelectric resonators rather than quartz.The filters are made in small packages, somein crystal packages and some in special pack-ages. Some of the special packages are smallerthan crystals and others are larger.

MECHANICAL FILTERS

Considerable improvement in filter action ispossible with the use of the mechanical fil-ter. These filters once were used in Collinshigh-end radio receivers and SSB transmit-ters but now are more widespread (althoughthe Rockwell/Collins company still makesthe filters).

The basic principle of operation is thephenomenon of magnetostriction; that is, thelength or circumference of a piece of mater-ial changes when it is magnetized. Nickeldoes this, although the effect is 1 part inabout 20,000. Other materials, such as theferrites or powdered iron type (61, Q1, or4C4), provide much stronger magnetostric-tion effects. In addition, these materials havea high electrical resistivity, so eddy currentlosses are minimized; and they have a me-chanical Q on the order of several thousand.These materials make far better filters thannickel. The typical ferrite material is formedat 1300–1400ºC and has a Q determined bythe proportion of oxides used in the forma-tion of the ferrite material.

Figure 9.14A shows a magnetostrictiveresonator, and Figure 9.14B shows the equiv-alent circuit. In Figure 9.14A, the ferrite rod iswound with a coil such that it is a slip fit. It isbiased magnetically with either a permanentmagnet or a DC component to the electricalsignal applied to the coil (L). When alterna-


Fig. 9.12Cohn filter.

C

Y1 Y2 Y3 Y4

C C

C

INPUTRS

C

RL

Fig. 9.13 Monolithic crystal filter.

Table 9.3 Cohn Three-Crystal Filter

Bandwidth C (pF) REND (Ω)

380 200 150

600 130 238

1000 70 431

1800 30 1500

2500 17 3300

tive current flows in the coil, L, it adds to orsubtracts from the magnetic field of the bias,causing the ferrite length to oscillate. But theparallel resonant component (Figure 9.14B)causes a sharp peak in the response at a fre-quency equal to the mechanical resonancefrequency. When the ferrite rod is wound intoa toroidal shape, the circumference (hence,the radius) or the toroid shape varies as it ismagnetized.

Figure 9.15 shows a mechanical filterbuilt using toroidal resonators. Various me-chanical filters are available in frequenciesbetween 60 and 600 kHz. A pair of trans-ducers are located at either end of the filterto translate electrical energy to mechanicalenergy and vice versa. The resonators sup-

ply a sharp shape factor of up to 1.2:1 (60 to 6 dB), with Q values of 8,000–12,000(this is up to 150 times the Q value of crys-tal filters). Over a temperature range of–25º to +85°C, the change of resonant fre-quency is as little as 1.5 parts per million. Inone test, the frequency shift for eightmonths was 1 ppm.

The mechanical filter consists of threebasic parts: the transducers, the mechanicallyresonant disks, and the disk coupling rods.The transducer is a magnetostrictive devicethat converts electrical energy to mechanicalvibrations and vice versa. The resonant disksform parallel resonant circuits, so increasingthe number of disks decreases the bandwidthof the circuit.


Fig. 9.14 Magnetostrictive (A) resonator and (B) circuit.

INDUCTANCEL

FERRITEROD

L

CR

A B

Fig. 9.15Mechanical filter. PINS

HEADER

TRANSDUCERTRANSDUCER TOROIDALRESONATORS

SAW Filters

The surface acoustic wave filter is a crystalin which mechanical energy is first con-verted to mechanical energy, and then backto electrical energy. This provides a verynarrow bandpass. A SAW filter consists of athin substrate sliced from a single crystal onwhich two sets of aluminum comb-shapedelectrodes are vapor deposited. When a highfrequency is applied to one of these elec-trodes, that power is converted to mechani-cal energy, transmitted across the crystal,and converted back to electrical energy atthe other end. The most common substratematerials include quartz, lithium niobate,lithium tantalate crystals, and langasite, withsome lithium tetraborate.

SAW filters tend to have a higher fre-quency than many IFs and are used inVHF/UHF/microwave receivers. SAW filterscan be had from about 21 MHz and up, sothey easily fill the bill for 30, 50, or 70 MHz IFs.

BIBLIOGRAPHY

The ARRL Handbook for Radio Amateurs, 1998(CD-ROM version), Chapter 16. Newington,CT: ARRL.

Bottom, V. E. Introduction to Quartz Crystal UnitDesign. New York: Van Nostrand ReinholdCo., 1982.

Carver, Bill. “High Performance Crystal FilterDesign.” Communications Quarterly (Winter1993).

Colin. “Narrow Bandpass Filters Using IdenticalCrystals Design by the Image ParameterMethod” [in French]. Cables and Transmission21 (April 1967).

Cohn, S. “Direct Coupled Resonator Filters.” Pro-ceedings of IRE (February 1957).

Cohn, S. “Dissipation Loss in Multiple CoupledResonators.” Proceedings of IRE (August 1959).

Dishal, “Modern Network Theory Design of Sin-gle Sideband Crystal Ladder Filters.” Proceed-ings of IEEE 53 (September 1965).

Gottfried, Hugh L. “An Inexpensive Crystal-FilterI.F. Amplifier.” QST (February 1958).

Hamish, J. “An Introduction to Crystal Filters.”RSGB Bulletin (January–February 1962).

Hardcastle, J. A. “Some Experiments with High-Frequency Ladder Crystal Filters.” QST(December 1978) and Radio Communications(January, February, September 1977).

Hardcastle, J. A. “Ladder Crystal Filter Design.”QST (November 1980): 22–23.

Hayward, W. “A Unified Approach to the Designof Crystal Ladder Filters.” QST (May 1982):21–27.

Hayward, W. “Designing and Building SimpleCrystal Filters.” QST (July 1987): 24–29.

Makhinson, Jacob. “Designing and BuildingHigh-Performance Crystal Ladder Filters.”QEX, no. 155 (January 1995).

Pochet, W. “Crystal Ladder Filters,” TechnicalTopics, Radio Communications (September1976) and Wireless World (July 1977).

“Refinements in Crystal Ladder Filter Design.”QEX, no. 160 (June 1995), CD-ROM version.

Sabin, William E. (1996). “The Mechanical Filterin HF Receiver Design.” QEX, March 1996.

Sykes. A New Approach to the Design of High-Frequency Crystal Filters. Bell System Mono-graph 3180. Bell Systems, n.d.

Van Roberts, Walter B. “Magnetostriction Devicesand Mechanical Filters for Radio Frequencies.Part I: Magnetostriction Resonators.” QST(June 1953).

Van Roberts, Walter B. “Magnetostriction Devicesand Mechanical Filters for Radio Frequencies.Part II: Filter Applications.” QST (July 1953).

Van Roberts, Walter B. “Magnetostriction Devicesand Mechanical Filters for Radio Frequencies.Part III: Mechanical Filters.” QST (August1953).

Vester, Benjamin H. “Surplus-Crystal High-Frequency Filters: Selectivity at Low Cost.”QST (January 1959): 24–27.

Zverev, A. L. Handbook of Filter Synthesis. NewYork: John Wiley and Sons, 1967.


Most of the gain and selectivity of a super-heterodyne radio receiver are in the interme-diate frequency amplifier. The IF amplifier,therefore, is a high-gain, narrow bandwidthamplifier. Typically, IF power gains run inthe 60–120 dB range, depending on the re-ceiver design. The IF amplifier usually has afar narrower bandwidth than the RF ampli-fier. The filters used are described in detail inChapter 9 and to some extent here.

FILTERS

The purpose of the IF amplifier is to providegain and selectivity to the receiver. The selec-tivity portion of the equation is provided byany of several different types of filter circuit,some of which are shown in Figure 10.1. Theclassic circuit is shown in Figure 10.1A. Thistransformer is shown with taps but also mayexist without the taps. The taps provide alow-impedance connection, while retainingthe overall advantages of high-impedancetuned circuits. Note that the capacitors usu-ally are inside the transformer-shielded can.A slightly different version is shown in Figure10.1B. This transformer differs from the pre-

vious one in that the secondary winding iscapacitor coupled to its load, and that capac-itor may or may not be resonating. Still athird version is had by making the secondarywinding an untuned low impedance loop.

A somewhat different approach isshown in Figure 10.1C. This transformer is aseries resonant tapped circuit on the inputend and parallel tuned and tapped on theoutput end. Two shield cans are needed toimplement this approach.

Finally, we have the representationshown in Figure 10.1D. This symbol repre-sents any of several mechanical or crystal filters. Such filters usually give a much nar-rower bandwidth than the L-C filters dis-cussed previously.

AMPLIFIER CIRCUITS

A simple IF amplifier is shown in Figure 10.2.A simple AM band radio may have one suchstage, while FM receivers, shortwave re-ceivers, and other types of communicationsreceiver may have two to four stages likeFigure 10.2. This IF amplifier is based on thetype of L-C filter circuits discussed in Figure

Chapter 10

IF Amplifier Circuits

163

10.1A. Transformer T1 has a low-impedancetap on its secondary winding, connected tothe base of a transistor (Q1). Similarly withT2, but in this case, the primary winding istapped for the collector of the transistor.

The bias for the transistor is providedby resistors R1 and R2, with capacitor C1used to place the cold end of the T1 sec-ondary winding at ground potential for ACsignals. Resistor R3 provides a bit of stability


Fig. 10.2 Traditional IF amplifier.

T2

T1

FROMMIXER

TODEMODULATOR

Q1

R1

R2R3

C1

R4C2 C3

C4

+12VDC

Fig. 10.1 L-C filters: (A) classic circuit; (B) circuit with the secondary winding capacitor coupled to itsload; (C) circuit that is series tapped on the input end and parallel tuned and tapped on the output end;(D) universal filter symbol.

FL1

FL1

A

C

B D

to the circuit. Capacitor C2 keeps the emitterof the transistor at ground potential for ACsignals, while keeping it at a potential of IER3for DC. The reactance of capacitor C2 shouldbe <R3/10. Resistor R4 forms part of the col-lector load for transistor Q1. Capacitors C3and C4 bypass and decouple the circuit.

Figure 10.3 shows a gain-controlledversion of Figure 10.2. This particular cir-cuit is designed for low-frequency use(240–500 kHz), although with certain com-ponent value changes it could be used forhigher frequencies as well. Another differ-ence between this circuit and that of Figure10.2 is this circuit’s double-tapped trans-formers for T1 and T2.

The major change is the provision forautomatic gain control (see Chapter 12). Acapacitor is used to sample the signal for

some sort of AGC circuit. Furthermore, thecircuit has an additional resistor (R3) to theDC control voltage of the AGC circuit.

CASCODE-PAIR AMPLIFIER

A cascode-pair amplifier is shown in Figure10.4. This amplifier uses two transistors (bothJFETs) in an arrangement that puts Q1 in thecommon source configuration and Q2 in thecommon gate configuration. The two transis-tors are direct coupled. Input and output tun-ing is accomplished by a pair of L-C filters(L2C1 and L3C2). To keep this circuit fromoscillating at the IF frequency, a neutraliza-tion capacitor (C3) is provided. This capaci-tor is connected from the output L-C filter onQ2 to the input of Q1.

IF Amplifier Circuits 165

Fig. 10.3Traditional IF amplifierwith an AGC circuit.

R4470

T1

T2

Q1

C10.047 µF

C20.047 µF

C3180 pF

AGCSAMPLE

AGCCIRCUIT

R215K

R383K

V+

C40.1 µF

R568

R147K

“UNIVERSAL” IF AMPLIFIER

The IF amplifier in Figure 10.5 is based onthe popular MC-1350P integrated circuit. Thischip is readily available through any of themajor mail order parts houses and manysmall ones. Basically, it is a variation on theLM-1490 and LM-1590 circuit but a little eas-ier to apply.

If you have difficulty locating MC-1350Pdevices, the exact same chip is available inthe service replacement lines such as ECGand NTE. These parts lines, sold at local elec-tronics parts distributors, are intended for ser-vice repair shops. I used actual MC-1350Pchips in one version and NTE-746 (same asECG-746) chips in the other, with no differ-ence in performance. The NTE and ECGchips actually are purchased from the sourcesof the original devices and renumbered.

The circuit is shown in Figure 10.5. TwoMC-1350P devices in cascade are used. Eachdevice has a differential input (pins 4 and 6).These pins are connected to the link wind-ings on IF transformers (e.g., T2 at device U1and T3 at U2). In both cases, one input pin isgrounded for AC (i.e., RF and IF) signalsthrough a bypass capacitor (C2 and C4).

In the past, I had difficulty applying theMC-1350P devices when two were used in cas-cade. The problem is that these are high-gainchips and any coupling causes oscillation. Ibuilt several good MC-1350P oscillators—un-fortunately, I was trying to build IF amplifiers.The problem was solved by two tactics that Ihad ignored in the past. This time I reversedthe connections to the input terminals on thetwo devices. Note that pin 4 is bypassed toground on U1, while on U2 it is pin 6. Theother tactic is to use different value resistors atpin 5.

Pin 5 is the gain control pin on the MC-1350P device. It provides either automatic ormanual gain control (MGC). The voltage ap-plied to this pin should be between +3 and+9 V, with the highest gain being at +3 V andnearly no gain at +9 V.

The outputs of the MC-1350P are con-nected to the primary windings of T3 and T4.Each output circuit has a resistor (R2 and R5)across the transformer winding. The trans-formers used are standard “transistor radio”IF transformers provided that the impedancematching requirements are met.

The DC power is applied to the MC-1350P devices though pins 1 and 2, which


Fig. 10.4 Cascode F/IF amplifier.

J1INPUT L1 L2 C1

Q1

Q2

C40.001

µFR1100

C3NEUTRALIZATION

C70.001 µF

R2100K

C2L3

L4

J2OUTPUT

C60.001 µF

R3270

+12VDC

are connected together. Bypass capacitors C3and C5 decouple the DC power lines andthereby prevent oscillation. All the bypass ca-pacitors (C2, C3, C4, and C5) should bemounted as close to the bodies of U1 and U2as possible. They can be disk ceramic de-vices or some of the newer dielectric capaci-tors, provided, of course, that they are ratedfor operation at the frequency you select.Most capacitors will work to 10.7 MHz, but ifyou go to 50 MHz or so, some capacitortypes might show too much reactance (diskceramic types work fine at those frequencies,however).

The DC power supply should be regu-lated at some voltage between +9 and +15VDC. More gain can be obtained at +15 VDC,but I used +10 VDC with good results. Eachpower line has a 100 Ω resistor (R3 and R6),which helps provide some isolation betweenthe two devices. Feedback via the power lineis one source of oscillation in high-frequencycircuits.

The DC power line is decoupled by twocapacitors. The C7, a 0.1 µF capacitor, decou-ples high frequencies that either get inthrough the regulator or try to couple fromchip to chip via the DC power line (which iswhy they are called decoupling capacitors).The other capacitor (C8) is a 10–100 µF de-vice used to smooth out any variations in theDC power or decouple low frequencies thatthe 0.1 µF does not take out effectively.

The RF/IF input circuit deserves somecomment. I elected to use a double-tunedarrangement. This type of circuit is coupledvia a mutual reactance. Various versions ofthis type of circuit are known, but I electedto use the version that uses a capacitive reac-tance (C1) at the “hot” end of the L-C tankcircuits. Coupling in and out of the networkis provided by the transformer couplinglinks.

The power and gain control connec-tions are brought through the aluminum boxwall via 1000 pF feedthrough capacitors, two


Fig. 10.5 Universal IF amplifier.

++

GAIN CONTROLVOLTAGE INPUT

INPUTT1 T2

C16 pF

U1MC-1350P

U2MC-1350P

T3

T4OUT

C20.1 µF

C30.1 µF

C40.1 µF

C50.1 µF

C647 µF

+12 VDC U378L10

C70.1 µF

C810 µF

R110K

R21K

R3100

R422K

R52.2K

R6100

10 VDC LINE

6

4

2 1

3 75

8

6

42 1

3 75

8

kinds of which are available, one solder-inand the other screw-thread mounted. Foraluminum boxes, the screw thread is neededbecause it is difficult to solder to aluminum.Both types are available at either 1000 or2000 pF values, either of which can be usedin this application. If you elect to use someother form of connector, then add disk ce-ramic capacitors (0.001 µF) to the connector,right across the pins, as close as possible tothe connector.

There are several ways to make this cir-cuit work at other frequencies. If you want touse a standard IF frequency up to 45 MHz orso, then select one with the configurationshown in Figure 10.5 (these are standard).

If you want to make the circuit operatein the HF band on a frequency other than10.7 MHz, then use the 10.7 MHz trans-former. If the desired frequency is less than10.7 MHz, add a small-value fixed or trimmercapacitor in parallel with the tuned winding.This will add to the built-in capacitance, re-ducing the resonant frequency. I do notknow how low you can go, but I have hadgood results at the 40 m amateur band (7–7.3MHz) using additional capacitance across a10.7 MHz IF transformer.

Frequencies higher than 10.7 MHz callfor more drastic action. Take one of thetransformers and turn it over so that youcan see the pins. In the middle of the bot-tom header, between the two rows of pins,will be an indentation containing the tuningcapacitor. It is a small tubular ceramic ca-pacitor (you may need a magnifying glassto see it well if your eyes are like mine). Ifit is color coded, then you can obtain thevalue using your knowledge of the stan-dard color codes. Take a small screwdriverand crush the capacitor. Clean out all thedebris to prevent shorts at a later time. Younow have an untuned transformer with aninductance right around 2 µH. Using this in-formation, you can calculate the requiredcapacitance:

(10.1)

where

C is the capacitance in picofarads (pF);f is the desired frequency in kilohertz

(kHz);LµH is the inductance in microhenrys

(µH).

Equation 10.1 is based on the standardL-C resonance equation, solved for C, andwith all constants and conversion factorsrolled into the numerator. If you know thecapacitance that must be used and need tocalculate the inductance, then swap the Land C terms in equation 10.1.

If the original capacitor was marked asto value or color coded, then you can calcu-late the approximate capacitance needed bytaking the ratio of the old frequency to thenew frequency and squaring it. The squareof the frequency ratio is the capacitance ra-tio, so multiply the old capacitance by thesquare of the frequency ratio to find the newvalue. For example, suppose a 110 pF capac-itor is used for 10.7 MHz and you want tomake a 20.5 MHz coil. The ratio is (10.7MHz/20.5 MHz)2 = 0.272. The new capaci-tance will be about 0.272 × 110 pF = 30 pF.For other frequencies, you might considerusing homebrew toroid inductors.

A variation on the theme is to make thecircuit wideband. This can be done for awide portion of the HF spectrum by remov-ing the capacitors from the transformers andnot replacing them with some other capaci-tor. In that case, IF filtering is done at the in-put (between the IF amplifier and the mixercircuit).

COUPLING TO OTHER FILTERS

Crystal and mechanical filters require certaincoupling methods. Figure 10.6 shows amethod for coupling to a crystal filter con-nected between two bipolar transistors.Each stage of the amplifier is a commonemitter bipolar transistor amplifier, biasedby R1/R2 and R5/R6. The connection to thefilter circuit is direct because the filter is not

Cf L

=×2 53 1010

2

.

µH

picofarads


sensitive to DC (such cannot be said of me-chanical filters).

Figure 10.7 shows a different approachthat accommodates mechanical filters as wellas crystal filters. The particular circuit shownis for very high-frequency IF amplifiers (e.g.,50 MHz); but with changes to the values ofthe components, this IF amplifier could beused from VLF through VHF regions. Theresonant frequency of this circuit is set by L1and C1 and the filter circuit. The amplifier isa MOSFET device connected in the commonsource configuration. Gate G1 is used for thesignal, and G2 is used for DC bias and gaincontrol.

The filter is connected to the filterthrough a capacitor to block the DC at thedrain of the MOSFET device (similar capaci-tors would be used in bipolar circuits aswell). The output of the filter may or may notbe capacitor coupled, depending on the de-sign of the circuits to follow this one.

IC IF AMPLIFIERS

The universal IF amplifier presented earlier isan example of an integrated circuit IF ampli-fier. In this section, we look at several addi-tional IC IF amplifier circuits.

MC-1590 Circuit

Figure 10.8 shows an amplifier based on the1490 or 1590 chip. This particular circuitworks well in the VHF region (30–80 MHz).Input signal is coupled to the IC through ca-pacitor C1. Tuning is accomplished by C2and L1, which form a parallel resonant tankcircuit. Capacitor C3 sets the unused differen-tial input of the 1590 chip to ground poten-tial, while retaining its DC level.

Output tuning in Figure 10.8 reflectsthe differential output as well as differentialinput of the 1590 chip. The L-C tuned circuit,consisting of the primary winding of T1 and


Fig. 10.6Coupling to a crystalfilter between twobipolar transistors.

FL1

FROMMIXER

C1

C2

C3

C4TO

DEMODULATOR

Q1

Q2

R1

R2

R3

R4

R5

R6

R7

R8

V+

Fig. 10.7Mixer postamplifiercircuit.

FROMMIXER

V+

FILTERC1

(SEE TEXT)

RFC1

L1(SEE TEXT)

C215 pF

AGC

R1150 C3

0.01 µF

C40.01 µF

R2220K

R3100K

RFC2

IFOUTPUT

Q1G1

G2

S

D

capacitor C6, is parallel resonant and con-nected between pins 5 and 6. A resistoracross the tank circuit reduces its loaded Q,which broadens the response of the circuit.

V+ power is applied to the chip boththrough the V+ terminal and pin 6 throughthe coil L2. Pin 2 is used as an AGC terminal.

SL560C Circuits

The SL560C is basically a gain block that can beused at RF and IF frequencies. Figure 10.9shows a circuit based on the SL560C. The inputof the SL560C is differential, but this is a single-ended circuit. That requires the unused inputto be bypassed to ground through capacitorC3. Because this is a wideband circuit, no tun-ing is associated with the input or output cir-cuitry. The input circuitry consists of a 0.02 µFcoupling capacitor and an RF choke (RFC1).

A tuned-circuit version of the circuit isshown in Figure 10.10. This circuit replaces

the input circuitry with a tuned circuit (T1)and places a transformer (T2) in the outputcircuit. Also different is that the V+ circuit usesa zener diode to regulate the DC voltage.

IF PROCESSING ICs

Several forms of IC are used for IF process-ing. The CA3189E is one used in broadcastand communications receivers. The input cir-cuitry consists of a filter, although L-C tunedcircuits could be used as well. In this version,the filter circuit is coupled via a pair of ca-pacitors (C1 and C2) to the CA3189E. The in-put impedance is set by resistor R1 andshould reflect the needs of the filter ratherthan the IC (filters do not produce the sameresponse when mismatched).

The CA3189E is an IF gain block anddemodulator circuit, all in one IC. Coil L1and C6 is used for the quadrature detector,


Fig. 10.8 MC1350P IF/RF amplifier.

U1MC-1590

1

3

2 4 8

76

5OUTPUT

INPUT

AGC

C122 pF

C210 pF

C30.02 µF

L1200 µH

C40.01 µF

V+

C50.01 µF

C610 pF

T1

L210 µH

R15.6K

R21K


Fig. 10.9 SL-560C IF amplifier/postamplifier.

U1SL560C

FROMMIXER

RFC1

C10.01 µF

C20.01 µF

C30.01 µF

C10.01 µF

C10.1 µF

R1

R2390

R333

OUTPUT

6

7

1

4

3

V+

Fig. 10.10 SL-560C IF amplifier with tuning.

U1SL560C

FROMMIXER

RFC1

C10.01 µF

T1

C20.01 µF

C30.01 µF

R110K

D16.8V

C40.1 µF

C510 µF

+

R2220K

V+

C60.01 µF

R233

T2

OUTPUT

6

7

1

4

3

Fig. 10.11 CA3189E IF subsystem.

U1CA3189E

FL1FROMMIXER

C10.01 µF

C20.01 µF

C30.01 µF

+C410 µF

V+

C50.01 µF

RFC1

L1(SEE TEXT)

C6(SEE TEXT)

+

C710 µF

C80.01 µF

C910 µF

+

AFC

AUDIOOUTPUT

R1470

R210 R3

3.9K

R46.8K

R53.9K

1

3 2 4 14

6

711 8 9 10

R668K

R720K

M1100 µA

13

and their value depends on the frequencyused. Three outputs are used on theCA3189E. The audio output is derived fromthe demodulator, as is the automatic fre-quency control output. Also, a signal strengthoutput can be used to drive an S-meter (M1)or left blank.

SUCCESSIVE DETECTIONLOGARITHMIC AMPLIFIERS

Where signal level information is required orinstantaneous outputs are required over awide range of input signal levels, a logarith-mic amplifier might be used. Linear amplifiershave a gain limit of about 100 dB but head

room of only 3–6 dB in some circumstances.One solution to the problem is the logarithmicamplifier. Radar receivers frequently use logamps in the IF amplifier stages.

The successive detection method isused because it is difficult to produce high-gain logarithmic amplifiers. The successivedetection method uses several log amps,then detects all outputs and totals them. Eachamplifier has an output voltage equal to

(10.2)

where

Vo is the output voltage;Vin is the input voltage;k is a constant.

V k Vo = log in


Fig. 10.12 Logarithmic amplifier: (A) circuit; (B) gain curve.

OU

TP

UT

(LI

NE

AR

)

INPUT LEVEL

DETECTOR DETECTOR DETECTOR DETECTOR

S OUTPUT

INPUT

AMPLIFIER AMPLIFIER AMPLIFIER AMPLIFIER

A

B

The circuit shown in Figure 10.12A issuch an amplifier; Figure 10.12B shows a hy-pothetical output vs. input characteristic forthe amplifier. Because the circuit uses four ormore stages of nonlinear amplification, sig-nals are amplitude compressed in this ampli-fier. Therefore, the amplifier sometimes iscalled a compression amplifier.

FILTER SWITCHING IN IF AMPLIFIERS

Switching filters is necessary to accommodatedifferent modes of transmission. AM requires4 or 6 kHz on shortwave and 8 kHz on theAM broadcast band (BCB). Single-sidebandrequires 2.8 kHz, RTTY/RATT requires 1.8kHz, and CW requires either 270 or 500 Hz.Similarly, FM might require 150 kHz for theFM BCB and as little as 5 kHz on landmobilecommunications equipment. These various

modes of transmission require different filters,and those filters have to be switched in andout of the circuit.

The switching could be done directly,but that requires either a coaxial cable be-tween the switch and the filter or placing theswitch at the site of the filters. A better solutionis found in Figure 10.13: diode switching. Adiode has the unique ability to pass a small ACsignal on top of the DC bias. In Figure 10.13,switch S1 is used to apply the proper polarityvoltage to the diodes in the circuit. In onesense of S1, a positive voltage is applied to D4through the primary winding of T2, throughRFC4 and RFC2, to D3, and then through thesecondary winding of T1 to ground. This cur-rent flow reverse biases diodes D1 and D2.The response is to turn on filter FL2. Similarlywhen S1 is turned to the other position—inthat case D1 and D2 are forward biased, se-lecting FL1, and D3–D4 are reverse biased.


Fig. 10.13 Diode switching circuit.

FL1FILTER No. 1

FL2FILTER No. 2

INPUT OUTPUT

T2T1RFC1

RFC2

RFC3

RFC4

R1 R2

R3 R4

C1 C2

C3 C4

R5

C5 C6

C7

C8

C9

C10

R7R6

D1 D2

D4D3

S1A S1B

V+

The purpose of detector or demodulator cir-cuits is to recover the intelligence impressedon the radio carrier wave at the transmitter.The process is called demodulation, and thecircuits used to accomplish this are calleddemodulators. They also are called seconddetectors in superheterodyne receivers.

In a superheterodyne receiver, the detec-tor or demodulator circuit is placed betweenthe IF amplifier and the audio amplifier(Figure 11.1). This position is the same inAM, FM, pulse modulation, and digital re-ceivers (although in digital receivers the de-modulator might be in a circuit called amodem).

AM ENVELOPE DETECTORS

An amplitude modulation signal consists of aslow audio signal that revolves around an av-erage radio frequency carrier signal. Both theRF carrier and AF signals are output in essen-tially a multiplication or mixing process,along with the (RF – AF) and (RF + AF) sig-nals. Because of the selectivity of the trans-mitter circuits, only the RF carrier and thesum and difference signals appear in the out-put. The AF signal is suppressed. The sumsignal (RF + AF) is known as the upper side-band (USB), while the difference signal (RF –AF) is known as the lower sideband (LSB).

Chapter 11

Detector and Demodulator Circuits

175

Fig. 11.1Position of the de-modulator circuit.

IF AMPLIFIERDEMODULATOR

CIRCUITAUDIO

AMPLIFIERS

Because of this action, the bandwidth of theAM signal is determined by the highest audiofrequency transmitted and is equal to twicethat frequency. A total of 66.67% of the RFpower in an AM signal is in the carrier, soonly 33.33% is split between two sidebands.

Figure 11.2 shows a simple AM enve-lope detector circuit; Figure 11.3 shows thewaveforms associated with this circuit. Thecircuit consists of a signal diode rectifier con-nected to the output of an IF amplifier. Thereis a capacitor (C1) and resistive load con-nected to the rectifier. When the input signalis received (Figure 11.3A), it is rectified(Figure 11.3B), producing an average currentoutput that translates to the voltage wave-form of Figure 11.3C.

The capacitor is charged to a value equalto 0.637 times the peak voltage (which is theaverage value), then scaled upward withmodulation to the full peak value. Low-passfilter R1C2 takes out the residual RF/IF signal.

The important attribute of a demodula-tor is for it to have a nonlinear response,preferably with a sharp cutoff. Vacuum tubes,bipolar transistors, and field effect transistorspossess these characteristics. So does the sim-ple diode rectifier. Figure 11.4 shows the in-put and output characteristics compared withthe diode’s I vs. V curve. At low signal levels,the diode acts like a square law detector, butat higher signal levels, the operation is some-what linear.

Consider a diode with an I vs. V charac-teristic of

(11.1)

Let

(11.2)

We then can write

(11.3a)

(11.3b)

where

m is the modulation index;fc is the carrier frequency;A is the peak amplitude;a0, a1, and a2 are constants.

... ( ) ( )

[ ( )] cos ( )

+ +

+ +

a A ms ta A

m S t

a Ams t f tc

22 2

2

22

22

2

2

21 4π

I a a A ms t f t

a A

d c= + +

+ +

0 1

22

1 2

2

[ ( )] cos ( )

..

π

.

V A ms t f td c= +[ ( )] cos ( )1 2π

I a a V a Vd o d d= + +1 22


Fig. 11.2Simple envelope detector.

FROMIF

AMPLIFIEROUTPUT

T1D1

C1

R1

C2

Fig. 11.3 Waveforms of envelope detector.

INPUT

RECTIFIED OUTPUT

FILTERED OUTPUT

A

B

C

The terms in the brackets are the mod-ulation and distortion products. The second-order terms are modulation and distortion,where the higher-order terms are distortiononly. What falls out of the equation is that, tokeep distortion low, the modulation index(m) must be low as well.

Figure 11.5 shows what happens at thecapacitor. The dashed lines represent theoutput of the diode, which is a half-waverectified RF/IF signal; the heavy line repre-sents the capacitor voltage (Vc). The capaci-tor charges to the peak value, then the diodecuts off. The voltage across the capacitor

Detector and Demodulator Circuits 177

Fig. 11.4 Waveforms of envelope detector superimposed on diode characteristic.

INPUT VOLTAGE

INPUTENVELOPE

DIODECHARACTERISTIC

OUTPUT

drops slightly to the point where its voltageis equal to the input voltage, then it turns onagain. The diode may be modeled as aswitch with resistance. During the noncon-duction period, the switch is open and thecapacitor discharges. During the conductionperiod, the “switch” conducts and chargesthe capacitor. The waveform across the ca-pacitor when there is no modulation is closeto a sawtooth at the carrier frequency (fc); itrepresents the residual RF. These are elimi-nated in the R-C filter to follow the envelopedetector (R1C2) and the response of the am-plifiers to follow the detector. These compo-nents typically are about 30 dB below thecarrier level.

The maximum time constant of the fil-tering action of C1, plus (R1 + R), where R isthe resistance to ground (typically a volumecontrol) that can be accommodated, dependson the maximum audio frequency that mustbe processed. A frequency of 3000 Hz and atime constant of 10 µS yields

(11.4)

That produces an output reduction of

(11.5)

which is 0.16 dB. At 10 kHz, these values are0.628 and 1.18, or 1.44 dB.

Figure 11.6 shows a version of the en-velope detector that uses a high-pass filter atthe output. The time constant just referred tois the time constant R1C1 and eliminates theresidual RF/IF signal. The time constantR2C2 is set to eliminate low-frequency humand noise. The speech requirement in com-munications receivers is 300–3000 Hz, so a60 Hz hum is easily accommodated. Forbroadcast receivers, the low-frequency au-dio is on the order of 100 Hz, so this methodworks less well.

In the linear mode of operation of thediode, the modulation index for which dis-tortion begins is

(11.6)

The distortion will be small if

(11.7)

where Zm is the impedance of the circuit at themodulating frequency. Equation 11.6 reducesto this form when Rd << R1 and Zm = R2.

Figure 11.7 shows three situations of in-put waveform and output waveform. Figure11.7A shows the situation with no distortionof the output waveform. Although this wave-form never occurs in real circuits, it is in-cluded for comparison. In Figure 11.7B, thecircuit clips the negative peaks of the modu-lating signal. In Figure 11.7C, we see anexample of diagonal clipping. This form ofclipping occurs when Zm is not resistive.

Another form of distortion occurs whenthe RF/IF waveform is distorted. Both in-phase and quadrature distortion can occur inthe RF/IF waveform, especially if the band-

Z

Rmm

1<

m

R R R R R

R R R Rd

d

= + ++ +

( )

( )( )

1 2 1 2

1 2 1

1 2 1 0182+ =( ) .πf Tm c

2 2 3000 10 0 18845π πf Tm c = × × =− s .


Fig. 11.5 Filtered output waveform.

TIME

VO

LTA

GE

Fig. 11.6Envelope detectorwith a high-pass filterat the output.

FROMIF

AMPLIFIEROUTPUT

T1D1

C1 R1

C2

R2

pass filters used in the RF/IF circuit havecomplex poles and zeros distributed asym-metrically about the filter center frequency orthe carrier is not tuned to the exact center ofthe filter passband.

NOISE

All radio reception is basically a game scoredby the signal-to-noise ratio. If the SNR is notadvantageous, then the receiver ultimately

fails. At low signal levels (Figure 11.8), theSNR is poor and the noise dominates the out-put. As the signal level increases, however,the noise level increases but at a slower ratethan the signal output. The noise level comesto rest (see dashed line in Figure 11.8) about3.7 dB over the no-carrier state, but the sig-nal level continues upward.

BALANCED DEMODULATORS

Figure 11.9 shows two examples of balanced,or full-wave, demodulator circuits. These cir-cuits generally work better than the half-waveversion shown earlier. Figure 11.9A shows thecircuit for a conventional full-wave circuit. Itdepends on two diodes and a center tappedtransformer (T1), working like the powersupply circuit of the same type. The centertap on the transformer establishes the zero, orcommon, point, so the polarities of the volt-ages at the ends of the secondary winding ofT1 are equal but opposite. When the top ofthe secondary winding is positive with re-spect to the bottom, diode D1 conducts andcharges capacitor C1. On the opposite halfcycle, the opposite occurs: The secondarywinding has reverse polarity, so the bottom ismore positive than the top. This turns off D1and turns on D2, causing it to conduct andcharge C1.

Figure 11.9B shows a bridge-rectifiedversion of the AM envelope detector. Whenthe top of the T1 secondary winding is posi-


Fig. 11.7 Waveforms: (A) perfect; (B) clipping ofnegative peaks; (C) diagonal clipping.

Fig. 11.8 AC power vs. input power.

INPUT POWER

AC

PO

WE

R O

UT

PU

T

NOISE OUTPUT

OUTPUTFOR 50% AM

A

B

C

tive with respect to the bottom, diodes D2 andD3 conduct, charging C1. When the bottom ofthe T1 secondary winding is positive with re-spect to the top, diodes D1 and D4 conduct,charging the capacitor. Since these conditionsoccur on alternate half cycles of the inputwaveform, full-wave rectification occurs.

SYNCHRONOUS AMDEMODULATION

A factor that affects the comfort level of lis-tening to demodulated AM transmissions isthe fading out of phase with each other of thecarrier and two sidebands. This can be over-come with quasi-synchronous demodulationor synchronous demodulation. Both requirethat the incoming carrier be eliminated. Thedifference is that in quasi-synchronous de-modulation the reinserted carrier is not inphase with the original, whereas in synchro-nous demodulation it is.

Quasi-synchronous demodulation ismuch like the demodulation of single-sideband

demodulators, discussed next. These have abeat frequency oscillator to replace the car-rier with an out-of-phase version. As long asthe signal does not drop to zero, the processworks fine. But when the signal drops tozero, as it does under deep fading, theprocess falls down and synchronous demod-ulation wins. In synchronous demodulation,the reinserted carrier is in phase with theoriginal carrier signal. The circuit must phaselock to the original carrier.

DOUBLE-SIDEBAND AND SINGLE-SIDEBAND SUPPRESSED CARRIERDEMODULATORS

Double- and single-sideband suppressed carri-ers are a lot more efficient than straight AM. Instraight AM, the carrier contains two thirds ofthe RF power, with one third split between thetwo sidebands. Interestingly enough, the entireintelligent content of the speech waveform isfully contained within one sideband, so theother sideband (and carrier) is superfluous.


Fig. 11.9 Full-wave envelope detector: (A) conventional; (B) bridge rectified.

FROM IFAMPLIFIER

R1

D1 D2T1

C1MODULATED

OUTPUT

D3D4

FROM IFAMPLIFIER

R1 C1MODULATED

OUTPUT

D1

D2

T1

A

B

Figure 11.10 shows a single-sidebandsuppressed carrier (SSBSC, usually shortenedto SSB) transmitter. The heart of the circuit is abalanced modulator circuit. This circuit is bal-anced to produce RF output from the crystaloscillator only when the audio frequency (AF)signal is present. As a result, the carrier is sup-pressed in the double-sideband output.

The next stage is a symmetrical band-pass filter circuit that removes the un-wanted sideband, leaving only the desiredsideband. The crystal oscillator determineswhich sideband is generated. By position-ing the frequency of the oscillator on thelower skirt of the filter, an upper sidebandsignal is generated. By positioning the fre-quency of the oscillator on the upper skirtof the filter, the lower sideband is gener-ated. This is shown in Figure 11.11. In thisfigure, Fc is the center frequency of the fil-

ter and the LSB and USB frequencies areshown.

Following the filter circuit is any amplifi-cation or frequency mixing needed to accom-plish the purposes of the transmitter. Contraryto AM transmitters, all stages following thebalanced modulator are expected to be linearamplifiers. This is because nonlinear stagesdistort the envelope of the SSB signal. Thismeans that heterodyning must be used totranslate the frequencies, rather than multipli-ers or other nonlinear means. Generally, SSBtransmitter designers generate the SSB signalat a fixed pair of frequencies, then heterodynethem to the desired operating frequency.There also is a phasing method of generatingan SSB signal. Double-sideband suppressedcarrier (DSBSC, usually shortened to DSB)transmitters are the same as Figure 11.10, ex-cept that the filter is not present.

The basis for SSB and DSB demodulationis the product detector circuit. Figure 11.12shows basis for product detection. In Figure11.12A, we see the SSB or DSB signal with thecarrier suppressed (it actually is a DSB signal,but an SSB signal would lack the other side-band). This signal is combined with a stronglocal oscillator signal (called a beat frequencyoscillator, BFO), Figure 11.12B, to produce thebaseband signal (Figure 11.12C).

Figure 11.13 shows a simple form ofproduct detector circuit. It is like the enve-lope detector except for the extra diode (D2)and a carrier regeneration oscillator (alsoconsidered a BFO). The circuit works by


Fig. 11.10 SSB transmitter using the filter method.

CRYSTALOSCILLATOR

Y1LSB

XTAL

Y2USBXTAL

BALANCEDMODULATOR

AUDIOAMPLIFIERSMICROPHONE

RF

AF

FILTER AMPLIFIER RFOUTPUT

RF RF

Fig. 11.11 USB and LSB crystal locations on thefilter response.

Fc

USBCRYSTAL

FREQUENCY

LSBCRYSTAL

FREQUENCY

FREQUENCY

AM

PLI

TU

DE

switching the diodes into and out of conduc-tion with the oscillator signal. When the out-put of the oscillator is negative, the diodesconduct, passing the signal to the output.But, on positive excursions of the local oscil-lator signal, the diodes are blocked from con-ducting by the bias produced by theoscillator signal. The residual RF/IF signal isfiltered out by capacitor C1.

A superior circuit is shown in Figure11.14. The circuit is a balanced product de-tector. It consists of a balanced ring demodu-lator coupled through a pair of center tappedtransformers (T1 and T2). Transformer T1 isthe last IF transformer. It has a center-tappedsecondary winding to receive the strong localoscillator signal used for carrier regeneration.The output transformer (T2) also is centertapped, but the center tap is grounded.

The circuit works by switching in andout of the circuit pairs of diodes on alter-nate half cycles of the SSB waveform.Consider first circuit action when no signalis present. That leaves only the local oscil-lator signal, which is applied to the com-mon point on the transformer T1 secondary


Fig. 11.12 Single-sideband demodulation.

LSB USB

CARRIER(SUPPRESSED)

SSB OR DSBSIGNAL

LOCALOSCILLATOR

BASEBANDSIGNAL

A

B

C

Fig. 11.14Balanced product detector.

FROM IFAMPLIFIER

D1

D2

T1

CARRIERREGENERATION

OSCILLATOR

MODULATEDOUTPUT

T2D3

D4

Fig. 11.13Simple product detector.

FROM IFAMPLIFIER

D1 D2T1

R1 C1MODULATED

OUTPUT

CARRIERREGENERATION

OSCILLATOR

winding. That means that the ends of thetransformer will be positive at the sametime and negative at the same time. Underthis circumstance, the diodes D1 and D4conduct, then diodes D2 and D3 conduct,but both signals are nulled in the primarywinding of T2. This occurs because bothsignals are the same.

When the top of the T1 secondarywinding is positive with respect to the bot-tom and the local oscillator signal is positive,diodes D1 and D2 conduct, creating an un-balanced situation, which results in output.Similarly, when the situation is reversed, onlythe D2 and D3 diodes conduct and the othersare cut off. Thus, the local oscillator controlsthe output of the product detector.

Figure 11.15 shows a circuit in which abipolar transistor is used as a product detec-tor. The circuit would act like any amplifier,except that the base of the transistor is con-trolled by the local oscillator. The SSB signalis applied through the low-impedance sec-ondary winding of T1 and ordinarily wouldbe amplified by Q1. But the quenching ac-tion of the local oscillator prohibits this ac-tion. The transistor is alternately cut off andcut on by the local oscillator circuit, and thatcreates the nonlinearity needed to demodu-late the waveform.

A differential pair of junction field effecttransistors is used in Figure 11.16 to producethe product detection. The SSB signal is ap-plied to the gate of Q1, while the local oscil-lator signal is used to disrupt the operation ofthe circuit from the Q2 side. Keep in mindthat the local oscillator signal is very muchgreater in amplitude than the SSB signal. Alow-pass filter tuned to the spectrum that is tobe recovered (typically audio) is connected tothe common drain circuits and thence to theoutput.

A dual-gate MOSFET transistor is thesubject of Figure 11.17. The normal signalinput of the MOSFET, Gate 1, is used to re-ceive the SSB signal from the IF amplifier.The MOSFET is turned on and off by the lo-cal oscillator signal applied to the Gate 2.Again, an audio low-pass filter is present atthe output (drain) circuit to limit the resid-ual IF signal that gets through to the modu-lated output.

Note, here, the two capacitors in thesource circuit of the MOSFET transistor (C1and C2). Typically, one of these will be forRF and the other for AF, although withmodern capacitors it might not be strictlynecessary.

An integrated-circuit SSB product de-tector is shown in Figure 11.18. This circuit


Fig. 11.15 NPN transistor product detector.

FROM IFAMPLIFIER

T1

CARRIERREGENERATION

OSCILLATOR

MODULATEDOUTPUT

+15 VOLTS

R1

R2

R3

R4C1

10 pF

C2

C3

C4Q1

is based on the MC-1496 analog multiplierchip. It contains a transconductance celldemodulator that is switched on and off bythe action of the local oscillator. The SSB IFsignal is input through pin 1, and the localoscillator through pin 10. The alternatepins in each case are biased and not other-wise used.

PHASING METHOD

The methods for demodulating an SSB orDSB signal thus far presented have beenproduct detectors. A phasing method also isused in some cases. Figure 11.19 shows thismethod in block diagram form. This methodsplits the SSB IF signal into two paths, I and


Fig. 11.16 JFET balanced product detector.

FROM IFAMPLIFIER

MODULATEDOUTPUT

CARRIERREGENERATION

OSCILLATOR

T1 T2

+15 VOLTS

LOW-PASSFILTER

Q1 Q2

RFC1

R1

R2

Fig. 11.17 Dual-gate MOSFET product detector.

FROM IFAMPLIFIER

CARRIERREGENERATION

OSCILLATOR

T1

LOW-PASSFILTER

+15 VOLTS

R2

MODULATEDOUTPUT

G2

G1

S

D

R1

C1

R3C1

C2

Q1

Q. They are mixed with a pair of signals thatare the same except for phasing. The I signalis mixed with the cos (2πnFc/F) version ofthe local oscillator signal, while the Q ismixed with the −sin (2πnFc/F) version of thesame signal. The Q channel is passed througha Hilbert transformer, which further phaseshifts it by 90º. The I channel signal is de-layed an amount equal to the delay of the Qchannel signal, (N − 1)/2 samples. The twosignals then are summed in a linear mixer cir-cuit to produce the output. This method willyield the lower sideband part of the signal. Ifwe subtract the two signals instead of addingthem, we get the upper sideband signal.

FM AND PM DEMODULATOR CIRCUITS

Frequency modulation and phase modula-tion are examples of angle modulation.

Figure 11.20 shows this action graphically.The audio signal causes the frequency (orphase) to shift plus and minus from the qui-escent value, which exists when no modu-lation is present. Frequency and phasemodulation are different but similar enoughto make the demodulation schemes thesame. The difference between FM and PMis that the phase modulation needs no pre-emphasis curve to the audio waveform (itdoes this naturally), while the frequencymodulation transmitter is pre-emphasizedfor noise abatement.

Another difference between FM and PMtransmitters is the location of the reactancemodulator used to generate the modulatedsignal. In the FM transmitter, the reactancemodulator is part of the frequency determin-ing circuitry; in the PM transmitter, it followsthat circuitry.


Fig. 11.18 LM-1496/MC-1496 product detector.

U1MC-1496MC-1596

CARRIERREGENERATION

OSCILLATOR

FROM IFAMPLIFIER

LOW-PASSFILTER

+15VOLTS

MODULATEDOUTPUT

R1

R2

R3

R4R5 R6

R7

R8 R9 R10

C1

C2

C3

C4

C5

14

4

1

10

82 3

6

12

5

Fig. 11.19 Phasing method of demodulation.

BANDPASSFILTER

HILBERTTRANSFORMER

S

S

COS (2pnFc /F)

-SIN (2pnFc /F)+ LOW-PASS

FILTERMODULATED

OUTPUTFROM

IFAMPLIFIER

DISCRIMINATOR CIRCUITS

A classic FM discriminator circuit is shown inFigure 11.21. This circuit uses a special trans-former that has two secondary windings.One secondary winding is tuned slightly

above the IF frequency, while the other istuned the same amount below the IF fre-quency. The two frequencies are spacedslightly more than the expected transmitterswing. Their outputs are combined in a dif-ferential pair of diodes (D1 and D2). The out-puts of the diodes are connected to loadresistors R1 and R2. Normally, when the sig-nal is unmodulated, the algebraic sum of thetwo diodes outputs is 0, resulting in no out-put. When the frequency or phase swingsabove the quiescent value, one diode willconduct higher than the other, resulting in animbalanced across R1 and R2. This producesoutput. Similarly, when the frequency orphase drops below the quiescent value, thenthe opposite situation exists. The other diodewill conduct harder and produces output ofthe opposite polarity.

The Foster-Seeley discriminator circuit isshown in Figure 11.22, and the waveforms areshown in Figure 11.23. This circuit requiresonly two tuned circuits, rather than the threerequired by the previous circuit. The outputvoltage is the algebraic sum of the voltagesdeveloped across the R2 and R3 load resis-tances. Figure 11.23A shows the relationshipof the output voltage and the frequency.

The primary tuned circuit is in serieswith both halves of the secondary winding.When the signal is unmodulated, the IF volt-age across the secondary winding is 90º out ofphase with the primary voltage. This makes


Fig. 11.20 Frequency modulation.

FREQUENCY

RF CARRIER FREQUENCY

AUDIOMODULATINGWAVEFORM

Fig. 11.21 Double-tuned FM detector.

MODULATEDOUTPUT

T1

D1

D2

C30.01 µF

FROM LASTLIMITERSTAGE

R1100K

R2100K

C10.001 µF

C20.001 µF

the voltages applied to each diode equal butout of phase (Figure 11.23B), resulting in nooutput. But consider what happens when thefrequency deviates (Figure 11.23C). The volt-

ages applied to the diodes no longer are equalbut opposite, and that creates output from thedetector that is frequency or phase sensitive.As the input frequency deviates back andforth across the frequency of the tuned circuit,an audio signal is created equal to the modu-lated frequency.

For the FM/PM transmitter to be re-ceived “noise free,” it is necessary to precedethe discriminator circuit with a limiter circuit.This circuit limits the positive and negativevoltage excursions of the IF signal, clippingoff AM noise.

RATIO DETECTOR CIRCUITS

The ratio detector circuit is shown in Figure11.24. This circuit uses a special transformer,


Fig. 11.22 Foster-Seeley discriminator circuit.

MODULATEDOUTPUT

R1100K

R2100K

R3100K

T1

50 pFC1

0.001 µF

C20.001 µF

D1

D2

C30.01 µF

FROM LASTLIMITERSTAGE

`

Fig. 11.23 Phase relationships in an FM detector: (A) output voltage vs. frequency; (B) unmodulated sig-nal; (C) frequency deviation.

RR'

PR

IMA

RY

VO

LTA

GE

SECONDARY VOLTAGE

FREQUENCY

VO

LTA

GE

0,0

+V

-V

+F-F

R

R'

PR

IMA

RY

VO

LTA

GE

SECONDARY VOLTAGE

A

B C

in which a small capacitor lies between thecenter tap on the primary winding and thecenter tap on the secondary winding. Notethat the diodes are connected to aid eachother rather than buck each other, as was thecase in the Foster-Seeley discriminator cir-cuit. When the signal is unmodulated, thevoltage appearing across R3 is one half theAGC (automatic gain control) voltage ap-pearing across R2, because the contributionof each diode is the same. However, that sit-uation changes as the input signal is modu-lated above or below the center frequency.Then, the relative contribution of each diodechanges. The total output voltage is equal totheir ratio; hence, the name ratio detector.

A ratio detector has several advantagesover a Foster-Seeley discriminator. First, alimiter amplifier ahead of the ratio detector isnot needed, unlike the Foster-Seeley discrim-inator. Furthermore, the circuit provides anAGC voltage, which can be used to controlthe gain of preceding RF or IF amplifierstages. However, the ratio detector is sensi-tive to AM variations of the incoming signal,so the AGC should be used on the stage pre-ceding the ratio detector to limit those AMexcursions. The capacitor, C3, also helpseliminate the AM component of the signal,which is noise.

PULSE COUNTING DETECTORS

The FM/PM detectors thus far consideredhave required special transformers to makethem work. In this section, we look at aspecies of coilless FM detector. The pulsecounting detector is shown in Figure 11.25.This circuit uses two integrated circuits, ahex inverter and a dual J-K flip-flop. Thehex inverter has six inverter stages. The firststage is used as an amplifier, and the nexttwo to produce an output free of AM noise(most noise is AM). This is followed by apair of divide-by-2 (total divide-by-4) stagesconsisting of a pair of J-K flip-flops. An in-verter at the output of the flip-flops is usedto drive the input of a half-monostable mul-tivibrator that has a period equal to aboutone half the period of the unmodulated in-put signal. The output of the half-mono-stable multivibrator is a time-varying pulsetrain that varies with the audio modulationapplied to the input signal. It is realized asaudio in the low-pass filter consisting of R3and C2.

Another circuit, shown in block form inFigure 11.26, uses a zero-crossing detectorand a limiter amplifier to eliminate the AMexcursions that are noise to an FM/PM signal.The output of the zero-crossing detector trig-


Fig. 11.24 Ratio detector circuit.

100 pF

FROM LASTLIMITER STAGE

MODULATEDOUTPUT

AGCOUTPUT

T1

R1100K

D1

D2

C10.001 µF

C20.001

µF

C310 µF

R2100K

C40.01 µF

R3100K

gers a monostable multivibrator circuit. Theoutput of the monostable multivibrator is apulse train that varies according the modulat-ing frequency applied. This is realized in alow-pass filter circuit.

PHASE-LOCKED LOOP FM/PMDETECTORS

The phase-locked loop circuit can be usedas an FM demodulator if its control volt-age is monitored. Figure 11.27A shows thebasic PLL circuit. It consists of a voltage-controlled oscillator (VCO), phase detector,low-pass filter, and amplifier. The FM signalfrom the IF amplifier is applied to one portof the phase detector, and the output of theVCO is connected to the other port. Whenthe two frequencies are equal, no outputemerges from the circuit (the value is quies-cent). When the FM IF signal deviates aboveor below the frequency of the VCO, an errorterm is generated. This error signal isprocessed in the low-pass filter and ampli-

fier to control the VCO. Its purpose is todrive the VCO back on the right frequency.This error signal becomes the modulatedoutput of the PLL FM demodulator circuit.Figure 11.27B shows a PLL based on the NE-565 PLL integrated circuit. The resonant fre-quency is set by R1 and C1, which shouldbe the center frequency of the FM signal. Asthe signal deviates up and down, the errorvoltage is monitored and becomes the mod-ulated output signal.

QUADRATURE DETECTOR

Figure 11.28 shows the quadrature detectorcircuit. This circuit is implemented in inte-grated circuit form (e.g., MC-1357P, CA-3189) and uses a single phase-shiftingexternal coil to accomplish its goals. Thisprobably is the most widely used form of FMdemodulator today.

The typical quadrature detector IC usesa series of wideband amplifiers to boost thesignal and limit it, eliminating the AM noise


Fig. 11.25 Coilless FM detector.

FROM IFAMPLIFIER

T Q T Q

+5 VOLTS

J-KFLIP-FLOP

J-KFLIP-FLOP

INVERTER INVERTER INVERTER

INVERTERINVERTERINVERTER

R110K

C10.001 µF

R26.8K

C2330 pF

R368K

C20.001

MODULATEDOUTPUT

Fig. 11.26 Coilless FM detector using a zero-crossing detector and a limiter amplifier.

LIMITERAMPLIFIER

MONOSTABLEMULTIVIBRATOR

FROM IFAMPLIFIER

ZERO-CROSSINGDETECTOR

LOW-PASSFILTER

MODULATEDOUTPUT

modulation that often rides on the signal.This signal is applied to the signal splitter.The two outputs of the signal splitter are ap-plied to a gated synchronous detector, but

one is phase shifted 90º. The output of thegated synchronous detector is the modulatedaudio.


Fig. 11.27PLL circuits:(A) basic;(B) based on theNE-565 PLL inte-grated circuit.

U1NE-565

FROM IFAMPLIFIER

R3560

C20.02 µF

R2560

C1

R120K

C30.001 µF

+5VOLTS

-5VOLTS

MODULATEDOUTPUT

2

3

1 9 4 5

10 87

6

114

1

CRf =

C4

q


LOW-PASSFILTER

AMPLIFIER

MODULATEDOUTPUT

FROM IFAMPLIFIER

PHASEDETECTOR

B

A

Fig. 11.28 Circuit from the CA-3189E FM detector.

FROM IFAMPLIFIER

SIGNALSPLITTER

GATEDSYNCHRONOUS

DETECTOR

90o

PHASESHIFTER

MODULATEDAUDIO OUTPUT

WIDEBANDAMPLIFIERS

In this chapter, we look at the other receivercircuits, including noise blankers and auto-matic gain control circuits.

NOISE BLANKERS AND LIMITERS

Noise tends to amplitude modulate signals,regardless of the mode type. Even “noisefree” FM will exhibit noise under the rightcircumstances if the receiver’s limiter thresh-old is not exceeded by the signal. There aretwo basic types of noise circuit: noise lim-iters and noise blankers. Noise limiters basi-cally are clipper circuits, because they clipthe amplitude of the signal. Noise blankerspoke a hole in the signal at the point wherethe noise occurs. Generally speaking, thedegradation of the intelligence this causes isless than the degradation caused by thenoise itself.

The simplest form of noise limiter cir-cuit is the peak noise limiter shown in Figure12.1. It consists of two batteries, back toback, with a pair of diodes. This particularcircuit works at a low level in the audiochain of the receiver. Signals are low leveland do not cause the diodes to conduct. But

impulse noise amplitude modulating the sig-nal does. The potentiometer (R1) adjusts thelevel at which clipping occurs.

Figure 12.2 shows the circuit for amore complex peak noise limiter circuit. Itconsists of an emitter follower or other low-gain audio frequency amplifier. The signalis input to the amplifier where it gains inpower but not in voltage. The output of the

Chapter 12

The Other Receiver Circuits

191

Fig. 12.1 Diode noise limiter circuit.

B2B1

S1A

AUDIOINPUT

AUDIOOUTPUT

S1B

R1CLIP

LEVEL

D1 D2

amplifier is applied to the noise limiter cir-cuit. Negative impulse noise is limited bydiode D1, while diode D2 limits the posi-tive pulses. The switch turns the limiter onand off.

The audio noise limiter works best inbroad IF bandwidth receivers, because narrowIF bandwidths tend to stretch out the impulses,rendering the noise limiter less effective.

IF Noise Limiters

IF noise limiting generally works better thanaudio limiting. Figure 12.3 shows a simple IFclipper circuit. Diodes D1 and D2 clip thenegative and positive impulses, respectively.The bias for the diodes is carried by capaci-tors C2 and C3, while capacitor C1 acts as a

DC blocking capacitor. Transistor Q1 turnsthe automatic noise limiter (ANL) on and off(when the switch is closed, the noise limiteris on).

Noise Blankers

Noise blankers poke a hole in the signal atthe point where the noise occurs. Figure12.4 shows a basic blanker circuit. TransistorQ1 serves as a noise amplifier and amplifiesboth signal and noise. Diodes D1 and D2serve as a negative impulse gate to triggertransistor Q2. The output of Q2 drives Q3into conduction, driving a hole in the outputsignal. The ANL level control sets the levelof signal that will trip the automatic noiselimiter.


Fig. 12.2 Turning diode noise eliminator circuit on and off.

AUDIOOUTPUT

AUDIOAMPLIFIER C2

2.2 µFC4

10 µFD1

D2

R15K R2

22K

R310K

R433K

C30.1 µF

R533K

R65.6K

AUDIOINPUT

C12.2 µF

S1

Fig. 12.3Gate circuit for turningnoise eliminator on and off.

IFAMPLIFIER

C10.1 µF

IFINPUT

IFOUTPUT

+12VOLTS

S1ANL

ON/OFFR110K

D1 D2R2100K

C20.1 µF

C30.1 µF

Q1

Figure 12.5 shows a somewhat morecomplex form of the noise blanker circuit inblock diagram format. The signal input is splitinto two paths. One path goes through a timedelay to the blanking gate. The other pathgoes to the noise amplifier circuit. The noisedetector picks out the noise and uses it totrigger a short-duration pulse generator (usu-ally a one-shot multivibrator). The output ofthe one-shot circuit drives the control ele-ment of the blanking gate circuit. When thissignal is active, the IF signal is not passed tothe output.

AUTOMATIC GAIN CONTROL

Automatic gain control stabilizes the gain ofthe receiver despite changes in input signalstrength. The AGC network keeps the vol-ume constant despite large variations in inputsignal level (in fact, it sometimes, in con-

sumer electronic receivers, is called auto-matic volume control, AVC). In a decent re-ceive, the AGC onset would be at 20 dB SNRat 1 µV and maintain the output within ±6 for120 dB changes in input conditions.

Figure 12.6 shows the block diagramof the superheterodyne receiver with theautomatic gain control. As with most super-heterodyne receivers, this one frequencytranslates the incoming RF signal to an IFfrequency for further processing (wheremost of the gain and the selectivity occur).Following the IF amplifier is a detector.This circuit represents one form of AGC cir-cuit: IF-derived AGC. There also is an audioderived AGC, which will be discussedshortly.

A sample of the IF signal is applied to theAGC network. This sample shows the strengthof the incoming signal. The signal is convertedto a proportional DC level used to control thegain of the IF or RF amplifiers or both.

The Other Receiver Circuits 193

Fig. 12.4 Noise blanker circuit.

IFSIGNAL

TO IFAMPLIFIER

C10.001 µF

C20.001 µFD1 D2

C368 µF

C40.1 µF

D3R1

1 MEG

R21 MEG

R31 MEG

R4100K R5

1 MEG

+12VOLTS

Q2

R610K

R72.2 MEG

R81 MEG

Q3

R910K D4

R1025K

ANL LEVEL

C50.02

C60.001 µF

Q1

R1110K

R12150

R131 MEG

Fig. 12.5Block diagram of morecomplex noise blanker.

TIME DELAYBLANKING

GATE

NOISEAMPLIFIER

NOISEDETECTOR

PULSEGENERATOR

SIGNALINPUT

SIGNALOUTPUT

Figure 12.7 shows how this circuit mightwork in a simple receiver. The IF sample istaken through capacitor C1, which is smallcompared to the other capacitors in the circuit(e.g., in the tuning network, if any). The signalsample is applied to a diode (D1), where it isrectified to DC level. This DC level is filteredby the combined effects of C2, C3, R1, and C2.Normally, capacitor C3 is quite large (50–500µF) and stabilizes the gain further by its timeconstant. The DC voltage then is applied tothe RF and IF amplifier circuits.

GAIN CONTROLLED AMPLIFIERS

If a transistor is used for the IF and RF am-plifiers, then the bias from the AGC circuit ismixed with the fixed bias to produce the to-tal bias on the transistor. By that means, thedesigner gets gain control. But few designersuse transistors singly as the front end of a ra-dio receiver anymore. They tend to use inte-grated circuits. These ICs can be differentialinput or single-ended input, but the princi-ple of operation is nearly the same. Figure


Fig. 12.6 IF-derived AGC circuit.

RFAMPLIFIER

IFAMPLIFIERS

LOCALOSCILLATOR

DETECTOR MODULATEDOUTPUT

AUTOMATICGAIN

CONTROL

IF S

AM

PLE

DC CONTROL VOLTAGE

MIXER

Fig. 12.7 Simple AGC circuit.

IFAMPLIFIER

RFAMPLIFIER

R1R2

C1

D1C2C3

12.8 shows the circuit for the input amplifieron an IC used for RF and IF amplification.This particular example is single ended, butyou could just as easily make it differentialby using the base of transistor Q2 as the al-ternate input.

This is a classic differential amplifier.Transistors Q1 and Q2 are a matched pairand transistor Q3 operates as a constant cur-rent source (CCS). Therefore,

(12.1)

The upshot of equation 12.1 is that anyincrease in the collector current of Q1 resultsin a decrease in the collector current of Q2.Therefore, when the signal goes positive, itincreases the collector current of Q1. This im-mediately causes a decreased collector cur-rent in Q2, which raises the output voltage.Similarly, when the signal input is negative,the collector current of Q1 is decreased andthe collector current of Q2 is increased (re-sulting in a decrease of output voltage).

Gain control is created by controllinghow much collector current is allowed fortransistor Q3. This figure is subject to the DCbias applied to the gain control input of theIC. Some amplifiers will prebias this amount

to a happy medium, then allow the designerto raise or lower it (not shown).

AGC Response Time

The AGC system will not respond directly toa change in input signal amplitude, rather, ashort delay is built into the system. Thefastest possible response time of the AGC cir-cuit depends on the filtering ahead of the de-tector, filtering of the detector, and the IFamplifier itself. For IF-derived systems, theminimum IF frequency that can be accom-modated is about 25 or 30 kHz, which resultsin a time of 4 µs. In audio-generated systems,with a minimum frequency of 50 Hz, thetime is 20 ms. It is considered good designpractice to use IF-derived systems rather thanbaseband or audio-derived systems in goodcommunications receivers.

The current practice is to use a re-sponse time of 20–50 ms. In actuality, therole of the response time is of primary in-terest when receiving CW or SSB signals,so that the background noise does not rise.The time constant for an AM signal de-pends on the lowest frequency in the AMsignal’s modulating waveform. A dual time-

I I IC Q C Q C Q( ) ( ) ( )3 1 2= +


Fig. 12.8Differential amplifier circuit.

DCBIAS

OUTPUT

DCBIAS

INPUT Q1 Q2

Q3GAIN

CONTROL

V+

R1 R2

constant system is needed if AM/FM is tobe accommodated in addition to CW/SSB.In some cases, three time constants areused: attack, hold, and decay. In AMbroadcast band receivers, time constants of60–100 ms are used.

In IF filters, there is the matter of delaythrough the filter. This filter could cause de-lays ranging from a few microseconds up to100 ms (for a mechanical filter). If the AGCattack time is shorter than the delay, thenAGC instability may result.

Attenuators

It is common practice, in modern receiver de-sign, to insert an attenuator into the signalpath. This is done to back the signal levelsaway from the –1 dB compression point orthe third-order intercept point. Often dramaticperformance improvement can be affectedthis way. Either single, switched (stepped), orvariable attenuators might be used.

Figure 12.9A shows a PIN diode, andFigure 12.9B shows its attenuator circuit. The

PIN diode is similar to the PN diode, exceptfor the resistive intrinsic section in the center,between the P and N sections. This intrinsicsection permits the diode to operate as a vari-able resistor. In the attenuator circuit of Figure12.9B, the diode is in series with the signalsource. It could be in parallel if another ap-proach were taken. DC bias, which sets theoperating point of the PIN diode, is appliedthrough resistor R2 from potentiometer R1.

S Meters

An S meter records the signal strength of theinput signal and is available on the front panelof the receiver. It is calibrated in S units, 0through 9, with as much as 60 dB over S 9possible. The definition of an S unit is open tosome ambiguity. Typically, an S unit is either a3 or 6 dB change in the input signal strength,with either 50 or 100 µV of signal being equalto S 9.

The S-meter circuit can be either audioor IF derived. Figure 12.10 shows an audio-derived S-meter circuit. It consists of a JFET


Fig. 12.9PIN diode: (A) structure;(B) attenuator circuit.

V+

R1

R2

C1 C2D1

R3

RFINPUT

RFOUTPUT

P I N

A

B

preamplifier, so that the high input imped-ance does not load the detector circuit toomuch. This circuit feeds a bipolar transistoramplifier and a voltage double-rectifier cir-cuit. The S meter itself is connected to theoutput of the rectifier circuit.

An IF-derived S-meter circuit is shownin Figure 12.11. This circuit uses an amplifierfrom the IF amplifier stage. Typically, thisamplifier will have a high input impedance,so it is likely to have a JFET or MOSFET de-sign. The output of the amplifier is fed to arectifier circuit. A DC amplifier consisting oftwo bipolar transistors actually drives the Smeter. Note that the AGC also is derived fromthis circuit. Any circuit that can drive a DCmeter with signal strength information is ca-pable of driving the AGC, too.

SQUELCH CIRCUITS

A squelch circuit (also called mute) is used tocut off the output of the receiver during times

when no signal is being received. When mon-itoring for signals, or between channels, con-siderable noise is generated that is fatiguing tolisten to. The squelch circuit will cut off the re-ceiver during times when no signal, or veryweak signals, is present to reduce this effect.

Figure 12.12 shows an AM squelch cir-cuit. AM receivers generally use the AGC cir-cuit to trigger the squelch circuit. The signalfor the AGC circuit may be taken from themixer or the IF amplifier, but the AGC detec-tor generates a DC voltage proportional tothe signal level. This voltage is large in thepresence of a strong signal and small in thepresence of a weak signal.

The signal flows from the demodulator,through an optional isolation amplifier, to asquelch gate. The inputs to the squelch gateare the audio and a control DC level.

Either the AGC voltage or a sample of itwill be used to trigger a Schmidt trigger circuit.This circuit will produce a binary output (highor low) dependent on the level set (shownhere as a squelch adjust control). If the level is


Fig. 12.10 Audio-derived S-meter circuit.

FROM IFAMPLIFIER

AGCOUTPUT

S-METER

+12VOLTS

AMPLIFIERC1

0.01 µF

D2

D1R1

2.2K

R210K

R31 MEG

D3

R44.7K

R547K R6

680

R712K

R810K

R9100

S-METER ADJ

C20.1 µF

C350 µF

C450 µF

exceeded, then the binary gate control signalis applied to the squelch gate, opening it.

Some FM receivers do not use an AGCcircuit, so they must generate the squelch com-mand some other way. A typical FM receiversquelch circuit is shown in Figure 12.13. Fromthe FM demodulator the signal is split into twopaths. One path is the squelched audio, andthe other path is through a 100 Hz low-pass fil-ter. The output of the 100 Hz low-pass filter islargely noise, which is amplified in the noise

amplifier and demodulated in the noise detec-tor. This output is applied to the Schmidt trig-ger. The output of the Schmidt trigger (level setby the squelch adjust control) operates thesquelch gate in the audio pathway.

AUTOMATIC FREQUENCY CONTROL

Automatic frequency control is used in someFM receivers to keep the tuning frequency


Fig. 12.11 IF-derived S-meter circuit.

FROMDETECTOR

+12VOLTS

TO AUDIOAMPLIFIERS

R1VOLUME

R21 MEG R3

560

R43.3K

R515K

R61.5K

R72.7K

R8220

D1

D2

C122 µF

C222 µF

C322 µF

C422 µF

C522 µF

C650 µF

S-METER

Fig. 12.12 AM squelch circuit.

IFAMPLIFIER

FROMMIXER

DEMODULATOR

AGCDETECTOR

ISOLATIONAMPLIFIER

SQUELCHGATE

AUDIOOUTPUT

AGCOUTPUT

SCHMIDTTRIGGER

R1SQUELCHADJUST

GATE CONTROL SIGNAL


stable. This type of circuit is confined largelyto older, analog FM receivers, because mod-ern digital frequency synthesizers preventthe kind of drift and outright frequency shift-ing common in analog receivers.

A typical AFC system is shown inFigure 12.14. The AFC output of the FM de-modulator is 0 V when the input signal fre-quency is directly on the resonant frequency

of the demodulator but positive or negativedepending whether the voltage is above orbelow resonance. This voltage is fed back toa voltage controlled oscillator circuit, whichfeeds the mixer. In practice, a varactor (vari-able capacitance diode) is used to shunt theL-C or quartz crystal in the oscillator, andthis varactor converts the oscillator into aVCO.

Fig. 12.13 FM squelch circuit.

IFAMPLIFIER

FROMMIXER

FMDEMODULATOR

AUDIOAMPLIFIER

SQUELCHGATE

AUDIOAMPLIFIER

AUDIOAMPLIFIER

LOW-PASSFILTER

NOISEAMPLIFIER

NOISEDETECTOR

SCHMIDTTRIGGER SQUELCH

ADJUST

Fig. 12.14 Automatic frequency control circuit.

RFAMPLIFIER

MIXERIF

AMPLIFIER


FMDEMODULATOR

AUDIOOUTPUT

Monolithic microwave integrated circuits(MMICs) and similar devices are used in awide variety of receivers. These devices maybe very wideband or relatively narrowband.Very wideband amplifiers have a bandpass(frequency response) of several hundredmegahertz or more, typically ranging fromsub-VLF to the low end of the microwavespectrum. An example might be a range of100 kHz to 1000 MHz (i.e., 1 GHz), althoughsomewhat narrower ranges are more com-mon. These circuits have a variety of practicaluses: receiver preamplifiers, signal generatoroutput amplifiers, buffer amplifiers in RF in-strument circuits, cable television line ampli-fiers, and many others in communicationsand instrumentation.

One reason why very wideband ampli-fiers are rarer than narrowband amplifier cir-cuits is that they were difficult to design andbuild until the advent of monolithic mi-crowave integrated circuit devices. Severalfactors contribute to the difficulty of designingand building very wideband amplifiers. Forexample, too many stray capacitances and in-ductances are in a typical circuit layout, and

these form resonances and filters that distortthe frequency response characteristic. Also,circuit resistances combine with the capaci-tances to effectively form low-pass filters thatroll off the frequency response at higher fre-quencies, sometimes drastically. If the RCphase shift of the circuit resistances and ca-pacitances is 180º at a frequency where theamplifier gain is ≥1 (and in very wideband cir-cuits that is likely) and the amplifier is an in-verting type (producing an inherent 180ºphase shift), then the total end-to-end phaseshift is 360º—the criteria for self-oscillation.

If you have ever tried to build a verywideband amplifier, it likely was a very frus-trating experience. Until now. New, low-costdevices, called silicon MMICs, make it possi-ble to design and build amplifiers that coverthe spectrum from near DC to about 2000MHz, using seven or fewer components.These devices offer 13–30 dB of gain (seeTable 13.1) and produce output power levelsup to 40 mW (+16 dBm). Noise figures rangefrom 3.5 to 7 dB. In this chapter, we use theMAR-X series of MMICs by Mini-CircuitsLaboratories as representative.

Chapter 13

Monolithic Microwave Integrated Circuits

201

Table 13.1 The MAR-X Series of MMICs byMini-Circuits Laboratories

Type Color Gain @ 500 Maximum Number Dot MHz (dB) Frequency

MAR-1 Brown 17.5 1000 MHz

MAR-2 Red 12.8 2000 MHz

MAR-3 Orange 12.8 2000 MHz

MAR-4 Yellow 8.2 1000 MHz

MAR-6 White 19.0 2000 MHz

MAR-7 Violet 13.1 2000 MHz

MAR-8 Blue 28.0 1000 MHz

Figure 13.1 shows the circuit symbol forthe MAR-X devices. Note that it is a very sim-ple device. The only connections are RF in-put, RF output, and two ground connections.The use of dual ground connections distrib-utes the grounding, reducing overall induc-tance and thereby improving the groundconnection. Direct current power is appliedto the output terminal through an externalnetwork. But more on that shortly.

The package for the MAR-X device is shown in Figure 13.2. Although an IC, the device looks very much like a smallUHF/microwave transistor. The body ismade of plastic and the leads are widemetal strips (rather than wire) to reduce thestray inductance that narrower wire leadswould exhibit. These devices are smallenough that handling can be difficult; Ifound that hand forceps (tweezers) werenecessary to position the device on a proto-type printed circuit board. A magnifyingglass or jeweler’s eye loupe is not out of or-der for those with poor close-in eyesight. Acolor dot and a beveled tip on one lead arethe keys that identify pin 1 (the RF inputconnection). When viewed from above, pinnumbering (1, 2, 3, 4) proceeds counter-clockwise from the keyed pin.


Fig. 13.1 MAR-X circuit diagram.

A1RF IN

GROUND1

GROUND2

RF OUT

Fig. 13.2 MAR-X and ERA-X device packages.

3

4

1

2

BEVEL

COLOR DOT

3

4

1

2

BEVEL

COLOR DOT

(A) MAR-X DEVICES (B) ERA-X DEVICES

INTERNAL CIRCUITRY

The MAR-X series of devices inherentlymatches 50 Ω input and output impedanceswithout external impedance transformationcircuitry, making it an excellent choice for gen-eral RF applications. Figure 13.3 shows the in-ternal circuitry for the MAR-X devices. Thesedevices are silicon bipolar monolithic ICs in atwo-transistor Darlington amplifier configura-tion. Because of the Darlington connection,the MAR-X devices act like transistors withvery high gain. Because the transistors are bi-ased internal to the MAR-X package, the over-all gains typically are 13–33 dB, depending onthe device selected and operating frequency.No external bias or emitter bias resistors areneeded, although a collector load resistor toV+ is used.

The good match to 50 Ω for both inputand output impedances (R), from the circuitconfiguration, is approximately

(13.1)

If RF is about 500 Ω and RE is about 5 Ω, thenthe square root of their product is the desired50 Ω.

BASIC CIRCUIT

The basic circuit for a wideband amplifierproject based on the MAR-X device is

shown in Figure 13.4. The RF in and RF outterminals are protected by DC blocking ca-pacitors C1 and C2. For VLF and MW appli-cations, use 0.01 µF disk ceramic capacitors,and for HF through the lower VHF (≤100MHz) use 0.001 µF disk ceramic capacitors.But, if the project must work well into thehigh-VHF through low-microwave region(>100–1000 MHz or so), then opt for 0.001µF (1000 pF) “chip” capacitors. If there is norequirement for lower frequencies, thenchip capacitors in the 33–100 pF range canbe used.

The capacitors for C1 and C2 should bechip capacitors in all but low-frequency(<100 MHz) circuits. Chip capacitors can be abit bothersome to use, but their use paysever greater dividends as operating fre-quency increases.

Capacitor C3 is used for two purposes.It prevents signals from A1 from being cou-pled to the DC power supply and from thereto other circuits. It also prevents higher-frequency signals and noise spikes from out-side sources affecting the amplifier circuit. Insome cases, a 0.001 µF chip capacitor is usedat C3, but for the most part, a 0.01 µF diskceramic capacitor suffices.

The other capacitor at the DC power sup-ply is a 1 µF tantalum electrolytic, which de-couples low-frequency signals and smoothesout short-duration fluctuations in the DC sup-ply voltage. Values higher than 1 µF may be

R R RF E=

Monolithic Microwave Integrated Circuits 203

Fig. 13.3MAR-X internal circuitry.

1

3

2 4

RF IN

RF OUT

GROUND1

GROUND2

RF

RB RO

RE

required if the amplifier is used in particularlynoisy environments.

Direct current is fed to the amplifierthrough a current limiting resistor (R1), via theRF out terminal on the MAR-X (lead 3). Themaximum allowable DC potential is +7.5 VDCfor MAR-8, +5 VDC for MAR-1 through MAR-4,+4 VDC for MAR-7, and 3.5 VDC for MAR-6. Ifa minimum voltage V+ power supply is used(e.g., +5 VDC for MAR-1), then make R1 a47–100 Ω resistor. Use only 1/4 W or 1/2 W non-inductive resistors, such as the carbon compo-sition or metal film types. If higher V+potentials (e.g., +9 or +12 VDC) are necessary,then use a higher-value noninductive resistorfor R1. To determine the value of R1, decide ona current level (I), and calculate by Ohm’s law:

(13.2)

In most cases, a good operating currentlevel for the popular MAR-1 is about 15 mA(0.015 A).

Example

Calculate a value for R1 in a MAR-1 circuitwhen a +9 VDC transistor radio battery is

used for the V+ DC power supply. Assume I = 15 mA.

Because 270 Ω is a nearby standard value, itwould be used instead of 267 Ω.

An optional inductor, RFC1, is shownin the circuit of Figure 13.4. This inductorserves two purposes. First, it improves thedecoupling isolation of the MAR-X outputfrom the DC power supply by blocking RFsignals. Second, it acts as a “peaking coil”to improve the gain on the high-frequencyend of the frequency response curve. Itdoes this by adding its inductive reactance(XL) to the resistance of R1 to form a loadimpedance that increases with frequency,because XL = 2 πFL. Suitable values of induc-tance range from less than 0.5 µH to about100 µH, depending on the application andfrequency range. Sometimes, however, thecoil forms the total load impedance. In those

R =V+ V

I

R =

R =

1( )( )

1( )(9 VDC) (5 VDC)

0.015 A

1( )4 V

0.015 A= 267

Ω

Ω

Ω Ω

−

−

R

V V

I1 ( )

( )Ω =

+ −


Fig. 13.4Basic circuit for a wide-band amplifier.

+

C41 µF

C30.01 µF

V+

R1

RFC1(OPTIONAL)

A1MAR-x

C1 C2

RFINPUT

RFOUT

F.B.

R1

I

VVR

-+=

)(1

cases, a decoupling capacitor is used at thejunction of RFC1 and R1.

Inductor coils are not without problemsin very wideband amplifiers because thestray capacitances between the coil windingsform unintended self-resonances with thecoil inductance. These resonances can distortthe frequency response curve and may causeoscillations. A popular solution to this prob-lem is to use a small ferrite bead (F.B., in theinset to Figure 13.4). The bead acts as asmall-value RF choke. These beads have asmall hole in them that fits nicely over the ra-dial lead of a 1/4 W resistor.

Alternative DC power schemes areshown in Figure 13.5. The circuit of Figure13.5A splits the load resistance into two com-ponents, R1 and R2. The value of R1 repre-sents most of the required resistance, with R2

typically being 33–100 Ω. This circuit, like thebasic circuit, works well to V+ voltages of 7–9VDC but is not recommended for V+ >9 VDC.

Power feed schemes that work well atV+ voltages greater than 9 VDC are shown inFigures 13.5B and 13.5C. Both use voltageregulation to stabilize the supply voltage tothe MAR-X device. In Figure 13.5B, a 68 ΩVDC zener diode holds the voltage applied toR1 constant, and within acceptable range, de-spite fluctuations in the source V+ potential.

OTHER MAR-X CIRCUITS

The simple circuit of Figure 13.4 will workwell in most cases, especially where the in-put and output impedance are reasonablystable. But, if the input source or output loadimpedances vary, then the amplifier may suf-fer a degradation of performance or showsome instability. One solution to the problemis to use resistive attenuator pads in the inputand output signal lines. Attenuators in an am-plifier circuit? Yes, a 1 or 2 dB attenuator inthe input and output signal lines will pseudo-stabilize the impedances seen by the ampli-fier, but only marginally affect the overallgain of the circuit. In vacuum tube days, wecalled this technique swamping.

Figure 13.6 shows the circuit of Figure13.4 revised to reflect the use of simple resis-tive attenuator pads in the input and outputlines. With resistor values of 6.2 Ω for the


Fig. 13.5 Alternative DC power schemes: (A) with optional ferrite bead output circuit; (B) with zenerdiode; (C) with 78L05.

RFOUT

R1

A1MAR-X

C2

R2820

C4

C3

F.B.(OPTIONAL)

D16.8 V/400 mWZener Diode

RFOUT

R1

A1MAR-X

C2

R2

C4

C3

F.B.(OPTIONAL)

RFOUT

R1

A1MAR-X

C2

C3

F.B.(OPTIONAL)

C4

IC178L05

A

B

C

series element and 910 Ω for the two shuntlines, the attenuation factor is 1 dB. A 2 dBversion uses 12 Ω and 470 Ω, respectively. If1 dB attenuators are used, then the overallgain is the natural gain of the MAR-X deviceless 2 dB (or 4 dB if 2 dB attenuator pads areused). The resistors used for these attenuatorpads must be noninductive types, such ascarbon composition or metal film units. If theamplifier is to be used at the higher end of itsrange, then chip resistors are preferable toordinary axial lead resistors.

An alternate approach is to use manufac-tured shielded RF 50-Ω attenuator pads. OtherMini-Circuits products are the AT-1 and MAT-11-dB attenuators; they are suitable for the pur-pose and match the frequency range of mostof the MAR-X products. These low-cost de-vices are similar except for size and are in-tended for mounting on printed circuit boards.

Keep in mind that the use of attenuatorsis not for free (the TANSTAFL principle: Thereain’t no such thing as a free lunch). The resis-tive attenuators reduce the gain (as mentioned)but also increase the noise factor by an amountset by the loss factor of the attenuator pad.

For VHF, UHF, and low-end microwaveamplifiers, it may be preferable to use aprinted circuit strip-line transmission line forthe input and output circuits. Figure 13.7A

shows such a circuit with input (SL1) andoutput (SL2) strip lines. Figure 13.7B showsdetails of how these lines are made. Thecharacteristic impedance (Zo) of the line is afunction of the relative dielectric constant ofthe printed circuit material (ε), the thicknessof the material (T ), and the width (W ) of thestrip line conductor. Common epoxy G-10printed circuit boards (ε ≈ 4.8) are usable to1000 MHz and work well to about 300 MHz.Above 300 MHz, the losses increase signifi-cantly. PTFE woven glass fiber printed circuitboards (ε ≈ 2.55) operate to well over 2000MHz, which is higher than the upper limit ofthe MAR-X devices. Widths required for 50 Ωstrip lines for various printed circuit boardmaterials are shown in Table 13.2.


Fig. 13.6 Use of input/output attenuator circuits.

+

C41 µF

C30.01 µF

V+

R1

RFC1(OPTIONAL)

A1MAR-X

C1 C2RF

INPUTRF

OUT

R26.2

R3910

R4910

R56.2

R6910

R7910

1 dBATTENUATOR

1 dBATTENUATOR

Table 13.2 Values for 50 Ω Strip Lines

Material ε T W

G-10 epoxy 4.8 0.062 in. 0.108 in. fiberglass (1.58 mm) (2.74 mm)

PTFE woven 2.55 0.010 in. 0.025 in. glass fiber (0.254 mm) (0.635 mm)

0.031 in. 0.079 in. (0.787 mm) (0.20 mm)

0.062 in. 0.158 in. (1.58 mm) (4 mm)

Figure 13.8A shows a typical printedcircuit board for a MAR-X wideband ampli-fier. The circuit layout is shown as an inset toFigure 13.8A. The printed circuit boardshould be double clad; that is, copper-cladon both top and bottom. The strip lines atthe input and output are etched from thecomponent side of the printed circuit mater-ial, not the bottom side as is common prac-tice in lower-frequency projects. The reasonfor this approach is to reduce the inductanceof the leads to the MAR-X device.

Strip lines should not contain abruptdiscontinuities, or else parasitic losses will in-crease. It is common practice to taper theline over a short distance from the strip lineto the width of the MAR-X leads right at thebody of the device.

Another tactic to keep stray lead induc-tances to a minimum is to drill a small hole inthe printed circuit to hold the body of the MAR-X (Figure 13.8B). The diameter of the MAR-Xpackage is 0.085 in. (2.15 mm), and the holeshould be only slightly larger than this value.


Fig. 13.7 Strip-line transmission lines: (A) input/output circuits; (B) implementation.

STRIP LINE

PRINTED CIRCUITMATERIAL

W

T

COPPER-CLADGROUND PLANE MATERIAL

R1

A1MAR-X

C2

RFOUTSL1 SL2

C3

V+

RFC1

C1

RFINPUT

A

B

The capacitors in the input and outputcircuits, as well as the decoupling capacitorat the junction of RFC1 and R1, are chip ca-pacitors. The break in the strip line to ac-commodate these capacitors should be justwide enough to separate the ohmic contactsat either end of the capacitor body. For the1000 pF (0.001 µF) chip capacitors I used inmaking a model in preparation for this chap-ter, the insulated center section between con-tacts on the capacitors averaged 0.09 in. (2.3mm) as measured on a vernier caliper set.

It is essential to keep ground returns asshort as possible, especially when the ampli-fier operates in the higher end of its range.If you opt to use the ground plane claddingfor the DC and signal return, then plated

through holes are required between the twosides of the board. These plated throughholes must be placed directly below theground leads of the MAR-X package.

MULTIPLE DEVICE CIRCUITS

The MAR-X devices can be connected in cas-cade, parallel, or push-pull. The cascade con-nection increases the overall gain of theamplifier, while the parallel and push-pullconfigurations increase the output poweravailable.

The simplest cascade scheme is to con-nect two stages such as Figure 13.4 in seriesso that the output capacitor of the first stage


Fig. 13.8Printed circuit board fora MAR-X wideband am-plifier: (A) layout;(B) mounting of theMAR-X device.

MAR-x

PRINTED CIRCUITTRACKS

GROUND PLANECOPPER CLAD

INSULATOR

HOLE

+

C1 C2

A1RFC1

C3 R1C4

C5

J2OUT

J1IN

31

2

4

TOPVIEW

A

B

becomes the input capacitor of the secondstage. Figure 13.9 shows a somewhat betterapproach. This circuit uses strip line match-ing sections at the inputs and outputs andbetween stages. Table 13.3 gives the dimen-sions of these lines for two different cases:Case A is for a 100–500 MHz amplifier andCase B is for a 500–2000 MHz amplifier. Inboth cases the MAR-8 device is used.

The parallel case is shown in Figure13.10. The MAR-X devices can be connecteddirectly in parallel to increase the outputpower capacity of the amplifier. In Figure13.10, four MAR-X devices are connected inparallel. Other combinations also are possi-ble. I built a two-up version for a signal gen-erator output stage. The output power inFigure 13.10 is four times the power availablefrom a single device.

Unfortunately, in the parallel amplifier,the input and output impedances no longerare 50 Ω but it is 50/N, where N is the num-ber of devices connected in parallel. InFigure 13.10, the four devices to the inputand output impedances are 50/4 or 12.5 Ω.An impedance-matching device must beused to transform the lowered impedances tothe 50 Ω standard for RF systems. Becausemost impedance transformation devices do

not have the same wide bandwidth as theMAR-X devices, there is an obvious degrada-tion of the bandwidth of the overall circuit.

The push-pull configuration is shown inFigure 13.11. This circuit has two banks oftwo MAR-X devices each. The two banks are


Fig. 13.9 Cascade UHF/microwave amplifier.

A1MAR-X

A2MAR-x

J1 C1

SL1 SL2 SL3

SL5

SL4

R3220

R4220

R1 R2

J2C3C2 C4

C50.1 µF

+12

Table 13.3 Dimensions of Strip Line Matching Sections for Two Cases

Component Case A Case B

R1 124 Ω 69.1 ΩR2 69.8 Ω 69.1 ΩC1, C4 470 pF 68 pF

C2 1.5 pF 2 pF

C3 7.5 pF 2 pF

SL1 0.10 × 0.10 in. 0.04 × 0.10 in.(2.54 × 2.54 mm) (1.02 × 2.54 mm)

SL2 0.10 × 0.05 in. 0.04 × 0.10 in.(2.54 × 1.27 mm) (1.02 × 2.54 mm)

SL3 0.10 × 0.20 in. 0.04 × 0.10 in.(2.54 × 5.08 mm) (1.02 × 2.54 mm)

SL4 0.10 × 0.10 in. 0.04 × 0.10 in.(2.54 × 2.54 mm) (1.02 × 2.54 mm)

SL5 0.05 × 0.20 in. 0.05 × 0.20 in.(1.27 × 5.08 mm) (1.27 × 5.08 mm)

Notes: Capacitors are chip type. Resistors are 1% chiptype. Strip lines are in width × length.

connected in push-pull, so the circuit is cor-rectly called a push-pull parallel amplifier.The circuit retains the gain and increase inpower level of the parallel connection but

improves the second harmonic distortion thatsome parallel configurations exhibit. Push-pull amplifiers inherently reduce even-orderharmonic distortion.


Fig. 13.11 Parallel/push-pull parallel amplifier.

T2

J2OUT

A1MAR-X

A2MAR-X

A3MAR-X

A4MAR-X

J1

T1

C4

C3

C2

C1

RFC1

RFC2

R1

C70.001 µF

C50.001 µF

C61 µF

V+

Fig. 13.10 Parallel connection to increase output power.

C5

RFOUT

A1MAR-X

A2MAR-X

A3MAR-X

A4MAR-X

RFIN

C1

C2

C3

C4

RFC1R1

C6

V+

The input and output transformers (T1and T2) for the circuit of Figure 13.11 arebalun (balanced unbalanced) types, used toprovide a 180º phase shift of the signals forthe two halves of the amplifier. The baluntransformers typically are wound on ferritetoroidal coil forms with #26 AWG or finerwire. Because the balun transformers are lim-ited in frequency response, the circuit typi-cally is used in medium-wave and shortwaveapplications. A common specification forthese transformers is to wind six or seven bi-filar turns on a toroidal form, the turns madeof #28 enameled wire wrapped together toform a twisted pair of about five twists to theinch (≈ two twists per cm). Suitable cores(and a catalog) are available on the market.

UNIVERSAL, FIXED-GAINPREAMPLIFIER

Often, a universal preamplifier that has thestandard 50 Ω input and output impedances(Figure 13.12) is needed. This preamplifiercan be used ahead of radio receivers or

wherever 15 dB of gain is needed. Based onthe Mini-Circuits MAR-1 device, it producesgain from the VLF region to a frequency ofseveral hundred megahertz. With correct se-lection of capacitors, the gain is available tofrequencies in the 1000 MHz region, the up-per limit of the MAR-1 device.

Input and output coupling is providedby a pair of 0.001 µF capacitors. For lower-frequency devices (up to 150 MHz), these ca-pacitors can be ordinary disk ceramic types.However, for all but the lowest-frequencyuse, make sure that the leads of the disk ce-ramic capacitors are straightened and the ex-cess ceramic material cleaned from them.

Power is applied to the MAR-1 devicefrom a +5 V DC power supply through a 100Ω resistor. This resistor can be a 1/4 W typebut must be of either carbon composition ormetal film construction to reduce inductance;no wire wound resistors are allowed.

A sample printed circuit board templateis shown in Figure 13.13. This design pre-serves the strip line needed for proper opera-tion at higher frequencies. The dimensionsassume that G-10 epoxy fiberglass printed cir-


Fig. 13.12Simple MAR-1 circuit.

A1MAR-x

C10.001 µF

C20.001 µF

J150-OHMINPUT

J250-OHM

OUT

R1

+

C41 µF

C30.01 µF

V+

1 3

2

4

31

2

4

TOPVIEW

cuit board is used. Other boards, with differ-ent dielectric constants, can be used, but thedimensions will change. A very crude proto-type, designed for operation at less than 100MHz, is shown in Figure 13.14. This devicewas built on ordinary perforated Vectorboardoverlaid with adhesive-backed copper foil.The foil can be purchased in some electronicsupplies stores, but it is rare today. For crudeprojects like this one, it is possible to usenon-adhesive-backed foil and glue it to theperf board. Hobby shops, especially thosethat cater to doll house builders, sell this typeof foil (you will want 40 or 44 gauge).

MAST-MOUNTED WIDEBANDPREAMPLIFIER

Preamplifiers like that just discussed are in-tended for mounting close to the radio re-ceiver or other circuit they serve. This projectdescribes a similar preamplifier, also basedon the MAR-1 device, that will work at re-mote locations, powered from its own coaxial cable feedline.

Figure 13.15 shows the circuit of the re-mote portion. It consists of a standard MAR-1device but with optional −1 dB attenuators inseries with input and output lines. The atten-


Fig. 13.13 Layout of the printed circuit board for the MAR-1 circuit.

14 mm 16 mm10 mm 10 mm

17 mm 22 mm

Notes:Tracks 2.75 mm wideTracks spaced from groundplane 4 mm all around

3 mm 3 mm3 mm

Fig. 13.14 Parts placement on the MAR-1 circuit.

uators are used to enhance stability, but notall builders will want to use them. If you arewilling to risk oscillation and need the extra2 dB of gain stolen by the attenuators, thendelete them.

Note that the power supply end of re-sistor R1 is connected not to the power sup-ply but to the output coaxial connector ( J2).A 1 mH RF choke (RFC2) isolates the DC cir-cuit path from the RF flowing in the coaxialcable. Similarly, a DC blocking capacitor (C5)isolates the MAR-1 and the attenuator fromthe DC voltage on the coaxial line.

The entire remote circuit must be builtinside a shielded box, and furthermore, that

box should be weatherproof. If no weather-proof box is available, then use an ordinaryaluminum box and coat the seams and jointswith silicone seal or some other goop.

A DC/RF combiner box is needed at thereceiver end of the coaxial line; a design isshown in Figure 13.16. The primary DCpower in this case is a 9 VDC battery. Again,an RF choke isolates the DC power supplyfrom the RF in the circuit, and a DC blockingcapacitor (C9) is used to keep the DC voltagefrom the receiver.

BROADBAND HF AMPLIFIER

Figure 13.17 shows the MAR-1 device used ina broadband preamplifier for 3–30 MHz high-frequency bands. Like the previous circuit, it isintended for mast mounting, but if the powercircuit is broken at X, it can be used otherthan remotely. It can be powered by the samesort of power box as in Figure 13.16.

The key feature that differentiates thiscircuit from the previous circuit is the band-pass filter in the input circuit. It consists of a1600 kHz high-pass filter followed by a 32MHz low-pass filter. The circuit keepsstrong, out-of-band signals from interferingwith the operation of the preamplifier.Because the MAR-1 is a very wideband de-vice, it easily responds to AM and FM broad-cast band signals.


Fig. 13.15 Shielded RF amplifier for mast mounting.

C10.001 µF

A1MAR-x

RFC1100 µH

J2RF OUT

TO RCVRINTERFACE BOX

1-dBATTEN

1-dBATTEN

J1RF INFROM

ANTENNA

C20.001 µF

RFC21 mH

C30.001 µF

C40.01 µF

R1150

Fig. 13.16 Remote power circuit for mast-mountedamplifier.

+

J3FROM

PREAMP

J4TO RCVR

C90.01 µF

R2120

S1

B19 VDC

BATTERY

RFC31 mH

The MAR-X devices are an extremelyeasy way to build RF amplifiers from fre-quencies near DC to the low microwave re-

gion. They are easy to use and well behaved.Hobbyists will find them very convenient fora wide variety of applications.


Fig. 13.17 Mast-mounted amplifier with filter circuit.

L1BL1A

IN

C10.002 µF

C20.001 µF

C30.002 µF

L33.3 µH

L23.3 µH

L40.3 µH

L50.3 µH

OUT

C4100 pF

C5100 pF

FILTER CIRCUIT

C60.01 µF

A1MAR-x

RFC11 mH

J2OUT

1-dBATTEN

1-dBATTEN

J1IN

C70.01 µF

RFC21 mH

C80.01 µF

C90.1 µF

R1150

EXT.POWER

FILTER(See Inset)

Signals can be represented in a number ofways. The most familiar is the time domainrepresentation shown in Figure 14.1A. Thisview of a pair of signals plots their ampli-tudes against time and so reveals their wave-shapes (in this case sinusoidal). From anamplitude vs. time display one can tell thefrequency (because F = 1/T), amplitude, andwaveshape. An oscilloscope normally is usedto view the time domain aspect of a signal.

Another view is the frequency domain,shown in Figure 14.1B. This display plots am-plitude vs. frequency, so the same two signalsseen in Figure 14.1A will plot as a pair ofspikes in Figure 14.1B. The comprehensiveview of signals requires that we look at boththe time and frequency domains. Becausethey share a common axis, amplitude, we canview them orthogonally, as in Figure 14.1C.

All continuous waveforms can be repre-sented mathematically by a series of sine andcosine functions. Only the sine wave is pure,in the sense that it contains only one fre-quency. All other waveforms, including sinewaves with even the smallest possibleamount of distortion, possess a number ofharmonically related frequency components.The specific harmonics, their amplitudes and

phases, determine the final shape of the over-all complex wave. The complex wave can bedescribed by a Fourier series of the form

where

an and bn represent the amplitudes ofthe harmonics (see later);

f is the frequency in hertz (Hz);ω is the angular frequency 2πf.

The amplitude coefficients (an and bn)are expressed by

The amplitude terms are nonzero at thespecific frequencies determined by the Fourierseries. Because only certain frequencies, deter-mined by the integer n, are allowable, thespectrum of the periodic signal is said to bediscrete.

b

Tf t n t tn

t= ∫2

0( ) sin ( )ω d

a

Tf t n t tn

t= ∫2

0( ) cos ( )ω d

f t

aa n t b n ttn nn

( ) cos ( ) sin ( )= + [ ] + [ ]=∞

∫o

2 1ω ω

Chapter 14

Spectrum Analyzer Receivers

215

The term ao/2 in the Fourier series ex-pression is the average value of f (t) over onecomplete cycle (one period) of the wave-form. In practical terms, it is the DC compo-nent of the waveform.

USING THE FREQUENCY DOMAIN

If we were certain that all signals in a systemwere pure sine waves, no modulation or het-erodyne mixing was taking place, and allstages in the system were perfectly linear,

then the time domain display seen on an or-dinary oscilloscope would suffice for practi-cal purposes. But that never happens. Realsignals have distortion, undergo both modu-lation and frequency mixing, and never see aperfectly linear signal processing stage.

The principal use of a spectrum ana-lyzer is to examine noise, distortion, mixingaction, and modulation. It is necessary tocharacterize signals going into and comingout of a system to understand how the sys-tem acts on the signal. By examining every-thing that goes into and out of a system, we


Fig. 14.1 Waveshapes: (A) time domain; (B) frequency domain; (C) time and frequency domains.

TIME

AM

PLI

TU

DE

FREQ

UENCY

TIME

AM

PLI

TU

DE

FREQUENCY

AM

PLI

TU

DE

A B

C

can characterize the system and determine itsperformance.

Noise

Figure 14.2A shows a frequency domain char-acterization of a noise signal. Understandingthe noise spectrum allows us to either evalu-ate or design the system to best overcome itseffects. The noise spectrum therefore permitsus to spot problems in system performanceand design accordingly.

Harmonic Distortion

When a pure sine wave is passed through anonlinear stage, harmonic components aregenerated. These new frequencies are integer

multiples of the fundamental frequency (2F,3F, 4F, . . ., nF). When an impure sine wave(which has its own harmonics) is processedin a nonlinear stage, additional harmonics orincreased harmonic amplitudes are created.Figure 14.2B shows the frequency spectrum ofa waveform with multiple harmonics present.The tallest spike represents the fundamentalfrequency sine wave, while the smaller spikesare the harmonics.

Intermodulation Distortion

While harmonic distortion occurs on a singlesignal, intermodulation distortion (IMD) oc-curs when two or more signals mix in a non-linear circuit. When this occurs, additionalfrequencies are generated according to the

Spectrum Analyzer Receivers 217

Fig. 14.2 Frequency domain signals: (A) noise; (B) harmonic distortion; (C) intermodulation distortion;(D) modulation.

A B

C D

rule mF1 ± nF2, where m and n are eitherzero or integers. Figure 14.2C shows this ac-tion when two equal amplitude signals (F1and F2) interact in a nonlinear manner. Thetwo small peaks are particularly interesting inamplifier and receiver designs because theyfall close to F1 and F2 (other products fall veryfar away). These are the 2F1 – F2 and 2F2 – F1products.

When not desired, this effect is calledIMD, but when desired to translate frequen-cies in a mixer circuit, the effect is called het-erodyning.

Modulation

A single-frequency unmodulated signal willhave a spectrum consisting of a single spike inthe absence of distortion. But, when informa-tion is imparted to the signal, additional prod-ucts are created. These show up as a spectrumsimilar to Figure 14.2D. Here we see the resultwhen a sine wave RF carrier is amplitudemodulated by a sine wave audio tone. In thiscase, mixing action takes place, as shownclearly on the spectrum trace. If RF carrier Fc ismodulated by audio tone Fa, then the twoproducts are Fc + Fa and Fc − Fa. These are theupper and lower sidebands, respectively.

SPECTRUM ANALYSIS

Spectrum analysis becomes possible whenthe various frequency components and noise

are measured and displayed. Over the years,several instruments have been used in spec-trum analysis: Fourier analyzers, tunable fil-ters, and spectrum analyzers.

The Fourier analyzer is depicted inFigure 14.3A, and its display is shown inFigure 14.3B. The analyzer consists of a se-ries of adjacent bandpass filters, each ofwhich passes a small amount of spectrum.When the outputs of these filters are poled, itis possible to build the display shown.

A number of problems are associatedwith the Fourier analyzer. First, it is not ter-ribly flexible, because the filters are fixedtuned. Second, the resolution depends onthe filter bandwidth, which may not beconsistent throughout the range of fre-quencies being measured. Third, only a restricted number of adjacent frequency filters can be accommodated, especiallywhere cost is a consideration. Finally, thefilters may interact, causing a loss of performance.

The tunable filter is shown in Figure14.4. A filter is designed to be tuned manu-ally across the entire range of frequencies.In most cases, these instruments actuallyare special-purpose radio receivers. Whenthe output is calibrated, the instrument iscalled either a wave analyzer or a tunableRF voltmeter.

Both approaches suffer from majorfaults, not the least of which is poor ease ofoperation. The modern spectrum analyzersolves these problems rather nicely.


Fig. 14.3 Fourier spectrum analyzer: (A) diagram; (B) display.

A B

The Spectrum Analyzer

The spectrum analyzer basically automatesand improves the tunable RF voltmeter bysweeping a superheterodyne receiver througha range of frequencies. Figure 14.5 shows thebasic block diagram for a generic spectrumanalyzer. It is a narrowband receiver swepttuned across the range of interest. A saw-tooth ramp waveform is used to sweep tunethe receiver and drive the horizontal deflec-tion system of the oscilloscope. The outputof the receiver is proportional to signalstrength and applied to the vertical input ofthe scope. The result is the spectrum plotshown.

To understand the operation of thespectrum analyzer, we look at each stage in

its turn. The heart of the spectrum analyzer isthe mixer and local oscillator (LO).

The LO is a voltage-controlled oscillatorthat produces an output frequency propor-tional to an applied input control voltage. Inthe spectrum analyzer, the input voltage is aramp, so the voltage will change as the rampvoltage rises. Because most VCO circuits havea quadratic relationship between the controlvoltage and frequency, it may be necessary toalter the tuning voltage waveform from a lin-ear ramp to a shape that makes the sweep ofthe VCO output frequency look linear.

The mixer is a nonlinear circuit thatmixes the RF input signal (F1) with the LOsignal (F2) to produce intermediate frequencyoutput. Any of the frequencies described bymF1 ± nF2 can be used, but it is not reason-able to use other than the second-order prod-ucts (F1 + F2 or F1 − F2).

The characteristics of the mixer are im-portant to the quality of the spectrum analyzer.Double-balanced mixers usually are preferredover single-ended or single-balanced mixers,because they tend to cancel the F1 and F2 sig-nals in the output, leaving only the sum anddifference products. Other forms of mixers in-variably have a residual F1 or F2 componentpresent in the IF output port.

A properly designed mixer circuitshould be terminated in its characteristic im-pedance (e.g., 50 Ω). Because only one ofthe two second-order products is needed,mixer performance is improved if the un-wanted second-order product is absorbedrather than reflected back toward the mixerIF port. A circuit called a diplexer often isused to separate the two second-order prod-ucts and route the desired product to the IFamplifier and the undesired product to amatching dummy load resistor.

It is very important to select a mixerwith a high third-order intercept point(TOIP) and a high dynamic range. One fail-ing of cheap spectrum analyzers is that themixer lacks these attributes, so it is possibleto generate both harmonic and intermodula-tion distortion products inside the mixer. Theproducts appear at the output of the spec-trum analyzer, and displayed on the screen,


Fig. 14.4 Tunable filter spectrum analyzer.

TUNE

even though they are spurious signals notpresent in the input spectrum.

The front-end of the spectrum analyzerconsists of the mixer/LO plus any prepro-cessing done. Two preprocessing resourcesare shown here: RF input attenuator and RFfilter. Some spectrum analyzers might alsohave a preamplifier. It is not unreasonable toexpect these stages to be switch selectable.

The RF attenuator is used to reduce theamplitude of all signals applied to the RF in-put of the spectrum analyzer by an equalamount. The input attenuator is used to pre-vent the mixer and any preamplifiers usedfrom going into gain compression. Once gaincompression is reached, intermodulationproducts begin to creep upward, distortingthe picture of the spectrum with spurs thatwere not in the original.

The input filter may be a bandpass,low-pass, or high-pass filter, used to preventunwanted frequencies from entering thespectrum analyzer. If you are looking at afairly limited range of frequencies, say themodulation around a transmitter signal, then

filtering can eliminate out-of-range signalsfrom interfering with the process. Those un-wanted signals could force the mixer intogain compression and create spurs.

The IF section handles the signal fromthe output of the mixer. Most of the gain andselectivity of the spectrum analyzer are pro-vided in the IF section. The principal stagesare IF gain amplifier, IF attenuator, narrow-band filter, and logarithmic amplifier.

The gain amplifier provides adjustablevoltage gain to permit adjustment and scal-ing. It is used to adjust the vertical displace-ment of the signals without changing theinput conditions. In some cases, the amplifiergain and input attenuator are adjusted in tan-dem to prevent shifts in the vertical display.

The resolution bandwidth (RBW) of thespectrum analyzer is set by the IF filter. Thesmallest resolvable frequency unit is deter-mined by the RBW, so making it too widewill cause smearing of close-in responses.

The logarithmic amplifier providesrange compression so that both high- andlow-amplitude products can be displayed at


Fig. 14.5 Spectrum analyzer block diagram.

FILTERRF INPUT

ATTENUATOR

RFINPUT

MIXER

IF GAINAMPLIFIER

NARROW BANDIF FILTER

IFATTENUATOR

ENVELOPEDETECTOR

VIDEOFILTER

LOG AMP


RAMPGENERATOR

CRT SCREEN

SWEEPRATE

SWEEPWIDTH

the same time. Otherwise, providing enoughgain to see low-amplitude signals will causehigh-amplitude signals to go off the top ofthe scale. Also, using a logarithmic amplifierpermits the use of decibel notation on theCRT screen.

The video detector is an envelope detec-tor that produces a DC output that is propor-tional to the signal strength at the frequencybeing measured. It produces the vertical de-flection signal seen on the CRT screen. Butthis signal often is not too clean and mustbe filtered. Figure 14.6 shows prefilteringand postfiltering versions of a noisy wave-form. The prefiltering version is barely us-able, if that.

RESOLUTION

Resolution is the ability to distinguish twothings, so in spectrum analyzers, we can de-fine resolution as the measure of the instru-ment’s ability to distinguish between twoadjacent frequencies. This concept is shownin Figure 14.7. Although Figure 14.7 overem-phasizes the issue for the sake of illustration,the concept is the same. In Figure 14.7A, thebandwidth of the system is set too broad.Adjacent frequencies F1 and F2 are not dis-criminated but appear as one smeared traceon the CRT screen. Narrowing the band-width a little produces a pattern such as inFigure 14.7B. A dip appears in the pattern,

indicating that two frequencies might bepresent but imparting little additional infor-mation. In Figure 14.7C, the bandwidth isnarrow enough to break out the two fre-quency spikes.

Several things affect the resolution ofthe spectrum analyzer. One is the selectivity(Figure 14.8). Selectivity is the ability to re-solve adjacent signals. While resolution usu-ally is stated in terms of breaking out twoadjacent equal amplitude signals, selectivityaddresses the ability to break out two adja-cent unequal signals. In the latter case, asmall signal nestled close to a larger signalcould be lost in the filter skirts of the largersignal. Figure 14.8 shows the definition of se-lectivity: the ratio of the −60 dB bandwidth tothe −3 dB bandwidth:

Typically, the value of selectivity will bearound 5:1 for digital filters and between 10:1and 15:1 for analog filters.

Consider an example. Two signals, F1and F2, are spaced 20 kHz apart. Suppose theresolution bandwidth is 3 kHz and the filterhas a selectivity of 14:1. The bandwidth at −60dB, therefore, is (−3 dB BW) × Selectivity = 3kHz × 14 = 42 kHz. Because the selectivity ismuch larger than the frequency spacing, onesignal is lost in the skirts of the other. If thereis a large amplitude difference (e.g., 60 dB),

Selectivity

60 dB BW

3 dB BW=

−−


Fig. 14.6Prefiltering and postfil-tering signals.

PREFILTERING

POSTFILTERING


F1 F2

BANDWIDTHCURVE

DISPLAY

INPUT

F1 F2

BANDWIDTHCURVE

DISPLAY

INPUT

Fig. 14.7Resolution of input-display waveforms:(A) system set too broad; (B) bandwidthnarrowed a little; (C) bandwidth nar-rowed enough.

F1 F2

BANDWIDTHCURVE

DISPLAY

INPUT

A

B

C

then the lower signal will be lost in the noise.A rule of thumb is that minimum discerniblefrequency separation for highly unequal sig-nals is one half the −60 dB bandwidth.

The bandwidth of the system is a keyplayer but not necessarily solely the IF band-width. For example, it is quite reasonable tomake the video resolution bandwidth onetenth the IF resolution bandwidth to improvediscrimination.

Other factors also affect resolution anddisplay; for example, sweep speed/measure-ment time, residual FM and phase noise, andnoise floor.

The RBW of the spectrum analyzer isset by internal filters, most profoundly in theIF and video amplifier sections. These filtershave an associated time constant required forthem to correctly respond to an input signal.If the spectrum analyzer LO sweeps by toofast for the set RBW, then we see smearing ofthe signal, amplitude error, and error in thefrequency of the signal being measured. It isalso possible for a too-fast sweep to cause“ringing” in the filters. The narrower the fil-ter’s bandwidth and the better the selectivitycurve of the filter, the more likely fast sweptLOs will cause ringing and therefore distor-tion of the signal being displayed. On theother hand, too slow a sweep will cause un-comfortable flicker in the display, making itmore difficult to either interpret or photo-

graph. Sweep speed, therefore, is a trade-offbetween the needs of the filter and the needsof the display.

Residual FM in the spectrum analyzerswept LO tends to smear the signals dis-played and usually is not detectable by theuser. The key is to specify very low levels ofresidual FM, which is an argument in favorof using a phase-locked loop frequencysynthesizer.

Another factor is the noise sidebands orphase noise of the system. Figure 14.9 showsthe phase noise problem. All signal sourcesproduce some amount of unintended phasenoise. The level of noise usually is measured


Fig. 14.8Filter response curve showingselectivity.

-3 dBBW

-60 dBBW

FREQUENCY

FO

AM

PLI

TU

DE

0 dB

-3 dB

-60 dB

Fig. 14.9 Phase noise interferes with display ofweak signals.

PHASE NOISELINE

RBW

WEAK SIGNAL

in decibels below carrier (dBc) and normal-ized to a 1-Hz bandwidth; that is, dBc/Hz.For example, suppose we need to measure adistortion product signal that is −60 dB be-low a carrier using a 3-kHz RBW. To normal-ize the phase noise in 3 kHz in terms of dB,

So, for 3-kHz bandwidth 10 × log10[3000/1]= (10)(3.48) = 34.8 dB. Therefore, to view asignal that is −60 dB down from a carrier atthe input, the spectrum analyzer will need aphase noise specification of −60 dBc − (34.8dB) = −94.8 dBc.

All electronic instruments producenoise, and spectrum analyzers are no excep-tion. If you either short-circuit the RF input orterminate it in a matched impedance, thenyou will still see a noise level present on theCRT display. In a properly terminated input(e.g., with a shielded 50 Ω resistor in place),the noise level will be higher than the ex-pected thermal noise (4KTBR). The excessnoise is created inside the instrument.

A practical aspect of the internal noisegenerated is that it is not affected by the RFinput attenuator, while the signal is so af-fected. Figure 14.10 shows the effect on aweak signal display (i.e., the signal is close to

the noise level). In Figure 14.10A, the signalsees 0 dB attenuation in the RF front end andis discernible above the noise floor. But, inFigure 14.10B, the RF attenuator is set to re-duce the input signal −10 dB, yet the noiselevel remains the same. The signal now isobscured by the noise floor of the spectrumanalyzer. The noise floor is specified in termsof displayed average noise level (DANL) andis a function of bandwidth.

Because the noise floor is a function ofbandwidth, the situation often can be im-proved in a practical sense by narrowing thebandwidth to the minimum necessary to cor-rectly show the signal (excess bandwidth in-creases DANL). In addition, it is wise to keepthe sensitivity low enough that the signal isreadable and measurable but the noise flooris reduced to a negligible level.

USING THE SPECTRUM ANALYZER

There are several different ways to use aspectrum analyzer. Figure 14.11 shows theuse of the spectrum analyzer to view the out-put of a signal source or signal generator.This will work but is not necessarily the bestway to approach the problem. For one thing,if the output level of the signal generator istoo high, it might drive the spectrum analyzer

Phase noise 10 log

BW(Hz)

1 Hz10=−


Fig. 14.10Attenuation vs. signal.

0 dBATTENUATION

-10 dBATTENUATION

A

B

into distortion. The displayed spectrum thenwould exhibit elements due to the test setupand not what actually is coming from thesignal generator. The problem can be par-tially overcome either by keeping the signallevel low enough or placing attenuators inthe line between the signal source and spec-trum analyzer.

Another application is to measure theoutput spectrum of some circuit or “deviceunder test” (DUT), as shown in Figure 14.12.This approach is used to measure both theharmonic and intermodulation distortionproduced by the DUT. Unfortunately, thespectrum analyzer may produce some distor-tion of its own, and because the same funda-mental signals are involved, it will look likedistortion produced in the DUT. A simple testto show whether an observed distortionproduct is due to the spectrum analyzer orthe DUT is to change the attenuation appliedto the RF front end and observe the effects. Ifthe input attenuation is changed by 10 dB

and the displayed distortion productschange, then at least part of the distortion isbeing created in the spectrum analyzer. If thelevel of the products does not change, thenthe products are due to the DUT, not thespectrum analyzer.

Small transmitters can be tested offthe air by keying them in close proximityto a spectrum analyzer fitted with a smallwhip antenna (Figure 14.13). Care must betaken to keep the transmitter close enoughto produce a readable display but farenough away to not overload the spectrumanalyzer. It is possible for higher powertransmitters or transmitters brought tooclose to the spectrum analyzer antenna todamage the analyzer input. If the transmit-ter is too far from the spectrum analyzerantenna, then other signals may be visibleon the display as well as the desired signal.This is a constant concern when doing off-the-air checks with a sensitive spectrumanalyzer.


Fig. 14.11Measuring the outputspectrum of a signal generator.

SIGNAL GENERATORSPECTRUM ANALYZER

Fig. 14.12Measuring a device un-der test with a spectrumanalyzer.

SIGNAL GENERATORSPECTRUM ANALYZER

DUT

Devices that produce RF output powerlevels too high for direct application to thespectrum analyzer can be tested in a mannersuch as Figure 14.14. In this arrangement, theoutput power from the source is connectedto a coupler with an isolated port as well as astraight-through or direct port. A dummyload is connected to the direct port, whilethe isolated port is fed to the spectrum ana-lyzer. In some cases, an additional attenuatoris needed to augment the isolation loss of thecoupled port. Additional information on cou-plers is found in Chapter 2.

Figure 14.15 shows a method used bysome instruments to sample a high-powersignal directly from a dummy load. Somedummy loads include the sampling loop(“gimmick”), while in others it has to bebuilt. The small wire sampling loop is usedto pick up a tiny sample of the RF power ap-plied to the dummy load resistor.


Fig. 14.13Off-the-air checks with a spectrumanalyzer.

HANDHELDTRANSMITTER

SPECTRUMANALYZER

Fig. 14.14 Using a spectrum analyzer with apower source.

SPECTRUM ANALYZER

POWER SIGNALSOURCE

ADJUSTABLEATTENUATOR

COUPLER

DUMMYLOAD

Fig. 14.15Dummy load.

NONINDUCTIVE DUMMY LOAD RESISTOR

SHIELDED ENCLOSURE

SMALL WIRE LOOP("GIMMICK")

The trend today in receiver design is towarddigital signal processing. This powerful tech-nique frees the receiver of drift problems in-volving temperature and power supplyvoltage excursions. Today, the small size oflarge-scale integration digital integrated cir-cuits makes it possible to deliver receivers thatcould be only dreamed of a generation ago.

Although the goal is to get the analog-to-digital converter (ADC) as close to the an-tenna as possible, most receivers today fit theschema shown in Figure 15.1. From the IFamplifier two paths can be taken by the digi-tal designer.

First, you could demodulate the IF sig-nal and do the signal processing at audio. Inthat way, a simple, low-cost ADC is used(indeed, maybe even your computer’ssound card). A digital-to-analog converter(DAC) is used to convert the signal back toaudio. About the only thing that can bedone after the demodulator is filtering andaudio AGC.

Second, the processing takes place at theIF frequency. The ADC must be considerablybetter than in the audio case, but this permitsutilizing the digital circuitry to demodulate theIF signal. It also permits using I-Q quadrature

Chapter 15

Digital Signal Processing

227

Fig. 15.1 Digital signal processing schema.

IFAMPLIFIER

DEMODULATOR

A/DCONVERTER

DIGITALPROCESSING

A/DCONVERTER

DIGITALPROCESSING

D/ACONVERTER

AUDIOOUTPUT

D/ACONVERTER

AUDIOOUTPUT

processing techniques. After the ADC, therewill be a digital processing stage and a DAC toreconvert the signal to audio.

It is worth noting that, in radar re-ceivers, a significant increase in sensitivity isavailable because we can integrate a numberof pulses (N). In coherent processing, the IFis converted and the increase in sensitivity isN, but in noncoherent processing (after thedemodulator), it is

Another advantage of using the micro-processor to control the receiver is that allthe tuning, bandwidth, gain, and controlfunctions can be implemented through themicroprocessor. The user’s decisions onwhich of these to invoke are registered in themicroprocessor and relayed to the circuitsunder control.

ADC NOISE

All reception is a problem of signal-to-noiseratio (SNR). The ADC must add minimally tothe noise produced in the receiver or thedesign will fail. The sampling rate (numberof samples per second taken by the ADC)must be high enough to demodulate thebandwidth being handled. Nyquest’s theo-rem tells us that the data sampling rate mustbe twice the nominal bandwidth being re-ceived, but because filters have no straightedges, the actual rate will be up to 2.5 timesthe bandwidth. The width of the samplingpulse must be small compared to the periodof the highest frequency within the receiverpassband.

The ADC will add little inherent noiseof its own at the IF frequency, but as it getscloser to the antenna, with its smaller signallevels, the percentage of the total becomeshigher and higher. As a result, IF processingprobably will be the standard for some timeto come.

Figure 15.2 shows the quantization er-ror noise that will be encountered in theADC. The top curve is an input/outputcurve, whereas the bottom curve is a repre-sentation of the error. Each step of the ADCoutput is QV high. If the input signal be-

comes too large, then there will be a hardclipping action taking place at ±NQ, and theautomatic gain control circuits must be bal-anced to eliminate this problem. The rootmean square value of the quantizing voltageis given in Table 15.1.

RECEIVER ARCHITECTURES

Figure 15.3 shows the layout of a simplesingle-conversion receiver based on digitaltechniques. The front end of the receiverconsists of the bandpass filtering, RF ampli-fying, and mixing functions. It is driven by a

N.


Fig. 15.2 Quantizing error of the signal.

+VMAX

-VMAX

-Q

-2Q

Q

2Q

-3Q

3Q

ERROR

Table 15.1 Quantization Noise Voltage

N Bit Decibels

6 −20

8 −41

10 −53

12 −63

16 −97

20 −125

number of signals. The RF signal from theantenna drives the input of the front-endcircuitry.

There will be control signals for select-ing the bandpass filtering and automatic gaincontrol for the RF amplifier. Sometimes, a selectable attenuator will be in the circuit as well.

Finally, there is the local oscillator cir-cuit. In this case, the LO is an agile digital frequency synthesizer controlled by the mi-croprocessor.

The microprocessor also controls the IFamplifier gain through an AGC function. TheIF amplifier produces an output signal thatalmost fills up the ADC in the digital receiversection when a strong signal is received atthe antenna. Finally, the digital receiver sec-tion demodulates the signal and applies it tothe audio amplifiers for output to the loud-speaker or earphones.

The microprocessor controls the frontend, the local oscillator (agile frequency syn-thesizer), IF amplifier, and digital receiversection. User inputs on the front panel of thereceiver determine which function is to beused.

Figure 15.4 shows a block diagram of ahigh-frequency, double-conversion digital re-ceiver. The microprocessor controls are notshown here but nonetheless are present. Thefront end of the receiver consists of an atten-

uator, low-pass or bandpass filter, and amixer/local oscillator. In this case, the LO is adigital frequency synthesizer.

At the output of the mixer are a band-pass filter and a noise blanker. The noiseblanker punches a hole in the signal at the+10 dB amplifier whenever a noise impulseis received. The output of the bandpass filterfeeds a +10 dB amplifier, and the output ofthat amplifier is mixed with a signal from alocal frequency standard (which also servesthe digital frequency synthesizer). This pro-duces the second IF signal, which is the highgain (50–90 dB) stage.

Following the second IF is the ADC.This stage outputs digitized signals to thequadrature multiplier stage, which producesthe I and Q signals needed by the channel fil-ter and demodulator stage. If the receiver isto be used digitally, then the I and Q signalsare available as data out lines, but if an audiooutput is required, the signal is fed to a DACto reform the audio.

Figure 15.5 shows a block diagram of asomewhat more complex single-sideband re-ceiver. Again, for the sake of simplicity, I donot show the microprocessor that controlsthe action of the receiver.

The front end consists of the preselec-tor, mixer stage and a digital frequency syn-thesizer. The preselector consists of an RFamplifier and the bandpass or low-pass filter-

Digital Signal Processing 229

Fig. 15.3 Digital receiver block diagram.

FRONT ENDIF

AMPLIFIER

AGILEFREQUENCY

SYNTHESIZER

DIGITALRECEIVER

AUDIO AUDIOOUTPUT

MICROPROCESSOR


Fig. 15.5 Block diagram of a complex single-sideband receiver.

PRESELECTOR MIXER

DIGITALFREQUENCY

SYNTHESIZER

IFAMPLIFIER

No. 1MIXER

FIXEDFREQUENCYOSCILLATOR

IFAMPLIFIER

No. 2

A/DCONVERTER

IFTRANSLATOR

DECIMATIONFILTER

DIGITALSSB

FILTERMIXER MIXER

DECIMATIONFILTER

DIGITALSSB

FILTERMIXER MIXER

S D/ACONVERTER

AUDIOOUTPUT

BEATFREQUENCYOSCILLATOR

ENVELOPEDETECTOR

AGCSYSTEM

D/ACONVERTER

I

Q

ANTENNA

Fig. 15.4 Block diagram of a high-frequency, double-conversion digital receiver.

ATTENUATORLOW-PASS

FILTERMIXER

DIGITALFREQUENCY

SYNTHESIZER

BANDPASSFILTER(15 KHz)

NOISEBLANKER

+10 dBAMPLIFIER

MIXER

FREQUENCYSTANDARD

IFAMPLIFIER

(50 dB)

A/DCONVERTER

QUADRATUREMULTIPLIER

I

Q

CHANNELFILTER &

DEMODULATOR

D/ACONVERTER

DATAOUT

AUDIOOUTPUT

I

Q

ANTENNA

ing needed to protect the receiver and im-prove the intermodulation distortion perfor-mance. IF amplifier 1 is at the first IFfrequency, which in HF receivers typically isabout 50 MHz. In VHF/UHF receivers, a firstIF of 70, 30, or 10.7 MHz may be used.

The second IF frequency is generatedby the mixer following the first IF amplifierand the fixed frequency oscillator. This signalusually is 10.7 MHz, 9 MHz, 8.83 MHz, or 455kHz, depending on the frequency of the re-ceiver and the design. The second IF ampli-fier is the amplifier with the highest gain.

At the beginning of the digital receiversection is the ADC. The output of the ADCis fed to an IF translator stage, which pro-duces the I and Q output signals. Followingthe IF translator, the I/Q signals are fed todecimation filters and a digital SSB filter.Following the digital SSB filters are a set ofmixers and an envelope detector, which de-velops the AGC voltage in a DAC circuit.Finally, the signal is mixed with a beat fre-quency, then combined to form the input tothe DAC. The output of the DAC is an audiooutput signal.

Digital Signal Processing 231

In this chapter, we look at certain specialpurpose receivers: the spread spectrum re-ceiver, scanning superheterodyne receivers,instantaneous frequency measurement re-ceivers, microscan compressive receivers,and Bragg cell receivers.

SPREAD SPECTRUM RECEIVERS

The spread spectrum (SS) mode of transmis-sion was invented in 1941 by the late actressHedy Lamarr (who was billed as the“world’s most beautiful woman”) and avant-garde composer George Antheils. It was in-vented as a result of a conversation thatLamarr had with a Navy officer at a party.The officer was lamenting that torpedoeswent off on their own, and Hedy Lamarr re-portedly asked him why they did not radiocontrol them. He responded that the enemycould jam the frequency of the torpedo,making it useless. Hedy Lamarr invented thefrequency hopping method of SS to solvethe problem.

Unfortunately, Hedy Lamarr was anAustrian, and that made her officially an “en-

emy alien” during World War II, so her patentwas confiscated by the U.S. Navy. Lamarr peti-tioned the U.S. Navy in the 1990s for the re-turn of her now useless patent, and herpetition was granted. Since 1941, her patentedmethod has become valuable as a method oftransmission.

There are two methods of spread spec-trum: frequency hopping and direct sequence.Figure 16.1A shows the block diagram for afrequency hopping spread spectrum receiver,and Figure 16.1B shows the spectrum to beexpected. Note in Figure 16.1B that the pulses,which represent frequency, are of constantamplitude. If s(t) is the baseband signal, thenthe output of the frequency mixer is s(t) cos([w + ap(t)]t). The PN generator generates thefrequency hopping code [p(t)] for either a mul-tifrequency synthesizer or a direct digital syn-thesizer (DDS). Figure 16.2A shows a blockdiagram for a direct sequence type of spreadspectrum generator, and Figure 16.2B showsthe spectrum of the same signal.

The principal receiver issues are the re-quirements for wide band of the front endand synchronization of the receiver andtransmitter or the system will not work.

Chapter 16

Special Purpose Receivers

233


SCANNING SUPERHETERODYNERECEIVERS

The scanning superheterodyne receiver basi-cally is a standard superhet receiver with avoltage-controlled oscillator for a local oscil-lator and some means to drive it. Typically,the means for driving the local oscillator willbe a stepped voltage. Whether the steps areof equal or differential amplitude dependson whether the receiver is intended to scan

many adjacent frequencies or only specificfrequencies, which may not be adjacent.Either a noise gate or AGC gate will be usedto stop the scanning on a live channel.

INSTANTANEOUS FREQUENCYMEASUREMENT RECEIVERS

The instantaneous frequency measurement(IFM) receiver measures the frequency of an

MIXER

PNGENERATOR

BASEBANDSIGNAL

s( t )

SPREADSIGNAL

s( t ) cos([w + ap( t )] t )

p( t )DDS

cos([w + ap( t )] t )

Fig. 16.1Frequency hopping spread spectrum re-ceiver: (A) block diagram; (B) output wave-form of signal. FREQUENCY

AM

PLI

TU

DE

A

B

MIXER

PNGENERATOR

BASEBANDSIGNAL

s( t )

SPREADSIGNAL

s( t ) p( t )

p( t )

Fig. 16.2Direct sequence spread spectrum receiver:(A) block diagram; (B) output waveform ofsignal. FREQUENCY

AM

PLI

TU

DE

A

B

incoming signal instantaneously, which pro-hibits the use of a scanned receiver design.Instead, the IFM receiver uses delay lines tocompare the phase of an incoming frequency.The IFM receiver has a wide bandwidth (pos-sibly more than one octave) and can measurevery short pulse frequencies. The receivergenerally handles only one frequency at atime, so if two or more frequencies arrive atthe receiver simultaneously, they will be mea-sured as one.

In general, IFM receivers have the fol-lowing characteristics:

1. Moderate sensitivity;

2. Extremely wide RF bandwidth (oneoctave);

3. Simultaneous signals may cause erro-neous frequency data;

4. Very fine frequency measurement evenon short pulses;

5. Very wide dynamic range.

The modern IFM receiver separates theamplitude and frequency measurement func-tions, although in primitive receivers, bothwere measured on a CRT. The IFM is consid-ered a wideband receiver even though it iscapable of measuring the frequency quiteaccurately.

In some IFM receivers, it is commonpractice to keep the design less than one oc-tave, because more than one octave may

tend to overload the mixer with high-levelsecond harmonics.

Figure 16.3 shows a block diagram of astandard IFM receiver. Two separate paths areprovided: one for frequency and one for am-plitude. The amplitude detector has a logarith-mic amplifier to account for a wide inputrange and an ADC to provide the amplitudeinformation. The second path consists of awideband RF amplifier and a amplitude limitercircuit. The signal then is provided to a powerdivider, where it is split into two paths. Onepath goes directly to the phase/frequency dis-criminator, while the other is delayed a time, τ,before being applied to the phase/frequencydiscriminator. The outputs are sin ωt and cosωt detected signals. Figure 16.4 shows a blockdiagram for an IFM receiver capable of sepa-rating pulse and CW signals, while encodingboth of them.

MICROSCAN COMPRESSIVERECEIVERS

The compressive receiver gets its name be-cause a dispersive delay line (DDL) is used tocompress the input RF signal. It is also referredto as a microscan receiver, because a fast-sweeping local oscillator is used to convert theinput RF into a frequency modulated signal.

The compressive receiver is a widebandreceiver. The key components are shown inFigure 16.5. A swept local oscillator and a

Special Purpose Receivers 235

Fig. 16.3 Standard IFM receiver system.

RFAMPLIFIER

LOGARITHMICAMPLIFIER

A/DCONVERTER

DIGITALOUTPUT

LIMITERPOWERDIVIDER

t

FREQUENCYDISCRIMINATOR

DETECTORS

sin w t

cos w t

ANTENNA

mixer stage are followed by a weighted filter.The DDL follows the weighted filter and, inturn, is followed by the video detector. Thetime of the output pulse is a measure of thefrequency at the RF input.

Figure 16.6 shows the detector. Follow-ing the DDL is a logarithmic amplifier for im-proved dynamic range. A DC comparator isat the output of the log amp. A fixed thresh-

old is provided by the DC bias on the com-parator, which in turn produces noise-freepulses for the output.

BRAGG CELL RECEIVERS

A Bragg cell receiver (Figure 16.7) is a kind ofoptical processor that performs a Fourier


Fig. 16.4 IFM receiver capable of separating pulse and CW signals, while encoding both of them.

FREQUENCYDISCRIMINATOR

RFINPUT

BUFFERAMPLIFIER

BUFFERAMPLIFIER

BUFFERAMPLIFIER

BUFFERAMPLIFIER

X PULSE

X CW

Y PULSE

Y CW

Fig. 16.5Weighted filter/DDLsystem.

MIXER

LOCALOSCILLATOR

WEIGHTEDFILTER

DDLVIDEO

DETECTOR OUTPUT

ANTENNA

transform and thereby obtains the frequencyof the input frequency. In the Bragg cell re-ceiver, the input RF signal first is converted toa spatial pattern that could modulate a lightbeam supplied by a laser. The light is trans-mitted through a Bragg cell, then through a

Fourier transform lens and an electronic arraydetector.

The principal aspect of a Bragg cell re-ceiver is its small size and low cost. Electronicwarfare is the primary use of Bragg cell receivers.

Special Purpose Receivers 237

Fig. 16.6 DDL/log amplifier system.

DDLFROMMIXER

LOGARITHMICAMPLIFIER

DCBIAS

COMPARATOR

OUTPUT

FIXED THRESHOLD

Fig. 16.7 Bragg cell receiver system.

LASER

BEAMEXPANDER BRAGG

CELLCOLLIMETER

FOURIERTRANSFORM

LENS

DETECTORARRAY

RFINPUT

In this chapter, we look at some tests andmeasurements used on radio receivers. Butfirst, we look at signal generators. These de-vices make the signals used in testing of ra-dio receivers.

SIGNAL SOURCES AND SIGNALGENERATORS

Signal generators and signal sources are in-struments that generate controlled signals foruse in testing and measurement. A distinctionis made by some people between signalsources and signal generators. The formerproduce continuous wave (CW) output sig-nals without modulation, while the latter willproduce one or more forms of modulatedsignal (AM, FM, SSB, PM) in addition to CWoutput. In many cases, however, the wordssource and generator are interchangeable inpopular usage.

Some signal sources produce a single-output frequency (or a discrete number offixed output frequencies). These instrumentssometimes are used for testing channelizedreceiver systems. Other signal sources pro-

duce output over a very wide range of fre-quencies. Add a modulator stage to these in-struments and a signal generator is created.Figure 17.1 shows a typical commercial sig-nal generator.

Grades of Instrument

Signal generators and sources come in sev-eral grades. Which to select depends on theuse. A service-grade instrument is used fortroubleshooting common broadcast band re-ceivers. Such instruments often lack a cali-brated or metered output level control, andthe frequency accuracy usually is low. Moreimportant for many sensitive measurements,at low output levels more signal will escapearound the flanges than comes through theoutput connector. Such instruments are use-ful for simple troubleshooting but useless foraccurate measurements.

Laboratory-grade signal sources andgenerators are very high-quality instrumentswith accurate frequency readout and outputlevel controls. These instruments are used inmaking lab measurements of receivers andother devices where high precision and ac-curacy is desired.

Chapter 17

Receiver Tests and Measurements

239

High-quality service-grade instrumentsfall somewhere between the two previousgrades. Several mainline manufacturers ofhigh-quality laboratory signal sources andgenerators also manufacture “economy” linesthat fit this category. They are of consider-ably higher grade than the simple service in-struments but are not up to the lab grade.They often are used for troubleshootinghigh-quality telecommunications and land-mobile systems, where the highly accurateand precise measurements are not needed.

Output Level

The output of a calibrated signal generator isusually expressed in either microvolts (µV)or dBm (decibels relative to 1 mW in 50 Ω)or both µV and dBm. It is useful to have afeel for both forms of output level indication.One microvolt (1 µV) is 10–6 V, so when ap-plied across a 50 Ω resistive load, it producesa power level of

V2/R = (10–6 V)2/(50 Ω) = 2 × 10–14 W (17.1)


Fig. 17.1 Typical signal generators: (A) Digital readout synthesizer; (B) Analog readout signal generators;(C) Sweep generator; (D) Service-grade signal generators.

The 0 dBm reference level is one milli-watt (1 mW) dissipated in a 50 Ω resistiveload. This represents an applied voltage of

(17.2)

If the output level set dial on a particu-lar instrument is not calibrated in the correct

units, then the required unit can be calcu-lated using these methods.

To find the output power level in wattsor milliwatts from dBm is similarly simple:

(17.3)

To find the level in watts divide equa-tion 17.3 by 1000.

P mWdBm= 10 10/V PR W V= = =( . )( ) .0 001 50 0 2236Ω

Receiver Tests and Measurements 241

OUTPUT SIGNAL QUALITY

It would be nice if all signal sources and sig-nal generators were ideal; that is, the outputfrequency and output level were noiselessand perfectly calibrated. This never occurs,although the differences in these specifica-tions involve a principal difference betweenhigh- and low-quality instruments.

FREQUENCY

The important considerations regarding fre-quency are the range, resolution, accuracy,and (in automatic test equipment applica-tions) the switching speed.

Range

The frequency range is a specification of thefrequencies covered. In some cases, only oneor some small number of discrete frequen-cies are covered. In other cases, one or morebands of frequencies are provided.

Resolution

Resolution is the statement of the smallest in-crement of frequency that can be set. Onanalog instruments that have no counter, theresolution is poor. The resolution may (butnot certainly) be improved by adding a digi-tal frequency counter to measure the outputfrequency. On modern synthesizers, it is pos-sible to set frequency with extremely goodresolution.

Accuracy

Accuracy specifies to how nearly the set fre-quency matches the actual output frequency.Accuracy is a function of the set frequency(and how closely it can be set), FSet, long-term aging (τAging), and the time since last cal-ibration (τCal). Mathematically,

Accuracy = ± FSet × τAging × τCal (17.4)

EXAMPLE

A signal generator is set to 480 MHz andhas an aging rate of 0.155 ppm/yr. It has

been six months (0.5 yr) since the last calibration.

Accuracy = ± FSet × τAging × τCal

= ± (480 MHz) × (0.155 ppm/yr) × (0.5 yr)

= ± 37.2 Hz

Also, some random variation may enterthe output frequency. Figure 17.2 shows theuncertainty band around the set frequency.The actual output frequency, Fo, will be FSet ±Accuracy. The general practice is to calibratea signal generator on a six-month or annualschedule, depending on the use.

Switching Speed (Settling Time)

The switching speed is the length of time,usually in milliseconds or microseconds, re-quired for a synthesized signal source orgenerator to move to a new frequency whendigitally commanded to change. It is calcu-lated as the length of time for the error of the


Fig. 17.2 Frequency response with an uncer-tainty band.

UNCERTAINTY

FREQUENCY

F

AM

PLI

TU

DE

FSET

frequency or output level (or both) com-manded by the change to come into specifi-cation range.

Output Level

The output level can be expressed in voltage,power, or dBm notation. All are equivalent,although one or another will be preferred inmost cases. The most common method ofdescribing the output level is dBm. A typicalsignal generator or CW signal source willproduce output levels from −136 dBm to +10dBm (some go to higher levels), with an out-put level accuracy of ±0.50 dBm and a reso-lution of 0.01–0.02 dB.

As with frequency, certain factors affectthe accuracy of the actual output vs. the setoutput. Figure 17.3 shows a zone of uncer-tainty around the output level set. Any givensetting has PMin and PMax values. For example,if the only error is the accuracy discussedpreviously (e.g., 0.50 dB), a level set of, say,−10 dBm will produce an actual outputpower level of −9.5 to −10.5 dBm.

SPECTRAL PURITY

The output signal is not always clean.Although the purity of the output signal is adistinguishing factor that differentiates lower-and higher-quality generators, all generatorsproduce signals other than the one desired.Figure 17.4A shows a typical spectrum out-put that might be seen on a spectrum ana-lyzer. The main signal is a CW sine wave, soideally we would expect only a single spike,with a height proportional to the outputlevel. But a lot of other signals are present.

First, note that the main signal is spreadout by phase noise, random variation aroundthe main frequency. When integrated over aspecified bandwidth, such as 300–3000 Hz,the phase noise is called residual FM (Figure17.4B).

Second, harmonics are present in Figure17.4A. If the main signal has a frequency of F,the harmonics have frequencies of nF, wheren is an integer. For example, the second har-monic is 2F, and the third harmonic is 3F. Inmany cases, the 3F harmonic is stronger than


Fig. 17.3Amplitude response with an uncer-tainty band.

SET LEVEL UNCERTAINTY

AM

PLI

TU

DE

FREQUENCY

FO

F

the 2F harmonic, although in general thehigher-number harmonics are weaker thanlower-number harmonics.

Sometimes, there are subharmonics aswell. These are integer quotients of themain signal. Again, if the F is the main sig-

nal frequency, nF/2 represents the subhar-monics. Typically, unless something inter-feres with the output signal, subharmonicsare not as prominent. One thing that makessubharmonics prominent, however, is theuse of frequency multiplier or divider


Fig. 17.4 Output signal noise: (A) spectrum output; (B) residual FM.

+300 +3000-3000 -300

FO

RESIDUAL FM ISINTEGRATED PHASE

NOISE OVERSPECIFIED

BANDWIDTH

F nFnF/2 FOther

FREQUENCY

AM

PLI

TU

DE

~ -45 dB~ -65 dB

~ 30 dB

MAIN SIGNAL

PHASENOISE

SUBHARMONIC

HARMONIC

SPUR

A

B

stages (which is the case in many moderngenerators).

Finally, miscellaneous spurious signals(spurs) are found on some generators. Thesemight be from power supply ripple modulat-ing the output signal, parasitic oscillations,digital noise from counter or phase-lockedloop circuits, or other sources.

Harmonics and spurs usually are mea-sured in terms of decibels below the carrier(dBc), where the carrier is the amplitude ofthe main output signal. In general, the fewerare the unwanted components, the better thesignal source.

Phase noise warrants some special con-sideration. It usually is measured in terms ofdBc/Hz; that is, decibels below the carrierper hertz of bandwidth. This noise is concen-trated around the main signal frequency andnormally graphed on a log-log scale to per-mit compressing both close-in and further-out noise components on one graph.

ARCHITECTURES

Despite many different configurations, withdifferent “block diagram” representations,only a few architectures are used in designing

signal generators. Figure 17.5 shows a simpleanalog architecture that once was commonon even high-grade instruments and still iscommon on service-grade instruments.

The signal is generated in an L-C con-trolled variable frequency oscillator. The VFOtypically has a band switch for selecting dif-ferent frequency ranges. A calibrated tuningdial gives the user an approximate idea of theoutput frequency. However, because of driftand the mechanical aspects of calibrating thedial, these dials are not terribly accurate.

Some instruments have an output am-plifier, although for many decades even qual-ity signal generators lacked power amplifiers.The output of the VFO was fed directly to theoutput level control.

Service-grade generators of this archi-tecture usually have a crude form of outputlevel control. Higher-quality instruments, onthe other hand, have some variant of the out-put circuit shown in Figure 17.5. High-leveloutput sometimes is provided to permit theuser to route the signal to a frequencycounter so that an accurate determination offrequency can be achieved.

Two attenuators are in the output level-setting circuit. A coarse attenuator is used to seta relative output meter to some calibrated


Fig. 17.5 Simple analog signal generator.

VARIABLEFREQUENCYOSCILLATOR

OUTPUTAMPLIFIER

BANDSWITCH

CALIBRATEDFREQUENCY DIAL

0OUTPUTMETER

VARIABLECOARSE

ATTENUATOR

CALIBRATEDFINE

ATTENUATOR

CALIBRATED LOWLEVEL OUTPUT

HIGH LEVELOUTPUT

point. In most cases, the meter would be cali-brated with 0 in the center of the analog scale.The coarse attenuator is adjusted to center themeter pointer over the 0 point in the center ofthe meter. When this is done, the settings of thefine output attenuator are valid.

A synthesizer architecture is shown inFigure 17.6. This type of signal generator ismore modern and can produce very accurate,high-quality signals. This signal source hasthree main sections: reference section, fre-quency synthesizer, and output section. Figure17.6 breaks down each section into furthercomponents for easy analysis.

Reference Section

The reference section is at the very core ofthe signal generation process. It is an accu-

rate, stable fixed-frequency source such as acrystal oscillator. The frequency of the refer-ence section must be precisely adjustableover a small range so that it can be comparedto a higher-order standard, such as a cesiumbeam oscillator or WWVB comparator re-ceiver, for purposes of calibration.

Because it controls the frequency syn-thesizer, the stability of the reference sectiondetermines the overall stability of the signalgenerator. The stability of ordinary crystal os-cillators is reasonably good for many pur-poses but not for use in the reference sectionof a signal source. For that purpose, eithertemperature-compensated crystal oscillators(TCXO) or oven-controlled crystal oscillators(OCXO) are used. The TCXO typically ex-hibits crystal aging of better than ±2 ppm/yrand temperature aging of ±1 ppm/yr. The


Fig. 17.6 Frequency synthesizer.

DIVIDE BY N

REFERENCEOSCILLATOR



SECTION

EXT.REF. IN

REF. OUT

f INTEGRATORVOLTAGE


DIVIDE BY N

REFIN

DCCONTROL

OUTPUTPOWER

AMPLIFIERSAMPLER

ALCDRIVER

ALCMODULATOR

ALCRECTIFIER


OCXO is capable of 0.1 ppm/yr for agingand 0.01 ppm/yr for temperature.

The crystal oscillator usually is oper-ated at a frequency such as 5 MHz, butlower frequencies often are needed. To gen-erate these lower frequencies, a divide-by-Ndigital counter is provided. This circuit dividesthe output frequency by some integer, N, toproduce a much lower reference frequency.

Many signal generators provide ref. outand ext. ref. in capability (these connectionsoften located on the rear panel of signal gen-erators, so may be overlooked). The ref. outconnector provides the reference signal toother instruments or is used for calibrationpurposes. The ext. ref. in allows the use of anexternal reference source in place of the inter-nal reference section. This sometimes is doneto lock two signal generators that must worktogether or to substitute a much more accu-rate reference such as a cesium beam clock.

Frequency Synthesizer

The actual signal is produced in the frequencysynthesizer section, which is generated by avoltage controlled oscillator whose output iscompared to the reference signal. Voltagevariable capacitance diode (varactors) can beused for the VCO, as can surface acousticalwave oscillators (which are used at higher fre-quencies and in the microwave bands).

The frequency of the VCO is set by aDC control voltage applied to its tuning inputline. This control voltage is generated by in-tegrating the output of a phase detector orphase comparator that receives the referencefrequency and a divide-by-N version of theVCO frequency as inputs. When the two fre-quencies are equal, the output of the phasedetector is 0, so the VCO tuning voltage is atsome quiescent value.

But, if the frequency of the VCO drifts,the phase detector output becomes nonzero.The integrator output ramps up in the direc-tion that will cancel the frequency drift of theVCO. The VCO frequency is continuouslyheld in check by corrections from the inte-grated output of the phase detector. Thistype of circuit is called a phase-locked loop.

Suppose, for example, a signal sourcehas a reference of 5 MHz and is divided by20 to produce a 250 kHz reference. If thefrequency synthesizer divide-by-N stage isset for, say, N = 511, then the VCO outputfrequency will be 0.25 MHz × 511 = 127.75MHz. Band switching, operating frequency,and frequency resolution are controlled bymanipulating the reference frequency andVCO divide-by-N settings. In some cases,the frequency is entered by keypad, andthis tells the signal generator the values toset. Alternatively, “tunable” signal genera-tors may have a digital encoder shaft con-nected to a front panel control.

The noise component of the output sig-nal is composed of thermal noise and phasenoise. Of these two, the phase noise tends to dominate the performance of the signalsource. Noise from both the reference oscilla-tor and VCO phase contributes to the overalloutput phase noise. There is a 20 log (N)degradation of the phase noise performance ofthe signal source because of the divide-by-Nnature of the PLL. Fortunately, the bandwidthof the PLL tends to limit the contribution of theVCO phase noise to the overall phase noiseperformance.

Output Section

The output section performs three basicfunctions: It boosts power output to a speci-fied maximum level, it provides precisioncontrol over the actual output level, and itkeeps the output level constant as frequencyis changed.

The power amplifier is a wideband am-plifier that produces an output level of somevalue in excess of the required maximumoutput level (e.g., +13 dBm). A calibratedprecision attenuator then can be used to setthe actual output level to any lower value re-quired (e.g., −136 dBm to +13 dBm).

The accuracy of the output power set-ting depends on keeping constant the RFpower applied to the attenuator input, eventhough oscillators (including VCOs) tend toexhibit output signal amplitude changes asfrequency is changed. In older, manual signal


sources, the RF voltmeter at the input of theattenuator had a 0 center and the outputcoarse attenuator level could be set manuallyto 0, making the calibration of the fine atten-uator meaningful. Modern signal sources,however, use an automatic level control cir-cuit to accomplish this job.

The automatic level control modulatoressentially is an amplitude modulator con-trolled by a DC voltage developed by rectify-ing and filtering a sample of the RF outputlevel. The ALC driver compares the actualoutput level with a preset value and adjuststhe control signal to the ALC modulator in adirection that cancels the error.

MODULATORS

A signal generator is said to differ from a sig-nal source in that it will provide modulationof the CW signal. Although I believe this is adistinction without a practical difference, it isnonetheless a commonly held usage.

Amplitude Modulation

Amplitude modulation (AM) conveys intelli-gence over a radio carrier by varying theamplitude in accordance with the appliedaudio. The sine wave RF carrier is describedby a(t) = Ac sin (2πfct), where a(t) is the in-stantaneous amplitude, Ac is the peak ampli-fier, fc is the carrier frequency, and t is time.When another signal is used to vary the am-plitude of the carrier, amplitude modulationresults:

(17.5)

where

µ is the depth of modulation; fm is the modulating frequency;

and all other terms are as previously defined.You will see the terms linear AM and

logarithmic AM. The former is used when thedepth of modulation is expressed as a per-centage and the latter, when it is expressedin decibels (dB). Figure 17.7 shows a block

diagram of the output section of a signal gen-erator that offers amplitude modulation. Thedepth of modulation for most signal genera-tors is adjustable over at least 0–30%, whilesome offer 100% AM.

The amplitude modulator is called aburst modulator in some signal generatorsand placed between the ALC modulator andthe output power amplifier stage. The inputsignal used to modulate the RF carrier can bederived from an external source or providedby an internal function generator that pro-duces at least sine waves. The function gen-erator also might produce square waves,sawtooth waves, triangle waves, or pulses,depending on the nature of the signal gener-ator. When the CW mode is desired, themodulating source is turned off.

Note that two modulators are in Figure17.7: ALC modulator and burst modulator.The ALC modulator is designed to keep theoutput RF level constant, while the burstmodulator wants to vary the output RF levelin accordance with the applied audio signal.These two circuits, therefore, are in conflictwith each other. The ALC hold-off circuit isused to moderate this conflict.

Frequency and Phase Modulation

There are two basic forms of angular modu-lation: frequency modulation (FM) and phasemodulation (PM). In FM, the carrier fre-quency is varied in accordance with themodulating frequency; while in PM, the car-rier frequency remains constant but its phaseis varied. Both FM and PM are considered es-sentially the same. Consider the case of FM.When no modulation is present, the RF car-rier ( fc) remains constant. But, as a sine waveaudio signal is applied, the carrier frequencywill deviate from fc to a lower frequency (F1)on one peak of the modulating frequency( fm), and to a higher frequency (F2) on thealternate peak of fm. The deviation is the dif-ference between the carrier and either ex-treme; that is, +Fd = F2 – fc and –Fd = fc – F1.In most cases, signal generators use symmet-rical sine wave modulation, so +Fd = –Fd andis called simply Fd.

s t A f t f tc c m( ) sin ( ) [ sin ( )]= × +2 1 2π µ π


The peak voltage of the carrier remainsconstant, defined by

(17.6)

In FM, the depth of modulation is ex-pressed as the modulation index (β), whichis defined as the ratio of the deviation to themodulating frequency, or Fd/fm. The FMprocess produces a large number of side-bands, and at certain values of β, the carrierwill go to 0. The sidebands are described bymathematical entities called Bessell functions.In phase modulation, the modulation index(β) is related to the variation in the phase;that is, β = ∆φpeak.

Straight frequency modulation is cre-ated by varying the frequency of the oscilla-tor in the frequency synthesizer (see Figure17.8). Phase modulation, on the other hand,

sometimes is generated by using a reactancemodulator to vary the phase of the oscillatorsignal.

SWEEP GENERATORS

Sweep generators (sweepers) are used toproduce a signal that changes frequency overa specified range. Although the sweepernominally resembles an FM generator, thereare differences. For one thing, the sweeprange usually is quite a bit larger than the de-viation of an FM signal generator. Anotherthing is that the sweeper tends to change fre-quency from one limit to the other, thensnaps back to the first limit.

There are three basic forms of sweep:linear (“ramp”) sweep, stepped sweep, and

V V t f m tc= +( ) sin [ ( )]2π β


Fig. 17.7 Output section of a frequency synthesizer.



SECTION

EXT.REF. IN

REF. OUT

ALCDRIVER

ALCMODULATOR

OUTPUTPOWER

AMPLIFIERSAMPLER

ALCRECTIFIER


BURSTMODULATOR

ALCHOLD-OFF

BURSTMODULATOR

CONTROL

FUNCTIONGENERATOR

EXTERNALMODULATION

CW

S1

RFOUT

RFIN

list sweep. All these forms use a voltage-controlled oscillator frequency synthesizer;the difference is in the waveform used forsweeping the output signal.

Figure 17.9A shows linear sawtoothwaveform sweep. At time T1 the ramp ap-plied to the VCO is at 0 and begins rampingup. The frequency begins to move upwardfrom 0 V in a linear manner until time T2, atwhich point it snaps back to 0 V.

Stepped sweep (Figure 17.9B) uses aseries of discrete voltage steps to change thefrequency of the VCO. This method does notproduce an output on every frequency in theoutput voltage range from T1 to T2 but onlyat specific values determined by the steps.The steps are produced in a circuit such asshown in Figure 17.10. A digital-to-analog

converter produces an output voltage pro-portional to the binary number applied to itsinputs. The maximum output voltage is setby an internal or external DC reference volt-age and the applied binary number. In thiscase, the DAC input is driven by a binarycounter, which in turn is incremented by adigital clock (square wave generator).

The maximum number of steps that canbe accommodated using any given binarycounter is 2N, where N is the bit length of thecounter. For the 16-bit counter shown inFigure 17.10, therefore, a total of 216 = 65,536different levels (including 0) can be created.The maximum output voltage is less than thereference voltage by the 1 lsb (least signifi-cant bit) voltage, 1/2N. If a 16-bit counter isconnected to a DAC with a 10.00 V reference


Fig. 17.8 Frequency modulation and phase modulation in a frequency synthesizer.



SECTION

EXT.REF. IN

REF. OUT

f INTEGRATOR

DIVIDE BY N

REFIN

DCCONTROL

FREQUENCYMULTIPLIER

xFSVOLTAGE


FUNCTIONGENERATOR

d/dt

EXTERNALMODULATION

S1

S2

source, then the 1 lsb step is 0.00015 V andthe maximum output voltage is 10.00 –0.00015 V = 9.99985 V.

Many signal generators are used to test ra-dio communications transceivers. Transceiverscontain both transmitter and receiver, which

share a common output connector to the an-tenna. If the signal generator is connected tothe antenna connector to test the receiver, andsomeone accidentally keys the transmitter,then serious damage can result to the signalgenerator. High-quality instruments provide


TIME

VO

LTA

GE

T

V

T1 T2

Fig. 17.9Waveforms used forsweeping the output sig-nal: (A) sawtoothed;(B) stepped. TIME

VO

LTA

GE

T

V

T1 T2T0

A

B

Fig. 17.10DAC-binary counter cir-cuit for generating astepped waveform.

DIGITAL-TO-ANALOGCONVERTER

16-BIT BINARY COUNTERCLOCK

DCREFERENCE

OUTPUT

architecture with output reverse power protec-tion. This circuitry prevents damage to the sig-nal source if a high RF power level is appliedto its output connector.

List sweepers also have a DAC to con-trol the VCO but drive it with a series ofvalues stored in a digital memory similar tothose used in computers. This permits arbi-trary waveforms to be created.

Figure 17.11 shows one way sweepersare built: open loop. The frequency synthe-sizer loop is broken and a linear sawtooth,stepped waveform, or list-generated wave-form is applied to its DC control voltage input.A closed-loop approach is shown in Figure17.12. The frequency synthesizer loop is notopen; rather, a binary counter or list memory

is used to drive the divide-by-N counter in thesynthesizer PLL feedback loop.

IMPROVING THE QUALITY OFSIGNAL GENERATOR USE

Improving Spectral Purity

Certain high-quality measurements are verysensitive to extraneous signals coming out ofa signal generator. Many signal generatorsput out harmonics that are –30 dB downfrom the main signal (carrier), while othersignals may be either higher or lower thanthis level. The way to get rid of these extra-neous signals is to place a frequency selec-


Fig. 17.11 Open-loop sweeper.

f INTEGRATORVOLTAGE


DIVIDE BY N

REFIN

DCCONTROL

SWEEPWAVEFORMGENERATOR

Fig. 17.12 Closed-loop sweeper.

f INTEGRATORVOLTAGE


DIVIDE BY N

REFIN

DCCONTROL

BINARYCOUNTER

CLOCK

tive filter between the output of the signalgenerator and the device under test.

Figure 17.13 shows the use of a low-passfilter to eliminate the harmonics and any spursabove the main signal. Select a filter with a –3dB point somewhere between the main signaland the first extra signal, and an attenuationslope steep enough to reduce the “bad” signalsas much as possible. If there are any subhar-monics (or spurs lower than the main signal),then either use a bandpass filter or add a high-pass filter with a –3 dB cutoff between themain signal and the subharmonic.

Take care, however. Real filters lack thenice flat response seen in some textbooks.They have passband ripple and some odd re-sponses out of band. Also, L-C filters are no-

toriously unpredictable when you terminatethem in an impedance other than the designimpedance. Understand the passband re-sponse and the insertion loss of the filter be-fore using it.

Improving Mismatch Loss

Mismatch error occurs because the load and the signal generator are not impedancematched. In any electronic circuit, the maxi-mum power transfer occurs when the imped-ances are matched. An inherent mismatchproblem may be in either the signal genera-tor or the load and almost certainly in the ca-bles or other devices connected in line withthe signal generator.


Fig. 17.13 Use of a low-pass filter to eliminate harmonics and spurs.

DEVICE UNDERTEST

SIGNAL GENERATOR

FILTER

For example, assume we have a signalgenerator with a VSWR of 1.9:1, and a deviceunder test with a VSWR of 1.6:1 connected inthe normal way (Figure 17.14A). The mis-match loss can be found once we know thereflection coefficients.

The source coefficient is

The device under test is

The mismatch loss is

ML = 20 log [1 + (ρS ρD)]

= 20 log [1 + (0.31)(0.23)] = 20 log [1.07]

= (20)(0.03) = 0.6 dB

Figure 17.14B shows the way of dealingwith this problem. Insert a 10 dB fixed attenu-ator in line with the line between the signal

generator output and the DUT. You have toadjust the signal generator output level control10 dB higher than normal to compensate forthe extra attenuation. This works becausefixed resistive attenuators tend to be designedwith very low reflection coefficients. Supposewe have the same components in Figure17.14B as in Figure 17.14A but add an attenua-tor with ρA = 0.31. The mismatch loss becomes

ML = 20 log [1 + (ρS ρD (ρA)2)]

= 20 log [1 + ((0.31)(0.23)(0.31)2)]

= 20 log [1 + 0.0069]

= 20 log [1.0069]

= (20)(0.003) = 0.06 dB

Improving Third-Order InterceptPerformance

One of the most important specifications foran amplifier or radio receiver is the third-order intercept point (TOIP). This specifica-tion tells us something about the device’s

ρD

SWR

SWR=

−+

= −+

= =1 1

1 1

1 6 1

1 6 1

0 6

2 60 23

.

.

.

..

ρS

SWR

SWR=

−+

= −+

= =1 1

1 1

1 9 1

1 9 1

0 9

2 50 31

.

.

.

..


Fig. 17.14 Use of an attenuator to improve range: (A) DUT connected in the normal way; (B) solution tothe problem.

DUT

DUT

A

B

dynamic performance, especially in the pres-ence of multiple input signals. If you listen toany shortwave receiver, AM or FM BCB re-ceiver, or any scanner receiver, you will real-ize that most areas of the country are pollutedwith too many radio signals. When multiplestrong signals are received at the same time,receiver (or amplifier) nonlinearity occursand heterodyne products are created.

If F1 and F2 are two input signals (oneof which might be the desired signal), these“intermodulation” products will have fre-quencies equal to mF1 ± nF2, where m and nare integers. The third-order harmonics arethose in which m = 2 and n = 1, or m = 1 andn = 2 (i.e., m + n = 3), and these are the mostdifficult to handle. The worst case is usuallythe 2F1 – F2 and 2F2 – F1 third-order prod-ucts because they fall close to F1 and F2 andmay be within the device passband. A prob-lem with these products is that they increaseat a rate three times the number of decibels asthe fundamental signal. If F1 or F2 goes up 1dB, then the third-order products go up 3 dB.

Figure 17.15 shows the basic setup formeasuring the third-order intercept point (alsocertain other parameters). The two signal gen-erators produce F1 and F2. They are set toidentical output levels, usually quite high, suchas −10 or −20 dBm. Initially, the receiver istuned to one frequency (e.g., F1) and a refer-

ence level established equal to the minimumdiscernible signal or, in some procedures, anS1 signal level. The receiver is tuned to thethird-order product frequencies and the atten-uator decreased (raising the signal level) untilthe same reference level is produced. TheTOIP can be calculated from these data points.

But, look what happens if the signalgenerator is working properly (Figure 17.16).The spectrum of Figure 17.16A is what wehope to see. Frequencies F1 and F2 stand upsmartly above the noise level (which ideallyis quite low). Figure 17.16B shows the samespectrum with the third-order IM products.When you see this at the output of an ampli-fier or receiver being tested, you might as-sume that the IM products are generated inthe DUT—but not always. Sometimes, thesignal from one generator gets into the out-put stages of the other generator and causesan IM response due solely to the test setup.

A couple things can be done to preventthe problem. First, if the combiner used tomerge the signals into one line is a resistivestar circuit, only 6 dB of isolation lies be-tween the ports. Using a hybrid combinerwith more port-to-port isolation helps tre-mendously, because it reduces the signalreaching the other generator’s output stages.

Another fix is to insert 10 or 20 dB fixedattenuators in each signal line (Figure 17.17).


Fig. 17.15 Combining two signal generators for measuring the third-order intercept point and other parameters.


HYBRIDCOMBINER

SIGNALGENERATOR No. 1


DUT

These attenuators provide additional isola-tion between the signal generators. Ofcourse, you have to adjust the output levelsof the signal generators to overcome the ex-tra loss.

Take care, however. Be certain that thesignal generator output can be cranked to ahigher level without producing spurious out-put signals, harmonics, and other extraneoussignals. One of my signal generators workswell from 0 to 90% full output; but at outputlevels >90%, the spectrum blossoms with un-wanted signals.

Extending Upper Output Range

Signal generator output controls are calibratedin terms of output voltage, usually microvolts(µV) or millivolts (mV), or the power level(e.g., dBm, decibels relative to 1 mW in 50 Ω).A typical generator produces output levels upto some value from about 0 dBm or perhaps+20 dBm (or some value in between). Whatdo you do if the signal generator maximum is,say, +10 dBm and you need a signal level of+30 dBm (1 W)? Or, what if you have a signalgenerator like mine that is wonderful at lowerlevels but falls apart at higher levels?

The solution is simple and obvious:Amplify the output. But take care. Figure 17.18shows an external amplifier used to boost theoutput level of the signal generator. Becauseall amplifiers can become nonlinear and pro-duce a bit of harmonic distortion in their ownright, a low-pass filter is inserted in the pathbetween the amplifier output and the DUT.

Also make sure that the selected ampli-fier can do the job. Make sure the TOIP speci-fication of the amplifier is sufficiently high thatthe signal generator cannot overdrive it. Themaximum input drive level (usually specifiedin dBm) and the output power level (also ex-pressed in dBm or possibly watts) must be suf-ficient to handle the job. Otherwise, addingthe amplifier might add problems as well.

Reducing the Output Level

You might want to reduce the signal genera-tor output level. One reason for doing this isthat you need a very small signal at the DUTbut a higher signal to act as a reference or befed to a frequency counter.

Sometimes this is done when using anold analog signal generator that has an inac-curate analog frequency dial but a well-calibrated output attenuator. To get a higherlevel for the counter while providing a low-level signal to the amplifier or receiver beingtested, use a setup like that in Figure 17.19.In other cases, you simply need a lower sig-nal level than the generator can provide.

The attenuator should be a calibratedtype. You can obtain a continuously variable


Fig. 17.16 Signal generator spectrums: (A) ideal;(B) with third-order IM products.

F1 F2

IMPRODUCT

IMPRODUCT

F1 F2

F1 F2

IMPRODUCT

IMPRODUCT

A

B


Fig. 17.17 Solution to spectrum with third-order IM products.


HYBRIDCOMBINER



D.U.T.

ATTENUATOR

ATTENUATOR

Fig. 17.18 Improving the output level of a signal generator with an external amplifier.

DEVICE UNDERTEST

SIGNAL GENERATOR

FILTERAMPLIFIER

Fig. 17.19 Using an economy signal generator to get a high level for the counter while providing a low-level signal to the DUT.

DEVICE UNDERTEST

ECONOMY SIGNALGENERATOR

ADJUSTABLESTEP

ATTENUATOR

SHIELDED CAGE

RFVOLTMETER

FREQUENCYCOUNTER

calibrated attenuator, but these are costly (al-though some tend to come on the surplusmarket). A lower-cost alternative is to use aprecision step attenuator. These devices haveswitch selectable attenuation levels in vari-ous steps. The total attenuation is the sum ofall the individual attenuations. You can buildstep attenuators, but for precision work, youare well advised to buy one (the resistors forprecision attenuation levels are really oddvalues, although they can be approximated).

RECEIVER SENSITIVITYMEASUREMENTS

Radio receiver specifications can be verifiedusing test equipment and a few simple pro-cedures. Such tests are made to evaluate re-ceivers, troubleshoot problems, and verifyperformance. A number of receiver parame-ters are important, but perhaps the mostcommonly discussed one is sensitivity. Wenow look at how these tests are done.

Sensitivity

Receiver sensitivity is a measure of how wellthe receiver picks up very weak signals. Aswith most engineering measurements, thenotion of sensitivity is an operational defini-tion. In other words, standard proceduresyield coherent results by which different re-ceivers (or the same receiver before and afterrepairs) can be compared.

Sensitivity basically is a problem of sig-nal-to-noise ratio or, more properly, the sig-nal-plus-noise-to-noise ratio [(S + N)/N ]. Everyreceiver or amplifier has a basic noise level,consisting of the noise produced external tothe receiver and noise produced inside the re-ceiver. Even a receiver with its antenna inputterminated in a shielded matching resistor,rather than an antenna or signal generator,shows a certain amount of thermal noise.

One important consideration when mak-ing sensitivity measurements (or comparingreceiver sensitivity specifications) is band-width. Thermal and other forms of noise areGaussian distributed over all possible band-

widths. The value of the noise at any given in-stant depends on the bandwidth of the chan-nel. For most receivers, this means the IFselectivity bandwidth, although in some casesthe audio bandwidth is less than the IF, so thatnumber dominates.

Figure 17.20 depicts two different defi-nitions of SNR. Basically, you cannot hearsignals down in the noise. The minimum dis-cernible signal is operationally defined as thesignal level that is the same as the noise flooror the signal level 3 dB above the receiver’snoise floor. But that sensitivity is not veryuseful for most applications. Perhaps thereare some people who can discern a signalonly 3 dB above the noise floor, but mostpeople require a higher SNR to notice a sig-nal. Although some definitions use 6, 12, or20 dB, the standard for practical sensitivity isthe signal level that produces a 10 dB SNR.This definition is found on most CW, AM,and SSB receivers.

SIGNAL GENERATOR

The signal generator selected to make sen-sitivity measurements must have very highisolation figures. Most service-grade signalgenerators are useful for troubleshooting butunsatisfactory for sensitivity measurement.The reason is that signal escapes around thecabinet flanges and control bushings. If you


Fig. 17.20 SNR for two levels: 3 dB and 10 dB.

3 dB

10dB

MinimumDiscernible

Signal (MDS)

PracticalSensitivity

have a sensitive receiver or spectrum ana-lyzer, then you can detect this signal.

Want to give it a try? Connect a shieldeddummy load to the output of the signal gen-erator and turn the signal generator’s outputto 0. Connect a whip or wire antenna to thereceiver’s antenna input and tune the receiveracross the signal generator frequency withthe RF gain cranked all the way up.

The signal generator also needs cali-brated output control. The correct calibrationsare either in dBm (power decibels relative to 1mW in a 50 Ω load) or microvolts (µV). Somesignal generators have an output meter thatcan set relative output but become “cali-brated” if a calibrated step attenuator is con-nected between the output of the signalgenerator and the receiver under test. You canfind the exact level, if you can measure thehigh level output of the signal generator.

Laboratory-grade signal generators maybe beyond the means of many people, butthere is a relatively vigorous market in usedor surplus equipment. A number of sources

of such equipment are listed on the WorldWide Web. If you do not need the latest dig-itally synthesized signal generators, you canfind good signal generators at low cost.

Test Setup

Figure 17.21 shows the test setup for mostreceiver sensitivity measurements. The atten-uator is optional and may not be needed ifthe signal generator is adequately equippedwith a good-quality calibrated output attenu-ator. When measuring an AM receiver, setthe signal generator modulation for 30%depth and 1000 Hz.

The receiver output level is measuredusing an audio AC voltmeter. Ideally, the in-strument should be calibrated in decibels aswell as volts and have RMS reading capability.

The receiver must be correctly set up orthe measurement will be in error. In most testsetups, the receiver’s RF and AF gain controlsare turned to maximum and the squelch isturned off. Further, the automatic gain con-


Fig. 17.21 Test setup for most receiver sensitivity measurements.

RECEIVER UNDER TEST

AUDIO ACVOLTMETER

OUTPUTLOAD

SIGNAL GENERATOR

HYBRIDCOMBINER

DUMMYLOAD

COAXIALSWITCH

ANTENNA

trol must be either turned off or, in somemodels, clamped to a DC level according tothe manufacturer’s directions.

MINIMUM DISCERNIBLE SIGNAL SENSITIVITY

To measure the minimum discernible signal(MDS), we need to find the signal level indBm or µV that is 3 dB above the receivernoise floor, using the following steps:

1. Connect the equipment as in Figure17.21 and set the receiver and signalgenerator to the same frequency.

2. Turn the signal generator output all theway down to 0.

3. Crank the RF gain and AF controls all theway up (you may want to set the audiooutput control to a convenient level ifyou have no dummy speaker load).

4. To measure the MDS, first measure therms value of the noise (“hiss”) outputon the AC voltmeter, then increase thesignal generator output level until thereceiver output level increases 3 dB.

You can determine the numerical valueof the receiver noise floor by the same ap-proach. Measure the output noise level, thenfind the MDS by this procedure. The receivernoise floor level will be the same as the sig-nal generator output level (less any attenua-tion in the line).

STANDARD OUTPUT CONDITIONS

A sensitivity specification used for consumerradio receivers uses a standard output ap-proach. A typical receiver sensitivity specmight read “xx µV for 400 mW in an 8 Ω loadwhen modulated 30% by 1 kHz.” The sameequipment setup (Figure 17.21) can be usedfor this measurement. A power of 400 mW(0.4 W) in an 8 Ω load is the same as 1.789 Vrms, which can be read on the AC voltmeter.Use an 8 Ω noninductive resistor for the loadrather than the loudspeaker—the sound lev-els are pretty annoying. Adjust the signalgenerator output level for an rms output volt-age of 1.789 V and read the output level fromthe signal generator controls.

FULL-POWER SENSITIVITY

Some older receivers use the full-power sen-sitivity figure. This is the signal level that willproduce the full rated audio output power.Set the signal generator for 1 kHz modulationwith 30% depth. Tune the radio and signalgenerator to the same frequency and crankup the output until the audio output power isat the full-rated power level (e.g., 400 mW, 1W). The signal level that produces this condi-tion is the “sensitivity” of the receiver.

10 DB (S + N)/N TEST

The 10 dB test method is the same as the 3dB MDS method, except that the signal gen-erator level is increased until the output is 10dB above the noise floor.

An alternate method sometimes is usedon AM receivers:

1. Set up the signal generator and receiveras discussed previously.

2. Set the output of the receiver to produceat least 0.5 W audio output, or if therated output power is lower than 1 W,set it for at least 50 mW audio outputpower.

3. Turn off the modulation. If the audiooutput drops at least 10 dB, then thesignal generator setting is at the 10 dBS + N/N level. If the level drops lessthan 10-dB, then readjust the signalgenerator output level upward a smallamount and try again.

ON-SITE EFFECTIVE SENSITIVITY TEST

This test is done on-site only when the re-ceiver is installed. It is intended to get someidea of how well the receiver performs in itsactual installed environment (Figure 17.22shows the test setup):

1. Measure the 10 dB S + N/N sensitivityas discussed previously (see Figure17.21 for setup) and write down thenumber.

2. Connect the hybrid combiner, two-position coaxial switch, antenna, anddummy load into the circuit.


3. Set the switch to the dummy load andmeasure the sensitivity. It will be consid-erably worse than the 10 dB sensitivity.

4. Set the coaxial switch to the antennaand again measure the 10 dB sensitivity.It should be lower still.

The effective sensitivity is SNR10dB −(SNRLoad − SNRANT), where SNRANT is the SNRwhen the antenna is connected. The figureSNRLoad − SNRANT is the degradation factor. Forexample, suppose the 10 dB SNR is −122dBm, the SNR when the load is connected is−77 dBm, and when the antenna is connectedit is −70 dBm. The effective on-site SNR is

SNREFF = SNR10 dB − (SNRLoad − SNRANT)

= −122 dBm − [(−77 dBm) − (−70 dBm)]

= −122 dBm − [−7 dBm]

= −115 dBm

The effective sensitivity is valid for onlythe given site and conditions present whenthe test is performed. If the site or the noisegenerators and other signals present change,then the test must be repeated.

FM Receiver Sensitivity

There are two basic methods for measuringthe sensitivity of FM receivers: 20 dB quietingand 12 dB SINAD. The 20-dB quieting methodtypically is used on FM broadcast band re-ceivers and once was popular for communica-

tions receivers as well. Today, the 12-dBSINAD method is preferred.

20 DB QUIETING METHOD

The 20-dB quieting method relies on the ca-pacity of the FM detector to suppress noiseonce the limiting signal level is reached. Thewell-known capability of FM to eliminate noiseis because most noise amplitude modulates thecarrier. If the amplitude can be clamped belowthe level where the noise is effective, then thefrequency variations can be detected to recoverthe audio. This effect is called quieting, the re-duction of noise as the signal level increases.

To measure the 20 dB quieting sensitivity,

1. Connect the receiver and signal gener-ator as in Figure 17.21. Keep the signalgenerator output at 0. The modulation(deviation) should be set to whateveris appropriate for the class of receiverbeing measured.

2. Turn the RF gain all the way up. Setthe audio output to produce a conve-nient reading in the high end of the AC voltmeter scale.

3. Measure the output noise level andwrite it down.

4. Turn up the signal generator outputlevel until the reading on the AC volt-meter drops 20 dB. The signal genera-tor output level that accomplishes thisis the 20 dB quieting sensitivity (typi-cally less than 1 µV).


Fig. 17.22 Setup for on-site test of receiver sensitivity.


RECEIVER UNDER TEST

AUDIO ACVOLTMETER

SIGNAL GENERATOR

OUTPUTLOAD

SINAD SENSITIVITY

The sensitivity of FM receivers is often ex-pressed in terms of SINAD (signal-noise dis-tortion). This approach to the signal-to-noiseratio recognizes that the problem of detec-tion depends on not simply signal and noiselevel but also distortion. The SINAD methodis described by equation 17.7:

(17.7)

In terms of decibels, the following equationis used:

(17.8)

where

SINAD(dB) is the SINAD sensitivity ex-pressed in decibels (dB);

VSignal is the output voltage due to thesignal;

VNoise is the output voltage due to noise;VDistortion is the output voltage due to

distortion.

The standard 12 dB SINAD sensitivitycorresponds to a 4:1 SNR, in which the sum

of noise and distortion is 25% the signal volt-age. As signal levels get higher, the SINADand 10 dB SNR values tend to converge.

Figure 17.23 shows a typical test setupfor the SINAD measurement. The output ACvoltmeter is augmented by a total harmonicdistortion analyzer, both of which measure theoutput signal across the audio load (speaker,load resistor, etc.). To test SINAD sensitivity,

1. Set the signal generator frequency andreceiver frequency to the same value.

2. Set standard conditions: modulating fre-quency 1 kHz sine wave, deviation setto 60% the peak deviation used for thatservice. For example, for an FM BCB re-ceiver, deviation is ±75 kHz, so set thesignal generator deviation to 0.6 × ±75kHz = ±45 kHz. For a communicationsreceiver designed for ±5 kHz deviation,set deviation to 0.6 × ±5 kHz = ±3 kHz.

3. Adjust the receiver audio output to ap-proximately 50% the receiver’s ratedaudio output.

4. Adjust the signal generator output untilthe input signal is high enough to pro-duce 25% distortion. This is the 12 dBSINAD sensitivity.

SINAD (dB)

Signal Noise Distortion

Noise Distortion

=+ +

+

log20

V V V

V V

SINAD

Signal Noise Distortion

Noise Distortion=

+ ++


Fig. 17.23 Setup for on-site test of SINAD sensitivity.


FM RECEIVER UNDER TESTAUDIO

TOTAL HARMONICDISTORTION ANALYZER

SIGNAL GENERATOR

AUDIO ACVOLTMETER

OUTPUTLOAD

Special SINAD sensitivity meters areavailable that combine the THD analyzer andaudio voltmeter functions in one instrument.

RECEIVER NOISE MEASUREMENT

Radio reception is a problem of the signal-to-noise ratio. For this reason, it sometimes isnecessary to measure noise levels and thenoise performance of receivers.

The noise floor of a receiver is thesame as the minimum detectable signal andmeasured in the same way. The MDS is de-fined as the signal level that causes an out-put 3 dB above the noise floor. Hence, whenyou measure the MDS you also measure thenoise floor.

Setting standard conditions is necessaryto take this measurement properly. Forexample, one caution is to use a signal levelat least 10 dB above the expected sensitivityof the receiver, then work down to find the 3dB increase level.

A standard condition is to ensure that thenoise floor measurements are made in a stan-dard bandwidth. Often, receivers with multiple

bandwidths specify the noise floor and sensi-tivity in the narrowest available bandwidth.However, that may not be useful if the normalmode for that receiver requires a wider band-width. For example, an H.F. communicationsreceiver may have filters for AM mode (BW = 6kHz), single-sideband (BW = 2.8 kHz), andCW (BW = 500 Hz). If the mode required foryour application is SSB, then do not rely onsensitivity and noise floor measurements madeon the 500 Hz bandwidth for the CW mode.

1 dB Compression Point

The 1 dB compression point is the input sig-nal level at which the receiver gain drops 1dB (Figure 17.24). The gain of the system isdepicted by the Pout/Pin line. This charac-teristic is linear up to a point; that is, a 1 dBincrease in input signal level causes a propor-tionally scaled output level change. At somepoint, however, the receiver is saturated andcannot accommodate any further input signal.The operational definition of where this oc-curs is the 1 dB compression point.

To measure this point, use the standardtest setup of Figure 17.21. Bring the input


Fig. 17.24The 1 dB compression point. Input Level (Pin )

Out

put L

evel

(P

out)

1 dB

power level applied to the receiver antennaterminals up from some low value in 1 dBsteps until the receiver output level drops 1dB. The input level at which this occurs isthe 1 dB compression point.

Dynamic Range

Dynamic range represents the total range ofinput signal levels that can be accommo-dated. The classical dynamic range measureis the blocking dynamic range, but the third-order IMD dynamic range is used as well.

BLOCKING DYNAMIC RANGE

The blocking dynamic range (BDR) mea-sures the difference between the receivernoise floor and the level of an off-channelundesired signal to reduce the sensitivity toon-channel signals by a specified amount(Figure 17.25). In other words, it is a measureof the range from MDS to a specified desen-sitization of the receiver.

Connect the test setup of Figure 17.26.Two signal generators are coupled throughan isolating hybrid to a step attenuator thento the receiver. An AC audio voltmeter isused to measure the output level of the re-ceiver. Figure 17.25 shows the signal situa-tion for the receiver input. Frequency F1 is

the desired signal, while F2 is the interferingsignal.

The amplitude of F1 should be set suchthat it is above the receiver MDS by at leastthe mount of the required minimum SNR(e.g., 10 dB) but below the point where it be-gins to increase IMD products. Using a higherlevel minimizes the noise error contribution.The exact level depends somewhat on thetest procedure. If the receiver’s automaticgain control can be disabled, then set the sig-


Fig. 17.25 The blocking dynamic range.

DF

F1 F2

DESIREDSIGNAL

INTERFERINGSIGNAL

Fig. 17.26 Setup for testing the blocking dynamic range.


HYBRIDCOMBINER



RECEIVER UNDER TEST

AUDIO ACVOLTMETER

nal level about 10 dB below the 1 dB com-pression point. If the AGC cannot be dis-abled, then the signal level should be lower,say, about 20 dB above the noise floor.

Frequencies F1 and F2 must have somespecified spacing (∆F). In most cases, an HFreceiver will call for a 20 kHz spacing, whilea VHF/UHF receiver will call for a 20 kHz or100 kHz spacing. Some special purpose mi-crowave receivers sometimes have consider-ably larger spacing. Whatever the case, thesame spacing should be used on different re-ceivers when comparisons are made, be-cause it seriously affects the results.

At the start of the measurement, the F1is turned on, F2 is turned off, and the re-ceiver is adjusted for maximum output of F1.Turn on F2 at some level, such as −100 dBm.Increase the level of F2 in 1 dB steps untilthe output level of the receiver drops 1 dB.The level at which this occurs represents theblocking dynamic range.

If this test is performed on automatic testequipment, then the output levels are auto-matically stepped. Also, in some cases, the fre-quencies also are stepped up and down theband, although maintaining the frequencyseparation ∆F. Adequate settling time betweenmeasurements must be programmed into theprotocol. The signal generator setting timerarely dominates this case, but be wary of theAGC settling time. AGC circuits have a timeconstant, and it might be a relatively long pe-riod (e.g., 5 sec). Understand the AGC timeconstant before programming a frequency orsignal level stepping protocol.

THIRD-ORDER IMD DYNAMIC RANGE

A test setup similar to that for BDR is used for the third-order IMD dynamic range(TOIMDDR). Set the signal generators for aconvenient output level such as −20 dBm andthe frequencies in band with a specified ∆Fspacing. Set the attenuator for a high degreeof attenuation (low signal level). Increase thesignal by decreasing attenuation in 1 dB stepsuntil a third-order response appears in theoutput. The actual signal level applied to thereceiver, as usual, can be calculated from thesignal generator and attenuator settings. The

difference between the MDS and the level de-termined by this procedure is the TOIMDDR.

A Cautionary Note

In both this measurement and the IMD mea-surement (discussed next), it is important toensure that only the receiver IMD productsare measured. Anytime a receiver or amplifieris overdriven into a nonlinear operating re-gion, the possibility exists for the creation ofunwanted distortion products. Even a singlesignal can produce first-order effects. If, forexample, frequency F is applied to the inputin sufficient power to overload the front end,then harmonics at 2F, 3F, and so forth mightbe generated. Similarly with normal mF1 ±nF2 IMD products higher than first order.

Several mechanisms exist and must beaddressed. First, the hybrid coupler could benonlinear. This can occur when broadbandferrite or powdered iron-core transformers areused in the hybrid coupler. The coupler mustproduce very low levels of IMD or these willaffect the results of the test. Typically, a third-order intercept point (TOIP) of >50 dBm is re-quired of the coupler. If the hybrid couplerbeing used has a lower TOIP or if the receiverhas a particularly high TOIP (>45 dBm), thenuse no test configuration that places attenua-tion between the receiver input and the out-put of the coupler. That configuration forcesthe coupler to operate at too high a level.

Or, the signal from one signal generatorcould enter the output circuits of the other.Most amplifiers produce IMD products whensignals arrive from the outside. This problem isreduced to negligible levels if the signal gener-ator outputs have a high degree of inherentisolation, if attenuation is used between thecoupler and the signal generators, or if the in-put-to-input port isolation of the hybrid ishigh. Try to get as much isolation as possiblebetween signal sources and between thesources and the receiver (>90 dB). Also pru-dent is using a signal generator with a high-level output (approximately +15 to +22 dBm).

Another problem with coupling betweensignal generators is that many high-quality sig-nal generators employ a feedback-controlled


automatic level control that samples and rec-tifies the RF output signal and then feedsback the derived DC control voltage to anamplitude modulator. When external signalsarrive, they can affect the ALC in two ways.First, the extra signal level may influencefeedback levels. Second and more likely, theIMD products may get into the feedback sys-tem, causing beat notes that modulate theregular output signal.

Intermodulation Distortion

Intermodulation distortion occurs when twofrequencies, F1 and F2, combine to produceheterodyne products that were not in theoriginal set of signals. Figure 17.27 showsthis behavior. When F1 and F2 are suffi-ciently strong, the receiver becomes nonlin-ear so mixing will occur. When this happensIM products rise up out of the noise.

Second-order IM products (F1 ± F2) tendto fall outside the band, so they are filteredout easily. Third-order IM products (2F1 ± F2and 2F2 ± F1) usually are outside the band.

Several methods are used to measure theIMD performance of a receiver. Figure 17.28shows one standard setup. Two signal gener-ators are used to provide the two different sig-nals required for the IMD test. Each signalgenerator is equipped with an adjustable at-tenuator, which may or may not be external tothe generator. In some cases, both internaland external attenuators may be used.

Optional bandpass filters sometimes areused to clean up the signal generator outputspectrum. These filters are used to suppressharmonics of the output frequency. If the sig-nal generator has sufficiently low harmonicoutput, then these filters can be eliminated.Keep in mind that some filters use ferrite orpowdered iron cores, so they may saturateand cause IMD products of their own. Thetwo signals are combined in a two-port hy-brid. Following the hybrid is another attenu-ator. This attenuator supplies signal to thereceiver input.

The output signal is monitored by anyof several means. Some procedures use theaudio AC output level, as measured by an ACvoltmeter. In others, the spectrum of the au-dio output signal is measured using a spec-trum analyzer. An alternative might be to usea frequency selective voltmeter (wavemeter).The last method is out of favor because spec-trum analyzer prices have fallen significantly.Some people will use the receiver’s S meter(if it has one) to make this measurement. Stillothers couple the IF signal to an RF/IF spec-trum analyzer, which may show more infor-mation but requires entry inside the receiver.The other methods treat the receiver as a“black box,” requiring no modification of, orentry into, the receiver. The IMD test is bestrun in one of the linear reception modes(SSB or CW), but that is not always possible(e.g., when the receiver is FM or AM only).

AUDIO SIGNAL LEVEL METHOD

The audio output level is monitored on anaudio spectrum analyzer (or measured on a

Tidbit: How do you tell if the IMD is due tothe receiver or the test fixture? In most cases,reducing the signal levels will give you theanswer. If the ratio of desired to IMD re-sponses changes when the input signal levelto the receiver is changed, then it is a reason-able assumption that the IMD is generated in-side the receiver. If the ratio does not change,then it is probably due to the test fixture.


Fig. 17.27 IMD products (close-in frequencies).

F1 F2

IMPRODUCT

IMPRODUCT

wavemeter). The signal levels are turned upuntil the IMD product being investigatedrises from the noise level.

The spectrum analyzer method can beparticularly useful for measuring productsbelow the noise floor of the receiver. Recallthat the noise floor is proportional to band-width. Typical bandwidths vary from 500 Hzfor CW receivers to 200 kHz for FM broad-cast receivers (and more for microwave radarreceivers). If the bandwidth filter on the au-dio spectrum analyzer is set to some narrowvalue, such as 5 or 10 Hz, then the noise ismuch lower, so low-level IMD problemsshow up better.

SIGNAL-TO-NOISE RATIO METHOD

The SNR method of measuring IMD uses ei-ther an audio signal-to-noise ratio meter or aSINAD meter. The audio output is set to pro-duce a 1 kHz signal for this method. Caremust be exercised to prevent excess noisecontribution from the signal generator outputnoise. This noise is indistinguishable from re-

ceiver noise, so it makes the IMD lookworse. Also, AGC action could interfere withthis test.

S METER METHOD

In this method, the level of the IMD productis noted on the receiver’s S meter. A refer-ence signal is then provided that matches theS meter reading. This yields the level of theIMD product. Problems with this method in-clude the tendency of some receivers tocompress gain when the signal level gets to alevel above S9 or S9 + 10 dB. However, thisis a better method than some for measuringthe IMD performance of receivers with veryhigh IMD performance.

STANDARD METHOD

The usual method for measuring the IMDperformance is to set the signal generators tosome convenient high-level output (e.g., −20dBm). Select test frequencies (F1 and F2) andcalculate the third-order products (2F1 + F2,2F1 − F2, 2F2 + F1, and 2F2 − F1).


Fig. 17.28 A standard setup for a complete receiver test.

OPTIONALADJUSTABLEATTENUATOR



HYBRIDCOMBINER

BANDPASSFILTER

BANDPASSFILTER



RECEIVER UNDER TEST

AUDIO ACVOLTMETER

OPTIONAL AUDIOSPECTRUM ANALYZER

OUTPUTLOAD

FREQUENCY SELECTIVEAUDIO VOLTMETER

(WAVEMETER)

Set the receiver to a channel frequency,F1. If possible, turn off the AGC or clamp it toa low value (highest receiver gain), if possible.If the receiver uses a front-end RF or IF atten-uator, then set that to 0 dB. If an RF preampli-fier is used, turn if off. Adjust both the receivertuning and F1 to the same frequency andmaximize the receiver output. Set the secondsignal generator to F2, a specified spacing(e.g., 20 kHz) away from F1. Set both signalgenerators to a convenient output level, suchas –10 dBm. Set the in-line attenuator to thehighest setting (most attenuation).

Once the setup is complete, turn off thesignal generators and measure the receiveroutput noise level on an AC audio voltmeter.Turn on signal generator F1 and decrease theattenuator setting in 1 dB steps until the outputnoise level of the receiver increases 3 dB. Thisis the minimum discernible signal referencelevel. Return the attenuator settings to maxi-mum. Record this signal level as PIM = −10dBm – (Attenuator setting).

Tune the receiver to either of the close-inthird-order product frequencies (2F1 − F2 or2F2 − F1), while leaving the signal generatorsat F1 and F2 (both −10 dBm output). Reducethe attenuator setting until the receiver outputresponse at this frequency increases 3 dB (thesame as the reference MDS). Record this levelas PA = −10 dBm – (Attenuator setting).

(17.9)

where

IPN is the intermodulation product oforder N;

N is the order of the intermodulationproduct;

PA and PIM are signal power levels indBm

EXAMPLE

A 162.55 MHz receiver is tested using fre-quencies F1 = 162.55 MHz and F2 = 162.57MHz. The close-in third-order products are2F1 − F2 = 162.53 MHz and 2F2 − F2 =162.59 MHz.

1. The minimum discernible signal at162.55 MHz requires an attenuator set-ting of –89 dB, so PIM = (−10 dBM – 89dB) = −99 dBm.

2. The response at 162.53 MHz required19 dB of attenuation for the third-orderresponse to equal the MDS level, so PA

= (−10 dBm – 19 dB) = −29 dBm.

Because this is a third-order response, N = 3 so

Once the PA and PIM points are found, any IPcan be calculated using equation 17.9.

Selectivity Testing

Selectivity is a measure of the receiver’s abil-ity to reject off-channel signals. It is mea-sured in terms of bandpass, so it has theunits of frequency. The operational specifica-tion of selectivity is the bandpass (Figure17.29A) between the two points (F1 and F2)where the frequency response drops −3 dBfrom its midband point (Fo).

CW METHOD

It is possible to use a standard CW generatorto take this measurement, although it is te-dious and may not yield the best informa-tion. Set up a signal generator and outputindicator as per Figure 17.21. Center the sig-nal generator frequency inside the receiverpassband and adjust the receiver output for aconvenient level indicated on the AC volt-meter. Adjust the signal generator below Fo

to a point where the output drops the speci-fied amount. This frequency is F1. Repeat theprocedure above Fo until F2 is determined.The receiver bandwidth is F2 − F1.

SWEEP METHOD

The CW method produces a raw indication butlacks certain information, at least in its simplest

TOIP[3 ( 29 dBm)] ( 99 dBm)

3 187 dBm 99 dBm

212 dBm

26 dBm

=× − − −

−

=− +

=+

= +

IP

NP P

NNA N=

−−

IM

1


form. We are not simply interested in the F2 −F1 value but also in the shape of the passband.Ideally, a receiver passband is flat inside thepassband, rolling off gently on the upper andlower ends. This may not be the actual case.By using the sweep method, we can deter-mine the shape of the passband as well as itsbandwidth. Surprises can be lurking here, sothe information is needed. To get shape databy the CW method requires collecting a lot ofdata points and manually graphing them. The

sweep method produces an oscilloscope tracethat can be photographed.

Figure 17.29B shows the test equip-ment setup for a simple sweep test. A sweepgenerator will sweep through a specifiedband of frequencies. In the simplest form, it consists of a voltage-controlled oscillatorwith its tuning voltage in the form of a saw-tooth waveform (modern sweepers are a bitmore sophisticated but the idea is the same).The same sawtooth can be used to control


Fig. 17.29 Selectivity testing: (A) frequency response; (B) use of a marker generator.

V

H

SAWTOOTHGENERATOR

SWEEP GENERATOR

MARKERGENERATOR

HYBRIDCOUPLER

RECEIVER UNDER TEST

OSCILLOSCOPE

F1 F2FO

A

B

the horizontal deflection of an oscilloscope.The sweep signal is applied to the receiver’santenna input, while the receiver output isapplied to the vertical deflection input of theoscilloscope. The result is a trace represent-ing amplitude vs. frequency.

A marker generator can be used to indi-cate specific frequencies within the passband(typically, F1, F2, and Fo). In some cases, dis-crete frequencies are used, while in others theharmonics of a standard frequency may beused. For example, a 1 kHz marker generatorthat is sufficiently rich in harmonics will pro-vide markers every 1 kHz throughout the pass-band. However, harmonic markers are limitedin that their upper harmonics may not bestrong enough to produce the desired display.

The sweep rate setting deserves somecomment. The sweep rate is the repetitionrate at which the signal passes through theswept band. This attribute is controlled bythe frequency of the sawtooth generator. Ifthe sawtooth rate is too slow, then the dis-play will flicker and be hard to read. If it istoo high, then there may be ringing in the re-ceiver’s IF filters, causing a distorted reading.Flicker fusion tends to occur in the 8–10 Hzrange for most human operators, so the min-imum sweep frequency will have to be atleast these values. When you get above 40Hz or so, the danger of causing a ringing re-sponse in high-Q IF filters becomes morelikely. Therefore, select a sweep frequencybetween roughly 8 and 40 Hz for manuallyoperated measurement systems.

Squelch Tests

Communications receivers often are equippedwith a squelch circuit. These circuits turn offthe receiver’s audio output when no signal isbeing received. Noise occupies the bandwidthwhen no signal is received, and it is uncom-fortable to hear. When the squelch circuit de-tects noise, rather than signal, it turns off thereceiver output. The operator sets the squelchfrom the front panel.

The two levels of squelch are criticalsquelch and tight squelch. A control set to

the critical squelch level just barely quiets thereceiver when no signal is being received.Indeed, when a particularly high level noiseburst is heard, it might degrade the squelchenough to pass through momentarily. Tightsquelch requires a much larger signal tocause break through; here, the squelch con-trol is set to maximum.

CRITICAL SQUELCH

Connect a signal generator to the antenna in-put of the receiver. Adjust the signal genera-tor frequency on channel.

For AM receivers, set the signal genera-tor modulation depth to 30% with a modulat-ing frequency of 1000 Hz. For FM receivers,use a deviation approximately 60% the nor-mal deviation for that receiver, with a 1000Hz modulating frequency. Then:

1. Set the signal generator output level tothe lowest possible level so that no sig-nal is heard in the receiver output.

2. Set the squelch control to the pointwhere the output noise just disappearswhen no signal is present.

3. Bring the signal generator’s outputlevel up very slowly until the squelchbreaks. Record the output level as thecritical squelch point (dBm or µV).

TIGHT SQUELCH

The tight squelch test is identical to the criti-cal squelch test but with the squelch controlon the receiver set to the maximum (mostsquelched) position. The signal level re-quired to break tight squelch will be muchlarger.

SQUELCH RANGE

The squelch range lies between criticalsquelch and tight squelch. It may be ex-pressed in signal level units (dBm or µV) orin dB. For example, the following signal lev-els lead to the squelch range listed here:

Critical Squelch: 0.4 µV (−115 dBm)Tight Squelch: 58 µV (−71 dBm)Squelch Range: 57.6 µV (44 dBm)


Planning a receiver system is one of thosephrases that has at least two meanings. First,it could mean selecting a receiver system foruse. Second, it could mean planning a radioreceiver design and architecture. It is the lat-ter meaning that we mean by planning a re-ceiver system.

WHAT IS IMPORTANT

No parameter of the receiver is unimportant,but several have predominance. First is thesensitivity required of the receiver. Whetherit is described in terms of microvolts (µV) ordBm to achieve a given signal-to-noise ratio,the sensitivity is important. Second, the dy-namic range of the receiver is very impor-tant. The dynamic range is the distance indecibels from the minimum discernible sig-nal to the –1 dB compression point. Third isthe matter of selectivity. The bandwidth ofthe IF amplifier comes into play at this point.Then it gets a little messy. The –1 dB com-pression point and the third-order interceptpoint (TOIP) certainly are very importantand related to the dynamic range. The SNR,

as well as the absolute noise level of the re-ceiver, is important—in fact, it may be themost important of all of these factors.Finally, where to locate the inevitable spuri-ous responses is a concern.

An ideal receiver would have a 60dBm TOIP, 0 dB noise figure (NF), and nospurious responses, but this is unreason-able to expect. The design of the receivermust trade off a number of factors toachieve its goals.

A SUPERHETERODYNE RECEIVER

Figure 18.1 shows a block diagram for a highfrequency shortwave receiver. The stagesfrom the RF input are discussed next, and thegains and noise figures to be expected ineach stage are shown in Figure 18.2.

20 dB Attenuator

A 6, 12, or more likely a 20 dB attenuator isprovided to reduce the signal strength of in-put signals. The attenuator typically can beswitched in and out of the circuit.

Chapter 18

Planning a Receiver System

271

LP or BP Filter

Some form of input filtering is needed to im-prove the image rejection of the receiver aswell as suppress spurious signals. Two de-signs are used. A 32–35 MHz low-pass filtermay be used to keep the response withinrange. Or, individual bandpass filters may beused instead of the low-pass filter.

RF Amplifier

If an RF amplifier is used, it will have a lowgain (3 to 20 dB) and relative broad selectiv-ity (maybe a wideband amplifier). There is aschool of thought that one should not be

adding unnecessary noise to the circuit priorto the selectivity, thus they eliminate the RFamplifier.

Mixer/High LO

This mixer will up-convert the 0.54–30MHz signals to some frequency in thevicinity of 45–77 MHz, with 50 MHz beingvery common.

Bandpass Filter

The bandpass filter is the high IF filter andwill be on the up-converted frequency (e.g.,50 MHz).


Fig. 18.1 HF shortwave receiver block diagram.

20-dBATTENUATOR

LP OR BPFILTER

RFAMPLIFIER

HIGH LOCALOSCILLATOR

BANDPASSFILTER

HIGH IFAMPLIFIER

LOW LOCALOSCILLATOR

2nd IF BPFILTER

LOW IFAMPLIFIER

DEMODULATOR

AUDIOPREAMP

AUDIO POWERAMPLIFIER OUTPUT

RFINPUT

MIXER

MIXER

Fig. 18.2 Gain and noise figures.

Power Gain

Noise Figure

IP2

IP3

LP/BP FILTER RF AMPLIFIER HIGH MIXER/LO BP FILTER HIGH IF AMPLIFIER LOW MIXER/LO 2nd IF FILTER LOW IF AMPLIFIER

-1

33

X

X

10

X

80

3.6

-6

X

X

-6

-6

X

X

-6

20

X

X

2

-1

X

X

X

-6

X

X

X

>90

X

X

4

dB

dB

dBm

dBm

High IF Amplifier

The high IF amplifier typically is low gain (20dB) and may be eliminated in some designs.

Mixer/Low LO

This mixer/LO circuit down-converts the45–77 MHz signal from the high IF to somefrequency such as 10.7 MHz, 9 MHz, 8.83MHz, or 455 kHz.

Second IF BP Filter

The second IF bandpass filter provides themain selectivity of the receiver, and its band-width dominates the performance of the re-ceiver. It may be switchable, with bandpassessuch as 270 Hz, 500 Hz, 2.8 kHz, 4 kHz, and6 kHz.

Low IF Amplifier

The low IF amplifier provides the bulk of thegain of the receiver. Typically, 60–100 dB ofthe overall gain will be in this amplifier.

Demodulator

The demodulator will demodulate or detectthe signal, recovering the impressed audio.The demodulator may be FM/PM, SSB/CW,or AM.

Audio Preamplifier

The audio preamp builds the small signaloutput from the demodulator circuit to apoint where it will drive the audio powerstage. Typically, the audio preamp and theaudio power amplifier are embedded in thesame chip today.

Audio Power Amplifier

The audio power amplifier builds the signalfrom the audio preamp to a level where itcan drive the loudspeaker or earphones.

NOISE IN THE RECEIVER

All radio reception faces the problem of sig-nal-to-noise ratio control. Indeed, when sig-nals are weak, the SNR is of paramount

importance to the success of the system. Acertain amount of noise is contributed fromsources outside the receiver, but here weconcern ourselves with the noise generatedinside it. A basic level of noise is inherent toany circuit, equal to the thermal noise of theresistive component of its impedance. Thisnoise is

(18.1)

where

PN is the noise power in watts (W);K is the temperature in Kelvin (K);B is the bandwidth in hertz (Hz);R is the resistance in ohms (Ω).

This is the least noise that will be pre-sent in the circuit. But, transistors and inte-grated circuits add noise of their own, andthis noise is likely to be much higher thanthe thermal noise. In general, in a cascadecircuit, the noise is given by Friis’s equation:

(18.2)

where

F is the overall noise figure;F1, F2, F3, F4 are the noise figures of

the successive stages;G1, G2, G3 are the gains of the succes-

sive stages.

Clearly, the noise factor of the initialstage in the cascade chain dominates thescene. It is important to keep the noise figureof the first stage or two in the chain very lowto lower the overall noise figure. This onefactor argues for a low noise amplifier (LNA)and a low noise mixer stage.

SENSITIVITY

The sensitivity of a radio receiver is proba-bly one of the most important aspects to itsdesign. You can specify sensitivity in termsof microvolts or dBm to achieve a given sig-nal-to-noise ratio in a given bandwidth.Therefore, a specification might read 0.5 µV

F FF

G

F

G G

F

G G G= +

−+

−+

−+1

2 1

1

3 1

1 2

4 1

1 2 3. . .

P KTBRN = 4

Planning a Receiver System 273

for 10 dB SNR in 2800 Hz bandwidth. Allgains (less losses) in the receiver contributeto the sensitivity figure, but the gain of theIF amplifier largely sets it.

SELECTIVITY

The selectivity of the receiver is set largely bythe IF amplifier filtering, but the filtering thatgoes up front is very important in reducingcertain spurious responses. After all, if nosignals are there to cause the spurious re-sponse, then they do not occur.

The IF filtering sets the bandwidth char-acteristics of the receiver, which in turn de-termines the selectivity characteristics. The IFfilter is single frequency, and as such, it canbe made with very steep side slopes and avery narrow bandwidth. Filters of 270 Hz arenot unheard of in shortwave receivers (forCW reception), but most are on the order of1.8, 2.8, 4, or 8 kHz. Multiple filters providemultiple bandwidths.

CALCULATING THE INTERCEPTPOINTS

Figure 18.3 shows the test setup for testingthe intermodulation distortion of the receiverunder test. The signal generators should beclean of spurious signals and well calibrated.The –3 dB hybrid couplers should producelow levels of IMD themselves, and theyshould be low-loss devices. It is importantthat the –3 dB hybrid coupler have equal lossto both input ports. The signal generators are

set to produce a comfortable level of output,say –10 or –20 dBm, a frequency 20 kHzapart. Shielding the signal generators, thecoupler, the attenuator, and all interconnec-tions is important to this test.

Perhaps the best procedure is to usethe minimum detectable signal to make thistest. The MDS is the signal level that pro-duces a +3 dB increase in the receiver audio output power level. Measure the re-ceiver’s noise output with no signal pre-sent, then adjust the attenuator to producethe 3 dB increase.

Once the level of the IMD product andthe levels of the signals are known, you cancalculate the intercept points as follows:

(18.3)

where

IPN is the nth-order intercept point;PA is the input power of one of the

inputs;PIMN

is the level of the IMD product sig-nal;

n is the order of the IMD product.

For example, to find the second-order in-tercept point (SOIP), the equation reduces to

(18.4)

And for the third-order intercept point (TOIP),

TOIP =−−

3

3 13

P PA IM

TOIP =

−−

2

2 12

P PA IM

IPnP P

nNA IMN=

−− 1


Fig. 18.3 Test setup for receivers.

SIGNALGENERATOR

NO. 1

SIGNALGENERATOR

NO. 2

3 dB HYBRIDCOUPLER

0–120 dBATTENUATOR(1 dB STEPS)

RECEIVERUNDER TEST

AUDIO POWERMETER

Several strategies can improve the electro-static interference and noise performance of aradio communications receiver. Some meth-ods are available for after-market use, whileothers are practical only when designing anew receiver.

One method for preventing the dynamiceffects of EMI is to reduce the signal level ap-plied to the input of the receiver at all fre-quencies. A front-end attenuator will helphere. Indeed, many modern receivers have aswitchable attenuator built into the front-endcircuitry. This attenuator will reduce the levelof all signals, backing the receiver circuitsaway from the compression point and, in theprocess, eliminating many of the dynamicproblems.

Another method is to prevent the of-fending signal or signals from ever reachingthe receiver’s front end. This goal can beachieved by using any of several bandpass fil-ters, high-pass filters, low-pass filters, or tun-able filters (as appropriate) ahead of thereceiver’s front end. These filters may not helpif intermodulation products fall close to thedesired frequency (e.g., the third-order differ-

ence products 2F1 − F2 and 2F2 − F1) but forother frequencies filters are quite useful.

When designing a receiver, some thingscan be done. If an RF amplifier is used be-tween the antenna and mixer, then use eitherfield effect transistors (MOSFETs are popular)or a relatively high-powered but low-noisebipolar (NPN or PNP) transistor intended forcable TV applications, such as the 2N5109 orMRF-586. Gain in the RF amplifier should bekept minimal (4–8 dB). It also is useful to design the RF amplifier with as high a DCpower supply potential as possible. That tac-tic allows a lot of “head room” as input sig-nals become larger. Slew rate symmetry is animportant characteristic in an RF amplifier.This means that the rise time and fall time ofthe amplifier are equal or nearly so.

You also might consider deleting the RFamplifier altogether. One receiver design phi-losophy holds that a high-dynamic rangemixer should be used with only input tuningor suboctave filters between the antennaconnector and the mixer’s RF input. It is pos-sible to achieve noise figures of 10 dB or sousing this approach, and that is sufficient for

Chapter 19

Improving Receiver Performancein a High EMI Environment

275

HF work (although it may be marginal forVHF and up). The tube-designed 1960s vin-tage Squires-Sanders SS-1 receiver used thisapproach and delivered superior perfor-mance. The front-end mixer valve was a 7360double-balanced mixer.

The mixer in a newly designed receiverproject should have a high dynamic rangeregardless of whether an RF amplifier isused. Popular with some designers is thedouble-balanced switching mixer. Althoughexamples can be fabricated from MOSFETtransistors and MOS digital switches, someintegrated circuit versions on the market areintended as MOSFET switching mixers. Pas-sive DBMs are available that operate up toRF levels of +20 dBm.

INTERMOD HILL: A TALE OF WOE

There is a hill not too far my home, whereradio receivers are sorely tested. I first ex-perienced this hill when a friend of mine,who lived only a block from the site, gothis ham radio operator’s license. This hill-top (bumptop is more like it) has a 2000 WAM BCB station, two 50 kW FM BCB sta-tions, and a microwave relay tower withscads of antennas on it. Not helping thingsmuch are the scores upon scores of two-way landmobile radio antennas. It seemsthat landmobile and cellular operators rentspace on radio station towers to gain theheight they need.

So what? Does that not just provide asignal-rich environment? No, the problem isthat a receiver can handle only a certainamount of radio frequency energy in its frontend. That area of the receiver is not very nar-rowband, so signals wander through thatotherwise might not make it. Indeed, even ifthe front end is tuned (usually with an “an-tenna tune” or “preselector” control), thebandpass is quite broad. Receivers with anuntuned front end may use a bandpass filter,and that is still a problem.

The offending signal need not be in thesame band as the signal being sought.

Indeed, the offending signal may be out ofband and not even heard on the receiver. If itgets to the RF amplifier or mixer, then it maytake up enough of the receiver’s dynamicrange to desensitize your receiver—making itless sensitive.

Another manifestation is that the of-fending signal may generate harmonics of itsown. Those harmonics are accepted as validsignals, so the receiver will tune them in as ifthey arrived on the antenna. The problem isthat the undesired signal drives the receiverfront end into a nonlinear operating region,and that has the effect of increasing har-monic distortion.

Still another problem caused by lettingstrong out-of-band signals reach the receiveris intermodulation products. When only twosignal frequencies are present and can drivethe front-end circuitry of the receiver intononlinearity, then a batch of new frequenciesis generated. After all, the front end containsa mixer, and if the signals hit the radio fre-quency amplifier first, they can make thatstage pretend it is a mixer. The frequenciesgenerated are defined by mF1 ± nF2, wherem and n are zero or integers (1, 2, 3, . . ., N).These frequencies are as follows:

FOR THIRD-ORDER PRODUCTS

F1 + 2F2F1 − 2F22F1 + F22F1 − F2

FOR FIFTH-ORDER PRODUCTS

2F1 + 3F22F1 − 3F23F1 + 2F23F1 − 2F2

and so forth.What does this mean? Suppose you live

near an AM broadcast band station that is op-erating on 1500 kHz. Suppose a local hamoperator starts broadcasting on a frequencyof 7200 kHz. These frequencies will providethird- and fifth-order intermodulation prod-


ucts of 4200, 9900, 10,200, 12,900, 15,900,18,600, 18,900, and 24,600 kHz.

Note where those intermodulationproducts fall? Right in the middle of some ofyour listening territory. And, when you con-sider that the ham operator can operate overa 300 kHz portion of the band, from 7000 to7300 kHz, a wide area 200 kHz below and100 kHz above those spot frequencies arevulnerable.

Keep in mind that AM and FM BCB sta-tions tend to have two attributes that helpthem mess up reception of other stations:(1) they are local, so their strength is veryhigh at the receiver’s location; (2) they arehigh powered (250 W–50 KW). As a result, ahuge signal from that BCB station is presentat the receiver. If the receiver antenna is onrented tower space (on a BCB tower), thensignal levels are huge. Add to the BCB sig-nals the dozens of microwave, two-way, re-peater, and cellular signals present, and youcan see how a simple equation like mF1 ±nF2 can explode into a huge number of sig-nals if IMD occurs. Is Intermod Hill rare?No, it is common for radio operators toseek height for antennas so they tend tocluster.

THE PROBLEM

What is the problem? Your receiver, no mat-ter what frequency it receives, is designedto accept only a certain maximum radio fre-quency energy in the front end. If more en-ergy is present, then one or more overloadconditions results. The overload could re-sult from a desired tuned station being toostrong. In other cases, simply too many sig-nals are within the passband for the re-ceiver’s front end to accommodate. In stillother cases, a strong out-of-band signal ispresent.

Figure 19.1 shows several conditionsyour receiver might have to survive. Figure19.1A is the ideal situation. Only one signalexists on the band and it is centered in thepassband of the receiver. This never hap-

pens, and the problem has existed sinceMarconi was hawking interest in his fledg-ling wireless company. Radio pioneersFessenden and Marconi interfered witheach other while reporting yacht races offLong Island around the turn of the cen-tury—not an auspicious beginning for mar-itime wireless. A more realistic situation isshown in Figure 19.1B. Here we see a largenumber of signals, both in- and out-of-bandsignals, both weaker and stronger than thedesired signal. In Figure 19.1C, an extremelystrong local station (e.g., AM BCB signal) ispresent but out of the receiver’s front-endpassband. The situation you probably faceis shown in Figure 19.1D: a large out-of-band AM BCB signal as well as the usualhuge number of other signals, both in andout of band.

Several problems result from the situa-tions in Figure 19.1, all of which are speciesof front-end overload-caused intermodula-tion or cross-modulation.

Blanketing

If you tune across the shortwave bands, es-pecially those below 10 or 12 MHz, andnote an AM BCB signal that seems like it ishundreds of kilohertz wide, then you arewitnessing blanketing. It drives the mixer orRF amplifier of the receiver into severe non-linearity, producing a huge number of spu-rious signals and apparently a very widebandwidth.

Desensitization

Your receiver can accommodate only a limitedamount of RF energy in the front-end circuits.This level is expressed in the dynamic rangespecification of the receiver and hinted by thethird-order intercept point and −1 dB com-pression point specifications. Figure 19.2Ashows what happens in desensitization situa-tions when a strong out-of-band signal is pre-sent. The strong out-of-band signal takes upso much of the dynamic range “head room”that only a small amount of capacity remains

Improving Receiver Performance in a High EMI Environment 277

for the desired signal, reducing the level ofthe desired signal. In some cases, the over-load is so severe that the desired signal be-comes inaudible. If you can filter out orotherwise attenuate the strong out-of-bandsignal (Figure 19.2B), then the head room isrestored and the receiver has plenty of capac-ity to accommodate both signals withoutcausing other problems.

Figure 19.3 shows two more situations.Figure 19.3A shows the response of the re-ceiver when output level is plotted as a func-tion of the input signal level. The idealsituation is shown by the dashed line fromthe (0, 0) intercept to an infinitely strong sig-nal. Real radio receivers depart from theideal and eventually saturate (the solid linebeyond the dot). The point denoted by the

dot on the solid line is the point at which theTOIP is figured. Important here is to considerwhat happens when signals are received thatare stronger than the input signal that pro-duces the flattening of the response.

Figure 19.3B shows the generation ofharmonics; that is, integer (1, 2, 3, . . .) multi-ples of the offending signal’s fundamentalfrequency. These harmonics may fall withinthe passband of a receiver and be seen asvalid signals even though they were gener-ated in the receiver itself.

The intermodulation problem is shownin Figure 19.3C. It occurs when two or moresignals are present at the same time. Strongintermodulation products are created whentwo of these signals heterodyne together.The heterodyne (mixing) action occurs be-


Fig. 19.1 Conditions faced by radio receivers: (A) ideal situation, one signal; (B) that faced by most radioreceivers; (C) strong local signal interference; (D) both types of interference combined.

A

B

C

D

cause the receiver’s front end is nonlinear atthis point. The frequencies produced by justtwo input frequencies (F1 and F2) are de-scribed by mF1 ± nF2, where m and n are in-tegers. As you can see, depending on howmany frequencies are present and howstrong they are, a huge number of spurious

signals can be generated by the receiver’sfront end.

What about IF selectivity? Suppose thereis an IF filter of 2.7–8 kHz (depending onmodel and mode). Why does the filter not re-ject the bad-guy signals? The problem is thatthe damage occurs in the front-end section of


STRONG OUT-OF-BAND SIGNAL

ACTUALSTRENGTH

APPARENTSTRENGTH

DESIREDSIGNAL

Fig. 19.2Desensitization:(A) before attenuation;(B) after attenuation.

STRONG OUT-OF-BAND SIGNAL

ACTUAL &APPARENTSTRENGTHTHE SAME

DESIREDSIGNAL

AFTERATTENUATION

A

B

the receiver, before the signals encounter theIF selectivity filters. The problem is caused byan overdriven RF amplifier, mixer, or both.The only way to deal with this problem is toreduce the level of the offending signal.

SOLUTIONS

The Attenuator Solution

Some modern receivers are equipped withone or more switchable attenuators in thefront end. Some receivers also include an RFgain control that operates in the same man-ner. Some receiver operators use external

switchable attenuators for the same purpose.The idea behind the attenuator is to reduceall the signals to the front end enough to dropthe overall energy in the circuit to below thelevel that can be accommodated without ei-ther overload or intermodulation at significantlevels. The attenuator reduces both desiredand undesired signals, but the perceived ratiois altered when the receiver front end is de-loaded to a point where desensitization oc-curs or intermods and harmonics pop up.

The Antenna Solution

The antenna that you select can make somedifference in EMI problems. Generally, a reso-


Fig. 19.3Output level as a function of the inputsignal level: (A) noise increases dra-matically above the 1 dB point orTOIP; (B) generation of harmonics;(C) intermodulation.

INPUT SIGNAL LEVEL

OU

TP

UT

SIG

NA

L LE

VE

L

A

B

C

nant antenna with its end nulls pointed towardthe offending station will provide better per-formance than an omnidirectional antenna.Also, vertical HF antennas are well known tobe more susceptible to EMI because they re-spond better to the ground wave electricalfield generated by the station.

The Filter Solution

One of the best EMI solutions is to filter outthe offending signals before they hit the re-ceiver front end in a manner that affects thedesired signals only minimally. This task isnot possible with the attenuator solution,which is an “equal opportunity” filter, be-cause it affects all signals equally. Figure19.4A shows what happens to a signal out-side the passband of a frequency selective fil-ter: It is severely attenuated. It does not dropto zero, but the reduction can be quite pro-found in some designs.

Signals within the receiver’s passbandare not unaffected by the filter, as shown inFigure 19.4B. The loss for in-band signals,however, is considerably less than for out-of-band signals. This loss, called insertionloss, usually is quite small (1 or 2 dB) com-

pared to the loss for out-of-band signals(lots of dB).

Several different types of filter are usedin reducing interference, depending on thecircumstances. A high-pass filter passes allsignals above a specified cutoff frequency(Fc). The low-pass filter passes all signals be-low the cutoff frequency. A bandpass filterpasses signals between the lower (FL) andupper cutoff frequencies (FH). A stopband fil-ter is just the opposite of a bandpass filter: Itstops signals on frequencies between FL andFH, while passing all others. A notch filter,also called a wave trap, will stop a particularfrequency (Fo) but not a wide band of fre-quencies like the stopband filter. In all cases,these filters stop the frequencies in the desig-nated band, while passing all others.

The positioning of the filter in the an-tenna system is shown in Figure 19.5. The ideallocation is as close as possible to the antennainput connector. The best practice, if space isavailable, is to use a double-male coaxial con-nector to connect filter output connector to theantenna input connector on the receiver. Ashort piece of coaxial cable can connect thetwo terminals if this approach is not suitable.Be sure to ground both the ground terminal on


BANDPASSFILTER

Fig. 19.4Bandpass filter operation: (A) sig-nals outside the band; (B) signalsinside the band.

BANDPASSFILTER

A

B

the receiver and the ground terminal of the fil-ter (if one is provided). Otherwise, depend onthe coaxial connectors’ outer shell making theground connection.

Wave Traps

A wave trap is a tuned circuit that causes aspecific frequency to be rejected. Two formsare used: series tuned (Figure 19.6A) and par-allel tuned (Figure 19.6B). The series-tunedversion is placed across the signal line (as inFigure 19.6A) and works because it produces avery low impedance at its resonant frequencyand a high impedance at frequencies removedfrom resonance. As a result, the interfering sig-nal will see a resonant series-tuned wave trapas a near short circuit, while other frequenciesdo not. The parallel-resonant form is placed inseries with the antenna line (as in Figure19.6B). Because it provides a high impedanceto its resonant frequency, it blocks the of-fending signal before it reaches the receiver.It provides a low impedance to frequenciesremoved from resonance.

Wave traps are useful when a single sta-tion is causing a problem and there is no de-sire to eliminate nearby stations; for example,if the receiver is close to a medium-wave AMBCB signal and the operator does not wantto interrupt reception of other AM BCB sig-nals. The values of components shown inFigures 19.6A and 19.6B are suitable for themedium-wave AM BCB but can be scaled tothe other bands if desired.

If two stations are causing significant in-terference, then two wave traps have to beprovided, separated by a short piece of coax-ial cable or contained within separate com-partments of the same shielded box. In thatcase, use a parallel-tuned wave trap for onefrequency and a series-tuned wave trap forthe other. Otherwise, interaction between thewave traps can cause problems.


Fig. 19.5 Location of filter for a receiver.

FILTERCOAX FROMANTENNA

RECEIVER

ANT IN

Fig. 19.6 AM BCB wave traps: (A) series tuned;(B) parallel tuned.

C1400 pF

L1220 µH

J1 J2

SHIELD

C1160 pF

L1700 µH

J1 J2

SHIELD

A

B

High-Pass Filters

One very significant solution to EMI from anAM BCB station is to use a high-pass filterwith a cutoff frequency between 1700 and3000 kHz. This filter passes the shortwave fre-quencies and severely attenuates AM BCBsignals, causing the desired improvement inperformance. Figure 19.7 shows a designused for many decades. It is built easily be-cause the capacitor values are 0.001 and0.002 µF (which some people make by paral-lel connecting two 0.001 µF capacitors). Bothinductors are µH, so it can be made withtoroid cores. If T-50-2 RED cores are used (AL = 49), then 26 turns of small-diameterenameled wire will suffice. Or, if T-50-15RED/WHITE cores are used (AL = 135), then

15 turns are needed. The circuit of Figure19.7 produces pretty decent results for littleeffort.

Absorptive Filters

The absorptive filter solves a problem withthe straight high-pass filter method and pro-duces generally better results at the cost ofmore complexity. This filter (Figure 19.8)consists of a high-pass filter (C4–C6/L4–L6)between the antenna input ( J1) and the re-ceiver output ( J2). It passes signals above 3 MHz and rejects those below that cutofffrequency. It also has a low-pass filter(C1–C3/L1–L3) that passes signals below 3MHz. What is notable about this filter, andthe source of its name, is that the low-passfilter is terminated in a 50 Ω dummy load.This arrangement works better than thestraight high-pass filter method because it ab-sorbs energy from the rejected band and re-duces (although does not eliminate) theeffects of improper filter termination.

Some of the capacitor values are notstandard but can be made using standarddisk ceramic or mica capacitors and the com-binations in Table 19.1. The other capacitorshave standard values.

The coils are a bit more difficult to ob-tain. Although it is possible to use slug-tuned


Fig. 19.7 Simple high-pass AM BCB filter.

J1 J2

SHIELD

C10.002 µF

C30.002 µF

C20.001 µF

L13.3 µH

L23.3 µH

Fig. 19.8Absorption AM BCB filter.

R150 Ω

C3270 pF

C21270 pF

C11820 pF

L14.1 µH

L24.1 µH

L32 µH

L41.5 µH

L52.2 µH

L610.2 uH

C61400 pF

C4680 pF

C5680 pF

J1ANT

J2RCVR

coils obtained from commercial sources(e.g., Toko) or homebrewed, this is not thepreferred practice. Adjusting this type of fil-ter without a sweep generator might provedaunting, due to interactions among thesections. A better approach is to use toroid-core homebrew inductors. The toroidalcores reduce interaction between the coil’smagnetic fields, so they simplify construc-tion. Possible alternatives are shown inTable 19.2. For all coils, use wire of a simi-lar UK SWG size to #24 to #30 AWG enamelinsulation.

The dummy load used at the output ofthe low-pass filter (R1 in Figure 19.8) can bemade using a 51 Ω carbon or metal film re-sistor or two 100 Ω resistors in parallel. In apinch, a 47 Ω resistor could be used but isnot preferred. In any event, use only nonin-ductive resistors, such as carbon compositionor metal film 1/4 –2 W resistors.

If you would like to experiment withabsorptive filters at cutoff frequencies other

than 3 MHz, then use the reactance values inTable 19.3 to calculate component values.

The exact component values can befound from variations on the standard induc-tive and capacitive reactance equations:

(19.1)

(19.2)

These component values are bound tobe nonstandard but can be made either usingcoil forms (for inductors) or series-parallelcombinations of standard-value capacitors.

C

F Xc cpF picofarads= 10

2

12

π

L

X

FL

cµ πH microhenrys= ×

2106


Table 19.1 Combinations for StandardizingCapacitors

Capacitor Value Directions

C1 1820 pF Use 1000 pF (0.001 µFor 1 nF) in parallel with820 pF

C2 1270 pF Use 1000 pF and 270 pFin parallel

C6 1400 pF Use 1000 pF, 180 pF,and 220 pF in parallel

Table 19.3 Reactance Values

Component X (XL or XC)

L1 28.8 ΩL2 78.4 ΩL3 38 ΩL4 28.8 ΩL5 42 ΩL6 193 ΩC1 28.8 ΩC2 42 ΩC3 193 ΩC4 78.4 ΩC5 78.4 ΩC6 38 Ω

Table 19.2 Alternative Cores

Coil Value Core AL Value Turns

L1 4.1 µH T-50-15 RED/WHITE 135 17

L2 4.1 µH T-50-15 RED/WHITE 135 17

L3 2 µH T-50-15 RED/WHITE 135 12

L4 1.5 µH T-50-2 RED 49 18

T-50-6 YEL 40 20

L5 2.2 µH T-50-2 RED 49 21

T-50-6 YEL 40 24

L6 10.2 µH T-50-2 RED 49 46

T-50-6 YEL 40 51

Transmission Line Stubs

At higher frequencies, where wavelengthsare short, a special form of trap can be usedto eliminate interfering signals. The half-wavelength shorted stub is shown in Figure19.9. It acts like a series-resonant wave trapand is installed in a similar manner, right atthe receiver. The stub is shorted at the freeend, so it presents a low impedance at itsresonant frequency and a high impedance atall other frequencies. The length of the stubis given by

(19.3)

where

Lcm is the length in centimeters (cm);FMHz is the frequency of the interfering

signal in megahertz (MHz);

VF is the velocity factor of the coaxialcable (typically, 0.66 for polyeth-ylene cable and 0.80 for poly-foam; for other types see themanufacturer’s data).

Shielding

Shielding is a nonnegotiable requirement offilters used for EMI reduction. Otherwise, thesignal enters the filter at its output and is notattenuated. Use an aluminum shield box ofthe sort that has at least 5–6 mm of overlapof tight-fitting flange between the upper andlower portions.

Expected Results

If the correct components are selected and a good layout practice is followed (whichmeans separating input and output ends, as

L

FcmMHz

VF=

1250


Fig. 19.9Half-wavelength shorted stub.

COAX FROMANTENNA

RECEIVER

ANT IN

well as shielding the low-pass and high-passsections separately), then the absorptive filtercan offer stopband attenuation of −20 dB atone octave above Fc, −40 dB or more at twooctaves, and −60 dB at three octaves. For a 3MHz signal, one octave is 6 MHz, two octavesare 12 MHz, and three octaves are 24 MHz.

Difficult Cases

The UHF and above bands are alive with sig-nals, and that makes them prime sources forEMI. Where the UHF or microwave receiver isexperiencing really serious amounts of EMI,the scheme of Figure 19.10 can be used. Inthis scheme, a circulator is used to split thesignal into two paths. The first path, consistingof a stub and a high-Q resonant cavity, istuned to the interfering signal. The third porton the circulator is connected to the receiver.

EMI TO TELEVISION AND CABLE TV EQUIPMENT

To many people, EMI to video and televisionequipment is all there is to the subject. Video

and TVI easily is seen or heard and terriblyirritating to a large number of people. Wenow look at video/TV interference and whatcan be done about it. Considered will be an-tenna-connected television receivers, cabletelevision receivers, and VCR equipment.

The Basic Television Receiver

Before we can deal with television interfer-ence we must examine the standard televi-sion system. The United States and Canadause NTSC television, while foreign countriesuse SECAM or PAL systems. Some wags tellus the USA/Canada standard means “nevertwice same color,” but in reality it means“National Television Steering Committee.”Figure 19.11 shows the basic NTSC color-TVsignal. The FCC has allocated space based onthis model for transmission of TV.

The NTSC color-TV standard requires avideo carrier 1.25 MHz from the lower end ofthe channel. Spaced 3.579 MHz from thevideo carrier is a color burst carrier, which isnot transmitted (it is suppressed in the trans-mitter). Spaced 4.5 MHz from the video car-rier is an audio carrier, which is frequency


Fig. 19.10Adjustable stub and a high-Q res-onant cavity.

1 2

3

RECEIVERANT

ADJUSTABLE STUB

CIRCULATORTEE

HI-QRESONANT

CAVITY

KEEP AS SHORT ASPOSSIBLE

ANTENNA

modulated (FM) with a 25-kHz deviation.The video carrier is vestigial sideband (aspecies of AM in which one sideband is pro-duced in full and one sideband is producedas a vestigial sideband). The chroma infor-mation (the color signal) is phase modulatedonto the signal.

A typical television receiver is shownin block diagram form (Figure 19.12). Thisreceiver is a superheterodyne, so it has afront end inside the tuner that is down con-verted to an IF frequency. An AM envelopedetector circuit, called a video detector, isused to demodulate the IF signal. The videodetector separates the video, color, andsound information.

The video amplifier section takes thecomposite black and white video and ampli-fies it to a point where it can drive the cath-ode ray tube (CRT). Part of the videoamplifier or detector (shown here as part ofthe amplifier) is a circuit called a sync sepa-rator. This circuit separates out the horizon-tal and vertical deflection pulses.

The deflection amplifiers control thevertical and horizontal aspects of the CRT.The vertical deflection is at 59.94 Hz, and the

horizontal deflection is 15,734 Hz (thesewere changed from 60 and 15,750 Hz to ac-commodate color). The high-voltage DC forthe second anode of the CRT is derived fromthe horizontal amplifier circuit.

The color information is passedthrough a 3.579 MHz IF amplifier from thevideo detector. From there, it is processedand displayed on the CRT in the form of(usually cathode-driven) red, blue, andgreen signals.

The sound information is passedthrough a 4.5 MHz IF amplifier and a 25 kHzFM detector to form the sound signal. Fromthere, the audio signal is built up in a chainof amplifiers and output through a loud-speaker.

The front end of the television set is thetuner. It consists of a mixer circuit, a local os-cillator, and (usually) an RF amplifier circuit.The front end is fed with the RF transmissionline from the antenna. Either 300 Ω twin-leador 75 Ω coaxial cable may be used. The 75 Ωcoaxial cable predominates today but youstill find older television receivers and low-cost modern television receivers that usetwin-lead cable.


Fig. 19.11 Basic American NTSC color-TV signal.

BLACK & WHITEVIDEO

COLORBURST

CHROMAINFORMATION SOUND

CARRIER

VIDEOCARRIER

1.25MHz

3.579MHz

4.5MHz

6MHz

250KHz

Fundamental Overload

Fundamental overload of a television re-ceiver occurs when a strong signal is presentat the input of the receiver. The television re-ceiver responds to the fundamental of the of-fending transmitter. A decent TV pictureneeds 25–30 dB of signal-to-noise ratio(SNR), and less than that magic number canresult in interference. The symptoms of fun-damental overload include a herringbonepattern or even a blanked out picture.

Harmonic Overload

While fundamental overload is the mostcommon form of EMI to TV receivers har-monics from the transmitter may be second(especially with HF or low-band VHF trans-mitters). Harmonics, integer multiples of thefundamental frequency of the transmitter,can reach considerable heights in transmit-ters that are maladjusted. The FCC requirestransmitter owners to keep the harmoniccontent of their emissions down –40 dBc(decibels below carrier) to –60 dBc depend-ing upon the service and frequency.

Harmonic overload may be caused bymaladjustment of the transmitter or gener-ated in the television receiver itself, if a PNjunction in the circuit can be biased into non-linearity by the transmitter signal. It may alsobe generated by dirty connections, corrosion,or even rusty bolts.

Audio Rectification

Audio rectification is heard rather than seen. Itis audio from the offending transmitter’s mod-ulation being picked up, envelope detectedand amplified by the TV’s audio amplifierchain. Interestingly enough, audio rectificationoccurs more often in cases where the antennaof the offending transmitter is vertically polar-ized. A good indication that interference is au-dio rectification is that it is audible regardlessof the setting of the volume control.

IMD Interference

A television receiver is an amplifier and de-modulator of a radio signal, so it is suscepti-ble to intermodulation distortion (IMD)interference. IMD interference occurs when


Fig. 19.12 Typical television receiver block diagram.

VIDEOAMPLIFIER

SYNCSEPARATOR

VERTICALDEFLECTION

HORIZONTALDEFLECTION

HIGHVOLTAGE

VIDEODETECTOR

IFAMPLIFIER

TUNERRFINPUT

COLORCIRCUITRY

3.579 MHzCOLOR IF

AUDIOSTAGES

SOUNDDETECTOR(25 KHz FM)

4.5 MHzSOUND IF

two or more signals are present in the inputof the receiver. They produce additional fre-quencies according to mF1 ± nF2, where F1and F2 are the frequencies and m and n areeither integers or zero. That means a lot offrequencies! If two or more frequencies hap-pen to follow this rule and produce a thirdfrequency that is on a television frequency,then IMD interference will result.

IF Interference

IF interference exists because of the IF am-plifier chain in the television set. It is identi-fied easily because it affects all the TVchannels. Furthermore, it produces a fineherringbone pattern on the screen that ro-tates with the setting of the fine-tuning con-trol. No other form of interference does thistrick (although, with other forms of interfer-ence, the intensity of the interference doeschange with the fine tuning).

Direct Pick-Up

This form of interference, which requirestransmitters of several kilowatts to accom-plish, is picked up directly by the televisionreceiver. It generally is due to internal wiringor printed circuit boards. To test for this typeof interference, terminal the antenna in itscharacteristic impedance and note whetherthe signal level decreases. If it does, then the

signal is arriving via the antenna terminals,and it is not a case of direct pick-up.

Common Mode vs. Differential Mode Signals

Very little has been written about the differ-ence between common mode and differentialmode interference. The difference is simple:Differential mode interference arrives entirelyinside the feedline, while common mode inter-ference does not. The common mode signalmay arrive on the transmission line or on theAC line cord. The treatment of each form of in-terference depends on whether it is commonor differential mode. Although both forms ofrepair may be required in a given case.

Common mode filters come in two vari-eties: transmission line and AC line cord. Bothare used to suppress EMI in TV systems. Thetransmission line variety is shown in Figure19.13. It consists of a toroid core wrappedwith the television antenna transmission line.This common mode choke should be in-stalled as close to the TV set as possible. Tomake an AC line cord type of common modechoke wrap the toroid with the AC line cordas close as possible to the TV set.

Filtering

The television can be filtered at the antennaterminals or the AC line cord. In this chapter,


Fig. 19.13Common mode transmis-sion line choke.

TO TVSET

FROMTV

ANTENNA

TV COAXIAL CABLEWRAPPED AROUNDTOROID CORE

we discuss the type of filtering that you placeon the transmission line.

Figure 19.14 shows the proper installationof the filter at the TV receiver. Figure 19.4Ashows a 300 Ω system, while Figure 19.4Bshows a 75 Ω system. The key in both casesis to make the transmission line between

the television and filter as short as possible.In fact, it would be preferable to install thefilter right at the antenna terminals of thetuner inside the TV set, but that usually isnot possible unless you are a television re-pair person.

The type of filter to install depends onthe nature of the transmitter causing interfer-ence to that TV set. Where the interference iscaused by an HF band transmitter (amateur,CB, or commercial) use a high-pass filter atthe television set, with a cutoff frequency below TV channel 2. For VHF or UHF inter-ference, use either a high-pass, low-pass,bandpass, or bandstop (i.e., notch) filter, de-pending on the situation. If the offendingtransmitter affects more than one channel, itmay be necessary to use a bandstop filtercentered on the transmitter’s frequency at theTV receiver.

Figure 19.15 shows a type of filter thatworks well for both common mode and differ-ential mode signals in coaxial cable type sys-tems. It consists of a 300 Ω twin-lead type ofhigh-pass filter sandwiched between two 4:1BALUN transformers. The 300 Ω twin-lead be-tween the transformers and the filter should beas short as possible, even of zero length.

Stubs for EMI Elimination

Stubs can be used for EMI suppression onTV receivers. The usual quarter- and half-wavelength stubs are used, but Figure 19.16shows an eighth-wavelength (λ/8) stub. Itis used to eliminate one particular fre-quency, so it functions as a notch filter.The frequency of the notch filter should bethe frequency of the offending transmitter.


Fig. 19.14 Position of filters on a TV set: (A) 300Ω; (B) 75 Ω.

FILTER

ANTENNA

ANTENNA

A

B

Fig. 19.15Use of a 300 Ω filter on a75 Ω system.

300 OHMHIGH-PASS TVI

FILTER

4:1BALUN

4:1BALUN

SHORT PIECE OF300 OHM TWIN-LEAD

(IF NECESSARY)

The stub has a capacitor (C1) at the end,and this capacitor is used to tune the stubto the correct frequency.

Faraday Shielded Coaxial Cable

Figure 19.17 shows a Faraday shielded coax-ial cable used to suppress EMI. This forms aspecies of common mode choke. To makethe choke, take a short length of coaxial ca-ble with a connector on one end. Wrap theend into a loop of less than 6 in. in diameter.Scrap away a contact point on the outer insu-lator, and connect the inner connector to it.Next, do the same thing to the cable from the

antenna or cable TV customer drop. Lay thetwo cables over each other in the mannershown on the inset in Figure 19.17 and se-cure with tape or cable ties.

Cable Television Systems

The cable television system in the UnitedStates is a closed system used to carry mul-tiple channels to homes, businesses, andother subscribers. It should be very clean,but often is not. Sharing certain channelswith other services (amateurs, navigation,and communications) makes the systemripe for interference.


Fig. 19.16Eighth-wavelength stub.

RFOUT

C1

L( l /8 )

RFIN

Fig. 19.17 Faraday shielded coaxial cable choke.

TV TUNER

TO TV ANTENNAOR CABLE DROP

A basic cable system is shown in Figure19.18. It can be coaxial cable or fiber optic innature, of which the fiber optic is by far mostfree of EMI problems. Although in some ar-eas, fiber has replaced coaxial cable, mostcable systems are made of coaxial cable.

The head end of the cable system takesprogramming from various sources (off-the-airchannels, satellite, local origin) and passesthem to various trunk lines. Only one trunk isshown for clarity. The signal on the trunk is ofrelatively low level but is boosted by trunk am-plifiers before being distributed further. Eachtrunk amplifier leads to a bridger network thatdistributes signal to the various subscribers.The subscriber tap into the system is called adrop. A line extender amplifier is used to boostthe signal lost in the coaxial cable and variousdrops to a level useful in the system.

The quality of the signal is determinedby two factors: noise and distortion. Thenoise mostly is from the amplifiers and de-grades picture quality. Distortion can beharmonic or IMD, but is mostly IMD. Thelength of each distribution leg is limited,principally by the distortion characteristicsof the amplifiers.

TWO-WAY CATV It is possible for the cable TV provider tocommunicate two ways on its systems. The5–40 MHz region is used to provide up-stream communication between the sub-scriber and the cable TV provider, and theinstitutional net (I-net) in the subsplit sys-tem (in the mid-split system 5–112 MHz isused, and in the high split 5–174 MHz is used).


Fig. 19.18 Cable TV distribution system.

HEAD ENDLOCALORIGIN

TELEVISION SET

TELEVISION SET

TELEVISION SET

TELEVISION SET

TELEVISION SET

TELEVISION SET

TELEVISION SET

TELEVISION SET

OTHERTRUNK

TRUNKAMPLIFIERS

TRUNKAMPLIFIER

BRIDGER BRIDGER

LINEEXTENDERAMPLIFIER

LINEEXTENDERAMPLIFIER

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

CUSTOMERDROP

Channelization

Three generally accepted channelizationschemes are used in cable TV systems: stan-dard, harmonically related carrier (HRC), andincrementally related carriers (IRC). Theseare defined in standard EIA-542. Of these,the standard scheme is by far the most com-mon. It uses TV channels assigned from 54MHz in 6n + 1.25 MHz schemes, up to 1002MHz. The HRC method uses TV channelslocked to a 6 MHz comb. The IRC uses thebasic 6n + 1.25 MHz carriers locked to an in-ternal comb.

Leakage

Cable TV is supposed to be a closed system.Cable TV leakage occurs, however. It iscalled ingress when an outside source inter-feres with the cable TV and egress when theRF energy gets out. Frequencies from 5 to750 MHz are used, including some assignedto other services. Leakage limits are relatedto distance as in Table 19.4.

A field strength of 15 mV/m at a dis-tance of 100 feet from the cable TV is a strongsignal. But the regulations also maintain thatthe cable TV system not cause harmful inter-ference to other services. The definition of“harmful” is not given, however.

Responsibility

The responsibility for EMI to and from cableTV systems is split among the cable TV sys-tem, the transmitter owner, and the owner ofthe afflicted television receiver. The transmit-ter owner’s responsibility is limited by theterms of his or her license. If the transmitteris operated according to the terms of that li-cense, including the emission of spurious sig-

nals, then you can do little to the transmitterowner. That leaves the cable TV system andthe subscriber. However, cooperation of theoffending transmitter owner is needed totroubleshoot the interference problem.

FINDING LEAKS

The problems with cable TV ingress andegress start with the effects of weathering onthe system as well as intentional acts by thecustomer. To protect against the former, theFCC requires cable TV operators to continu-ously patrol the system to find leaks. Findingthe leaks is an exercise in radio directionfinding. The technician drives along the linewhile monitoring the leakage levels on a sen-sitive receiver. The mobile unit will locate theleak within about 80–100 ft, after whichhandheld receivers of lower sensitivity arerequired.

WHAT TO DO WHEN THE INTERFERENCE IS AT THE

SUBSCRIBER END

The problem of what do you do when theproblem is at the subscriber end is difficult.Some cable TV companies are willing to trou-bleshoot the situation to keep the subscriberhappy, while others are not. Modifications toa television set should be done by competentpersonnel, but beyond that it is possible totroubleshoot the system.

First, diagnose the problem. Is the setor the converter really involved? The inter-ference may be experienced by only onesubscriber. In that case, assume that the sub-scriber’s system is at fault (multiple faultsalong a line do not necessarily mean that thesystem is at fault but do point to a systemproblem).

The basic solution for most problems isa common mode choke. The common modechoke can be made using the cable and atoroidal core per instructions given in Figure19.13. It consists of a toroid core wrappedwith coaxial cable. Figure 19.19 shows theplacement of the common mode choke inthe system. This does not preclude placing acommon mode choke and a differentialmode high-pass filter at the television re-ceiver’s antenna input terminal.


Table 19.4 Leakage Limits

Frequency (MHz) Distance (ft) Field Strength

5–54 MHz 100 15 mV/m

54–216 MHz 10 20 mV/m

216–750 MHz 100 15 mV/m

Interference can occur through twopaths: power line conduction and direct radi-ation. The power line conduction schemecan be eliminated at the transmitter by plac-ing either a common mode choke or a differ-ential filter in the power line. In other cases,however, the common mode choke shouldbe placed in the AC line cord of the con-verter or susceptible TV receiver. The com-mon mode choke is made by winding the ACline cord around the toroid core.

WHAT TO DO WHEN THE CUSTOMER IS AT FAULT

The customer might be at fault in two cases:(1) the existence of a susceptible TV and (2)the use of multiple sources. The susceptibleTV usually is found by disconnecting theconverter from the antenna terminals and re-placing it with a carbon resistor of either 300Ω or 75 Ω, depending on the impedance ofthe set. If the interference persists, then theset is responsible. In that case, commonmode chokes on the antenna terminal andAC line cord and differential mode filtering atthe antenna terminals may help.

The use of multiple sources for their TVsignals may be the problem. Look for con-nections that should not be present. If theconnections are made through an A/Bswitch, then disconnect the A/B switch andconnect the converter directly. If the interfer-ence disappears, then you have solved theproblem. If not, keep trying.

THE BEST SOLUTION

What is the best solution. In all the casesmentioned, one solution works best of all:Prevent the offending signal from enteringthe receiver. Whether the problem is desen-sitization from front-end overload, harmonicgeneration, or intermodulation problems,the solution is to get rid of the bad signal. Inthe case of intermodulation products, get-ting rid of just one of the two signals is allthat is needed. A filter in the antenna lineahead of the receiver will work nicely forthis purpose.


Fig. 19.19Common mode choke from cable TV system.

34

CABLE TVCONVERTER

TELEVISIONSET

SHORTCABLETO SET

FROMCABLE TVSYSTEM

COMMONMODE CHOKE

Ionospheric fading affects shortwave propa-gation more than the other bands. Becauseof the vagaries of ionospheric propagation,high-frequency shortwave signals cannot bedepended on for highly critical applications.But, in amateur radio, international broad-casting, and a host of other services, thehigh-frequency shortwave bands are as stillpopular as ever.

Perhaps the main mechanism forionospheric distortion of the signal onshortwave radio is fading. Unfortunately, onamplitude modulation (AM) stations, thetwo sidebands fade out of phase with eachother and also with the RF carrier, produc-ing a hollow, rolling fade. Added to ordi-nary amplitude fading, this makes desiredsignals difficult to receive.

How do the big international broadcast-ers and other users of spectrum space dealwith fading? Well, truthfully, most of them to-day use satellite reception. It is much morereliable. But in their heyday, shortwavebroadcasters and local relaying signals from ashortwave source used diversity receptiontechniques. Even today, commercial users ofthe shortwave spectrum use diversity recep-tion as a matter of course.

DIVERSITY RECEPTION

The best method for ridding ourselves of fad-ing in the shortwave is diversity reception.There are three versions of diversity recep-tion: frequency diversity reception, spatial di-versity reception, and polarization diversityreception. We look at all three methods inthis chapter.

FREQUENCY DIVERSITY

Frequency diversity reception is based onthe differential fading of different shortwavefrequencies. This is seen when the side-bands and carrier fade out of phase witheach other. By its nature, AM separates thelower sideband, carrier, and upper sidebandin frequency equal to the audio spread ofthe input signal to the transmitter’s modula-tor. The signals do not fade the sameamount and at the same time. By using dif-ferent frequencies, and then voting on theoutput received, the desired reception canoften be obtained.

Figure 20.1 shows a frequency diversityscheme. Three receivers are used in this

Chapter 20

Diversity Reception Techniques

295

scheme, although two, four, or five may beused in actual practice. Each receiver istuned to a different frequency or, perhaps, adifferent band. The antenna inputs of the re-ceivers are tied together in a single multifre-quency antenna that covers all the bands.

Note the outputs of the receivers. Theygo to some sort of combiner that votes onwhich has the best signal. This circuit may beat the IF or audio. In IF-based systems, thecombiner also includes the demodulator cir-cuitry, so that audio signals come out of thecombiner.

SPATIAL DIVERSITY

Spatial diversity reception depends on themovement of the wave from place to place asit fades. This is due largely to the ionospherebeing unstable heightwise, and therefore, thesignal walks about a bit. The spatial diversityreception system is shown in Figure 20.2.

The key to spatial diversity reception isthe antenna field. Although three antennas

and receivers are shown here, real systemsmay have two to five antenna/receiver com-binations. The key to the antenna field’s per-formance is that they are spaced nλ/2 apart,where n is an integer (including 1). Thisspacing is dictated by the physics of the situ-ation. Any closer spacing would nullify theoperation considerably.

Three receivers are shown in the spatialdiversity reception scheme of Figure 20.2.Note that the same audio or IF combiner cir-cuitry is used as in the frequency diversity reception method (why mess up a goodthing?). The IF/audio combiner outputs thehighest signal automatically.

Note that the variable frequency oscil-lators of the three receivers are linked to-gether. More correctly, a designated “master”receiver drives a VFO input on the othertwo receivers. This configuration permitsthe user to adjust just one receiver, whilecontrolling all three. One sure sign that a re-ceiver is designed for the diversity receptionis the existence of VFO in/out connectorson the rear panel.


Fig. 20.1 Frequency diversity system.

RECEIVER NO. 1

RECEIVER NO. 2

RECEIVER NO. 3

COMBINERAUDIO

OUTPUT

MULTI-FREQUENCY ANTENNA

POLARIZATION DIVERSITYRECEPTION

The polarity of the transcendental electromag-netic wave that forms the shortwave signal ismessed up, to say the least. Despite as muchas 30 dB difference between the vertical andhorizontal polarization (i.e., if you cross-polar-ize your receiver antenna you will suffer up toa 30 dB loss), it rarely matters on shortwavewhether or not the polarity of the receiver an-tenna matches that of the transmitter antenna(normally good engineering practice). Thereason is that the polarity of the incoming sig-nal keeps shifting and rotating.

The solution to the problem is shown atFigure 20.3. Polarization diversity receptionuses two or more receivers tuned to the samefrequency, but fed with colocated verticallyand horizontally polarized antennas. The an-tennas are located at the same site but are of

opposite polarization. That way, when thepolarity shifts from more vertical to morehorizontal, the proper receiver takes over.

The same IF/audio combiner present inthe previous two methods is used again in polarization diversity reception. As in Figure20.2, the VFO of the master receiver is drivingthe VFO in terminal on the slave receiver.Again, this allows a cochannel receiver to beoperated by the master receiver.

IF/AUDIO COMBINER

The IF/audio combiner might be a simplevoting logic signal selector on the audio sig-nal. It will select whichever of the two tofive receivers is putting out the strongestsignal, or if two or more are putting outequally strong signals, it will select accord-ing to a protocol.

Diversity Reception Techniques 297

Fig. 20.2 Spatial diversity system.

nl / 2

nl / 2

VFOMASTER RECEIVER

SLAVE RECEIVER NO. 1


COMBINERAUDIO

OUTPUT

ANTENNA NO. 1

ANTENNA NO. 2

ANTENNA NO. 3

Another type of combiner operates atthe IF frequency of the receivers. This com-biner takes the signal and suppresses the car-rier and one sideband, then reinserts a stronglocal carrier from an oscillator circuit. The re-created single-sideband signal is more free offading than any of the input signals, so it isused to create the audio output on top of theadvantages provided by diversity reception.

This, at least, eliminates the problem ofthe sidebands fading out of phase with each

other and the carrier. The method is calledthe Farnsworth method by some authoritiesand the Crosley method by others.

Still another method for the combineris synchronous AM reception. This is an up-dated version of the Farnsworth or Crosleymethod as nearly as I can tell, because it re-quires the carrier to be nulled out throughphasing, then uses an oscillator to reconsti-tute the carrier.


Fig. 20.3 Polarity diversity system.

VFOMASTER RECEIVER


COMBINERAUDIO

OUTPUT

ANTENNA NO. 2(HORIZONTAL)

ANTENNA NO. 1(VERTICAL)

AAbsorptive filters, 283–284Acoustic wave filters, surface, 66ADC. See Analog-to-digital converterAFC (automatic frequency control), 198–199, 259AGC. See Automatic gain controlALC (automatic level control modulator), 148, 248AM. See Amplitude modulationAmplifiers. See also Preamplifiers; Postamplifiersaudio, 11blood pressure, 100broadband RF, 82, 86–89, 213–214buffer, 73, 138, 139, 201cable television and, 292cascode, 18–19, 165, 166, 209circuits, 163, 164–165classifying by common element, 72–73compression, 173configurations, 69Darlington, 203DC, 143deflection, 287differential, 194–195gain controlled, 194–197, 220intermediate frequency (IF), 11, 163, 165,

166–168, 169, 173, 220, 223, 227limiter, 188linear, 172, 181logarithmic, 172–173, 220, 221, 235, 236low noise (LNA), 6, 19, 68, 69MC 1350P IF/RF, 170output, 245phase-locked loop (PLL) circuit and, 189, 190push-pull, 82–86, 210

radio frequency, 38, 43, 57–59, 67–89, 213,235, 272

receiver architecture and, 231signal generator and, 255, 256, 257superheterodyne receiver and, 273transfer function, 29tuned, 83wideband, 83, 148, 189, 201, 203, 204, 205, 207,

208, 247Amplitude modulation, 9, 23, 32, 33, 57, 148,

175, 176–179, 180, 188, 248, 261, 266, 295Amplitude versus frequency, 215Amplitude versus time plots of signal and noise,

3–4Analog-to-digital converter, 227, 228IFM receiver and, 235noise, 228receiver architecture and, 229, 231

ANL (automatic noise limiter), 192Antenna, 24coils, 57directional, 6, 31electromagnetic inference problems and,

280–281height and, 277mounting, 69omnidirectional, 6polarization diversity reception and, 297spatial diversity reception and, 296sweep generators and, 251tuning problem, 57–59whip, 225

Architectures of radio receiver. See Receiverarchitecture

299

Index

Armstrong oscillator, 110, 111Attenuator(s), 196, 206, 212, 213, 28020 dB, 271calibrated precision, 148coarse, 245, 246, 248continuous variable calibrated type, 256, 258dynamic problems and, 275dynamic range and, 264filter position in, 281, 282input, 31, 220, 247, 248intermediate frequency (IF), 220intermodulation distortion (IMD) and, 266, 268mismatch loss and, 254output, 245, 246, 248, 256pads, 205, 206precision, 247, 258radio frequency (RF), 224receiver architecture and, 229signal generator and, 255, 256, 259TOIMDDR and, 265

Audioamplifiers, 11rectification, 288

Automatic frequency control, 198–199, 259Automatic gain control, 9, 165, 188, 193–194, 227components of, 9dynamic range blocking and, 264, 265intermodulation distortion (IMD) and 266,

267, 268radio frequency amplifier, 229radio frequency/intermediate frequency

transformers, 50, 52–53, 54range and threshold, 27receiver architecture and, 229, 231scanning superheterodyne receiver and, 234sensitivity measurements and, 259–260S-meter circuit and, 196–197speed, 27squelch circuits and, 197–198

Automatic level control modulator, 148, 248Automatic noise limiter, 192Automatic volume control, 193

BBandspreading, 60–61Bandwidthfilters with narrow and wider, 23front-end, 23, 24instantaneous frequency measurement (IFM)

receivers and, 235

intermediate frequency (IF), 25, 223noise measurement and, 263noise voltage versus, 16phase-locked loop (PLL), 148resolution (RBW), 220, 223sensitivity and, 20, 258

Barkhausen’s criteria, 110, 125, 126, 131Beat frequency oscillator, 11, 181Bessell functions, 249Bifilar winding, 83, 85, 87, 99, 107, 155, 211Bit error rate, 5“Black box,” 266Blanketing, 277Blocking, 32, 33dynamic range (BDR), 264–265

Bragg cell receivers, 236, 237Branly coherer, 6Burst modulator. See Amplitude modulationButler oscillators, 128–131, 134

CCable television, 286, 291, 292–294Capacitors, 165, 182air variable, 109bypass, 125, 167ceramic, 203, 283chip, 203, 208crystal phasing, 155DC blocking, 203, 213decoupling, 167, 205, 208dielectric, 78, 141, 167feedthrough, 64fixed, 113, 115, 141mica, 55, 109, 115, 141, 283MOSFET transistor and, 183neutralization, 78reactance and, 167selection, 141small-value, 138standardizing, 283stray, 57, 58, 113, 205trimmer, 57, 59–60, 113, 141variable, 113, 115

Cathode ray tube, 221, 224, 235, 287Channelization, 293Chebyshev response, 62, 152, 158Circuitsamplitude limiter, 235aperiodic oscillator, 129base, common, 73


bias, 74–75broadbands, 83, 84classification by common elements, 72–73clipper, 191collector, common, 73configuration, basic, 110, 111–119demodulator, 179, 180, 185–186diplexer, 91, 92–93, 219discriminator, 186–187double-balanced diode mixer, 101–103emitter, common, 72–73Foster-Seeley discriminator, 186, 187, 188frequency converters, 91–108front-end, 11inductor-capacitor resonant tank, 45–48integrated (IC), 169, 170, 171–172, 194inverter, 72linear mixer, 185multiple device, 208–211multivibrator, 189mute, 197–198parallel tuned, 113radio detector, 187, 188resonant, 45–48Schmidt trigger, 197, 198series tuned, 113SL560C, 170, 171S-meter, 196–198squelch, 197–198tank, 57, 58, 78, 93tempco, 141–142VHF/UHF, 22, 63, 64voltage-controlled oscillator (VCO), 143

Coaxial cable, 281, 282, 287, 290, 291, 292, 293

Coil core selection, 140Collector-to-base bias network, 75–76Colpitts and Clapp oscillators, 58, 110, 111, 116,

117, 124–125, 128, 131–135, 141Common mode choke, 289, 291, 293, 294Common mode signal, 289Compression point, –1 dB, 29Continuous wave signals, 9, 11, 20, 106, 195, 235,

239, 243, 248, 263, 266–268, 273Crosley method, 298Cross-modulation, 32, 33CRT (cathode ray tube), 221, 224, 235, 287Crystal filters, 153–159, 163, 168, 169Crystal oscillators, 105, 109, 143, 146, 246, 247Butler oscillators, 128–131, 134

Colpitts crystal oscillator circuit, 110, 111, 116,117, 124–125, 128, 131–135, 141

Colpitts oscillators, 131–133long-term stability, 124Miller oscillators, 125–127ovens, 137overtone oscillators, 134–135Pierce oscillators, 127–128piezoelectric crystals, 119, 120, 153piezoelectricity, 119, 120–122, 123short-term stability, 124sideband determination and, 181tempco, 26, 141–142temperature performance, 122, 123–124

Crystal video receivers, 6–7, 23Cutoff frequency, 290Cut-plate capacitor method, 60, 105CW. See Continuous wave signals

DDAC. See Digital-to-analog converterdBc (decibels below the carrier), 146, 245dBm, 13, 14, 240, 256, 259DBM. See Double-balanced mixersdBmV, 14DC load lines, 73–74DC power supply, 138, 139, 203, 204, 205,

213, 275DCR. See Direct-conversion receiversDeForest, Lee, 6De-Qing resistor, 53Desensitization, 68, 277, 278–280, 294Detector, 11amplitude, 235balanced product, 182envelope, 11, 21, 175, 176–179, 180, 181, 221,

231, 287noise, 193phase, 247phase-locked loop (PLL) FM/PM, 189, 190product, 11, 183, 184pulse counting, 188–189purpose of, 175quadrature, 170, 189, 190square law, 176video, 221, 236, 287zero-crossing, 188, 189

Detector and demodulator circuits, 175–190AM envelope detectors, 175, 176–179balanced demodulators, 179–180

Index 301

Detector and demodulator circuits (continued)definition, 175discriminator circuits, 186–187double-sideband and single-sideband suppressed

carrier demodulators, 180–184, 185FM and PM demodulator circuits, 185–186noise, 179phased-locked loop FM/PM detectors,

189, 190phasing method, 184, 185pulse counting detectors, 188–189quadrature detector, 189, 190radio detector circuits, 187, 188synchronous AM demodulation, 180

Device under test, 225, 254, 255, 256, 257Differential mode signal, 289Digital signal processing, 227–231ADC noise, 228receiver architectures, 228–231schematic, 227

Digital-to-analog converter, 148, 149, 227, 228,231, 250, 251, 252

Direct-conversion receivers, 12Directional antenna, 6, 31Dispersive delay line, 235–237Diversity reception techniques, 295–298frequency diversity, 295–296IF/audio combiner, 297, 298polarization, 297, 298spatial diversity, 296–297

Double- and triple-conversion receivers, 11–12Double-balanced diode mixer circuits, 101–103, 117Double-balanced mixers, 10, 91, 92, 104–106,

107, 276Double-balanced modulators and oscillator

integrated circuits, 117, 118–119Double sideband suppressed carrier, 9, 11,

180–184, 185Drift, 135, 141, 147, 199, 245, 247cause, 27digital signal processing and, 227long- and short-term, 26preventing, 138warm-up, 136

Dummy load, 226, 283, 284Dynamic range, 31–32, 264–265

EElectrocardiogram, 86, 166Electromagnetic interference

receiver performance improving in environmentof, 275–294

source increase, 27television and cable television equipment and,

286–294Extraterrestrial noise, 14

FFaraday shielded coaxial cable, 291Farnsworth method, 298Feedback oscillator, 109, 110Feedthrough capacitors, 64Ferrite beads, 127, 131, 205Filters, 163. See also Intermediate frequency filtersabsorptive, 283–284bandpass, 23, 37–40, 62, 63, 105, 179, 181, 213,

218, 220, 228, 229, 253, 266, 272, 273, 275,276, 281, 290

ceramic, 25Cohn, 158, 159coupling to other filters, 168–169crystal, 25, 153–159, 163, 168, 169electromagnetic inference problems and, 281–282half-lattice, 155, 156, 157helical, 63, 64high-pass, 275, 281, 283, 286, 290, 293L-C IF, 151–153, 164, 165, 253low-pass, 143, 183, 189, 201, 213, 220, 229, 253,

256, 272, 275, 281, 283, 284, 286, 290mechanical, 25, 159–161, 163, 168, 169narrowband, 115, 220notch, 34–35, 281, 290passband, 152resonant frequency, 23SAW filters, 161single sideband, 231single-frequency versus bandpass, 61–63stopband, 281strip line, 65surface acoustic wave, 66switching, 173television, 289, 290tunable, 218, 219, 275vintage, 155

Formulaeaccuracy, 242amplitude modulation (AM), 248dBm, 13dBµV conversion to dBm, 14capacitance, 58, 64, 113, 158, 168


current, 195current gain, 71dielectric constant, 65distortion, 178emitter current, 71frequency accuracy, 144Friis’s noise, 19, 40, 273full duplex image, 41gain-bandwidth product, 76half duplex image, 41heterodyning, 10, 41intercept points, 274intermediate frequency (IF) rejection, 42intermodulation distortion (IMD), 268mismatch loss, 254mixer, 91noise, 4, 17, 18, 40, 68, 252, 273nth-order intercept points, 42Ohm’s law, 204oscillation frequency, 113padder capacitor, 60power, 240, 241Q, 53, 63resonance frequency, 45selectivity, 221signal-to-noise ratio (SNR), 5SINAD, 262spurs occurrence, 41stub length, 285temperature coefficient of frequency (TCF), 142TOIP, 42, 268transistor gain, 71voltage, 172, 241, 249voltage divider network, 13white noise, 16

Fourier series, 215, 216, 218, 236Frequency, 144accuracy, 144audio (AF), 181characteristics, 76considerations, 242–243diversity, 295–296domain usage, 216–218low, 138modulation (FM), 242, 261–263output level, 243range, 144, 242resolution, 144, 242stability, 135switching speed, 144, 242, 243

synthesizers, 143–150, 246, 247very low (VLF), 8, 211

Friis’s noise equation, 19, 40, 68, 273Front end overview, 37–43architectures, 37–39intercept points, 41, 42mixer and local oscillator performance, 39–40mixer attributes, 40noise performance of system, 40–41nth-order intercept point, 42, 43RF amplifier, 43second-order intercept point (SOIP), 41, 42spurious responses, 41third-order intercept point (TOIP), 42

Front-end bandwidth, 23,24Front-end circuits, 11, 229Front-end filtering, 45–66bandspreading, 60–61bandwidth of RF/IF transformers, 50, 52–53, 54component values for L-C resonant tank

circuits, choosing, 55, 56–57construction of RF/IF transformers, 49–50, 51cut-plate capacitor method, 60inductor-capacitor resonant tank circuits, 45–48local oscillator problem, 59–60padder capacitor method, 60parallel resonant circuits, 48, 49problems with IF and RF transformers, 53,

54–55, 56RF amplifier/antenna tuner problem, 57–59series resonant circuits, 45–48single-frequency filtering versus bandpass

filtering, 61–63surface acoustic wave filters, 66tracking problem, 57trimmer capacitor method, 59–60turned RF transformers, 48, 49UHF/microwave frequencies, 64, 65–66VHF/UHF circuits, 63–64

Fundamental overload, 288

GGain-bandwidth (GBW), 129, 130Gaussian (white) noise, 3, 4, 16, 258

HHam radios, 276, 277Harmonics, 130, 133, 134, 135, 146, 265, 276,

278, 280cable television and, 292, 294

Index 303

Harmonics (continued)complex wave and, 215distortion, 68, 143, 210, 217, 219, 225, 256,

262, 276IFM receiver and, 235IM distortion and, 266overload, 288push-pull amplifier, 82–83, 210selectivity testing and, 270sine waves, 111spectral purity and, 243–245, 252, 253third-order, 255TOIP and, 256

Hartley oscillators, 110, 111, 112, 113–116“Head room,” 275Heterodyning, 9–11, 106, 138, 181, 218attribute, main, 9IF notch rejection and, 34mixer and, 91, 278spurious response, 41TOIP and, 255

High-fidelity coupling, 52

IIC (integrated circuits), 169–172, 175IF. See Intermediate frequencyIFM (instantaneous frequency measurement

receiver), 234–236Image rejection, 24Image response, 67IMD. See Intermodulation distortionInductor-capacitor resonant tank circuits, 45–48Input attenuator, 31Insertion loss, 85, 100, 281Instantaneous frequency measurement receiver,

234–235, 236Intercept points, 41–43second-order, 41–42third-order, 42

Interference, at subscriber end, 293–294Intermediate frequencyamplifier, 11analog-to-digital converter (ADC) noise and, 228audio combiner, 297, 298noise limiters, 192notch rejection, 34–35passband shape factor, 25, 26rejection, first, 24–25selectivity, 279, 280spectrum analyzer and, 220

television and interference of, 289Intermediate frequency amplifier circuits, 163–173cascode-pair amplifier, 165, 166coupling to other filters, 168–169filter switching in, 173filters, 163, 164IC, 169–170, 171processing ICS, 170, 171–172successive detection logarithmic amplifiers,

172–173“universal” amplifier, 166–168

Intermediate frequency filters, 151–161bibliography, 161crystal filters, 153–159L-C, 151, 152–153mechanical filters, 159–161SAW filters, 161

Intermediate frequency/radio frequencytransformers

bandwidth, 50, 52–53, 54construction, 49, 51problems with, 53, 54–55, 56repairing, 55schematic of, 51tuned, 48, 49, 50

Intermodulation distortion, 68, 265, 266–268, 277audio signal line method, 266, 267cable television and, 292, 294dynamic range and, 264example, 268levels, 28receiver architecture and, 231signal-to-noise ratio (SNR) method, 267S-meter method, 267spectrum analyzer, 217–218, 219, 225standard method, 267–268television and, 288–289testing, 274

Intermodulation products (IPs), 27–29, 255, 266,275–278, 294

Internal spurii, 35

JJFET and MOSFETbalanced product detector, 183, 184connections, 76–77double-balanced mixer circuits, 98–100singly balanced mixers, 94–98

JFET preselector, 77–78Johnson noise, 16


LLamarr, Hedy, 233Leakagecable television, 293currents, 71–72

LNA (low-noise amplifier), 6, 19, 273Local oscillator (LO), 184microscan receiver and, 235mixer performance and, 39–40problem, 59–60product detector and, 183receiver architecture and, 229scanning superheterodyne receivers and, 234spectrum analyzer and, 219, 223television, 287types of, 109use for, 109

Local oscillator (LO) and frequency synthesizercircuits, 109–150

Barkhausen’s criteria, 110basic categories of VFO circuit, 110–111basic circuit configurations, 110, 111–119crystal oscillators, 119–135, 136frequency synthesizers, 143–150L-C variable frequency oscillators, 109oscillator accuracy and stability, improving,

135–143power supplies for VFO circuits, 110RF oscillator basics, 109–110

Lower sideband (LSB), 105, 106, 175, 181, 218Low-noise amplifier, 6, 19, 273

MMagnetostriction, 159, 160Manual gain control (MGC), 166Medical devicesblood pressure amplifiers, 100electrocardiogram, 86

Metal oxide semiconductor field effect transistor.See MOSFET

Microscan compression receivers, 235, 236Microstrip technology, 65Miller oscillators, 8, 125–127Mini-Circuits MAR-1 through MAR-8 devices, 85,

107, 108, 202, 204, 211, 212, 213Mini-Circuits SRA-1 and SBL-1, 101–103Minimum detectable signal (MDS), 21, 260, 263,

264, 265, 268, 274–1 dB compression point, 29, 263–264, 277Mismatch loss, improving, 253, 254

Mixersattributes, 40definition of, 91double-balanced, 98, 104–106, 107, 219, 276frequency converter circuits and. See RF mixer

and frequency converter circuitsJFET and MOSFET double-balanced, 98–100local oscillator performance and, 39–40postamplifier circuit, 169receiver architecture and, 229single-ended, 93–94, 95, 96, 219singly balanced, 94, 96–98spectrum analyzer, 219, 220superheterodyne receiver and, 272, 273types, 93–98

Modulation index, 249Monolithic microwave integrated circuits

(MMICs), 201–214basic circuit, 203–205broadband HF amplifier, 213–214internal circuitry, 203MAR-X circuits, 201, 202, 203, 205, 206–208mast-mounted wideband preamplifier, 212–213multiple device circuits, 208–211silicon, 201, 203universal fixed-gain preamplifier, 211uses, 201

Morse telegraphy, 11MOSFET and JFETbalanced product detector, 183, 184connections, 76–77double-balanced mixer circuits, 98–100dual gate MOSFET transistor, 183, 184singly balanced mixers, 94–98

MOSFET preselector, 80Motional reactance values, 154

NNoise, 14–19, 135, 179. See also Signal-to-noise

ratioamplitude modulation, 188analog-to-digital converter (ADC), 228background, 195bandwidth versus noise voltage, 16blankers and limiters, 191–193cable television and, 292in cascode amplifiers, 18–19classes of, 14detector, 193digital, 146, 245

Index 305

Noise (continued)factor, 17–18figure, 18floor, 5, 19, 223, 224, 258, 260, 263, 267gate, 234Gaussian, 3, 16, 258measurement, 263–270performance of system, 40–41phase, 91, 135, 146, 148, 223, 243, 247preselectors or preamplifiers and, 69quantization error, 228in receiver, 273receiver sensitivity and, 261sources through the bands, 15spectrum analyzer, 217spikes, 203temperature, 18thermal, 4, 16, 148, 247, 258, 273white, 16wideband, 143

Noise, signals, and reception, 3–6power of noise, in resistor, 4reception problem, 4–5strategies to improve, 5–6

Notch filter, 34–35, 281, 290Nyquest’s theorem, 228

OOhm’s law, 204Origins of radio receivers, 6Oven-controlled crystal oscillator (OCXO), 26,

122–124, 146, 246, 247Overtone oscillators, 134–135

PPadder capacitor method, 60Parasitic oscillations, 146, 245Passband shape factor, IF, 5, 26Performance, temperature, 122, 123–124Phase modulation, 248, 249Phase noise, 91, 135, 146, 148, 223, 243, 247Phase-locked loop (PLL) frequency synthesizer,

143, 147, 148, 150, 223, 245, 247Phase-locked loop (PLL) FM/PM detectors,

189, 190Pick-up, direct, 289Pierce oscillators, 127–128Piezoelectricity, 119, 120–122, 123crystal packaging, 121, 122, 123

definition, 120equivalent circuit, 120, 121, 122

Planning receiver system, 271–274calculating intercept points, 274noise, 273selectivity, 274sensitivity, 273, 274superheterodyne receiver, 271

PM (phase modulation), 248, 249Polarization diversity reception, important

parameters, 271Postamplifiers, 106–108, 169Preamplifiers, 31, 68, 106–108, 220. See also RF

amplifiers and preamplifiersaudio, 73broadband, 82, 213JFET, 196–197mast-mounted wideband, 212–213MMICs and, 201noise and, 69universal fixed-gain, 211–212

Preselectors, 6, 24, 67, 68, 276active, 68JFET, 77–78MOSFET, 80noise and, 69preamplifier versus, 68receiver architecture and, 229VHF receiver, 78, 79–80voltage tuned receiver, 80, 81

Printed wiring board (PWB), 65Pseudo-Gaussian (pink) noise, 3, 4

RRadio astronomy, 3, 14, 16“Radio Luxembourg effect,” 33Receiver(s)architectures, 3–12, 228–231, 245–248digital, 229direct-conversion, 12double- and triple-conversion, 11–12functions of, 6noise floor, 19noise measurement, 263–270role of, 1–2special purpose, 233–237spectrum analyzer, 215–226

Receiver architecture, 3–12, 228–231, 245–248audio amplifiers, 11


crystal video receivers, 6–7detector, 11direct-conversion receivers, 12double- and triple-conversion receivers, 11–12frequency synthesizer, 247front-end circuits, 11heterodyning, 9–11intermediate frequency amplifier, 11origins, 6output section, 247–248reception problem, 4, 5reference section, 246–247signals, noise, and reception, 3–6strategies, 5–6superheterodyne receivers, 8–9thermal noise, 4tuned radio frequency receivers, 7–8

Receiver performance factors, 1, 13–35bandwidth versus noise voltage, 16dBm, 13, 14dBµV, 14dBmV, 14dynamic performance, 27–35input signal voltage, 13, 14measurement units, 13–14noise, 14–19selectivity, 19, 22–26sensitivity, 19–22stability, 26–27static measures of, 19–27

Receiver performance in high EMI environment,improving, 275–294

Receiver tests and measurements, 239–270architectures, 245–248frequency, 242–243grades of instruments, 239improving the quality of signal generator use,

252–258modulators, 248–249, 250noise measurement, 263–270output signal quality, 242sensitivity measurements, 258–263signal sources and signal generators, 239–241spectral purity, 243–245sweep generators, 249, 250–252

Receivers, special purpose, 233–237Bragg cell receivers, 236, 237instantaneous frequency measurement

receivers, 234–235, 236

microscan compressive receivers, 235, 236scanning superheterodyne receivers, 234spread spectrum (SS) receivers, 233–234

Reciprocal mixing, 33–34Resonant frequency filter, 23Responsibility, cable television EMI, 293–294RF amplifiers and preamplifiers, 67–89, 272broadband 50 Ω input and output, 86–87broadband for VLF, LF, and AM BCB, 82broadband or tuned suing MC-1350P, 87–89classification by common element, 72–73configuration, 69DC load lines, Q point, and transistor biasing,

73–74frequency characteristics, 76JFET and MOSFET connections, 76–77leakage currents, 71–72MOSFET preselector, 80noise and preselectors or preamplifiers, 69problems with, 67push-pull, 82–86transistor biasing, 74–76transistor gain, 71transistors, simple bipolar, 69–71VHF receiver preselector, 78, 79–80voltage tuned receiver preselector, 80, 81

RF choke, 170, 205, 213RF mixer and frequency converter circuits,

91–108bipolar transconductance cell DBMs,

104–106, 107diplexer circuits, 92–93double-balanced diode mixer circuits, 101–103double-balanced mixers, 98JFET and MOSFET double-balanced mixer

circuits, 90–100mixer definition, 91note, 108preamplifiers and postamplifiers, 106, 107–108single-ended mixer, 93–94, 95, 96singly balanced mixer, 94, 96–98types of mixer, 93–98

RF oscillator basics, 109–110RF preamplifiers. See RF amplifiers and

preamplifiersRF/IF transformersbandwidth, 50, 52–53, 54construction, 49, 51problems with, 53, 54–55

Index 307

RF/IF transformers (continued)repairing, 55schematic of, 51

Room temperature crystal oscillators (RTXO),122, 124

SSchottky hot carrier diodes, 101Selectivity, 19, 22–26, 40, 274aspects of, 23–26definition, 67densitization and, 279, 280distant frequency (“ultimate”) rejection, 25–26first IF rejection, 24–25front-end bandwidth, 23, 24IF bandwidth, 25IF passband shape factor, 25image rejection, 24spectrum analyzer and, 221, 223testing, 268–270

Semiconductors, 69Sensitivity, 19–22, 40, 67, 228, 258–263,

273, 274Shape factor, IF passband, 25, 26Shielding, 285, 286SIDs (sudden ionospheric disturbances), 8Signal generator usage, improving, 252, 258Signal quality output, 143Signal sources and generators, 239–241, 242Signal-plus-noise-plus-distortion-to-noise

(SINAD), 19, 261, 262–263, 267Signals, noise, and reception, 3–6forms of signals, 3reception problems, 4–5strategies to improve, 5–6

Signal-to-noise ratio (SNR), 5, 14, 17, 19, 179, 258antenna mounting and, 69control, 273dynamic range blocking and, 264expressing, 5IMD measuring using, 267manipulating, 40preamplifiers and, 31reception and, 228, 263sensitivity and, 258, 261, 273signal-plus-noise-plus-distortion-to-noise

(SINAD) sensitivity and, 262strategies to improve, 5–6television and, 288

SINAD (signal-plus-noise-plus-distortion-to-noise), 19, 261, 262–263, 267

Sine waves, 29, 111, 130, 135, 145, 215, 216, 217,218, 248

Single sideband, 9, 11, 20, 21, 26, 40, 105, 106,139, 143, 153, 159, 180–185, 195, 231, 263,266, 273

Single-sideband suppressed carrier, 181S-meters, 196–197, 266, 267Spatial diversity reception, 296–297Spectral purity, 243–245, 252–253Spectrum analyzer receivers, 41, 103, 148,

215–226, 243analysis, spectrum, 218–221block diagram, 220frequency domain usage, 216–218harmonic distortion, 217intermodulation distortion, 217–218, 266, 267modulation, 218noise, 217resolution, 221–224signal generator and, 259uses, 216, 224, 225–226

Spread spectrum (SS) receivers, 233–234Spurious responses, 41, 103, 143, 146, 157, 220,

256, 274Square jaw detector, 9Squelch circuits, 270Squires-Sanders SS-1 receiver, 38, 276SSB. See Single sidebandSSBSC (single-sideband suppressed carrier), 181Stability, 26–27, 124, 135Standing-wave ratio, 92Stopband filter, 281, 286Stubs, 207, 290, 291Sudden ionospheric disturbances, 8Superheterodyne receivers, 8–9, 55, 56, 163,

271–27320 dB attenuator, 271audio power amplifier, 273audio preamplifier, 273bandpass filter, 272block diagram of, 9, 194demodulator, 273front end of, 23high IF amplifier, 273image rejection, 24image response, 67LB or FB filter, 272


low IF amplifier, 273mixer/high local oscillator, 272mixer/low local oscillator, 273mixers and, 91, 93purpose of, 8reciprocal missing, 33, 34RF amplifier, 272scanning, 234schematic of, 56second detectors, 175second IF BP filter, 272spectrum analyzer and, 219television, 287VFOs and, 109

Sweep generators (sweepers), 148–149, 249,250–252, 268–270, 284

SWR (standing-wave ratio), 92Synthesizer architecture, 146–150

TTCXO. See Temperature-compensated crystal

oscillatorTelevision, 286, 287–288audio rectification, 288cable systems, 291–293channelization, 293common mode versus differential mode

signals, 289direct pick-up, 289Faraday shielded coaxial cable, 291filtering, 289, 290fundamental overload, 288harmonic overload, 288IF interference, 289intermodulation distortion interference, 288–289leakage, 293responsibility, 293stubs for EMI elimination, 290, 291

Temperature, 136changes, 135, 136crystal ovens, 137digital signal processing and, 227self-heating, 137, 138thermal isolation, 136–137

Temperature-compensated crystal oscillator, 26,122, 124, 141–142, 146, 246

Tests and measurement. See Receiver tests andmeasurements

TGTP (tuned-grid-tuned plate oscillators), 133

THD (total harmonic distortion), 21, 263Thermal noise, 4, 16, 148Third-order intercept point (TOIP), 29, 42, 196,

219, 254–257, 265, 268, 274, 277, 278Third-order intermodulation distortion dynamic

range (TOIMDDR), 265Toroidal cores, 81, 84, 87, 99, 112, 115, 138, 140,

160, 211, 284, 289, 293, 294Total harmonic distortion (THD), 21, 263Tracking problem, 57Transformers, 163, 164, 166balun, 211, 290Hilbert, 185hybrid coupler and, 265IF, 182, 183wideband transmission line, 84

Transistors, 164basic diagram, 70biasing, 74–76bipolar, 168, 169, 176, 183, 197, 275definition, 69dual gate MOSFET, 183field effect, 176gain, 71JFET, 183MOSFET, 183origin of, 69simple bipolar, 69–71UHF/microwave, 202

Transmission line stubs, 285Trifilar winding, 83, 99, 155Trimmer capacitor, 57, 59–60, 113, 141Tuned radio frequency (TRF) receivers, 7–8Tuned-grid-tuned plate oscillators, 133Tuning scheme, 57“Tweet filter,” 35

UUHF/microwave frequencies, 64, 65–66“Ultimate” rejection, 25–26Universal wound, 50Upper sideband (USB), 105, 106, 175, 181, 218

VVaractors, 115, 142–143, 147, 199, 247Variable frequency oscillator (VFO) circuits, 26,

27, 109, 245, 296, 297basic categories, 110, 111power supplies, 110

Index 309

VCRs, 286Very low frequency (VLF) solar flare/SID

monitoring, 8VHF receiver preselector, 78, 79–80VHF/UHF circuits, 22, 63, 64VHF/UHR receivers, 231Vibration isolation, 139, 140

Voltage-controlled oscillator (VCO) circuit, 143,147–149, 189, 199, 219, 234, 247, 250, 252, 269

WWave analyzer, 218Wave trap(s), 281, 282, 285White noise, 3, 16


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