transmission and decoding in colour … bound...investigations qndecoding in colour television ......

JUNE 1964

IPhilips 'Research ReportsI ' EDITED BY THE RESEARCH LABORATORY. OF N.V. PHILlPS' OLOBILAMPBNFABRlEKBN. EIl'p)HOVEN. NETHERLANDS .

m 505 Philips Res. Repts 19, 195-280, 1964

TRANSMISSION AND DECODINGIN COLOUR TELEV:~S~ON*)

by J. DAVIDSE

3. INVESTIGATIONS qN DECODING IN COLOUR TELEVISION

AbstractThis chapter is concerned with decoding techniques in colour televi-sion. First an analysis is made of the essential functions of the decodingsystem. In particular, methods of separating the luminance and chromi-nance information are considered. Then the mechanism of self-decodingin one-gun display systems is dealt with. The errors inherent to thisdecoding method are analyzed. Subsequently the problem of how totranslate the NTSÇ signal into a driving signal for a one-gun displaysystem working in continuous colour sequence is discussed. A new typeof signal-translator circuit is dealt with, in which the overall decodingcircuitry is simplified. A subsequent section discusses the possibilitiesof modifying the transmitted signal to the needs of sequential displays.It is found that a modification in which a symmetrical subcarrier signalis transmitted in place of the conventional NTSC subcarrier signalpermits simplification of one-gun decoding techniques while not de-tracting from transmission quality. Accordingly, a proposal is put for-ward for a subcarrier signal employing quadrature modulation withmodified J- and Q-signals. A subsequent section of this chapter dealswith possibilities of proper utilization of the, vestigial-sideband compo-nents in the subcarrier signal. Practical methods are surveyed and anovel method is presented which is particularly suited for applicationin decodingsystems for one-gun tubes. This method employs two mixerstages both acting as elliptical-gain amplifiers, of which the first is fed bythe complete subcarrier signal including the vestigial-sideband compo-.nents, and the second by the double-sideband portion of the subcarriersignal only. It is shown that the elliptical amplification of the secondmixer can be modified in such a way that without extra circuitry, theconversion of 'the NTSC.signal into a CCS dot-sequential-signal isincorporated in the circuit. It is also shown that the Y-to:M conversion,needed for translating the luminance signal into the monochrome signalfor a one-gun tube, can conveniently be incorporated Into this circuit.The last part of this chapter is devoted to methods of improvingluminance rendering in colour transitions. Because of imperfect adheren-ce to the constant-luminance principle, luminance rendering is erroneousin transitions. Since the magnitude of the' errors depends on the sub-carrier signal, correction methods can be devised by proper utilizationof this signal. Various groups of such methods are discussed. It is con-cluded that there are certain promising solutions which provide partialcorrectiori but do not require complex additional circuitry.

3.1. IntroduetionThe discussion of transmission systems revealed the outstanding features of

*) Continued from Philips Res. Repts 19, .112-194, ,1964.

196 J. DAVIDSE

the NTSC system and led to the conclusion that this system was preferable focolour-television transmissions.Our study of principles and methods of decoding will therefore be based 0

the assumption that an NTSC-type colour-television signal is available in threceiver. This does not mean that our considerations will be without any valuif other transmission systems are concerned; the results will also be applicablin part to other band-sharing transmission systems.

3.2. Essential elements of the decoding system

3.2.1. Filtering of the chrominance signals

The available signal is band-shared, that is luminance and colour informatioare both transmitted within the video band. The luminance information is distributed over the entire video band while the chrominance information is contained in the subcarrier signal which occupies only the upper part of the videband. Inorder to prevent it from being disturbed by the luminance informatiofiltering is essential.The composite signal can be written

Et = Er' + Ac cos (ws + 1fJ), (3.1where Ac denotes subcarrier amplitude and 1fJ subcarrier phase. Decoding thchrominance information requires synchronous detection of the subcarriesignal. If no filtering is applie~ we have at the output of a synchronous detectoworking in the cosine phase

Eo = [Ey' + Ac cos (wst + 1fJ)] cos wst == Ey' cos wst + t Accos (2wst + lp) + t Ac cos 1fJ (3.2

The contribution t Ac cos(2wst + 1fJ) is outside the video band and can bremoved simply by a suppression filter. The term tAccoslp describes the wantechrominance signal, while theterm Ey'coswst is an unwanted contributio~due to "synchronous detection of the luminance signal". Splitting up Ey' intEyz' and EYh', where EYh' denotes the band-shared part of the video band, whave Eyz'coswst + EYl/coswst. The term EYh'COSWs! describes the unavoid-l

fromvldeodetector

r-----------~----~--~Ey,--___.

Fig. 3-1. (a) Block diagram of decoder employing chrominance filtering.

TRANSMISSION AND DECODING IN COLOUR TELEVISION 197

ble crosstalk of the band-shared luminance components into the subcarrier'Ignal. Itwas considered in detail in sec. 2..6.2. T.he te~ EYI'cosw8t containsnly frequency components above the passbands of the chrominance channels.[ence these components can be :filtered out in the receiver, either by having aand-pass :filterin front of the detectors or by having low-pass :filtersbehind theetectors. In practice a combination of the two is commonly employed (fig.-la). In-the case of "I-Q demodulation" :filters of different bandwidth areeeded for both detectors.The delay of the chrominance signals introduced by these filters must beompensated for by passing the luminance signal through a delay line. Thiselay line, which is found in every colour-television receiver, is a componenthose proper design and application is a source of considerable trouble. Inractice, delays between 0·5 fLsecand 1 fLsecare involved depending on the typef filters employed in the chrorninance channels.The required delay time can considerably be reduced if an alternative filteringethod is employed. This method consists of separating the chrorninanceformation from the composite signal by subtracting the output signals of twow-pass filters. The first low-pass filter passes the entire composite colour-elevision signal, the second passes solely the components below the subcarrierand. Figure 3-Ib shows the block diagram of a decoder based on this methodor the case of equi-band modulation. In the case of I-Q demodulation withmequal bandwidths a third low-pass filter is needed, the cutoff-frequency ofhich should be about 3 Mc/s.

Fig. 3-1. (b) Block diagram of decoder employing luminance filtering.

The delay line which is needed must compensate the difference in delay ofthewo low-pass :filters.A rough calculation can give some idea of the reduction inompensating delay. Let us assume that a subcarrier band from 4 Mc/s to- Mc/s must be separated from the composite signal and that the signal delayntroduced by a low-pass :filtersection is inversely proportional to its bandwidthhile the delay introduced by a band-pass filter section is inversely prop or-ional to its half-bandwidth. Assuming in both cases that simple constant-klters are employed it can be shown that for equal attenuation the low-passlters must contain three sections for every section of the band-pass filter.

. f the compensating delay in the case of the band-pass filter is 1 usec, the low-ass filters only require about 0·075 fLsec.

198 J.DAVIDSE

From vldeod.l.clor

Fig. 3-1. (c) Basic circuit of decoder employing luminance filtering.

Figure 3-1c shows schematically a convenient decoding circuit employing t .filtering method explained above. The corn.po site colour-television signal is fi~'•.to three detector circuits in parallel producing directly the primary-colour sinals ER', EG' and EB'. Each detector consists of two mixer stages. The le .hand stage is fed by the composite colour-television signal, It is designed .such a way that the direct amplification through this stage and the conversi~product of the input signal and the appropriately phased reference signal fro I

the subcarrier regenerator together produce the wanted primary-colour signThe unwanted conversion product of the low-frequency components of t~luminance signal and the reference subcarrier are,cancelled by the output of t~right-hand tube which is fed by the output signalof the 4 Mcfs low-pass fil~~'and tbe phase-inverted reference subcarrier. The subcarrier signal and t .reference-subcarrier signal are removed from the output signal by a suppressiolfilter, This suppression filter has th~ 'same function as the suppression filter i'the luminance channel of the conventional circuit shown in fig. 3-la.

3.2.2. Decoding withsequential display devices

3.2.2.1. Three-gun and one-gun decoding systems

In the preceding section it was tacitly assumed that a three-gun displa.system was employed. The detection has then to be accomplished by synchrnous detectors which derive the chrominance signals- from the sUbcarri1signap,2).. '"., Display systems of-the one-gun type can reproduce only one primary colouat a certain moment. They can therefore -be said to operate sequentially. In)certain given sequence, depending on the type of tube, samples of the primar

, . I


.2.2.2. Basic principles of one-gun display tubes

The described method of forming the driving signal by full detection of theiansmitted signal with subsequent gating of the output signals is quite un-ractical and leads to very complicated receiver circuitry 3). It is never appliedits.full strength in decoders for one-gun tubes. ln practice a simplified signalalways formed that approximates to the theoretically correct signal. We shallalyze this technique for the case-of picture tubes working with continuous

olour sequence (CCS) as this' type of operation is the one most commonlyplied. In this mode of operation the beam hits the colour phosphors alwaysthe same sequence, e.g. RBGRBG etc., in contrast to "reversed colour se-

uence" (RCS) where the colour sequence alternates, e.g. RBGGBRRBGGBR,tc. 4). For the latter case considerations can be put forward analogous to thosethe ces case.In order to make sure that the samples of the primary-colour signals are fedthe gun at the right moment it has to be known at any instant what colour

hosphor is hit by the electron beam. Existing one-gun picture tubes can beivided into two classes that solve this problem in an essentially different way.In the first group the beam is forced by an external signal to hit a certain

olour phosphor at certain predetermined instants. The best known and mostighly developed example ofthis type of tube is the one-gunrChromatron'<ê-").ere control of the beam position is accomplished' with a switching-grid exactlyarallel to and at a short distance from the phosphor screen that containsorizontal or vertical phosphor stripes. The signal at the switching-grid causesost-deflection of the beam, which therefore hits the different colour phosphorsa sequence determined by the waveform of the switching-grid signal and the

tructure of the screen. To prevent loss of information available in the colour-elevision signal, switching has to be carried out at a sufficiently high frequency.n order to simplify decoding circuitry it is in practice commonly done atbcarrier frequency.If the electron spot has, to move instantly from one colour phosphor to thether the switching waveform has to be very steep. However, feeding a steepectangular waveform at subcarrier frequency to the switching-grid with its,arge input capacitance is quite unpractical. Therefore in practice a sine-wavewitching signal is employed. The switching-signal amplitude needed for p~oper

lour signals have to be fed to the gun. These samples can of course be formed!y appropriately gating the three primary-colour signals obtained by full detec-;on of the composite NTSC signal, In this way a driving signal is produced,rhich can be looked upon as being a new composite signal. For every type of'ne-gun tube such a composite driving signal can be formulated. The picturebe can be said to decode this composite signal. For this reason the one-gunbe' is often called a self-decoding device.

200_ 1. DAVIDSE

post-deflection is so large that even with sine-wave 'switching the problempreventing unwanted radiation ofthe switching signal is considerable.The employment of a "smooth" switching signal causes the electron spot

move smoothly from one phosphor to the other. Moreover, in practice the pósfocussing will not be free from error. More than one colour phosphor would thbe hit at the same instant if the colour lines were not separated by bla"guard lines". In a typical "Chromatron" tube these black lines are ofabout t~•.same width as the colour lines. .In the second class of one-gun tubes the electron beam is allowed to find i

own way. It is not externally forced to hit a certain phosphor at a certain mment. However, means are provided that continuously detect the Position. of t~electron beam. For this purpose index lines are available in the screen stnictuin addition to the colour-phosphor lines. These index lines can, for instancconsist of a material with large secondary emission. The secondary electro~are collected by a collector. The index signal thus obtained is utilized to sychronize the se arming of the colour-phosphor lines and the gating of the prmary-colour signals 8,9,10,11,12,13,14). An alternative solution is to employ faindexing an extra light-emitting phosphor. Since it must be possible to di~tinguish the separate indexing pulses the decay time of this phosphor has to bivery small. Only phosphors emitting ultraviolet light meet this reqUiremen~The emitted u.v. light is collected by a photocell whose output signal serv~.again as the index signal P). '.

In tubes of this general type focussing of the electron beam must be vergood over the entire phosphor screen. The spot is not permitted to hit marthan one colour line at the same moment. As in practice the spot nevcan be infinitely small black guard lines are indispensable here also. In praeticindex tubes they are of about the same width as the colour-phosphor lines.

3.2.2.3. Formulations ofthe driving signalfor one-gun tubes

If a one-gun tube works correctly, in that the beam never hits the wronphosphors when a certain gated primary-colour signal is fed to the gun, threproduetion can be exactly right. The writing signal can for this case be writte

Ew = ER' GR + EB' GB + EG' GG , (3.3where GB, GB and GG denote gating functions working on the gamma-correted primary-colour signals ER', EB' and EG' (fig. 3-2).

~.=n--··-··-n------n------n---- GR

1 -----n--~---n------n-----rr 6o .. B

~ ------n------n-----n------ G6

_tFig. 3-2. Gating functions working on the primary-colour signals.

TRANSMISSION AND DECODING IN COLOUR TELEVISION

The usual simplification consists in the employment of ,a writing signal con-aining a d.c. term and a first harmonic term only. A more elaborate approxi-Ination employing in addition a second harmonic component has also been~roposed 16). The associated circuitry is then considerably more complicated., e shall not extend our considerations to this latter method.If the efficiencies of the colour phosphors are matched in such a way that.ith constant beam current the colour of the screen is white and if the instantshat the three colour phosphors arehit are separated by equal time .intervals,hich is commonly the case, we have, by virtue of the time-relationships be-een GR, GG and GB, .

w = a (ER' + EB' + EG') + fl [ER'COSwwt + EB'COS (wwt-2n/3) + EG'os (wwt-4n/3)], (3.4)

here a and {3 are constants and Ww denotes the writing frequency.If the d.c. terms and the a.c. terms in (3.4) are taken together we can rewriteis expression as

Ew = aEM' + fl Aw cos (wwt + rpw),

here Aw and rpwdenote amplitude and phase of the a.c. component of thevriting signal. From (3.4) it is clear that Aw and rpware functions of ER', EB'nd EG' and hence of time.

.2.2.4. Analysis of the mechanism of self-decoding in one-gun tubesLet us now analyze what the light output of the tube is if the gun is fed by a

ignal of the form (3.5). The electron beam, which is driven by the writing signals intercepted by the screen structure; the interception by the red-phosphorattern produces red light and similarly for green and blue. We can describehis process by multiplying the expression for the beam current by a functionepresenting the scanning ofthe screen structure. We sball call the latter functionhe "scanning function". It can be defined as the light output, due to the exci-ation of a certain colour phosphor, as a function of time if the beam current isrnstant. It may be noted that our assumption that the electron spot neverrits two different colour phosphors at the same time implies that the three

(

canning functions never overlap each other.First we consider the case that the scanning function consists of equally

{

paced pulses of infinitely short duration (fig. 3-3a). In the case of a post-ocussing tube this means that the switching signal is assumed to consist ofiuch very narrow pulses or that the colour-phosphor stripe width and the spot-Widthare assumed to be infinitely small. In the case of an index tube it means~lso that we assume the width of the phosphor lines and the width of the elec-\ron spot to be infinitely small.

201

(3.5)

202 1. DAVIDSE

Fig. 3-3. Typical scanning functions.(a) Scanning function if the width of the phosphor stripes and of the electron spot is infinitel

small. ,(b) Scanning function if width of phosphor stripes is finite while electron spot width is sti

infinitely small.. .(c) Practical scanning function if electron spot has finite width and non-uniform curren

density.

Ifthe tube characteristic obeys a power law we can write

Ib = k Ew."I.

Hence the red-light output can be written

(3.6

LR = k [Ew"l]* SR, . (3.71

where SR denotes the "red" scanning function. The notation [0]* denotes thainegative parts of the function between brackets have to be taken zero. In thi:way the fact is taken into account that below cutoff the tube current is zero

As for a scanning function of the type of fig. 3-3a

(3.8:

and in addition, by its very nature SR is never negative, (3.7) can also be writter

(3.9'

Fig. 3-4. Phasor representation of the writing signal.

EwSR can be found by taking the value of Ew at the moments determined by SRFrom the phasor representation of Ew (fig. 3-4) we see that if SR is taken sucl


lat it is in phase-synchronism with the phasor PER', the value of EwSR at the~ments determined by SR equals ' . .

us equals ER' if a = tand P = t.According to (3.9) the red-light output is

at the moments that Ew equals ER' obviously the function EwSR 'is positive.Likewise it is found that 2nf3ww seconds later the light output equals kEGid 4nf3ww seconds later kEB. We conclude that for the ideal case that theanning function is infinitely narrow, the reproduetion is exactly right, alsoen y =1= 1.We shall now consider what happens if the scanning function is still infinitelyeep but has finite width (fig. 3-3b). In the case ofan index tube this pertains toosphor stripes of a certain width while the spot is still infinitely small.For this scanning function too the relation SR')' _ SR is valid, hence (3.9) isso valid. However, EwSR can now not be evaluated as simply as in the formerse. We therefore expand SR into a Fourier series, obtaining

SR = [a(ER' + EB' +EG') + P{ER' cos Wwt+ EB' cos (wwt:- 2n/3) +EG' cos (wwt- 4n/3)}] [so + S1 cos wwt + S2 cos 2wwt + ...] _:_So a (ER' + EB' + EG') + s1P/2 (ER'-tEB'-t EG') + a.c. terms. (3.10)

If the tube characteristic is linear (y = 1) eq. (3.9) can be written LR = kEwSR) ]*.80 long as the tube is never driven beyond cutoff, which is the caser not too saturated colours this equals kEwSR, hence the red-light outputin this case proportional to EwSR and given by (3.10). As will be shown in

ec, 3.2.2.5, the a.c. terms in this formula form the mathematical expression ofe fact that the picture shows a striped structure. They do not contribute toe total light output which is entirely determined by the d.c. terms. From.10) we see that the light output is exactly proportional to ER' if

a/S1 = P/4so. (3.11)

In the case of infinitely narrow pulses SI/SO = 2, so here we have a/p = ,!,hich is in accordance with the res~lt of the calculation pertaining to that case.Ifthe tube is driven beyond cutoffthe relation [EwSR]* = EwSR is no longeralid. The beam current now also contains higher harmonics of the writingrequency and new d.c. terms will be introduced in the expression fot LR 'de-ending on the conduction angle of the picture tube. We conclude that with a

204 J. DAVIDSE

scanning function consisting of steep pulses"of finite width the reproductiis exactly right if r = 1 and the tube is not driven-beyond cutoff. If the tubedriven beyond cutoff the reproduetion is erroneous: The smaller the conductiangle of the tube·the larger the errors ..If the tube is non-linear, which in practice is always the case, then k(EwSR

no longer is proportional to EwSR. Inspection ofthe expression for EwSR (3.1shows that (EwSR)'Y will contain additional d.c. terms even in the case that ttube is always conducting. The errors in the reproduetion take the formcrosstalk between the primary-colour signals, as will be shown later.

Subsequently we shall consider the case that the scanning function does nconsist of steep pulses (fig. 3-3c). For an index tube this pertains to the praeticcase that the electron spot has finite width and current-density distributionnot uniform. Now (3.8) is no longer valid, so we have to work with (3'1Again we can expand SR into a Fourier series and once again EwSR is describby (3.10), although of course with new values for So, SI, ••• Sn.

If y = 1 and the tube is never driven "beyondcutoff, again a value for al {J cibe fou~d yielding exact reproduction. With y i= 1 crosstalk errors are preent agam.In practice the. situation is still further complicated by the fact that the Wid~

of the electron spot depends on the deflection of the beam and on the mome .taneous value of the beam current ("spot blow-up") .. As a consequence tscanning function depends also on the beam current and hence on the writiJsignal. This causes areas of high brightness to be reproduced with too sm5saturàtion. To improve the rendering of bright colours usually a correcti~signal is added to the luminance signal. Such a correction signal can for i~1stance be obtained by envelope detection of the a.c. component ofthe writi~signal ("diode correction") 17). "

Itwill be clear that exact numerical computations on the reproduetion errotaking into account all effects occurring in practice, will necessarily be vecumbersome. Such numerical computations ha~e been carried out for t~"Apple"-tube (an index tube with secondary-emission indexing) emPIOYi~certain decoding instrumentation 18.19). However, such numerical computtions, though indispensable as a final check of colour fidelity for a completeone-gun display design, do not give much insight into the general nature of t~colour distortions. To obtain such general insight a formal mathematical aj~waCh, dealing with a simplified but sufficiently ,.ealistic case, can be of valu

~.2.2.5. Mathematical analysis of one-gun decoding errors

Let us assume again y =2 in order to siinplify the calculation. Furthermolilwe assume that the tube is not driven beyond cutoff, which is valid for not tosaturated colours. Finally we assume the scanning function not to depend 01the beam current (no "spot blow-up"). Equation (3.7) then reads


-j

xpanding SR into a Fourier-series:

R = k[aEM' + PAw cos (wwt + IPw)]2 (so + SI COSwwt + S2COS2wwt + ...)k[so a2EM'2 + tsoP2Aw2 + SlapEM'Aw cos IPw+ tS2p2Aw2 cos 2IPw]+.a.c. terms. (3.12)

Employing the definitions of EM', Aw and IPw (3.5), we can express theseantities in ER', ~G' and EB'. We obtain

LR = ER'2 (a2so + tp2so + iP2S2 + apsI) +(EG'2 + EB'2) (a2so+ tP2so-tp2s2-taPsI) +(ER' EG' + ER' En') (2a2so~tp2so-tP2s2 +tapsI) +EG'EB' (2a2so-tP2so + tP2S2-apsI) + a.c. terms: (3.13)

s a check we can substitute the values of So, SI and S2 for infinitely narrowulses (SI = S2 = 2so). It is then found that the coefficients of the crosstalkrms vanish, so that here the decoding proves to be ideal, as was to be expected.Expression (3.13) describes the red-light output as a function of time. How-

ver, what interests us is the distribution of the red light over the screen structure.o learn this we note that, no matter the spot size and shape, the red light islways emitted where red phosphor is available. Hence, all the light emitteduring one period of the scanning function is emitted by one phosphor line.ence, to find the light output of one red colour stripe we must integrate (3.13)ver one period of the scanning function. If we do so, the a.c. terms of course10 not contribute. We can therefore say that the a.c. terms in the time function3.13) are the mathematical expression of the fact that the reproduced imageecessarily has a striped structure.By the way, we note that the visual effect of the light contribution describedy the a.c. terms also includes secondary phenomena such as beat patternsetween the stripe structure ofthe screen and periodic structures in the pictureontent (moiré effect). We shall not deal with such phenomena in the presentiscussion.It is interesting to look at this conclusion from yet another point of view.

nvestigating the general properties of decoders for three-gun tubes we haveeen that an essential element in the decoding system is a filter preventing inter-erenee between the luminance and chrominance components in the compositeignal. Itmay be noted that this apparently essential element is not available in

f,'he self-decoding one-gun system. Inspection of the derivation of (3.13). revealshy this is so. It appears that the terms corresponding to the interference effectn the three-gun decoder contribute in the one-gun system only to the a.c.terms in the output signal. So the remarkable result is obtained that the inter-Iference effect in the three-gun system, in the one-gun system is included in the

206 J. DAVIDSE

.... striped structure ofthe repr.oduced image. The striped structure of the.pictureunavoidable and essential for the one-gun system. It can not be eliminatedthe employment of electric filters as in the three-gun system. If filtering is eployed, for instance, in a one-gun decoder for gated tube operation, where :firthe primary-colour signals are formed with the aid of synchronous detector]the image reproduetion reintroduces the phenomenon.As the output of the decoding element in the self-decoding one-gun displ~

system is not an electric signal but light, the only way to reduce the visible effe~of the striped structure is the employment of optical filtering. This can be dO~W.iththe aid of a so-called lenticula.r screen, that is a screen consisting of parallcylindrical lenses of optimally chosen design placed in front of the phosphscreen. However, if such optical filtering is not employed the stripe structuin the one-gun reproduetion will in..general be less annoying than the interfi~ence effect .would be in three-gun reproduction. This is because there are twdifferences in the decoding mechanism. In the case of the three-gun systemusually have sine-wave detection, whereas in the one-gun system we have essetially pulse-type detection. A more important difference is that in the three-system the demodulation as such is followed by a non-linear operation on tsignal (the non-linearity of the picture tube), while in the one-gun system thnon-linear operation precedes the demodulation of the composite signal by thscanning of the screen structure. Theoretically, if the demodulation in the thre9gun case were carried out with sharp pulses of infinitely small width rath9than with sine-wave signals, a similar reproduetion as with the one-gun systeJwould be obtained.Let us now analyze the expression found for the output light, viz. the no~

periodic terms in (3.13). Ifthe detection were ideal it would equal ER'2. Inspeetion of the formula shows that this would require a and fl to be such that simujtaneously the following conditions are fulfilled:

a2so + 1-(12so- tfl2s2 - tafls! = .0 )~a2so -1:f12so - ·;l:f12S2+ tafls! =.0 .2a2so - tfl2so + tfl2s2 ~ afls! = 0 (3.14

The quantities so, s! and S2 are determined by the shape of the scanning function. It is at once clear that in general a and fl can not be, chosen such that thrthree equations (3.14) are fulfilled. Thus the coefficients of the terms (EG'2 +EB'2), ER'(EG' + EB') and EB' EG' in (3.13) are in general not zero. Apparentljthe decoding errors take the form of a certain crosstalk between the primarycolour signals, which can be minimized by a judicious choice of the ratio fllaThe best choice depends on the values of so: s! and S2 and hence on the shape olthe scanning function.For convenience we write (3.13) in an abbreviated notation:

LR = mER'2 + n!(EG'2 + EB'2) + n2ER'(EG' + EB') + nSEB'EG';' (3.13a:

Ij


.G and LB are given by similar expressions, which can be found by simple cy-lie alteration of the expression for LR. '. '.

lweshall now compare the transfer of two colours, differing in luminancenly. Let the input quantities for the first colour be ER, EG and EB. and those,fore second colour pER, pEG, pEB. For the first colour we have . 'JR= mER +m(EG +EB) + n2ER'(EG' + EB') + n3EB'EG', since, for y = 2.JG'= Ei, etc. LG and LB are again found by cyclic alteration.For the second colour we haveR :_ pmER +pnl(EG '+ EB) +pn2ER'(EG' + EB') +pn3EB'EG': etc.e conclude that the output colours differ by their luminance only, so satura-

.on and hue are not affected.Let us now compare two colours differing in saturation, but not in hue. The

nput quantities be denoted by ER, EG, EB and (ER +c), (EG+ c), (EB :t c).r~~output quantities for the first colour areR= mER + nl (EG + EB) + n2ER'(EG' + EB') + n3EB'EG',G= mEG + nl (EB + ER) + n2EG'(EB' + ER') + n3ER'EB', " (3.15a)B = mEB + m(ER + EG) + 1l2EB'(ER' + EG') + n;EG'ER' ...

For the second colour we haveR = m(ER + c) + nl(EG + c + EB + c) + n2(ER + c)' [(EG + c)' +EB + c)'] + n3(EB + c)'(Ee. + c)',

LG= m(EG + c) + nl(EB + c + ER + c) + n2(EG + c)' [(EB + c)' +ER + c)'] + n3(ER + c)'(EB + c)',LB = m(EB + c) + nl(ER + c + EG + c) + n2(EB + c)' [(ER + c)' +(EG + c)'] + n3(EG + c)' (ER + c)'. . (3.15b)

What interests us now is whether both colours have still the same hue. Inorder to investigate this point we determine what colours of full saturation havethe same hue as the given colours. This can be done by subtracting equalquantities from LR, LG and LB (that means, we subtract a certain amount ofwhite light) until the smallest of them becomes zero. If we assume for instancethat LB is the smallest ofthe three we obtain in this way for the first colourLR = m(ER-EB) + nl(EB-ER) + n2EG'(ER'-EB') + n3EG'(EB~-ER'),LG = m(EG-EB) + nl(EB-EG) + n2ER'(EG'-EB') + naER'(EB'-EG'),LB ~ O. .' (3.16a)

And for the second colourLR = m(ER-EB) + nl(EB-ER) + n2(EG + C)'[(ER + C)'-(EB + c)']+ n3(EG+ c)' [(EB + c)' -(ER + c)'],L~ = m(EG'--EB) + 1l1(EB-EG) + n2(ER + c)'[(EG + C)'-(EB +c)']+ n3(ER + c)' [(EB + c)' ~(EG + c)'], ' . .LB = O. _:_ ~(3.16b)

207

208 J. DAVJDSE

These colours are of equal hue only if the ratio LRILa for both colours is tsame. Obviously this is generally not the case. However, if ER == EB: orEa =' EB the required condition is fulfilled. It can hence be concluded that funsaturated primaries and their complementaries the transfer process does nintroduce a hue shift, while for colours off the lines joining the primary colouand their complementaries in the colour triangle increasing hue shifts occur.

So much for the general properties of the derived expression for the coloerrors due to the detection method. For a numerical evaluation the values so,and S2 pertaining to the scanning function under investigation must be, sustituted. As an example we consider the case of the idealized scanning functio

lof jig. 3-3b with steep edges. Let the duty cycle of the waveform be 1 : 6. Wihave then So = 1;, SI = l/n and S2 = v'3;2n.It seems reasonable to choose the ratio alP such that the coefficient nl i

(3.13a) is minimum. Substituting the values for Soh and 'S2 into the equatiofor ni we find

2 (1 1 fJ2 v'3p2 ..1 P)nl = a - +------ -- . (3.1

1

. . 6 12a2 16n a2 2na

By differentiation ofthis function we find that nl is minimum for Pia = 1·6 I

As could be expected this value differs from the value found for the case ofscanning function consisting of infinitely narrow pulses, where we founPia = 2.

Substituting Pia = 1·61 into (3.1 3) we obtain

LR = a2 [1'06 ER + 0'03(Ea + EB) +0·23 ER' (Eo' +EB')-0·03 EB'Ea']."

Choosing a2 = 0·94 we obtain

LR = ER + 0'03(Ea + EB) + 0'21ER' (EG' + EB')-0·03EB'EG'. (3.18

This expression indicates that the crosstalk effect caused by the non-lineadetection system is quite moderate which is confirmed by experiments.

Summarizing we conclude that the combined effect of tube non-linearityfinite phosphor stripe width and spot size gives rise to er_rors in the reproduce.colours if sine-wave driving of a one-gun picture tube is employed. Howeverthese errors are small enough that they may be tolerated in practical applica

•. '. . 1.

tions where very exact colorimetrie accuracy is not required (i.e. for home re.--ceivers).

3.3. Decoding circuits for one-gun display tubes3.3.1. General theory of signal translators

In the preceding section we have found that one-gun picture tubes operatinin continuous colour sequence (CeS) work satsifactorily if they are fed bydriving signal consisting of a linear combination of the three primary-colou


gnals and a component at colour-switching frequency. The composite writingIgnal therefore resembles the composite NTSC signal. This suggests that the~mposite writing signal can be formed from the available detected NTSC!gnal in the receiver more directly than by full splitting-up into the three prima-~-colour signals., .: .IFor RCS operation a similar derivation can be made: It then turns out that,l addition, a component at two times the writing frequency should be available.

the resemblance to the NTSC signal is therefore much less than in the CCSse. It is true that in the 'RCS case also the required writing signal can b.ermed by transformations of the composite NTSC signal, but here the total(fort is so considerable that the overall circuit is hardly less complicated thanat employing full detection and remodulation or gating.The transformation in the CCS case is certainly interesting enough to beiscussed in greater detail. The NTSC signal can be written

,t = Ev' + 0'88(ER'-Ey') cos rost+ 0'49(EB'-Ey') sin rost.

he CCS signal can be written

w= a(ER' + EB' + EG') + P [ER' cosrowt + EB' cos(rowt - 2n/3) +EG' COS(roiot- 4n/3)]. ,. (3.4)

(3.19)

or an ideally shaped scanning function consisting of infinitely narrow pulses,= t ang.p = t.We shall base our further considerations on these values.Applying simple trigonometrie operations we can rewrite (3.4) as .

w= t(ER' + EB' + EG') + 0·89 (ER'-lfy') cos(rowt-19°) ++ 0·74 (EB'-Ey') sin (rowt-21 0). (3.20)

, .hasor representations of the a.c. terms of the NTSC signal and the CCS signalre given in fig. 3-5. .Comparison of (3.19) and (3.20) shows that the transformation ofthe NTSC. .

ignal into the CCS signal requires three steps 20,21,22).

a) The subcarrier frequency has to be replaced by the writing frequency. Ifhe writing frequency equals the subcarrier frequency; as in one-gun focus-ask tubes, this step can be omitted.

b) The signal Ey' has to be transformed into t(ER' + EB' + EG'). For con-enience weshall from now on denote the latter signal by EM' (not by aEM' ase did in the preceding calculations where this was more convenient). We shallall EM' the "monochrome signal". 'c) The a.c. terms of the NTSC signal have to be transformed into the a.c.erms of the CCS signal. .At present we shall confine ourselves to points (b) and (c), assuming that theIriting frequency equals the subcarrier frequency. As to (b): translation from.Ey' to EM' requires the addition of EM' - Ey' to the available signal Ev': The

210 J.DAVIDSE

~[O.li9(£S-£Y)

Ey: 0.59 EG+o.JOE~+o.11E;

g'

O.74(Eá-S;)

fh=O.33(EG +ER +E8)11

Fig. 3-5. Phasor representations of (a) NTSC signal and (b) CCS writing signal.The right-hand phasor diagrams are equivalent to the left-hand diagrams.

signal EM'-Ey' can be written as a linear combination of ER'-Ey' a~EB'-Ey',

EM'-Ey' = 0·17 (ER'-Ey'). + 0·27 (EB'-Ey'). . (3.2

It ·can. hence. be obtained by synchronous detection of the subcarrier sign'Calculation shows that the demodulation should be carried out at a phase of 1and a gain of 0·58.

As to (c), translation from the NTSC subcarrier signal into the CCS sucarrier signal requires a change ofthe lengths, ofthe (ER'-Ey') and (EB'-Eyphasors and ofthe angle between them (see fig. 3-5). This can be done by appl)ing the sub carrier signal to a so-called elliptical-gain amplifier, that is a circin which the gain is governed by the phase of the.input signal. Elliptical ampfication can be brought about by mixing.the input signal with a second-harmnic subcarrier of proper phase, If the _input signal is Accos( rost + 1jJ),the ratioconversion gain to direct gain is given by m, and () is the phase of the seconharmonic of !he subcarrier frequency, we have for the output of the amPlifi1

Ac cos(rost + cp) [1 + 2m cos (2rost + ())]. Ac cos (rost + 1jJ) + .+ mAc cos (ro~t+ ()-1jJ), . '. . (3.22

if we omit the t~rm at 3ro8 which can be removed by filtering. By properl!

ioosing the values of m an~8Vtl signal,can be made to be equivalent to the>.siredces signal 22). .

3.2. Simplified signa~-!ransla!or circuit

We can: now draw the.block diagram of a complete signal translator. It is -

~

own in fig. 3-6. The delay' line in the Y:cbannel has to compensate for theJay introduced by the band-pass filters in the translator sections. The delayne in the M-Y channel compensates for the delay introduced by the extraand-pass filter in the elliptical-gain channel. For proper operation of the cir-it, two refererice signals, one at/s and one at 2Js have to be provided by thebcarrier regenerator. The signal at/s must be entirely free from componentst 2/s, while the signal at ~/s must be entirely free from components at/s.

Composite NTSC .

from videodefector

Fig. 3-6. Block diagram of conventional decoder for ces one-gun picture tube.

Itmay be noted that the overall circuit is rather complicated. The splitting-upf the various functions demands the use of several filters and delay lines while,noreover, the requirement that both reference signals should be free of un-vanted components introduces a complication.As in the case of the three-gun decoder, here too a simpler design is obtained'the separation of the luminance signal and the subcarrier signal. is carriedut by low-pass filters. Figure 3-7 shows a possible design based on this prin-iple ~3)employing two mixer tubes. .. The left-hand tube amplifies the !-signal, acts as a synchronous detectorreducing EM' - Ev' by .the multiplication of Accos( wsl + 'IJ!) and the reference

lignal A cos(ws! + ffJ) and acts 3:s,an elliptical gain amplifier by virtue of theI'econd reference signal Bcos(2 Ws!+ B) .. The only unwanted components\vhich .are not automatically removed by the output low-pass filter are thebirectly .transferred reference signal Aeo's( ws! r+ ffJ) and the term desèribed by\he product E';' Acos(ws! +; ffJ). These terms,' however, are cancelled by the,~utPu~ ~f the right-hand tube. . '

,

212 J. DAVIDSE

+

r---'.------4-"---,r------l, Adder'

t!F~)r-J L...:.._....:oIII ,I ."II

, I

r----~-rE~y---+-----J

Composite NTSC

E; +A,cosr""l+tp)

Fig. 3-7. Simplified decoder for ces one-gun picture tube.

-An analysis of the operation of the .circuit shows that the circuit in its basform produces the Ey' -signal at its output with an amplitude which is two timétoo large. There are several ways of compensating for this error. One possibiliis shown in fig. 3-7 by the connection drawn in broken lines: at the output tproper amount of Ev' -signal is subtracted from the output signal. A secopossibility is the application of "chromaboost" somewhere in the circuit, this, the amplification of the circuit is made to depend on frequency in suchmanner that signal components around the subcarrier are amplified' twias much as the low-frequency components. Various designs of decoders of tgeneral type have been built and investigated experimentally by Van Odehoven 24).

The advantages of this type of decoder are obvious, it requires fewer tuband filters while, moreover, the only delay line needed has to compensate for tsmall difference in delay of both low-pass filters. In addition, no "cleanreference signal is required.

3.4. Modification of the transmitted signal for simplüying one-gun detection

3.4.1. Can a one-gun tube be devised needing the NTSC signal as the drivinsignal?

In the preceding section it was shown that the composition of the NTSsignal resembles the writing signal for a ces one-gun display tube. Thanks t·this similarity no full detection to the three primary-colour signals is necessarbut simpler detection systems, known as signal translators, can be emPIoye~The problem now arises,whether it is possible to £, acilitate the translation procesby modifying the transmitted signal slightly. .. An obvious question is whether a different construction of the phosphoscreen and the colour-scanning mechanism would enable a' tube to be madirequiring the actual NTSC signal as its driving signal. Such a modification oi


f required CCS writing signal can, for instance, be evoked by having unequal•aces between the red, green and blue colour stripes of the screen so that thefee scanning functions are not 120° apart. Though the writing signal neededr such a modified tube differs from the symmetrical signal needed for a sym-~trical screen construction, one property of the writing signal remains valid:lring one cycle of the writing frequency the signal has at three different mo-~nts to be precisely equal to the red, the green and the blue primary signal.~et the ~o~ulation for t!Ie, writing signal, of a generalized CCS one-gull,be be WrItten' I

I~aIER' + a2EG' + aaEB' + (cIER' + C2EG' + caEB') cos wwt +(dIER' + d2EG' + daEB~) sin wwt. (3.23)

t tI, t2 and te be the moments at which the signal should represent the re~_, -- - ' 'en and blue primary-colour signals. We have then \

, • • t

R' + a2EG' + aaEB' + (cIER' +C2EG' + CaEB') cos WwtI + (dIER' + .,d2EG' + daEB') sin WwtI == krER' (3.24)

d similar expressions for t z and te. The. quantities k-, kg and kb depend onluminous efficiencyof each of the colour phosphors and on the width of theosphor stripes.The coefficients of the left-hand terms in these identities should be equal toe corresponding coefficients of the right-hand terms. In this way nine equa-ns are obtained in the quantities al, a2, aa, CI~ C2, Ca, dl, dz, dz, tI, tz; ta,kg, kb. The first nine of these quantities describe the composition of theen writing signal, while the remaining six describe the composition of thereen. Hence we have nine' equations in six unknown parameters. By substitu-g the parameters of the NTSC system we can easily see that there is nolution to our problem. However, it may be asked whether our problem mightsolved if the NTSC signal were slightlymodified, in such a way that its essen-Iproperties as a transmission system were not affected. The most essentialtures of the NTSC signal are its adherence to tbe constant-luminance prin-le and modulation of the subcarrier by linear combinations of colour-dif-rence signals. The general form of a composite colour-television signal com-ing with' these basic requirements can be written= Ev' + ao(ER'-Ey') cos wst + PO(EB'-Ey') sin (wst +x). . (3.25)The question is now whether ao, Po 'and X can be chosen such that Et atree.different moments within one cycle of the subcarrier frequency representse three, primary-colour signals. This requires

, +ao(ER'-Ey') cos Wstl + PO(EB'-Ey') sin (Wsil + x) == k-Es', (3.26a)

1:" + ao(ER'-Ey') cos wst2 + PO(EB'-Ey') sin (Wst2+ x) == kgEG', (3.26b)

II:" + ao(ER'~Ey') cos wsta+ PO(EB'-Ey') sin (wsta + x) == kbEB'. (3.26c)

k- = 1 and cos x = ± l/ao.

Similarly (3.26b) yields,

kg= I andPo [±{1'- (0'51/ao)2}t cos x-O'5l sinx/ao] , 0'19, .(3.

(3.

214 , J.DAVIDSE

Substituting Ev' = 0·59 EG' + 0·30ER' + 0·11EB' and putting coefficiein the left-hand terms of (3.26a) equal to corresponding coefficients in the righand terms we have

while (3.26c) yields

kb = .1 and cos X = ± I/Po .' (3.

Eliminating ao and Po from the equations, we obtain an equation in sireading

'. . (0'59)2 - (0'30)2 - (0'11)2sm X = ---::-----::-:-----::------'-± 2 X 0·30 X 0·11

(3.

Hence,

sin x =.± 3·8.

This means that no value of X can be given complying' with the requiremeIt is therefore impossible to design a phosphor screen requiring a driving sigthat exhibits the essential properties of the NTSC signal.. .

3.4:2. Modification of certain design parameters of the transmission system

3.4.2.1. Modification of primary colours and choice of reference white;

A different approach to the problem of how to adapt the transmitted sigto the requirements of a ces display tube is to maintain the requirements,symmetry for the writing signal while modifying one or more of the parametof the transmission system.' The symmetry of the writing signal requirestransmitted luminance signal to be of the form tER' + fEB' + tEG'. Itpossible to give the luminance signal this form by changing the point of reilence white or by changing one ormore ofthe primary colours of the transmissisystem. Itmust be noted that only very slight .shifts of these quantities, whchoice is based on the properties of colour perception, can be allowed. -Ktarëv 25) has calculated how the point of reference white should be shifted if tprimary colours remain' what they are in the NTSC system. It turns out tha

. considerable shift towards blue is required which is certainly unacceptable.further possibility, investigated by the same author, is to change one of t

TRANSMISSION AND DECODING IN COLOUR TELEVISION . 215

mary colours. Shifting solely the green primary leads to a non-existing colourhtside the CIE colour triangle). The same applies. for a shift of the red pri--try. Shifting the blue primary turns out to be possible in principle. However,~ shift required (from ~ = 0'14; y ---:0·08 to x = 0,12; y = 0'20) is so large~t the gamut of reproducible colours would decrease considerably, whichrkes this method unpractical. . ' .:'

t2.2. Modification of the subcarrier sig~alrom the foregoing it can be concluded that changing the colour points of

erenee white or the primary colours would detract too much from the qualitythe transmission system. If these parameters remain what they are the com-sition of the luminance signal also remains unaffected. Any modification ofe luminance signal aiming at simpler one-gun decoding would detract fromconstant-luminance properties, It is therefore not possible to transmit thenal EM' instead of Ev', This means that it is impossible to discard the neces-y of having ~ Y-to-M converter in the decoder. Thus, if anything can be doneimprove the adaptation of the transmitted signal to one-gun decoders it hasbe found in a modification of the subcarrier signal. ,The obvious modification of the subcarrier signal is to choose it such that theed for the elliptical-gain conversion is eliminated in a decoder for a symmetri-I CCS one-gun tube. This requires transmission of the symmetrical CCS sub-rrier signal instead ofthe NTSC subcarrier signal 26). Such a modifica.tion isrmitted only if the transmission features of the neW signal are not inferiorthose of the original signal. We have therefore to investigate whether theodified signal complies with the requirements' to be met in general by theiboarrier signal. . ,As we have seen earlier (sec. 2.2) the composition of the' subcarrier signaldictated by the following considerations:) The subcarrier must be modulated by linear combinations of ER'- Ey' andB'-Ey'.~) The sub carrier signal should not extend too much in the blacker-than-lack and the whiter-than-white regions of the dynamic signal range. Common-tan "overswing" of 33% ofpeak-white is'assumed to be tolerable. . ,

~

') The effect of improper phase of the subcarrier signal should be such thate noticeability of Ph.a.se erro.rs. depends not too much on the hue to be rendered.) The signal-to-noise ratio and the wanted-to-unwanted signal ratio for thebcarrier signal should' be satisfactory." ,,'.,"

tTOgether with (b), our additional requirement that the signal be symmetricallly.determines the. composition ofthe chrominance signal.Jfthe symmetricalbcarrier signal is transmitted together with the standard .luminance signal;'s amplitude for saturated primary and complementary colours.should equal,·44 if the maximum overswing. which occurs for blue and y~llow, :is p.ot~o,

216 J. DAVIDSE

exceed 33%. Figure 3-8 shows the phase and the amplitude of the subcarrfor saturated colours together with an equivalent representation with tcolour-difference phasors. This figure also makes clear that "the symmetri

O.S9([~-m 111~o.44.iRed

Yel/ow171°0.44

291°:C0.44 i yon

Fig. 3-8. Phasor representation of modified subcarrier signal.

signal can be formed by modulating linear combinations of ER' - Er' a~EB' - Ey' onto the subcarrier, so that requirement Ca)is also fulfilled. ,IAs to point (c), De Vrijer27) has investigatedthe optimum composition oftj

subcarrier signal if the effectof phase errors on the hue should be as indepenent as possible of the phase of the subcarrier signal. His investigation takinto account the differentlal sensitivity of the human eye to colour and Jconcerned with a subcarrier signal composed of linear combinations of ER'Ey' and EB' - Er' while the "overswing" should not exceed 33% of peak whitFigure 3-9 gives this optimum subcarrier signal. If we now compare fig. 3-

RedO.66(ER-EY)

O.SI(Eá-EY)Fig. 3-9. Subcarrier signal for' best adaptation to noticeability of hue errors.

with figs 3-5 and 3-8 we see that the symmetrical signal is even closer to tln"flat-hue-error" signal than the NTSC subcarrier signal. It can be conclude,that, from this point of view, there are no objections against modifying tlusignal in the way proposed. < •

TRANSMISSION AND DECODING IN COLOUR TELEVISION.•• ' .p ._

ifhe only point remaining at this stage concerns the signal-to-noise ratio for'~ subcarrier signal and the mutual crosstalk of the luminance signal and theI . .bcarrier signa1.We note that the proposed modification consists mainly of a~uction of the amplitude of the ER'-Ey' component in the signa1. This.kes the signal-to-noise ratio for this component smaller than in thé NTSCtnal. However, it is well known that the standard NTSC system-is much moresitive to noise in the luminance signal than to noise in the subcarrier signal'e sec. 2.9, numerical data are also given by Burgett 28)). It seems, therefore;lly permissible to transmit one of the components of the subcarrier signalth smaller amplitude than in the NTSC signa1.The crosstalk of the subcarrier signal into the luminance channel will beghtly smaller than in the standard NTSC system since the average amplitude.the subcarrier signal is smaller. On the other hand, the crosstalk of the high-equency components of the luminance signal into the subcarrier channel willsomewhat greater in the modified system.In conclusion we can state that the confrontation of the proposed signalmposition with the basic requirements to be met by the subcarrier signalows that transmission of the' symmetrical signal does not jeopardize any ofe essential features of the NTSC signa1.

11-2.3. Choice of the 1- and.Qssignals in the modified sub carrier signalHaving justified the proposed modification of the NTSC signal we have nowinvestigate how the broad-band I-signal and the narrow-band Q-signal,

hich are modulated in quadrature on the subcarrier, should be chosen.IIn the NTSC signal the 1- and Q-axes are selected such that their loei coverrrtai~ colours in :he CIE colour diagr~~. If the co~position .of t~e subcar~iergnal is changed ID the proposed way It IS not possible to mamtam the loci ofbth the I-axis and the Q-axis in the CIE diagram. When one of these axes hasren chosen the other is determined by the already fixed composition of theibcarrier signa1. Since the latter differs from the composition of the NTSCIlbcarrier signal it is obvious that no more than one of the axes can be made to

~

incide with the corresponding axis in the NTSC system. If the locus of theaxis is maintained the Q-axis turns towards a line connecting points quitear to green and magenta. If, on the other hand, the Q-axis is maintained, the~xis turns to near the line red-cyan. It is, however, not necessary to choose

fE:~cili:.:e: ::::::::'~: :~::o:~n:_d::ri:~=:~rtb:::::;rstem was analyzed. Itwas found that this choice is governed by the statisticalroperties of the chrominance signals. It was concluded that the I-axis shouldbincide .with the axis of maximum "average signal excursion" (sec. 2.5.2).lhe measurements concerning these statistical properties of the chrominance

217

218 1. DAVJDSE

information were carried out with an mSC subcarrier signal: Owing to tasymmetrical nature of the NTSC subcarrier signal their results cannotapplied directly to the case of the proposed symmetrical signal. However,correction accounting for this known difference can easily be made. If thisdone, the measuring results can be used to select appropriate modulatiaxes for the proposed subcarrier signal. Since the measured maximum appearto be rather flat, this choice is not very critical. Working on these lines one fin .(hat it is appropriate to choose the axes such that the Q-axis is 30~from tB-Yaxis. The I-axis will then be located 28° from the R-Yaxis (fig. 3-1. From these data the composition of the corresponding 1- and Q-signals ceasily be calculated. The result is presented in the table below; for easecomparison the formulae for the NTSC system have been included 26).

Fig. 3-10. Choice of 1- and Q-axes for the symmetrical subcarrier signal.

,NTSC Modified signal

Ey' =0'59EG' +0'30ER' +O'IIEB'Er' =-0~i7(EB'-Ey')+0'74

(ER'-Ey')EQ' = 0'41(EB' -Ey')+0·48

(ER'__:_Ey')~Er' =-0'28EG' +0·60ER' -0'32EB'.È_Q'=-0'52EG' +0·21ER'+

0'31EB'

Composite signal:Ey',f-EQ'sin(w;t+33°)+ "Er'cos(wst+33°) if:phase 0° repre-sents the phase óf (EB'-~~') .

Ey' =0'59EG' +0:30ER' +0'11E1/EI* ' -:-0'25(EB',-Ey')+O'52

(ER'-Ey')EQ* ' : 0·42(E~'_:Ey')+O·28

(ÉR'-Eyi) , ,'EI*" -O'l~EG' +0'44ER' ....::.0·28E1EQ* ."':_0·42EG'+0·07ER' + .

0'35EB" .,'

Compo~ite signal:, . Ey' +EQ* sin(~si+3ÖO)+' '.EI* cos(wst+3ÓO) if phase 0° represents the phase of (EB' __:'Ey') ,

-- ----- ------------------------,


F.~.~;nstant-l;..i~a~c~ errors with the modified signaljAs we have seen earlier, the luminance signalof the NTSC system does notovide full constant-luminance operation due to the method óf gamma cor-~tion. Part of the luminance information is transmitted by the narrow-bandIrominarice signals. Fot saturated colours this portion can be considerable;fhe modified signal employs the same luminance signal as the NTSC system.Î a consequ~nce th~ portion of the luminance information that is carried by thebcarrier signal is the same for both systems. However, the broad-band and~ narrow-band components (1- and Q-signals) in the sub carrier signal areerent. This causes the luminance errors in transitions between saturatedours to be slightly different from those in the conventional NTSC system. ,.Assuming r = 2, we have found that for the standard NTSC system theproduced luminance obeys the equation

, L = Ey'2 + .12,

ere(2.4)

The loci of constant .12 in the I- Q plane are ellipses. The ratio of the lengthsthe major and minor axes is 1,27, the angle between the major axis and thexis being 17°.In, the same manner .12 can be expressed in the 1- and Q-signals of the sym-etrical subcarrier signal. One finds '. . ,

(,.12 = Q'94E[*2 + Q'93EQ*2 + Q'88E[*EQ* . _. (3.31)

e ratio of the lengths of the major and minor axes is 1,68, the angle between

t major 'axis and the I-axis being 45°. We see that the symmetrical signal isore elliptical than the NTSC signal. This is a slight disadvantage of the symmet-al signal. , .' , ", , .

lcompensation of luminance errors by forming and utilizing the envelope ofe subcarrier signal in the receiver will be less effective. As further effect theo.sstalk of the luminance signal into the subcarrier channel (cross-colour) willenhanced. According to (3.31), spurious signals in the I-channel and in the

[

-channel donate roughly equal luminance con.tributions. We have seen (sec.6.2.5) that in the standard NT~C system advantage is taken of ~he fact thatlie I-signal contributes less to theluminance than the Q-signal,does. This isivourable because the I-channel has the larger bandwidth so that the amplitu-:es of the spurious sign~ls inJhis channel are larger, As 'a consequence a veryrvourable compromise between the amount of useful information and the

r:~::~~l~~o~st~lk ~s~b~ained. _In:the. ~~etr~cal signal,t,~is ~ompromise is less

I Finally we note that the luminance contribution of the subcarrier signal i~ alack-and-white receiver, due to rectification of the subcarrier signal, dependsI ' '. .. .

220 0·"· .••.. ~;. "J. DAVIDSE • _o' '.~ • • ': ~J."

on the amplitude of thê sub carrier only, so the compensating èffect of t .rectification on the constant-luminance errors will be less the larger the elliticity of the subcarrier signal is.It must be noted that these disadvantages of the symmetrical signal are

small that in practice they will hardly give rise to visible effects. For the sakecompleteness they should, however, be mentioned. .

3.5. Utllizatlon of the vestigial-sideband components of the I-signal

3.5.1. Introduetion

In the preceding sections the principles of three-gun and one-gun decodinmethods have been analyzed. So far the analysis did not include refinemeniwhich, though within the capabilities of the transmission system, at the preseftime are never utilized in practical designs. We shall now deal with some (t?ese possible improvements of the decoding system.

The first point we shall pay attention to is the possibility of better utilizatioof the information in the vestigial-sideband of the I-signal.

It is true that taking advantage of the possibilities of improving the renderinof colour transitions provided by the extra bandwidth makes sense only if it iin balance with other aspects of the system which have a bearing on the qualitof transitions. In present-day receivers the rendering of colour transitions idetermined mainly by errors of the picture-tube assembly, but it can be expectethat in future designs these practical imperfections will be overcome.

3.5.2. Practical methods for utilizing the vestigial-sideband information

If the vestigial-sideband components are uti~zed at all the usual way to do sis to demodulate the sub carrier signal along the 1- and Q-axes. In that way th1- and Q-signals are obtained which are subsequently combined to colourdifference signals in a matrix network 29). However, with this technique thvestigial-sideband components are obtained with half the intended signal levelEqualization of the spectral amplitudes requires a "boosting filter" (fig. 3-11:In practice such a filter can not be made satisfactorily because the discontinuitin the gain-frequency characteristic cannot be obtained without serióuslaffecting the phase response.

t6r------"'1 j~1.5 MC/5 0.5 1.SMC/s

-freq. - freq •. Fig. 3-11. Theoretical characteristics of "boosting filter".

0.5

With self-decoding one-gun picture tubes the problem is even more difficultTheoretically the 1- and Q-signals may be formed here first by full demodula

TRANSMISSION AND DECODING IN COLOUR TELEVISION- "

221

'On,and subsequently remodulated in the proper way on the writing frequency.Is a rule this method is unattractive in practice. But the conventional techniquer translating the NTSC signal into a ces writing signal "directly does not ac-funt for the vestigial-sideband components of the I-sign~l. If the vestigial-

f:

'de,'",band is filtered out the extra infOrmatl,·on is lost; on the other hand, if it isnserved, then full quadrature crosstalk of the sideband components into the-signal will be introduced by the self-decoding of the picture tube. The onlyay to avoid this dilemna is to transfer the subcarrier signal into a"full double-deband signal. This means that the missing upper sideband has to be added toe writing signal.By heterodyning against a second harmonic carrier frequency it is possible torm a signal with solely an upper sideband from a signal having solely a lowerdèband, Hence, by combining the original signal and the heterodyned signal aouble-sideband signal is obtained. Figure 3-12 shows a method based on thisrinciple.

compo_sltesubcarrier si9oo1El=r.:st+Eó.sin c..s

Double sldebardcomponents af _suli:orrier signal

Input

.£Fig. 3-12. (a) Method of forming subcarrier signal with full sidebands.

(b) Frequency spectrum of input signal.(c) Frequency spectrum of output signal.

Though theoretically sound, this method has the disadvantage that spectralomponents in the region of 4·0 Mcts are on the slope of both filters. The

oiitplJtslfTdhaving MIsld.bands

Fig. 3-13. Practical method of forming double-sideband signal.

222 J.DAVIDSE

amplitude and phase ,characteris!ics .of both filters must therefore match ça9other very precisely. T.his comPlication. origin..ates from the separate proCe,SSij.of the vestigial-sideband component. Figure 3-13 shows a method which avoithe separation of this component from the _total signal by the employmenttwo mixer stages. The filter in. the upper branch passes the full input signJincluding the vestigial-sideband components, the filter in the lower branpasses only tile double-sideband components in the input signal (see fig. 3-12bThe different delay of both filters is equalized by the delay line.

3.5.3. Analysis of the two-mixer circuit

In order to analyse conveniently the operation of the circuit of fig. 3-13assume that the Q-signal consists of a single sine wave while the I-signal c01Sist.Sof two sine waves, one in the dou ble-sideband region and one in the Singl.sideband region. So we can write '

Er' = cos Il-It + cos Il-Ivt , ' .EQ' = COS'Il-Qt. (3.32

For the subcarrier signal wehave

Ei = (cos Mt + cos Il-Ivt) cos oost+ cos Il-Qtsin oost= t cos (ws + f.lI)t +t cos (Ws-M)t + t cos (Ws-Mv)t + t sin (ws + Il-Q)t+ t sin (ws-Il-Q)t,

(3.3Jif, for convenience, we omit the phase shifts of 33° in relation to the referen9phase in the NTSC signal. JThe wide-band filter passes this signal fully whereas the narrow-band filt~

rejects the vestigial-sideband component cos (ws-f.lIv)t. If k« denotes t~1direct gain of mixer I and ki its co.n.version gain, its .output signal can be Writte]omitting the terms at 3 Ws,

EMI = Es(k2 + 2klcos2wst) = t(k2+kI) cos (ws + f.lI)t + t(k2 + kl) cos(Ws-Il-I)t+ tk2COS(Ws_:_:/l-Iv)t+tklCOS(Ws + Mv)t + t(k2-k1) sin (ws + /J-Q)+ t(k2-kI) sin (ws-Il-Q)t.' . (3.341If k2 = kl, the sine terms cancel while the cosine terms describe a full dOUbl]sideband signal. . , " " , ' . I, .' :

If k, denotes the direct gain of mixe.r II and ka its conversion gain, the outpuof this mixer can be written ,

EMQ = Es(k4-2kacos2wst) = t(k4~ka)cos(Ws-Il-I)t + t(k4-ka)COs(ws +Il-I)t + t(kil + ka)sin(ws +'flQ)t -i- t(k4 + k3)sin(ws-Il-Q)t.' (3.35:

Now ifka = k4, the cosine terms cancel; thus, combining the signals EMI ancEMQ yields the original subcarrier signal to which the missing upper sidebancis added. However, the high-frequency components of the I-signal have stilhalf the intended signallevel. This is.because all I-components are produce? b)one and the s,amemixer, viz. ~i~er~. _

•• a'

_ • _ TRANSMI~SION AND DE<?ODING IN COLOUR TELEVISION 223IWe notethat mixer IT does riot affect the vestigial-sideband components in.Dyway~ So if mixer II is made to' contribute not only low-frequency Q-compo-Je~ts, as is the case if ka = ~4;but also low-frequency I-components, the ampli-oude relationships'between the high-frequency and the low-frequency compo-\ents in the I-signal may be changed. From (3.34) and (3.35) we see that with

11= k2 = =k« and ka = 0 a double-sideband signal with flat spectrum isbtained, A further possibility requiring no negative value of one of the gain 'actors is ka = k: = k2 and k4 = o. The phasor diagrams in fig. 3-14 illustratehis case. Itmay be noted that the direct gain of mixer n has to be zero, whichequires afully balanced mixer. However, this disadvantage can easily be avoided

Fig. 3-14. Phasor 'diagrams of the operation of the mixers for the case k, = k« = k3-, and k, = o.

(a) signal delivered by mixer I.,(b) signal delivered by mixer 1I.(c) addition of (a) and (b).

224 J. DAVIDSE

because it is not necessary to maintain the ratio of the signallevels of the 1- aQ-components of the NTSC subcarrier signal. For instance, if 2ki = 2k2 == 2k4, we get the output signal shown in fig. 3-15. Again a full double-sidebansignal is obtained. The amplitude of the Q-signal is larger than in the origin

1J.lQ.Fig. 3-15. Phasor diagram of output signal if kt = ka = k, = -!k3'

signal but for the further signal processing of the full double-sideband signathis is no disadvantage at all.

3.5.4. Application to one-gun display systems

If we have to do with a one-gun tube the double-sideband signal delivered bthe two-mixer circuit can be fed to an elliptical amplifier in order to convert ijinfo a ces writing signal. By the self-decoding action of the picture tube thiwriting signal is then decoded without quadrature crosstalk being introducedwhile full advantage is taken of the high-frequency components of the I-signalFigure 3-16 shows a block diagram of the essential elements of a ces decodeworking along the lines described above. This figure reveals a remarkable peculiarity ofthis decoding method: both mixers I and ITand the elliptical amplifie

£,'y

Inputcomposite N7SC

composIte cesItf'itlng signal

output

Fig. 3-16. Block diagram of ces decoder utilizing the high-frequency components of theI-signal. '

TRANSMISSION AND DECODING IN COLOUR TELEVISION 275I . ....... -_.---...--.---....~ ." _.-- ... .re fed by auxiliary signals at the second harmonic of the subcarrier frequency;le it at different phase angles. This suggests the possibility that these two oper~-~onsmay be combined in some wayor other, _th:u~substantially simplifyingi1eoverall signal-processing chain. We shall now analyze whether such a sim-i lification is possible and under what conditions.IFirst of all we note that essentiallythe function of mixer I is to produce the.ussing upper sideband of the !-signal. This requirement fixes the phase of its~uxiliarysignal at 2fs. Changing this phase wou1d introduce an unwanteduadrature upper-sideband component. Mixer II operates on a full double-ideband 'signal. It can therefore' not introduce any quadrature crosstalk,respective of the phase of its second-harmonic auxiliary signal. It is clear thate functions of mixer I and ofthe elliptical-gain amplifier cannot be combined.f any combination of functions is possible, it must consist of incorporatinghe elliptical amplification into mixer 11.We note that in fact mixer 11can beoked upon as a special elliptical amplifier, hence what is needed is a change ine parameters of the elliptical amplification. To this purpose the amplitudend phase.of the second-harmonic auxiliary signal have to be changed. Bearing's insight in mind we can proceed to a mathematical analysis of the problem.As the numerical formu1ation of the NTSC signal will be of interest in thenalysis it is convenient to reintroduce here the phase angle of 33° with respecto the reference phase, occurring in the NTSC signal specifications. Of course,he choice of the reference phase is unessential in our problem but it may be andvantage that numerical results are obtained which are directly applicable toractical design considerations.The input signal to the circuit is, , EI' cos (rost + 33°) + EQ' sin (rost+ 33°).

mitting the terms. at 3ros, which are removed by, the,output filter, the outputignal of mixer I can be written .[EI' cos (rost + 33°) + EQ' sin (rost + 33°)] [k2 + 2kl cos 2( rost + 33°)] =klEI' cos (rost + 33°) = 2kl [(j·15(EB'- Ey') sin rost- 0'23(EB' - Ey')os rost-0-40(ER' - Ey') sin rost + 0-62(ERi - Ey') cos rost],' (3.36)

ince le: = k2, as was derived earlier.Expression (3.36) is valid for the double-sideband components in the input

ignal: For the high-frequency (originally single-sideband) components theoutput is half this value.If cp denotes the phase angle of the second-harmonic signal fed to mixer 11,

the output of this mixer can be written .[0'49(EB'-Ey') sin rost+ O'88(ER'-Ey') cos rost] [k4 + 2kacos (2rost + cp)]= k« [0'49(EB'-Ey') sin rost + 0-88(ER'-Ey') cos rost] + ka [-0-49(EB'-Ey') sin (rost+ cp) + 0-88(ER'-EB') cos (rost + cp)]. (3.37)

J. DAVIDSB "226

-This output signal contains no high-frequency components thanks to the ha'pass filter at its input. Adding the, signals described by (3.36) and (3.37)obtain the complete output signalof the circuit so far, as the low-frequen'components are concerned. .

E1) = 0·49k4(EB'~Ey') sin C:O~t+0:88k4(ER'_:_Ei') 'cos wst.' ":"'0'49ka(EB' -Ey') (sin wst cbs f{!.+ cos 'wst"sin rp),+0'88k~(ER'-Ey') (cos wst cos rp~sin wst sin rp)... . .. .,' ~-0'45k!(EB'-:-Ey') cos wst + 0'29k!(EB"-Ey') sin wst+ l'24k!(ER' -Ey') CÓS wst-;-0'80k~(ER' - Ey') sin :nst. (3.3. . , '

~.According to equation (3.20) and fig. 3-5 the symmetrical CCS subcarri'signal can be written

0·66 [ER' cos wst + EB' cos (wst-1200) + ÈG' cos '(wst-2400)] = '0'89(ER'-Ey') cos (wst-19°) + 0'74(EB' -:-,Ey') sin (wst-21°).

We note that this formulation assumes that phase angle zero corresponds wired. The formulation can be made more general by introducing an arbitraphase angle ~ and by multiplying the expression by an amplitude-scalirig factks, Doing so we obtain as a generalized expression for the CCS subcarrisignal

Ez = k» [0:89(ER'-Ey') cos (wst-~) + 0'74(EB'-Ey') sin (wst-!5-2°)]

or,

" .

Ez = k5 [0'89(ER' - Ey') cos wst cos ~ + Q'89(ER' - Ey') sin wst sin ~ +0'74(EB' - Ey') sin wst cos (~ + 2°)-0'74(EB' ~Ey') cos wst sin (!5 + 2°)].

o (3.3

o •• ICorrect signal translation requires E1) and Ez to be identical: Comparinequivalent coefficients of (3.38) and (3.39) we Obiai~ the following e~uatio,

Q'49k4-Q'49ka cos rp+ 0'29k! 0 ~'74k5 cos (~,+ 2°), (3.401Q'88k4 + 0·88ka cos rp+ l'24k! = 0'89k5 cos ~ , (3.41

, -0'49ka sin f{J -0·45k! = Q'74k5 sin (~ + 2°)' , (3.42-0'88ka sin f{J -Q'80k! = Q'89k5 sin ~ . (3.431

I~ th~se' ~~~ equations six q~~~tities occur, viz. k!, ka, k4, k5, s and rp.~Ithese, k5 occurs in the same manner in all right-hand terms, it can thus be' co~1sidered to be a gain factor that is unessential toour problem. Without loss ogenerality it °c~~be put equal to 1. Or'the remaining parameters obviously o~can be chosen at will, the other four parameters are then fixed by the equation(3.40) to (3:~3J. However, the proper solution ~o our signal-translation pr?bleforces us to choose two parameters. To s~e this we note that the output sIgnal)of mixers Iand 11have to be added and hence have to be matched to each othei

I

TRANSMISSION AND DECODING IN COLOUR TELEVISION . 227

•n the correct way. The eperation of mixer I is entirely determined by the fact.hat is has to process the vestigial-sideband components of the input signal.r=components are not subjected to elliptical amplification. In consequence,hey remain components of a NTSC-type subcarrier signal, not of a CCS

tlUb.carrier signallike the low-frequency components, due to their simultaneousrocessing in both mixers. Figure 3-17 illustrates the situation. .Exact matching of both contributions of the output signal requires that theomponents of the ces signal along the I-axis obey the same equation as theTSC I-signal.

r: ,i. "} ,,~JDirection otk'CG-27,£,-£; +O'l.s{£;-EJ, E' -E'

= k,El 1/ R Y

ig. 3-17. (a) Phasor representation of high-frequency components delivered by mixer I.(b) Phasor representation of low-frequency components delivered by the combination of

mixers I and 11.

Mathematically (see fig. 3-17)

0.89(ER'~Ey') cos (330 + c5)-0'74(EB'-Ey') sin (350 + 15)=kl [-0·27 (EB'-Ey') + 0'74(ER'-Ey')] (3.44)

or,

0·89 cos (330 + 15)= 0·74 kland 0·74 sin (350 + 8) = 0·27 k: .

(3.45)(3.46)

Solution ,of these equations yields 8 = _110 and k i = H2. Inspection ofeqs (3.42) and (3.43) shows that these equations can only be obeyed if 8 =-1015'. This means that exact matching is not possible; the vestigial-sidebandcomponents are about 100 out of phase with the double-sideband components.However, this slight phase shift leading to a very slight hue shift in colour tran-sitions cannot be called a serious drawback. Itcould be avoided by introducingone extra degree of freedom. An obvious method of obtaining this would be toshift the entire input subcarrier signal to mixer ITin phase. However, the effort.would not be worth the slight advantage gained in this way.

Assuming 0 = -1015' we can solve ka, k« and cp from eqs (3.40) to (3.42).

{228 J. DAVIDSE

We find then

k4 = .1'26-1-00 k1

0·45 k1tan rp = ------

0·12 + 0·20 k1 '

- 0·23 k1 1k3-----

0·25 sin tp

Putting k: = 1·12 we find k« = 0'14, k3= 1·21 and rp= 238°.Figure 3-18 shows schematically the subcarrier translator according to t

results of our calculations.

Fig. 3-18 Schematic representation of the subcarrier signal translator.

3.5.5. Incorporation of Y-to-M conversion in the two-mixer circuitAn obvious further possibility is to incorporate the required Y-to-M conver

sion in the translator stage of fig. 3-18 also. We have seen that the requirecorrection signal EM' - Ey' can be produced by synchronous detection of th

lsubcarrier signal. The required detection phase is 19° and the required conversion gain for this conversion process is 0'58,30).Figure 3-19 shows how the circuit can be modified to this purpose. The syn-

chronous detection of the input signal by the signal at Ws at the second grid 0

mixer I yields the desired signal EM' - Ev', But the second grid signal at Ws isalso available in the output. As the common impedance Z2 has to be tuned tothe subcarrier frequency this unwanted component cannot be removed byfiltering. To cancel this unwanted component a signal at Ws of opposite polarityis fed to the second grid of mixer II. The impedance Z 1, which is supposed tohave negligible response at subcarrier frequency but full response at the fre-quencies composing the EM' -e« signal (0 - 1·3 Mc/s), is inserted into theanode lead of mixer I only, so that the wanted EM' - Ey' signal is not cancelledbya signalof opposite polarity produced by mixer II. The required value ofthe


MixerIfdirect gain-O'14conv.galn for ellipt.ampl. =1'21conv.gain for YioM cmv =0'58

ig. 3-19. Circuit providing both double-sideband elliptical amplification and Y-to-M .con-version.

onversion gain can be obtained by choosing suitable values of the amplitudef the reference signal and of the impedance ZI.If ZI is inserted in the anode lead of mixer I, as indicated in the figure, the

ignal EM' - Er' contains quadrature crosstalk due to the vestigial-sidebandomponents in the input signal. If, alternatively, Zl is inserted in the anodead of mixer H, the EM' - Ev' signal is free of quadrature crosstalk, but lackshe vestigial-sideband components. Small errors in the EM' - Ey' signal, whichtself is usually small, have negligible influence on picture quality, so the placef ZI is rather immaterial. Itmay be noted that the decoding axis for the signalM' - Ey' is quite near the Q-axis, so the contribution of the I-signal to theM'-Ey' signal is small. It might therefore be preferred to take this signalrom mixer II rather than from mixer I. 'A further simplification of the overall processing circuitry would be obtained

f the circuit could at the same time be employed for the amplification of theuminance signal Ey', so that the circuit could be driven by the compositeTSC signal. With the signal-translator circuit of fig. 3-7 it appeared that thiseature could be incorporated simply. However, with the present circuit this

Suppressionat 2ts and 3ts

:::?:..:.::..o.;.=:=;-'f---+~-"""--'---:------+M""'ixe~r-If........j1lg~~~it~CCSsignai

-sn(Wst+19o)+ms(2uJ"t+238")

Ei cos("'stiJ3")+ EQ sin (wst+33")

Fig. 3-20. Circuit providing full translation of composite NTSC to composite CCS signal.

230 J.DAVIDSE

is not so easy. The difficulty here is to prevent the production of unwantintermodulation products between the luminance signal and the referensignal at subcarrier frequency: Compensation of these unwanted signal coponents is of course possible by the addition of a third mixer, for instanceindicated in fig. 3-20. Here the extra tube is driven at its first grid by the Iunance signal only, while the signal at its. second grid is the opposite of the coresponding signalof mixer r. Though such a circuit might work properly, .practical value is doubtful since separate amplification of the luminance signismore obvious and hardly more complicated.

3.6. Compensation of luminance errors3.6.1. Introduetion

This section aims to introduce a second topic in the field of receiver techniquto which until now little attention has been paid in the technicalliterature.concerns the possibilities of compensating for luminance errors caused Dimperfect constant-luminance operation.We have seen that the NTSC system, though basically a constant-luminan

system, does not provide full constant-luminance operation. The aberrationcaused by the method by-which the signal is gamma-corrected. In consequencpart of the luminance information is. transmitted by the narrow-band colachannel. For the case y = 2 we found in sec. 2.3.2 for the luminance contribtion of the subcarrier signal

Replacing by. approximation the elliptical subcarrier signal by a circulsubcarrier signal we found in sec. 2.8.2.2 that (3.47) can be simplified to

(3.48

where Ac is the amplitude of the subcarrier signal. Hence we have by approximation for the total reproduced luminance

(3.49The portion 0·52Aè2 is transmitted through the narrow-band colour channel

lts lack of high-frequency components manifests-itself in transitions betwee~saturated colours. The luminance errors caused are commonly called "Iumilnance notches".

3.6.2. Analysis of the luminance errors

For the analysis ofthe luminance errors it is useful to represent (3.48) graphi-cally, For this purpose we draw (fig. 3-21) lines of equal Ll2 using EQ' and Er' asthe quantities plotted along the x- and y-axis (so-called I-Q planej.These linesare of course circles.thanks to our approximation of circularity of the subcarrier


g. 3-21. Different types of transitions between saturated colours. The circles represent lociof constant subcarrier amplitude.

gnal. If we assume for simplification of the problem that the bandwidths ofe J- and Q-chann!,:lsare equal and that no ringing occurs in these channels,e transitions between complementary colours will follow straight lines. Inactice neither condition is fulfilled, which means that as a rule the transitionill not follow a straight line but a more complicated curve. In our furthernalysis we shall assume equal bandwidths for the chrominance channels. .We can distinguish three essentially different types of transition:) From a certain colour to its complementary colour. or to a colour near thismplementary colour (A-B in fig. 3-21).) From a certain colour to white (C-D in fig. 3-21).) From white to a certain colour (D-E in fig. 3-21).igure 3-21 shows also a fourth type of transient (H-G), from a certain colouranother nearby; we can see that in this case not many circles are intersected

r, in other words, that the luminance contribution of the colour signal changesnly slightly during the transition. In principle the process may be comparedthe transition from A to B, but the effectsare much smaller.The time required for each transition equals the rise time of the chrominanceannels, .chr.With the aid of fig. 3-21 and eq. (3.49) it is easy to see which luminance

rrors Le can occur during the transitions drawn. Figures 3-22a to 3-22d showèese luminance errors. In practice the case of fig. 3-22a is the most trouble-ome. This is because the errors shown in figs 3-22b and 3-22c have a smalleraximum (0·39 Ac2 against 0·52 Ac2) while the most important part of therror takes up a shorter time (t .chr against .chr).The ensuing considerations will only cover the cases represented: by figs-22a, band c, because these are the most serious ones. The other cases may, if.ecessary, be dealt with in an analogous manner.Whether a luminance error is present or not does not follow from the value of

232 1. DAVIDSE

@ ® ©

tLe0·52 -t

. tLeA2c

I :'position of trueI",position of true luminance transition

luminance transition(a) (b)

1:dJ@ ® ® ®

tLe -ttLe

_t

',position of trueI,position of true I luminance transition

luminance transi tion

(e) (d)

Fig. 3-22. Luminance errors in transitions between saturated colours.

A c since in large areas luminance is always correct. Errors can only occuduring transitions. During a transition involving a luminance error, Ac wilshow a "notch" (compare fig. 3-22). Hence, in any case that a transition occurdAc/dt differs from zero. We can therefore say that the value of dAc/dt can bused as a quantity to reveal whether a transition occurs. It does not disclose tlnmagnitude of the luminance error during the transition which, as we have seenis determined by the value of Ac. It is a serious drawback that the correcvalue of Ac is not known the moment we need it, due to the occurrence of tlu"notch". As it is, we have to estimate the value with the aid of the knowrvalues, that is, from preceding and following moments.It will be clear that if all transients in the chrominance signals were steepenec

so that they had the rise time of the transients in the luminance signal,' the luminanee errors would be satisfactorily compensated. As is well known, suclsteepening of signal transients which are unsteep due to bandwidth limitintcan be obtained by a technique called "signal crispening" 31). In accordanewith the considerations put forward above the chrominance signals have to bidifferentiated in order to derive a criterion for the occurrence of a transient irthe signals. Furthermore, a suitable non-linear operation governed by thedifferentiated signal has to be performed on the chrominance signals. In thaiway all transients in the signal can be given a uniform short rise time. Of course.in this way overcompensation can occur: transitions which are not intended tebe steep will nevertheless be rendered as steep transitions since no discrimination is available between transitions that have lost their steepness by bandwidthrednetion and those which are unsteep in themselves. Unhappily enough this

TRANSMISSION AND DECODING IN COLOUR TELEVISION 233.

lhnique requires rather complicated extra circuitry. Moreover it can only be'~ployed if the fully demodulated chrominance signals are available in theIceiver. In general such is only the case in three-gun decoding systems, It iserefore desirable to look for methods which require less complicated circuitryan full cri spening does and which act more directly. If such methods turnIt to be simple enough they might be applied even if only partial compensa-n could be obtained with them.Differentiation of the subcarrier signal is not the only way to obtain infor-ation whether a transition occurs at a certain time or not. As a rule a transi-n in chromaticity will be accompanied by a luminance transition. Hence theesence of high-frequency components (mixed highs) in the luminance signaln also be employed as a criterion for the occurrence of a colour transition.

6.3. Possible practical methods of luminance correction

6.3.1. Addition of a correction signalto the luminance signal

We shall now briefly survey possible methods of luminance correction. So fartheoretical considerations are needed we shall basé them on the approxima-ns assumed in the preceding paragraph.To substantiate the correction it will be necessary to form a correction signal.s the intended correction concerns the luminance of the reproduetion it isvious to apply the correction signal to the luminance channel. Let us assumeat a correction signal S is added to the luminance signal, then the renderedminanee can be expressed by

L = (Ey' + S)2 + kAcz ,

here k is about 0'52, according to (3·48). Hence,

L = Ey'2 + 2Ey'S + S2 + kAc2 • (3.50)

During the transients the term kAc2 is too small or even becomes zero (see. 3-22) and it must be corrected by the terms S2 and 2Ey' S; in other words,e luminance notch must just be filled by the contribution from these terms.he moment the luminance notch is deepest (:fig. 3-22a), the de:ficit equals kAc2•or this reason S should be so chosen that at this moment

(3.51)

his shows that S has to depend on Ey' as well as on Ac. Similar considera-ons apply to any value of .dAc (the aberration of the proper value of Ac arisingom the finite rise time ofthe chrominance channel).It is theoretically possible to produce a signal S that is properly governed byoth Br' and Ac, but the practical solution is complicated, let alone the essen-al difficulty that Ac is not well known at the required moment. Moreover.51) tells us only how large S should be at its maximum; nothing definite is

234 J. DAVIDSE

known about the shape of the required signal. Furthermore we have to masure that the correcting signal is only effective during transitions. As we haseen this requires the formation of a signal representing dAc/d!, whichmust woon the correction signal. This suggests deriving the correction signal.not froAc but directly from dAc/dt. Figure 3-23 shows the waveforms representidAc/dt in the idealized cases of fig. 3-22a, band c.Comparison of figs 3-23a and 3-22a shows that the simple signal represen

ing dAc/dt cannot serve, even approximately as a correction signal. The ter2 Ey' S, which isUSUallY.the principal component of the correcting signal alwa Ihas the same sign as S. The type of luminance notch shown in fig. 3-22a mube compensated for by an ever-positive luminance contribution. We migltherefore make the dAc/dt signal positive by rectification. However, by dOi~so, we encounter erroneous signs in the cases represented by figs 3-23b an3-23c, which can be seen in the associated figs 3-22b and 3-22c where, apafrom the negative portions, positive ones a.re present. No easy measure.s can ~taken to avoid this situation, but the errors in the cases of figs 3-22b and c a~1not very troublesome, so that in practice this simple correction method migwell be useful.

-t

I.-\Q_-t

I·

J?-o --t

. representation of

Ac

-t

(a) (b) (c)

Fig. 3-23. Representation of Ac and dAc/dt for the cases of fig. 3-22a, b and c.

. We notice that the straight portions of fig. 3-23 will have to be rounded o~slightly if the more flowing lines of fig. 3-22 are to be correctly imitated. Ir

. Ipractice we shall get more or less flowing lines as a result of the bandwidtlI

limitations of the circuits involved in producing the correcting signal. .The compensation method developed so far requires in the receiver three

extra circuits for deriving the compensating signal. First, envelope detection o:the sub carrier signal is needed in order to produce a signal representing AcThis signal has to be differentiated and subsequently rectified; then it is addecto the luminance signal. It is true that these operations do not require man)extra components. On the other hand it must not be forgotten that only partial


Epensation can be obtained in this way and that the luminance errors under.cussion, though usually noticeable, are rarely annoying.I'he compensation method indicated here must be looked upon as suggesting} direction experimental investigations in this field might take. Many possible{iations based on the general outline will occur to the reader. An obviouse is the employment of a signal representing Ac2 instead of Ac. When dif-lentiated this gives 2Ac dAc(dt, which also differs from zero only when dAc/dt~'ers from zero, i.e., during a transition. A signal representing Ac2 can Deily obtained by squaring the sub carrier signal.

.3.2. Luminance' correction by controlled amplification of the "mixed highs"

e have already seen that the presence of "mixed highs" in the luminancenal can be taken as a criterion for the occurrence of a transition.Of the luminance information transferred in the frequency range below thetoff-frequencies of the 1- and Q-channels nothing is lost, though there are twoys by which the information reaches the reproducing device, viz. the lumi-nee channel and the chrominance channel. Luminance information conveyedthe frequency band lying above the cutoff-frequencies of the chrominancearmels gets lost as far as the portion that is suppressed in the narrow-bandlour channel is concerned. But the portion fed through the luminance chan-1will still reach its destination. Now, it must be possible to arrange the receiv-in such a manner that any losses of luminance information in the colourannel are compensated by an additional amplification of the portion passedthe luminance channel. This additional amplification must be higher as the

sses are higher. Again, the amplitude of the subcarrier is a measure of thesesses. We shall now first try to find the relation between the amplification ofe mixed highs in the luminance channel and the amplitude of the subcarrier.e totalluminance is given by Ey'2 + kAc2. For the mixed highs only thertion Ey'~ is present. The additional amplification G of the mixed highsould therefore satisfy the equation

ence

(3.52)

It cannot be said that the generation of a signal obeying this equation is easy.o simplify the matter we might overlook the dependence on Er' and use a, lue of G belonging to a conveniently chosen "average" value of Ey'. Let thisalue be EYm', then

[ ( Ac)2]!G= 1+ k EYm" . (3.53)

235

236. J. DAVIDSE

ir, in addition, k ( Ac ,)2 ~ 1, then G will be approximately equal toEYm

1+~(~)2.2 EYm'

The indicated condition will, however, not always be fulfilled.WhenEy'islargerthanEYm', Gwillbetoolarge, giving rise to overcomp~

sation, and vice versa. A further problem is presented by the fact that we ne.Ac at the moment its correct value is not available: as we have seen, Ac. shoa notch during a transition. We can eliminate this notch by applying the sigrepresenting Ac2 to a "crispening circuit" 31). However, such an additioJcircuit would make the overall circuitry rather complicated. A simpler approimation is to use the value of Ac found before or after the transition, whimay be achieved by introducing a suitable delay into the circuit.Figure 3-24 shows the block diagrams of the video section of a receiver e

ploying luminance correction in the suggested way.

(3.

to synchroMIIJsdetedors

Fig. 3-24. Luminance correction by controlled amplification of "mixed highs".

3.6.3.3. Luminance correction employing the existing crosstalk of the subcarriinto the luminance channel .

Hitherto we have based our considerations on the assumption that only t~luminance signal adds to the luminance contribution of the.signal in the lumnance channel. Actually there will be a fraction of the subcarrier signal le.in theluminance channel, since the suppression filter in the luminance channdoes not fully suppress the sidebands of the subcarrier signal. This fraction ~rectified by the picture tube so that an extra luminance contribution OCCUl1

during transitions.In sec. 2.12 we have found that the luminance contribution of a subcarrie

signal with amplitude Ac in the luminance channel is t Ac2, for the case y = 2Hence, the totalluminance of the reproduetion will be given by , . I

L = -Ey'2 + kAc2 .-t t Ac2 • (3.55


Let USnow examine the type of waveforms to be expected for the remainder

titheSUbcarn.·erSigna.Ia.:tthe outp~t of the suppression filte.r.We assume that.frequency response IS symmetrical around the 'subcarner frequency (fig.

(5). The subcarrier signal can be written Aceos(wst + 1p); Ac and 1p are bothictions of time. However, in the transitions which are of special interest toEr' IEQ' is constant so that during the transition 1p is either constant orws a 180°-jump. 'If 1p is constant there remains amplitude modulation of the

Va 1Vil

8III11I1111I

fs -f

Fig. 3-25. Assumed gain-frequency characteristic of suppression filter.

bcarrier by Ac only. The effect of a band-suppression filter with bandwidth Ban amplitude-modulated signal is the same as the effect of a high-pass filterth cutoff-frequency tB on the modulating signal before it is modulated on thebcarrier, provided the group-delay characteristics are the same for bothstems. We can employ this rule to good advantage when determining thetput signalof the suppression filter. In the case that the transition passes thehite point so that a 180°-phase jump is present w~ can not employ this rulerectly. However, it is easy to see how we can deal with such transitions. Thesponse of the suppression filter to a transient signal in which a 180°-phasemp is present is equal to its response to the same transient signal to which abcarrier signalof constant amplitude and constant phase is added as theter is supposed to have infinite attenuation at the exact subcarrier frequency.the phase of the input signal is 0 before the transient and ()+ :Tt after theansient and we add a constant subcarrier signalof phase () and sufficientplitude we obtain a transient signalof constant phase yielding the sametput signal as the original signal did.Figure 3-26 illustrates the addition of the auxiliary subcarrier signal. ltsplitude is chosen to be equal to A c so that after the transient the resulting

gnal has zero amplitude.At this stage it is not difficult to construct the shape of the subcarrier signalfter it has passed the suppression filter. We simply take the envelope of thebcarrier or of the "adapted" subcarrier signal described above, construct

238 J.DAVIDSE

phase angle+9

phase angle : phase angle+9 : 9+180° becomes

~--- ---------~--: I

Fig. 3-26. Addition of auxiliary constant-phase subcarrier signal.

2Ac

phase angle +9

then the response of a low-pass filter to this envelope signal, and subtract toutput signalof this filter from the original signal, thus finding the outpsignalof the complementary high-pass filter. To simplify the matter we assthat the low-pass filter does not produce ringing and only magnifies the ritimes which, for our purpose, is a satisfactory first-order approximation. Tenables us to construct the envelope of the resulting output signals by meanssimple drawings: Figure 3-27 presents the results for the transients depictin figs 3-22a, band c. In the. case of fig. 3-22a first the transformation of fi3-26 is applied. Fig. 3-27a shows that

a = 't'lp-Tchr 2 Ac ,TIp

(3.5

original signal

position of the trueluminance transitionoriginal signalresponse of low-pass filler tothe original signalo

--t ~ response of high-pass filter to

(a)

o. -A

o

(b) {e}

Fig. 3-27. Construction of envelope of output signalof the suppression filter in response te. input signal transients according to fig. 3-22; I

TIp is the rise time of the low-pass filter complementary to the hypothetical high-passfilter;Tchr is the rise time of the chrominance signals (equi-band case),


!tilefrom figs ;-27b and 3-27c can be derived that

I TIP.-TehrAa = c •TIp

If IJ denotes the bandwidth of the suppression filter (fig. 3-25) by approxima-,n the relation

(3.57)

, 0·8. lidTIp = -- IS va .

B

e maximum contribution to the luminance is, according to eq. (3.55), equalt a2 or.t a/2• '

Assuming Tehr = 0·4 !Lsec(bandwidth of the chrominance channels aboutcis) and assuming B = 1Mc/s, we find with the aid of (3·56) and (3'57)

ta2 = tAc2 and ta/2 = t Ac2 .

om this we see that, when a normal suppression filter is used, the compensa-

tg contribution is of the correct order of magnitude. ,The waveforms depicted in fig. 3-27are the envelopes of the filtered subcar-r signal. Hence the output signals for the three cases are as shown in fig. 3-28,

1I, .position of trueluminance transition

(a)

position of trueluminance transition

position of trueluminance transition

(b) (c)

Fig. 3-28. Practical output signals of suppression filter.

here the shapes have been rounded off slightly. We see that around the tran-ent we have always a "hump" with a peak value a or a', and a well-defined{ail". As the bandwidth of the suppression :filter becomes narrower, a be-omes larger and the tai1longer. .I To prove that these approximating theoretical considerations are accurate

~

ough for practical uses the d.etailsof the filtered,subcarrier signalswere stud-d with a line-selector oscilloscope. It was found that the actual outputgnals are remarkably similar to the "constructed signals". By' employing appression :filterwith variable bandwidth (see sec. 2.6.1.2) the influence of

te suppression bandwidth could be investigated, which turned out to be inood agreement with the expectation.When we compare the luminance contributions of fig. 3-28 with the lumi-ance errors offig. 3-22, we cansee that, whereas in fig. ~-22 the true transition

240 J.DAVIDSE

is in the centre, the main part of the correcting signal in fig. 3-28 is shifto the right. For this reason the compensation is generally rather poor.situation wou1d improve considerably if we could shift the phenomenonfig. 3-28 forward in time. Figure 3-29 shows once more the luminance errorsour three types of transitions. However, the luminance contribution 'ofcrosstalking subcarrier signal is now shown shifted forward in time overt-r

ILuminanceerror causedby lack ofconstantlumifJlJnce

XILuminanceerror causedby residualaossfalk(shifted forward

in time)

IIIRemalningoverallluminanceerror

,

~I~----~rI

tLe . . ...

--~--~----'--~~ W ~ .

Fig. 3-29. Residuallurninance errors after compensation by rectification of residual subcarrsignal.

. ,The sketches show that the annoying pronounced "luminance notch" disapears and the overall error is distributed more evenly. The forward shift of tcrosstalk contribution is especially effective in case a, which is by far the metroublesome in practice.In order to shift forward the effect cáused by the residual crosstalk of ti

subcarrier signal into the luminance channel, it is apparently necessary to haall chrominance transitions in the composite NTSC signal preceding the assciated transitions in the luminance signal.An obvious method of doing this is .give the luminance signal the required additional delay in the encoder beforeis combined with the chrominance signal, but this method has two dra'backs. First the crosstalk effects of the luminance signal in the chrominanchannel (see sec. 2.6.2) and effects due to differential phase arid differentigain (sec. 2.10) no longer coincide with the transitions. Secondly, in thee omptible black-and-white picture the transitions in the dot pattern fail to ooinekwith the luminance transition. The first disadvantage cannot be called troiblesome, as was shown during experiments. The second effect turned out to 1rather troublesome, at least with the monochrome receivers used in our expeiments. It shou1d be noted, however, that the seriousness of this drawbacdepends largelyon the type of receiver used, more specificallyon its response 1

the highest video frequencies and on its group-delay characteristics; the effe.


ay' even become favourable in some types of receivers where group-delay-rors are present which counteract our delay shift. ." ...IThe experimental investigation of the described method is very simple. Oneily has to increase the delay of the Y-signal in the encoder by the requiredInount and to decrease the delay in the decoder correspondingly in order tolaintain the correct overall delay. The experiments showed that the desired

inance compensation is clearly perceptible.Should we decide not to change anything at the transmitting end, we coulden try to obtain the required situation as well as possible at the receiving end.is could be done by shortening the delay for the highest frequenciesthe video amplifier or in the i.f. amplifier. It is true that the same delay isen also given to the components with the highest frequencies in the luminancenal, but most probably this effect will not be very troublesome in practice.he signal components affected by this procedure are in the high-frequencygion whereas the errors to be compensated are of a low-frequency nature andill be far more troublesome.Itmay be noted here that the luminance contribution, due to rectification ofe remainder of the subcarrier signal will always be present when the subcar-r is not fully suppressed in the luminance channel of the receiver. This factould be borne in mind also when other methods of compensation are applied.The main disadvantage of the method discussed so far is that the compen-ting signal has a "tail", as was shown previously. This is because we have notfferentiated the signal representing Ac but have passed it effectively through agh-pass filter instead. It may therefore be expected that the employment ofe differentiation could obviate this drawback. The required shift in the delaystribution could then be smaller or even zero (see below). A similar resultight be obtained by the employment of a more complicated suppressionter behaving more like an ideal differentiator.The results of our investigation justify the conclusion that compensation ofnstant-luminance errors by judicious utilization of residual subcarrier cross-Ik is certainly promising .•

6.3.4. Luminance correction by the addition of a differentiated subcarriersignal

In: the preceding section we discussed a method of luminance correctionploying the residual subcarrier signal as a correcting signal. The correctionaccomplished by the rectification of the subcarrier signal in the picture tube,hich explains that no separate subcarrier envelope detector is required: Thisethod suggests the alternative possibility of employing not the residual sub-rrier in the luminance channel but a signal derived directly from the subcarriergnal proper without preceding envelope detection of this signal': It may be.' pected that in this way the employment of true differentiation will be more

I

242 J. DAVIDSE

. easily effected than by the formermethod. We have seen earlier that some kindifferentiation is at any rate necessary since the correction signal shouldeffective during transitions only.Let the subcarrier signal be written

Es = Ac cos (oost + 1J!).

Differentiating (3.58a) we find

dEs dAc (d1J!) .- = -. cos (oost + 1J!)-Ac .OOs+ - sm (oost + 1J!)dt dt dt .

(3.5

(3.5

and

d2Es _ [d2Ac _ (oos + d1J!)2Ac lcos (rust + 1J!)

dt2 dt2 dt.

- .[d21J!Ac + 2 (rus + d1J!)dAc] sin (rust + 1J!). (3.5

dt2 dt dtAccording to (3.50) the luminance contribution of a correction signal Sin

luminance channel isS2 +2SEy'. (3.

From (3.58) we see that, if the correction signal is a linear combination of I

and its derivatives, then 2 SEy' contains only terms describing modulasignals at subcarrier frequency, i.e. terms whose visible effect is negligiblevirtue of the dot-interlace principle. Hence, the only term which can giveminanee correction is the term S2. Now we want S2 to be composed in suemanner that a luminance contribution occurs during transitions only. Trequires that S should contain only terms which contain derivatives of Ac anas a factor. From (3.58) we see that the simplest signal complying with thisquirement is of the form

S = 'YJ [Es + _1 d2Es].rus2 dt2

The signal Es is already available in the luminance channel since this chatransmits .in principle the composite colour-television signal. That Es in tluminance channel is employed as a correction signal means that no subcarrisuppression filter is needed in this channel. The second component in the creetion signal must be formed by processing of the subcarrier signal.A simple calculation may give an impression of the order of magnitude

the correcting luminance contribution and ofthe amplification 'YJ needed for tcorrection signal. We are interested in particular in transitions through refence white. For such transients either 1J!is independent of time or we can 'rid of the time dependence of 1J!by adding an appropriate subcarrier signal

TRANSMiSSION AND DECODING IN COLOUR TELEVISION 243

(nstant amplitude and phase to the signal Es. That this áddition is permittedn be explained by a reasoning analogous to that given for the same problemI the compensation method discussed in the preceding section. Hence we~n disregard the time dependence of 1Jiin the case of the transients which are'special interest to us. If d1Ji/dt= 0, eq. (3.60) simplifies to

[2 dAc] . 1/ d2AcS=1/ --- sm(wst+1Ji)+---cos(wst+1Ji)=Ws dt ws2 dt2

1/ [~(dAc)2 + _2_ (d2Ac)2]t cos (wst + 1Ji-O),

ws2 dt ws4 dt2(3.61)

heredAc

-2ws-dt

(3.62)tanO=----d2Acdt2

According to (3.59) the luminance contribution of this correction signal is

t 1/2 [~ (dAc)2 + _1 (d2Ac)2]

ws2 dt wé dt2

again the term at 2 Ws is neglected.or a transition from a saturated colour to the complementary saturated coloure have approximately (see fig. 3-30)

(3.63)

-t(a) (b) (c)

(d) (e)

ig, 3-30. (a) Subcarrier amplitude Ac for transition from a colour to its complementaryolour.b) The same transition in Ac after addition of a constant-amplitude subcarrier signal.c) Idealized representation of dAc/dt for transient of fig. 3-30b.d) dAc/dt for a practical circuit.e) Shape of d2Ac/dt2 for t,he case of fig. 3-30d.

244' J. DAVIDSE

dAc lAo---.dt .chr

(3.6

Hence, the luminance contribution due to the first term of (3.63) i8TJ2A2o/ws2.chr2. Substituting jç = 4 Mc/s and .chr = 0·4 usec as practic~values we find TJ2Ao2/12·8. • JThe luminance contribution of the second term in (3.63) can only be est]

mated with some exactness if the circuits employed are known in detail; it cabe expected that the contribution in generalwill be small (see fig. 3-30e).The luminance deficit in the "luminance notch" is, as we have seen, 0'52 A,o

If we disregard the contribution of the second term in (3.63) we see that thcompensation is only partial if the correction signal is not amplified (that is, iTJ= 1).The compensating contribution would equal the deficit for TJ= 2·5.Thneed for amplification of Es.in the luminance channel militates against a verattractive aspect of the method as it means that the composite colour-television signal can no longer be employed without extra processing. A·moderatamplification can however be accompliehed by introducing some "chromaboost" in the luminance channel. A further point to be taken into account ithat rigorous application of the described method is even undesirable becausthen the mixed highs of the luminance signal, which are also transmitted in thregion of the subcarrier frequency will be attenuated too much. In the casthat the. amplification factor TJ = 1 the suppression of the mixed highs icomparable to that caused by a broad suppression filter. To make clear thipoint let us assume that the luminance signal contains a component cos oostThis component is treated by the compensation circuit like a subcarrier signalThe addition of the second derivative with amplitude l/ws2 cancels the signal entirely. Let us now suppose that a component at angular frequency differinfrom Ws is available. The addition of the second derivative gives no full cancellation. Obviously the resulting signal is TJ(1- w2/ws2) cos wt. Assuming j,4 Mc/s, for f = 3 Mc/s the resulting signal is 0·44 TJ cos oot and for f ==. 5 Mc/the resulting signal is - 0·56 TJ cos wt. Thus, with TJ = 1 the response of thequivalent suppression filter is 44% at 3 Mc/s and 56% at 5 Mc/s (with invertephase). The suppression obtained in this way is more than in general will bwanted. Itmay therefore be desirable to deduce the differentiated signal onIfrom the part of the subcarrier signal situated in the direct neighbourhood 0

the subcarrier frequency. The compensation ofthe luminance errors will then 0

course be less effective since the rise time of the employed subcarrier signal ilonger. But, on the other hand an extra compensation is afforded by the reetification ofthe remaining subcarrier sideband components originating from EsWhat is ultimately the best practical compromise and. what is the best practicadesign can only be found with the aid of suitable experiments based on 'thoutlines presented in this section. . <.

- ------------------------,

g'y


IThe needed differentiation may be accomplished, by means of simple' RC-ctions. It is true that such sections give considerable attenuation but this can~fully compensated by the gain of the sub carrier band-pass amplifier. Figure'131gives the block diagram of the video section of a receiver employing theIscussed compensation method. '

Delay T,

Sub carrierband-pass ampli-fier: delay '72

to synchronousdetectors

Correction: delay T2 in right-hand block must read: delay T.

g. 3-31. Block diagram of video section of receiver employing differentiated subcarriernal for luminance correction. If T2 - Tl is not zero, it must be a multiple of the subcarrier

period TB = 1/2/s.

6:3.5. Final remarksOur discussion on luminance-correction methods has shown that several

, ethods can be devised which promise improvement of the rendering of lumi-lance detail. Until now practical circuits aiming at luminance correction havelot been published. But it is probable that future development of receiving and,'splay techniques will create the need for attacking the residual errors due toe imperfections of the transmission system. The present investigation mightell serve as a starting point for further work in this field.

REFERENCES

~

l)~D. H. Pritchard and R. N. Rhodes, R.e.A. ReV. 14, 205-226, 1953.D. e. Livingston, Proc. Inst. Radio Engrs 42, 284-287, 1954.K. G. Freeman, J. Brit. Inst. Radio Engrs 19, 667-677,1959.

~

) K. McIl wain and e. E. Dean, Principles of color television, Wiley and Sons,New York,, 1956, pp. 435-436.) R. Dressier, Proc. Inst. Radio Engrs 41,851-858,1953. '

,6) J. M. Lafferty, Proc. Inst. Radio Engrs 42, 1478-1495, 1954.o C. J. Hirsch, Acta Electronica 2,132-142, 1957/1958.,8) R. G. Cl app et al., Proc. Inst. Radio Engrs 44,1108-1114, 1956.,0) G. F. Barnett et al., Proc. lnst. Radio Engrs 44,1115-1119, 1956.

1

°) R. A. Bloomsburgh et al., Proc. Inst. Radio Engrs 44, 1120-1124, 1956.1) - D. Payne et al., Convention Record Inst. Radio Engrs 5, rn, 238-242, 1957.12) R. A. Bloomsburgh et al., Convention Record lnst. Radio Engrs 5, rn, 243-249,1957.3) D~ G. Fink et al., Television engineering handbook 15, McGraw-Hill, New York, 1957,

pp.54-57.4) D. G. Fink et al., Television engineering handbook 16, McGraw-Hill, New York, 1957,

pp. 234-246.

246 J. DAVIDSE

16) R. Graham, J. W. H. Justice and J. K. Oxenham, Proc. Inst. elect. Engrs 108511-523, Paper 3468E, 1961.

16) G. S. Ley, Trans. Inst. Radio Engrs BTR 1, 1, 36-43, 1955.l1~ D. G. Fink.et al., Television engineering handbookl6, McGraw-HilI, New York, 195

pp. 251-252.18) J. B. Chatten and R. A. Gardner, Convention Record Inst. Radio Engrs 5, nr, 23

237, 1957.19) R. G. Cl app , Acta Electronica 2, 181-188, 1957/1958.20) S. K.Altes and A. P. Stern, Convention Record Inst. Radio Engrs 2, VII, 46-52, 195

121) B. D. Loughlin, Proc. Inst. Radio Engrs 42, 299-308, 1954.22) K. McIlwain and C. E. Dean, Principles of color television, WHey and Sons, Ne]

York, 1956, p. 16.23) J. Davi dse and F. F. Th. van Odenhoven, Dutch Patent Application 260429, 19624) F. F. Th. van Odenhoven, Internal Report Philips Research Labs No. 3364, 196i26) A. K. Kus tar ëv, Tekhnika Kino i Televideniya 5,9, 38-45, 1961. I26) J. Davidse, Electronic Technology 38, 388-392, 1961. . . I21) F. W. de Vrijer, Acta Electronica 2, 103-109, 1957/1958. I28) M. J. Burgett, Proc. Inst. Radio Engrs 41, 979-983, 1953.29) L. R. Kirkwood and A. J. Torre, R.C.A. Rev. 15,445-460,1954.30) K. McIlwain and C. E. Dean, Principles of color television, Wiley and Sons, Ne1

York, 1956, p. 445.31) P. C. Goldmark and J.M. Hollywood, Proc. Inst. Radio Engrs 39, 1314-1322, 1951

"


4. DECODING WITH BEAM-INDEX DISPLAY TUBES

AbstractThis chapter is devoted to decoding: methods when a one-gun beam-index tube is employed for picture reproduction. First the general princi-ples of beam indexing are discussed. The stability of the index loop, theoccurrence of cross-modulation between indexing information andwriting information and various methods for reducing this cross-modu-lation are investigated. In particular a theoretical and experimentalinvestigation is described on non-linear compensation techniques (pre-modulation) in connection with the employment of a high-frequencypilot carrier. It is concluded that practical application of methods of thistype is not attractive. The same conclusion holds for separation in timeof the writing and indexing functions of the beam and for the employ-ment of a separate index beam. The best method for loop stabilizationand elimination of cross-modulation turns out to be the related-frequen-cy method in which the number of index stripes in the screen structurediffers from the number of colour triplets. Then an analysis is presentedof essential problems in the processing of tbe indexing information ifthis method is employed. Methods for compensation of phase errors dueto frequency variations of the index signal are studied and the dynamicbehaviour of sucb phase-compensation systems is analyzed .. Further-more, methods are investigated both theoretically and experimentallyfor direct conversion of the subcarrier signal into the writing-frequencysignal needed for driving the display tube. Subsequently attention ispaid to the operation of the frequency divider converting the indexfrequency into the writing frequency; in particular the conditions forproper synchronization, when the index frequency is not constant, arestudied. Finally the transient behaviour of the signal processing systemis briefly considered .

.1. IntroductionIn the preceding chapter we discussed the principles and methods of decoding

techniques common to all types of display systems. The only distinction madein the treatise concerned the employment of three-gun and one-gun displayystems. For all known types of three-gun tubes and for focus-mask one-guntubes the discussion presented the essential functions of the decoding system forTSC type colour-television signals. However, one-gun display devices of the

beam-index type (see sec. 3.2.2.2) present a number of additional problems DO

less essential than those discussed before. To :fill up this gap our last chapter willtherefore be devoted to the discussion of some fundamental aspects of decodingwith beam-index tubes. :Here again emphasis will be laid upon problems of ageneral nature while matters of practical technology will only be touchedincidentally.

~.2. Basic principles of beam-index display tubesIn beam-index picture tubes. the phosphor screen, in addition to the colour

phosphors, contains indexing material which provides an index signal when it ishit by the electron beam. All types of index tubes developed or proposed

248 J. DAVJDSE

hitherto have the colour phosphors and the indexing material in the formvertical or horizontal stripes.When horizontal stripes are proposed 1) the number of these stripes is relat

to the number of scanning lines. Colour selection is then provided by auxiliavertical deflection ("wobbling"). The indexing information is employedcorrect the wobbling in order to ensure that the horizontal scanning is centrjcorrectly around the colour triplet. However, tracking of the main verticjscanning so that certain colour triplets are not scanned twice and others nJat all, presents a major problem for which no satisfactory solution has yet bejproposed. Practical developments have therefore centred upon ncreen structurwith vertical stripes, so we shall confine the further discussion to this typescreen.In tubes employing such screens the colour-phosphor lines are hit by t~

beam in a fixed sequence, e.g. R B G, etc. A group of three colour stripetogether with the associated black guard lines separating them, is common!called a triplet. The function of the guard lines has been discussed in se3.2.2.2.The index stripes must be placed such that their position is uniformly relate

to the position of the colour-phosphor stripes. Their capability for yieldinindexing information may originate in several physical properties. A methoextensively practised is the employment of indexing material having a higlsecondary-emission factor (for references see sec. 3.2.2.2). A method alsapplied with success in practice is the employment of fast ultraviolet phosphorsF~rther methods are the employment of conductive stripes or the utilization 0

the light from the colour phosphors proper as a means of indexing. The problems associated with the latter methods are so great that they have got nfurther attention of any importance.Compared to one-gun focus-mask tubes, three additional basic decoding

problems arise with beam-index tubes. First, there is a fundamental stabilityproblem. The index signal obtained from the screen is employed to derive th~proper writing signal which is fed to the gun. If the writing signal in one way oi

Ianother influences the index signal a closed loop is formed, It is.clear that thisleads to a precarious situation. The closed loop can give rise to instabilities andeven ,oscillations.. ,Secondly, derivation of the writing signal from the index signal requireselaborate signal processing. The complex circuitry involved can cause phaseshifts and introduces delay of the information stream, which is a fundamentalsource of indexing inaccuracy.Thirdly, the writin¥ frequency is usually not equal to the subcarrier frequency.

In the index tubes of practical importance, i.e. tubes with vertical stripes notemploying high-frequency wobbling the writing frequency is moreover notperfectly constant during scanning. As a consequence, extra processing of the


lbcarrier signal is required in addition to the transformations discussed in theIreceding chapter. '. .

1.3. Conversion of subcarrier modulation to writing frequency

IA carrier at writing frequency can be derived from the index signal. This

~

riting.:frequency carrier must then be modulated with the colour informationvailable in the subcarrier signal. Basically there are two solutions to this prob-m. The first method is to demodulate the available sub carrier signal with theid of synchronous detectors and remodulate the chrominance signals obtained11 this way on the colour writing frequency. This last has to be performed withalanced modulators. The. stability of the balancing of the modulators has toneet stringent requirements. The method is straightforward but cumbersome.In the second method the subcarrier signal is transformed into a symmetrical

lot-sequential signal according to the theory developed in secs 3.3 and 3.5,~hile the modulation is transferred by a heterodyning process from subcarrier'requency to writing frequency. In practice the frequency conversion is compli-ated by the fact that usually the writing frequency is rather near to thesub-arrier frequency. Practical circuits depend on the method of indexing and willpe discussed later.

"4. Stability of the index loop

~.4.1.General considerationsAt first sight it seems obvious to build the screen of the index tube such that

here is one index stripe per colour triplet. The index frequency is then equal tothe writing frequency. Figure 4-1 gives the block diagram of a basic indexingsystem employing this type of tube. In general such a .simple system cannot

Index I Amplifier 11-+--'1sIgnal

rCcIour processiry 1I-0Il1---circuits r Suf?carfiert slSJna

..--__,/Gun r

..._,.----..." Screen

I Adder II-<~'_-1 MonochromeL..-_""II_---I signal·

L- ------~

Fig. 4-1. Basic indexing system.

work properly. The circuit forms a c1osedloop which will give rise to instabili-ties unless special measures. are taken. Furthermore there is full crosstalkbetween the. writing signal and the index signal. This ,means that phase and

250 J.DAVIDSE

amplitude of the index signal depend on the phase and amplitude of the writisignal whereas for proper working the index signal should depend solely on tj'scanning of the indexing structure. Hence large phase and amplitude errors coccur giving rise to large colour errors.' .To get rid of the amplitude errors a limiter can be inserted in the circuit

front of the chrominance-processing circuits. The system then behaves likeoscillator which is synchronized by the index signal' generated in the systeitself by the scanning of the index stripes. The amplitude of the index signaldetermined by the limiting level while the amplitude of the writing signaldetermined solely by the available video signals. Instead of a limiter an ampltude-dependent electronic switch (e.g. a Schmitt trigger) can also be employeThough amplitude errors can be effectively eliminated in this way nothingdone against the phase errors in the index signal caused by the crosstalkivideo information. Calculation shows that these phase errors are so large ththey greatly disturb colour rendering. It should be noted that this crosstaloccurs even if the tube has linear characteristics since the screen structugenerating the indexing information works as a multiplicative device.

4.4.2. Use of high-frequency pilot carrier

In the light of the foregoing it will be clear that special measures should btaken to get rid of the unwanted crosstalk effects between driving signal anindex signal. A well-known method of attacking such a problem is to separat

Sidehandampffier 48-SMcf

Ccmposifewr:it1ngsl!JIIal

4O'5Mc/s

Pilotfrequeocy

4O-SMcjs

from suócarrierregenerator 4'5Mcls(reference phas~

Suhcarriersignal

4'5Mc/sFig. 4-2. Use of high-frequency pilot signal.

both functions in the frequency domain. To this end a low-level high-frequencypilot carrier can be added to the writing signal. Figure 4-2 shows a solutionbased on this method 2). A Iocal oscillator generates' a signal at 36 Mc/s. A


er stage adds to this signal the reference subcarrier at 4·5 Mc/s produced by .regenerator. This 40·5 Mc/s signal is employed as the pilot signal. A second

~

er 'stage adds the chrominance-modulated subcarrier signal to the localllator signal. In this way a signal at pilot frequency is obtained which isdulated by the chrominance information. When the screen is scanned, thet signal is amplitude modulated in the rhythm ofthe scanning ofthe indexing

~ucture so that sidebands will occur at index-frequency distance from the·rier. By filtering out one of these sidebands and mixing it with.the chromi-ce-modulated pilot signal the writing signal is obtained. Crosstalk due tomultiplicative behaviour of the indexing structure is eliminated in this

nner, as the unwanted signal components can be removed by the filters insideband amplifier. However, due to the non-linearity of the driving char- ,eristic of the tube, modulation of the pilot signal by the writing signal willeady occur in the gun. As a consequence cross-modulation will again existween video signal and index signal and additional measures have to be takeneliminate it.

.3. Particular problems of secondary-emission indexing

ow attention must be paid to a general disadvantage of the use of a pilotrier. To provide sufficient frequency separation between video signal andot signal the frequency of the pilot carrier has to be high. A practical valueabout 40 Mc/s. The indexing mechanism has to follow the high-frequencyot-current fluctuations. In practice this means that only secondary-emissionexing is feasible. Indexing by means of a light-emitting phosphor cannot beployed because of the relatively large decay times of existing phosphors.n secondary-emission indexing two classes can be distinguished. First; trueondary emission can be employed. To this end several "soft" materials canused. A suitable material is magnesium oxide. As is well known, secondary-ission electrons have low velocities, they must hence be collected with theof an accelerating field. The electron transit times involved in the travel

m screen to collector are large and differ considerably for electrons emittedthe centre of the screen and those emitted by the edges of the screen. Theseerences in transit times are interpreted as phase shifts. To compensate form the index stripe pattern must be shifted with respect to the colour-phosphortterns. This complicates the technology of the tube considerably. A furtheradvantage is that the sensitivity of secondary emission is poor. As a result theex signal must be considerably amplified. This enhances the susceptibility ofsystem to spurious signals within the passband of the indexing amplifierked up by the large screen acting as a receiving aerial for such interferingmals.The second method employs no true secondary emission but reflected.·ctrons. The velocity of such electrons is near that of the primary incident .

252 J. DAVIDSE,

electrons. Hence the differences in transit times are much less than with s~secondaryelectrons, so the phase errors associated with this effect are alsom,1smaller. The remaining phase errors cannot be compensated by shiftingindexing pattern as a sufficient yield of reflected electrons can only be obtaifrom heavy materials like, e.g. bismuth oxide, that would produce vis'shadowing-off if the indexing pattern crossed the colour-phosphor patteHowever, since the remaining errors are small they can be compensatedphase modulation of the indexing signal with an extra phase modulator driby suitable signals derived from the line-and frame-timebases 3) as has bshown during experiments.The elimination of the requirement of pattern shifting is certainly an im~

tant advantage from the technological point of view. This advantage is mover stressed by the fact that heavy materials are more easily handled in scrmaking than soft ones. However, the employment of reflected electrons hasimportant disadvantage. Witb true secondary emission the secondary-emissfactor can be well above I but the reflection factor necessarily must be beloHence, the sensitivity obtained witb this indexing metbod is still lower twith true secondary electrons. As a consequence susceptibility to interfersignals is untolerably large. An extensive tbeoretical and experimental invegation on the feasibility of this indexing method has been carried outCornelissen and the present writer. Itwas found that the electron yield waslow that even signal-to-noise conditions for the indexing amplifier became ccal (for numerical data see sec. 4.4.5). Moreover it was shown that due tovery low efficiency of tbis indexing method, even when a separate pilot bewas employed (see sec. 4.4.4) the relative magnitude of tbe remaining ertalk from the writing signal cou1d be so large that instabilities occurred agFor these reasons the method was dropped as impractical.The conclusion from these considerations must be tbat the employmen

secondary-emission indexing suffers from great general drawbacks whicb hto be accepted if a pilot carrier is used.

4.4.4. Employment of separate pilot beamA method providing effective reduction of the crosstalk between index

'and writing information (for references see sec. 3.2.2.2) is the introductio~a second beam, driven by a separate grid fed by the pilot signal. Theoreticamutual crosstalk between pilot signal and video signal is thus eliminated, hea "clean" index signal can be obtained from the tube. This method comPliC~the tube design seriously. Much care has to be taken to assure uniform betracking all over the screen as in fact tbe position of the pilot beam is used asindication of the position of the writing beam. The present writer has alsoried out many experiments with beam-indexing tubes ofthis type, wbichcleashowed the very stringent requirements imposed by this method.

.-----------~ --

TRANSMISSION AND DECODING lN COLOUR 'tELEVISION'. - . .. 253

.5. Application of compensation techniques with single-beam pilot carriersystem

i': different method of reducing cross-modulation between the video signalCl the pilot signal is the employment of compensation techniques. Before~nssing such techniques we shall analyze first how severe the unwanted cross-rdulation is in order to get an impression ofthe requirements to be met by the.mpensation.. . .Suppose that in the simple indexing system of fig. 4-2 the sideband amplifiersses only the upper sideband of the index signal. Let the pilot signal atquencyfp be

Ep = A sin wpt ,

d the writing signal

Ew.= EM' + Aw cos (wwt + 1J!).e assume furthermore that the tube characteristic is quadratic. For the beamrent we have then

I b = k (Ep + Ew)2 . (4.1)

o calculate the screen current Is we have to multiply the beam current I b byunction representing the scanning of the indexing stripes. We shall call thisnetion the indexing function. In analogy to the definition of the scanningnetion (sec. 3.2.2.4) it can be defined as the output signalof the screen if it isnned by a beam of constant current. If a + fl is the secondary-emissiontor of the indexing stripes and fl that of the parts of the screen between theexing stripes (see fig. 4-3), the indexing function can be written as 1 -fl-aSI,

indexing stripes

. 4-3. Secondary-emission factors of index lines and of metal backing of the phosphorscreen.

ere SI is a switching function describing the position of the indexing stripesmpare sec. 3.2.2.4). Assuming that the width of the indexing stripes is ol of thelour-triplet width and assuming that the spot width is infinitely small we canite, expanding SI into a Fourier series

° [1 1 v'3 ]1-fl-aSI = 1-fl-a - +-coswwt + - cos 2 wwt + ... .6 ~ 2~

(4.2)

254 J.DAymsE

EM' Aa .- k sin (wp + ww)t ,:;r; (

The useful term in the screen output current Is is

'while the main cross-modulation component is given by

(l-p--:-i) kAAwsin [(wp+ww)t+V'].

The phase angle V' can of course accept any value, depending on the colobe reproduced. The ratio F of the amplitudes of the unwanted cross-modulat

I signal and the wanted index signal is

(

:;r;(l- P -~)6 Aw

F= -.EM'

(4.a

For saturated colours EM' = tAw holds good. Substituting this into (4.5a) i

obtain

a1-P-6

F=2:;r;----(4.

a

The actual values of a and P depend on the type of screen. For the screen 0experimental tube with bismuth-oxide indexing stripes the followingvalues wfound by measurements: a = 0·08 and P = 0·10. If we substitute these valinto (4.5b) we find F = 70. To assure that the phase of the index signal is sficiently independent of the phase of the writing signal the ratio of unwanto wanted signal has to be well below l. The compensation must therefqreduce the crosstalk by a factor of about 103• This is certainly a very stringrequirement.We note that even if a = 1 and P = 0, which is the theoretical optimu

reflected electrons are employed, the compensationmust reduce the crosst.considerably.

4.4.6. Compensation methods

4.4.6.1. Premodu!ation of the pilot signa!

We shall now analyze practical possibilities for crosstalk compensation.obvious method which, as far as is known, was first proposed by ,Crrner 4) is to apply predistortion of the pilot signal by the writing signal in suemanner that the beam current contains no cross-modulation products betwpilot signal and writing signal.

TRANSMISSION AND DECODING IN COLOUR TELEVISION 255.

Lt the beam current be given by

t,= kEav , (4.6)

ere Ea is the total driving signal (pilot signal + writing signal) and k and y..constauts. Equation (4.6) is only valid if the driving signal always exceeds.black level so that the tube is never driven beyond cutoff. It is always possi-

J' to. ~ssure tha.t this condition is fulfilled in practice by clipping off the writi~glnal at a level Just above the black level. We want the beam current to contam, proper video information and a "clean" pilot component. This can beressed .mathematically as

(4.7)

viously (4.7) can be satisfied if the driving signal is,Ea = (EwV + Ep)l/v . (4.8)

It is not easy to derive such a signal from the available signals Ew and Ep asiswould require two successive non-linear operations. To obtain a more con-ient type of driving signal we expand (4.8) into a binominal series

1 I- (--I)

. 1 [' 1 s; Y Y e; 2(EwV + Ep) tv = Ew 1+- - + --- (-)

" Ewv 2 ! Ewv+ ...1. (4.9)

king the first two terms we get

1 EpEa=Ew+---·

y EwV-1

e signal described by (4.10) can be formed by only one non-linear eperation.can for instance be done by applying Ep to an impedance, which is controlledEw in a prescribed wày.We must now analyze whether this simplification is permissible. To thisurpose we again write the new expression for the beam current as a binomialries. We find

(4.10)

i, = k(Ew+!_~)V =kEwV( 1 +!_ Ep)V =" Ewv-1 " Ewv

k [EwV+Ep+'''-1 Ep2 +(,,-1)(,,-2) Ep3 + ... ].2" Ewv 6,,2 Ew2

"

Itmay be noted that the expansion is only permissible if the series is conver-ent, that is if

(4.11)

II'Ep I < 1" Ew"

(4.12)

256 J.DAVIDSE

This condition is always satisfied as we have assumed that the driving sigalways exceeds the black level. As a consequence Ew has to be such that Ealways positive.All components in the screen output signal produced by the third and hig

terms in (4.11) can be removed by simple filtering since they represent molated harmonies of the pilot frequency. So (4.10) represents the simplest foof driving signal yielding the desired output signal. We note that this simplication of the driving signal is permitted because the index signal is filtered in tindex-signal amplifier.If the non-linearity of the tube transfer characteristic were the only ca

of cross-modulation between video and indexing information a driving sigobeying (4.10) would, ifit could be formed accurately enough in practice, prvide an indexing signal free of cross-modulation. However, there is a second,less important, cause of this cross-modulation. The indexing function whiwas defined earlier (sec. 4.4.5), depends on the width of the scanning electrspot and on the current distribution within it, whieh in turn depend on tbeam current and hence on the driving signal. Thus the indexing functionmodulated by the writing signal. Since the index signal is the product of tindexing function arid the beam current, the index signal is also modulatedthe writing signal. Further analysis of this phenomenon shows that adaptatiof the premodulation technique to this effect is almost impossible. Hence, ev .

. if the driving signal is perfectly formed in accordance with (4.10) a certaresidual cross-modulation will exist. The analysis ofthe performance requirments for the premodulation technique (sec. 4.4.5) has shown that operatiowithin very narrow tolerances is essential. The occurrence of the spot-dependecrosstalk decreases the available margin still more.Nevertheless, some experiments were set up to learn to what extent the sp

rious modulation of Ep can be suppressed with practical circuits: In theexperiments the signal Ep/EwY-1 was formed by applying Ep to an impedanwhich was properly controlled by e; The latter could be realized by utiliZi~the internal anode or cathode impedance of a vacuum tube with the signalapplied to its grid. It can be shown theoretically that in this way a good approimation of the demanded operation can be obtained. uwas found that t~overall circuitry; including the amplifiers which are needed, is very compleand 'must meet stringent stability requirements. But' even Wit.hc.ircuit.sdesignevery carefully the performance obtained was far below what was required.must therefore be concluded that premodulation of the pilot signal does nprovide a practical solution to the problem of preventing cross-modulatiobetween writing and indexing information.

4.4.6.2. Alternative techniques

If no pilot signal is added to the writing signal the screen current is


Is = k(l-f3-aSI)EwV •. '

(see sec. 4.4.5).

.: require a signal representing SI uncontaminated by the writing signal.~ViOUSlythis might be achieved by multiplying the screen output signal by anal of the type .

lEw-v cos wat ,

ere Wa is an auxiliary frequency. This provides a signalof the type SIcos wat.demodulation we obtain the desired signal SI. The problem of deriving, signal Ew-v cos wat resembles that of achieving premodulation. It is there-e probable that this technique will not make things easier. The need for an·ditional non-linear operation for achieving the multiplication of the screennal and ~heauxiliary signal makes this method evenmore complex., other possibility is to employ a pilot carrier which is not premodulated'le a suitably chosen harmonic ofthe pilot carrier is actually employed as theot signal. In certain cases crosstalk of the writing signalon certain harmo-s of the pilot carrier will be less than on its fundamental component. Fortance, in the case of an exactly quadratic characteristic, the second harmonicthe pilot carrier is free of crosstalk by the writing signal. However, practicalplication of this method suffers from two important disadvantages. First,actical tube characteristics never exactly obey power functions and moreoverey differ from tube to tube. So here too a difficult tolerance problem has to belved. Secondly, pickingup a small amplitude harmonic of a frequency, whichelf is very high, from an extended structure like a phosphor screen is hardlyssible in practice. The pilot frequency has to be very high in order to reducevisibility and to prevent interference patterns. Practical values may be in, thenge from 30 Mcfs to 50 Mcf~. The wavelength of harmonics will therefore. bethe order of magnitude of the dimensions of tube and screen.A different method, which seems promising at first sight is to linearize the~am-current characteristic for the high-frequency pilot signal by feedback.r particular, feedback obtained by inserting a cathode impedance Zk (cathoderedback) seems to offer the advantage of simplicity. However, the practicallossibilities of this method are very limited. By approximation, the transcon-juctance gm of the gun is detemiined only by the writing signal Ew, since inèneral Ep -e Ew. We have therefore

(4.13)gm = vk Ewv-1 •

he pilot current is then given by

. vk Ep E~V-llp=-------

1+ yk Zk Ewv-1

r Zk is chosen such that it represents a high impedance for the pilot signal11-.14)is reduced to . .

(4.14)

258 J. DAVIDSE

. Eplp=-.

Zk(4.1

This requires vk Z« EwV-1 ~ 1 in the frequency range which is of interest fthe signal Ep EwV-1, that is, a broad band around the pilot frequency. The tranconductance of practical guns is in the order of 0·02mA/V. To provide tenfofeedback, which is presumably not sufficient, Zk has to be of the order of 5(k.o, which is far beyond practical possibilities. The method might be-saved 1employing amplified feedback. From the foregoing analysis it is clear that cosiderable amplification of the feedback signal would be needed. Assuming tha cathode impedance of 500.0 can be realized with the bandwidth required fproper operation, amplification by a factor of 1000 would be essential. Tldesign of an amplifier meeting the requirements is so complicated that it mube concluded that application of the feedback method is not feasible in praetic

4.4.7. Related-frequency methodWe have.seen that the writing and the indexing functions of the tube must 1

separated without mutual crosstalk being introduced. In the foregoing we dicussed a method of separation in the frequency domain by modulating tlbeam with.arhigh-frequency pilot signal. It was shown that a disadvantage (this method is that modulation of the pilot signal by the writing signal occuin the gun. Elimination 'of the crosstalk required far-reaching countermeasunofwhich only.the.double-beam method proved to be practically usable, witholbeing attractive, however.The cross-modulation caused by non-linearity of the tube characteristic coul

be avoided if frequency separation were achieved without modulating the beai'current by a pilot signal. This is made possible by using an indexing structuihaving a periodicity other than that of the phosphor triplets. The index fnquency will then inherently be different from the writing frequency 5.6). In ordethat the position of the beani can be located with respect to the position of ttcolour-phosphor stripes and in order that the writing frequency can converiently be derived from the index, frequency, there must be a simple relatiobétween the index frequency and the writing frequency. In choosing the ratio (index frequency to writing frequency residual cross-modulation errors must 1:accounted for. Moreover, it is very desirable to have the index stripes in tbblack "guard bands" between the colour-phosphor stripes. The number (suitable index-stripe àrrangements is therefore limited. This question is discu:sed at length by Graham, Justice and Oxenham 5). Their conclusion is that twstripe arrangements are promising, viz. three index stripes in every two COIOli

triplets and three index stripes in every four colour triplets. In the first case tbindex frequency is ~ times the writing frequency, in the second case it is i timethe writing frequency. In the latter case there is more risk of crosstalk betwee


le high-frequency part of the monochrome component in the writing signal~d the index signal. For this reason the 3 : 2 ratio is to be preferred. Ourirther discussion of this method will therefore be based on this choice.IAn important advantage of this method is that it can employ photo-electricidexing with fast ultraviolet phosphors because the frequency of the indexgnal is relatively low, thanks to the fact that no high-frequency pilot modula-on of the beam is needed. Compared with secondary-emission indexing, photo-lectric indexing has several advantages. First, no spurious electric signals suchs short-wave broadcast transmissions are picked up by the indexing system.econd, the complications due to transit-time differences are avoided. Third,w-noise amplification of the index signal can be conveniently obtained with ahotomultiplier.A disadvantage of the related-frequency method is that the indexing infor-ation obtained is not unambiguous. To derive the writing frequency from the"dex frequency a frequency divider is needed. If the division has to be by aactor of 3/n (because the index frequency is 3/2 times the writing frequency),here n is an integer being not a multiple of 3,the output signal can have one ofee different phases, mutually 120° apart. To overcome this difficulty the

equency divider is started at the start of each line scan by index pulses 0btain-d from "run-in" index stripes 5). These "run-in" stripes are located at theeft-hand edge of the screen. During the time they are scanned the writing signals suppressed, so no unwanted cross-mod.ulation can occur. Their spacing is suchhat the index frequency when they are scanned equals the writing frequencyr a submultiple of it. Once the divider is started by the run-in signal the indexignal has to be maintained until the end of the scanning line is reached inrder to maintain the proper phasing of the index signal. For this reason thelack level of the writing signal is not permitted to decrease below a certaininimum level, otherwise index pulses are lost.The run-in system and the associated circuitry complicates the overall

ndexing system considerably. However, experiments have shown that withppropriate circuits stable and reliable operation is quite within the practicalpossibilities. This system is therefore interesting and promising enough to beorth a more detailed discussion, which will be presented in a subsequenteetion .

.4.8. Time-domain separation of writing and indexing functions

A logical alternative to separation in the frequency domain of the writing and.ndexing functions of the picture tube is separation of these functions in time.n that case a gating system switches the tube between writing operation andindexing operation. As these functions. are effectivelyseparated in time, crosstalkbetween writing information and indexing information cannot occur. Hence.there is nothing against having one indexing stripe in each colour triplet. During

260 . . . J. DA VIDSE •

à. time interval around the scanning of the index stripes' the driving voltage ahence the beam current is clamped at a fixed value. Simultaneously, the indsignal is gated so that 'no crosstalk with the writing information can occur.should bé noted that ideal separation in time requires infinite bandwidth of tgating circuits.This method seems at a first glance logical and attractive. The writer h

carried out an extensive theoretical and experimental investigation into tpractical aspects of this method. It turned out that realization of the concerequires very complicated circuitry working within very close tolerances. Tcause ofthis disappointing result is essentially that the gating has to be extremIy rapid in order to avoid confusion between the two functions of the tubThis requires the. employment of millimicrosecond pulse techniques at higlsignal levels. In view of the disappointing final results, we shall neither deal hewith the detailed analysis of the problems involved in the realization of thsystem nor with the experiments carried out.

. . .4.4.9. Conclusion on methods concerning stabilization and elimination of cros

modulation .

At this stage it is useful to formulate a conclusion on the practical usefulneof the various methods aiming at reduction of crosstalk between writing a;~indexing information. The foregoing analyses show that of the methods discu~sed, i.e. employment of pilot carrier on separate beam, employment of pilocarrier in c~nne~tio? with a compensation method, related-frequency indexin~and separation ID time, only the related-frequency method proves to be praqtically feasible. The other methods complicate the design of the tube and/othe circuitry impermissibly while all of them, perhaps with the exception of thdouble-beam system, require circuits working within very close tolerances. Ithe remainder of this chapter emphasis will therefore be laid upon the relatedfrequency method of indexing.

4.5. Indexing errors due to phase shifts and time delay in the signal-processincircuits

4.5.1. Needfor integration ofthe indexing informationThe third problem of a general nature we have to discuss pertains to th

indexing errors associated with the processing of the indexing informatie(see sec. 4.2).We have already" seen that the rough indexing signal as obtained from th,

screen of a secondary-emission tube or from the anode of the photomultipIie~. in the case of u.v, indexing, is contaminated with signals originating from th,:vrit~ng signa.l and with cross-modulation pr~ducts bet~een writing and inde.xjmg information. We have further seen that, If appropriate measures are taken(e.g. application of related-frequency indexing), things can be arranged suchl

I

TRANSMISSION AND DECOD~NG IN COLOUR TELEVISION 261

at disturbing signal compönents at the same frequency as the wanted signalle greatly reduced. The remaining unwanted signal components must be~moved with the aid-of filters. From another point of view we can also say thatIe have to average the available indexing information over a certain time inrder to get rid of the unwanted periodic components. This requires integrationfthe information .

.5.2. Consequences of the integration of information

There are several methods of carrying out the integration in practice. A well-nown 'method is the employment of an automatic phase-controlloop (ape-oop). We then have an "active" integrator. A filter can be regarded as a passivetegrating device. Its integration time is inversely proportional to its bandwidth.Averaging will essentially delay the information. We must therefore concludeat a certain minimum time delay isunavoidable. Ofcourse this delay has conse-

[uences for the picture reproduction: the indexing information is "stale" athe moment of its actual use. When changes in the indexing information arelow this does not present difficulties, but rapid changes in the indexing infor-ation can cause trouble. In practice such rapid changes can occur due toringing" of the line-timebase, that is, contamination of the sawtooth wave-orm ~y sine-wave oscillations of a rather high frequency. The scanning veloc-ty shows these variations also and hence the frequency of the index signaloes so. If the period of the oscillations is in the order of the time delay of therocessing circuitry difficulties can be encountered. .A further cause of rapid changes in the index signal is the remaining cross-odulation from the writing signal into the index signal. The disturbing effectf such signal changes can be enhanced by the delay in the processing chain.he cross-modulation causes a phase transition in the writing signal to giveise to a phase transition in the index signal. This transition is fed back with aertain delay to the gun by the processing circuitry. In consequence the transi-ion in the writing signal is followed by a "ghost". We shall consider this phe-omenon in a subsequent section in more detail.The employment of an integrating device has a second important effect.uch a device will introduce a phase shift as a function of frequency. With anpc-loop this phase shift can be kept very small by providing sufficient loopIgain.However, if such an ape-loop is to be applied in practice serious trouble~s encountered from deviations of the index frequency from its nominal value.Euch deviations occur due to inaccuracies of the line-timebase. The design of an~apc-Ioopwi.th~wide working-frequency range, a high loop gain and the properintegration time is very difficult and hence in practice unattractive. The em-toyment of a passive integrating system, i.e. a band-pass amplifier, is thereforecommon practice. But then the phase shift as a function of frequency can be

262 J. DAVIDSE

considerable. As is well known, this phase shift is closely related t.o the del~.time for the information that is passed through the filter.If cp(00) denotes the phase characteristic of the filter the time delay produc

by the filter is given by Tg = dcp/dw. If the phase shift at the nominal indfrequency fo is cpo, the phase shift as a function of frequency is given by

00

cp(w) = CPO+ f Tgdw •wo

(4.1

If 7:gis constant over the frequency band of practical importance (linear phacharacteristic), then

cp(w) = CPo + 7:g(w-wo). (4.1

If we denote the phase shift when changing the frequency from Wo to 00 bLlcpand 00-000 by Llw, we can write (4.17) simply as

LIep= 7:gLlw. (4.1&Let us now conside~ an indexing system in which the writing frequency il

8 Mc/s.Let us assume that the index frequency ana hence the writing frequencchanges by 5% of its nominal value. In practice one has to expect frequencvariations of this magnitude if a line-timebàse of common design is employeAccording to (4.18) the phase shift introduced by the frequency change is the144°per {Lsecdelay time. The phase shift which can be tolerated is in the ordeof 5°. The delay of practical processing circuits is usually' 1 to 2 (Lsec,so thphase error is much larger than can be tolerated.

4.5.3. Compensation of phase errors due to frequency variations of the indesignal

4.5.3.1. Application of frequency-dependent phase modulation

. It is obvious that compensation of the phase errors due to frequency variations of the index signal is indispensable. One method of achieving this phascompensation is by measuring the momentaneous frequency with a frequencjdiscriminator whose output is employed to control a phase modulator. Suchphase modulator can be devised for example with an ordinary reactance-tubcircuit. This method is straightforward but it has the disadvantage of requirinrather much extra circuitry which has to meet high stability requirements.

4.5.3.2. Phase compensation by providing two signal paths

. Another method of phase compensation consists in adding an equal phasshift, but opposite in sign. This can be done with the aid of a mixer circuit. Pigeneral theory of phase-compensation methods of this type is given by Rihac~zek 7). In practice this type of phase compensation can often be applied withoutmuch extra circuitry. '


A fInpUt

Output

F

ig. 4-4. Phase-compensation system for which the output frequency is equal to the inputfrequency.

We shall illustrate the method here with two examples. Figure 4-4 shows astem in which the output frequency is equal to the input frequency. The signaldelayed by 0U1 in the first part of the signal-processing system. It is subse-uently fed to two signal paths in parallel. The first one contains a frequencyultiplier, the second one provides the compensating delay 0U2. The mixer stagedesigned in such a way that at its output only the component at the differenceequency (2/1-/1 = f1) is present. The delay introduced by the output ampli-er is denoted by og3.

The phase shifts occurring under steady-state conditions due to deviationsom the nominal frequency are:

at point B, LI!p= 0u1 L1w,at point C, L1!p= (OUI + 0u2) Llw,at point D, L1!p= 2ou1 Llw,at point E, LI!p= «U1-ou2) zlco,at point F, LI!p= (ou1 + 0u3-'--0u2) !.Iw.

Hence, if 0u2 = 0U1 + 0u3 a change in frequency has no influence on thehase. However, it should be noted that this is only valid for the quasi-station-ry case. In the dynamic case when the signal frequency changes duringe time that the signal is processed (that is OOI + og2 + 0u3) the effect of theelay of the indexing information (the "staleness of the indexing information")ill become apparent. An approximating calculation may well illustrate thisffect,Let the input index signal be given by sin !pet). Let wet) denote the frequency ofe signal. We can write wet) = Wo [1 + g(t)], where wo is the nominal index

requency and, get) describes the variation of index frequency with time. Foronvenience we assume that g(t) = 0 for i <0. We then have '

t!pet) == f w(t)dt = wot + wó [G(t)-G(O)],

o(4.19)

t.

here G(t) denotes the primitive .of get). Hence the index signal can be written

et = sin [wot + wo{ G(t)-G(O)} ]. (4.20)

264 J.DAVIDSE

This signal can be regarded as a phase-modulated carrier of frequency wo. Tmodulation is described bythe second term in the.argument, thatis, wo{G(t)G(O)}. This modulation gives rise to the occurrence 'Ofsidebands of the indsignal. If the bandwidth of the index amplifier is so large that all sideband coponents are within its passband, the effect of traversing the index amplifier cbe described as a delay of the modulation content by a time Tgl. The signalpoint B in fig. 4-4 may then be written

eB = sin [wot + Wo {G(t-Tgl)-G(O)} 1, (4.2since G (-Tgl) = G(O).Hence, by traversing' the index amplifier the phase modulation of the indsignal has changed by an amount

.dtp= Wo[G(t)-G(t-Tgl)]. (4.2j

Assuming that the dynamic behaviour of the frequency multiplier can be appro .imated as in the stationary çase, that is, by doubling of .dtp, and assuming ththe bandwidths in all branches of the circuit are so large that any irnportasideband components are transmitted, we find for the further points in fig. 4-

point C, .dtp= Wo[G(t)-G(t- Tgl- Tg2)],point Dç áip = 2wo[G(t)-G(t-Tgl)],point E, .dtp= woG(t)-2WOG(t-Tgl) + WOG(t-Tgl-Tg2),point F, .dtp= woG(t)-2WOG(t-Tgl-Tg3) + woG(t-Tgl-Tg2-Tg3)'

·lfwe again choose Tg2= Tgl + Tg3the final phase error is.dtp= Wo[G(t)-2G(t-Tg2) + G(t-2Tg2)]. (4.2

It must be noted that, due to the simplifying assumptions we have madthis expression does not present more than an approximation of the behaviaof an actual signal-processing circuit. However, .itmay well serve t.oincrease~".insight into the phenomena we are studying here.It is interesting to consider, with the aid of (4.23), what happens in cei:tai

special cases. First, the case that the index frequency increases linearly wittime during a certain time interval, that is get) = at, where a is a constant.then have for values of t > 2 ~g2

G(t) = tat2 and .dtp(t)= WOaT2g2= woa(Tgl + Tg3)2..

If no phase compensation had been applied the phase error would beLltp(t) ==: Wo[at(Tgl + Tg3)-ta(Tgl + Tg3)2].

So for values of t ~ Tgl + Tg3 the phase compensation still provides a versubstantial reduction ofthe phase error.

. , 1Secondly, weconsidetthecaseg(t) = sin at. Wefind then GCt)= -cos at an

afor t > 2 Tg2


,'-wo 'à('r9'l + Tga) -: { , "!}.dtp = - 4 - sin2. cos a ,t -: (;91 + Tga) .

a . ',2 'I

«(26) ,

ithout phase compensation the, phase error ,'o/.ould be '

A 2000 . a (T,91 + Toa) . '{ (Tgl + Toa)}LJtp = - - sm ' sm at,,' •, a 2", " 2

y comparison of (4.26) and (4.27) we see that the phase compensation reduces'le ampli,tude of the oscillating phase error so long as

. (4.27)

, a (Tgl + Tga) , no2sm <1or a (Tgl + Toa) <-.

2 3

ence, as could be expected, if the period of the ringing frequency is of therder of magnitude ofthe circuit delay the phase compensation no longer works ..can then even magnify the phase error. '

f

.g. 4-5. Phase-compensation system for which the output frequency is two thirds of the inputfrequency.

Figure 4-5 shows the second circuit 5) illustrating the phase-compensationethod. Here the output frequency is two thirds of the input frequency, so thisrcuit conveniently meets the needs of the related-frequency-indexing method.Listing once again the phase errors due to deviations from the nominal indexequency at various points we have

at point B,

at point C,

.dtp = TOl.dw,Tgl

.dtp = _ .dw,3

A _ (Tgl + Tg2) ALJtp - -- LJW,3

.dtp = (j-Tgl-tT02) zlco,

.dtp ={!(Tgl + Tga)~tT02}Liw.

at point D,

at point E,at point F,

o the condition for phase compensation is Tg2 = 2 (.Tgl + Tga). For the dynam-~ case a calculation similar to that pertaining to the first example can be ear-l. d ''le out. .: , ,,', ,

265

4.6. Signal-processing methods with related-frequency indexing

4.6.1. Inventarization ofthefunctions ofthe signal-processing circuits

The general considerations on beam-indexing principles and methods hamade it clear that among the thinkable methods the related-frequency methoffers the best practical realization. In particular it has the advantage ofquiring a comparatively simple display tube employing only one beam w .the index stripe pattern can easily be derived from the colour-phosphor stripatterns.It is therefore worth while considering in some more detail the signal-pr

cessing problem for this method of indexing. For giving these considerationpractical background we shall first discuss the question of how to choose twriting frequency. Theoretically, ifthe luminance detail comprised in the lU~nance signal should be reproduced to its full extent the width of a colour triplshould equal the width of one picture element. Hence the colour-writing frquency should be twice the cutoff frequency of the luminance channel, which5 Mc/s for the 625-line standard. However, focussing of the beam has to be Igood over the entire screen that the beam never hits more than one phosph/stripe at the same time. Hence, any increase in the width of the coloïtriplets facilit~tes the desi~ of t,he focussin~ and deflection .syst~m. I

The transmitted composite Signal contains the subcarrier Signal at aboiI

4·5 Mc/s. In the receiver the components of the luminance signal around tlsub carrier are suppressed. Moreover, even if the colour triplet is slightly widithan one picture element, so that the luminance signal can change within tlscanning of one triplet, the luminance information is not entirely lost, thougit is rendered slightly erroneous. Takin~ all these considerations into .accouiit may be concluded that choosing the colour writing frequency about 8 Me,is a good compromise, The index frequency will then be about 12 Mc/s in tl3 : 2 related-frequency system. Where numerical data are needed we shall en,ploy these figures in the further discussion.

Let us now draw up what is to be expected from the signal-processing chaiiWe have two signals comprising the input information: the index signal obtaiied from the photomultiplier and the composite colour-television sign:obtained from the video detector in the receiver. The output information mube delivered in the form of the composite writing signal.

The index signal at 12 Mc/s obtained from the photomultiplier has first to 1integrated in order to eliminate the influence ofunwanted periodic componentFurthermore, its amplitude variations have to be eliminated as the indexirinformation is solely contained in the phase of the index signal. Then the indefrequency has to be converted to the writing frequency by means of a frequentdivider. Finally, phase compensation (sec. 4.5.3.2) has to be provided.

The detected colour-television signal has to be transformed into a symmetric

266 J._DAVIDSE


tS signal (sec. 3.3.1), while the modulation has also to be brought on to the!iting frequency. The lumina~ce signal has to be translated into a symmetricalrnochrome signal (sec. 3.3.1). '

f.2. Demodulation-remodulation method!Aswe have cseen ear1ie~,two basic methods for the processing of the colourformation can be distinguished. In the first method the NTSC signal is fullymodulated so that two colour-difference signals are obtained which are sub-uently remodulated onto the writing frequency. The processing circuit mayén take the form presented in' the block diagram of fig. 4-6, in which phasempensation according to the method of fig. 4-5 is incorporated, The ampli- .

• l • •

'--_--' EM '------'Fig. 4-6. Index-tube decoder employing demodulation-remodulation method.

er following the frequency divider must contain a delay line to adjust the delayf the compensating circuit to the correct value Tg2 (see fig. 4-5). The running-ingnal must ensure that the frequency divider starts in the right phase. ,Thenning-in system will be considered in more detail in a subsequent section.The modulators must be of the suppressed-carrier type. Incomplete balancingIill lead to colour errors ("colour set-up"). Balancing has therefore to be good~ithin close tolerances. The overall circuit is. straightforward but rather cum-ersome.

.6.3. Direct conversion of subcarrier signal into writing-frequency signól

.6.3.1. Elimination of unwanted conversion products

We shall now c~msider the second method of signal processing, which con-

;?68 J. DAVIDSE. . ..,~

'sists in direct conversion of the subcarrier-freqüency signal into the writhfrequency signal. We shall assume that the NTSC sub carrier signal has fbeen converted into a ces signal at subcarrier frequency (see sec. 3.3).We must first pay attention to an essential problem in the conversion of 1

subcarrier-frequency signal into the writing-frequency signal. Assuming that 1index frequency and hence the writing frequency can vary by 5% due to devtions of the scanning velocity, the writing frequency can vary between 7·6 a8·4Mc/s. The filters passing this signal must have a still wider passband to allefor fast variations of the writing frequency, When the writing-frequency sigris modulated with the colour information the total occupied bandwidth i'creases considerably. Assuming that the chrominance signals have a bandwidof 1 Mc/s, the passband for the writing frequency has to be at least from (to 9·4 Mc/s.These figures make clear that the frequency conversion is not so easy. T

passbands of input and' output signals tend to overlap each other, hence filning requirements are very high. In addition we have to note that frequenconverters are inherently non-linear devices, so the production of harmonicsthe input signals and higher-order conversion products can hardly be avoidePart of these unwanted components can again be within the passband of toutput signal. This increases the filtering difficulties still more. In many casthe unwanted components cannot be removed by filtering. Only balancecircuits can then provide a solution, but such circuits are inherently criticand therefore unattractive.

4.6.3.2. Phase compensation in systems employing direct conversion

The practicability 'of circuits yielding direct conversion of the subcarrisignal depends strongly on the ease of removing the unwanted components :the mixer outputs. There are two further criteria which have to be taken imaccount when judging a particular practical circuit: the circuit must incorprate phase compensation and the overall delay must be as slight as possiblLet the index frequency be denoted by f, the writing frequency by i f. 1

general terms we can say that the signal at i/must be produced by mixing t"signals at frequencies mfl3 and n//3,. where m and n are positive integers all'm-n = 2, as shown in fig. 4-7 (compare fig. 4-5 where m:__ 3 and n = 1). (course this circuit must be extended in order to incorporate the conversion «

Fig. 4-7. Generalized indexing-processing circuit.

TRANSMISSION AND DECODING IN COLOUR TELEVISION , 269

, subcarrier signal. To this end at least two further mixers are needed: one to:d'the modiilated subcarrier signal and one to add the reference subcarrieriich.is derived from the colour synchronization signal (subcarrier "burst").~nce at least three mixers are needed to provide the' entire signal processing~ludiDgphase compensation.fhe two subcarrier mixers roay in principle be located anywhere bebind the'quency division. Since there are three branches in the basic circuit there are

it1

ft

3

ft5

'it ,

qt

Fig. 4-8. Survey of the possible circuit arrangements employing.direct conversion of subcarrier signal. .

IA denotes index-frequency amplifier.WA denotes writing-frequency amplifier.f denotes index frequency.Is denotes subcarrier frequency.M denotes mixer stage.

270 J. DAVIDSE

six different circuit arrangements. Furthermore, in each of these cases we luagain two different possibilities: the subcarrier frequency can first be ad:and then be subtracted and 'alternatively the subcarrier can first be subtracand then be added. This leads to 12 different arrangements in total. Figure,gives a comprehensive survey.

In figure 4-9 the group delays in the branches of the generalized circuit:indicated. In the same manner as was done in connection with the circuits

2


Fig. 4-9. Group delays in the generalized circuit ..

s 4-4 and 4-5, for the circuit of fig. 4-9 the generalized condition for phasepensation can be derived. One finds

(4.28)

e signal delay from input to output through the upper branch is Ta + Tb +while the signal delay through the lower branch is Ta + Tc + Td. Substi-ting (4.28) we find '

Ta + Tc + v« = (1 + 2/n) (Ta + Td) + mln Tb. (4.29)

As m> n, it is always true that Ta + Tc + Td > Ta + Tb + Td, hence theerall delay of the circuit is determined by the delay of the lower branch. Wen call the lower branch the compensating branch since the phase shift causedTChas to compensate for the phase shifts caused by Ta, Tb and Td.From (4.29) we see that for minimum overall delay Ta, Tb and va must beinimum, However, the minimum values of Ta and Td depend on the selec-ity requirements for the indexing and writing amplifiers. Increasing Td

yond its minimum value has obviously an unfavourable influence on the over-I delay. Hence, if possible, locating the subcarrier mixers with their inherentnd-pass filters in the writing-signal branch must be avoided.To keep Tb small, none of the subcarrier mixers should be inserted in the m-anch. We must therefore conclude that from the viewpoint of minimizationoverall delay, circuit arrangements in which both subcarrier mixers are in thempensating branch (n-branch) are generally more favourable. In the survey,fig. 4-8 circuits 3 and 4 are of this general type.Considering (4.29) further we see that for small overall delay the value of nould be large. At a first glance one might think that the larger n is, the smallere overall delay. However, it has to be noted that ifboth subcarrier mixers areserted in the compensating branch a certain minimum delay in this branch isnavoidable, From (4.29) we know that Tc = 2/n (Ta + Td) if Tb ' O.In conse-uence there is a practical limit to the delay minimization by increasing the

alu~ of n: increasing n beyond the value 2 (Ta + Td) ,determined by the. Tc mln

inimum values of Ta, va and Tc which can be attained, yields no furtherecrease in the overall delay. In practice it holds that (Ta + Td)mln is aboutqual to 2(Tc)mln, so n = 4 can be considered to be the practical optimum.

272 . J. DAVIDSE

Sincem-n = 2 must be valid, optimum overall delay is attained with m-=n = 4. Of course common factors ,inm and n can be taken together, as is doin the circuit of figure 4-10, where the index frequency is first multiplied b)1(the common factor in'm and n) and then divided by i. It is assuined that ~delay produced by the filters in the compensating branch is half the delay in 5index amplifier and the writing amplifier. Ifthis is not so, delay has to be ad 'in one of these branches.

Inpvt 4'5Mcls

Subcarriersignal

4'511<:/5

Input

Index signal .....__---'at 12Mcjs

Inpvt

£YFig. 4-10. Index tube decoder employing direct conversion of subcarrier signal.

Critical consideration of the operation of the mixers shows that all unwantcomponents in the mixer outputs can be removed by filtering. Filtering ~quirements are rather severe, though not beyond practical feasibility. They Cjbe alleviated when the unwanted components are attenuated by employing simpbalanced mixers. Balancing need not be critical since only moderate attenuatiof the unwanted components is aimed at.

4.6.3.3. Alternative solution; experiments

The described solution was found in a straightforward and logical way.complies with all the requirements set forth in advance. However, if we arebe certain that there are no alternative solutions yielding equal or better perfO,lance, the full scale of possible circuits for usable values of m and n has toconfronted with these requirements. This demands the separate investigationabout 100 possible circuit arrangements. .In an investigation of this type no better circuit was found as far as filterin

1requirements and overall delay are concerned. One circuit seemed to be on]little inferior in this respect while it had the advantage of not requiring an extt

" , I

.---~~-----~- -----_-- -- -~ -- -


~quencymultiplier for the index signal (m = 3, n = 1). Figure 4-11 shows thesic circuit. It may be noted that the second subcarrier mixer is not in thempensating branch. This is not a great drawback since the delay associatedth this mixer is but small because ofthe great bandwidth needed in the output

rUit.

~out

f (12Mc/s)

Fig. 4-11. Signal-processing circuit with m = 3, n = 1.

The overall delay in this circuit is more than in the former one because of thewer value of n. In addition, two of the three mixers are rather critical. Inixer I the second harmonic of both input signals is within the passband of the

~.

tPut Signal..ln mixer III there is a corn.parable situation. The application oflanced mixers is therefore essential. Whether this complication can be accept-or not depends on the visible effect of errors due to imperfect balancing.

he only method of investigating this point is to build a practical circuit. Anperimental decoder based on this method was therefore built and measure-ents carried out on the effects of imperfect balancing of the mixers. Thesult was disappointing: it turned out to be possible to produce pictures fullyee of the second-harmonic errors but the slightest disturbance of balancingoved to be disastrous.In order to make possible a critical comparison, an experimental circuit basedthe first method (n = 4, m = 6) was built as well. The mixers were roughly

alanced to alleviate the filter requirements. Here the result was much better; as! as to be expected, balancing was less critical here so that stable operationould easily be obtained. .

IWe ~on~lude that among the. syste~nswhich can be derived from the ~eneral-ed CIrCUltof figure 4-7, that III which m = 6, 12 = 4 and all subcarner pro-essing in the compensation branch, is by far the best.

IFor the sake of completeness we should note that the generalized circuit ofg. 4-7 does not cover all conceivable methods of converting the subcarrier"lgnalinto the writing signal. We have restricted the discussion to systems in~hich the subcarrier signal is added behind the frequency divider. Formally it~ also possible to add the subcarrier in front of the frequency divider but thenhe added frequency is also .divided, This means that not the subcarrier fre-

273

·'. :- 274 J.DAVIDSE

quency proper has to be added but a multiple of it. Moreover, in the runningcircuit too an auxiliary signal has to be added at the proper harmonic of tsubcarrier frequency. In practice such circuits are so complicated that tbcannot compete with the systems presented in figures 4-6 and 4-10. The salholds for systems in which the frequency conversion is effected with the aidan auxiliary frequency generated by a local oscillator (compare the circuitfig.4-2).

A further range of possible circuits is found if the reference subcarrier is fto a frequency multiplier yielding a suitable harmonic of the subcarrier fquency while the colour information too is converted to this frequency. Tsignals obtained in this way can be applied to the circuit of the general typefig. 4-7. Here also the overall circuitry becomes so complicated that praeticapplication is quite unattractive.

4.6.4. Frequency division and the running-in problem

4.6.4.1. Basic principles of frequency dividers

In the related-frequency. indexing method the employment of a frequendivider is essential. We have seen that the indexing frequency must be mulplied by nl3 and by ml3 where at least one of the quantities n and m is notmultiple of 3, so in fact, frequency division is needed.

In the frequency range of the index signal regenerative frequency dividepresent the simplest solution to this problem. Their operation has been tisubject of many investigations 8,9,10,11), so here a simple explanation of tigeneral principle may suffice. Figure 4-12 shows the basic circuit. The incomiifrequency fis multiplied by q. On the other hand, the output frequency nll3multiplied by p. Both q and p are integers. The signals at frequencies qf alpnfl3 are fed to a mixer stage, whose output is tuned at the frequency qf - pnfJ

Fig. 4-12. Basic circuit of frequency divider.

The condition for correct operation of the divider is that this frequency equathe wanted output frequency n113. So the relation

(4.3

where n, pand q are integers, must be valid. For n = 1 the simplest solutionq = 1 and p = 2, for n = 2 the simplest solution is q :== 2, p - 2.

An alternative solution is obtained if the output signal at frequency nfis mixed with the signal at qf Multiplication of the difference frequency qf·

TRANSMISSION AND DECODING IN COLOUR TELEVISION ,275

Jlf3

Fig. 4-13. Alternative circuit of frequency divider.

1/3 by p must then produce the output signal at nf/3 (see fig. 4-13). Hence thebndition for operation reads

n 'pq = _ (1 +p). (4.31)

3'or n = 1no solution exists, for n = 2 the simplest solution is q = 1,P = 2.

.6.4.2. Phase relations in the frequency divider

We must now consider in some detail the phase relations within the fre-uency divider in order to find conditions for the ruiming-in system, Foronvenience we shall carry out the analysis for a numerical example, to wit,îvision of the frequency by 3 according to the method of fig. 4-12. In order toeparate the wanted and unwanted components in the, divider the employmentf filters between the composing elements is essential. Hence frequency-depend-nt phase errors will occur.

J H

In/)l1t Atndex freg. f

Inl!.ut DRunning-insignal f!3Fig. 4-14. Group delays in frequency divider and running-in circuit.

As in our considerations on phase compensation we shall assume that groupelay is constant over the passband of the filters. Figure 4-14 shows the circuitf the frequency divider including the group-delay times. Omitting first the

~unning-in channel we can write for the signals at the points indicated in the

r,~u::~(wt + 2kn).(k = 0, 1,2,3, ... ).

~

~: cos (wt-.Llw + 2kn). , . ': cos (twt + ç),where ç is the unknown phase shift of the output signal.

~: cos (lwt-.Llw-ç + 2kn).G: cos (twt-.Llw-t."Llw-; + 2kn).IH: cos (twt-2.Llw-t." Llw-2ç + 4kn). "IJ: cos (twt-~.Llw-t." Llwxt."'Llw-2; + 4kn).

276 '. J. DAVIDSE

The found signal at J must be equivalent to the assumed signal at the sapoint, so ç follows from . .

ç = - i.Llro-ÖLlw(-r" + ."') + lkn,Substituting this value in the expression for the signal at point F (the outpsignal), we find

cos [twt-t .Llro +ÖLlw(." + ."')+ Fen]. (4.3

From this formula we can draw two important conclusions. First we nothat the frequency-dependent phase shift is Llw(-i. + Ö." + Ö.'''). Thmeans that the delay within the frequency divider compensates partly for tdelay outside it. This is nöt so surprising; in fact, the frequency divider in itSjforms a phase-compensation system. In designing the main phase-compenstion system in the signal-processing circuit this effect has to be taken inaccount. .

Secondly we note that, as was to be expected, there is an ambiguity as to t~phase of the output signal. Substituting k = 0, k = 1 and k = 2 we finithat there are three possible values for the output phase mutually 12apart. To be sure that the frequency divider starts in the right phase a runninin signal is needed that starts the divider unambiguously. The divider onstarted, the running-in signal is switched off and the frequency divider runs 0in the phase nearest to the starting phase. From the expressions describing tbsignals at points F and J we see that the running-in signal may be both at fIand at 21j3. This signal can be derived by the scanning of special "running-istripes" at the left-hand edge ofthe screen. Such running-in stripes can conve~iently be obtained from the normal index-stripe pattern by omitting one ijevery three stripes. In this way a running-in signal is obtained containing noonly Ij3 but also 21j3 and f, so switching over from running-in operation tnormal frequency division can be smooth.

4.6.4.3. Phase relations during running-in

We shall now consider the phase relations during running-in. We assume thathe frequency divider is started by a signal atlj3, hence the running-in channeis tuned at this frequency. Returning now to fig. 4-14 for the signal in thrunning-in channel at point D we can write cos trot and at point E, cos (twtt.'LIw). During running-in the mixer operates as an amplifier for the running-i~signal, thus at pointFwe also have the signal cos (trot-lr' zlco). Afterrunning-i~the divider will continue in that one of the possible output signals according tal(4.32) which is nearest in phase to the running-in signal cos (twt-t.' LIw).

Let us now suppose that the index frequency changes, for instance, due to ~change in the scanning velocity. According to (4.32) the phase of the outputsignal will then change by -tdro +H." + ."')Llw, while the phase of the


Inning-in signal changes by -lr'LIw. If these two phase shifts differ by a~rtain amount, the frequency divider will switch over to a different value of k) that a sudden phase jump of 120° occurs. To avoid this intolerable situation~viously the relation

-c' =.-c-t(-c" + -c"') (4.33)ust be valid. This means that the delay of the running-in channel has to bess than the delay of the main indexing channel. This condition, which at ast glance may seem strange, is caused by the inherent phase compensation. ailable in the frequency divider.Making the delay in the running-in channelless than in the main channel hasremarkable consequence. If, during running-in only starting signal (at fre-uencyf/3) is available there will be no signal at all available at the input of theequency divider during a time t (-c" + -c'''). When the main index signalrives the divider may then already have "forgotten" the starting signal, soat starting In a wrong phase can occur. For the case of a running-in signal at

.'(/3 considerations can be put forward not essentially different from those given~re. It may be noted that we have based our considerations on the assump-Ionthat the filters produce a pure delay of'the information which is comprehend-in the running-in signal and the index signal. In fact they also introduceoothing of the information in time. By proper design of the circuitry it iserefore possible to ensure that during the "dead time" still enough run-information is available. A better solution is to design the running-in stripeattern such that it produces simultaneously the running-in signal and theain index signal. The described stripe pattern complies with this requirement.

.6.5. Phase errors in the index signal and transient behaviour of the processingcircuit

.6.5.1. Phase errors due to cross-modulationThe information on the position of the electron-spot scanning the screen isomprehended solely in the phase of the fundamental component of the indexgnal. As a consequence any phase error originating during the signal proces-ng leads to misinterpretation of the positional information and hence toolour errors. We shall here deal briefly with some causes of such phase errors.In the first place a fundamental cause of wrong index phase has to be men-oned, i.e. residual cross-modulation effects within the picture tube due to itson-linearity.Let the composite writing signal be represented by

Ew = EM' + Aw cos (wwt + IPw).ue to the non-linear transfer characteristic of the tube the beam current alsoontains harmonics of the writing frequency. H nee, in general the beam ur-ent can be written

278 J. DAVIDSE

1b = ao + al cos (wwt + rpw) +a2 cos (2wwt + 2rpw) + as cos (3wwt + 3rpw)... (4.;

The coefficients ao, al etc. depend on EM', Aw and the transfer characteristicthe tube ..The index signal is found by multiplying (4.34) with the indexing function ~

which describes the scanning of the index-stripe pattern (see sec. 4.4.5). Fortube in which there are 3 index stripes in every two colour triplets we can wri

SI = bo + b i cos IWwt + b2 cos 3wwt + be cos £ Wwt + ... (4.3

As the index amplifier is tuned to IWw, only components in the index signalthat frequency are passed. Hence, the index signal is given by

acb : cos Iwwt + tasbl cos (~wwt + 3rpw) +lasb« cos (~wwt-3rpw) + ... (4.3

The first term represents the wanted indexing signal, the other terms descrilthe cross-modulation of the a.c. components of the writing signal in the indisignal. A more detailed analysis is given in an article by Graham et al. 5), whealso the r~sults of numeri~al calculations are given.It turns out that the resuling phase errors aresmall; they do not exceed roughly 6°..Itmust be noted that similar phase errors may occur in the photomultiplier

this device operates in a non-linear way. This is so because it is essentialimpossible to apply filtering in front of the photomultiplier, the informatiethere being available in the form of a light signal, rather than an electric signs

4.6.5.2. Conversion of amplitude modulation into phase modulationA second cause of phase errors is the conversion of amplitude modulatie

into phase modulation in the photomultiplier and in the index amplifier. Tlindex signal contains strong amplitude modulation because of the modulatieof the beam by the writing signal. Conversion of amplitude modulation in1phase modulation can among other things be due to voltage-dependent inpiand output impedances of amplifying devices. Moreover, it originates fromsecond cause: the index frequency can vary by about 5%. If a deviation from tlnominal index frequency occurs the signal is not at the centre frequency of tlpassband of the amplifier. Hence sidebands originating from the. amplitucmodulation are attenuated asymmetrically so that phase modulation OCCUI

during transients. Finally, the limiter, which has to cope with a very largdynamic range of the signal, can introduce phase shifts. With a careful desigof the index amplifier it is possible to reduce satisfactorily these phase erroidue to the amplitude variations of the signal. .

4.6.5.3. Imperfections ofphase compensationA further important cause of phase errors is imperfect phase compensatior

The phase-compensation system must work over the entire range of possibl

';fRANSMISSION AND DECODING IN COLOUR TELEVISION 279

,equencies of the index signal. However, in the compensation branch the se-~ctivecircuits, like filters, usually differ in structure from those to be compen-I\.tedfor. As a consequence the associated group-delay characteristics are of a

Ifferentnature. In order to overcome these difficulties in practical designork on indexing circuits the phase'relations throughout the frequency rangeif the index signal have to be analyzed thoroughly to find the best possible

tmpro~se on this point. .The conclusion from these general considerations may be that full masteringphase relations in the entire signal-processing circuitry is essential. It mustI noted, however, that we have to do here with a problem of a practical nature,ot with an inherent shortcoming of the indexing system as such.

6.5.4. Transition effects of phase errors,The effect of phase shifts will in the first place manifest itself by erroneouslour rendering, i.e. by hue shifts in coloured areas in the picture. In the secondace there will be an effect on transitions.This can easily be seen with the aid of fig. 4-5. Letus assume that a phase'ft occurs in the input signal. For convenience we assume that the phaseansition is a linear function of time and fully free of ringing. Furthermore wesume as a first-order approximation that passing the signal through a filterly delays the transient and possibly adds to its rise time .

o

.<IQa

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.<I'p

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o

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Fig. 4-15. Phase transitions at various points in the circuit of fig. 4-5.(a) transition at A,(b) transition at B,(c) transition at C,(d) transition at D,(e) transition at E,(f) transition at F.

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, .

J. DAVIDSE280

For the purpose of gaining insight into the basic phenomena these verough approximations may suffice. By way of example we assume a seriesnumerical values for delay times and rise times which are Dot too unrealistiviz. rise time of phase transition at point A in fig. 4-5, 0·25 (J.sec,rise timeB 0·8 (J.sec,at C, .D, l! and F also 0·8 (J.sec;for the group delays: Tgl = O·S(J.seTg3= 0·25 (J.sec,'g2 = 2·1 (J.sec.Figure 4-15 shows the phase transitions at the various points in the CirCUi·

The original phase transition Llp in the indexing signal is of course convertinto a phase transition t Llp in the writing signal. The figures ShO.w clearly ththe final state of the output phase is preceded by a long smear in which tphase shift is Llp instead of t Llp. It will be evident that such long smears aj'very annoying. They are basically caused by the fact that the indexing informtion reaches the gun by two ways which have different delay times. We haseen earlier that signal processing along two separate paths which differ in del~time serves to provide phase compensation. We can therefore conclude that tprice of improving the colour rendering in large areas by the applicationphase compensation is enhancement of phase transitions in the index signaFor different phase-compens~tion circuits (see e.g. fig. 4-8) the smearing e

feet is numerically different, which can easily be seen by applying the constrution of fig. 4-15 for each separate case, but the effect as such is of course alwapresent. The only method of getting rid of t~e smears effectively is therefore titake care that no phase transitions occur in the index signal. This means thin the design of the photomuItiplier circuits, the index amplifier and the frquency divider, great emphasis must be laid upon the prevention of unwantecross-modulation and amplitude-dependent phase shifts.

Eindhoven, May 196

REFERENCES

1) D. S. Bond, F. H. Nicoll and D. G. Moore, Proc. Inst. Radio Engrs 39, 1218-1231951. _

2) D. G. Fink eta!., Television engineering handbook 16, McGraw-HiII, New York, 195

1

pp. 246-253. .'3) B. H. J. Cornelissen and J. Davidse, Dutch Patent Application 247559, 1960.4) E. M. Craemer, USA-Patent 2674651.5) R. Graham, J. W. H. Justice and J. K. Oxenham, Proc. Inst. elect. Engrs 10SB

511-523, 1961.6) R. C. Moore and R. G. Cl app, USA-Patent 2771504.7) A. W. Rihaczek, Proc. Inst. Radio Engrs 4S, 1756-1760, 1960.8) R. L. Miller, Proc. Inst. Radio Engrs 27, 446-457,1939.9) R. L. Fortescue, Journ. Inst. elect. Engrs 84, 693-698, 1939.

10) P. Thiessen, Technische Hausmitteilungen NWDR 7,77-87 and roi-uo, 1955.11) I. T. Turbovich, Radio Engineering (Radiotekhnika) 16, 4, 14-26, 1961.

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