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Recent Advances in Theory and Applications of Substrate-Integrated Waveguides: A Review
Abhishek Sahu(1), Vijay K. Devabhaktuni(1), R. K. Mishra(2) and P. H. Aaen(3)
(1)Electrical Engineering. & Computer Science Department, The University of Toledo, OH 43606, USA.
(2)Electronic Science Department, Berhampur University, Ganjam, Odisha, India.
(3)Advanced Technology Institute, University of Surrey, Guildford, Surrey, UK.
E–mail: [email protected]
Abstract – The use of substrate integrated waveguides (SIW) for microwave and millimeter wave integrated components has increased dramatically over the last decade. They mimic the performance of conventional metallic waveguides and they are fabricated using printed circuit boards using the top and bottom metallisation with two rows of vias forming the side walls. This creates a low profile, compact, and light weight alternative to conventional metallic waveguides, and they allow a direct interconnection with printed circuit boards and active components. This paper reviews the fundamental theory, documents the research that has been performed over the past decade, and summarises progress up to the recent state-of-the-art including novel SIW structures for passive circuits and antennas as well as new applications for reconfigurable and printed circuits using SIW technology.
Keywords–Millimeter–waves, Substrate integrated circuits (SICs), Substrate integrated waveguide (SIW).
1. Introduction
The exploitation of millimeter waves for next generation mobile communications has
fostered significant advances in the development of new high-performance microwave
components to support applications ranging from mm-wave handsets to microwave back-haul [1-
2]. There has also been a considerable interest in a wide range of applications beyond mobile
communications that includes imaging sensors [3], automotive radars [4] and biomedical devices
[5]. All of these applications require microwave/mm-wave components to be integrated into a
multilayer circuit and existing manufacturing techniques, that use precision machining have
difficulties to economically achieve the required precision and they are also difficult to scale for
mass production [2]. To address this, innovative integrated waveguides, such as the substrate
integrated waveguide [6–8], post–wall waveguide [9], and laminated waveguide [10] have been
proposed. These waveguide technologies attempt to maintain the advantages of rectangular
waveguide but are significantly smaller in size, with lower profiles, and are straightforward to
fabricate using existing printed-circuit board (PCB) technology. Substrate integrated waveguide
(SIW) structures are fabricated by embedding two rows of connecting vias or slots in a dielectric
to connect the top and bottom metal planes as shown in Fig. 1. In this way, the non-planar
rectangular waveguide can be made in planar, and the SIWs are then compatible with existing
fabrication techniques
Fig. 1. Illustration of a substrate integrated waveguide (SIW) showing the via arrangement that
forms the side-wall of the waveguide.
such as printed circuit board (PCB) or low–temperature co–fired ceramic (LTCC).
Substrate-integrated waveguides exhibit the propagation and dispersion characteristics
similar to rectangular waveguides. As SIW structures provide a platform to integrate all the
components including active circuits, passive circuits, planar circuitry and even antennas, there
is a possibility of mounting one or more chip-sets on the same substrate. Moreover, there is no
need of transition between elements fabricated with different technologies, thus reducing the
losses and extending the concept of system-in-package (SiP) to system-on-substrate (SoS) [11].
Over the last decade SIWs have been used in design of many microwave components including
post and cavity filters [51–56], power dividers [77], phase shifters [78, 79], attenuators [82],
oscillators [84], various antennas (slot, array and conformal) [101–122], and reconfigurable
microwave resonators, antennas and filters [127–140]. Figure 2 shows the number of
publications using SIWs over the past decade showing the cumulative annual growth. This
paper summarizes the present state-of-the-art and the applications and uses of SIW technology.
2005 2006 2007 2008 2009 2010 2011 2012 2013 20140
500
1000
1500
2000
2500
0 160 330511
748972
12481583
1826
2258
Fig. 2. Cumulative plot of number of publications on SIW technology in last decade (source:
web of science: webofknowledge.com).
2. Recent Progress in Waveguide Structure and Analysis
The original SIW design was conceived in 1998, when it was known as the laminated
waveguide [10-11]. It is also noteworthy that integrated waveguide technique was proposed
much earlier through a patent [12] that was aimed at facilitating the processing of and improving
the productivity of waveguide based microwave components. The design used two rows of
plated through holes to connect the top and the bottom conductors. In each of the rows the
separation between two consecutive holes is smaller than the cut-off wavelength. But it was not
until 2000 when SIW technique was unified with substrate integrated circuits (SICs) [13], which
refers to the synthesis or conversion of any non-planar structures in planar form (such as
rectangular waveguide, coaxial line, non-radiating dielectric waveguide, and image waveguide),
and their full integration with other planar structures such as microstrips and coplanar
waveguides on single or multi-layered substrates. Thereafter, numerous attempts have been made
to formulate design equations for primitive SIW structures to simplify their design. In their
review work, Dr. Wu et. al. [11] provide an elaborate study on the operating principles and
development of different empirical relations to calculate the effective width for SIWs. In this
Section we will discuss recent approaches to design of different SIW structures and review their
operation principles.
A. Operation Principles and Loss Analysis:
Substrate-integrated waveguides have propagation characteristics similar to rectangular
waveguides when the metallic vias are closely spaced; as this minimizes radiation leakage. Their
TEno (n=1,2,…) modes coincide with a subset of the guided modes of the rectangular waveguide.
TM modes are not supported by SIW structures due to spacing between metalized vias [11]. The
metallic posts establish the boundary conditions for electromagnetic waves; as they conductively
connect the surface currents between the top and bottom planes. Various empirical relations
have been proposed relating geometrical dimension of SIW and rectangular waveguide. Two
such relations proposed in [14, 15] are
(1)
(2)
Where d is the diameter of the metal vias, w represents their transverse spacing and s represents
their longitudinal spacing. The effective width enables the SIWs to be analyzed as rectangular
waveguide, thus reducing the design complexity.
An improved closed form formula derived from an analytical method was proposed in [16].
(3)
Very recently, in [17] the authors showed that the accuracies of the above relations depend on
via diameter to via pitch (d/p) ratio. Hence, a new method to design the SIW width , in
terms of its equivalent dielectric waveguide width Wequi, based on the mode matching technique
was proposed in [17].
(4)
Where d and s are the diameter and pitch of the via respectively. The above relation covers all
practical SIW applications for which 0.5< <0.8.
Numerous attempts have been made for rigorous determination of the propagation and
dispersion characteristics of SIW structures based on numerical and modal analysis methods.
Analysis of the propagation characteristics of the EH1 mode of the HMSIW was proposed in
[18]; using method of auxiliary sources (MAS), which is an alternative numerical technique to
the widely used surface-integral formulation, that is suitable for solving elliptical boundary value
problems, which appear in electromagnetic-scattering analysis, antenna modeling and waveguide
structures. Addition theorems employing Bessel and Hankel functions were used to analyze the
full–wave behavior of the SIW devices [19]. In this approach, the equivalent circuit parameters
were extracted using the hybrid mode matching between guided and cylindrical modes.
Interested readers can refer to [17] and [19] to learn more about the mode matching technique.
An approximate analysis of a slotted substrate integrated waveguide with periodically loaded
elements using the transverse technique was proposed in [20]; for designing a travelling wave
attenuator where pin diodes are capacitively coupled to the waveguide slot.
The minimization of losses is a significant consideration in the design of SIWs. There are
three major mechanisms of loss: conductor losses, dielectric losses, and radiation losses [21].
SIWs conductor losses closely resemble those exhibited in rectangular waveguides and they are
primarily affected by the substrate thickness; as the conductor losses depend on the surface
integral of on the metal surface, more specifically on the top and bottom metal surfaces of
the SIW. Hence, increasing h results in a reduction of proportional to , further scaling the
conductor loss proportional to 1/h. Conversely, the dielectric losses depend on the volume
integral of . Since increasing h results in a reduction of proportional to , but linearly
increases the volume of the substrate, so overall there is no effect on the dielectric loss.
Analyzing the quality factor related to dielectric loss shows that it is independent of h and
depends on the substrate dielectric material, but not on the geometry of the SIW [21]. Earlier
research indicated that radiation losses can be reduced [21] if p/d<2.5, with p/d=2 as the
recommended value. Recently, a formula based on a semi–analytical model to calculate the
attenuation constant due to radiation leakage in SIW interconnects has been proposed [22] based
on the decomposition of fundamental mode of SIW into two plane waves. Lateral radiation
leakage can result in crosstalk between adjacent SIWs and in [23], an analytical formula has been
developed for the crosstalk. Considering a case of two adjacent SIW structures sharing common
row of metalized vias, as shown in Fig. 3, the authors formulated the forward crosstalk using
attenuation constant due to lateral leakage (αr), and the attenuation constants due to dielectric and
ohmic losses, αd and αc, respectively.
Fig. 3. Two adjacent substrate integrated waveguide layout for crosstalk determination [23].
The forward crosstalk C can be expressed as:
(5)
Where d, w and s are the diameter, transversal spacing and pitch of the via respectively. N is the
number of apertures and is the wavelength of operation. The number of apertures is directly
related to the SIW as (l= N×s).
The scalar parameters in (5) are extracted through a least square curve fitting and hence,
limits of validity have to be considered while using the same, in terms of operating frequency
and geometrical parameters. Equation (5) has been derived for validity in the single-mode band
of the fundamental SIW mode, according to the nominal band ranges from 1.25 f0 to 1.90 f0 (f0
being the cutoff frequency of the fundamental SIW mode). Finally, the geometrical limits are
directly described by the attenuation constants and SIW length, by the approximation (αr + αd +
αc) l <<1.
The insertion loss, which accounts for conductor, dielectric and radiation losses, can be
significantly affected by surface roughness. Although analytical models of losses due to surface
roughness have been developed for classical waveguides and incorporated in electromagnetic
simulators, no publications have been reported so far for insertion losses in SIW structures [11].
A detailed comparison of losses in SIW structures, microstrip lines and coplanar waveguides is
reported in [24], which shows that SIW structures losses are comparable or lower than those of
planar transmission lines. Furthermore, the recent advancements in waveguide technologies for
millimeter- and submillimeter- wave applications have led to development of new topologies
such as gap waveguides [25], hard-wall waveguides [26], microstrip-ridge gap waveguides [27].
Gap waveguides offer very low loss and manufacturing cost compared to SIWs, since there is no
need of dielectrics or any conductive joint between the top and bottom metal plates, which have
led to investigation of microwave circuits with improved performance over SIWs at frequencies
above 40 GHz. A comparison among the major features between SIW, gap waveguides,
rectangular waveguides and microstrip lines is presented qualitatively in Fig. 4.
Fig. 4. A spider chart showing SIW's performance qualitatively compared to gap waveguide,
ideal rectangular waveguide and microstrip line based on different criteria.
B. Recent configurations of SIWs:
The compactness of SIWs is one of the key attributes considered in designing efficient
commercial mm-wave components; different topologies have been proposed to improve the
compactness of SIW structures. Substrate integrated folded waveguide (SIFW) was proposed in
[28]; where a metal septum is introduced to permit folding of the waveguide, thus, reducing the
size by a factor of two at the cost of slightly larger losses, as illustrated in Fig 5(a). The half
mode substrate integrated waveguide was proposed in [29] based on the approximation of the
vertical cut of the waveguide as a virtual magnetic wall permitting reduction in size of almost
50% , as illustrated in Fig. 5(b).
(a) (b)
Fig. 5. (a) Substrate-integrated folded waveguide [28] (b) Half-mode SIW [29].
A quarter–mode substrate integrated waveguide is proposed in [30], based on the
approximation that centre symmetrical plane of HMSIW can also be regarded as magnetic wall
for several modes, allowing the cross section to be further bisected along the symmetrical plane,
as illustrated in Fig. 6(a); resulting in a size reduction of 75%. A miniaturization method is
proposed in [31], based on the concept of backward wave propagation; where a SIW when
loaded with SRRs shows uniaxial negative transversal permeability and the width becomes less
than half a wavelength at cut–off frequency (Fig. 6(b)). This miniaturized structure also provides
flexibility for tuning the cut–off frequency, provided the SRRs are designed accordingly.
(a) (b)
Fig. 6. (a) QMSIW topology [30] (b) An equivalent infinite parallel plate waveguide (PPWG) containing
the SRRs to analyze proposed SIW loaded with SRRs [31].
However, none of the above topologies focus on longitudinal dimension reduction. As
discussed before, the dielectric loss in a SIW depends on the substrate dielectric constant. This is
one of the reasons, that printed circuit board (PCB) technology is too often used for SIW
component fabrications. Alternatively, when SIW is realized using other fabrication technologies
such as low temperature co-fired ceramic (LTCC) and semiconductor IC’s, the relative high
dielectric constant often makes the width of SIW shrink to a size which is impossible for
practical use at mm-waves. Another important aspect to be accounted for the design of SIWs, is
the ease of integration of active components. Unlike traditional rectangular waveguides, SIW
avoids the requirement of complex mechanical structures, such as diode mount and RF
mechanical coupling between waveguide and active devices; consequently, the manufacturing
cost for these structures is reduced. To address these critical design issues, some SIW topologies
have been developed. The corrugated substrate integrated waveguide (CSIW) utilizing open–
circuit quarter–wavelength microstrip stubs, in place of vias, to form the electric sidewalls of
SIW (Fig. 7(a)) was proposed in [32]. This results in a structure that is completely dc isolated
from the ground plane and permits the shunt connection of active devices. The half–mode buried
corrugated substrate integrated waveguide (HMBCSIW) [33], is almost identical to HMSIW; but
offers a floating top conductor, thereby permitting ease in integration of active devices with an
additional advantage of a mono–mode bandwidth of 3:1. The empty substrate integrated
waveguide (ESIW) proposed in [34], removes the dielectric substrate, and a wideband transition
allows the excitation of the waveguide accessing the microstrip line so that waveguide can be
connected to traditional planar circuits (Fig. 7(b)). This approach enables very low loss circuits
with up to 4.5 times greater quality factor as compared to an equivalent design in SIW.
Extending the approach, a low-loss hollow substrate integrated waveguide (HSIW) was proposed
in [35], where the HSIW is realized by removing the inner dielectric of an SIW and therefore
incorporating a hollow cavity inside (Fig. 7(c)). As this structure incorporates a hollow cavity
inside, the losses again significantly reduce and the structure eases the integration with
rectangular waveguides of equal heights and design of antennas with better radiation
performance. A slow–wave substrate integrated waveguide (SW–SIW) was proposed in [36],
which requires a double layer substrate with a bottom layer including internal metalized via–
holes connected to the conducting ground plane (Fig. 7(d)).
(a) (b)
(c) (d)
Fig. 7. Topologies for (a) Half–mode corrugated SIW [33] (b) Empty SIW [34] (c) Hollow SIW
[35] (d) Slow–wave SIW [36].
Based on a double-layer substrate technology, these SW–SIWs exhibit a significant slow-
wave effect leading to a reduction in both transverse and longitudinal dimensions and thus, a
noteworthy surface miniaturization ratio.
3. Recent Progress in Passive Circuits:
SIW based filters and couplers are extensively documented in the literature. Various filter
topologies have been proposed which include filter with an inductive post, with irises, and cavity
filters with circular and rectangular cavities [7, 37, 38]. Subsequently, dual mode SIW filters
[39–41], wideband SIW filters [42], multi–band SIW filters [43–45], filters based on multi–
layered structures with cavities [46], and compact and super–wide band–pass filters using
electromagnetic band–gap structures [47] were reported.
With respect to passive components, the development of bandpass filters is an essential
component of a communication system. Several attempts have been made to design filters with
substrate integrated waveguides exhibiting a desired passband and high rejection level in the
stopband. Table 1 presents a qualitative comparison of recently published bandpass filters in
SIW technology. In addition to these bandpass filters, several other topologies including
introduction of complementary split–ring resonators (CSRRs) and composite right–left handed
(CLRH) transmission line [55, 56], a triple mode filter on quarter–mode SIW [57], balun filters
based on SIW technology [58, 59], and filters with controllable electric and magnetic coupling
[60, 61] were reported.
TABLE 1 QUALITATITVE COMPARISON OF RECENTLY DEVELOPED SIW BANDPASS FILTERS
Reference Size (mm2) Bandwidth Insertion loss (dB)
Center Frequency(GHz)
Technology
[48] 61 × 32.8 4% 1.8 5.05 Inclusion of planar resonators
63 × 33.13 4.2% 2.05 5.1
[49] 62.36×64.92 5.7% 2.4 5.25 Using frequency dependent coupling
[50] 12×19 10.1% 1.2 4.65 Using HMSIW and avoiding
the need of vias
[51] 30×15 42% <1.1 8.5 U shaped slot etched in SIW cavity to form a multiple mode resonator for wide band
[52] 28.13×28.13 10.9% 1.8 5.5 Hybrid structure of SIW and coplanar waveguide (CPW)
[53](Circular SIW)
1515.55 9.7% 0.29 2.06 Using a triple mode dielectric resonator implemented in SIW
[54](triple passband)
74.73×21.54 4%, 3.4%, 3.3%
0.33, 0.45, 0.3
9.72, 10.76, 11.76
Inclusion of six cascaded singlets in SIW
Fig. 8. The relationship between SIW filter and its substrate characteristics [64].
Very recently, a series of three papers summarizing the various filter topologies, design
considerations and practical aspects of SIW filters, have been published in [62–64]: basic design
rules and fundamental electrical characteristics to indicate the superior performance of SIW
structures are reported in [62]. Advanced design techniques such as cross–couplings realized by
physical and non–physical paths, SIW filters with dual–mode and multi–mode techniques,
miniaturization enabled techniques such as LTCC, wideband and multiband filters have been
presented in [63]. In [64] the authors have summarized a series of practical aspects and design
considerations of SIW filters. Critical aspects such as substrate selection, fabrication tolerance,
thermal stability, and power handling capability are discussed in depth with suitable examples.
The discussion showed that substrate characteristics play a crucial role on the electrical and
physical performance of the SIW filters, as shown in Fig. 8. The size, insertion loss, power
handling, temperature stability and fabrication cost of an SIW filter are all related to the
characteristics of its substrate.
Additionally, several coupler topologies have been developed in SIW. Among them, two
directional couplers were proposed. The first one uses two adjacent SIWs with a common wall
on which an aperture is utilized to realize the coupling between two SIWs working in TW10
mode. This topology was used to design and fabricate 3, 6, and 10-dB couplers [65]. The second
configuration presents a cruciform shape, and was adopted to design a super–compact 3-dB
directional coupler [66]. An efficient design technique for accurate design of wideband couplers
was proposed in [67]. In this design, the even mode propagation constant for tapered slot section
was first accurately extracted with the help of numerical thru–reflect–line calibration method.
Then, it was fitted into the model of a dielectric filled rectangular waveguide and thereafter
extrapolated to the operation range of odd-mode. Based on the extraction of the equivalent
circuit models of the waveguide bifurcation effects along with parametric values, a 90° 3–dB
coupler was developed to validate the approach. Design and fabrication of a 3–dB coupler based
on a hybrid HMSIW–slotline guided wave structure is proposed in [68]. With this approach, the
coupler achieved a relative bandwidth of 17%~22% with 0.5 dB amplitude imbalance. A variety
of ring coupler topologies have been proposed (Fig. 9). Among them, a SIW-ring coupler and
HMSIW ring coupler adopting closed side to closed side designing, for connections of HMSIW
[69] with 24.6% bandwidth for amplitude imbalance, a dual band ring coupler based on CLRH
folded substrate integrated waveguide (FSIW) [70] and a hybrid ring coupler based on compact
ridge substrate integrated waveguide (RSIW) [71] are noteworthy.
(a) (b)
Fig. 9. Layout for (a) Hybrid ring coupler based on HMSIW [69] (b) Lumped-element
equivalent of dual ring coupler unit cell based on CLRH folded SIW [71].
The design of SIW-based power dividers and phase shifters has also received significant
interest from the research community. Development of three configurations of SIW power
dividers were proposed in [77], including a compact radial cavity power divider developed in
[72], a frequency–selective power divider based on single–layer proposed in [73], and a multi–
layer four way out of phase power divider developed in [74]. Two topologies of ring power
divider were proposed in [75, 76]. Also, two configurations of phase shifters were proposed in
[78, 79]. One of these is based on several phase channels made by SIW resonators loaded with
additional metallic posts, to design a phase shifter with 10% fractional bandwidth and the other
design proposes a compact phase shifter using omega particles; thus, providing bandwidths of
around 55% for 90° and 45° versions of the phase shifter, with the accuracy of 3° and 60% with
the accuracy of 2.5°. A broadband magic T [80] and X–band SIW attenuator [81] were also
implemented and experimentally verified. The analysis of loaded SIW attenuators was presented
in [82].
4. Recent Progress in SIW Active Circuits
Complete system on substrate (SoS) integration is possible with integration of active devices
in SIW technology, which is yet to be a focus area of research in this field. The development of
SIW technologies have stimulated the possibility of design and optimization of components like
oscillators, mixers, and switches etc [11]. This section highlights several recent developments of
these components including new SIW-based switch topologies.
The first reported SIW oscillator used a rectangular SIW resonator appropriately placed in
the feedback path between the input and output nodes of an amplifier circuit designed using
Agilent ATF36077 pHEMT transistor [83]. The feedback transmission lines and the input and
output lines of the oscillator circuit were fabricated in microstrip technology and were connected
to the SIW cavity using appropriately designed transitions. This design demonstrated the
fabrication of low cost, high Q, oscillator circuits using SIW technology. Successively, two
oscillators based on Gunn diodes were proposed in [84, 85]. In [85], the authors achieved
frequency tuning for the oscillator by introducing a varactor diode along with a Gunn diode.
These reflection oscillators were designed based on linear simulation techniques in order to
estimate the resonance frequency of the oscillators. Linear simulation techniques are very
intuitive and provide a means for estimating the oscillation frequency using S-parameters
obtained from measurements of the passive resonator. However, as they rely on small-signal
operating conditions, they are unable to estimate the output power of the harmonic content of the
oscillator [11]. Addressing this problem, two X–band oscillators using rectangular cavity were
reported in [86]. In that work, the oscillators were designed by harmonic balance simulation,
where the S-parameters of the cavities were imported from an electromagnetic simulation. A new
technique based on self–compensation of the resonant frequency of SIW cavities was applied to
the oscillator design in [87], in which a nominal ratio of thermal expansion and permittivity
coefficients was defined to generate temperature compensation. In that work, a cavity of specific
substrate was designed, measured and compared to cavities realized on other substrates. Finally,
the cavity was integrated with an amplifier to build an oscillator which showed a stability of 2
ppm/°C in the temperature range of –40 to 80° C. Another Gunn oscillator design based on
HMSIW was reported in [88]. The circuit was composed of a Gunn diode, resonant cavity, direct
current power supply circuit and a transition of HMSIW to microstrip. The HMSIW cavity acts
both as a tuned resonant circuit and an energy coupling device of GUNN diode, coupling energy
from cavity to transmission line. A single–layer cavity backed antenna oscillator was
implemented in [89]. The active circuit and radiator were placed on the opposite side of the
substrate and within the cavity area allowing a compact configuration with 11.87–12.36 GHz
tuning range and phase noise better than –107 dBc/GHz. A push–push and push–pull oscillator
(Fig. 10 (a)) based on SIW and substrate integrated coaxial line (SICL) was presented in [90].
Design of a tunable oscillator based on mechanically tuned resonant cavity was reported in [91]
that allowed varying the oscillation frequency by 2%–3% by means of a flap (Fig. 10 (b)).
(a) (b)
Fig. 10. Schematic of (a) SIW push–push push–pull oscillator [90] (b) SIW oscillator with tuning
element [91].
While, considerable work on oscillators based on SIW technology have been reported in
literature, still there are several areas for future research. For example, phase noise can be
minimized by increasing the unloaded quality factor of the resonator, matching the noise
impedance or achieving a balanced characteristic of the nonlinearities. Similarly, recent
published results show a 2%-3% tuning range through the use of varactor diodes or
mechanically tuned resonant cavities, which can be potentially further improved by inserting
more diodes or introducing various resonant circuits as CSRRs or CLRH topologies. Further,
there is also scope for fabricating low cost frequency sources for high frequency range (100 GHz
or above).
As one basic building block of communication system, mm–wave mixers have widely been
studied, especially with regard to the aspects of port isolation and spurious rejection. Two
singly–balanced mixers, one: X–band single–balanced diode mixer and the other a part of 24
GHz automotive radar system–on–package front–end, were proposed [92, 93]. Further, a
broadband single–balanced mixer using a standard H–plane 3–dB coupler and a novel 90°
broadband SIW phase shifter, was proposed in [94]. The mixer had a conversion loss better than
10 dB across the frequency range 20–26 GHz and LO–IF RF–IF isolations as –55 dB and –45 dB
respectively. Additionally, a single balanced mixer based on a SIW Magic–T structure was
proposed in [95]. The structure showed conversion gain higher than –9.5 dB in the case of
applying –20 dBm RF signal for both high and low LO power configurations at 10.6 GHz, over
the entire RF frequency band of 10–10.55 GHz. Self–oscillating mixers are compact circuits
providing the functionality of both the oscillator and the mixer. They are designed by
appropriately biasing and loading oscillator circuits in order to optimize conversion gain [11].
Two prototypes of self–oscillating mixers were proposed: the first one, a self–oscillating mixer
based on a feedback oscillator that used a rectangular SIW cavity placed in the feedback path of
a field–effect transistor (FET)–based amplifier circuit [96], and the other, an X–band active
antenna self–oscillating down–converter SIW mixer [97], with IF frequency of 3.15 GHz for
conversion gain in at least 600 MHz bandwidth around the IF frequency. As in case of
oscillators, there are several areas where SIW technology can be explored for mixer design.
Some such areas include implementation of mixers in high frequencies (100 GHz and more);
integration of active devices in SIW circuits [11]; new topologies for self-oscillating mixers etc.
Besides oscillators and mixers, switches are essential components that are found in many
microwave applications and a switchable SIW was first proposed in [98]. This research
introduced a switch based on slotted–SIW (SSIW), whose mode of propagation can be switched
between two different modes controlled by integrated pin diodes (Fig. 11 (a)). The switch was
fabricated using microwave laminated SIW, and showed an isolation of approximately 50 dB
over the operating bandwidth with 3-dB insertion loss. In [99], a magnetically controlled switch
based on ferrite–loaded substrate integrated waveguide was proposed. Through an external
transverse magnetic field, the permeability of the ferrite slabs loaded inside the SIW cavity (Fig.
11 (b)) is tuned, thereby making the SIW mode cut–off frequency change, to enable the
switching function. This switch had a 1.1 GHz bandwidth centered at 1.1 GHz with an insertion
loss less than 1 dB in the on state and an isolation of 20 dB in the off state. The design and
implementation of a SIW SPDT switch for X–band applications was presented in [100].
Inductive posts with rectangular slots were embedded in the SIW to control the travelling EM
wave. By adding pin diodes on the top slots of inductive posts, the SPDT switch achieved
isolation (S31) greater than 10 dB, (S32) greater than 15 dB and an insertion loss less than 2.55 dB
in 8.24–10.36 GHz band.
Fig. 11. Layout for (a) Switchable SIW [98] (b) Geometry of a ferrite loaded SIW section [99].
5. SIW Antennas:
The development of millimeter wave technologies has led to a growing interest in SIW
antennas. Several early developed antenna topologies have been discussed in [11]. SIW based
antennas offer advantages similar to microstrip antennas as they can be fabricated with the same
planar printed technology. Furthermore, SIW as a waveguide structure, does not suffer from
unintentional radiation and surface wave loss, which reduces the limitation of thin substrates.
These merits make SIW a better candidate for antenna design. In this Section we will focus on
the recent advancements for several SIW antennas.
A. Slotted antennas:
In this technique, the radiation in SIW antennas is obtained by etching slots in the top metal
surface of the SIW structure (Fig. 12 (a)). The first slotted SIW antenna was based on a four–by–
four slotted SIW array operating at 10 GHz [101]. Recently, many topologies of SIW slot
antennas have been proposed for back lobe suppression. A SIW slot antenna array based on
comb–shaped chokes of an array of quarter–wavelength short–end parallel microstrip lines at the
bottom surface of array was reported for lowering the backward radiation by 10–dB [102]. A
SIW slot antenna with folded corrugated stubs was proposed in [103] for suppressing the back
lobes. Bandwidth broadening of SIW slot antenna was reported in [104]. The work proposed the
use of two unequal slots to increase the bandwidth up to twice of standard slot antennas. Two
design methods for SIW travelling wave slot antennas and planar slotted array antennas were
also reported in [105, 106]. Both design methods are based on the method of least squares and
Elliot’s design formulas using the concept of equivalent waveguide to formulate the error
function. Using the concept of hollow SIW, a slotted waveguide antenna array was designed in
[107]. Using classical log-periodic theory, SIW slot antennas with broadband performance were
presented in [108]. Based on the orientation of the slots a transverse SIW slot antenna and a
longitudinal SIW slot antenna were designed and results indicated a much wider impedance
bandwidth as compared to traditional slotted waveguide antennas.
B. Leaky–wave SIW Antennas:
In the SIW leaky–wave antennas, the radiation is obtained by increasing the longitudinal
spacing s of the side wall metal vias, as shown in Fig. 12(b).Two leaky wave antennas based on
CLRH SIW were presented in [116, 117]. In [116] a double periodic CLRH substrate integrated
waveguide is proposed and the equivalent circuit, the dispersion behavior and the expression for
cut–off frequencies were studied. Two SIW leaky wave antennas composed of periodic set of
transverse slots were proposed by Long et al. [118, 119]. Such structures were found to support
three kinds of modes (a leaky mode, proper waveguide mode, and a surface–wave–like mode).
Furthermore, a SIW leaky wave antenna with tapered transverse slots on top and bottom sides
was proposed by the same group for investigation of end–fire radiation with a narrow beam and
side lobe suppression [120]. The synthesis of one dimensional SIW–leaky wave antennas with
modulated geometry was presented in [121] using holographic concepts; its synthesis technique
demonstrated the capability of flexibly tailoring the radiated field’s pattern, both in near and far–
field regime.
C. Cavity backed Antennas:
Generally, there are two categories of SIW cavity backed antennas: SIW cavity backed
patch antenna (Fig. 12 (c)) and SIW cavity backed slot antenna. Numerous SIW cavity backed
antennas, with different configurations, different feeding structures, and different performances
have been developed by researchers around the globe. In 2013, Dr. X. H. Zhang et. al. reviewed
SIW cavity antennas [109]. Thereafter, some new topologies have also been introduced to this
area. A wideband SIW cavity backed antenna was proposed in [110]. In order to enhance the
bandwidth and radiation efficiency, the cavity was designed to resonate at its TE210 mode.
Another broadband SIW cavity backed antenna was introduced in [111], where the broadband
performance is achieved by introducing a bow–tie–shaped slot instead of a conventional
rectangular slot. A SIW–fed circularly polarized antenna array with a broad axial–ratio
bandwidth was proposed in [112]. The antenna array consisted of 16 sequentially rotated
elliptical cavities fed by slots on the SIW acting as radiating elements, four 1–to–4 SIW power
dividers, and a transition from a coaxial cable to SIW. Two dual band SIW cavity backed
antennas were introduced in [113, 114]: the first one, integrates cavity backed SIW and
triangular slot (TLS), where the dual mode of TLS is achieved by simultaneously exciting the
mode of the slot and the mode of the patch inside the slot; the other one uses a dumbbell shaped
slot along with thin substrate integrated waveguide cavity backing. The SIW–slot feeding
structure is introduced in a fully substrate–integrated thin Fabry–Perot cavity (FPC) antenna to
achieve low backward radiation levels [115].
D. Conformal Antennas:
A significant effort has been devoted very recently to the development of conformal SIW
antennas. There are a few concerns regarding analysis of the SIW conformal antennas and it
requires the synthesis of planar antenna to be carried over to conformal antennas. Unless a
systematic analysis method is developed to study the effect of conformal shape on the properties
of antenna, it is impossible to achieve satisfactory performance or special shaped beam
requirements with conformal antennas because the curved shape of the conformal antenna
determines its characteristics to a great extent [122]. However, some attempts have been made to
come up with accurate synthesis design for SIW conformal antennas. A 35 GHz slot array
antenna conforming to a prescribed curved surface with shaped–beam was investigated in [122],
which had low side lobe level in H–plane and a flat–topped fan–beam in E–plane (Fig. 12 (d)). A
conformal coplanar feed network was designed for the desired excitation. The structure had a –
27.4 dB side lobe level beam in H–plane, and a flat–topped fan beam with –38° ~ 37° 3–dB beam
width in E–plane, along with a cross–polarization lower than –41.7 dB at the beam direction.
Subsequently, a travelling wave slot array on a cylindrical substrate integrated waveguide
(CSIW) at K–band was presented in [123]. A 16 element longitudinal slot array on the broad
wall of CSIW was designed using Elliot’s procedure. For the structure, about 10° beam steering
is achieved when the frequency is swept from 24 to 26 GHz with an antenna gain of 14 dB. The
capability to conform a substrate integrated leaky wave antenna along an arbitrarily curved line
by suitably tapering the leaky mode along the antenna length was demonstrated in [124]. It was
shown that, by means of locally adjusting the pointing angle of the radiated wave, a coherent
plane waveform at the far–field region can be obtained. A conformal wideband SIW H–plane
horn antenna was presented in [125].
E. Active Antennas:
The term active antenna here refers to ‘circuit–antenna module’ and hence it includes active
integrated antennas, where an active device is integrated in the same substrate with the radiating
antenna structure. In addition, it includes antenna elements where an active device is used to
modify or reconfigure the properties of the antenna such as beam direction, polarization or
bandwidth [11]. Compact, single–substrate cavity–backed slot and patch oscillator antennas were
proposed in [126, 127], where a square cavity was used along with antenna and the feeding
networks, etched on the top and bottom layers respectively. Furthermore, a tunable oscillator was
presented in [128] by removing one via hole from the cavity wall and introducing a varactor
diode in its place.
Significant focus has been placed on design of frequency reconfigurable SIW antennas. A
frequency reconfigurable antenna based on HMSIW was presented in [129]. Its resonant
frequency was electronically controlled by a varactor–loaded interdigital capacitor on the top
plate of HMSIW and the bias network was designed on the bottom plate. A reconfigurable SIW
cavity–based antenna was presented in [130] by loading the SIW cavity with shorting posts and
manipulating the field distribution within the cavity. The frequency shift was controlled by the
number of posts and their location. Using this technique a frequency tuning ratio as high as an
octave (1.1–2.2 GHz) was achieved. A frequency reconfigurable SIW interdigital capacitor
antenna on a composite right/left handed (CRLH) resonator was presented in [131]. A varactor
diode was embedded on the interdigital slot of the short–ended SIW, whose zeroth–order
resonance varied from 4.13 GHz to 4.50 GHz by changing the bias voltage from 0 to 36 V. A
varactor loaded complementary split ring resonator (CSRR) was used to design a compact
eighth–mode substrate integrated waveguide antenna [132] operating in dual band. A broadband
frequency tuning was achieved in a ferrite loaded SIW antennas by simultaneously changing the
location of ferrite slabs in the antennas (mechanical tuning) and bias magnetic fields (magnetical
tuning) [133].
(a) (b)
(c) (d)
Fig. 12. Layout for (a) Slotted SIW (b) Leaky–wave SIW (c) Cavity Backed SIW (d) Conformal
SIW [122].
6. Re–configurability with SIW:
Reconfigurable components are essential for millimeter–wave multifunctional radio and
radar systems, such as smart and cognitive radio and radar techniques for better use of the
electromagnetic spectrum as they facilitate the simplification of multiband and wideband
wireless systems architecture [59]. These techniques can eliminate interference while preserving
good signal receiving condition. Through a dynamical reconfiguration of operational frequency
and bandwidth, tunable resonators efficiently cope with time and regional variation of traffic
demands. Hence, they are crucial building blocks for design and realization of tunable RF and
microwave components. One of the preliminary tuning solutions was proposed in [134] based on
the insertion of vertical capacitive posts integrated within SIW cavities. Further, a low–loss
tunable resonator based on combline–SIW cavity loaded with GaAs varactor was proposed in
[135]. A 2.6–3.1 GHz tunable band was obtained with a Qu between 180 and 70, a capacitance
variation between 0.25 and 1.25 pF. An ultra–wideband two–port resonator based on CSRR and
varactor based SIW was presented in [136]. The resonant frequency was varied between 0.83 and
1.58 GHz and has a wide tuning ratio of 90%. Subsequently, electrically tunable evanescent
mode HMSIW resonators were presented [137]; HMSIW loaded with CSRRs where a variable
capacitor connected to one of the conductors of CSRR changes its effective capacitance with
respect to ground, resulting in frequency tuning of the resonator. A novel approach for providing
SIW tunable resonators by means of placing an additional metalized via–hole on the waveguide
cavity was presented in [138]. The via hole contains an open loop slot on the top metallic wall.
The tuning range was defined by the dimension, position and orientation of the open–loop slot.
A novel tunable second–order filter was implemented on three–layer Rogers RT/Duroid
substrate using p–i–n diode switching elements [139]; the filter provided six states ranging from
1.55 to 2 GHz (25% tuning). A two pole filter has been demonstrated on a low–cost substrate
showing a tunable center frequency between 2.64 and 2.88 GHz with 1.27–3.63–dB insertion
loss across the tuning range [140]. Tuning a SIW cavity by embedding a frequency agile material
into an SIW cavity was presented in [141]. A specific switchable post constructed using plasma
(argon) was introduced in the SIW cavity. A compact tunable filter integrated in ferroelectric
ceramic substrate was reported [142] based on an evanescent–mode dielectric cavity loaded by a
pair of tunable–mushroom–type complementary split–ring resonators.
7. System on substrate (SoS):
The use of microstrip or coplanar waveguides is convenient for frequencies below 30 GHz,
but these techniques becomes impractical at high frequencies due to the incurred losses. The
previous sections highlighted the possibility of SIW technology in the development of various
active and passive components. These possibilities provide opportunities for design of system on
substrate (SoS), where all the components not included in chip-set are fabricated using SIW
technology. Two methodologies have been reported for the deployment and integration of SoS.
The complete front end of a 24 GHz frequency-modulation continuous-wave (FMCW) was
reported in [144]. All building blocks including up-converters, down-converters, power dividers,
one transmitting and one receiving 16-element slotted antenna array were integrated in the circuit
(Fig. 13 (a)). This approach showed that the surface hybrid integration scheme enables the
complete integration of planar and non-planar microwave circuits.
The other approach was based on deployment of multi-chip modules. Here, the active
portions of the circuit were implemented in the chipset whereas antennas, filters and other
passive components were developed off-chip on a multilayer substrate (Fig. 13 (b)). This
methodology offers a much simpler design, while exploiting the SIW technology for those
components which cannot be fabricated on-chip. A 60 GHz receiver based on this methodology
in GaAs technology was proposed in [145].
Fig. 13. Demonstration of SoS (a) 24-GHz FMCW radar front end [144] (b) 60-GHz multi-chip
module receiver [145].
8. Fabrication Technologies and Novel Materials
The fabrication is one of the key aspects in design of SIWs, especially for millimetre-wave
operation. PCB techniques is one of the most common technologies adopted for implementing
SIWs as they provide advantages in terms of low cost and good design flexibility. In the early
versions of SIW components, the holes made by a drilling machine and the metal vias were
implemented by using metal rivets. More recently, the metalized holes are usually obtained
either by micro–drilling or laser cutting, and their metallization is subsequently performed by
metal plating or by filling the holes by a conductive paste [146]. Apart from PCB, LTCC
technology has also been used in SIW implementation. The possibility of using several layers
and the tiny dimension of the via holes, provides a platform for implementation of extremely
compact SIW components.
The implementation of SIWs on novel materials have also been explored. The design of
SIW components and antennas based on polyethylene terephthalate was proposed in [145]. In the
work, initially SIW interconnects were designed and fabricated, which finally led to design of a
two cavity filter and a slotted waveguide SIW antenna (Fig. 14 (a)). The design and
implementation of an inkjet printed flexible broadband multilayer coupler on a polyimide film
(Kapton) in a SIW technology was demonstrated in [148] (Fig. 14 (b)). In [149], the flexible
liquid crystal polymer (LCP) was used to implement broadband bandpass filters, that features
7.5% size reduction compared to conventional SIW (Fig. 14 (c)).
(a) (b)
(c)
Fig. 14. Photograph of (a) SIW Filter based on PET substrate [147] (b) Inkjet-printed multilayer
broadband coupler on polyimide film [148] (c) Fabricated 3 pole filter on LCP [149].
Popovic and her research team are exploiting a new area of research on the micro-coaxial
array fed components at the University of Colorado, Boulder. Performance of broadband
Wilkinson dividers covering 2-22GHz (11:1 bandwidth) implemented in micro-coaxial lines
along with integration of active devices with micro-coaxial components for 20W transmitters
covering the 4-18GHz band was shown, and challenges associated with high power densities and
interconnect parasitic are addressed in [150]. Although, this technology requires high precision
and complex fabrication process, it could be one of the research area for SIW based components
operating in high frequencies. This technology provides advantages in terms of low loss, high
isolation, and large range of characteristic impedances.
9. Conclusions
The present state-of-the-art of the SIW technology is summarized in this paper, with a
focus on recent advancements. Design rules that are important for the development of
active and passive SIW circuits are discussed to highlight the attractiveness of this
emerging technology. The SIW technology has been compared with other recently
developed techniques from the view point of different critical aspects such as loss,
packaging and design complexity. In addition to the traditional components, the recently
explored areas in SIW technology such as switches, conformal antennas, have been
discussed. Furthermore, we have found that the attractive features of this technology make
it the ideal tuning platform for reconfigurable filters, antennas and impedance matching
networks. Finally, issues related to fabrication technologies and using novel materials for
SIW based components have been presented.
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