Improved RF Performance of Travelling Wave MR with aHigh Permittivity Dielectric Lining of the Bore

A. Andreychenko,1,2* J. J. Bluemink,2 A. J. E. Raaijmakers,2 J. J. W. Lagendijk,2

P. R. Luijten,1 and C. A. T. van den Berg2

Application of travelling wave MR to human body imaging is

restricted by the limited peak power of the available RF ampli-fiers. Nevertheless, travelling wave MR advantages like a large

field of view excitation and distant location of transmit ele-ments would be desirable for whole body MRI. In this work,improvement of the Bþ1 efficiency of travelling wave MR is

demonstrated. High permittivity dielectric lining placed next tothe scanner bore wall effectively reduces attenuation of the

travelling wave in the longitudinal direction and at the sametime directs the radial power flow toward the load. First, this isshown with an analytical model of a metallic cylindrical wave-

guide with the dielectric lining next to the wall and loaded witha cylindrical phantom. Simulations and experiments also reveal

an increase of Bþ1 efficiency in the center of the bore for travel-ling wave MR with a dielectric lining. Phantom experimentsshow up to a 2-fold gain in Bþ1 with the dielectric lining. This

corresponds to a 4-fold increase in power efficiency of travel-ling wave MR. In vivo experiments demonstrate an 8-foldsignal-to-noise ratio gain with the dielectric lining. Overall, it is

shown that dielectric lining is a constructive method toimprove efficacy of travelling wave MR. Magn Reson Med70:885–894, 2013. VC 2012 Wiley Periodicals, Inc.

Key words: travelling wave; body MRI; ultra high fields;

dielectric lining

Ultra high fields (�7T) provide an increased sensitivityresulting in higher spatially resolved imaging of thehuman body. Nevertheless, a pronounced advantage of7T MRI has only been shown for neuro applications(1,2), whereas whole body MRI at ultra high fieldsremains restricted to a research setting. At ultra highfield, MRI image quality is degraded due to several limit-ing physical and safety factors. This complicates a rou-tine application of ultra high field whole body MRI in aclinical setting (3,4). First of all, a strong interaction ofradio frequency (RF) signals and human tissue occurs atultra high fields because the RF wavelength becomessmaller (e.g., 14 cm at 7T in brain) than the size of thehuman body. The resulting RF interferences cause a

heterogeneous RF signal distribution in the human bodydegrading image quality (5), whereas constructive elec-tric field interferences can lead to regions of elevatedlocal high specific absorption ratio (SAR) (6). Addition-ally, it becomes more technically challenging to con-struct a proper functioning coil, e.g., parts of an RF coilwith large physical dimensions become self-resonant (7).Moreover, the close proximity of the conducting ele-ments and body inevitably exposes human tissue tostrong electric fields and as a result, causes high SARlevels in the body parts close to the coil (4). Therefore,distant and more distributed transmit elements would bedesirable to avoid these high local SAR depositions.

Travelling wave MRI can be achieved by exploiting thewaveguide action of the whole-body MR scanner at thesmall RF signal wavelength for proton MRI at 7T (8,9).Above the so-called cut-off frequency the scanner borestarts to act as a metallic cylindrical waveguide and theRF signal (emitted by a distant antenna at the bore’s end)propagates along its longitudinal axis as an electromag-netic wave. One of the attractive advantages of travellingwave MR is that there is no need for a large volume coilwith lengthy conductors (which can self-resonate at highfields) to transmit and receive RF signals but a distantantenna is used instead. In a waveguide, travelling wavepropagation occurs in the form of modes characterized bytheir own distinct electromagnetic field patterns and cut-off frequencies. Both features depend on the waveguidegeometry. In an empty 60-cm-diameter scanner bore, thefirst transverse electric mode (TE11) has a cut-off fre-quency around 300 MHz (proton Larmor frequency at 7T),and thus, in 7T travelling wave MR, the TE11 mode elec-tromagnetic field pattern is propagating in the scannerbore (8). The RF signal in the travelling wave MR is dis-tributed over the whole cross-section of the scanner boreand, consequently, excites a large volume. Therefore, thewhole human body in the scanner bore is exposed to theRF signal. Since the body is conducting it inevitably leadsto high power losses. As a result, relatively large powersare needed to achieve an acceptable level of Bþ1 (10,11).These factors make the travelling wave technique as itwas proposed by Brunner et al. (8) not a very efficienttechnique for body MRI at 7T.

High dielectric constant materials have been demon-strated to significantly reduce standing wave effects inthe RF field distribution when placed next to the bodyor head (12–14) or in the coil-to-shield space (15). Thiswork proposes to use high permittivity dielectric liningof the scanner bore to improve in vivo RF performanceof travelling wave MR imaging. The high dielectric con-stant material is placed on the wall of the bore instead,leaving a large air space between the subject and high

1Department of Radiology, University Medical Center Utrecht, TheNetherlands.2Department of Radiotherapy, University Medical Center Utrecht, TheNetherlands.

*Correspondence to: A. Andreychenko, M.S., Department of Radiology andRadiotherapy, University Medical Center Utrecht, Q.04.4.303, PO Box85500, 3508 GA, Utrecht, The Netherlands. E-mail: a.andreychenko@umcutrecht.nl

Received 2 June 2012; revised 15 August 2012; accepted 8 September2012.

DOI 10.1002/mrm.24512Published online 8 October 2012 in Wiley Online Library (wileyonlinelibrary.com).

Magnetic Resonance in Medicine 70:885–894 (2013)

VC 2012 Wiley Periodicals, Inc. 885

permittivity material. The dielectric lining and the boreform a closed waveguide with a reactive wall (16). Inwaveguide literature such a dielectric lined waveguide isknown as a low attenuation waveguide (17–19). Toinvestigate the effect of dielectric lining to improve trav-elling wave MR efficiency, the modes in the scannerbore with a load and with the dielectric lining werestudied with an analytical model for a range of relativepermittivity values of the dielectric lining. The modelshowed that the relatively high permittivity dielectric lin-ing increased Bþ1 field strength in the load. There are twofactors that contribute to this Bþ1 gain. First, the dielectriclining reduces losses of the wave. Second, the permittiv-ity of the dielectric lining modifies transmission proper-ties of both dielectric lining and air layers and, as a con-sequence, alters the radial power flow from the wall tothe load. By optimizing the relative permittivity value ofthe dielectric lining, the transmission of radial power tothe load is maximized resulting in higher Bþ1 values. Toimplement the dielectric lining in practice, an array ofplastic tubes filled with distilled water was assembled.Numerical simulations of the setup including a humanload were performed before the experiments. MR experi-ments were performed with a phantom and in vivo. Inboth cases, the dielectric lining showed the improvementof Bþ1 efficiency of the travelling wave when the dielectriclining was placed in the scanner bore.

METHODS

Electromagnetic mode behavior in a metallic waveguidewith dielectric lining was investigated in this work. Forthe analytical model, the scanner bore (loaded with aphantom and with dielectric lining) geometry was sim-plified to three concentric, cylindrical layers, namely thecylindrical core resembling a load (human subject orphantom), the air spacing, and the dielectric lining nextto the metallic wall. A similar approach was describedby Foo et al. (15,20) to design RF resonators for 1.5 and4.0 T MR imaging. Their model had two concentric me-tallic boundaries: RF shield and RF coil. In our case,only the RF shield is present in the model, and the RFsource (antenna to launch the travelling wave) isexcluded from the model because of the large distancebetween the antenna and the imaging area in the travel-ling wave MR.

Analytical Three-Layered Closed Waveguide Model

The scanner bore with the dielectric lining and loadinside was modeled as a three-layered concentric closedcylindrical waveguide. The core of the waveguide repre-sented a load (phantom), followed by a layer of air and,finally, the dielectric lining. Figure 1a illustrates themodel. The waveguide’s metallic wall was modeled as aperfect electric conductor. The waveguide was assumedto be infinitely long to exclude the effect of reflectionsfrom the waveguide’s ends. In a source free region, thepropagating electromagnetic modes can be found as solu-tions to the homogeneous Helmholtz equation. For acylindrical geometry, the longitudinal components of theelectric field (Ez) and the magnetic field (Hz) in each

layer can be expressed as a linear combination of Besselfunctions of the first (Jm) and second (Ym) kind (21):

Ez ¼ An;1 2;3ð ÞJm qn;1 2;3ð Þr� �

þ Bn;1 2;3ð ÞYm qn;1 2;3ð Þr� �� �

� sin muþ hmð Þ � ejvt�jbz ð1Þ

and

Hz ¼ Cn;1 2;3ð ÞJm qn;1 2;3ð Þr� �

þ Dn;1 2;3ð ÞYm qn;1 2;3ð Þr� �� �

� cos muþ hmð Þ � ejvt�jbz; ð2Þ

where

q2n;1 2;3ð Þ ¼ v2e0e1 2;3ð Þm0m1 2;3ð Þ � b2 ½3�

is the radial wave number in each layer, the indices1,2,3 indicate the layers, m and n are the mode indices,v is the angular RF frequency, b is the longitudinal wavenumber (the common wave number in three layers), e1,2,3

are the relative permittivity values of each layer, andm1,2,3 are the relative permeability values of each layer.Bn,1 and Dn,1 are set to zero to assure finite fields in thecenter (r ¼ 0) of the waveguide. The radial (Er, Hr) andazimuthal (Eu, Hu) components of fields can beexpressed in terms of the longitudinal components (Ez,Hz). The expressions are well known and listed in manytext books, e.g., (22). Here we skip them for brevity. Asystem of 10 linear homogeneous equations was formedbased on the boundary conditions: eight equations fromthe continuity of the tangential electric and magneticfields across the dielectric–dielectric boundaries betweencore and air layer and between the air and the dielectriclining; two equations from setting the tangential electricfields to zero at the dielectric lining-metallic wall bound-ary. This system of 10 linear homogeneous equationshad eleven unknowns: 10 unknowns are the fieldsamplitudes: An,1(2,3), Bn,(2,3), Cn,1(2,3), Dn,(2,3) and the elev-enth unknown is the longitudinal wave number b:

M bð Þð Þ �An;...

Bn;...

Cn;...

Dn;...

0BB@

1CCA ¼~0; ½4�

where (M(b)) - is a 10�10 matrix. The only case inwhich a system of homogeneous equations additionallyadmits a set of nontrivial solutions (nonzero electromag-netic fields) is when M is a singular matrix, i.e.,

M bð Þj j ¼ 0: ½5�

Therefore, for every propagating mode in the three-lay-ered waveguide the equation should be solved to findthe longitudinal wave number (b) of the mode. In thiswork, only modes with m ¼ 1 azimuthal dependencewere investigated because a sinusoidal azimuthal de-pendence was impressed by the excitation antenna usedin the actual MR experiments. The equation was solvednumerically by a grid search method on a complex planeof longitudinal wave number (b) values for each relativepermittivity value of the third layer. The relative dielec-tric permittivity of the third layer (dielectric lining) was

886 Andreychenko et al.

varied from 2 up to 56 in 10 steps and a set of propagatingmodes with their longitudinal and radial wave numberswas determined for each relative permittivity value. Theelectrical conductivity of the third layer was 10�3 S/m.

To calculate the resulting electromagnetic fields in thewaveguide for every mode, the longitudinal and azi-muthal sinusoidal surface current densities wereimposed on the outer metallic wall of the waveguide,because the sinusoidal dependence is setup by the patchantenna used for the waveguide excitation in practice:

Hz;3 ¼ � ~J s

� �u

½6�

Hu;3 ¼ � ~J s

� �z

½7�

In this way, a system of nonhomogeneous equations wasformed. This system of nonhomogeneous equations wassolved to obtain amplitudes of electromagnetic fields.Subsequently, for simplicity the fields of each modewere normalized assuming the same power was carriedby each mode. However, in reality efficiency of eachmode excitation depends on the RF probe used to excitethe waveguide. The fields were weighted with theirattenuation constants (imaginary part of b) assuming 1 mdistance from the source region. Finally, the normalizedand weighted fields of all the present modes for thegiven relative permittivity of the dielectric lining weresuperimposed. A Bþ1 field average over the phantomcross-section was determined for different dielectric rela-tive permittivity values of the dielectric lining.

Considering the transmit efficiency of MR experiment,the Bþ1 field in the center region of the waveguide, where asubject or phantom is located, is the most important.

Because of the high RF frequency (298 MHz) and the dis-tant location of the RF probe used to excite the waveguide,we operate outside the near field regime (23). Therefore,power flow (described by Poynting vector) drives thebuildup of the magnetic (and electric) fields. There are twocomponents that are of direct importance for the accumu-lation of the fields in the centre: (1) the longitudinal powerflow in the center layer(load) which is heavily attenuateddue to conductive losses in the load and (2) radial powerflow from the peripheral regions to the center.

In a metallic cylindrical waveguide, the dielectric lin-ing next to the wall creates a reactive surface that sup-ports surface longitudinal, electromagnetic power flow(16). Outside this reactive surface, in the air layer, elec-tromagnetic fields are evanescent (their radial wave num-bers have a nonzero imaginary part). However, theseevanescent fields may still couple into the core layerwhere load is located and create sufficiently high mag-netic fields there (24). This coupling of the evanescentfields of the reactive surface to the center layer isdefined by the transmitting and reflecting properties ofthe air layer. These transmitting/reflecting properties ofthe air layer are a combined effect of refractions at thetwo interfaces (dielectric lining/air and air/load) and ra-dial attenuation of the evanescent field in the air layer.Thus, to optimize the magnetic field in the load oneshould alter the transmitting and reflection properties ofthe waveguide air layer to maximize the radial powerflow of electromagnetic waves from the waveguide’s wallto its center (22,23). Based on the radial and longitudinalwave numbers of the dominant mode (mode with thelowest attenuation for a given relative permittivity valueof the dielectric lining and/or the highest contribution tomagnetic fields in the core layer) in the three layers,

FIG. 1. a: 2D model of the three-layered cylindrical metallic waveguide. The external wall of the waveguide is a perfect electric conduc-tor. The wall is coated with a 5-cm-thick dielectric lining with variable relative permittivity. In the core of the waveguide, the phantom is

located. There is an air layer of 13 cm between the dielectric lining and the phantom. b: The scanner bore with the designed dielectriclining inside. The dielectric lining was formed with 32 PVC tubes filled with distilled water. The tubes were arranged along the circumfer-

ence of the bore and supported with four wooden frames (inset). c: The simulated and experimental setup with the dielectric lining. Avolunteer placed in the scanner bore together with the water tube array. The patch antenna is placed at the end of the bore. [Color fig-ure can be viewed in the online issue, which is available at wileyonlinelibrary.com.]

Dielectric Lining to Improve Travelling Wave MR 887

resulting from solving Eq. 3, the transmitting and reflect-ing properties of the middle layer (air) were investigated.The power reflection and transmission coefficient(http://www.ece.rutgers.edu/~orfanidi/ewa, p. 163) ofthe air layer were calculated for each relative permittiv-ity value of the dielectric lining. The spatial distributionof the radial component of the Poynting vector (powerflow) was calculated for a number of relative permittivityvalues to visualize the radial power flow through thewaveguide cross-section.

MR Experiment

MR experiments were performed on a Philips 7T Achievaplatform scanner (Philips Healthcare, Cleveland, OH,USA). A patch antenna was used to transmit and receiveRF signal. The patch antenna consisted of a circular patch(diameter 35cm) on a dielectric substrate (Acrylic glass, 3cm thickness), backed by a square ground plane (42 � 42cm2). The circular patch and ground plane consisted of 1-mm-thick brass sheets. The antenna had two orthogonalports: one vertical and one horizontal. The ports werelocated at equal radial positions (7 cm radius) to match theports to 50 Ohm in free space. External matching networkswith variable capacitors were created for both ports tomatch their input impedance to 50 Ohm when the antennawas placed in the scanner bore. The horizontal port wasused in the experiments. The port was driven with 4 kWpeak power (excluding cable loss) through a single channelTx/Rx switch box. The antenna was placed at the patientend of the scanner bore. Two travelling wave setups werecompared: with and without the dielectric lining of thebore. Presence of the dielectric lining altered the imped-ance at the patch antenna port and the external networkwas used to match the port impedance to 50 V.

The dielectric lining of the bore was realized in practiceby 32 PVC tubes (Ø 4 cm, length 120 cm, thickness 0.2cm) filled with distilled water (relative permittivity 80).The water tubes were densely packed along the scannerbore circumference and supported with a wooden frame(Fig. 1b). The effective relative permittivity of this tubestructure was calculated (25) to be 3.5 (Maxwell-Garnetmixing rule), the average relative permittivity was 46.

For the phantom measurements, the two setups wereloaded with a cylindrical water phantom (diameter 25 cm,length 26 cm) containing 3g/L saline solution. For in vivoexperiments, a healthy male volunteer (height 178 cm,weight 75 kg) was placed with the head toward the patchantenna (Fig. 1c). Written informed consent was given bythe volunteer before the examination. RF safety of the invivo experiment was evaluated with the FDTD simula-tions. Low flip angle gradient echo images were acquiredin three orthogonal planes (TE ¼ 1.7 ms, TR ¼ 200ms(phantom) and 150 ms (in vivo), acquisition voxel 3 � 3 �5 mm3). The RF signal was transmitted and received withthe horizontal port of the antenna. The field-of-view incor-porated the water tubes in phase encoding direction toavoid signal aliasing from the water tube array. Absolutesignal intensities in the gradient echo images were dividedby the noise to calculate signal-to-noise ratio (SNR) maps.The noise was determined as the standard deviation ofimage intensities in an air region free from artifacts.

FDTD Simulations

A finite difference time domain (FDTD) method (SEM-CAD X64 v14.2, Speag, Zurich, Switzerland) was used tosimulate the electromagnetic field distribution in thebore of the MRI scanner. The patch antenna was mod-eled as it was built in practice: a circular patch (diameter35 cm, perfect conductor) on a dielectric substrate (thick-ness 3 cm, relative permittivity 2.5, el. conductivity 10�3

S/m) backed by a square ground plane (42 � 42 cm2, per-fect conductor). The male model ("Duke") from the "Vir-tual Family" (26) was used to simulate the in vivoexperiment. The model was scaled to fit the shouldersinside the water tube array that corresponded to theactual experimental setup. The simulations were per-formed without and with the water tube array (Fig. 1c).The water tube array was modeled similar to the experi-mental design: plastic tubes filled with distilled water(relative permittivity 80, conductivity 10�3 S/m). Thewooden support was not modeled. For the in vivo case,the FDTD simulations were also performed with zeroconductivity of the antenna’s substrate.

The simulated domain size was 160 � 160 � 293 cm3.At the domain’s edges, perfectly matched absorptive boun-daries were imposed. Adaptive gridding was applied andindividual voxel size varied between 0.2 � 0.2 � 0.2 cm3

and 3 � 3 � 3 cm3 with �20 million cells in total. Har-monic excitation of a soft voltage source was performed at298 MHz. Voltage and current sensors were placed at thesource location. The simulations were considered to bestable if both the voltage and the current amplitudesremained constant over at least 30 periods at the end ofthe simulation cycle. Calculated electromagnetic fieldswere normalized to 1 W power delivered to the domain. Apower balance was evaluated over the whole domain tocheck that all the power delivered to the source was dissi-pated either in the human model, lossy materials of thebore and in the absorbing boundaries at the domain edgesand not reflected back to the source.

RESULTS

Model of the Three-Layered Waveguide

The three-layered waveguide model showed that therelative permittivity of the dielectric lining modifiedtransmission and reflection properties of the second (air)waveguide layer. As shown in Figure 2a, a gradualincrease in transmission and decrease in reflectionoccurred when the relative permittivity of the dielectriclining rose until value around 15. Above this value, thetransmission of the air layer started to decline slightly.Between 30 and 34 a strong transition in the transmis-sion/reflection plot occurred: a drop and a followinggain which resulted in a maximum of the transmissionfor a relative permittivity value of 34. For higher permit-tivity values, the transmission decreased slightly butstayed higher than the initial transmission in the lowpermittivity (<30) range. The increase of the transmis-sion properties of the air layer resulted in a higher radialpower flow toward the waveguide center. Figure 2bshows the real and imaginary parts of the longitudinalwave number for the dominant mode with respect to the

888 Andreychenko et al.

relative permittivity of the dielectric lining. The dielec-tric lining gradually reduced attenuation of the propagat-ing wave (imaginary part of the longitudinal wave num-ber) and decreased the longitudinal wavelength (inverseof the real part of the longitudinal wave number).Between the relative permittivity values of 30 and 34, atransition of both real and imaginary parts was presentthat indicated a change of the dominant mode. Figure 3bshows the radial Poynting vector within the waveguidecross-section for a number of relative permittivity valuesof the dielectric lining. The electromagnetic penetration

into the center of the waveguide (where the phantomwas located) increased considerably with the relativepermittivity rise (due to increased transmission of the airlayer). The maximum radial power flow to the centerwas achieved with a relative permittivity value of 34.The highest radial power flow into the phantom resultedin the maximum Bþ1 field in the phantom (Fig. 3a). Theaverage transverse Bþ1 field in the phantom rose whenthe relative permittivity value was increased, with amaximum (4-fold gain in comparison with the lowest rela-tive permittivity) around a relative permittivity of 34 (Fig.

FIG. 3. a: Relative transverse B1þ field in the phantom for several relative permittivity values of the dielectric lining: 2, 13, 23, 34, 45,

56. B1þ field rises in the phantom with the increased radial power flow. In contrary to the altering radial power flow distribution, the B1

þ

pattern does not change with the relative permittivity; b: Radial component of the Poynting vector in the waveguide’s cross-section for

several relative permittivity values of the dielectric lining: 2, 13, 23, 34, 45, 56. The radial power flow toward the waveguide center isnegligible for low relative permittivity of the dielectric lining. However, it increases considerably for higher relative permittivity values and

has its maximum at a relative permittivity of 34. A reduced radial flow toward the center is observed for higher relative permittivity val-ues but it is still much higher than for the low relative permittivity dielectric lining. Whereas the radial power flow distribution in the cen-ter layer stays similar for all relative permittivity values it changes drastically in the air layer. [Color figure can be viewed in the online

issue, which is available at wileyonlinelibrary.com.]

FIG. 2. a: Reflection and transmission coefficient of the air layer between the phantom and dielectric lining depends upon the relativepermittivity value of the dielectric lining. The air layer becomes more transparent for the electromagnetic wave when the relative permit-tivity of the dielectric lining rises from 2 until 15. Above 15, the transmission slightly drops. Between relative permittivity values of 30

and 34 a strong transition of the transmission occurs. The maximum transmission of the air layer is around a relative permittivity valueof 34 of the dielectric lining. The strong transition in the transmission graphs coincides with a modulation in the real and imaginary parts

of the longitudinal wave number b. As the relative permittivity value of the dielectric lining rose from 2 to 30, the attenuation of the prop-agating wave (imaginary part of the longitudinal wave number) and the longitudinal wavelength (inverse of the real part of the longitudi-nal wave number) decreased. Between relative permittivity values of 30 and 34 a strong modulation of the longitudinal wave number

occurred (a sudden drop of the real part and rise of imaginary part) that indicated a change of the dominant mode. For higher relativepermittivity values, the real part continued to grow and imaginary parts continued to decrease.

Dielectric Lining to Improve Travelling Wave MR 889

4). The change of the dominant mode around relative per-mittivity values between 30 and 34 for the dielectric liningresulted in a strong modulation of the Bþ1 gain around thispermittivity value. A significant drop of Bþ1 occurredaround a relative permittivity value of 29 because the firstdominant mode became confined to the dielectric liningwith a very limited coupling to the core layer. Between rel-ative permittivity values of 30 and 34, another modebecame dominant with a stronger coupling to the core layerthat resulted in a maximum gain of Bþ1 . For relative permit-tivity values above 34, a drop of Bþ1 gain occurred (to 2.5-fold gain; Fig. 4). Interestingly, whereas the radial Poyntingvector distribution altered with the relative permittivityvalue and despite a change of the dominant mode, thetransverse Bþ1 pattern in the phantom was preserved. Theasymmetry of the Bþ1 pattern in the phantom can bereduced if a second orthogonal mode with a 90 degreephase shift is excited in the waveguide. For relative permit-tivity above 34, the radial power flow into the phantomdecreased and the Bþ1 in the phantom was also lowered.Remarkably, for 56 relative permittivity value radial powerflow was confined mainly to the air region and did not pen-etrate into the phantom due to strong reflections of thephantom/air interface. However, the residual leakage of theradial power flow into the phantom was sufficient to createreasonable level (70% of the maximum) of Bþ1 in the phan-tom and still higher Bþ1 values were obtained in the centerin comparison with the low relative permittivity. Note thatthe average relative permittivity of the designed water tubearray used as the dielectric lining in the simulations andMR experiments was around 46 and is slightly higher thatthe optimal value according to the analytical model.

FDTD Simulations

FDTD simulations were performed to evaluate the Bþ1efficiency and the RF safety of the two set-ups: withoutand with the dielectric lining (water tube array) in place.The presence of the dielectric lining increased signifi-

cantly the Bþ1 field in the head as was predicted by the2D waveguide model (Fig. 5a). For the head center, a 3-fold gain of Bþ1 field strength occurred with the dielectriclining. Similar to the modeled Bþ1 patterns in the phan-tom, the in vivo Bþ1 patterns were not altered with thedielectric lining. It indicates that also in vivo the samemode was excited and propagated without and with thewater tube array in the scanner bore.

Presence of the water tube array increased travellingwave power delivered to the head by raising both longi-tudinal and radial power flows (Fig. 5b,c). Approxi-mately, a 5-fold rise of the longitudinal power flowoccurred in the middle of the scanner bore with thewater tube array (Fig. 5b, included conductivity loss inthe antenna’s substrate). However, the simulation of theempty bore (without the water tube array) and zero con-ductivity of the excitation antenna’s substrate showedcomparable longitudinal power flow strength to the sim-ulation with the water tube array and nonzero conductiv-ity of the antenna’s substrate (Fig. 5b, excluded andincluded conductivity loss in the antenna’s substrate).This demonstrated that the 5-fold gain in longitudinalpower flow with the water tube array was mainly associ-ated with reduced conductivity losses in the excitationantenna’s substrate. These lower losses in the antennaindicated a better coupling of the antenna to the borecontaining the water tube array.

The water tube array altered power flow distributionin the bore space from the antenna to the body by creat-ing a reactive surface next to the wall that partially sup-ported longitudinal power flow. This resulted in a rela-tively high power flow in the water tube array beyondthe shoulders. Without the water tube array, longitudinalpower flow was focused in the center of the bore anddecayed towards the bore wall. As it was predicted withthe model, presence of the dielectric lining increased ra-dial power flow toward the center of the bore where thehead was located (Fig. 5c, green arrows). The water tubearray increased penetration of power flow into thehuman body, whereas without the water tube array thepower flow stayed mainly in the air spacing between thehuman body and the bore wall that is clearly seen in Fig-ure 6c (excluded conductivity loss in the antenna’ssubstrate).

The peak 10g-averaged local SAR (peakSAR10g)increased when the water tubes were placed in the borefrom 0.05 to 0.45 W/kg. However, the (Bþ1 )2/peakSAR10g

ratio for the middle brain region did not alter signifi-cantly when the water tube array was used: 0.28 mT2/W/kg without the water tube array and 0.27 mT2/W/kg withthe water tube array.

MR Experiment

Phantom Measurements

The dielectric lining significantly increased the SNR inthe travelling wave MR phantom experiments (Fig. 6).For some areas, a 4-fold gain was observed. Assuminglow flip angle conditions (signal is proportional to theproduct of Bþ1 and B1

�), approximately a 2-fold Bþ1 fieldincrease would correspond to this SNR gain. Similar Bþ1patterns were established in transverse, coronal and

FIG. 4. Average B1þ in the center layer (phantom) of the three-

layered waveguide, depending on the relative permittivity of thedielectric lining. The B1

þ was normalized to the average B1þ value

in the core layer for the lowest relative permittivity of the dielectric

lining. The maximum 4-fold gain occurred for the relative permit-tivity between 30 and 34 that relates to the maximum radialenergy flow to the center layer. Around a relative permittivity of

30, a strong modulation of B1þ field is present due to the change

of the dominant mode.

890 Andreychenko et al.

sagittal planes with and without water tube array. How-ever, when the dielectric lining was present, longitudinalcoverage was improved because of the reduced losses ofthe travelling wave.

In Vivo Measurements

Figure 7a demonstrates the great advantage of the dielec-tric lining. In vivo image quality with the water tubearray benefited from the increased SNR. Signal inten-sities showed distributions similar to the calculatedB1þ*B1

� patterns (Fig. 7b): a strong amplification in thetemporal lobe and neck and signal void in frontal andparietal lobes. Presence of the dielectric lining raisedmeasured SNR by a factor of 8, especially in the head

center. The calculated B1þ*B1

� patterns predicted similargain (Fig. 7, bottom).

DISCUSSION

The relatively novel concept of travelling wave MRI hasnot yet been routinely applied for in vivo human MRimaging. The high power that is demanded to achieve anacceptable flip angle is one of the limiting factors. How-ever, the relative simplicity of the setup and the largeimaging coverage of travelling wave MR are very attrac-tive. Here, we have demonstrated that waveguide princi-ples from microwave engineering could be adopted toelevate Bþ1 efficiency of travelling wave MR. In thiswork, we used dielectric lining of the waveguide wall toreduce wave attenuation and simultaneously increase

FIG. 5. FDTD calculated B1þ fields in human head (a) and Poynting vector distributions in the bore in coronal plane (b: longitudinal

component, c: radial component) without and with the water tube array normalized to 1W input power. The water tube array (average

relative permittivity of 46) increases B1þ efficiency of travelling wave MR significantly (for some regions a 3-fold gain in B1

þ occurs). Thegain in B1

þ is associated with increased power delivered to the head by means of both longitudinal (b) and radial (c, green arrows indi-cate direction of total Poynting vector) power flows. Remarkably, longitudinal power flow in the bore strongly depends on the coupling

of the antenna to the bore (b, excluded and included conductivity of the antenna’s substrate). The water tube array improves couplingof the antenna to the bore that reduces losses in the antenna substrate and boosts longitudinal power flow (b, included conductivity of

the antenna’s substrate). Next to it, the water tube array directs the flow inside the body whereas in case of the empty bore only very lit-tle power penetrates into the body independently from the antenna’s substrate conductivity (c). [Color figure can be viewed in the onlineissue, which is available at wileyonlinelibrary.com.]

Dielectric Lining to Improve Travelling Wave MR 891

the radial power flow from the wall region to the core ofthe waveguide where the load/subject is located. Adielectric lining with relative permittivity above 30 effec-tively reduced reflection and increased transmissioncoefficients of the air layer between the bore’s wall andthe subject. This resulted in an increased Bþ1 fieldstrength in the subject. The simultaneous, strong modu-lation of the transmission properties of the air layer andthe longitudinal wave number between permittivity val-ues of 30 and 34, indicates a change of the dominantmode in the three-layered waveguide. The Bþ1 in the corelayer followed the trend of the transmission properties ofthe air layer and also showed the strong modulationaround 30. However, some deviations were present prob-ably due to the mutual effect of the transmission of theair layer, longitudinal attenuation and differencebetween the dominant modes for the permittivity valuesbelow and above 30. Comparing RF safety of the setupswith and without the dielectric lining, they showed sim-ilar (Bþ1 )2/peakSAR10g ratios. Thus, the dielectric liningis not advantageous if RF safety is concerned. However,it is favorable when little RF peak power is availablewhich is typically the case for ultra high field MR.

Increase of Bþ1 with the dielectric lining originatedfrom two effects: improved wave penetration into thebody by means of radial power flow and gain in the lon-gitudinal power delivery from the antenna to the body.The dielectric lining modified the power flow and par-tially directed it into the body, whereas without the lin-ing the power was flowing in the air spacing betweenthe body and bore wall with very little penetration intothe body (Fig. 5b,c). The dielectric lining of the scannerbore reduced attenuation of the wave and increased lon-gitudinal power flow. The wave cut-off frequency in theempty bore is higher than in the bore with the dielectriclining, and thus, the wave is closer to its cut-off (i.e.,closer to the evanescent regime) in the bore without thedielectric lining. This evanescence nature is in fact dis-tributed reflections occurring in the empty bore andattenuating the wave. These reflections sweep up the

fields in the antenna substrate creating significant lossesin the antenna (despite its relatively low conductivity10�3 S/m). When no losses in the antenna’s substrate are

FIG. 6. Calculated SNR maps inthe phantom for the transverse,coronal, and sagittal planes with-

out and with the water tubearray. In the left down corner of

each map, a corresponding gra-dient echo image is present.Note the almost three time

higher color scale of the SNRmaps for the water tube array.

Considerable SNR improvementwas observed when applying thewater tube array: both in terms

of the absolute values and spa-tial coverage.

FIG. 7. a: In vivo low flip angle gradient echo head imagesobtained without and with the water tube array (TE ¼ 1.7 ms,TR ¼ 150 ms, acq. voxel 3 � 3 � 5 mm3). A significant improve-

ment of image quality (eight times higher SNR) can be seen whenthe water tube array is used because of the increased B1

þ effi-

ciency of the travelling wave MR. b: Simulated product of B1þ

and B1� in the head. The product and obtained in vivo head

images correspond well to each other in terms of patterns and rel-

ative SNR gain with the water tube array (bottom profiles). Theprofiles were taken along the dashed line.

892 Andreychenko et al.

included in the FDTD simulations (which is not a realis-tic scenario), the reflected wave is not absorbed by theantenna but reflects back to the human body. It resultsin higher total longitudinal power flow in the bore thanwith the antenna conductive losses included and theadvantage of the water tube array in terms of longitudi-nal power flow is not significant (Fig. 5b). However, as itis apparent from Figure 5, the wave still does not pene-trate the head/body efficiently since the radial powerflow is poorly matched. Similar to the phantom results,it can be seen that more efficient power incoupling tothe load takes place when the water tube array is presentin the bore.

The dielectric lining could be exploited only when avery low loss material was used. Otherwise, the wavewould rapidly attenuate due to conductive losses in thedielectric lining. In this work, distilled water was used.However, for an MR experiment water has a disadvant-age as it creates very intensive unwanted background sig-nal and a large field of view has to be imaged to avoidthe water signal from folding into the imaging area.Another disadvantage of the here proposed dielectric lin-ing is the reduction of the scanner bore by 8 cm. Theinvestigation performed in this work was limited to afixed thickness of the dielectric lining and alternatingrelative permittivity. However, both thickness and rela-tive permittivity define an optical thickness of thedielectric lining that influences mode propagation char-acteristics in the waveguide. Therefore, a thinner mate-rial with higher relative permittivity that has no protonsignal, could be an attractive alternative to the proposedwater tube array. Recently, some materials with veryhigh dielectric relative permittivity values have beentested for use in MRI, that have these features (13). Inaddition, geometry of the dielectric lining is not optimalsince a low dielectric filling factor is achieved with theplastic tubes.

As known from microwave theory, a high dielectricfilling of the waveguide reduces cut-off frequencies ofthe propagating modes. In this work, only modes withthe m ¼ 1 azimuthal variation of the electromagneticfields were excited by the antenna. However, the dielec-tric lining increases the number of propagating modes(27) and modes with higher azimuthal dependenciescan be established. Presence of several modes could beexploited for RF shimming and parallel imaging as wasinitially demonstrated by Brunner et al. (28). To per-form body torso imaging, the dielectric lining could becombined with coaxial waveguide (11) body MRI.

CONCLUSIONS

An increase of Bþ1 efficiency of travelling wave humanMRI by means of a high relative permittivity dielectriclining was demonstrated by an analytical model,simulations, and experiments. The dielectric liningimproved penetration of the travelling wave into theimaging subject resulting in a locally three times higherBþ1 field and an 8-fold in vivo SNR gain. Lower waveattenuation along the longitudinal direction wasachieved by the dielectric lining. The dielectric liningcreates a reactive surface next to the wall of the scanner

bore and maximizes the radial power flow from the wallto the load by modifying the transmission/reflectionproperties of the waveguide air layer between the liningand the load. The relative dielectric permittivity ofthe lining should be chosen carefully as it definesthe amount of coupling of the reactive fields in the lin-ing to the load and, thus, the amount of Bþ1 field createdthere.

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