integrated magnetic component of a transformer and a

IEEJ Journal of Industry ApplicationsVol.9 No.6 pp.713–722 DOI: 10.1541/ieejjia.19005951Translated from IEEJ Transactions on Industry Applications, Vol.140 No.2 pp.107–116

Paper(Translation ofIEEJ Trans. IA)

Integrated Magnetic Component of a Transformer and a MagneticallyCoupled Inductor for a Three-Port DC-DC Converter

Shuntaro Inoue∗ Member, Kenichi Itoh∗ Member

Masanori Ishigaki∗ Member, Takahide Sugiyama∗ Member

J-STAGE Advance published date : October 1, 2010

This paper discusses a design method for a proposed integrated magnetic component for an isolated bidirectionalthree-port DC-DC converter (TPC). TPC comprises a dual active bridge converter (DAB) and a non-isolated bidi-rectional DC-DC converter (NBC); each converter is independently controlled with a transformer and a magneticallycoupled inductor. To reduce the size of the magnetic components, an integrated magnetic component that can integratea magnetically coupled inductor and a transformer is implemented. A 750-W magnetically integrated TPC prototypewas constructed and tested to validate the operation. The experimental results show that the efficiency of the integratedTPC is above 90% for the entire output power range, which is nearly equal to that of the conventional magnetic com-ponent. As a result, the proposed component was 10% smaller than the conventional magnetic components, and theoverall size of the integrated TPC was 33% smaller than that of the conventional one.

Keywords: magnetic integration, multiport, bi-directional, galvanic isolation, DC48V, electric vehicle

1. Background

Due to the strengthening of CO2 emission regulations ineach country, the mainstream in the powertrain of vehicles isexpected to shift from gasoline-powered vehicles to electri-cally driven vehicles such as hybrid vehicles (HV) and elec-tric vehicles (EV) (1). As a consequence, the introduction ofon-board DC-DC converters and AC-DC converters is pro-gressing. Non-isolated DC-DC converter for inverter voltageboosting (2), isolated AC-DC converters for vehicle chargingcircuits (3)–(5), and isolated DC-DC converter for 12 V auxiliarysystems (6) (7) can be given as examples. On the other hand, inEurope, the introduction of 48 V auxiliary equipment is be-ing studied as one of the measures towards CO2 emissioncontrol (8). If a system requiring multiple inputs and multipleoutputs, such as a 12 V/48 V mixing system, is realized us-ing a conventional power conversion circuit, the number ofconversion circuits, their size and cost will increase. How-ever, multiport converter technology is being developed toreduce the size and cost by sharing circuit components (9)–(14).In this paper, we propose a technology (15) (16) that integratesthe two magnetic components used in the three-port DC-DCconverter (TPC) discussed in Reference (13) and (14). In-stances of the usage of magnetic component integration tech-nology in the inductor for filters (17) can be seen as early asin the 1930s. In the power conversion circuits, the magneticcomponent integration technology has been addressed start-ing from the technology to integrate the transformer and re-actor of a forward converter (18) reported in the 1970s, to theCuk converters (19), push-pull converters (20), high-boost tapped

∗ Toyota Central R&D Labs., Inc.41-1, Yokomichi, Nagakute, Aichi 480-1192, Japan

inductor circuits (21), LLC circuits (22), and various other cir-cuits (23)–(37). In this paper, we have used the above knowledgeto study the technology for integrating magnetic componentsin conventional TPCs. In Chapter 2, we describe the oper-ating principle of the conventional TPC, and in Chapter 3,we describe the operating principle and design method of theproposed integrated magnetic component.

In Chapter 4, we compare the experimental results of a cir-cuit using the integrated magnetic component and the con-ventional circuit, and show that for the same efficiency, thecore volume of the magnetic component can be reduced by10%, and the circuit volume can be reduced by 33%. InChapter 5, we present the conclusion.

2. Three-Port DC-DC Converter

Figure 1 shows a conventional TPC that can realize twocircuit operations with a single circuit, using a magnetic cou-pling component.

Conventional TPCs have a center tap transformer and amagnetic coupling reactor as the magnetic coupling compo-nents, which are connected to the center points Mu, Mv, Mα,

Fig. 1. Conventional three-port dc-dc converter

c© 2020 The Institute of Electrical Engineers of Japan. 713

Integrated Magnetic Component of a Transformer and a Magnetically Coupled Inductor（Shuntaro Inoue et al.）

Fig. 2. Dual active bridge converter (DAB)

Fig. 3. Non-isolated bi-directional dc-dc converter (NBC)

and Mβ of a full-bridge circuit. Taking the DC voltage partsof the two full-bridge circuits as Port A and Port B, and theDC voltage part obtained from the center-tap of the trans-former as Port C, one circuit is made to have three DC ports,by which it becomes a power conversion circuit capable ofbidirectional power transmission between the ports. Whenfocusing on Port A-B, the conventional TPC can be consid-ered as a bidirectional isolated DC-DC converter (Dual activebridge converter: DAB) as shown in Fig. 2. The power trans-mitted PO due to DAB operation is expressed as the follow-ing equation, where N is the turns ratio of the transformer,fSW is the switching frequency, ϕ is the phase difference be-tween the rectangular wave voltages generated from the pri-mary and secondary side full-bridge circuits, and δ is the ONratio of the lower arm of the full-bridge circuit. The trans-mitted power between ports A and B can be controlled by thephase difference ϕ (38)–(40).

PO =2NVAVB

fSWLeqφ

(2π − δ − |φ|

2

)· · · · · · · · · · · · · · · · · · (1)

Here, Leq is the total leakage inductance when the magneticcoupling component is converted to a T-type equivalent cir-cuit. If kL and kTr denote the coupling coefficients of themagnetic coupling reactor and the center tap transformer re-spectively, and Lu and L1 denote the self-inductance of themagnetic coupling reactor and the primary of the transformerrespectively, Leq is given by the following equation.

Leq = 2N2{(1 − kL)Lu + (1 − kTr)L1}· · · · · · · · · · · · · · (2)

When focusing between the ports A and C, conventional TPCcan be regarded as a non-isolated bidirectional DC-DC con-verter (NBC) as shown in Fig. 3. The voltage ratio betweenVA and VC can be expressed in terms of switching ration δby the following equation.

VC =

(1 − δ

2π

)VA · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (3)

Therefore, the voltage ratio between ports A and C can becontrolled by the duty ratio δ.

Figure 4 shows the direction of change in magnetic flux inthe core of each magnetic coupling component according tothe direction of change in current.

Figure 4(a) shows the change in the reverse-phase se-quence current iDD (= (iu − iv)/2) related to the DAB oper-ation mode. The magnetic coupling reactor and the center

(a) DAB mode

(b) NBC mode

Fig. 4. Magnetic flux behaviors of magnetic compo-nents for the conventional TPC

tap transformer operate as a small inductance Leq shown inEquation (2). Since the transmitted power PO due to the DABoperation shown in Equation (1) is inversely proportional tothe total leakage inductance Leq, the transmitted power ofDAB operation can be designed to have a large value usingthe small inductance of the magnetic coupling reactor. Fur-ther, at this time, due to the excitation of the center tap trans-former, isolated power is transmitted through the transformerwinding. Figure 4(b) shows the change in the in-phase cur-rent iC (= iu + iv) related to the NBC operation mode. Thecenter tap transformer does not function, as the magnetic fluxinside the core is cancelled. At this time, the magnetic flux ofthe magnetic coupling reactor reinforces and operates as thenecessary reactor component Lb in the NBC operation moderepresented by the following equation.

Lb =(1 + kL)Lu

2· · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (4)

Since the inductance of the magnetic coupling reactor due tothe strengthening of the magnetic flux can be designed to belarge, the ripple current flowing through the capacitor con-nected to Port C can be decreased, and hence, the capacitorcan be designed to be small.

Figure 5 shows the ideal operation waveform when thepower is transmitted from Port B to Port C. Since the reactorcurrents iu and iv have a DC offset component due to NBC op-eration, a DC offset component occurs in the reactor core fluxϕL. However, since the DC offset component of the reactorcurrent flows only from the center tap to Port C, no DC offsetcomponent occurs in the transformer core magnetic flux ϕTr.

Although conventional TPCs can reduce the number ofcomponents compared to a system configured with DAB andNBC by circuit integration, the problem is that since twomagnetic components are required, the reduction in circuitsize is hindered by the component size and the dead spacebetween the magnetic components.

3. Magnetic Component Integrated TPC

3.1 Operating Principle Figure 6 shows the circuitdiagram of a TPC using the proposed integrated magneticcomponent. The primary winding n1, the secondary wind-ing n2 and the primary winding nL coming out of the cen-ter tap are magnetically coupled, and the coupling coefficient

714 IEEJ Journal IA, Vol.9, No.6, 2020


Fig. 5. Ideal wave forms of three-port dc-dc converter

Fig. 6. Magnetically integrated three-port dc-dc converter

between the windings n1 and n2 is kX.Figure 7 shows the change in magnetic flux of the inte-

grated magnetic component for each operation mode, andFig. 8 shows the equivalent circuit for each operation mode.In the DAB operation mode shown in Fig. 7(a), the changein magnetic flux occurs in the outer peripheral path of thecore. A self-induced electromotive force is generated in theprimary winding n1 wound on both ends of the three-leggedcore, and a mutually induced electromotive force is generatedin the secondary winding n2. Since the magnetic flux is can-celled in the center leg, no mutually induced electromotiveforce is generated in the center leg winding nL, and the centerleg does not operate as a reactor. Therefore, for the change inthe reverse-phase current, the integrated magnetic componentoperates as the small inductances N2(1−kX)L1 and (1−kX)L2,and the transformer excitation inductance kXL2 necessary forthe DAB operation shown in Fig. 8(a). In the NBC operationmode shown in Fig. 7(b), the change in magnetic flux occursalong the path through the center leg. At this time, althougha mutually induced electromotive force is generated in thesecondary winding n2 wound around both legs, since the di-rection is reversed between the two secondary windings n2, itdoes not operate as a transformer. On the other hand, since aself-induced electromotive force is generated in the primarywindings n1 and nL, they act as reactors. Therefore, when the

(a) DAB mode

(b) NBC mode

Fig. 7. Magnetic flux behaviors of the proposed inte-grated magnetic component

(a) DAB mode

(b) NBC mode

Fig. 8. Equivalent circuit for each operation mode

in-phase current changes, the integrated magnetic componentoperates as the reactor component necessary for NBC oper-ation shown in Fig. 8(b) by the primary windings n1 and nL.Thus, for the two operating modes, the integrated magneticcomponent functions as a transformer and a reactor withoutinterfering with each other.

While the trapezoidal current due to the DAB operationmode flows through both the transformer winding and themagnetically coupled reactor winding in the conventional cir-cuit, in the integrated magnetic component, since the wind-ing n1 functions as both the transformer and reactor winding,and nL functions as a reactor winding in a path through whichthe trapezoidal current does not flow, the winding loss in theDAB operation mode can be reduced. Further, the dead spacebetween the magnetic components can be reduced comparedto separate magnetic components.

Figure 9 shows the analysis model of the integrated mag-netic component. Figure 9(a) shows the parameter positionsof the integrated magnetic component, and Fig. 9(b) showsthe equivalent magnetic circuit model of the integrated mag-netic component.

As, Ac, Rs, and Rc represent the core cross-sectional areaand magnetoresistance of the outer legs and the center leg,respectively.

If the magnetic fluxes ϕ1 and ϕ2 are defined by theloop shown in Fig. 9(b), the magnetic circuit equation is as



(a) Magnetic structuer

(b) Equivalent magnetic circuit model

Fig. 9. Analysis model of the integrated magnetic com-ponent

follows:[φ1

φ2

]=

1Rs(Rs + 2Rc)

[Rs+Rc Rc

Rc Rs+Rc

]

·[n1iu − n2iα + nLiCn1iv+n2iβ−nLiC

]· · · · · · · · · · · · · · · · · · · · · (5)

First, we derive a theoretical formula relating to the trans-former operation using Equation (5). By taking the sum ofthe first and second rows and differentiating both sides, Equa-tion (5) can be rearranged as follows:

dφ1

dt+

dφ2

dt=

n1

Rs

ddt

(iu−iv)− n2

Rs

ddt

(iα−iβ) · · · · · · · · · · (6)

Applying Faraday’s law of electromagnetic induction to theterminals Mu-Mv and terminals Mα-Mβ respectively, it can beexpressed as

vu − vv = −n1dφ1

dt− n1

dφ2

dt· · · · · · · · · · · · · · · · · · · · · · (7)

vα − vβ = −n2dφ1

dt− n2

dφ2

dt· · · · · · · · · · · · · · · · · · · · · (8)

where vu, vv, vα, and vβ are the electrical potential at Mu, Mv,Mα, and Mβ, reprectively.

In order to focus only on the inductance component of thereverse-phase current change, we assume (iu − iv)/2 = i1 and(iα − iβ)/2 = i2. Substituting Equation (6) into Equations (7)and (8), we get,

vu − vv = −2n2

1

Rs

di1dt+

2n1n2

Rs

di2dt· · · · · · · · · · · · · · · · · (9)

vα − vβ = −n1n2

Rs

di1dt+

n22

Rs

di2dt· · · · · · · · · · · · · · · · · · (10)

According to Kirchhoff’s second law, Equations (9) and (10)can also be written as follows.

vu − vv = −L1di1dt+ kX

√L1L2

di2dt· · · · · · · · · · · · · · · (11)

vα − vβ = −kX

√L1L2

di1dt+ L2

di2dt· · · · · · · · · · · · · · (12)

Here, L1 and L2 denote the self-inductance of the primary

winding n1 and secondary winding n2, respectively. Compar-ing Equations (9) and (10) with Equations (11) and (12), L1

and L2 are obtained as follows.

L1 =2n2

1

Rs, L2 =

2n22

Rs· · · · · · · · · · · · · · · · · · · · · · · · · · · (13)

In the proposed integrated magnetic component, since theleakage inductance component 2(1−kL) generated in the con-ventional magnetic coupling reactor, is eliminated, the leak-age inductance component Leq used in the DAB operationshown in Equation (2) becomes as follows.

Leq = 2(1 − kX)N2L1 · · · · · · · · · · · · · · · · · · · · · · · · · · · (14)

Next, we derive the theoretical expressions relating to the re-actor operation. Applying Faraday’s law of electromagneticinduction between the terminals Mu-C and terminals Mv-Crespectively, it can be expressed as,

vu − vC = n1dφ1

dt− nL

ddt

(φ1 − φ2) · · · · · · · · · · · · · · (15)

vv − vC = −n1dφ2

dt− nL

ddt

(φ1 − φ2) · · · · · · · · · · · · · (16)

where vC is the electrical potential at Port C.To focus only on the inductance component of the change

in the in-phase current, we take

vu = vv = 0, iu = iv = iC/2 · · · · · · · · · · · · · · · · · · · · · · (17)

Hence, it can be consolidated as,

vC =(n1 + 2nL)2

2(Rs + 2Rc)diCdt· · · · · · · · · · · · · · · · · · · · · · · · · · · (18)

By Kirchhoff’s second law, Equation (18) can also be writtenas below.

vC = LbdiCdt· · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (19)

Lb denotes the inductance that functions as a reactor for NBCoperation. By comparing Equations (18) and (19), Lb can beexpressed as follows.

Lb =(n1 + 2nL)2

2(Rs + 2Rc)· · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (20)

Since the numerator of Equation (20) includes the windingsof the two outer legs n1 and the winding of the center leg nL,it can be seen that, while the separate magnetic componentcreates Lb only with the winding of the magnetic couplingreactor, the integrated magnetic component creates Lb withboth n1 and nL.

Hence, it can be designed to have a reduced core volumecompared to the conventional separate core.

Further, under the conditions of Equation (17), from Equa-tion (11), we get,

di1dt=

di2dt= 0 · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (21)

Substituting Equation (21) in Equation (12), we get

vα − vβ = 0 · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · ·(22)

Thus, in the NBC operation mode, no induced electromotive



Fig. 10. Relationship curves of Lb, L1 and As

voltage is generated in the secondary winding, indicating thatthe integrated magnetic component operates only as a reactor.

From the above, since kX does not influence Lb, but con-tributes only to the change in Leq, it is possible to design Leq

and Lb independently using kX.Next, we derive a theoretical formula for the magnetic flux

density generated inside the core, which is required for de-signing the integrated magnetic component. Using Equation(5), the magnetic flux density generated at the center leg ofthe three-legged core is obtained as follows.

φ1 − φ2 =n1(iu + iv) + 2nLiC

Rs + 2Rc· · · · · · · · · · · · · · · · · · · (23)

Therefore, if the saturation magnetic flux density of the corematerial used is BSAT, the maximum magnetic flux densityof the center leg is Bc,MAX, and the maximum current valueof Port C is iC,MAX, the following equation must be satisfiedinside the core of the center leg.

AcBSAT ≥ AcBc,MAX =(n1 + 2nL)iC,MAX

Rs + 2Rc· · · · · · · · ·(24)

On the other hand, the condition for BSAT and iC,MAX at thetwo outer legs of the three-legged core can be expressed asfollows, where the maximum flux density at both the outerlegs of the core is Bs,MAX.

AsBSAT ≥ AsBs,MAX

=TTr,ON

4n2VB+

n1 + 2nL

2(Rs + 2Rc)iC,MAX · · · · · · · · (25)

However, TTr,ON is the time of application of transformer ex-citation voltage during one cycle. If the duty ratio is δ andthe switching frequency is fSW, when δ ≥ π, it becomes,

TTr,ON =2π − δ2π fs

· · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · (26)

When δ < π, it becomes,

TTr,ON =δ

2π fs· · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · · ·(27)

Figure 10 shows the relationship between Lb and L1 withrespect to As, plotted using Equations (13) and (20). It showsthe Lb values for three types of Ac values when the number ofturns is set to nL = 2 and n1 = 1. Lb increases linearly withAc, increases sharply with As in the range of As ≤ Ac/2, andhas almost no sensitivity in the range of As > Ac/2. On theother hand, L1 increases linearly with As, but has no sensitiv-ity with respect to Ac, as seen from Equation (13). Therefore,Lb and L1 can be independently designed by adjusting Ac andAs.

(a) Integrated core

(b) Windings

Fig. 11. Dimensions of the integrated magnetic component

3.2 Design of the Core Shape of the Magnetic Com-ponent This section shows a core shape design methodwhen the winding is a rectangular wire and a copper wire. Itshould be noted that, for simplicity, only the DAB operationmode current is considered for the winding copper loss.

Figure 11 shows the definition of each dimension of theintegrated magnetic component. Figure 11(a) is the dimen-sional drawing of the core part of the integrated magneticcomponent. The length of the center leg is taken as lc, thelength of the two outer legs is ls, the depth of the core is d,the gap length of the center leg is lg, width of the core windowis w, and its height is h. As the distance between the primaryand secondary windings increases, the core window height hincreases, and as the winding width increases, the core win-dow width w increases. The core depth d is determined fromthe viewpoint of the core volume V integ and copper loss of theprimary winding Ps,integ. Using the above variables, the coresize V integ is expressed by the following equation.

Vinteg=d

(h +

2As

d

) (Ac+2As

d+2w

)−2d ·w·h−Aclg

· · · · · · · · · · · · · · · · · · · (28)

Figure 11(b) is the dimensional diagram of the windingpart of the integrated magnetic component. Assuming thewinding width of the integrated magnetic components aswinteg, the winding lengths of conventional components aslconv, the winding width of the integrated magnetic compo-nents as wconv, and the sum of the AC resistance losses of thetransformer winding and reactor winding on the primary sideof the conventional TPC for trapezoidal current as Pconv, thesum of the AC resistance loss in the primary winding n1 ofthe integrated magnetic component Ps,integ is expressed by thefollowing equation.

Ps,integ =4n1

lconv· wconv

winteg

(d +

As

d+ 2winteg

)Pconv

· · · · · · · · · · · · · · · · · · · (29)

However, Ps,integ considers only the change in the AC resis-tance with respect to the change in the winding length fromthe winding loss of the conventional separate magnetic com-ponents Pconv, and the influence of the proximity effect be-tween the windings is ignored.



Fig. 12. Calculated core size and winding loss

Table 1. Parameters of magnetic component for calcu-lated curves

Description Symbol Value

Height of core window h 5.0 mmWidth of core window w 6.4 mm

Length of core gap lg 0.6 mmCross-sectional area of core Ac 97 mm2

Cross-sectional area of core As 223 mm2

Total size of conventional core Vconv 16.6 cm3

Chosen total size of proposed core Vinteg 14.9 cm3

Total resistance of conventional components at 175 kHz Rconv 16.9 mΩWinding length of conventional component lconv 437 mm

Width of conventional windings wconv 7.0 mmWidth of proposed windings winteg 5.0 mm

Figure 12 shows the variation of the core volume V integ andthe primary winding copper loss Ps,integ with respect to thecore depth d. The magnetic component parameters used inthe plot are shown in Table 1. Although V integ decreases asthe core depth d increases, since the cross-sectional shape ofthe core becomes elongated, winding length increases, andthe winding copper loss increases. Figure 12(a) is a relationaldiagram of V integ to d, plotted using Equation (28). Since thesize of the core window is fixed, V integ decreases as d in-creases, as the core cross-sectional area is kept constant. Thetotal core volume Vconv for a separate core designed with thesame maximum magnetic flux density and number of turns isalso shown.

If the depth d of the integrated magnetic component ismore than 30 mm, a core volume reduction effect can be ob-tained. Figure 12(b) is a plot of the winding loss Ps,integ ver-sus core depth d plotted using Equation (29). As the coredepth d is increased from 0, the winding loss Ps,integ decreasessharply, reaches a minimum at a point where the winding be-comes a square, and then gradually increases. The figure alsoshows the primary-side conventional winding loss Pconv. Ifthe depth d of the integrated magnetic component is between5 mm and 65 mm, the effect of winding copper loss reductioncan be obtained. From Fig. 12, it can be seen that if the depthd of the magnetic component is between about 30 mm and65 mm, the winding loss can be suppressed while reducing

(a) Proposed integrated magnetic component

(b) Convention coupling inductor (c) Conventional center tappedtransformer

Fig. 13. 3D model for FEM simulation

Table 2. Simulation parameters of magnetic components

Description Symbol Conventional Integrated

Transformer turn number n1, n2 1, 4 -Inductor turn number nLu, nLv 2, 2 -

Integrated magneticcomponent turn number

n1, n2,nL

-1, 4,

2

Effective cross section Ac, As 170 mm2 97 mm2,

223 mm2

Gap length lgap 0.5 mm 0.6 mmWindow size of core w × h - 6.4 ×5.0 mm

Depth of core d - 37.2 mmCore size of tranformer - 8.3 cm3 -

Maximum flux density - Trans. : 118 mTInductor : 318 mT

Bs,MAX : 109 mTBc,MAX : 309 mT

Core size of inductor - 8.3 cm3 -Total core size Vconv, Vinteg 16.6 cm3 14.9 cm3

the volume, as compared to the conventional magnetic com-ponent. In the prototype of this paper, depth d = 37. wasselected as the depth that can reduce the winding loss whilereducing V integ, compared to the conventional core volume.3.3 Examination of Winding Configuration In the

proposed integrated TPC, the leakage inductance componentLeq required for DAB operation, which was adjusted usingthe coupling coefficient kL of the magnetic coupling reactorpreviously, must be adjusted with the coupling coefficient kX

of the primary winding n1 and the secondary winding n2. Aneffective means to obtain the desired Leq value by reducingkX is to increase the distance between the primary windingn1 and the secondary winding n2. However, if the distancebetween the windings is simply increased, the height of thecore will increase, which will result in the decrease of the sizereduction effect of magnetic component integration.

Therefore, in this paper, secondary winding n2 of the inte-grated magnetic component was wound perpendicular to theprimary winding n1 to reduce kX, and by simulation, it wasconfirmed that the required Leq value was obtained.

Figure 13 shows the cross-sectional view of the three-dimensional model of the integrated magnetic componentand the separate magnetic component. Table 2 shows eachparameter. Using this model, the leakage inductance Leq ofthe integrated magnetic component and separate magneticcomponent was compared by three-dimensional electromag-netic field simulation. The total leakage inductance of the in-tegrated magnetic component shown in Fig. 13(a) is 3.2 μH,



which is found to be equivalent to the sum (3.0 μH) of theleakage inductances of the magnetic coupling reactor modelshown in Fig. 13(b) and the center tap transformer modelshown in Fig. 13(c). Hence, in the prototype, we adopteda winding configuration where the secondary winding waswound perpendicular to the primary winding. It should benoted that if a leakage inductance larger than the value ob-tained by the vertical winding is required, it is necessary tofurther increase the core height of the integrated magneticcomponent and increase the distance between the primaryand the secondary windings.3.4 Trial Calculation of the Items in the Magnetic

Component Losses Using Three-dimensional Model Inthe above, a design method considering the winding loss dur-ing DAB operation of the integrated magnetic componentwas shown. However, accurate loss estimation of the en-tire magnetic component requires analysis considering theiron loss, proximity effect of the windings, and the DAB andNBC operations. Therefore, the loss composition of the mag-netic component was analyzed by three-dimensional electro-magnetic simulation using JMAG FEM. In the analysis, theloss of the proposed integrated magnetic component shown inFig. 13(a) was compared with that of the conventional trans-former/magnetic coupling reactor set shown in Figs. 13(b),(c).

Figure 14 shows the results of the analysis of the loss com-position of the magnetic component. Figure 14(a) shows theloss itemization of PA single load (transmission from Port

(a) PA single mode (PA.out = 750 W)

(b) PC single mode (PC.out = 750 W)

Fig. 14. Result of loss analysis in the experimental circuits

B to Port A only) mode. By using the integrated magneticcomponent, the total value of the magnetic component loss isreduced by 3.4 W from 12.1 W to 8.7 W. This is because, thewindings n1 of the two outer legs of the integrated magneticcomponent functions as a transformer winding and a reactorwinding as shown in Equation (20), leading to a reduction inthe winding loss generated by the trapezoidal current in DABoperation. Figure 14(b) shows the loss itemization of PC sin-gle load (transmission from Port B to Port C only) mode. Byusing the integrated magnetic component, the total value ofthe magnetic component loss increases 1.9 W from 18.2 Wto 20.1 W. This is because, the reactor windings, which wereconventionally two parallel windings, are consolidated intothe center leg winding. From the above, if the integratedmagnetic component is designed to be equal to the sum of theiron losses of the separate magnetic components as shown inFig. 14, a reduction in the copper loss in PA single load modecan be expected. Furthermore, when the thickness of cop-per in the center winding is designed such that the coppervolumes of the reactor winding and center winding of the in-tegrated magnetic component are equal, an improvement inefficiency in the PC single load mode can be expected. How-ever, increasing the center winding thickness increases thecore height and reduces the volume reduction effect.

4. Experimental Results and Discussion

4.1 Experimental Circuit Configuration Figure 15shows the configuration of the experimental circuit, and Ta-ble 3 shows the list of circuit parameters. A DC power supplywas connected to Port B, an electronic load was connected toPort A and Port C, and the voltages of Port A, Port B, and PortC were set to VA,out = 48 V, VB,in = 200 V, and VC,out = 12 V,respectively. Figure 16 shows the appearance of two pro-totypes designed at 750 W and 175 kHz, and Table 4 showsthe measured magnetic characteristics of the prototype of the

Fig. 15. Experimental setup

Table 3. Circuit parameters of the prototypes

Description Symbol Value

Switching frequency fSW 175 kHzRated power PO,MAX ± 750 W

Rated DC Voltage VA,out, VB,in,VC,out

48 V, 200 V,12 V

Core material - Ferrite PC95Saturation magnetic flux density BSAT 380 mT (120◦C)Maximum magnetic flux density Bc,MAX, Bs,MAX 309 mT, 109 mT

Primary Si MOSFET S1 ∼ S4 TPW4R50ANH(Toshiba)Secondary GaN FET S5 ∼ S8 TPH3006LD(Transphorm)

DC capacitor CA, CB,CC

44 μF, 4 μF,286 μF

Dead time Td 200 ns



Fig. 16. Prototypes of the integrated TPC and the conventional TPC

Table 4. Measured parameters of magnetic components

Description Symbol Conventional Integrated

Inductance of transformer L1, L245 μH,723 μH

48 μH,767 μH

Coupling rate of Integrated core kX - 0.997Coupling rate of transformer kTr 0.999 -

Coupling rate of inductor kL 0.97 -Inductance for NBC function Lb 1.70 μH 1.89 μHInductance for DAB function Leq 4.33 μH 4.85 μH

integrated magnetic component and the separate magneticcomponent.

The prototype on the left side of Fig. 16 is the magneticcomponent integrated-type TPC. As the saturation magneticflux density of ferrite core material at 120◦C is 380 mT,the design value of the maximum flux density Bc,MAX ofthe center leg of the integrated magnetic component wasset to 309 mT, and the design value of the maximum fluxdensity Bs,MAX of its two outer legs was set to 109 mT.The low-voltage side switching element used a Si MOS(Toshiba, TPW4R50ANH) with a withstand voltage of 100 V,and the high-voltage side switching element used a GaNFET (Transphorm, TPH3006LD) with a withstand voltageof 600 V, a next-generation semiconductor component, whichwas adopted with the purpose of increasing the efficiency ofthe circuit. The core material of the magnetic component wasMn-Zn ferrite (TDK, PC95). Although kX takes a very highvalue of 0.997, since it is possible to change only the leakageinductance Leq while keeping the self-inductance (L1, L2, andLb) of the integrated magnetic component almost constant bychanging the distance between the primary and secondarywindings, fine adjustment of kX according to the design re-quirements is possible. The prototype on the right side ofFig. 16 is conventional TPC created to verify the applicationeffects of the integrated magnetic component technique.

The circuit parameters other than the magnetic componentswere same as those of the magnetic component integrated-type TPC, and PQ core (PC95PQ32/20Z-12) was used forthe magnetic core of the transformer and the magnetic cou-pling reactor. Each magnetic component parameter is shownin Tables 2 and 4. The total core volume was 16.6 cm3 for theconventional type but 14.9 cm3 for the integrated type, show-ing a 10% reduction. Further, since the wiring space betweenthe magnetic components can also be reduced compared tothe conventional type, the overall circuit size is reduced by

Fig. 17. Experimental waveforms in case of PA,out =375 W and PC,out = 375 W

33%, from 300 cm3 (2.50 W/cm3) to 200 cm3 (3.75 W/cm3).4.2 Experimental Results Figure 17 shows the mea-

sured waveforms of Vuv, Vαβ, and iu, in the combined loadmode when 375 W of power was simultaneously transmittedfrom Port B to Port A and Port C. Similar to the ideal op-eration waveform of Fig. 5, the phase-U current waveformwas such that the DC current with the ripple due to NBC op-eration overlapped with the trapezoidal current due to DABoperation.

Figure 18 shows the comparison of the efficiencies of themagnetic component integrated-type TPC and the conven-tional TPC in the single load mode. Figure 18(a) shows theefficiency curves of the proposed circuit and the conventionalcircuit in PA single load mode, and it can be seen that atthe rated output of 750 W, the efficiency improved by 0.3%(93.4%) in the proposed circuit compared to the conventionalcircuit. Figure 18(b) shows the efficiency curves of the pro-posed circuit and the conventional circuit in PC single loadmode, and it can be seen that at the rated output of 750 W,the efficiency of the proposed circuit was reduced by 0.4%(90.8%) compared to the conventional circuit.

Table 5 shows the comparison between the experimentalresults and the simulation results. The experimental results ofthe difference in circuit loss Δ Loss when changing from theconventional magnetic component to the integrated magneticcomponent were −2.3 W for PA single load mode and +3.5 Wfor PC single load mode, which was consistent with the in-crease/decrease trend of loss obtained by simulation (−3.4 Win PA single mode and +1.9 W in PC single mode), thus con-firming the validity of the simulation results. Hence, it canbe applied in future applications and design. From the above



(a) PA single mode

(b) PC single mode

Fig. 18. Comparison of measured efficiencies of inte-grated TPC and conventional TPC

Table 5. Loss comparison between experimental andanalitical results

Control mode ΔLoss in Exp. ΔLoss in FEM.

PA single mode (PA,out = 750 W) −2.3 W −3.4 WPC single mode (PC,out = 750 W) +3.5 W +1.9 W

results, it was confirmed that the magnetic component inte-grated TPC can reduce the core volume by 10% and the cir-cuit size by 33%, while maintaining the conversion efficiencyat the same level as the conventional type.

5. Conclusion

In this paper, we have proposed a transformer/reactor in-tegrated TPC. The proposed circuit realizes the integrationof the transformer and magnetic coupling reactor used in theconventional TPCs, using a three-legged core and a verticalwinding structure. We created the prototypes of the proposedcircuit and the conventional circuit at 750 W, 175 kHz, andmeasured their efficiencies to verify the validity of the pro-posed circuit. From the measurement results, it can be seenthat the efficiency is 90% or more over a wide output range,and the same efficiency as the conventional type is obtained.Further, with the proposed circuit, we were able to demon-strate the possibility of reducing the core volume by 10% andthe circuit volume by 33%, while maintaining the same effi-ciency. Since the 33% volume reduction of TPC will makeit easy to replace the conventional single-port output powersupply unit with a multiport power supply unit, it is expectedto promote the spread of 12 V and 48 V vehicle accessorysystems aiming at higher efficiency. In the future, we aim tofurther reduce the size of the integrated magnetic componentby increasing the frequency of the circuit.

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Shuntaro Inoue (Member) received the B.S. degree in mechanical en-gineering from Osaka University, Japan, in 2011, andthe M.S. degree in precision engineering from theUniversity of Tokyo, Japan, in 2013. Since 2013, hehas been with Toyota Central Research & Develop-ment Laboratories, Inc. His research interests includepower electronics and mechatronics. He receivedthe Japan Society of Mechanical Engineers (JSME)Kansai-branch Best Presentation Award in 2011, theInstitute of Electrical Engineers of Japan (IEEJ) Best

Presentation Award in 2016, the IEEJ Encouragement Award in 2017, andAPEC 2019 Outstanding Presentation Awards.

Kenichi Itoh (Member) received the B.S. and M.S. degrees in ap-plied physics from University of Tsukuba, Japan, in2005 and 2008, respectively. Since April 2008, hehas been with Toyota Central Research & Develop-ment Laboratories, Inc. His research interests includeDC/DC converter and power devices. He is a mem-ber of the Institute of Electrical and Electronics Engi-neers (IEEE) and the Institute of Electrical Engineersof Japan (IEE-Japan).

Masanori Ishigaki (Member) received the B.S. degree in electri-cal engineering from Tokyo Metropolitan University,Tokyo, Japan, in 2005 and the M.S. degree in elec-trical and electronic engineering from Tokyo Insti-tute of Technology, Tokyo, Japan, in 2007. Since2007, he has been with Toyota Central R&D Labs,Inc., Nagakute, Japan. He was with the Electron-ics Research Department, Toyota Research Instituteof North America, from 2014 to 2017. His researchinterests include electrical systems for vehicle elec-

tronics and power converter circuits. Mr. Ishigaki was the recipient of theIPEC 2010 Second Prize Paper Award and the APEC 2012 and 2017 Out-standing Presentation Award.

Takahide Sugiyama (Member) received his B.S. and M.S. in Elec-trical, electronics and information engineering fromNagoya Institute of Technology, Japan, in 1989 and1991. Since 1991, he has been with Toyota Cen-tral R&D Laboratories, Inc., Nagakute, Japan. Cur-rently, R&D team manager for power supply systemsfor electrified vehicles. His research interests includepower semiconductor devices. He is a member of In-stitute of Electric Engineers of Japan (IEEJ). He isalso a member of Japan Society of Applied Physics

(JSAP).


integrated magnetic component of a transformer and a

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